Multi-phase exciting and multi-phase scanning with zero-sum sequences

Abstract

To provide apparatuses and methods of multi-phase exciting and multi-phase scanning with zero-sum sequences for a touch panel.SOLUTION: A method of multi-phase scanning of a touch panel includes: selecting a sequence having a number of one values, negative one values, and zero values, the one values corresponding to an in-phase drive signal, the negative one values corresponding to an opposite-phase drive signal, and the zero values corresponding to a reference signal (e.g., reference voltage or ground), a sum of the sequence being equal to zero; applying one of the in-phase drive signal, the opposite-phase drive signal, or the reference signal to each of a first set of electrodes at a first stage according to the sequence; rotating the sequence to obtain a rotated sequence; applying one of the signals according to the rotated sequence; and receiving sense signals to detect a presence of an object on the touch panel.SELECTED DRAWING: Figure 15

Description

[Technical Field]

[0001] Embodiments of the present invention relate to the field of user interface devices, and more particularly to multiphase excitation and multiphase scanning using zero-sum sequences in touch detection devices. [Background technology]

[0002] Computing devices such as notebook computers, personal digital assistants (PDAs), and mobile phones have user interface devices, also known as human interface devices (HIDs). One type of user interface device that has become relatively common is touch sensing devices such as touch sensor pads (commonly called touchpads), touch sensor sliders, touch sensor buttons, touch sensor keyboards, touchscreens, and touch panels.

[0003] A basic notebook-style touch sensor pad emulates the functionality of a personal computer (PC) mouse. Touch sensor pads are typically embedded within PC notebooks for built-in portability. They replicate the x / y movement of a mouse using two defined axes, which include a collection of sensor elements that detect the position of conductive objects such as fingers. Right / left mouse button clicks can be replicated by two mechanical buttons located near the touch pad, or by tapping commands on the touch sensor pad itself. Touch sensor pads provide a user interface device for performing functions such as pointer positioning or item selection on a display.

[0004] Another user interface device that has become relatively common is the touchscreen. Also known as a touchscreen, touch panel, or touchscreen panel, a touchscreen is typically a display overlay that is pressure-sensitive (resistive), electrically sensitive (capacitive), acoustically sensitive (SAW - surface acoustic wave), or light-sensitive (infrared). The effect of such an overlay makes it possible to use the display as an input device, eliminating the need for a keyboard and / or mouse as the primary input device for interacting with the display's content. Such displays can be attached to a computer or connected to a network as a terminal. There are several types of touchscreen technology, such as optical imaging, resistive, surface wave, capacitive, infrared, dispersed signal, and strain gauge technology. Touchscreens are becoming common in retail environments, point-of-sale (POS) systems, ATMs, mobile phones, game consoles, and PDAs, where a stylus is sometimes used to interact with the graphical user interface (GUI) and input data.

[0005] Generally, capacitive sensing devices are intended to be used as a replacement for mechanical buttons, knobs, and other similar mechanical user interface controls. Capacitive sensing devices eliminate complex mechanical switches and buttons and provide reliable operation under harsh conditions. Furthermore, capacitive sensing devices are widely used in modern customer applications and provide new user interface options for existing products. Capacitive touch sensor elements can be arranged in the form of a sensor array for touch sensing surfaces. When a conductive object, such as a finger, touches or comes into close proximity to the touch sensing surface, the capacitance of one or more capacitive touch sensor elements changes. This change in capacitance of the capacitive touch sensor elements can be measured by an electrical circuit. An electrical circuit supporting one operating mode converts the measured capacitance of the capacitive touch sensor elements into a digital value.

[0006] Capacitive sensing circuits have two main operating modes: self-capacitive sensing mode and mutual-capacitive sensing mode. Self-capacitive sensing mode is also called single-electrode sensing mode because each sensor element requires only one connecting wire to the sensing circuit. In self-capacitive sensing mode, when the sensor element is touched, the touch capacitance of the finger is added to the sensor capacitance, so the sensor capacitance increases. In mutual-capacitive sensing mode, a change in mutual capacitance is detected. Each sensor element uses at least two electrodes, one of which is the transmitter (TX) electrode (also called the transmitter electrode herein) and the other is the receiver (RX) electrode. When a finger touches or comes close to the sensor element, the finger diverts a portion of the electric field to ground (e.g., chassis or earth), thus reducing the capacitive coupling between the receiver and transmitter of the sensor element.

[0007] This disclosure is illustrated in the attached drawings as examples, not limitations. [Brief explanation of the drawing]

[0008] [Figure 1] This is a schematic diagram of a capacitance detection device according to one embodiment. [Figure 2] This is a block diagram of a processing device, according to one embodiment, which includes a capacitance detection circuit (CSC) for measuring mutual capacitance or self-capacitance using multiphase scanning. [Figure 3A] This is a block diagram of the capacitance detection circuit shown in Figure 2, according to one embodiment. [Figure 3B] This is a block diagram of a sequence generation circuit according to one embodiment. [Figure 4A] This is a schematic diagram of the detection grid of a capacitance detection device according to one embodiment. [Figure 4B] This is a schematic diagram of the detection grid of a capacitance detection device according to one embodiment. [Figure 5A] This figure compares the electromagnetic emission related to the capacitance detection device in each embodiment when the sum of the excitation sequences is 1 and when it is 0. [Figure 5B] This figure compares the electromagnetic emission related to the capacitance detection device in each embodiment when the sum of the excitation sequences is 1 and when it is 0. [Figure 6] This is a schematic diagram of the detection grid of the capacitance detection device shown in Figure 4, according to one embodiment. [Figure 7] This is a schematic diagram of a detection grid for a capacitance detection device having a first scanning region, a second scanning region, and an overlapping region, according to one embodiment. [Figure 8] This graph shows the baseline correction across the TX electrode scanning area of ​​the detection grid of a capacitance detection device 800 according to one embodiment. [Figure 9] This is a flowchart of one embodiment of a method for multiphase scanning. [Figure 10] This is a flowchart of an analytical ambiguity reduction method for reconstructing finger response on a capacitive touch panel, according to one embodiment. [Figure 11A] This is a capacitance sensor matrix showing deconvolved data for multiphase transmission using eight electrodes excited by a zero-sum excitation sequence, according to one embodiment. [Figure 11B] This is a capacitance sensor matrix showing filtered data obtained by applying a common mode filter according to one embodiment. [Figure 11C] This is a capacitance sensor matrix showing filtered data after restoring common-mode values ​​along a set of RX electrodes to correct the baseline, according to one embodiment. [Figure 11D] This is a capacitance sensor matrix showing filtered data after applying an orthogonal common-mode filter according to one embodiment. [Figure 12A] This graph compares the relationship between noise reduction coefficients and peak emission for three methods of multiphase transmission according to one embodiment. [Figure 12B] Figure 12A is a graph comparing the relationship between the noise reduction coefficient and the average emission for the three methods described. [Figure 13] A series of schematic diagrams showing a zero-sum polyphase receiver 1300 according to an embodiment. [Figure 14] A graph showing a noise comparison between a detection signal received when zero-sum polyphase reception is performed and a detection signal received when zero-sum polyphase reception is not performed according to an embodiment. [Figure 15] A flowchart of a zero-sum polyphase transmission method according to an embodiment. [Figure 16] A flowchart of a zero-sum polyphase reception method according to an embodiment. [Figure 17] A diagram showing an embodiment of the core architecture of a PSoC (registered trademark) processing device, such as that used in the product group of PSoC3 (registered trademark) provided by Cypress Semiconductor Corporation (San Jose, California).

Embodiments for Carrying Out the Invention

[0009] Touch panels (e.g., touchscreens) used in consumer electronics and automotive environments are increasingly utilizing relatively large screen sizes, such as those exceeding 12.3 inches. Simultaneously, these large screen sizes impose relatively stringent specification requirements regarding electromagnetic emissions, particularly in the long-wave frequency band (e.g., frequencies below approximately 540 kHz) and the medium-wave frequency band (e.g., frequencies between approximately 530 kHz and 1700 kHz). Existing capacitive sensing devices utilizing square wave excitation emit electromagnetic emissions exceeding current limits. Furthermore, the operation of capacitive sensing devices can be affected by external common-mode noise from sources such as liquid crystal display (LCD) screen operation, electrical ballast, handheld transceivers, amplitude modulation (AM) radio, and test procedures (e.g., testing in electromagnetic immunity (EMI) anechoic chambers and / or electromagnetic compatibility (EMC) anechoic chambers). Therefore, there is a need for capacitive sensing touch panels that perform excitation and scanning while minimizing electromagnetic emissions and improving electromagnetic immunity.

[0010] Prior art for reducing electromagnetic emissions includes frequency spreading (e.g., TX spreader) and sine wave excitation. Frequency spreading is a technique for reducing (e.g., spreading) the harmonic peaks in the emission spectrum, but modulates a square wave excitation signal. Frequency spreading mitigates electromagnetic emissions, but can limit the detection frequency range and cause electromagnetic immunity problems (e.g., reduce the signal-to-noise ratio (SNR) and degrade robustness against LCD noise) when the panel is large and the scan time is short. As the harmonic width increases, the noise transfer function of the receiving channel increases, thereby increasing the system's vulnerability to broadband noise. In the case of sine wave excitation, only a single harmonic exists in the emission spectrum, and the system can be configured such that the main harmonic is within the frequency range to minimize noise. However, this can lead to increased power consumption, increased cost, and more stringent requirements for scan time and phase. Furthermore, sine wave excitation does not reduce electromagnetic emissions compared to square wave excitation. This is because the radiated energy is similar but concentrated in the main harmonic. Additionally, since the frequency range is limited, the SNR decreases.

[0011] This specification describes various embodiments of apparatuses and methods for multiphase excitation and multiphase scanning using a zero-sum sequence for a touch panel. According to the described apparatuses and methods, a touch panel with sufficiently low electromagnetic emissions and sufficiently high SNR becomes possible. In particular, according to the methods described herein, multiphase transmission for reducing electromagnetic emissions and multiphase reception for suppressing external common-mode noise become possible. The methods described herein support the mitigation of white noise (such as multiphase scanning), are applicable to both mutual capacitance sensing and self-capacitance sensing, and support the multi-touch characteristics of the touch panel.

[0012] Apparatus and methods for multiphase transmission and multiphase reception are described. Multiphase transmission refers to an electrode excitation method in which multiple electrodes are simultaneously excited by excitation signals having different phases. Multiphase reception refers to a panel scanning technique for systems in which the number of sensors is greater than the number of receiver channels. In one embodiment, a capacitance sensing circuit generates a common-phase drive signal and a reverse-phase drive signal, generating a sequence having multiple positive 1s, negative 1s, and 0s. The positive 1s correspond to the common-phase drive signal, the negative 1s correspond to the reverse-phase drive signal, and the 0s correspond to a reference signal derived from a voltage potential or ground potential. The sum of the sequence (e.g., the sum of each of the positive 1s, negative 1s, and 0s in the sequence) is zero. In one embodiment, the capacitance sensing circuit applies one of the common-phase drive signal, the reverse-phase drive signal, or the reference signal to a first set of transmitter (TX) electrodes in a sequence at a first stage of a plurality of scanning stages. The capacitance sensing circuit rotates the sequence to obtain the rotated sequence. The capacitance sensing circuit, in the second stage of a series of scanning stages, applies one of the following to a second set of transmitter (TX) electrodes according to a rotated sequence: a common-mode drive signal, an out-of-mode drive signal, or a reference signal. The capacitance sensing circuit receives a detection signal from a set of receiver electrodes to detect the presence of an object on the touch panel. The detection signal represents the capacitance associated with the set of receiver electrodes. It should be noted that the capacitance sensing circuit can detect not only conductive objects but also other objects (also called touch objects). An object or touch object is any object that interferes with the electric field for mutual capacitance sensing technology, thereby reducing the coupling between the receiver and transmitter electrodes. For example, if a user touches the touch surface while wearing gloves, the capacitance sensing circuit may not detect the user's fingers as a conductive object, but the user's fingers still interfere with the electric field, reducing the coupling between the electrodes, so the capacitance sensing circuit can still detect the user's fingers.It should also be noted that the embodiments described herein can be used on a touch panel having three or more transmitter and receiver electrodes, as described below.

[0013] The embodiments described herein provide multiphase TX and multiphase RX technologies that reduce electromagnetic emission and improve electromagnetic immunity, thereby enabling the application of capacitive sensing to large touchscreens (e.g., larger than 12.3 inches). The technology is based on the requirement to minimize electromagnetic emission by zeroing the sum of sequences corresponding to the in-phase drive signal, out-of-phase drive signal, and reference signal. Furthermore, the technology is based on determining an inverse convolution cyclic matrix, which is the pseudo-inverse of a singular (e.g., non-invertible) excitation matrix, computing the inversely convolved data using the inverse convolution cyclic matrix and the sensing signal (digital data), and reconstructing the touch signal on the capacitive sensing touchscreen. In some cases, the technology relies on applying appropriate filtering techniques (common-mode filtering, orthogonal common-mode filtering, neural network-based filtering, etc.) and / or post-processing algorithms (in which case filtering may not be required). The embodiments described herein are applicable to a variety of touch sensing devices, including indium tin oxide (ITO) based designs.

[0014] The following description includes numerous specific details, such as examples of particular systems, components, and methods, in order to provide a good understanding of some embodiments of the present invention. However, it will be obvious to those skilled in the art that at least some embodiments of the present invention can be carried out without these specific details. In other examples, well-known components or methods are not described in detail or are presented in the form of simple block diagrams to avoid unnecessarily obscuring the invention. Thus, the specific details described are merely illustrative. Certain implementations may differ from these exemplary details, but may still be considered to fall within the spirit and scope of the present invention.

[0015] Any reference in this specification to “one embodiment” or “embodiment” means that certain features, structures, or characteristics described in relation to that embodiment are included in at least one embodiment of the present invention. The phrase “in one embodiment” appearing in various places in this specification does not necessarily refer to the same embodiment.

[0016] Figure 1 is a schematic diagram of a capacitance detection device 100 according to one embodiment. The capacitance detection device 100 is a zero-sum multiphase capacitance detection device. The capacitance detection device 100 provides at least one example of multiphase excitation and multiphase scanning using a zero-sum sequence to support various capacitance detection modes such as mutual capacitance or self capacitance. Figure 1 shows a capacitance detection device having a touch controller architecture having a detection grid 102 (e.g., detection panel or capacitance matrix) having a rectangular array of detection electrodes. The rectangular array of detection electrodes may include an integer M TX electrodes 104 and an integer N RX electrodes 106. TX multiplexers 108 and RX multiplexers 110 can connect the panel electrodes to one or more detection channels and can multiplex signals between a common-mode drive signal, an out-of-mode drive signal, and a reference signal.

[0017] The capacitance detection device 100 may include a capacitance detection circuit that can transmit and / or receive. In some embodiments, the TX signal generator 112 can generate a common-mode drive signal and a reverse-mode drive signal, and can select an excitation sequence having a plurality of positive 1s, negative 1s, and 0s corresponding to the common-mode drive signal, the reverse-mode drive signal, and the reference signal, respectively. The excitation sequence can be selected such that the sum of the excitation sequences is zero. If the excitation sequence has a sum of zeros, it can be called a zero-sum excitation sequence. In the first scanning stage, the TX signal generator 112 may simultaneously apply at least one of the common-mode drive signal, the reverse-mode drive signal, or the reference signal to the TX electrode 104 according to the excitation sequence. The common-mode drive signal, the reverse-mode drive signal, and the reference signal are applied to the TX electrode 104 according to the excitation sequence. The excitation sequence can be selected such that the sum of the excitation sequences is zero. The excitation sequence can be rotated, and in the second scanning stage, the TX signal generator 112 simultaneously applies at least one of a common-mode drive signal, an out-of-mode drive signal, or a reference signal to the TX electrode 104 according to the rotated excitation sequence. Furthermore, the RX signal receiver 114 can receive a detection signal from the RX electrode 106 to detect the presence of an object (such as a finger or other conductive object) on the touch panel of the capacitance detection device 100. The detection signal represents the capacitance associated with the RX electrode 106.

[0018] In some embodiments, the RX signal receiver 114 can detect at least one of a common-mode drive signal, a reverse-mode drive signal, or a reference signal in a first scanning stage according to the excitation sequence. The RX signal receiver 114 detects these signals substantially simultaneously at the RX electrode 106. In a second scanning stage, the RX signal receiver 114 can detect at least one of a common-mode drive signal, a reverse-mode drive signal, or a reference signal according to the rotated excitation sequence. The detected signal may represent the capacitance associated with the receiver electrode 106 and is receivable by the RX signal receiver 114 to detect the presence of an object on the touch panel.

[0019] As illustrated in Figure 1, in the first scanning stage, the excitation sequence (e.g., excitation pattern) selected by the TX signal generator 112 is {+1,-1,0,0,···,0}, which is multiplexed by the TX multiplexer 108, indicating that an in-phase drive signal is applied to TX electrode 104(1), an out-of-phase drive signal is applied to TX electrode 104(2), and a reference signal is applied to TX electrodes 104(3) to 104(M). Each of the signals corresponding to the excitation sequence can be applied to the TX electrodes substantially simultaneously. In the second scanning stage, the excitation sequence can be rotated (or cycled), so that the rotated excitation sequence is {0,+1,-1,0,0,···,0}. In other words, the "0" applied to TX electrode 104(M) can be shifted so that it is applied to TX electrode 104(1), the "1" applied to TX electrode 104(1) can be shifted so that it is applied to TX electrode 104(2), and so on. The same rotation can be applied to each subsequent scanning stage. Such a rotation can generally be called a cyclic rotation. This rotation can be generated by a sequencer (not shown in Figure 1).

[0020] An excitation matrix (H) can be established using a given excitation sequence in each scanning stage. The excitation matrix H can represent a pattern of excitation sequences for exciting the TX electrode 104. The product of the excitation matrix (e.g., via matrix multiplication) and the finger response (x) (such as a direct response map of a finger or conductive object on a capacitive touchscreen panel) returns a measured signal (s), such as a detection signal measured by the RX electrode 106 and received by the RX signal receiver 114. Thus, the reciprocal of the excitation matrix (H) (e.g., via matrix multiplication) -1 The finger response can be reconstructed by multiplying the measured signal(s) by the value of (). Additional details of this process are described in more detail, at least in relation to Figure 6.

[0021] In some embodiments, the rotation in each scanning stage may be a shift of a number (or position) other than 1. For example, in some embodiments, a "0" applied to TX electrode 104(M) can be shifted to be applied to TX electrode 104(2), a "1" applied to TX electrode 104(1) can be shifted to be applied to TX electrode 104(3), and so on. It should be noted that the number of shifts may be a number other than 1 or 2. In other embodiments, the sequencer can create various different patterns for the shifts. In such cases, the excitation matrix can be determined, for example, via a machine learning algorithm.

[0022] Figure 2 is a block diagram of a processing unit 200, according to one embodiment, which includes a capacitance sensing circuit (CSC) 202 for measuring mutual capacitance or self capacitance using multiphase scanning. The processing unit 200 includes the capacitance sensing circuit 202, a CPU core 206, and one or more communication interfaces, such as an I2C interface 208, which can communicate with a host processing unit 240 via an I2C bridge 230. Alternatively, other types of communication interfaces may be used to communicate with the host processing unit 240. In other embodiments, the capacitance sensing circuit 202 can be implemented in other types of processing units having components similar or dissimilar to those of the processing unit 200. In other embodiments, the CPU core 206 can communicate with the host processing unit 240 using other communication devices and protocols other than I2C, as can be understood by those skilled in the art who are interested in this disclosure. In some embodiments, the processing unit 200 does not communicate with the host processing unit 240 via the I2C bridge 230, but may communicate with other peripheral devices or may not communicate with other external devices.

[0023] In this embodiment, the capacitance detection circuit 202 can measure the mutual capacitance or self-capacitance of the sensor elements of the capacitance detection device 100. As described above, the capacitance detection device 100 has M (rows) of TX electrodes 104 and N (columns) of RX electrodes 106. Each intersection of the TX electrodes 104 and the RX electrodes 106 represents a sensor element. The capacitance detection circuit 202 can measure mutual capacitance by applying a plurality of TX signals (e.g., in-phase drive signals, out-of-phase drive signals, or reference signals) onto the drive line 210 according to an excitation sequence, and can receive detection signals on the detection line 220. The capacitance detection circuit 202 can use the measured capacitance to detect the presence of an object and the object's position, motion, velocity, and / or acceleration. In one embodiment, the capacitance detection circuit 202 can measure mutual capacitance or self-capacitance and convert these into digital values. In one embodiment, the capacitance detection circuit 202 is configured to use digital values ​​to detect a single touch and a corresponding single gesture. In other embodiments, the capacitance sensing circuit 202 is configured to detect multiple touches and corresponding multi-touch gestures using digital values.

[0024] In another embodiment, the CPU core 206 can process digital values ​​received from the capacitance sensing circuit 202 to detect the presence of an object, the movement of the object such as its velocity, acceleration, and distance, and single and multi-touch gestures that can be understood by those skilled in the art who are interested in this disclosure. The CPU core 206 can also be used to receive raw data from the capacitance sensing circuit 202 and convert that raw data into digital values, etc. In another embodiment, the CPU core 206 can transmit the raw data to the host processing unit 240 for processing.

[0025] In one embodiment, the host processing unit 240 may represent the processing unit of one or more machines such as a desktop computer, laptop computer, other types of portable computers, portable devices, set-top boxes (STBs), personal digital assistants (PDAs), servers, network routers, switches, or bridges. In one embodiment, the processing unit 200 is a PSoC® processing unit provided by Cypress Semiconductor Corporation (San Jose, California), as described in relation to Figure 17. Alternatively, the processing unit 200 may be another type of processing unit as understood by those skilled in the art who are interested in this disclosure. Alternatively, the capacitance sensing circuit 202 is incorporated into the host processing unit 240. Furthermore, although only a single machine is illustrated with respect to the host processing unit 240, the term “machine” should also be understood to include any set of machines that individually or collectively execute one or more instruction sets for implementing any one or more of the methodologies discussed herein. In other embodiments, the electronic system may include one or more host processing units 240, the processing unit 200, the capacitance sensing circuit 202, and one or more peripheral devices.

[0026] Figure 3A is a block diagram of the capacitance detection circuit 202 of Figure 2 according to one embodiment. The capacitance detection circuit 202 includes a first signal source 302, a second optional signal source 304, a reference signal source 303, a common-mode circuit 306, a reverse-mode circuit 308, a selection circuit 310, a detection circuit 312, and an analog-to-digital converter (ADC) 314. The common-mode circuit 306 can receive a signal from the first signal source 302 and generate a common-mode signal to be applied to one or more TX electrodes of the capacitance detection device 100. The reverse-mode circuit 308 can receive the same signal from the signal source 302 and generate a reverse-mode signal to be applied to one or more TX electrodes of the capacitance detection device 100. The reference signal source 303 can be derived from ground potential or other voltage potentials.

[0027] The capacitance sensing circuit 202 can be implemented in either the analog or digital domain. In one embodiment, the common-mode circuit 306 is an analog circuit having an amplifier 307 with a gain coefficient of +1. The amplifier 307 can be used to generate a common-mode signal. In this embodiment, the phase-reverse circuit 308 is also an analog circuit having an amplifier 309 with a gain coefficient of -1. The amplifier 309 is used to generate a phase-reverse signal. In another embodiment, an optional signal source 304 can be used to generate a second signal that is already out of phase with the first signal. In this embodiment, the phase-reverse circuit 308 does not need to use an amplifier with a gain coefficient of -1. Alternatively, three or more signal sources can be used to generate multiple signals, including at least a common-mode signal, one or more phase-reverse signals, and a reference signal. In these embodiments, signal sources 302 and 304 generate analog signals to be received by the common-mode circuit 306 and the phase-reverse circuit 308. Signal sources 302 and 304 may be passive components that generate one or more carrier signals having one or more frequencies. In other embodiments, the signal sources 302 and 304 may be components controlled externally, for example, by the CPU core 206 of the processing unit 200. It should also be noted that the common-mode circuit 306 does not have to include an amplifier, but other analog circuits can be used to generate the common-mode signal, as will be understood by those skilled in the art who are interested in this disclosure.

[0028] In another embodiment, the common-mode circuit 306 may include a digital circuit or digital processing logic that receives a first signal from the signal source 302 and is configured to generate a common-mode signal using, for example, a digital buffer. In this embodiment, the inverting-mode circuit 308 includes an inverter that receives the same signal from the signal source 302 and inverts this signal to generate an inverting-mode signal. In another embodiment, a second optional signal source 304 can be used to generate a second signal, which can be used for the inverting-mode signal as described above. In these embodiments, the signal sources 302 and 304 generate digital signals to be received by the common-mode circuit 306 and the inverting-mode circuit 308. For example, the signal source 302 can generate a digital sequence (e.g., a pseudo-random sequence) to be applied to the common-mode circuit 306 and the inverting-mode circuit 308. Alternatively, the signal source 302 can provide a modulated digital signal. In another embodiment, signal sources 302 and 304 generate analog signals, and common-mode circuit 306 and inverse-mode circuit 308 include analog-to-digital converters that convert analog signals to digital signals. As described above, signal sources 302 and 304 may be passive components that generate one or more carrier signals having one or more frequencies. In another embodiment, signal sources 302 and 304 may be components that are externally controlled, for example, by the CPU core 206 of the processing unit 200.

[0029] In the illustrated embodiment, the selection circuit 310 receives common-mode and out-of-mode signals and applies the appropriate signal to the TX electrode of the capacitance sensor 100. The selection circuit 310 is externally controllable, for example, by one or more control signals 318 received from the CPU core 206. For example, a phase modulation function selection routine can be executed on the CPU core 206, and this routine uses the control signals 318 to control the selection circuit 310. Alternatively, the selection circuit 310 can be used to control which TX electrode receives the common-mode signal, which TX electrode receives the out-of-mode signal, and which TX electrode receives the reference signal. In the illustrated embodiment, three signals are provided: one common-mode signal, one out-of-mode signal, and one reference signal. In other embodiments, three or more signals may be provided: one common-mode signal, multiple out-of-mode signals, and one or more reference signals. The reference signal (0) can be interpreted as both nodes of a capacitor (e.g., a sensor) having the same potential, and therefore no current flows into the RX channel for measurement. This is the opposite of the case where current flows through the capacitor to the RX channel, which may be true for the measured common-mode or out-of-mode signal (+1 or -1).

[0030] In one embodiment, the selection circuit 310 includes one or more multiplexers, which are controllable to select the appropriate signal for the appropriate TX electrode. In one embodiment, the selection circuit 310 is configured to select one of the alternating TX electrodes to receive a common-phase signal and the other of the alternating TX electrodes to receive a reverse-phase signal. Alternatively, the selection circuit 310 can drive the TX electrodes in other patterns described herein (e.g., excitation sequences) and other patterns that can be understood by those skilled in the art who are interested in this disclosure.

[0031] In the illustrated embodiment, the selection circuit 310 is also coupled to the detection circuit 312. The detection circuit 312 is configured to receive a detection signal from the RX electrode of the capacitance detection device 100. The detection signal results from a zero-sum excitation sequence. The detection signal represents the mutual capacitance or self-capacitance at each intersection of the panel electrodes. In one embodiment, the detection circuit 312 includes an analog circuit, a digital circuit, or a digital processing logic configured to measure capacitance. For example, in one embodiment, the detection circuit 312 includes an analog circuit for measuring the current associated with each of the RX electrodes. The detection circuit 312 provides the measured value to the ADC 314, which is to be converted to a digital value 316. The capacitance detection circuit 202 provides the digital value to the CPU core 206 or the host processing unit 240, as described above.

[0032] In one embodiment, the capacitance sensing circuit 202 is configured to drive the TX electrode 104 using phase modulation function selection. For example, the coefficients of the phase matrix (F), also called the excitation matrix (H), can be selected in several ways to form a non-singular matrix. Several criteria are available for phase modulation function selection: 1) The matrix must be a well-defined matrix that obtains a stable set of linear solutions. 2) Since the matrix coefficients determine the filtering characteristics of the system, the matrix can be selected to obtain the best filtering characteristics. 3) The receiver input current can be minimized in "no-touch" mode. 4) If the receiver has a limited dynamic range, it would be desirable to have a receiver current that is close to constant for multiple different scanning stages.

[0033] Figure 3B is a block diagram of a sequence generation circuit 350 according to one embodiment. The sequence generation circuit 350 includes a programmable frequency divider 352, a shift register 354, and an array of XOR gates 356 used for binary phase shifting of the TX signal. In the initialization phase, a sequence (e.g., an excitation sequence) is loaded into the shift register 354. A clock pulse is generated for the shift register 354 every n cycles, and the shift register 354 shifts the sequence at its output. Each register output is connected to a second XOR gate input. A common clock signal is applied to the first XOR gate input. Depending on the state of the register outputs, the XOR gates pass the clock signal with or without inversion to form different polyphase output signals.

[0034] Several methods exist for filling the phase matrix. In one embodiment, the selection function uses a pseudo-random sequence of +1, -1 and rotates this sequence through multiple scanning stages to completely fill the matrix. In some embodiments, the sequence can be selected such that its sum is zero. In this embodiment, included for illustrative purposes herein, The sequence has a sum of -2. For example, if M=10, selecting the following sequence:

number

number

number

[0035] F V These can be selected so that different operating phases can be easily distinguished. Using phase modulation sequences approximates a pseudo-random process, and the cyclical shift in process time makes it possible to obtain phase modulation sequences that have the smallest cross-correlation function and provide a stable solution to linear equation (1). Therefore, the cross-correlation coefficients between different phase modulation sequences should be significantly smaller than the peaks of the autocorrelation function. When using cyclic rotation of phase modulation sequences, the autocorrelation function of this sequence should have the smallest side peaks and the largest signal peaks.

[0036] Figures 4A and 4B are schematic diagrams of the detection grid 402 of a capacitance detection device 400 according to one embodiment. The detection grid 402 and the capacitance detection device 400 may be the same as or similar to the detection grid 102 and capacitance detection device 100 in Figure 1. As shown, the capacitance detection device 400 has eight TX electrodes 404(1) to 404(8). Each TX electrode 404 may have a value of positive 1, a value of negative 1, or a value of 0, respectively, corresponding to an in-phase drive signal, an out-of-phase drive signal, or a reference signal, according to a given excitation sequence. The excitation sequence may refer to values ​​sequentially assigned to each TX electrode. For example, in Figure 4A, the excitation sequence is {-1,+1,0,0,0,0,0,0}, while in Figure 4B, the excitation sequence is {-1,+1,+1,-1,+1,-1,+1,-1}. In either case, the sum of the excitation sequences is zero.

[0037] To minimize electromagnetic emission, the TX electrodes (e.g., 104 and 404) should be excited such that the sum of their excitation sequences is zero. In open space, each TX electrode of a capacitive touch panel can emit electromagnetic waves with the same power. Therefore, minimizing electromagnetic emission can be achieved by exciting the TX electrodes such that the excitation of the in-phase drive signal cancels out the excitation of the out-of-phase drive signal. In some embodiments, electromagnetic emission can be proportional to the absolute value of the sum of the excitation sequences and is therefore minimized when the sum of the excitation sequences is zero. It should be noted that making the sum of the excitation sequences zero is achievable by a combination of positive 1, negative 1, and 0, as shown in Figure 4A, or by a combination of only positive 1 and negative 1, as shown in Figure 4B. In some cases, the specific sequence used may affect the SNR and the emitted emission. It should also be noted that a capacitive sensing device can have an integer number of TX electrodes, and any suitable combination of positive 1, negative 1, and 0 can be used in the excitation sequence. In particular, in some cases, the capacitance sensing device may have an odd number of TX electrodes and still have an excitation sequence that has a sum of zeros.

[0038] In some embodiments, electromagnetic emissions may not be identical for each TX electrode due to specific hardware configurations and ground locations (e.g., ground potential or reference potential) in consumer electronics or automotive applications. As a result, even if the sum of the excitation sequences is zero, there may be cases where electromagnetic emissions are not exactly zero. In some embodiments, the optimal excitation sequence depends on the hardware configuration and the location of the specific TX electrode relative to the ground. Particularly in automotive applications, the ground is often located below the sensor, close to the lower TX electrode. In such cases, an optimal configuration for reducing electromagnetic emissions may have two upper electrodes excited by a common-phase drive signal (e.g., positive 1) and three lower electrodes excited by an out-of-phase drive signal (e.g., negative 1).

[0039] In automotive applications, a small electromagnetic emission gradient may exist between the upper and lower TX electrodes. For example, in the first scanning stage, adjacent TX electrodes may emit substantially the same electromagnetic force, but not exactly the same. This can result in a small electromagnetic emission gradient between adjacent TX electrodes, which can lead to very low electromagnetic emission during differential scanning (e.g., scanning of two electrodes). However, in another scanning stage, if the excitation sequence is cyclic, the upper TX electrode (excited by the in-phase drive signal) can be differentially scanned by the lower TX electrode (excited by the out-of-phase drive signal), which can result in a large electromagnetic emission gradient and, consequently, strong electromagnetic emission. In such cases, some scans, especially those with large gradients, can be omitted to minimize electromagnetic emission.

[0040] Other applications may have various hardware configurations and ground positions, and therefore, performing scans across all TX electrodes (with excitation sequences having a sum of zeros) can support the mitigation of large electromagnetic emission gradients and minimize electromagnetic emission. Similarly, some scans can be omitted to further reduce electromagnetic emission. Partial scanning (e.g., scanning only some TX electrodes) and excitation of specific TX electrodes (e.g., exciting only some TX electrodes) may result in ambiguity as further described herein.

[0041] Figures 5A and 5B compare the electromagnetic emission of a capacitance sensor according to each embodiment, with the sum of the excitation sequence being 1 and with the sum being 0. Figure 5A shows a capacitance sensor having an excitation sequence of {+1,+1,+1,-1,+1,+1,-1,+1,-1,-1,-1} and a sum of +1. As illustrated by the graph, the electromagnetic emission of the capacitance sensor exceeds the required limits for automotive applications. Figure 5B shows a capacitance sensor like the capacitance sensor 400 in Figure 4, having an excitation sequence of {-1,+1,0,0,0,0,0,0,0,0,0} and a sum of 0, as in Figure 4A. As illustrated by the graph, the electromagnetic emission of the capacitance sensor is within the required limits for automotive applications.

[0042] Figure 6 is a schematic diagram of the detection grid 402 of the capacitance detection device 400 of Figure 4, according to one embodiment. Figure 6 includes symbols for the excitation matrix (H) 601, the finger response (x) 603, and the measured signal (s) 605. As an exemplary example, the excitation sequence illustrated in Figure 6 is {+1,+1,+1,-1,+1,-1,-1,-1}, which corresponds to applying a drive signal to the TX electrode 404 in the first scanning stage. For each subsequent scanning stage, the excitation sequence can be rotated cyclically until each rotation results in the initial sequence. Each subsequent scanning stage can add an additional column to the excitation matrix H, so that the excitation matrix has dimensions L × M, where M is the number of TX electrodes and L is the number of rotations. In some embodiments, L may be the same as M, resulting in a square matrix; however, in other embodiments, L may be different from M, resulting in a non-square matrix. The excitation matrix H can represent the excitation pattern of drive signals (in-phase drive signal, out-of-phase drive signal, and reference signal) for exciting the TX electrodes. The finger response x may be a matrix corresponding to an electrode map, in which case, for example, when a user's finger interacts with a capacitive sensing touch panel, several electrodes may receive signals from the touch. The finger response x may have dimensions M × N, where N corresponds to the number of RX electrodes receiving sensing signals corresponding to the touch. The measured signal s may be a matrix corresponding to a measured sensing signal, such as one measured by scanning the RX electrodes. The measured signal s may have dimensions L × N. The measured signal s is the result of multiplying the excitation matrix H by the finger response x: H·x=s (4) That is the case.

[0043] For a capacitive sensing device, the finger response x should be determined so that the user can interact with the capacitive sensing touch panel. The excitation matrix H is predetermined or pre-calculated and therefore known. Furthermore, when the user interacts with the capacitive sensing touch panel, the sensing signal can be measured (e.g., the measured signal s is known). By inversion, the finger response x is determined. x=H -1 ·s (5) It can be decided in this way.

[0044] In some embodiments, the excitation matrix H can be inverted when the sum of the excitation sequences is not zero (e.g., 1 or 2), and the finger touch x can be reconstructed. However, this may result in relatively large electromagnetic emission. Pseudo-inverse matrix H -1 It can have dimensions of M × L.

[0045] If the sum of the excitation sequence is zero, electromagnetic emission can be made smaller, but the excitation matrix H is singular (e.g., has an infinite number of roots) and therefore invertible. In such cases, equation (5) can be narrowed down to a partial decomposition and solved for the case where touch is present (but not where touch is absent or where all sensors are covered) by finding a suitable pseudoinverse of the excitation matrix. In one embodiment, the pseudoinverse is the Moore-Penrose pseudoinverse. In other embodiments, the pseudoinverse may be one of the following: a one-sided inverse (right or left inverse), a Bott-Duffin inverse, a Drazin inverse, or a reflective generalized inverse.

[0046] To locate the touch, an analytical ambiguity reduction method (process) can be applied. One potential drawback of the analytical ambiguity reduction method is that common-mode levels may be lost. However, by scanning one or more regions such that each sequential scanning stage includes at least one common TX electrode, common-mode levels can be matched to form a baseline. The analytical ambiguity reduction method is described below in relation to Figure 7.

[0047] Figure 7 is a schematic diagram of a detection grid 702 of a capacitance detection device 700 according to one embodiment, having a first scanning region 701, a second scanning region 703, and an overlapping region 705. The detection grid 702 and the capacitance detection device 700 may be the same as the detection grids 102, 402, 702, and 802 and the capacitance detection devices 100, 400, 700, and 800 in Figures 1, 4, and 7-8. The overlapping region 705 represents a region in which one or more TX electrodes are included in both the first scan in the first scanning stage and the second scan in the second scanning stage.

[0048] In one case (not shown in Figure 7), the first and second scanning stages do not contain common electrodes (e.g., overlap of TX electrodes in the first and second scanning stages). For example, in the first scanning stage, TX electrodes 704(1) to 704(4) can be simultaneously excited by a common-mode drive signal, an out-of-mode drive signal, or a reference signal according to a zero-sum excitation sequence. In the second scanning stage, TX electrodes 704(5) to 704(8) can be simultaneously excited by a common-mode drive signal, an out-of-mode drive signal, or a reference signal according to a rotated zero-sum excitation sequence. Subsequent scanning stages can be excited by a common-mode drive signal, an out-of-mode drive signal, or a reference signal according to further rotation of the zero-sum excitation sequence. In this case, the common-mode level corresponding to the baseline across multiple scanning stages may be lost due to the singularity of the excitation matrix H, resulting in a discontinuous baseline.

[0049] To recover the common-mode level and a continuous baseline, the first and second scanning stages should include at least one common TX electrode (e.g., an overlap of TX electrodes in both the first and second scanning stages). For example, in the first scanning stage, a first scan can be performed including a first scanning region 701. The first scanning region 701 includes a first set of TX electrodes 704(1) to 704(4). In the second scanning stage following (or following) the first scanning stage, a second scan can be performed including a second scanning region 703. The second scanning region 703 includes a second set of TX electrodes 704(4) to 704(7). The first set of TX electrodes and the second set of TX electrodes have a common TX electrode 704(4) in an overlapping region 705. In some embodiments, the overlapping region 705 includes a single common TX electrode. In some other embodiments, the overlapping region 705 may include two or more common TX electrodes. Furthermore, although the scanning regions 701 and 703 are illustrated in Figure 7 as including four TX electrodes, in other embodiments, the scanning regions may include fewer than four or five or more TX electrodes. As will be described in more detail in relation to Figure 8, such overlapping scanning methods can be used to align common mode levels and form a continuous baseline.

[0050] Figure 8 is a graph showing baseline correction across the TX electrode scanning area of ​​the detection grid 802 of the capacitance detection device 800 according to one embodiment. The detection grid 802 and the capacitance detection device 800 may be the same as the detection grids 102, 402, and 702 and the capacitance detection devices 100, 400, and 700 in Figures 1 to 7.

[0051] As illustrated in Figure 8, a first scan including a first scanning region 801 includes a first set of TX electrodes 804. The first detection signal resulting from the first scan may have a first common-mode level. A second scan performed following the first scan includes a second scanning region 803 including a second set of TX electrodes 804. The second detection signal resulting from the second scan may have a second common-mode level different from the first common-mode level. In one embodiment, the first common-mode level can be considered a reference for the baseline. In other words, the common-mode levels measured on the overlap 805 of the TX electrodes by the RX electrodes 806(1) to 806(3) may be s1 = {s11, s12, s13}, respectively. The second common-mode levels measured on the overlap 805 of the TX electrode by the RX electrodes 806(1) to 806(3) may be s2 = {s21, s22, s23}, respectively. The difference between the first common-mode level and the second common-mode level can be determined as ds12 = s1 - s2 = {s11 - s21, s12 - s22, s13 - s23}. The difference value ds12 can be added to the second common-mode level, i.e., s2 + ds12 = {s21, s22, s23} + {s11 - s21, s12 - s22, s13 - s23} = {s11, s12, s13}, and applied to the data obtained from the measurement of the second scanning area 803 (e.g., TX slot 2) to achieve a shifted second common-mode level equal to the first common-mode level, thereby establishing a continuous baseline between the first and second scanning stages. A similar operation can be performed for subsequent scanning stages to achieve a single common-mode level for each scanning stage. In another embodiment, instead of adding the difference value to the second common-mode level, the difference value can be subtracted from the first common-mode level. In another embodiment, the first and second common-mode levels can be averaged between each scanning stage to achieve a single common-mode level and thus a continuous baseline.

[0052] Figure 9 is a flowchart of one embodiment of a method for multiphase scanning. In some embodiments, processing logic can be used to carry out method 900. The processing logic can include hardware, software, or any combination thereof. In one embodiment, the processing unit 200 of Figure 2 carries out method 900. In another embodiment, the capacitance sensing circuit 202 of Figure 2 carries out method 900. Alternatively, other components may be used to carry out some or all of the operation of method 900. In block 902, the processing logic reads a zero-sum excitation sequence for the TX line signal from a table and loads the signal generation parameters into the TX signal generator. In block 904, the processing logic applies the zero-sum TX signal sequence to the TX electrode and detects the 1-sensor overlap response from the receiver electrode substantially simultaneously. In block 906, once the conversion stage is complete, the processing logic performs analog-to-digital conversion of the received response. The conversion result is stored in a data buffer (block 908). In block 910, the rotation state is checked. Once all rotations of the zero-sum excitation sequence are complete, the excitation matrix H can be determined. Since the excitation sequence has a sum of zeros, the excitation matrix H is singular. In this case, the analytical ambiguity reduction method (process) is initiated in block 912, and method 900 is completed. The analytical ambiguity reduction method is further explained in relation to Figure 10. In the reverse case, the excitation sequence is rotated in block 909, and blocks 902-910 are performed again.

[0053] It should be noted that in other embodiments of the multiple embodiments of the present invention, the implementation flow may be modified. For example, in some embodiments of the present invention, blocks 902 and 904 can be implemented in parallel by using multiple analog-to-digital conversion channels. There are various possible arrangements for the deconvolution procedure. Given that the analytical ambiguity reduction method may involve multiple iterations, a rearrangement into multiple steps is possible, and each step can be implemented once the results of each conversion stage are supplied. Since the analytical ambiguity reduction method can be implemented in parallel with the scan, this implementation provides an advantage in terms of overall execution time.

[0054] In one embodiment, the processing logic generates a common-mode drive signal and a reverse-mode drive signal. The processing logic selects a zero-sum excitation sequence having a plurality of positive 1s, negative 1s, and 0s corresponding to the common-mode drive signal, the reverse-mode drive signal, and the reference signal, respectively. The reference signal originates from a voltage potential or ground potential. The zero-sum excitation sequence has a sum equal to zero. During the first scan stage of a set number of scan stages, the processing logic applies one of the common-mode drive signal, the reverse-mode drive signal, or the reference signal to each of the first set of TX electrodes at substantially the same time (e.g., simultaneously) according to the zero-sum excitation sequence. The processing logic rotates the zero-sum excitation sequence to obtain a rotated zero-sum excitation sequence. During the second scan stage of a set number of scan stages, the processing logic applies one of the common-mode drive signal, the reverse-mode drive signal, or the reference signal to each of the second set of TX electrodes at substantially the same time (e.g., simultaneously) according to the rotated zero-sum excitation sequence. The processing logic receives a detection signal from a set of RX electrodes to detect the presence of an object on the touch panel. The detection signal represents the capacitance associated with the set of RX electrodes.

[0055] In one embodiment, the TX electrode and the RX electrode are the same electrode. In this case, the detection signal represents the self-capacitance of the RX electrode set. In another embodiment, the TX electrode and the RX electrode are different electrodes. In this case, the detection signal represents the mutual capacitance at the intersection of the TX electrode and the RX electrode.

[0056] In some embodiments, there is an odd number of TX electrodes, and the zero-sum excitation sequence includes one or more positive values ​​of 1, one or more negative values ​​of 1, and one or more values ​​of 0, and the sum of the zero-sum excitation sequence is zero (for example, equal to zero).

[0057] In one embodiment, the processing logic uses binary phase modulation to implement a method for receiving or generating a carrier signal as an in-phase or out-of-phase drive signal. The processing logic applies a carrier signal or reference signal to a first set of TX electrodes according to a zero-sum excitation sequence.

[0058] In another embodiment, the processing logic selects a pattern of zero-sum excitation sequences for the TX electrodes and drives each of the TX electrodes according to this pattern with either a common-mode drive signal, a reverse-mode drive signal, or a reference signal. In another embodiment, the processing logic selects a pseudo-random number sequence having a plurality of "1" values, "negative 1" values, and "0" values. The "1" values ​​correspond to the common-mode drive signal, the "negative 1" values ​​correspond to the reverse-mode drive signal, and the "0" values ​​correspond to the reference signal, and the number of values ​​in the pseudo-random number sequence (with a sum of zeros) is equal to the number of TX electrodes. The processing logic applies either a common-mode drive signal, a reverse-mode drive signal, or a reference signal to the TX electrodes according to the pseudo-random number sequence. Once this is done in one scanning stage, the processing logic then rotates the pseudo-random number sequence in subsequent scanning stages and applies either a common-mode drive signal, a reverse-mode drive signal, or a reference signal to the TX electrodes according to the rotated pseudo-random number sequence in subsequent scanning stages.

[0059] In one embodiment, the processing logic applies an in-phase drive signal to one of the alternating TX electrodes and substantially simultaneously applies an out-of-phase drive signal to the other of the alternating TX electrodes. Alternatively, the processing logic may apply one or more in-phase drive signals, one or more out-of-phase signals, and one or more reference signals to the TX electrodes in other patterns that can be understood by those skilled in the art who are interested in the present disclosure.

[0060] In the embodiments described herein, a multiphase TX driving method (i.e., multiphase scanning) makes it possible to improve noise immunity to white noise and narrowband noise by several times without increasing the TX voltage. Multiphase scanning can provide the same improvement in noise immunity to noise caused by liquid crystal displays (LCDs). This noise improvement can be increased with increasing M, which is the number of panel transmitter electrodes driven substantially simultaneously, by approximately M 1 / 2 It improves. Multiphase scanning has a computational requirement (N*M for each scan). 2 It should be noted that the MAC operation (of the TX electrode) can be relatively high. Multiphase scanning can also reduce power consumption by driving the TX electrode substantially simultaneously, and can also reduce the current of the RX electrode. If special phase sequencing and decoding techniques are used, the MAC operation may only require addition and subtraction to perform.

[0061] Figure 9 illustrates the application of three signals to the TX electrode array, and it should be noted that all RX electrodes can be detected substantially simultaneously. In other embodiments, signals with various phases, amplitudes, or frequencies can be applied substantially simultaneously to two or more TX electrodes, such as applying an in-phase signal to one of the alternating TX electrodes and an out-of-phase signal to the other of the alternating TX electrodes. In other embodiments, as can be understood by those skilled in the art who are interested in this disclosure, two or more receiver signals can be detected substantially simultaneously or sequentially.

[0062] FIG. 10 is a flowchart of an analytical ambiguity reduction method 1000 for reconstructing finger responses on a capacitance sensing touch panel according to an embodiment. The method 1000 can be implemented by processing logic including hardware, firmware, or any combination thereof. The method 1000 can be implemented by the capacitance sensing device 100 of FIG. 1. In another embodiment, the method 1000 can be implemented by the processing device 200 and / or the capacitance sensing circuit 202 of FIG. 2. In another embodiment, the method 1000 can be implemented by the capacitance sensing devices 400, 700, or 800 of FIGS. 4 or 6-8.

[0063] Returning to FIG. 10, at block 912 of FIG. 9, the analytical ambiguity reduction method 1000 begins. The method 1000 begins with processing logic to find a convolution inverse circulant matrix (block 1002). The convolution inverse circulant matrix H -1is the pseudo-inverse of the excitation matrix H, which is singular due to the zero-sum excitation sequence having a sum of zeros. The processing logic calculates the deconvolved data (block 1004). The deconvolved data refers to calculating the finger response x as a deconvolved cyclic matrix multiplied by the measurement signal s, for example, measured by the RX electrode, using equation (5). The processing logic applies a common-mode filter to the deconvolved data on the TX electrode line to obtain filtered data (block 1006). The common-mode filter can block high-frequency noise that may be common to two or more electrodes, but still allows signals of the desired frequency to pass through. The processing logic corrects the baseline in the filtered data by merging overlapping scan regions (block 1008). The first scan stage scans over a first region containing a first number of TX electrodes, and the second scan stage scans over a second region containing a second number of TX electrodes. In some cases, a first scan across a first region may have a first common-mode level that sets a baseline for the sensing signal from the first region. A second scan across a second region may have a second common-mode level that sets a baseline for the sensing signal from the second region, but the first common-mode level and the second common-mode level may not be the same, and therefore the baselines of the first region and the second region may be different, causing a discontinuity in the baseline between the two regions. To solve this problem, the first and second regions can be scanned with overlap, which means that the first and second regions share a common TX electrode. A difference value can be calculated as the difference between the first common-mode level and the second common-mode level, and this difference value can be added to the second common-mode level to achieve a continuous baseline. Similar steps can be included for subsequent scanning stages as needed. The processing logic applies an orthogonal common-mode filter to the RX electrode line (block 1010).A quadrature common-mode filter is the same as a common-mode filter, but it is applied to the RX electrode line instead of the TX electrode line. Method 1000 is now complete.

[0064] Figure 11A is a capacitance sensor matrix 1100 showing deconvolved data for multiphase transmission using eight electrodes excited by a zero-sum excitation sequence according to one embodiment. The capacitance sensor matrix 1100 shows both the deconvolved data and LCD noise. Figure 11A corresponds to block 1004 in Figure 10.

[0065] Figure 11B shows a capacitance sensor matrix 1100 with filtered data obtained by applying a common-mode filter according to one embodiment. A common-mode filter blocks high-frequency noise that may be common to two or more electrodes, while still allowing signals of the desired frequency to pass through. As illustrated in Figure 11B, the common-mode filter filters out a number of values ​​to read zero. Figure 11B corresponds to block 1006 in Figure 10.

[0066] Figure 11C shows the capacitive sensor matrix 1100, which, according to one embodiment, shows filtered data after the common-mode values ​​have been restored along the set of RX electrodes to correct the baseline. Figure 11C corresponds to block 1008 in Figure 10.

[0067] Figure 11D shows a capacitance sensor matrix 1100 with filtered data after applying an orthogonal common-mode filter according to one embodiment. The orthogonal common-mode filter removes artifacts along the set of RX electrodes to obtain a capacitance map of the touch panel. Figure 11D corresponds to block 1010 in Figure 10. As illustrated by Figure 11D, the capacitance map clearly maps finger touches on the touch panel.

[0068] Figure 12A is a graph comparing the relationship between noise reduction coefficients and peak emission for three methods of multiphase transmission according to one embodiment. The three methods of multiphase transmission are as follows: 1) MPTX11 Sum1, which refers to multiphase transmission using 11 TX electrodes and an excitation sequence with a sum of 1s; 2) MPTX8 Sum0, which refers to multiphase transmission using 8 TX electrodes and an excitation sequence with a sum of zeros (this is the method described herein); and 3) Differential MPTX Sum0, which refers to multiphase transmission using 2 electrodes and an excitation sequence with a sum of zeros. A relatively large noise reduction coefficient corresponds to a relatively large desirable SNR. As mentioned above, a relatively low peak emission (electromagnetic emission) is desirable. It should be noted that the emission is measured with respect to the fundamental frequency. As illustrated in Figure 12A, MPTX11 Sum1 provides the best SNR but also provides the largest peak emission. The peak emission for MPTX11 Sum1 exceeds the electromagnetic emission limit. MPTX8 Sum0 provides a noise reduction factor of approximately 2 and a peak emission of approximately 41 dBμV / m. It should be noted that a noise reduction factor of 2 means, for example, that white noise is reduced by half. Differential MPTX Sum0 has the lowest peak emission but also the lowest noise reduction factor and therefore the lowest SNR. The SNR for differential MPTX Sum0 may be too low for actual touchscreen applications. Therefore, multiphase excitation and multiphase scanning with zero-sum excitation sequences for touch panels described herein provide both a good SNR and sufficiently low peak emission.

[0069] Figure 12B is a graph comparing the relationship between the noise reduction coefficient and mean emission for the three methods described in Figure 12A. Here again, multiphase excitation and multiphase scanning using the zero-sum excitation sequence (MPTX8 Sum0) provide both a good SNR and a sufficiently low mean emission.

[0070] Figure 13 is a series of schematic diagrams showing a zero-sum polyphase receiver 1300 according to one embodiment. Zero-sum polyphase reception uses the same or similar ideas as zero-sum polyphase transmission, including analytical ambiguity reduction methods similar to those used in zero-sum polyphase transmission. It should be noted that touch cannot be detected if all sensors are covered. In the embodiment illustrated in Figure 13, four receiving sensors are provided, but in other embodiments, more or fewer sensors may be provided.

[0071] The zero-sum polyphase receiver 1301 may include a zero-sum receiving sequence corresponding to touch or capacitance from the capacitors 1320 that constitute the sensor array. The receiving sequence shown in 1301 is {+1,+1,-1,-1} (with a sum of zeros), and this receiving sequence corresponds to the reception of a common-mode signal from capacitor 1320(1), a common-mode signal from capacitor 1320(2), an out-of-mode signal from capacitor 1320(3), and an out-of-mode signal from capacitor 1320(4). The signals corresponding to the receiving sequence can be transmitted to ABusB line 1324 and ABusA line 1325 and can be received by an RX signal receiver such as the RX signal receiver 114 in Figure 1.

[0072] Similarly, zero-sum polyphase receiver 1303 includes a zero-sum receiving sequence corresponding to touch or capacitance from capacitors 1320 constituting the sensor array. The shown receiving sequence is {-1,+1,+1,-1} (having a sum of zeros), and this receiving sequence corresponds to the reception of an out-of-phase signal from capacitor 1320(1), a common-phase signal from capacitor 1320(2), a common-phase signal from capacitor 1320(3), and an out-of-phase signal from capacitor 1320(4). Zero-sum polyphase receiver 1305 has a receiving sequence of {-1,+1,-1,+1}, and this receiving sequence corresponds to the reception of an out-of-phase signal from capacitor 1320(1), a common-phase signal from capacitor 1320(2), an out-of-phase signal from capacitor 1320(3), and a common-phase signal from capacitor 1320(4). The zero-sum polyphase receiver 1307 has a receiving sequence of {+1,-1,-1,+1}, which corresponds to the reception of a common-mode signal from capacitor 1320(1), an out-of-mode signal from capacitor 1320(2), an out-of-mode signal from capacitor 1320(3), and a common-mode signal from capacitor 1320(4).

[0073] Figure 14 is a graph showing a noise comparison between a detection signal received with zero-sum polyphase reception and a detection signal received without zero-sum polyphase reception, according to one embodiment. In one case, an external noise source is applied and the detection signal is received via the RX electrode. In the case of curve 1401, zero-sum polyphase reception is not applied, and in the case of curve 1403, zero-sum polyphase reception is applied. At the fundamental frequency to which zero-sum polyphase is applied, the noise can be reduced to about 1 / 20th by reception, thereby demonstrating excellent electromagnetic immunity.

[0074] It is worth noting that multiphase RX and multiphase TX may be used together, or either multiphase RX or multiphase TX may be used separately. Both multiphase RX and multiphase TX use similar algorithms (e.g., analytical ambiguity reduction methods) to reconstruct the signal. Furthermore, the scanning of multiphase RX and multiphase TX can be applied in any order, and this order can be selected based on implementation flexibility or optimization requirements. With multiphase TX / RX technology, it is possible to achieve a theoretical value of an equivalent integration time comparable to the panel scan time using only a single measurement channel.

[0075] Figure 15 is a flowchart of a zero-sum polyphase transmission method 1500 according to one embodiment. Method 1500 can be implemented by processing logic including hardware, firmware, or any combination thereof. Method 1500 can be implemented by the capacitance sensing device 100 of Figure 1. In another embodiment, Method 1500 can be implemented by the processing device 200 and / or capacitance sensing circuit 202 of Figure 2. In yet another embodiment, Method 1500 can be implemented by the capacitance sensing devices 400, 700, or 800 of Figure 4 or Figures 6-8.

[0076] Referring back to Figure 15, Method 1500 begins with processing logic that generates a common-mode drive signal and an out-of-mode drive signal (block 1502). The processing logic selects a sequence (block 1504). The sequence may have a number of positive 1s, negative 1s, and 0s corresponding to the common-mode drive signal, the out-of-mode drive signal, and the reference signal, respectively. The reference signal may originate from a voltage potential or ground potential. The sum of the sequences is zero. The processing logic applies at least one of the common-mode drive signal, the out-of-mode drive signal, or the reference signal according to the sequence. In the first scanning stage, the common-mode signal, the out-of-mode signal, and the reference signal are applied substantially simultaneously to a first set of TX electrodes (block 1506). The processing logic rotates the sequence to obtain the rotated sequence (block 1508). In the second scanning stage, the processing logic applies at least one of the common-mode drive signal, the out-of-mode drive signal, or the reference signal according to the rotated sequence (block 1510). The in-phase, out-of-phase, and reference signals are applied substantially simultaneously to a second set of TX electrodes. The processing logic receives a detection signal from the RX electrode set (step 1512). The detection signal may be for detecting the presence of an object on the touch panel, and the detection signal may represent the capacitance associated with the set of receiver electrodes of the touch panel. Method 1500 ends.

[0077] In some embodiments, the TX electrode and the RX electrode are the same electrode, and the detection signal represents the self-capacitance associated with the set of RX electrodes. In other embodiments, the TX electrode and the RX electrode are different electrodes, and the detection signal represents the mutual capacitance at the intersection between the TX electrode and the RX electrode.

[0078] In some embodiments, the number of TX electrodes is even, and the sum of sequences including one or more positive 1s, one or more negative 1s, and one or more 0s is zero. In other embodiments, the number of TX electrodes is odd, and the sum of sequences including one or more positive 1s, one or more negative 1s, and one or more 0s is zero.

[0079] In further embodiments, the first set of TX electrodes and the second set of TX electrodes include a common electrode, particularly for overlapping scanning. The first and second scanning stages may be sequential. The processing logic may implement analytical ambiguity reduction methods to align common-mode levels (e.g., between the first set of TX electrodes and the second set of TX electrodes) to form a baseline. Overlapping scanning can enable the formation of a continuous baseline between the first and second scanning stages.

[0080] The processing logic can implement an analytical ambiguity reduction method, which is described in relation to Figure 10. The processing logic determines an inverse convolution cyclic matrix, which is the pseudo-inverse of the excitation matrix, including at least the sequence and the rotated sequence. It is noteworthy that the excitation matrix is ​​a singular matrix due to the sum of the sequences being zero and the sum of the rotated sequences being zero. The processing logic uses the inverse convolution cyclic matrix and the sense signal to compute the inverse convolution data. The processing logic can apply a common-mode filter to the inverse convolution data to obtain filtered data. The processing logic can reconstruct common values ​​along the set of RX electrodes to correct the baseline in the filtered data. Correcting the baseline can refer to establishing common-mode values ​​between the first and second scanning stages to obtain a continuous baseline. The processing logic can apply an orthogonal common-mode filter to remove artifacts along the set of RX electrodes to obtain a capacitance map of the touch panel. A capacitance map refers to a direct image of a finger (or other conductive object) touching a touch panel.

[0081] Figure 16 is a flowchart of a zero-sum polyphase reception method 1600 according to one embodiment. Method 1600 can be implemented by processing logic including hardware, firmware, or any combination thereof. Method 1600 can be implemented by the capacitance detection device 100 of Figure 1. In another embodiment, Method 1600 can be implemented by the processing device 200 and / or capacitance detection circuit 202 of Figure 2. In yet another embodiment, Method 1600 can be implemented by the capacitance detection devices 400, 700, or 800 of Figure 4 or Figures 6-8.

[0082] Referring back to Figure 16, Method 1600 begins with processing logic that generates a common-mode drive signal and a reverse-mode drive signal (block 1602). The processing logic applies a signal to each of the TX electrodes in the set of TX electrodes (block 1604). The processing logic selects a sequence (block 1606). The sequence can have a number of positive 1s, negative 1s, and 0s corresponding to the common-mode drive signal, the reverse-mode drive signal, and the reference signal, respectively. The reference signal can originate from a voltage potential or ground potential. The sum of the sequences is zero. In the first stage of the set of scanning stages, the processing logic detects substantially simultaneously one of the common-mode drive signal, the reverse-mode drive signal, or the reference signal in accordance with the sequence at each of the first set of receiver electrodes. The processing logic rotates the sequence to obtain the rotated sequence. The processing logic, in the second stage of the scanning stage set, substantially simultaneously detects one of the following signals—in-phase drive signal, out-of-phase drive signal, or reference signal—at each of the second set of receiver electrodes according to a rotated sequence.

[0083] In some embodiments, the TX electrode and the RX electrode are the same electrode, and the detection signal represents the self-capacitance associated with a first set and a second set of RX electrodes. In other embodiments, the TX electrode and the RX electrode are different electrodes, and the detection signal represents the mutual capacitance at the intersection between the TX electrode and the RX electrode.

[0084] In some embodiments, the number of RX electrodes is even, and the sum of a sequence containing one or more positive 1s, one or more negative 1s, and one or more 0s is zero. In other embodiments, the number of RX electrodes is odd, and the sum of a sequence containing one or more positive 1s, one or more negative 1s, and one or more 0s is zero.

[0085] In further embodiments, the first set of RX electrodes and the second set of RX electrodes include a common electrode, particularly for overlapping scanning. The first and second scanning stages may be sequential. The processing logic may implement an analytical ambiguity reduction method to align the common-mode levels (e.g., between the first set of RX electrodes and the second set of RX electrodes) to form a baseline. Overlapping scanning can enable the formation of a continuous baseline between the first and second scanning stages.

[0086] The processing logic can implement an analytical ambiguity reduction method, which is described in relation to Figure 10. The processing logic determines an inverse convolution cyclic matrix, which is the pseudo-inverse of the excitation matrix, including at least the sequence and the rotated sequence. It is noteworthy that the excitation matrix is ​​a singular matrix due to the fact that the sum of the sequence and the sum of the rotated sequence are zero. The processing logic uses the inverse convolution cyclic matrix and the detection signal (digital data corresponding to the detection signal) to compute the inverse convolution data. The processing logic can apply a common-mode filter to the inverse convolution data to obtain filtered data. The processing logic can restore common values ​​along a first set and a second set of RX electrodes to correct the baseline in the filtered data. Correcting the baseline can refer to establishing common-mode values ​​between the first and second scanning stages to obtain a continuous baseline. The processing logic can apply an orthogonal common-mode filter to remove artifacts along the set of RX electrodes and obtain a capacitance map of the touch panel. A capacitance map refers to a direct image of a finger (or other conductive object) touching the touch panel.

[0087] Figure 17 shows an embodiment of the PSoC® processing unit core architecture 1700, such as that used in the PSoC3® product family provided by Cypress Semiconductor Corporation (San Jose, California). In one embodiment, the core architecture 1700 includes a microcontroller 1702. The microcontroller 1702 includes a CPU (Central Processing Unit) core 1704, flash program storage 1706, DOC (Debug-on-Chip) 1708, prefetch buffer 1710, private SRAM (Static Random Access Memory) 1712, and special function registers 1714. In one embodiment, the DOC 1708, prefetch buffer 1710, private SRAM 1712, and special function registers 1714 are coupled to the CPU core 1704, while the flash program storage 1706 is coupled to the prefetch buffer 1710.

[0088] The core architecture 1700 may also include a CHub (core hub) 1716, which includes a bridge 1718 and a DMA controller 1720, and is coupled to the microcontroller 1702 via a bus 1722. The CHub 1716 can provide a primary data and control interface between the microcontroller 1702 and its peripherals and memory and the programmable core 1724. In one embodiment, the capacitance sensing circuit 202 shown in Figure 3A can be implemented in the core architecture 1700 as part of the programmable core 1724. The DMA controller 1720 can be programmed to transmit data between system elements without burdening the CPU core 1704. In various embodiments, each of these subcomponents of the microcontroller 1702 and the CHub 1716 may differ depending on the respective selection or type of CPU core 1704. The CHub 1716 can also be coupled to a shared SRAM 1726 and an SPC (system performance controller) 1728. The private SRAM 1712 is independent of the shared SRAM 1726, which is accessed by the microcontroller 1702 via the bridge 1718. The CPU core 1704 can access the private SRAM 1712 without going through the bridge 1718, thereby enabling local register and RAM access to occur simultaneously with DMA access to the shared SRAM 1726. Although labeled as SRAM herein, these memory modules may be any suitable type from the wide variety of (volatile or non-volatile) memory or data storage modules in various other embodiments.

[0089] In various embodiments, the programmable core 1724 may include, but is not limited to, various combinations of subcomponents (not shown), including digital logic arrays, digital peripherals, analog processing channels, global routing analog peripherals, DMA controllers, SRAM and other suitable types of data storage, I / O ports and other suitable types of subcomponents. In one embodiment, the programmable core 1724 includes a GPIO (General Purpose I / O) and EMIF (Extended Memory Interface) block 1730 for providing a mechanism for extending off-chip access to the microcontroller 1702, a programmable digital block 1732, a programmable analog block 1734, and a special function block 1736, each of which is configured to perform the function of one or more subcomponents. In various embodiments, the special function block 1736 may include dedicated (non-programmable) function blocks and / or include one or more interfaces to dedicated function blocks such as USB, crystal oscillator drives, JTAG, etc.

[0090] The programmable digital block 1732 may include a digital logic array that includes an array of digital logic blocks and associated routing. In one embodiment, the digital block architecture consists of UDBs (Universal Digital Blocks). For example, each UDB may include an ALU along with CPLD functionality.

[0091] In various embodiments, one or more UDBs of the programmable digital block 1732 can be configured to perform a variety of digital functions, including, but not limited to, the following functions: basic I2C slave, I2C master, SPI master or SPI slave, multiwire (e.g., 3-wire) SPI master or slave (e.g., MISO / MOSI multiplexed on one pin), timers and counters (e.g., a pair of 8-bit timers or counters, a 16-bit timer or counter, an 8-bit capture timer, etc.), PWM (e.g., a pair of 8-bit PWMs, one 16-bit PWM, one 8-bit deadband PWM, etc.), level-sensitive I / O interrupt generators, quadrature encoders, UARTs (e.g., half-duplex), delay lines, and any other suitable types of digital functions or combinations of digital functions that can be implemented in multiple UDBs.

[0092] In other embodiments, groups of two or more UDBs can be used to implement additional functions. While not limiting, multiple UDBs can be used to implement the following functions: an I2C slave supporting hardware address discovery and the ability to process complete transactions without intervention from a CPU core (e.g., CPU core 1704) and to prevent forced clock stretching for any bit in the data stream; an I2C multimaster that can include slave options in a single block; PRS or CRC of any length (up to 32 bits); SDIO; SGPIO; a digital correlator (e.g., having up to 32 bits with 4x oversampling and supporting a configurable threshold); a LIN bus interface; a delta-sigma modulator (e.g., for a Class D audio DAC with differential output pairs); I2S (stereo); and LCD drive control (e.g., a UDB can control an LCD drive). It can be used to perform timing control of blocks and to provide addressing of display RAM); full-duplex UART (e.g., 7-bit, 8-bit or 9-bit with 1 or 2 stop bits and parity and RTS / CTS support); IRDA (transmit or receive); capture timer (e.g., 16-bit, etc.); deadband PWM (e.g., 16-bit, etc.); SMbus (including formatting of SMbus packets with CRC in software); brushless motor drive (e.g., to support 6 / 12 step commutation); automatic baud rate detection and generation (e.g., automatically determining the baud rate for standard rates of 1200 to 115200 baud and, after detection, generating the clock required to generate the baud rate); and any other suitable type of digital function or combination of digital functions that can be implemented in multiple UDBs.

[0093] The programmable analog block 1734 may include, but is not limited to, analog resources such as comparators, mixers, PGAs (programmable gain amplifiers), TIAs (transimpedance amplifiers), ADCs (analog-to-digital converters), DACs (digital-to-analog converters), voltage references, current sources, sample-and-hold circuits, and any other suitable types of analog resources. The programmable analog block 134 may support a variety of analog functions, but is not limited to, analog routing, LCD drive I / O support, capacitance sensing, voltage measurement, motor control, current-to-voltage conversion, voltage-to-frequency conversion, differential amplification, optical measurement, inductive position monitoring, filtering, voice coil driving, magnetic card reading, acoustic Doppler measurement, echo ranging, modem transmit and receive encoding, or any other suitable types of analog functions.

[0094] It should be noted that the embodiments described above use common-mode, out-of-phase, and reference signals. Common-mode and out-of-phase signals can be used when an inverter or complementary output stage is used to generate these signals. Common-mode and out-of-phase signals can also be used to simplify ADC measurements as +1 or -1 data codes. However, in other embodiments, other different arbitrary phase signals may be used. For example, a common-mode signal and one or more out-of-phase signals may be used.

[0095] The embodiments of the present invention described herein include a variety of operations. These operations can be implemented by hardware components, software, firmware, or a combination thereof. As used herein, the term “coupled to ~” may mean directly coupled or indirectly coupled through one or more intervening components. Any signal provided through the various buses described herein can be time-multiplexed with other signals and can be provided through one or more common buses. Furthermore, interconnections between circuit components or blocks may be shown as buses or single signal lines. Each bus may alternatively be one or more single signal lines, and each single signal line may alternatively be a bus.

[0096] Certain embodiments can be implemented as computer program products that include instructions stored on a computer-readable medium. These instructions can be used to program a general-purpose or dedicated processor to perform the operations described above. The computer-readable medium includes any mechanism for storing or transmitting information in a form readable by a machine (e.g., a computer) (e.g., software, processing applications). The computer-readable storage medium may include, but is not limited to, magnetic storage media (e.g., floppy diskettes), optical storage media (e.g., CD-ROMs), magneto-optical storage media, read-only memory (ROM), random access memory (RAM), erasable programmable memory (e.g., EPROM and EEPROM), flash memory, or other types of media suitable for storing electronic instructions. The computer-readable transmission medium may include, but is not limited to, other types of media suitable for transmitting electrical signals, optical signals, acoustic signals or other forms of propagating signals (e.g., carrier waves, infrared signals, digital signals, etc.) or electronic instructions.

[0097] Furthermore, some embodiments can be realized in a distributed computing environment in which a computer-readable medium is stored and / or executed by two or more computer systems. In addition, information transferred between computer systems can be pushed or pulled via a transmission medium connecting the computer systems.

[0098] Although the operations of the methods described herein are illustrated and described in a specific order, the order of the operations of each method may be changed so that certain operations can be performed in reverse order, or so that certain operations can be performed at least partially simultaneously with other operations. In another embodiment, the commands or suboperations of multiple distinct operations may be intermittent and / or alternating.

[0099] The above specification has described the present invention in relation to its specific exemplary embodiments. However, it will be apparent that various modifications and changes may be made to those embodiments without departing from the relatively broad idea and scope of the invention as described in the appended claims. Accordingly, this specification and the drawings should be considered illustrative rather than restrictive.

Claims

1. generating a in-phase drive signal and an anti-phase drive signal; selecting a sequence having a plurality of values of 1, values of -1, and values of 0, wherein the value of 1 corresponds to the in-phase drive signal, the value of -1 corresponds to the anti-phase drive signal, the value of 0 corresponds to a reference signal, the reference signal is derived from a voltage potential or a ground potential, and the sum of the sequence is equal to zero; in a first one of a plurality of scanning stages, applying to each of a first set of transmitter electrodes, substantially simultaneously, (i) at least one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal according to the sequence; rotating the sequence to obtain a rotated sequence; in a second one of the plurality of scanning stages, applying to each of a second set of transmitter electrodes, substantially simultaneously, (i) at least one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal according to the rotated sequence; receiving a sensing signal from a set of receiver electrodes to detect the presence of an object on the touch panel, the sensing signal representing a capacitance associated with the set of receiver electrodes; A method comprising the above steps.

2. The transmitter electrodes and the receiver electrodes are the same electrodes, The sensing signal represents the self-capacitance of the set of receiver electrodes, The method according to claim 1.

3. The transmitter electrodes and the receiver electrodes are different electrodes respectively, The sensing signal represents the mutual capacitance at the intersection of the transmitter electrodes and the receiver electrodes, The method according to claim 1.

4. When there are an odd number of transmitter electrodes, the sequence includes one or more values of 1, one or more values of -1, and one or more values of 0, The method according to claim 1.

5. The first set of the transmitter electrodes and the second set of the transmitter electrodes include a common electrode, The first stage and the second stage are sequential, The method further includes performing an analytical ambiguity reduction process to align a common mode level to form a baseline. The method according to claim 1.

6. The step of performing the analytical ambiguity reduction process is Determining a deconvolution cyclic matrix that is a pseudo-inverse matrix of an excitation matrix including the sequence and the rotated sequence; Calculating deconvolved data using the deconvolution cyclic matrix and the detection signal; Applying a common mode filter to the deconvolved data to obtain filtered data; Restoring a common value along the set of receiver electrodes to correct a baseline in the filtered data; Applying an orthogonal common mode filter to remove artifacts along the set of receiver electrodes to obtain a capacitance map of the touch panel; comprising; The method according to claim 5.

7. Generating an in-phase drive signal and an anti-phase drive signal; Applying a signal to each of a set of transmitter electrodes; Selecting a sequence having a plurality of values of 1, values of -1, and values of 0, wherein the value of 1 corresponds to the in-phase drive signal, the value of -1 corresponds to the anti-phase drive signal, the value of 0 corresponds to a reference signal, the reference signal is derived from a voltage potential or a ground potential, and the sum of the sequence is equal to zero; In a first one of a plurality of scan stages, at each of a first set of receiver electrodes, substantially simultaneously detecting (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal according to the sequence; Rotating the sequence to obtain a rotated sequence; In a second one of the plurality of scan stages, at each of a second set of receiver electrodes, substantially simultaneously detecting (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal according to the rotated sequence; A method comprising.

8. The transmitter electrode and the receiver electrode are the same electrode; The detection signal represents self-capacitances of the first set of receiver electrodes and the second set of receiver electrodes; The method according to claim 7.

9. The transmitter electrode and the receiver electrode are different electrodes respectively; The detection signal represents mutual capacitances at intersections between the transmitter electrodes and the receiver electrodes; The method according to claim 7.

10. When there are an odd number of receiver electrodes, the sequence includes one or more 1 values, one or more negative 1 values, and one or more 0 values. The method according to claim 7.

11. The first set of the receiver electrodes and the second set of the receiver electrodes include a common electrode. The first stage and the second stage are sequential. The method further includes the step of performing an analytical ambiguity reduction process to align the common mode levels to form a baseline. The method according to claim 7.

12. The step of performing the analytical ambiguity reduction process includes: determining a convolution-backward cyclic matrix that is a pseudo-inverse matrix of an excitation matrix including the sequence and the rotated sequence; calculating the convolution-backward data using the convolution-backward cyclic matrix and the detection signal; applying a common mode filter to the convolution-backward data to obtain filtered data; restoring a common value along the first set of the receiver electrodes and the second set of the receiver electrodes to correct the baseline; applying an orthogonal common mode filter to remove artifacts along the first set of the receiver electrodes and the second set of the receiver electrodes to obtain a capacitance map of the touch panel. including The method according to claim 11.

13. An apparatus including a capacitance detection circuit, the capacitance detection circuit is configured to select a sequence having a plurality of 1 values, negative 1 values, and 0 values, the 1 values corresponding to in-phase drive signals, the negative 1 values corresponding to anti-phase drive signals, the 0 values corresponding to reference signals, the reference signals being derived from a voltage potential or a ground potential, and the sum of the sequence being equal to zero; in a first scanning stage, the capacitance detection circuit is configured to apply, to each of a first set of transmitter electrodes of a touch panel, according to the sequence, (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal; the capacitance detection circuit is configured to rotate the sequence to obtain a rotated sequence. The capacitance detection circuit is configured to apply (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal to each of a second set of transmitter electrodes of the touch panel in a second scanning stage following the first scanning stage, according to the rotated sequence. The capacitance detection circuit is configured to receive a detection signal from a set of receiver electrodes of the touch panel to detect the presence of an object on the touch panel, and the detection signal represents the capacitance associated with the set of receiver electrodes. Device.

14. The capacitance detection circuit A selection circuit configured to be coupled to the first set of transmitter electrodes and the second set of transmitter electrodes and to the set of receiver electrodes; A signal generator coupled to the selection circuit for generating the in-phase drive signal and the anti-phase drive signal; A reference potential or ground potential for providing the reference signal; A detection circuit coupled to the selection circuit; An analog-to-digital converter (ADC) circuit coupled to the detection circuit; A processing core coupled to the selection circuit; comprising The device according to claim 13.

15. The device A signal source; An in-phase circuit coupled to the signal source for generating the in-phase drive signal; An anti-phase circuit coupled to the signal source for generating the anti-phase drive signal; further comprising The selection circuit includes a first set of multiplexers configured to be coupled to the first set of transmitter electrodes in the first scanning stage, and each of the first set of transmitter electrodes is coupled to the in-phase circuit, the anti-phase circuit, and the reference potential or the ground potential. The selection circuit includes a second set of multiplexers configured to be coupled to the second set of transmitter electrodes in the second scanning stage, and each of the second set of transmitter electrodes is coupled to the in-phase circuit, the anti-phase circuit, and the reference potential or the ground potential, and the first set and the second set include common electrodes. The device according to claim 14.

16. The processing core further receives digital data corresponding to the detection signal from the ADC circuit, determines a deconvolution cyclic matrix that is a pseudo-inverse matrix of an excitation matrix including the sequence and the rotated sequence. Using the inverse convolution circulant matrix and the digital data, calculate the inverse-convolved data, Apply a common mode filter to the inverse-convolved data to obtain filtered data, Restore a common value along the set of receiver electrodes to correct the baseline in the filtered data, Apply an orthogonal common mode filter to remove artifacts along the set of receiver electrodes to obtain the capacitance map of the touch panel, which is for, The device according to claim 14. **Claim 17** A device including a capacitance sensing circuit, The capacitance sensing circuit is configured to apply a signal to each of a set of transmitter electrodes, The capacitance sensing circuit is configured to select a sequence having a plurality of values of 1, values of -1, and values of 0, where the value of 1 corresponds to a in-phase drive signal, the value of -1 corresponds to an anti-phase drive signal, the value of 0 corresponds to a reference signal, the reference signal is derived from a voltage potential or a ground potential, and the sum of the sequence is equal to zero, In a first stage of a plurality of scanning stages, the capacitance sensing circuit is configured to detect, at each of a first set of receiver electrodes, according to the sequence, (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal, The capacitance sensing circuit is configured to rotate the sequence to obtain a rotated sequence, In a second stage of the plurality of scanning stages, the capacitance sensing circuit is configured to detect, at each of a second set of receiver electrodes, according to the rotated sequence, (i) one of the in-phase drive signal or the anti-phase drive signal and (ii) the reference signal, Device. **Claim 18** The capacitance sensing circuit, A selection circuit configured to be coupled to the set of transmitter electrodes, the first set of receiver electrodes, and the second set of receiver electrodes, A signal generator coupled to the selection circuit for generating the in-phase drive signal and the anti-phase drive signal, A reference potential or ground potential for providing the reference signal, A detection circuit coupled to the selection circuit, An analog-to-digital converter (ADC) circuit coupled to the detection circuit, A processing core coupled to the selection circuit, including The device according to claim 17.

19. The device comprises: a signal source; an in-phase circuit coupled to the signal source for generating the in-phase drive signal; an anti-phase circuit coupled to the signal source for generating the anti-phase drive signal; further comprising: The selection circuit includes, in the first stage, a first set of multiplexers configured to be coupled to a first set of the receiver electrodes, each of the first set of the receiver electrodes being coupled to the in-phase circuit, the anti-phase circuit, and the reference potential or the ground potential; The selection circuit includes, in the second stage, a second set of multiplexers configured to be coupled to a second set of the receiver electrodes, each of the second set of the receiver electrodes being coupled to the in-phase circuit, the anti-phase circuit, and the reference potential or the ground potential, and the first set of the receiver electrodes and the second set of the receiver electrodes including a common electrode. The device according to claim 18.

20. The processing core further: receives digital data corresponding to the detection signal from the ADC circuit; determines a deconvolution cyclic matrix that is a pseudo-inverse matrix of an excitation matrix including the sequence and the rotated sequence; uses the deconvolution cyclic matrix and the digital data to calculate deconvolved data; applies a common-mode filter to the deconvolved data to obtain filtered data; restores a common value along the first set of the receiver electrodes and the second set of the receiver electrodes to correct a baseline in the filtered data; applies an orthogonal common-mode filter to remove artifacts along the first set of the receiver electrodes and the second set of the receiver electrodes to obtain a capacitance map of the touch panel. for The device according to claim 18.

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