Method for resolving time ambiguities, associated system, associated transmitter and associated receiver

Abstract

An embodiment of the present invention relates to a method and associated apparatus for resolving time ambiguities between a wireless transmitter of a plurality of transmitters having a first time scale and a wireless receiver of a plurality of wireless receivers having a second time scale, the wireless transmitter being coupled to the wireless receiver, the wireless transmitter transmitting a wireless signal to the wireless receiver, the method comprising the steps of generating, by the wireless transmitter, an overlay sequence including a set of symbols for each time ambiguity interval, the set of symbols having a predetermined length, the overlay sequence satisfying a condition of a single occurrence of a subset of symbols within the set of symbols of the time ambiguity interval, each of the time ambiguity intervals including an implicit time marker; transmitting, by the wireless transmitter, the wireless signal to the wireless receiver, the wireless signal including the overlay sequence modulated onto a carrier of the wireless signal; receiving, by the wireless receiver, the wireless signal; and 1 capturing a snapshot of the wireless signal by a wireless receiver, the snapshot comprising a subset of symbols comprising N symbols of the overlay sequence; processing the snapshot by the wireless receiver to identify relative positions of implicit time markers of the wireless signal based on positions of the subset of symbols included in the snapshot within a set of symbols of a time ambiguity interval; and resolving time ambiguities between the first time scale and the second time scale by evaluating a delay between implicit time markers based on the processing of the snapshot, expressed at the first time scale, and implicit time markers in the overlay sequence generated based on the second time scale, the overlay sequence comprising an M-ary sequence based on an M-ary de Brown sequence.

Description

[Technical field]

[0001] SUMMARY OF THE DISCLOSURE Embodiments of the present invention relate to a method for resolving time ambiguities in a radio navigation system, an associated system, a transmitter, and an associated receiver. [Background technology]

[0002] Currently, in radio navigation systems such as the Global Navigation Satellite System (GNSS) radio navigation system comprising multiple radio transmitters and at least one radio receiver, the at least one receiver is adapted to receive radio navigation signals transmitted by each of the multiple transmitters, and such received signals can be used for position location and synchronization purposes.

[0003] Recent years have seen the emergence of new types of location (also called positioning) and timing services resulting from the convergence of new trends. First, the rapid development of terrestrial networks, especially using the new fifth generation (5G) communication standards to provide 10 to 1000 times the data throughput, positions mobile devices such as smartphones as the primary interface between users and communities or ecosystems. This change in outlook reduces the importance of the role played by the old Position Navigation Device (PND) or non-communicative “GPS swatch” in the mass market segment. To become “smartphone-centric”, devices with communication capabilities may include all kinds of “things” that can facilitate daily life, such as communication-enabled keys, household devices, etc. It should be noted that these “things” do not necessarily directly interfere with the daily life of the user, but may also include “micro-elements” (e.g. sensors such as “motes” or “smart dust”) or “macro-elements” (e.g. drones) as part of the new layer of transparency in their respective services. Second, there is general agreement that a significant increase in these connected elements could significantly increase the overall electricity required to power all of these end devices if no measures are taken, which is especially important in an era when global warming and the non-renewability of raw materials such as fossil fuels cannot be ignored.

[0004] Thus, a new category of not only positioning services but also timing services will emerge from millions, if not billions, of these communicating "elements," most of which also need their rough location and time on an absolute basis. Since meter or even decimeter accuracy can already be achieved with other GNSS signals and enhanced services, it has also been shown that the key performance measure for this new type of application is not precise or high precision positioning and timing performance, but rather the rapid provision of both time and position, preferably with minimal power required to access this information.

[0005] Radio navigation systems, such as global navigation satellite systems, can still play a role in the face of this evolution, as GNSS can provide an absolute time and position reference. However, current radio navigation signals of such existing systems, e.g. Global Navigation Satellite System (GNSS) systems, are not designed and optimized to support fast and sensitive synchronization between a first time scale, e.g., a GNSS time scale, of a user device, also called a radio receiver, and a second time scale, e.g., a time scale of a terrestrial network to which the radio receiver, e.g., the user device, is connected, or a local time scale generated by a receiver clock of such a radio receiver. In fact, it has been shown that the receiver's local time scale may rapidly deviate from the first time scale due to drift of the receiver clock, depending on the type of local oscillator implemented in the receiver.

[0006] In the following, as part of the background art, some mathematical descriptions are disclosed that formulate the methods that are typically implemented in a wireless receiver, e.g., a user device, to estimate the position and time of the user device. This description will in particular help the unskilled person to understand the prior art methods used to derive pseudoranges from the content of the received GNSS signals and navigation messages embedded in the GNSS signals, the prior art methods used to estimate the position and time of a wireless receiver connected to a communication network as part of Assisted GNSS (A-GNSS), or the prior art methods, also called "snapshot" positioning, used to estimate the position and time based on short parts of the signal. Some mathematical elements introduced in the description of these methods are also used to support the description of the invention that will be presented later in this application. For a description of these background art methods, reference is made to the following publications: [Reference document 1]: Non-patent document 1 [Reference document 2]: Non-patent document 2 [Reference document 3]: Non-patent document 3 [Reference document 4]: Non-patent document 4 [Reference document 5]: Non-patent document 5

[0007] Below is shown a prior art method used to calculate pseudoranges based on the reception of received signals and modulated navigation messages, as well as a prior art method used to calculate the position of a GNSS radio receiver based on the corresponding pseudoranges. A GNSS radio receiver needs to process at least four GNSS signals to derive its position and time. Here, it is assumed that the receiver is capable of demodulating the navigation message during tracking. Note that at least four satellites are needed to ensure that a solvable position equation takes into account the three coordinates (x, y, z) and the user receiver time offset Δb. Pseudorange ρ i is expressed as the following equation: i and the "physical" distance r between the user device iand an offset Δb that accounts for the clock alignment error between the user receiver and the GNSS time scale, and c0 denotes the speed of light. JPEG2024535752000002.jpg9150 Pseudo distance ρ i is also defined as the difference between the time of transmission at the satellite, expressed in the GNSS time scale, and the time of reception at the user device, expressed in the receiver time scale. JPEG2024535752000003.jpg9150

[0008] To derive the pseudoranges, existing radio satellite navigation signals are equipped with so-called time markers that indicate when the signal left the satellite with a Time of Transmission (ToT). Such time markers can take various forms. For example, in the case of GPS signals, the time markers include a Telemetry Word (TLM) and a Handover World (HOW) that contains the Time of Transmission (ToT). The TLM words are encoded in legacy signals several seconds apart in the navigation message, which forces radio receivers, such as user devices, to process such signals for a longer time in order to retrieve these TLM words, which is not optimal to reduce the power consumption of such radio receivers, such as user devices. The TLMs shall be transmitted synchronized to the GNSS time scale. The corresponding synchronized transmission is shown in the left part of Fig. 3. The satellite clock offset Δb Sat Note that because of this, perfect synchronization of the ToT is not achieved between the satellites, but the user device can correct the pseudoranges for this additional contribution based on the satellite clock correction models provided in the navigation messages.

[0009] There are at least two main techniques for computing pseudoranges that have a "common time of reception" on one side and a "common time of transmission" on the other side (see [1], [3], [4]). Both are equivalent and for the purposes of the discussion we will choose the "common time of reception". When expressed in the GNSS time scale it is called t GNSSrx,i or t if expressed in the receiver time scale. Rec rx,i All pseudoranges are calculated at the same epoch, denoted by ρ. On reception, the corresponding TLM words are not received at the same epoch due to different distances between the satellites and the user device, resulting in different propagation times. This is shown in the right part of Figure 3. The pseudorange ρ i To calculate , the receiver waits for the receipt of the TLM word, demodulates it, and determines when the signal was transmitted at epoch t GNSS ToT,i All you need to know is Epoch t GNSS rx,i The fractional part of the spreading code in t Rec rx,i By summing up the cumulative number of full spreading code periods (e.g., 1 ms for GPSC / A, 4 ms for Galileo E1-B / E1-C) between the TLM "reading" and the time (represented by the receiver time scale), the receiver can determine the relative reception time offset δ between the satellite and the user device. i The transmission time at satellite i is given by JPEG2024535752000004.jpg10152To construct the absolute pseudorange, the measured time t Rec rx,i This is because the transmission time t GNSS tx,i and the estimated distance between the satellite and the user, ρ est i For both the common time-of-transmission and time-of-reception methods, typically the first channel of the four to receive and demodulate the TLM is considered the basis for the construction of all other (e.g., three) pseudoranges.

[0010] For the first epoch (k=1) (i.e., at initialization), we usually use ρ est Set a rough value of 1[1] = ρ1, where ρ1 is the minimum travel time between the satellite and the user: for GPS, it is set to about 65 ms to about 85 ms, and for Galileo, it is set to about 77 ms to about 96 ms. For epochs (k>1) below JPEG2024535752000005.jpg9170, ρ est 1[k] is the information provided by the demodulated navigation message and the estimated user position (x est, y est ,z est ):ρ est 1[k]=r est 1[k-1] and r, the distance from the satellite to the user est i Based on the latest estimate of [k-1]. Finally, the time of reception in both the GNSS time scale and the receiver time scale can be expressed based on the receiver clock offset Δb. In the following description, for ease of explanation, the epoch index [k] is omitted. JPEG2024535752000007.jpg9147It is then possible to construct absolute pseudoranges for all four satellites by reusing equations (Eq. 2), (Eq. 4), and (Eq. 6) as follows: JPEG2024535752000008.jpg9157 Pseudo distance ρ i Once N (N ≥ 4) lines of sight are available, the absolute position solution x est, y est, z est is obtained from the linearization of the pseudorange equation as follows: JPEG2024535752000009.jpg8157In formula, -e i : represents the normalized vector connecting the user device position and the position of satellite i. -δX=[δx est , δy est , δz est ] is x est =x0+δx est , y est =y0+δy est and z est =z0+δz estand [x0,y0,z0] is the reference position used for linearization, which can be either a rough position estimate at the time of initialization of the position filter, or the latest state of the estimated position in the iterative solution. The reference position is the relative position solution [δx est , δy est , δz est ] based on the absolute position solution [x est ,y est ,z est ] is possible to express. −δΔb similarly represents the residual of the clock bias estimate. -ε i represents the additive measurement noise to the pseudorange. δ at each iteration X ext =[δ X , δΔb]=[δx est , δy est , δz est, The solution for the relative position of [δΔb] is given by the following equation (from [Reference 2]): JPEG2024535752000010.jpg8147In formula, -δ ρ is a vector of N residual pseudoranges according to (Equation 8). - H is of the form [ -e i 1], which is the so-called design matrix.

[0011] Below, we present a prior art method used to calculate pseudoranges in an assisted GNSS context. Many wireless receivers are connected to terrestrial communication networks that provide important prior information and information about both the user's position and time relative to a second time scale, such as the receiver's time scale. This feature represents an opportunity to accelerate the provision to the final user or associated application of precise position and time relative to a first time scale, such as the GNSS time scale. Such a scenario, called Assisted GNSS (A-GNSS), differs from standalone GNSS by the fact that the receiver does not have access to the navigation data modulated in the GNSS signals. It only tracks the GNSS signals to derive the pseudoranges. This means that the user device does not have access to the satellite Clock and Ephemeris Data (CED), which is usually modulated in the navigation messages, nor to the TLM and HOW that mark the transmission time of the navigation signals. To alleviate this lack of information, communication networks provide some of the information, such as clock and ephemeris data, but cannot provide all the necessary information, such as TLM and HOW. It has further been shown that other types of information can also be provided to the communication-enabled device, such as its rough location (e.g., using the dimensions and locations of cellular cells), or any other type of data that can facilitate signal acquisition, tracking or pseudorange calculation. Assisted GNSS typically contemplates the case where such information is conveyed to the communication-enabled device.

[0012] Both the satellite position and the clock offset are related to the transmission time t Ntx ToT,iNote that the time offset between the receiver and the GNSS time scale is calculated as ΔT, where ΔT = 2 × ΔT. However, it may differ from the GNSS time scale by a few milliseconds, in which case "fine time aiding" is considered. It may also differ by hundreds of milliseconds up to a few seconds, in which case "coarse time" aiding is considered. In the following, the time offset between the receiver and the GNSS time scale, i.e. the synchronization error, is called ΔT, where ΔT = 2 × ΔT. max A typical value for ΔT on a rough time scale is ±2 seconds, in which case ΔT max =2 seconds.

[0013] The first implications of A-GNSS for pseudorange calculations can be deduced: First, since the user device does not demodulate the TLM and HOW, this information required for pseudorange construction (see Equation 7) is not available and must somehow be conveyed by other means. Second, since there is no time marker, the time offset δ i To estimate the TLM and the reception time t Rec rx,i As a result, it is not possible to count the complete symbol periods that are completed in fractions between the received signal and the relative time offset δ i It is not possible to calculate Finally, the satellite positions and clock offsets calculated using models of the clock and ephemeris data provided by the network can differ by hundreds of meters or even kilometers for large ΔT values ​​due to the offset ΔT. As an example, assuming a maximum range change rate of 800 m / s for GPS orbits (900 m / s for Galileo orbits, respectively), this results in 1.6 km (1.8 km, respectively) (see [Reference 2]). Mathematical forms in which the magnitude of the pseudorange error can be expressed are described later in this section. Such errors in the satellite positions at the time of transmission will certainly propagate to the position and, if no mitigation measures are taken, will result in user device position errors of the same order of magnitude.

[0014] In [Reference 2] it is shown how the pseudorange, and in particular the pseudorange residual, is calculated in the particular case of A-GNSS. In the absence of TLM information, the pseudorange residual is given by the following formula: JPEG2024535752000011.jpg9153In formula, -ρ meas i represents the measured pseudorange reduced to a fractional part of the primary code in the case of A-GNSS, according to [Reference 2]: "Because the receiver has only measured the C / A code-phase offset and has not yet detected the data bit end or decoded the HOW, the measured pseudorange will be a sub-millisecond value (i.e., between 0 and approximately 300 km)." -ρ pred i represents the predicted pseudorange, which is constructed as follows: JPEG2024535752000012.jpg15159In formula, -X UD (t GNSS,est ToT,i ) is the estimated transmission time t GNSS,est ToT,i represents a rough location of the user device available to the user (which may be provided to the user device by the network) at -Δb Sat (t GNSS,est ToT,i ) is the estimated transmission time t GNSS,est ToT,i and represents the satellite clock bias offset provided by the network to the user device. -Δb Pred represents a rough estimate of the receiver clock bias. -X sati (t GNSS,est ToT,i ) is the estimated transmission time t GNSS,est ToT,i t represents the position of satellite i (provided by the network to the user device) in GNSS , est ToT , iis the actual transmission time t as a result of the receiver time synchronization error ΔT with the GNSS time scale GNSS ToT , i It has already been shown that this can differ from This error ΔT can lead to an error of several kilometers in the satellite position, as will be demonstrated below. It has been shown in [Reference 2] that the synchronization error ΔT leads to an additional error in the residual pseudorange (Equation 10) compared to the ideal case when ΔT = 0. This additional pseudorange error is expressed as the radial velocity or pseudorange rate of change v i is proportional to. The right part of Figure 4 shows that this additional contribution to the pseudoranges is not the same for all lines of sight. In contrast to the receiver clock bias, which is common to all pseudoranges, this non-common contribution introduces position errors that can reach hundreds of kilometers for a second level of synchronization error.

[0015] To address this situation, we introduce another variable ΔT in the extended state vector, in addition to the user position and clock offset: δ X ext =[δ X , δΔb]=[δx est , δy est , δz est, A new extended state vector [δΔb,ΔT] has been proposed. The aim is to estimate the synchronization between the receiver time, which may be synchronized to the network time or another local time scale, and the GNSS time scale. Different algorithms exist to solve this extended state vector, such as [Reference 2] and [Reference 5]. These algorithms solve the five unknowns by solving the extended state vector δ X ext We propose to introduce a fifth pseudorange to generate the determined simultaneous equations, to estimate as part of . This fifth pseudorange, derived from the fifth line of sight, is also shown in Figure 4 (compared to Figure 3, which shows only four pseudoranges).

[0016] In the following, a prior art positioning method based on a snapshot of the navigation signal, also called "snapshot positioning", is presented. Snapshot positioning is first introduced in an A-GNSS context. A-GNSS positioning is not only applied to receivers that continuously track satellite navigation signals. Another important subcategory of A-GNSS applications includes so-called snapshot positioning. In this case, the receiver "punctures" only a part of the received signal, also called a "signal snapshot", whose duration may include a few milliseconds to a few seconds (e.g., 1 or 2 seconds). Several designations exist for this kind of positioning application, such as "snapshot positioning", "instant positioning", or "single shot positioning". The short duration of the snapshot signal means that neither the satellite clock and ephemeris data (CED) nor the TLM words can be extracted and demodulated. For A-GNSS, the corresponding CED information may be provided by the terrestrial communication network or any other communication channel.

[0017] Note that if CED information previously extracted from satellite navigation signals (e.g., minutes or hours ago) is still valid or applicable, it is also possible to apply them to the pseudoranges derived from the snapshot, in which case the snapshot positioning is no longer assisted but stand-alone.

[0018] Therefore, for both A-GNSS and standalone snapshot positioning, the main problem is related to the lack of time synchronization information from the TLM words that are not part of the signal snapshot. To solve this problem, solutions such as the "millisecond integer ambiguity" that are encountered mainly in the literature dedicated to A-GNSS / A-GPS have been disclosed. The "millisecond integer ambiguity" is a measure of the time lag between the measured pseudorange ρ meas iThis is related to the fact that θ is not an absolute pseudorange as in conventional standalone positioning, but only a fraction of the code period (for GPS C / A, one code period is 1 ms), and the integer value of the code period is not part of the measured pseudorange. The lack of an absolute reference results in multiple solutions of the simultaneous equations, i.e. suboptimal solutions, of which only one is the correct optimal solution. Different solutions exist to resolve the corresponding millisecond ambiguities, which are close to the problem of integer ambiguities encountered in the case of carrier positioning. For example, in [Reference 5], other applications of the "lambda method" are disclosed. Finally, it should be noted that the "millisecond integer ambiguity" method and the method based on the fifth unknown mentioned above also share some commonalities in the sense that they are based on the use of more than one (at least five) different lines of sight to guarantee the resolution of the pseudorange ambiguities.

[0019] In conclusion, the following main factors explain the inferior performance of A-GNSS positioning and snapshot positioning compared to conventional standalone positioning: The effect of the synchronization error ΔT can result in satellite position errors of hundreds of meters or even kilometers. This motivates the introduction of a new variable to be estimated, besides the user position and clock offset, namely the fifth unknown ΔT, in order to "clean" this additional performance penalty. - The introduction of a fifth variable requires a fifth pseudorange, i.e. line of sight, which can impact availability and accuracy performance, especially in urban environments where satellites are poorly visible. "Millisecond ambiguity" (closely related to methods that utilize a fifth unknown) requires additional mitigation techniques to resolve the correct absolute optimum.

[0020] A typical application of such radio navigation systems is therefore the case of global navigation satellite systems, where the satellite clock and orbit correction models applied to the estimated satellite-user device pseudoranges are provided to the radio receivers via a terrestrial communication network, as in the A-GNSS context. As explained above, one of the main problems cited regarding the seamless synergy between the processing of radio signals and the aforementioned network information is that the corresponding models are not necessarily based on the actual time epochs "t" expressed in a first time scale, i.e. the GNSS time scale. gnss ” is expressed in a second time scale, i.e. the receiver time scale (which may be synchronized with the terrestrial network time scale), which may differ from the ground network time scale by a few seconds. Rx ". In this case, it has been shown previously that the corresponding deviations can have a significant, if not detrimental, effect on the positioning solution derived with incorrectly corrected pseudoranges. Therefore, there is a strong need to solve the time synchronization between the first time scale, i.e. the GNSS time scale, and the second time scale, i.e. the user device time scale, for the purpose of calculating pseudoranges based on information provided by the communication network (e.g., CED) in a manner that matches the receiver satellite range observations and with the shortest signal snapshot duration to support applications where the receiver power consumption must be as low as possible. Moreover, this property must be achieved over long symbol durations to support high sensitivity applications.

[0021] Therefore, a problem with such navigation systems and associated radio receivers is that radio receivers applying snapshot positioning, which is affected by a second time scale, exhibit large synchronization errors with respect to a first time scale, such as the GNSS time scale. Another drawback of current radio signals, e.g., GNSS signals, is that while they have the ability to achieve this synchronization with the shortest portion of the radio signal to reduce the number of operations, also to reduce the power consumption of the radio receiver, they are unable to maintain long symbol times without losing sensitivity.

[0022] Non-Patent Document 6 further discloses that spread spectrum technology, originally conceived to counter the effects of noise and interference, has not only enabled the development of advanced mobile, multi-user, and satellite-based solutions that are one of the most common and widespread communication technologies today, but also provides a preliminary performance analysis of direct sequence spread spectrum signals obtained by using an innovative binary spreading sequence, the de Brown sequence, in scenarios where large Doppler shifts and relative rates of change occur as a result of the sometimes variable high speed of aircraft and, in the worst case, the loss of the on-board frequency offset estimation capability. These results show that the use of the binary de Brown sequence can improve signal recovery at the receiver even in the presence of large distortions due to the uncompensated Doppler effect. Patent document 1 discloses a pseudorange measuring device for providing pseudorange information representing an estimate of a distance between a transmitter and a receiver based on a modulated signal having a sequence of symbols, the primary code sequence being modulated according to a secondary code sequence, the primary code sequence being configured to progressively correlate a portion of a received signal having at least two symbols with at least two reference sequences, the first reference sequence representing at least two subsequent symbols having the same phase and the second reference sequence representing at least two subsequent symbols having a different phase, and depending on the result of the correlation, progressively acquiring a portion of the secondary code sequence. The pseudorange measuring device is configured to provide the pseudorange information based on acquiring a meaningful portion of the secondary code sequence.

[0023] Non-Patent Document 7 discloses a possible concept raised within the framework of the Galileo Evolution activity, called Quasi-Pilot (QP) signals, designed to also meet the needs of IoT devices, where the main driver for the design of the QP signal is the reduction of acquisition complexity, allowing fast and robust resolution of time ambiguities, coupled with the ability to allow long coherent integration to achieve sufficient sensitivity in challenging environments when necessary. Furthermore, the QP signal design should allow handover to existing legacy signals in order for users to also take advantage of the high-precision capabilities. [Prior art documents] [Patent documents]

[0024] [Patent Document 1] US Patent Application Publication No. 2016 / 161614 [Non-patent literature]

[0025] [Non-Patent Document 1] “Using GNSS Raw Measurements On Android Devices(White Paper)”, Raw Measurements Task Force, European GSA [Non-Patent Document 2] “A-GPS:Assisted GPS,GNSS,and SBAS, Frank Van Diggelen, GNSS Technology And Application Series, Artech House [Non-Patent Document 3] "Code Tracking Pseudoranges.How can pseudorange measurements be generated from code tracking?", M.Rao, G.Falo, InsideGNSS, January / February 2012 [Non-Patent Document 4] “Estimation of Satellite-User Ranges Through GNSS Code Phase Measurements” by Marco Pini [Non-Patent Document 5] "GPS Position Can Be Computed without the Navigation Data", N. Sirola, ION GPS 2002, pp. 24-27, September 2002, Portland [Non-Patent Document 6] PELLICCIONI GIOVANNI et al.: "DE BRUIJN SEQUENCES AS SPREADING CODES IN EXTREME DOPPLER CONDITIONS: ANALYSIS AND RESULTS" [Non-Patent Document 7] WALLNER STEFAN et al.: "NOVEL CONCEPTS ON GNSS SIGNAL DESIGN SERVING EMERGING GNSS USER CATEGORIES:QUASI-PILOT SIGNAL" Summary of the Invention

[0026] It is an object of embodiments of the present invention to provide a method for a radio navigation system of the above known type, an associated system, a time ambiguity resolution device in a radio transmitter and a radio receiver of such a radio navigation system, in which the above mentioned drawbacks or disadvantages of the known solutions are reduced or overcome. In particular, it is an object to provide such a method, a system and an associated radio receiver applying snapshot positioning, which overcomes synchronization errors with respect to a first time scale when influenced by a second time scale.

[0027] The object is to first generate, by a radio transmitter, an overlay sequence comprising a set of symbols for each time ambiguity interval, the set of symbols having a predetermined length, the overlay sequence satisfying a condition of a single occurrence of a subset of symbols within the set of symbols of a time ambiguity interval, each of the time ambiguity intervals comprising an implicit time marker; subsequently, transmitting, by the radio transmitter, a radio signal to a radio receiver, the radio signal comprising the overlay sequence modulated onto a carrier wave of the radio signal; receiving the radio signal by the radio receiver; and capturing a snapshot of the radio signal by the radio receiver, the snapshot comprising a subset of N symbols of the set of symbols of the overlay sequence. The method is accomplished by: capturing a snapshot of the wireless signal, and within a time ambiguity interval of the wireless signal, by the wireless receiver to retrieve values ​​of N symbols of the overlay sequence and determine a relative position of an implicit time marker of the wireless signal represented at a first time scale based on a position of a subset of symbols included in the snapshot within the set of symbols of the time ambiguity interval, and then determining by the receiver a time ambiguity between the first time scale and a second time scale by evaluating a delay between the implicit time marker obtained from processing the snapshot and an implicit time marker in the overlay sequence generated based on the second time scale, where the overlay sequence comprises an M-ary sequence based on an M-ary de Brown sequence. The general process of the method of the present disclosure is shown in FIG.

[0028] An overlay sequence based on an M-ary sequence means that the overlay sequence may comprise either a de Brown sequence modulated onto the carrier wave of the radio signal, or a truncated de Brown sequence, or an integrated de Brown sequence, or a combination of two or more de Brown sequences and / or truncated sequences and / or integrated de Brown sequences.

[0029] If the overlay sequence contains a combination of V de Brown sequences and / or truncated sequences and / or integrated de Brown sequences, then the given number of symbols L corresponds to a symbol periodicity expressed in units of symbols of the aggregated overlay sequence obtained by the combination of the V constituent sequences. If the V constituent sequences contain binary symbols, then the aggregated overlay sequence obtained for each combination contains M-ary symbols, where M=2×V. Furthermore, the number N of symbols contained in a snapshot of a subset of symbols of the aggregated overlay sequence is such that it satisfies the single occurrence property SO(L,N) and also guarantees the maximization of the ratio L / N. These definitions of the parameters L and N are based on the symbol periodicity expressed in units of time T s is the same for the V different constituent sequences that are combined to form the aggregated overlay sequence. If the corresponding symbol durations differ among the V different constituent sequences that are combined to form the aggregated overlay sequence, then we can extend the definition of the periodicity of the aggregated overlay sequence in defining the symbol duration of the aggregated overlay sequence as the greatest common divisor of the symbol durations of the V constituent sequences. The periodicity L of the aggregated overlay sequence is the sum of its duration T sLet be denoted by the symbols just defined. With this extended definition of the aggregated overlay sequence L, which is applicable when the symbol durations vary between different constituent sequences, the snapshot duration again contains N symbols of the aggregated overlay sequence, which on the one hand satisfies the SO(L,N) property and on the other hand guarantees the maximization of the ratio L / N.

[0030] In addition, since it is further recognized that fast time provision or synchronization based on the shortest duration of signal snapshots results in lower power consumption of the user device for signal snapshot processing, the application of an overlay sequence based on an M-ary de Brown sequence ensures the further advantageous single occurrence property of the subset in the overlay sequence. Therefore, an additional design constraint is that the ratio of the overlay sequence to the snapshot duration proportional to L / N should be as large as possible. This property ensures the most efficient snapshot length for a given time ambiguity interval. To achieve this further objective, the overlay sequence modulated onto the wireless signal is based on a de Brown overlay sequence.

[0031] In the following, it is assumed that acquisition of the primary code has already been achieved and that embodiments of the present invention are independent of the type of acquisition scheme. Furthermore, the processing steps of the present invention assume that the code delay and Doppler offset obtained from the acquisition step are known with sufficient accuracy so as not to degrade the performance of those further processing steps.

[0032] The radio receiver may be embodied by any kind of radio receiver and is not limited to receivers that derive the binary values ​​of the N symbols by implementing a Phase Locked Loop (PLL), but may also derive the values ​​by exploiting relative phase changes (i.e., by implementing a Frequency Locked Loop - FLL). The exact implementation details of both PLL and FLL techniques, as well as any other type of demodulation technique used to derive the corresponding overlaid sequence of M-ary symbols, shall be known to those skilled in the art.

[0033] The required duration of a signal snapshot including N symbols required to retrieve the values ​​of N overlay symbols exceeds the exact duration of the N symbols, i.e. N times the symbol duration, by a small fraction of the total snapshot duration including one time guard located on each side of the signal snapshot. The total duration of these time guards depends on the exact symbol retrieval process and other configuration parameters such as the signal to noise power spectral density ratio (C / N0), and the duration of this additional snapshot portion is usually much shorter than the exact duration of the N symbols. Therefore, in the following, the signal snapshot duration is incorrectly specified with respect to the duration of N symbols, but the signal snapshot duration shall be interpreted as the sum of the duration of N symbols and the additional duration of both time guards. Some numerical examples providing specific orders of magnitude of the corresponding snapshot and time guard durations are given later in the detailed description section.

[0034] In this way, the correct location of the implicit time marker in the time ambiguity interval relative to the snapshot location can be determined based on a single piece of information contained in a snapshot of the wireless signal, the snapshot comprising a subset of N symbols of the overlay sequence. The location of the snapshot relative to the time ambiguity interval can be determined based on information derived from the wireless signal, i.e., the subset of N symbols. Based on the location of the snapshot within the time ambiguity interval, the location of the implicit time marker can be estimated, and this information can be used for synchronizing the first and second time scales.

[0035] Note further that the location of the implicit time marker is also known in the relative time frame of the received signal. The overlay sequence contains a set of L symbols for each time ambiguity interval, each of which contains an implicit time marker. The location of the implicit time marker within the time ambiguity interval is known (according to a rule) and can be, for example, the first symbol of the sequence.

[0036] Thus, the derivation of implicit time markers based on information contained in this short snapshot of the received signal makes it possible to perform a time transformation to synchronize the second time scale of the user device to the first time scale, i.e. the absolute GNSS time scale of the radio transmitter.

[0037] The set of symbols of the overlay sequence consists of a predetermined number L of symbols, and the snapshot of the signal consists of a number N of symbols, N being less than L. L can also be understood as the periodicity of the overlay sequence expressed in units of overlay sequence symbols.

[0038] The derivation and processing of implicit time marker information represents an alternative to existing solutions such as the "fifth unknown" or "millisecond integer ambiguity" techniques previously described in the context of A-GNSS / A-GPS. This alternative, in comparison to the "fifth unknown", avoids "sacrificing" one line of sight since the required information is an inherent part of each signal, thus improving availability. Such a time marker indicates the time of transmission of the signal and may be embodied differently in different types of systems. In the Global Positioning System (GPS), the (explicit) time marker comprises a TLM word that explicitly encodes the time of transmission, whereas in embodiments of the disclosed solution, the time marker word is implicit since it corresponds to the beginning of the overlay sequence (first symbol) according to a rule. It should be noted, however, that the rule for the location of the implicit time marker can be defined elsewhere in the sequence, for example at the last symbol, as long as this rule is known by both the sender and the receiver.

[0039] Moreover, while in the case of (legacy) global navigation satellite systems the TLM word is the absolute time reference ("time scale") of the GNSS providing the complete date and time within the week since midnight of the most recent Saturday (Saturday 24:00 is the reference time for weekly TLMs): 3 days, 7 hours, 36 minutes, 40 seconds... + week number, in embodiments of the disclosed solution the processing of a subset of symbols is done for their relative position to the start of the sequence represented by an implicit time marker. Thus only the relative time within the time ambiguity interval with a duration equal to the overlay sequence duration is provided. Nevertheless, some embodiments disclose extending the duration of the time ambiguity interval to values ​​much more than minutes or even hours by appropriately selecting the parameters N and L when assuming an overlay sequence based on a single de Brown sequence or by assuming an overlay sequence based on a combination of several de Brown sequences.

[0040] In the above disclosure, it is assumed that the first time scale is shared within a global navigation satellite system that transmits signals to a device equipped with a GNSS receiver and is synchronized to the second time scale of the device. However, alternative applications are possible in which the first time scale is shared by a space-based communication network, or by a system that transmits signals via a terrestrial communication network or a base station or a beacon, or the first time scale is shared by another communication-enabled device, for example in a "machine-to-machine" communication link, such as vehicle-to-vehicle (V2V), vehicle-to-exchange (V2X), or device-to-device (D2D). In the latter case, the second "slave" device is synchronized to the first "master" device by the method of the present disclosure.

[0041] Such a radio navigation system may comprise a number of transmitters with a first time scale, meaning that such a transmitter of the number of transmitters deals with a time scale that is global across the number of transmitters. For ease of understanding, it is considered that the transmitters are perfectly synchronized to a global time scale or that a model such as a clock correction model allows the time scales of the number of transmitters to be estimated with sufficient accuracy relative to the global time scale. In the case of GNSS, the satellite clock correction model allows the alignment of each local time scale of the satellites to a global time scale, i.e. the GNSS time scale. This first global time scale is therefore quite different from the second time scale dealt with by radio receivers communicating with other systems where a second time scale applies.

[0042] It is further recognized that fast time provision or synchronization based on a minimum duration of signal snapshots reduces the power consumption required for signal snapshot processing in user devices. Therefore, an additional design constraint is that the ratio of the overlay sequence to the snapshot duration proportional to L / N should be as large as possible. This property ensures the most efficient snapshot length for a given time ambiguity interval. To achieve this further objective, the overlay sequence modulated onto the wireless signal is based on a de Brown overlay sequence.

[0043] [Reference 6]: "Generalizing the classic Greeding and Nicklace Constructions for De Bruijn and Universal Cycles", Joe Sawada, Aaron Williams, and Dennis Wong. [Reference 7]: "A problem in arrangements", M. H. Martin, Bulletin of the American Mathematical Society, 40:859-864, 1934.

[0044] The objective is to provide enough information in the snapshots that it is possible to position them relative to an implicit time marker in the time ambiguity interval. For this purpose, a particular type of overlay sequence called a "de Brown" sequence is applied. Such a "de Brown" sequence guarantees a single occurrence of any subsequence of length N in an overlay sequence of length L (including on the boundary). This property satisfied by a "de Brown" sequence is called the single occurrence of N symbols in L symbols or the SO(N,L) property. Such a "de Brown" sequence may contain binary symbols, but does not inherently contain them. Alternatively, other M-ary sequences may be applied to embody a de Brown sequence. For example, if we consider a quaternary alphabet containing the symbols 0, 1, 2, and 3, then "003" and "213" represent two examples of quaternary sequences of length 3. The definition of a de Brown M-ary sequence is given in [Reference 6]: "Let T(n;k) be a set of k-ary character sequences of length n. For example, T(2;3)={11;12;13;21;22;23;31;32;33}. A de Brown sequence of T(n;k) is a set of k-ary character sequences of length n such that, when viewed circularly, each character sequence in T(n;k) appears as a subcharacter sequence exactly once. n , a sequence of length M N If we denote the de Brown sequence of M by B(M,N), then the number of distinct de Brown sequences B(M,N) equals M^(M^(N-1)-N). Certain cases of de Brown sequences contain binary symbols, and in such cases the "de Brown" sequences are called binary "de Brown" sequences. A binary "de Brown" sequence is a sequence such that L=2^N and the number of "de Brown" binary sequences satisfying the SO(N,L) property equals 2^(2^(N-1)-N) (see [Reference 6]).

[0045] Furthermore, the "de Bruijn" sequence also satisfies the cyclic property that even if it is a sub-sequence of length N constructed by concatenating the last k (k < N) symbols of the sequence with the first [N - k] symbols, it occurs only once within the complete "de Bruijn" sequence. One important property of the "de Bruijn" sequence is the large ratio of (L / N) = (2^N / N), which represents an excellent advantage for snapshot positioning. In fact, this means that for a small number N of symbols (i.e., short snapshot duration), the overlay sequence length (i.e., time ambiguity interval) can be large. Some examples of de Bruijn sequences with different values of length L are shown in the table of Figure 7 for illustration.

[0046] Various methods enable the generation of de Bruijn sequences. The object of the present invention is not to examine in detail all the references that describe methods for generating such "de Bruijn" sequences, but rather to utilize such "de Bruijn" sequences, particularly to generate a large pool of candidate "de Bruijn" sequences from among the M^(M^(N - 1) - N) existing M-ary B(M,N) "de Bruijn" sequences, from which a specific "de Bruijn" sequence that provides specific characteristics advantageous for time ambiguity resolution is selected. As an example, citing [Reference 6], "Martin showed in 1934 in [Reference 7] that the de Bruijn sequence of T(n;k) can be constructed by a simple greedy algorithm. The algorithm starts with the sequence k n-1 (where the exponentiation represents iteration), and then the following rule is repeatedly applied: add the smallest symbol to {1; 2;....; k} so that the partial character sequences of length n in the resulting linear sequence are different."

[0047] As a result of the SO(N,L) property fulfilled by the "de Brown" sequence, the position of this unique symbol sequence within the interval of a radio signal such as a GNSS signal or any kind of terrestrial signal can be unambiguously identified and, based on the position of this unique sequence, the (relative) distance between the position of the unique sequence of N symbols contained in the snapshot and the position of an implicit time marker can be precisely measured. Furthermore, the SO(N,L) property achieved by "de Brown" provides an optimized ratio between snapshot duration and time ambiguity interval and is therefore the most efficient in terms of power consumption of the user device.

[0048] In another related embodiment of the present invention, the sequence generation means of the radio transmitter is further configured to generate a plurality of different overlay sequences, which may each be modulated onto a different primary code or chip stream multiplexed onto the same carrier signal.

[0049] An advantage of this further embodiment is that it allows the time ambiguity interval to be extended by joint processing at the receiver side of multiple overlay sequences. In the special case where the multiple overlay sequences consist of at least a first untruncated M-ary de Brown overlay sequence and at least one second truncated M-ary de Brown overlay sequence, differences in overlay sequence lengths ensure that corresponding snapshots do not occur more than once in an "implicit" aggregated overlay sequence whose length is obtained by summing the length of the untruncated sequence and the length of the subsequent truncated sequence, the length of this aggregated overlay sequence corresponding to the extended ambiguity period. Each of the multiple overlay sequences may be modulated onto a dedicated signal component following the same techniques as the modulation of a single overlay sequence onto that dedicated signal component. In yet another embodiment of the invention, the subset of symbols included in the snapshot is extended with an additional (adjacent or non-adjacent) subset of symbols of the overlay sequence, the additional subset being N Ext symbols, the expanded symbol subset is P=N+N Ext Contains symbols.

[0050] In other words, the subset of symbols included in the snapshot may be a subset of symbols that includes N symbols from the set of symbols of the time ambiguity interval, and may further include a subset of N symbols that may be adjacent to the first subset of N symbols or may be separated by Q symbols from the subset of N symbols. Ext a second subset of symbols including P=N+N symbols in the overlay sequence, and the processing means (23) processes the expanded subset of symbols included in the snapshot and a second subset of symbols including P=N+N symbols in the overlay sequence. Ext Calculate the Hamming distance between each of the L possible subsequences of the overlay sequence that contains symbols (which are the same length as the subset of extended symbols contained in the snapshot) and the subset of extended symbols contained in the snapshot, P=N+N Ext and further configured to detect an error in the extended subset of symbols if a minimum over all L Hamming distances calculated between each subsequence in the overlay sequence containing the symbols is non-zero or is zero and occurs multiple times;

[0051] Finally, the processing means (23) calculates a subset of the expanded symbols contained in the snapshot and P=N+N Ext The method is further configured to determine a relative position of an implicit time marker of the wireless signal based on the subset of extended symbols included in the snapshot if the minimum over all Hamming distances calculated between each subset of the overlay sequence containing the symbol is 0 and occurs once. The position of the extended subset of symbols that allows time ambiguity resolution is P = N + N Ext symbols, resulting in a Hamming distance of zero to the extended subset of symbols. Furthermore, the subset of symbols included in the snapshot is expanded and P=N+N Ext If the minimum of all L Hamming distances calculated between each subsequence of the overlay sequence containing N symbols is 0 and occurs only once, then the symbol of the extended subset included in the snapshot is N err,max It has been shown that it is possible to guarantee with a 100% confidence level that the demodulation error will be less than (or equal to)

[0052] Furthermore, a predetermined minimum value N err,max is up to N err,max The error detection of P errors is deduced from an iterative process for the selection of an overlaid de Brown sequence that supports error detection of P errors, whereby any extended subset of P symbols in the overlaid de Brown sequence is randomly located within the P symbols, err ≦N err,max is the maximum N err Even if contaminated by individual errors, the overlaid De Brown sequence is guaranteed to be error-free and not occur once.

[0053] In yet another embodiment of the present invention, the processing means (23) determines a subset of the extended symbols contained in the snapshot and P=N+N Ext The minimum value over all the Hamming distances calculated between each subsequence of the overlay sequence containing the symbols is determined to be a second predetermined minimum value depending on the selected overlay sequence. JPEG2024535752000014.jpg11166, in which case the receiver is further configured to correct the error if the error does not exceed the maximum Hamming distance between the subsequence of the overlay sequence containing P=N+NExt symbols and the extended symbol subset included in the snapshot. JPEG2024535752000015.jpg 11166 symbols are corrected. In this embodiment, JPEG2024535752000016.jpg12166 refers to the integer part just below the value x.

[0054] In yet another embodiment of the present invention, the receiving means of the radio receiver RX1 is further configured to receive a first radio signal from a first radio transmitter and at least a second radio signal from a second radio transmitter, the first radio signal comprising an overlay sequence having a length of L symbols and the at least second radio signal having a length of L1 symbols, the first overlay sequence and the at least second overlay sequence being different, the receiving means subsequently combining the overlay sequence of the first radio signal with the overlay sequence of the at least second radio signal into an aggregated overlay sequence, and the snapshot capturing means captures a snapshot of the aggregated overlay sequence of the first radio signal and the at least second radio signal, the snapshot comprising a subset of symbols of the aggregated overlay sequence.

[0055] The processing means may then determine relative positions of implicit time markers of the wireless signal based on positions of a subset of symbols of the aggregated overlay sequence included in a snapshot comprising N symbols, and the processing means may then further resolve time ambiguities between the first time scale and the second time scale by evaluating delays based on the implicit time markers and processing of snapshots in the aggregated overlay sequence generated based on the implicit time markers represented at the first time scale and the second time scale.

[0056] An advantage of this further embodiment is that it allows the time ambiguity interval to be extended by joint processing at the receiver side of multiple overlay sequences. If overlay sequences with different lengths are transmitted by the first and second radio transmitters, it is guaranteed that a corresponding snapshot does not occur more than once in an "implicit" aggregated overlay sequence whose length is obtained by summing the length of the untruncated sequence and the length of the subsequent truncated sequence, the length of this aggregated overlay sequence corresponding to the extended ambiguity period.

[0057] In another related embodiment of the present invention, a radio transmitter (T x) is further configured to generate a truncated transition sequence based on an original sequence consisting of an original de Brown sequence having a length of L symbols by first removing N symbols containing "0" from the original sequence, followed by removing a single symbol containing "1" from the original sequence to result in a truncated sequence, and optionally removing additional K symbols from the truncated sequence to result in a truncated transition sequence of length LN-1-K; generate a first integrated sequence indicative of a phase transition of the truncated transition sequence and a second integrated sequence indicative of a phase of the inverted truncated transition sequence, the first integrated sequence being in an opposite phase to the second integrated sequence; and then generate a concatenated integrated sequence by concatenating the first integrated sequence and the second integrated sequence, the concatenated integrated sequence being configured to be modulated onto a carrier wave of the wireless signal.

[0058] In yet another related embodiment of the present invention, the snapshot capture means is adapted to take a snapshot of the radio signal, the snapshot being captured by the radio transmitter (T) according to claim 8. x ), the snapshot comprising a subset of symbols of an overlay sequence consisting of a concatenated aggregate sequence generated by a time ambiguity interval (TAI) of the wireless signal, the snapshot comprising N+1 symbols, the processing means being further configured to identify N transitions from the subset of symbols of the overlay sequence comprised in the snapshot and subsequently identify a position of the subset of symbols comprised in the snapshot relative to an implicit time marker of the wireless signal based on the N transitions from the subset of symbols comprised in the snapshot in an entry of the repository (25), the repository (25) comprising for each entry a plurality of symbols of the snapshot and a plurality of relative positions of the snapshot relative to the time markers within a time ambiguity interval of the wireless signal.

[0059] A further related embodiment relates to a wireless receiver for resolving time ambiguities, wherein the processing means (23) of the wireless receiver is further configured to determine a relative position of an implicit time marker represented at a first time scale in the wireless signal by searching a subset of symbols contained in snapshots in entries of a repository, the repository including, for each entry, a plurality of symbols of the snapshot and a relative position of the plurality of symbols of the snapshot with respect to the implicit time marker within a time ambiguity interval of the wireless signal.

[0060] The repository can function as a lookup table that associates a subset of N symbols of the sequence with its relative position within the complete sequence of L symbols, and therefore an implicit time marker: the N symbols are input into the repository and the relative position is output as a result.

[0061] In other words, this subset of N symbol values ​​is used to obtain an entry in a repository in which it is possible to find a subset of N symbols according to the snapshot, based on a subset of N symbols of the overlay sequence retrieved from the snapshot content using either a PLL, or an FLL, or any other type of demodulation technique aimed at estimating symbol values, the repository also containing information about the relative positions of these N symbols contained in the snapshot within a time ambiguity interval, or equivalently, the relative positions of the N symbols contained in the snapshot relative to an implicit time marker whose position in the overlay sequence is known according to a rule.

[0062] Such a repository may contain L subsets of N symbols, allowing to locate a snapshot of N symbols within a complete sequence of L symbols, thus resulting in L×N lookup tables. Another related embodiment relates to a wireless receiver for resolving time ambiguities, further comprising: generating a snapshot sequence from a wireless signal comprising a subset of N symbols of a set of L symbols corresponding to the wireless signal transmitted by a transmitter; the snapshot receiver further configured by the processing means to determine a relative position of an implicit time marker represented at a first time scale in the wireless signal by applying a partial autocorrelation between the snapshot sequence and the entire set of L symbols to estimate a position of the subset of N symbols within the entire set of L symbols enabling to determine a relative position of the N symbols comprised in the snapshot sequence within the time ambiguity interval. Here, the term partial autocorrelation function is used because only the subset of N symbols is multiplied and added with the entire overlay sequence of L symbols and the remaining part is completed with zeros, i.e. by applying zero padding, as shown in FIG. 8. The offset between the snapshot sequence and the overlay sequence corresponding to the autocorrelation maximum makes it possible to identify the position of a subset of N symbols contained in the snapshot sequence within a set of L symbols of the overlay sequence in a time ambiguity interval, or equivalently, to identify the relative position of the N symbols contained in the snapshot sequence with respect to an implicit time marker whose position in the overlay sequence is known according to a rule.

[0063] Two methods may be disclosed for generating the snapshot sequence. A first method, which can be classified as part of the general soft decoding technique, generates a snapshot sequence incorporating samples derived from a signal snapshot and obtained after wiping off both the Doppler offset and the code delay estimated from the acquisition process, i.e. without an intermediate step aimed at retrieving the values ​​of the N symbols contained in the signal snapshot. More precisely, this first method consists in concatenating samples derived from a signal snapshot comprising a subset of N binary symbols and additive reception noise to the signal samples, and concatenating them with another subset of "0 samples" obtained after code delay and carrier Doppler wipe-off, with zero padding, to complete a snapshot sequence with a length equal to the overlay sequence L multiplied by the number of samples per symbol duration. This snapshot sequence is then correlated with a spread overlay sequence based on the overlay sequence corresponding to the snapshot sequence, and whose length is equal to the overlay sequence length L multiplied by the number of samples per symbol duration. The term spread is used because each symbol of the spread overlay sequence is repeated as many times as the number of samples in one symbol duration. The type of samples and the number of samples per symbol are configurable and can correspond directly to RF samples or to samples after correlation, where this first correlation operation is performed with the primary code during the signal acquisition process. Thus, the type of samples depends on the receiver implementation, but the radio receiver needs to remove the Doppler offset and code delay in all cases. Thus, both the snapshot sequence and the spread overlay sequence have the same length and can therefore be processed with an autocorrelation operation.

[0064] The second method consists of concatenating a subset of N binary symbols retrieved from the signal snapshots by using a PLL, or FLL, or any other type of demodulation technique aimed at estimating symbol values, with another subset of LN "0's" obtained with zero padding to complete a snapshot sequence of length L. Due to this intermediate step of symbol value retrieval in the snapshot sequence generation, this second method can be classified as a general hard decoding technique. This snapshot sequence of length L is then correlated with an overlay sequence of length L corresponding to the snapshot sequence.

[0065] When the number of L symbols in the overlay sequence becomes too large, applying this partial autocorrelation method rather than a repository (i.e., a look-up table) is advantageous to avoid applying an oversized look-up table (repository) using excessive storage memory and to avoid excessively long access times if the look-up table maintained by such a repository is too large. For example, for N=7 and L=2^7=128, the memory requirements are smaller assuming the second option, since instead of storing 128×7 lookup tables, we generate a single snapshot sequence that includes snapshots of N=7 symbols, complete with 128-7=121 symbols set to 0.

[0066] However, alternative applications may be envisaged in which the first time scale is shared by a space-based communications network, or by a ground communications network or a system transmitting signals via base stations or beacons, or in which the first time scale is shared by another communications-enabled device, for example in a "machine-to-machine" communications link such as vehicle-to-vehicle (V2V), vehicle-to-exchange (V2X), or device-to-device (D2D). In the latter case, a second "slave" device is synchronized to the first "master" device by the methods of the present disclosure.

[0067] The radio receiver may be embodied by any kind of radio receiver, and is not limited to receivers implementing a Phase Locked Loop (PLL) to retrieve the symbol values, but may also retrieve the symbol values ​​by exploiting relative phase changes (i.e. by implementing a Frequency Locked Loop - FLL) or by implementing any other type of demodulation technique aimed at estimating the M-ary symbol values. In yet another alternative embodiment of the present invention, the radio receiver (RX1) implements a phase locked loop to recover the phase of the radio signal. In yet another alternative embodiment of the present invention, the radio receiver (RX1) implements a frequency locked loop to extract the phase changes of the radio signal. The present invention is further explained by the following description and the accompanying drawings. [Brief description of the drawings]

[0068] [Figure 1] 1 illustrates a system for resolving time ambiguities in a radio navigation system that includes multiple radio transmitters and a radio transmitter and a radio receiver. [Diagram 2] 2 shows functional elements of a radio transmitter TX1 and a radio receiver RX1 according to an embodiment of the present invention. [Diagram 3] This shows how to reference the pseudoranges corresponding to four satellites based on "common reception" to calculate the GNSS receiver position. [Figure 4] The impact of synchronization errors on pseudorange estimation and final position accuracy is illustrated and explained, and a mitigation technique based on the use of a fifth line of sight to address the synchronization errors is presented. [Diagram 5] A concept is presented for deriving the position of a signal snapshot of a transmitted overlay sequence based on a look-up table (or repository) relative to an implicit time marker that is regularly placed at the beginning of the overlay sequence. [Figure 6] 1 shows a signal structure including an overlay sequence modulated onto a primary code. [Figure 7] A table containing examples of "de Brown" sequences as overlaid binary sequences is shown. [Figure 8] We present a so-called hard decoding method based on partial autocorrelation between a zero-padded subset of N=5 symbols taken from the signal snapshot and the overlay sequence to resolve time ambiguities. [Figure 9] We present a so-called soft decoding method based on partial autocorrelation between a zero-padded signal snapshot sequence and an overlay sequence to resolve time ambiguities. [Figure 10] A method based on implicit time markers for snapshot positioning to resolve synchronization between a first time scale as one of the GNSS and a second time scale as one of the receivers that may be synchronized to the network is shown. [Figure 11] This paper shows the shortcomings of a method based on implicit time markers for snapshot positioning to resolve synchronization between a first time scale as one of the GNSS and a second time scale as one of the receivers that may be synchronized to the network when the time ambiguity interval is shorter than the synchronization between the first time scale and the second time scale. [Figure 12] We present a table expressing the relationship between the time ambiguity interval and the overlay symbol duration as a function of snapshots for different values ​​of the de Brown sequence length L and the number of overlay symbols in a snapshot N. [Figure 13] We show the application of the concept of multiple de Brown sequences transmitted by the same source, e.g., a satellite, to improve the time ambiguity interval. Here, two de Brown sequences are shown, where the second one is obtained from the first one per single symbol truncation. [Figure 14]FIG. 1 illustrates the time ambiguity interval achieved when processing two constructive overlay sequences having different lengths and transmitted by two different components from the same satellite, where the constructive second overlay sequence is truncated by K symbols relative to the first overlay sequence as a function of the number of symbols included in the snapshot. [Figure 15] We show the case where a sequence of P=8 symbols taken from a signal snapshot and corrupted by a demodulation error (here Nerr=8 corrupted symbols) occurs at another position in the overlay sequence. [Figure 16] We show the case where a sequence of P=8 symbols taken from a signal snapshot and corrupted by demodulation errors (here Nerr=8 corrupted symbols) does not occur at any other position in the overlay sequence. [Figure 17] We show the case where an extended subset of P=10 symbols taken from a signal snapshot and corrupted by one demodulation error does not occur anywhere else in the overlay sequence and can be corrected by evaluating the Hamming distance between this corrupted subset of length P=10 and any subset sequence of the original overlay sequence. [Figure 18] We show the values ​​obtained by correlating a snapshot containing the first N=7 symbols of the overlay sequence with any subset sequence of N=7 symbols in the overlay sequence when the overlay sequence is not truncated (length L=128) or when the overlay sequence is truncated by 8 symbols (length L=120). [Figure 19] Represents a satellite-to-user device geometry that allows for estimating the minimum duration of an overlay symbol to provide time synchronization with different overlay sequences transmitted by different satellites. [Figure 20]1 shows a flowchart describing a method and steps used to identify an overlay sequence to be modulated onto a signal processed in a receiver implementing FLL based on a truncated transition sequence to result in an integrated de Brown sequence modulated onto a signal carrier. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0069] The present invention will be described with respect to particular embodiments and with reference to certain drawings but the invention is not limited thereto but only by the claims. The drawings described are only schematic and are non-limiting. In the drawings, the size of some of the elements may be exaggerated and not drawn to scale for illustrative purposes. The dimensions and relative dimensions do not necessarily correspond to reductions to actual implementations of the invention.

[0070] Moreover, the terms first, second, third, etc. in the specification and claims are used to distinguish between similar elements and are not necessarily intended to describe a sequential or chronological order. These terms are interchangeable under appropriate circumstances, and embodiments of the invention may operate in sequences other than those described or illustrated herein.

[0071] Moreover, terms such as top, bottom, upper, lower, etc. in the specification and claims are used for descriptive purposes and not necessarily for describing relative positions. The terms so used are interchangeable under appropriate circumstances, and the embodiments of the invention described herein may operate in orientations other than those described or illustrated herein.

[0072] The term "comprising" used in the claims should not be interpreted as being limited to the means listed thereafter, and does not exclude other elements or steps. It should be interpreted as specifying the presence of the mentioned and described features, integers, steps or components, but without excluding the presence or addition of one or more other features, integers, steps or components, or groups thereof. Thus, the scope of the expression "a device comprising means A and B" should not be limited to a device consisting only of components A and B. This means that in the context of the present invention, the only relevant components of the device are A and B.

[0073] In the following paragraphs, an implementation of a system for resolving time ambiguities between a radio transmitter and a radio receiver in a radio navigation system according to an embodiment of the present invention is described with reference to the drawing in Fig. 1. In the further paragraphs, all connections between the mentioned elements are defined. Next, multiple radio transmitters TX1...TX x and all associated functional means of a radio transmitter according to an embodiment of the invention, followed by a description of all the interconnections of these functional means. In the following paragraphs, a practical implementation of a system for resolving time ambiguities between a radio transmitter and a radio receiver in a radio navigation system according to one embodiment of the present invention is described. A radio navigation system uses multiple radio transmitters (TX1...TX x ), each radio transmitter being configured to transmit a radio signal by radio signal to at least one radio receiver RX1 of the radio navigation system, inter alia for navigation and synchronization purposes. Such a radio transmitter may be a GNSS transmitter, which is a satellite transmitting a radio navigation signal, or a satellite part of a satellite communication network, or a pseudolite, or a transmitting device implemented in a terrestrial communication network, for example a base station (BTS), a fixed or mobile radio transmitter in the case of a wireless communication network, or a device implemented in a V2V or V2X communication network. Such a radio receiver is not limited to a receiver that derives the binary values ​​by implementing a Phase Locked Loop (PLL), but may be a GNSS receiver embodied by any kind of radio receiver that is also able to derive the binary values ​​by exploiting relative phase changes (i.e. by implementing a Frequency Locked Loop - FLL) or by implementing any other type of demodulation technique aimed at estimating the M-ary symbol values.

[0074] Such a radio receiver may be a GNSS receiver integrated into a user device, such as a navigation device or a personal mobile device like a smartphone, which is a device comprising a processor coupled to a memory and interface means such as a display and a keyboard. Such mobile computing devices are configured to have installed thereon many different types of applications, the execution of each such application intended to perform a different type of task, such as navigation. The radionavigation system according to an embodiment of the invention may be a satellite radionavigation system such as the Global Navigation Satellite System GNSS, or a single positioning beacon, such as a pseudolite, or a network of positioning beacons, or it may be a terrestrial system such as a wireless communication network requiring synchronization from user terminals.

[0075] An alternative embodiment of such a system according to the invention may be an application where the first time scale is shared by a terrestrial communication network or a system transmitting signals via base stations or beacons, or where the first time scale is shared by another communication-enabled device, for example in a "machine-to-machine" communication link such as vehicle-to-vehicle (V2V), vehicle-to-exchange (V2X), or device-to-device (D2D). In the latter case, the second "slave" device is synchronized to the first "master" device by the method of the present disclosure.

[0076] The first essential element of a radio navigation system is a number of radio transmitters TX1...TX x The radio transmitter TX1 may comprise a transmitting means 12 configured to transmit a radio signal via a radio network RN to a radio receiver for navigation and synchronization purposes, the transmitting means 12 comprising a transmitting means 12 configured to transmit a radio signal via the radio network RN to a radio receiver, the transmitted radio signal comprising an overlay sequence such as a de Brown sequence modulated onto a carrier wave of the radio signal, or a truncated de Brown sequence, or an integrated de Brown sequence, or a combination of two or more de Brown sequences and / or truncated sequences and / or integrated de Brown sequences.

[0077] Such carrier signals may employ waveforms to modulate a primary code with, for example, Binary Phase Shift Keying (BPSK) for GPS C / A signals, or Binary Offset Carrier (BOC) for Galileo E1-B / -C. FIG. 6 shows a typical GNSS signal structure including the overlay sequence of the present disclosure. At the top of the figure, an example of a binary overlay sequence including 32 overlay symbols is shown. Here, logic levels [0,1] are used to represent the corresponding symbols. This overlay sequence can then be represented by signal levels [1,-1] corresponding to the logic levels, as shown below. Each symbol of this overlay sequence is then modulated or spread with a primary code including chips. Note that in the case of the Galileo signal structure, the secondary code plays the role of the overlay sequence. Finally, the overlay symbol duration T s and chip duration T c is also shown. The type of waveform modulated onto each chip is not shown in Figure 6. This may be, for example, binary phase shift keying (BPSK) for GPS C / A signals, or binary offset carrier (BOC) for Galileo E1-B / -C. The use of the overlay sequence for synchronization is independent of the type of waveform modulation.

[0078] FIG. 10 introduces elements that are useful for understanding the invention of this disclosure. In FIG. 10, an example of the value of the synchronization error ΔT is shown. In this case, the GNSS satellite is considered to transmit a signal that includes an implicit time marker, ITM. An implicit time marker differs from an explicit time marker in that it does not encode the time of transmission in the navigation message. However, both implicit and explicit time markers aim to provide information about the time of transmission. The telemetry words (TOW and HOW) encoded in the GPS navigation message are an example of an implicit time marker. Implicit time markers rather utilize overlay sequences, i.e., repeating binary sequences that can be modulated onto the primary code, providing indirect time information about the transmission of the message. These overlay sequences are periodic, and the ITM is also repeated at different positions within the entire signal transmitted by the GNSS satellite. Nevertheless, the position of the ITM within each overlay sequence can be unambiguously defined according to rules. The local position of the ITM in the overlay sequence generated by the receiver on its receiver time scale, also called the second time scale, is identifiable according to the synchronization error ΔT, with a span ±ΔT referenced to the GNSS time scale (e.g. GPST for GPS and GST for Galileo), also called the first time scale. max Note that each receiver (potentially synchronized to a different network) will generate a different local ITM position. Only one local position is shown, highlighted with a thick black border, representing the correct position that would be obtained if the receiver were perfectly synchronized to the GNSS time scale. The user device receives and processes the signal snapshots delimited by the bold dashed box, and from processing the signal snapshots it is possible to determine the relative positions of the ITMs of the wireless signal represented in the first time scale based on the positions of the subset of symbols contained in the snapshots within the overlay sequence. It should also be noted that since the receiver has already acquired the signal and is in tracking mode, it is synchronized to the received signal with primary code period granularity, assuming that the overlay symbol duration is limited to an integer multiple of the primary code period. Any position of the ITM is therefore represented on the receiver time scale with the granularity of the symbol duration. The difference in the relative position of the ITM position, represented on the receiver time scale, to the position of the ITM derived from the signal snapshot allows to identify and resolve the synchronization error ΔT. This alternative approach based on the transmission of a GNSS signal including an ITM makes it possible to avoid "sacrificing" one line of sight from which a fifth pseudorange can be derived, as in the previous approach disclosed for A-GNSS. This allows to improve the availability of the positioning service. Since the GNSS signal is transmitted continuously, the implicit time markers are repeated and transmitted periodically. Thus, the time ambiguity still persists, as shown in the upper part of Figure 10. The distance between the repeated ITMs is defined as the Time Ambiguity Interval (TAI). Here, the objective is to achieve a maximum span of the synchronization error 2 × ΔT max The aim is to increase the TAI value as much as possible beyond ΔT. For GPS C / A signals, TAI is expressed in milliseconds (1 ms if only the spreading code sequence is considered, 20 ms if the symbol edge is considered). TAI shall be expressed in seconds, with 2×ΔT being the minimum for unambiguous time synchronization. max span. To understand the design constraints that guarantee unambiguous time synchronization, Fig. 11 represents the situation when the time ambiguity interval is shorter than the synchronization error span. For the same snapshot position, the "relative" ITM derived from the received signal will be placed at a different position than the "absolute" ITM, leading to errors in synchronization. This shows why it is essential that the time ambiguity interval must be larger than the synchronization error span.

[0079] The overlay sequence comprises a set of L symbols for each time ambiguity interval, each of which comprises an implicit time marker. The transmitting means may be a GNSS transmitter, a positioning beacon transmitter such as a pseudolite or a satellite in a communication network, or a networked vehicle in a V2V / V2X architecture, or a fixed or mobile radio transmitter in case of a wireless communication network having a first time scale. Such overlay sequences may include, but do not essentially include, binary symbols, or other non-binary sequences, i.e., any kind of M-ary symbols, may be used to implement the overlay sequences. Additionally, the overlay sequence may include, but does not essentially include, real symbols, or other complex symbols may be used to embody the overlay sequence. The radio transmitter TX1 further comprises signal processing means 11 configured to generate an appropriate intended radio navigation signal, which includes an overlay sequence satisfying the condition of a single occurrence of a subset of N symbols within a plurality of L symbols of a time ambiguity interval. Such signal processing means 11 may comprise a microprocessor, in particular for processing the signals to be transmitted, and the processing means may further comprise a memory device coupled to the microprocessor for storing electronic information such as computer instructions, results of the signal processing including final and intermediate results, as well as further information. The signal processing means 11 may be configured to generate a de Brown sequence modulated onto a carrier wave of a radio signal, or a truncated de Brown sequence, or an integrated de Brown sequence, or an overlay sequence consisting of a combination of two or more de Brown sequences and / or truncated sequences and / or integrated de Brown sequences. The radio transmitter TX1 further comprises transmitting means 12 arranged to transmit the radio navigation signal generated by the signal processing means 11. In addition, the radio transmitter TX1...TX x has the same functional configuration as the wireless transmitter TX1. The radio receiver RX1 is configured to resolve time ambiguity between a radio transmitter having a first time scale and the radio receiver RX1 having a second time scale based on a radio signal transmitted by a radio transmitter of the multiple radio transmitters and received at the radio receiver RX1.

[0080] The radio receiver RX1 firstly comprises signal receiving means 21 arranged to receive a radio signal which is a GNSS radio signal transmitted by a radio transmitter TX1. The radio receiver RX1 may be any kind of device incorporating a GNSS receiver and synchronised to its second time scale, which may be based on a local clock or on the clock of a communication network to which the device is connected. The radio receiver RX1 is not limited to a receiver that derives the binary values ​​by implementing a Phase Locked Loop (PLL), but may be embodied by any kind of radio receiver that is also able to derive the binary values ​​by exploiting relative phase changes (i.e., by implementing a Frequency Locked Loop - FLL) or by implementing any other type of demodulation technique aimed at estimating the M-ary symbol values. The radio receiver RX1 further comprises a snapshot capture means 22 configured to obtain a snapshot of the radio signal received from the radio transmitter TX1, and a signal processing means 23 configured to determine a relative position of an implicit time marker expressed on a first time scale within the radio signal based on a position of a subset of N symbols included in the snapshot within a set of L symbols of the overlay sequence of the time ambiguity interval.

[0081] The processing means 23 of the radio receiver RX1 is further configured to determine the relative position of the implicit time marker in the radio signal by searching a subset of the symbols of the snapshot in entries of a repository, the repository comprising for each entry a number of symbols of the N symbols taken of the snapshot and a relative position of a number of symbols of the N symbols of the snapshot relative to the implicit time marker within a time ambiguity interval of the radio signal. The radio receiver may additionally or alternatively comprise a snapshot sequence generating means 24 configured to generate a snapshot sequence corresponding to the radio signal transmitted by the radio transmitter. In a first option, the snapshot sequence may be generated from a noisy snapshot signal of the radio signal, where both the Doppler offset and the code delay estimated from the acquisition process are wiped off and finally completed with 0 samples. Alternatively, in a second option, the snapshot sequence may be generated with N retrieved symbols contained in the snapshot of the radio signal, completed with 0. Furthermore, the processing means 23 of the radio receiver RX1 is configured to determine the relative position of the implicit time marker expressed in the first time scale in the radio signal by partially autocorrelating the snapshot sequence with a spread overlay sequence corresponding to the snapshot sequence, the length of which is equal to the overlay sequence multiplied by the number of samples per symbol duration, if the signal snapshot is generated according to the first option, or by partially autocorrelating the snapshot sequence with an overlay sequence of length L corresponding to the snapshot sequence, if the signal snapshot is generated according to the second option. The snapshot capture means 22, the processing means 23, the snapshot sequence generation means 24 and the repository 25 may further comprise hardware, software or any combination thereof, such as a microprocessor with associated electronic memory for storing instructions, results and intermediate results of the processing of received radio signals. This may be a local processor with associated memory for performing all functions or may be dedicated to each mentioned function. The sequence generating means 11 of the radio transmitter TX1 is coupled at an output-terminal to an input-terminal of transmitting means 12, which has an output-terminal that is also an output-terminal O1 of the radio transmitter TX1. The radio receiver RX1 has an input-terminal I1 which is also an input-terminal of receiving means 21 which is coupled at an output terminal to an input-terminal of snapshot capturing means 22 which is coupled at an output terminal to an input-terminal of processing means 23. The snapshot sequence generating means 24 is coupled at an output terminal to an input-terminal of the processing means 23.

[0082] To illustrate an embodiment of the present invention, it is assumed that at least one radio transmitter TX1 configured to resolve time ambiguities between a radio transmitter TX1 having a first time scale and a radio receiver RX1 having a second time scale first generates, by means of a signal generating means 11, an overlay sequence satisfying the condition of a single occurrence of a subset of N symbols of a plurality of L symbols over the entire time ambiguity interval. This overlay sequence is characterized in that it comprises a set of L symbols for each time ambiguity interval and that each of the time ambiguity intervals comprises an implicit time marker. The length of such an overlay sequence is a predefined length L. The resolution of the time ambiguities can then be used internally in the device, for example to estimate the position and time of the device based on the time ambiguity resolved ranging signal, or externally in the device, for example to indicate the timing of the timing receiver device and to provide an output O2. Such a radio signal is then generated by modulating the generated overlay sequence onto a carrier wave of a radio signal, and the generated radio signal is then broadcast by the transmitting means 12 via the coupled radio network RN towards the at least one radio receiver RX1, this broadcasted radio signal then comprising the generated overlay sequence modulated onto the carrier wave of the radio signal. Alternatively, the overlay sequence may be modulated onto a primary code that comprises chips modulated onto a carrier wave of a radio signal. The overlay sequence contains a set of symbols for each time ambiguity interval, each of which contains an implicit time marker, whose position within the time ambiguity interval is known (according to a rule) and can be, for example, the first symbol of the sequence. Subsequently, the radio receiver RX1 receives the transmitted radio signal including the generated overlay sequence modulated onto the carrier of the radio signal by the receiving means 21. The overlay sequence is characterized in that it includes a set of L symbols for each time ambiguity interval and each of the time ambiguity intervals includes an implicit time marker. The length of such an overlay sequence is a predetermined length L. The snapshot capture means 22 obtains a snapshot of the overlay sequence retrieved from the received radio signal. Upon reception of the radio signal, the received signal snapshot is demodulated to retrieve from the received radio signal a subset of N symbols in the overlay sequence modulated onto the carrier signal. The snapshot of the overlay sequence included in the received radio signal includes a predetermined amount of N symbols, which is smaller than the amount of L symbols included in the overlay sequence as shown in FIG. 5. Furthermore, the processing means 23 determines the relative position of the implicit time marker, represented in the first time scale of the radio signal, based on the position of the subset of symbols contained in the snapshot within the set of symbols of the time ambiguity interval. The snapshot captures N, for example N=5, symbols from an overlay sequence comprising L symbols, where L is for example 32 symbols. Since the overlay sequence is characterized by a property (including a cyclic property) that guarantees that any subsequence of length N occurs once within a sequence of length L based on the subset of N symbols, the position of this mentioned subset within this set of L symbols of the time ambiguity interval of the corresponding overlay sequence can be determined by this property. As an option for determining this position, the processing means 23 determines the relative position of the implicit time marker in the wireless signal by searching a subset of, e.g., N=5, symbols of the snapshot in an entry of the repository 25. This repository 25 may comprise a table or database which contains, for each table or database entry, a number N of subsequent symbols contained in the snapshot together with the relative position of the symbol of the snapshot with respect to the implicit time marker in the time ambiguity interval of the wireless signal.

[0083] Based on the extracted symbol combination "01001" as contained in the snapshot (see Figure 5), it is possible to extract the relative position of the implicit time marker expressed in the first time scale of the wireless signal based on the position of the subset of symbols contained in the snapshot within the set of symbols of the time ambiguity interval, and therefore it is possible to resolve the time ambiguity. The table or database of repository 25 may include, for each entry in the table or database, a number N of subsequent symbols contained in the snapshot, together with information about the relative positions of these symbols contained in the snapshot within the time ambiguity interval.

[0084] In another related alternative embodiment, the radio receiver RX1 generates a snapshot sequence corresponding to the radio signal transmitted by the radio transmitter TX1 by a snapshot sequence generating means 24, where in a first option, the radio receiver RX1 generates the snapshot sequence from noisy samples of the radio signal by completing with “0” samples after wiping off both the Doppler offset and the code delay derived from the snapshot signal and estimated from the acquisition process, or in a second option, the radio receiver RX1 generates the snapshot sequence by concatenating a subset of the removed symbols of the N symbols comprised in the snapshot of the radio signal with another subset of LN “0”s obtained with 0 padding to complete the snapshot sequence of length L.

[0085] The processing means 23 of the radio receiver RX1 then determines the relative positions of the time markers in the radio signal by (partial) autocorrelating the generated snapshot sequence with a complete overlay sequence comprising a number of samples corresponding to the number of samples comprised in the snapshot sequence in order to estimate the position of the subset of N symbols comprised in the snapshot within the full set of L symbols allowing time ambiguity resolution, when considering the second option for snapshot sequence generation based on extracted symbols, as shown in FIG. 8, or the first option for snapshot sequence generation based on samples derived from the snapshot signal, as shown in FIG. 9. It is advantageous to apply this partial autocorrelation when the number of symbols in the overlay sequence is too large (e.g., when N=7, L=2^N=128 symbols), and then preferably identify the relative position of the N symbol snapshot to the implicit time marker (start of the overlay sequence) by using a partial autocorrelation of N=7 removed symbols in the 128 symbol overlay symbol stream, completed with 128-7=121 symbols set to 0. This solution is introduced when N is large to avoid oversized lookup tables (repositories) using excessive storage space memory and excessively long lookup times if the tables maintained by such repositories are too large. A first approach to generate a snapshot sequence for the partial autocorrelation process consists of completing a subsequence of N extracted symbols with LN "0", i.e. zero-padding, for example using a PLL or FLL implementation. It has been shown that the performance of the extraction of symbols from the snapshot is significantly improved if the code delay and carrier Doppler offset obtained from the acquisition step are first wiped off from the snapshot signal before applying the extraction demodulation step. In this first option, one zero is applied per symbol. This snapshot sequence of L symbols is associated with a complete overlay sequence of L symbols. The position of the subsequence that results in the maximum partial autocorrelation is then used to place the snapshot subsequence at the beginning of the overlay sequence. The principle of this first approach is illustrated in FIG. 8 for the special case of N=5. Also shown are the partial autocorrelation values ​​obtained at the boundaries (for k=1,2,3,4).

[0086] A second approach for generating the snapshot sequence consists in obtaining preprocessed samples directly from the snapshot signal, i.e. without symbol take-off demodulation, and completing it by padding the corresponding samples again with zeros. The preprocessing step consists in wiping off the Doppler estimated from the acquisition step (i.e. by "derotation"). Furthermore, different options for the type of samples to be assumed for the signal snapshots can be disclosed. The first option assumes raw "I / Q samples", which, when derotated with the application of Doppler, are measured with an effective sampling frequency equal to the sample rate, resulting in a snapshot sequence with a large amount of samples, which is difficult to support processing in low-power devices. Another option assumes samples after correlation (correlation performed in the acquisition stage), also derotated with Doppler, in which case the effective sampling frequency is reduced to the primary code rate, resulting in a much smaller number of samples. It should be noted that in this second option, the number of zeros padded per symbol must assume the effective sampling frequency. This second approach is particularly suitable when the overlay sequence is modulated onto a primary code modulated onto the radio signal. The principle of this second approach is illustrated in FIG. To identify the corresponding peaks of the partial autocorrelation, implementations similar to those used for GNSS signal acquisition may be disclosed. One possible implementation relies on the use of self-generated snapshots and serial correlation between the padded sequence and the overlay sequence, where each symbol position is tested consecutively. Another possible implementation relies on the use of FFT, in such a way exploiting the periodicity property of the overlay sequence. As explained above, a snapshot subsequence of N symbols is first zero-padded to generate a snapshot sequence to reach length L. Then, the partial autocorrelation ACF p The following formula is applied: JPEG2024535752000017.jpg15163In formula: - FFT and IFFT stand for Fast Fourier Transform and Inverse Fast Fourier Transform, respectively. -Conj represents the conjugate operation. -Seq Overlay denotes a binary overlay sequence of length L. -Seq N,0 represents a zero-padded snapshot sequence.

[0087] A related embodiment relates to a method in which the overlay sequence modulated onto the wireless signal comprises a de Brown overlay sequence. Such an overlay sequence, consisting of a de Brown sequence or "de Brown" overlay sequence, guarantees a single occurrence of any subsequence of length N within an overlay sequence of length L (including on the boundaries). This property satisfied by "de Brown" sequences is referred to as the single occurrence of N symbols within L symbols or the SO(N,L) property. As a result of the SO(N,L) property fulfilled by the "de Brown" sequence, the position of this unique symbol sequence within the time ambiguity interval of radio signals, such as GNSS signals, or any kind of terrestrial radio signals, such as signals transmitted by satellites in a communication network, or radio signals transmitted by pseudolites, or radio signals transmitted by transmitting equipment of a terrestrial communication network, such as a Base Station Transmitter (BTS), a fixed or mobile radio transmitter in the case of a wireless communication network, or a radio signal transmitted by a device implemented in a V2V or V2X communication network, can be unambiguously determined and, based on the position of this unique sequence, the (relative) distance between the unique symbol sequence contained in the snapshot and the position of an implicit time marker can be precisely determined.

[0088] Furthermore, the overlay sequence may be a de Brown sequence modulated onto a carrier wave of the radio signal, or a truncated de Brown sequence, or an integrated de Brown sequence, or a combination of two or more de Brown sequences and / or truncated sequences and / or integrated de Brown sequences. Such a "de Bruijn" overlay sequence may include binary symbols, but essentially does not. Alternatively, other non-binary sequences, i.e., M-ary sequences, may be applied to implement the de Bruijn sequence. Furthermore, the overlay sequence may include real symbols, but essentially does not. Alternatively, other complex symbols may be used to implement the overlay sequence. Furthermore, the "de Bruijn" sequence is also a subsequence of length N constructed by concatenating the k (k < N) last symbols of the sequence with the first [N - k] symbols, and also satisfies the cyclic property that guarantees a single occurrence within the complete "de Bruijn" sequence. One important property of the "de Bruijn" sequence is the large (L / N) = (2 N / N) ratio, which represents an excellent advantage for snapshot positioning. In fact, this means that for a small number N of symbols (i.e., short snapshot duration), the overlay sequence length (i.e., time ambiguity interval) can be large.

[0089] Some examples of de Bruijn sequences of different lengths L are shown in the table of FIG. 7 for illustration. Therefore, in an advantageous embodiment of the present invention, at least one wireless receiver RX1 is configured to resolve the time ambiguity between a wireless transmitter TX1 having a first time scale and the wireless receiver RX1 having a second time scale. The wireless transmitter RX1 first generates an overlay sequence based on a "de Bruijn" sequence that satisfies the condition of single occurrence of a subset of symbols within a plurality of symbols over the entire time ambiguity interval by means of signal processing means 11. This overlay sequence is characterized in that it is based on the "de Bruijn" sequence, includes a set of L symbols for each time ambiguity interval, and each of the time ambiguity intervals includes an implicit time marker. The length of such an overlay sequence is a predetermined length L.

[0090] Below, we describe a method for the design, dimensioning, and selection of de Brown sequences, but also highlight the potential performance that can be gained by applying de Brown sequences. We show how to achieve the same performance for different values ​​of N and L (L=2^N), overlay symbol duration (T s ), and snapshot duration (T Snp = N × T s ) are also disclosed and illustrated in FIG. 12. The corresponding action points are shown in the table of FIG. 12. Such action points are selected with the main constraint to have a signal snapshot duration shorter than 500 ms, which is easy to support the processing of low power devices.

[0091] The table in FIG. 12 shows that it is possible to ensure that the time ambiguity interval (TAI) is larger than the typical synchronization error (±2 s) of a receiver with a snapshot duration of 384 ms. The snapshot duration (T Snp ) is the duration T before and after the N overlay symbols. Grd It is important to note that we do not assume two signal processing time guards. These time guards are actually needed to estimate the polarity (in the case of PLL processing) or polarity change (in the case of FLL processing) of the corresponding de Brownian subsequence. Grd is essentially dependent on the received signal-to-noise power spectral density ratio (C / N0) and varies from 30 to 40 dB-Hz for typical GNSS applications and decoding techniques (PLL or FLL based). Grd Typical magnitude scales vary from a few milliseconds up to tens or even tens of milliseconds. From the above exemplary configurations, the following relationships between the main requirements and design parameters can be deduced: JPEG2024535752000018.jpg9163JPEG2024535752000019.jpg9163Note that in the above formula, the division by a factor of 2 is to express TAI as "one-sided" (e.g. ±0, 16 seconds). If TAI is expressed as a "span" (e.g. 0, 32 seconds), this division by a factor of 2 disappears.

[0092] As a result, the time ambiguity interval (TAI), symbol duration (T s ), and the processing time guard duration (T Grd ) is specified as a requirement, it is possible to easily estimate the length L of the de Brown sequence and therefore the number of de Brown symbols in the snapshot according to Note that in the above formula, the de Brownian sequence length, which is directly related to the TAI, is expressed here as a span (e.g., 0, 32 seconds) and not as a side (e.g., ±0, 16 seconds). JPEG2024535752000021.jpg12169 specifies the ceiling function (the smallest integer greater than or equal to x). s = 40 ms, L optim =2^( JPEG2024535752000022.jpg11169=8) and N optim = 3. Time Guard lasts for T Grd = 4ms, the snapshot time becomes 128ms.

[0093] Another design scenario is to reduce the time ambiguity interval (TAI), symbol duration (T s ), and the processing time guard duration (T Grd ) is given and we need to estimate the de Brown sequence length L, and the number of symbols in the snapshot N. Then, we can use (14) and (15) again to find the two unknowns T s and we obtain a system of simultaneous equations for N: JPEG2024535752000023.jpg9170JPEG2024535752000024.jpg9170This system can be reduced to a single equation for the unknowns N: JPEG2024535752000025.jpg9170 Solution for number of symbols per snapshot N optim If there exists an optimal symbol duration T s,optim can be easily estimated from (Equation 14). Now that the main principles for applying "de Brown" sequences have been described, we disclose presenting additional applications that offer higher performance based on the combination of several "de Brown" sequences.

[0094] In another alternative and advantageous embodiment of the invention, each GNSS signal transmitted by the same satellite comprises two (or more) signal components, each modulated with a different constituent "de Brown" sequence, resulting in two (or more) constituent "de Brown" sequences transmitted by the same satellite. The corresponding constituent "de Brown" sequences, when combined, form an aggregated overlay sequence. In the following, V denotes the number of constituent "de Brown" sequences transmitted by the satellite.

[0095] In the following, we consider for illustration the special case of two signal components (V=2), each modulated with a constructive "de Brown" sequence, resulting in two constructive "de Brown" sequences transmitted by the same satellite. One possible implementation consists in modulating these sequences with two different primary code streams, which can be in quadrature or in phase, and thus two different signal components. In a subcase of this embodiment, the first constituent sequence is considered to be an untruncated de Brown sequence of length L1, also called the original de Brown sequence, and the second constituent sequence is considered to be a truncated de Brown sequence of length L2=L1-1. The latter is obtained by removing one bit, for example the last bit, from the original de Brown sequence of length L1. In the example of the present disclosure shown in FIG. 13, L1=32 symbols and L2=32-1=31 symbols. In this example, both sequences have the same duration (T s =T s1 =T s2 ) symbols. Both sequences are shown in the upper part of Fig. 13a. In Fig. 13b, the GNSS signal is represented over a long period of time and shows two overlay sequence streams, the first obtained by concatenating a first constitutive overlay sequence of length 32 symbols and the second by concatenating a second constitutive overlay sequence of length 31 symbols. In this example, it is also conceivable that the edges of corresponding symbols are transmitted synchronously. On reception, the untruncated and truncated constitutive "de Brown" sequences can be combined per "juxtaposition" into a so-called aggregated overlay sequence, which can be seen as a sequence containing complex symbols (assuming symbols modulated onto in-phase and quadrature-phase signal components) or, alternatively, a sequence whose quaternary symbols are of the same duration T sand can be viewed as a quaternary sequence that can be associated to the combined binary symbols of the untruncated and truncated constructive de Brown sequences as follows, where the quaternary symbol "0" represents "00", "1" represents "01", "2" represents "11" and "3" represents "10". In this notation, the first binary symbol "a" of the pair "ab" arises according to the rule from the first (untruncated) constructive de Brown sequence and the second binary symbol "b" arises according to the rule from the second (truncated) constructive de Brown sequence. In the example of the present disclosure shown in FIG. 13b, the duration N×T s It is assumed that a user device processes snapshots of a signal having N=5 binary symbols of each sequence, and thus including 5 quaternary symbols of the aggregated overlay sequence. The difference in the lengths of the constituent overlay sequences ensures that a corresponding snapshot does not occur more than once in an aggregated overlay sequence of length L=L1×L2=992 symbols. This aggregated overlay sequence length corresponds to the periodicity of the aggregated overlay sequence, expressed in units of quaternary symbols. The period of the aggregated overlay sequence, expressed in seconds, is L×T s This period corresponds to an extended TAI. As an example, the overlay symbol duration is equal to T s1 =T s2= 44 ms, then the TAI is 992 × 44 ms = 43.648 s. Furthermore, the location of the implicit time marker of this aggregated overlay sequence can again be defined according to a rule at the location in the aggregated overlay sequence where the first symbol of the first sequence and the first symbol of the second sequence are coincident and in phase. This implicit time marker again serves to estimate the relative position of the snapshots in the second time scale. Identification of the relative position of the implicit time marker of the wireless signal based on the position of a subset of symbols contained in the aggregated overlay sequence snapshots may be based on a lookup in a repository, as in the case of transmitting and processing a single de Brown-based sequence, or based on calculating a partial autocorrelation function.

[0096] Based on the exemplary configurations of the present disclosure, the following design scenarios may be disclosed. First, consider that the extended TAI is obtained by a single occurrence within a snapshot of a combination of N symbols as part of an original constructive de Brown sequence of length L1 modulated with a first signal component and N symbols as part of a second constructive de Brown sequence of length L2=L1-K obtained from the original constructive de Brown sequence by truncating K symbols and modulating with a second signal component. Furthermore, consider that the same symbol duration T s Based on these design assumptions, the (extended) TAI satisfies the following set of equations: JPEG2024535752000026.jpg8167JPEG2024535752000027.jpg8167In contrast to the design represented by equations (14) and (15), based on a single de Brown sequence, two other degrees of freedom are introduced. The first corresponds to the number K of truncated symbols from the original de Brown sequence, while the second, which varies between 1 and (L11), corresponds to the position of the truncated symbols in the original "de Brown", assuming the constraint that the K truncated symbols are adjacent, thus preserving the property of the original "de Brown" once truncated. Following a mathematical derivation similar to that described in the case of the single de Brown design, several solutions for L1 (which yields N), K and truncated symbol positions can be found. One optimal solution (L 1,optim , K optim , as well as the optimal truncated symbol position) favors the smallest N to reduce snapshot duration.

[0097] The table shown in Figure 14 shows the symbol duration T s We calculate the extended TAI when we assume that the number of truncated symbols is K, and the columns indicate the original de Brown sequence length L1 (derived from N as L1 = 2^N). Each cell indicates the TAI based on equation (21). In this example, the position of the K truncated symbols is at the end of the original de Brown sequence. In this example, assuming a minimum required TAI of 5.12 seconds, the table in FIG. 14 shows that two configurations can meet this TAI, using either (L1=16, K=8) or (L1=32, K=28). In this case, the first configuration (L 1,optim =16,K optim = 8) is chosen to guarantee a minimum snapshot duration (without time guard), with N equal to 4 × 40 ms = 160 ms.

[0098] The optimal configuration is designed for symbol duration T sIf not provided, then a parameter L that satisfies TAI with the minimum snapshot duration constraint (assuming a time guard) is 1,optim (Thus, N optim ), K optim , the truncated symbol position and T s,optim A parameter analysis should be performed to determine the symbol duration T s In this case, the minimum snapshot duration (L1, K, T s and the truncated symbol position) is the optimal solution (L 1,optim , K optim , T s,optim ) is held for The principle described in Figures 13a and 13b used an original de Brown sequence of length L1 = 32 symbols, which is truncated by one symbol to form a second overlay and a shorter sequence. This principle can be extended to other values ​​of the length L1 of the original de Brown sequence, for example L1 = 64, 128, .... Furthermore, the principle can be generalized in the following subcases of the embodiment: - a combination of two constructive "de Brown" sequences, each constructed using two different elementary de Brown sequences of the same length, where the first sequence is based on a first elementary de Brown sequence without truncation and the second sequence is based on another elementary de Brown sequence with a truncation of K symbols. -The combination of two de Brownian sequences is constructed from original sequences of different lengths. For example, the first sequence is constructed from an original de Brownian sequence of length L1 = 32, and the second sequence is constructed from an original de Brownian sequence of length 2 4 It can be thought of as being generated by truncating K symbols from the original de Brownian sequence of length L2 = 16 − K. - If the second sequence is obtained by truncation from the first sequence, the number K of truncated symbols may vary between 1 and (L1-1). Nevertheless, to effectively guarantee the extended TAI, care must be taken to ensure that the lengths of both sequences after the possible truncation are not multiples of each other. For example, assuming two original sequences of length L1=L2=32 symbols, if 16 symbols are truncated from the second original sequence of length L2=32, the aggregated overlay sequence obtained by combining the first original constituent sequence of length L1=32 with the second truncated sequence of length L2=32~16=16 symbols will exhibit a periodicity of L=32 symbols with the same TAI as the first original constituent sequence. It is noted that cases where the lengths of the constituent sequences are identical to each other or multiples of each other are envisaged later in another sub-case of the embodiment covering the processing of combined constituent sequences transmitted by the same satellite and disclosed to provide other advantageous performances. - Instead of transmitting two sequences, the GNSS transmitter can transmit three or more overlay sequences. V represents the number of constituent overlay sequences transmitted and is in the "de Brownian" basis. By doing so, the length of the aggregated overlay sequence (L = L1 × L2 × L v ...×L V ), it is possible to further extend the TAI. - The symbol duration of two or more overlay sequences (total V) may differ as "de Brown" based sequences: s1 (≠ or =) T s2 (≠ or =) T sv ...(≠ or =) T sV In this particular case, the symbol duration of the aggregated overlay sequence is equal to the largest common divisor (lcd) of the different symbol durations of the original and constituent overlay sequences: s =lcd(T s1, Ts2, T sv ,···,T sV ). Furthermore, the following steps are applied to obtain the aggregated overlay sequence. First, R v By repeating the doubling, we obtain a so-called interpolated constructive sequence, where R v is a constructive sequence T sv represents the ratio of the symbol duration of the aggregated overlay array to the symbol duration of the s :R v =T sv / T sSecondly, each interpolated constructive sequence is concatenated with itself in a similar manner to the case shown in Fig. 13b to generate a stream of concatenated interpolated constructive sequences. Thirdly, an aggregated overlay sequence is obtained by combining the streams of concatenated interpolated constructive sequences. Assuming the case of binary constructive sequences, the symbols of the aggregated overlay sequence are M-ary symbols, where M = 2 × V. The length L of the aggregated overlay sequence represents the periodicity represented by the M-ary symbols of the aggregated overlay sequence. Finally, the snapshot duration shall include a subset containing N M-ary symbols that on the one hand satisfies the SO(L,N) property and on the other hand guarantees the maximization of the ratio L / N. A description of the procedure for obtaining an interpolated constructive sequence is disclosed below based on an example assuming two constructive sequences with different symbol durations. The first original "de Brown" sequence contains L1 = 4 symbols (for N = 2) with symbol duration T1 = 20 ms and is equal to [1,1,0,0], while the second original "de Brown" sequence contains L2 = 4-1 = 3 symbols (i.e., N = 2 and K = 1 symbols are truncated) with symbol duration T2 = 15 ms and is equal to [0,1,1]. In this example, the symbol duration of the aggregated overlay sequence is equal to 5 ms and 5 = lgcd(15,20). Furthermore, by repeating each of the four symbols of the corresponding first constituent sequence 20 / 5 = 4 times, a first interpolated constituent sequence is obtained with the sequence [1111,1111,0000,0000]. Similarly, by repeating each of the three symbols of the corresponding second constituent sequence 15 / 5 = 3 times, a second interpolated constituent sequence is obtained with the sequence [000,111,111]. Then, as shown in FIG. 13a, both of the aforementioned interpolated constructive sequences are concatenated to form a stream of concatenated interpolated constructive sequences, and an aggregated overlay sequence is obtained by combining the streams of concatenated interpolated constructive sequences.

[0099] In another subcase of the corresponding embodiment, the same satellite transmits the same length L and symbol duration T s It is disclosed to transmit two or more constructive de Brown sequences with V = 1, V = 2^N, ... s1 =T s2 =T s Furthermore, these constructive de Brown sequences are selected such that an interval with a small number of transitions (the consecutive symbols

[01] and

[10] constitute a transition, and the consecutive symbols

[00] and

[11] do not constitute a transition) of the first constructive de Brown sequence corresponds to an interval with a larger number of transitions of the second constructive de Brown sequence. By applying this selection and design rule, we can obtain a N×T sFor a given snapshot duration of 1 and containing 2N symbols (N symbols from the first de Brown sequence and N symbols from the second de Brown sequence), the average number of transitions per snapshot duration is larger, which allows to retrieve the corresponding 2N symbols by applying the previously mentioned better performing soft or hard decoding techniques. As a direct consequence, the retrieval performance of the corresponding 2N symbols is improved when compared to the case of transmission of a single de Brown sequence with the same aggregated power, i.e. the transmission power allocated to the signal components modulated with the single de Brown sequence is equal to the aggregated power allocated to both signal components modulated with both de Brown sequences. In addition, the same number of symbols is obtained for a snapshot applied to a combined signal containing two constituent overlay sequences and having half the duration of that applied to a signal containing a single overlay sequence (N). Thus, at a higher signal-to-noise ratio allowing error-free demodulation, the latency is reduced by a factor of 2. In this alternative scheme, both constituent sequences have the same length L and symbol duration T s, the location of the implicit time marker can again be defined according to the rule to be the location of the first symbol of the first constituent sequence which is identical to the location of the first symbol of the second constituent sequence. This implicit time marker is again used to find the synchronization error between the first and second time scales. Note that the case where two binary de Brown sequences are modulated can be similar to the case where a single quaternary de Brown sequence is modulated onto a single signal component. Thus, the modeling and formulation assuming an aggregated overlay sequence as described above in the case where the constituent sequences have different lengths can also be used in this case when the constituent sequences have the same length. An aggregated overlay sequence is again obtained for each combination of both constituent overlay “de Brown” based sequences. The method of the present disclosure can be extended to more than two constituent de Brown sequences (up to V constituent sequences) or when the symbol durations of both constituent de Brown sequences are different (T S1 ≠T S2 ) is inversely proportional to the length of each sequence (L1 / L2=T s2 / T s1 ), which can be extended to the case L1 × T1 = L2 × T2, which has the same sequence duration when expressed in seconds.

[0100] Further embodiments utilize a large ensemble of candidate de Brown sequences (equal to 2^(2^(N-1)-N) for binary sequences) in order to introduce new features such as the ability to detect and correct errors in the overlay symbol retrieval process (i.e., demodulation). While the objective of using de Brown sequences is to minimize the L / N ratio, in the special case of binary de Brown sequences, alternative processing approaches can be envisaged by utilizing a longer snapshot that contains more than a minimum number of N symbols as part of a subset within the entire overlay sequence of length L equal to 2^N. Specifically, instead of processing N overlay symbols, the radio receiver may use P=N+N symbols contained in the longer snapshot in order to increase the robustness of the synchronization. Ext Processes N overlay symbols. Ext Extending the snapshot duration by additional symbols firstly probabilistically increases the number of transitions supporting time synchronization, making it possible to improve the synchronization performance for retrieving the P symbol values. In addition, to improve the demodulation performance, errors in the demodulated symbols, e.g. due to a signal received at low (C / N0), can be reduced by N Ext The following example is presented to illustrate the principle of exploiting the property that results in the detection of the error: P=N+N Ext Suppose we assume an extended subset containing P overlay symbols, and N of these P symbols are err Pieces are damaged, N err Assuming that ≥ 1, then P = N + N Ext The newly obtained corrupted subset containing overlay symbols is P=N+N Ext This extended subset cannot occur in the same position of an uncorrupted subset of overlay symbols (i.e. there is no demodulation error). However, this extended subset can still occur in an uncorrupted stream of overlay symbols in other positions. Here, we need to distinguish between different cases: - The first case assumes that the corresponding corrupted subset of P overlay symbols does not occur at all in the uncorrupted stream of overlay symbols. In this case, the receiver does not rely on this snapshot for synchronization, since, per design, the uncorrupted subset occurs once according to the SO(N,L) and thus the SO(P,L) property. - The second case assumes that the corresponding corrupted subset containing P overlay symbols occurs multiple times in the uncorrupted stream of overlay symbols. Since, per the design, any subset of N or more (i.e., P) symbols occurs once, according to the SO(N,L) and thus the SO(P,L) property, the receiver also does not rely on the corresponding snapshot to provide synchronization. - The last case considers a single occurrence of the corresponding corrupted subset containing P overlay symbols. This situation leads to ambiguity, since the receiver may interpret the sequence as uncorrupted even though it is corrupted.

[0101] Two examples are disclosed below to illustrate the circumstances in which a particular overlay (i.e., de Brown) sequence may or may not support the detection of, for example, 8 errors in a subset of the N symbols taken. - Figure 15 shows the situation where the overlay sequence cannot help to detect the 8 erroneous and removed symbols. First, an overlay sequence containing L = 16 symbols (for N = 4) is assumed: [1000011001011110]. Then, a longer snapshot of length P = 8 symbols is received and processed. Furthermore, under extreme demodulation conditions, i.e. low or very low (C / N0), N errAssume that all 01010000 symbols are demodulated in error. In this situation, the demodulated subset undergoes a complete inversion (assuming that PLL-based demodulation allows for the retrieval of the symbol polarity). Figure 15 shows that the corresponding erroneous subset of the retrieved symbols [11010000] can be found at another position (concatenated with the following overlay sequence) in the entire overlay sequence. This example shows that one single occurrence of an erroneous subset can result in an ambiguity in the retrieval of time, since it results in an incorrect position relative to the implicit time marker (14th symbol) compared to the actual and true subset position relative to the implicit time marker set according to the rule at the first symbol of the overlay sequence (8th symbol). Therefore, this particular overlay sequence [1000011001011110] is difficult to detect 8 errors. - Figure 16 shows another example of a particular overlay (i.e., de Brown) sequence capable of supporting the detection of eight errors in a given extended subset, and illustrates the processing logic in a radio receiver. In this example, the overlay sequence is equal to [0001001101011110], again containing L = 16 symbols corresponding to N = 4. Furthermore, an extended subset of length P = 8 is assumed, where N Ext N = 4 is taken from the fourth symbol position [00110101] from the beginning of the overlay sequence. err Assuming that the =8 erroneous symbols are in the taken-out subset of 8 symbols [11001010], it can be shown that this erroneous subset of 8 symbols cannot be found anywhere else in the (concatenated) overlay sequence. Thus, the receiver is prepared to not accept the corresponding subset of erroneous symbols in order to resolve the time ambiguity.

[0102] The distinction between the three aforementioned cases is that the condition required for the de Brown sequence to detect demodulation errors is that at any position, at most N err,maxis corrupted by erroneous symbols, contains P symbols, and err,max allows the realization that any extended subset that varies between 1 and P will not occur in the uncorrupted stream of overlay symbols, or if it does occur, it will occur multiple times. If both conditions are met, it means that the extended subset of the corresponding de Brown sequence is at most N err,max If a signal is corrupted by erroneous symbols, meaning that they do not occur or occur multiple times in the stream, it is possible to decide to reject the synchronization obtained with the longer snapshot containing P symbols. The radio receiver therefore discards the demodulated extended subset. Then, by assuming that the receiver can simultaneously receive different signals transmitted by different satellites, as is the case in GNSS navigation systems, the receiver can try to extract a longer snapshot containing P symbols transmitted by another satellite contained in the same signal snapshot. Alternatively, the receiver can select N to N * Ext >N Ext N * Ext By including additional symbols, P * >P is P * The signal snapshot duration can be further extended to include P symbols. Alternatively, the receiver can take another longer snapshot including P overlaid symbols transmitted by the same satellite to attempt to resolve time ambiguities at the same satellite. Thus, the wireless receiver can receive up to N err,max With this detection capability, it is possible to demodulate P symbols instead of the minimum N symbols. Such processing logic, combined with the properties of a subset of de Brown sequences, allows the detection of the maximum N symbols in a sequence. err,max It provides error detection capability for time synchronization of demodulation errors.

[0103] N err,maxTo support the detection of N symbols, an iterative selection process of de Brown overlay sequences is performed. In a first step, design parameters are defined. This corresponds to the overlay sequence L and therefore the minimum (i.e., non-extended) snapshot duration containing N symbols. The number of additional symbols adjacent to the N symbols, N, is Ext also contains a longer signal snapshot P=N+N Ext It is set to define an extended subset containing up to N symbols. ext = LN is assumed to be the extreme case with an extended subset having the same length as the overlay sequence. Finally, the maximum number of detectable errors N applied to an extended subset containing P symbols err,max,test is initialized to the value L (for the extreme case where the snapshot has length P, the same as the length L of the overlay sequence, and all P symbols are erroneous). In the second step, candidate binary de Brown overlay sequences from a pool containing 2^(2^(N-1)-N candidate de Brown overlay sequences are successfully tested. For each candidate de Brown overlay sequence, an extended subset containing P symbols is selected (we also assume the periodicity property). Thus, L such extended subsets containing P symbols can be selected from the complete de Brown overlay sequence. Up to N err,max,test Errors (1, or 2, or ..., N err,max,test error) is applied within the corresponding expanded subset, and the application of the error is performed by removing the 0 (respectively 1) symbols of the first binary de Brown overlay sequence from these up to N err,max,test The method consists of substituting selected 1 (respectively 0) symbols at specific positions, resulting in an erroneous extended subset containing P symbols. err,max,testAll possible combinations of N possible positions are considered in sequence. Then, if it is shown that the corresponding erroneous extended subset can be found only once in the first de Brown overlay sequence, the candidate de Brown overlay sequence is rejected, otherwise, if the corresponding erroneous extended subset cannot be found or can be found but multiple times, another erroneous extended subset is generated to continue the process for this candidate de Brown overlay sequence. This process continues for each selected extended subset in the de Brown overlay sequence and for all L possible extended subsets containing P symbols in the de Brown overlay sequence, with up to N possible combinations of P possible positions. err,max,test After this selection process, the de Brown overlay sequence contains P symbols, and the maximum N obtained at any position within P symbols is err,max,test If it can be shown that any extended subset contaminated with errors does not occur once in the original de Brown overlay sequence, then the corresponding de Brown overlay sequence will have at most N err,max,test is chosen to support the detection of N err,max is N err,max,test otherwise, N err,max,test is decremented by 1. This process is repeated until N err,max,test Continue until a value is found, in which case N err,max is N err,max,test Equivalent to N err,max,test If L decreases to 0 (i.e., the condition is not met even if only one error is applied to the extended overlay sequence), another de Brown overlay sequence from the pool is selected as a candidate. This process continues until successful completion of the condition by at least one candidate de Brown overlay sequence among all candidate de Brown overlay sequences of length L.

[0104] [Reference 8]: Robinson, Derek JS (2003) "An Introduction to Abstract Algebra", Walter de Gruyter, pp. 255-257 In addition to the error detection capability, the selection of a particular de Brown sequence as the overlay sequence shall also enable the radio receiver to correct a particular amount of errors. An extended subset of the P symbols can be considered as codewords in a particular set. Such a set consists of all possible subsequences of P symbols starting from all the different L positions in the overlay sequence. With this assumption, it is possible to apply concepts from coding theory to the demodulated sequence. Coding theory allows the radio receiver to determine the maximum N err,max If a set of errors can be detected, then the minimum Hamming distance between any two codewords is (N err,max +1), then the maximum JPEG2024535752000028.jpg specifies that 8132 errors can be corrected [Reference 8]. The radio receiver can apply a minimum distance decoding technique to the processed sequence with errors. Specifically, the radio receiver examines all L possible subsequences of P symbols in the entire overlay sequence. It then finds from this set the subsequence of P symbols that has the minimum Hamming distance to the extended subset of the demodulated P symbols. By definition, the Hamming distance between two codewords of equal length is equal to the number of positions where the corresponding symbols differ. The number of errors is determined by the Hamming distance between the two codewords of equal length. JPEG2024535752000029.jpg8132, the wireless receiver can select the correct subsequence of P symbols, correct the erroneous symbols, and achieve synchronization, regardless of the demodulation error. If the Hamming distance appears to be 0 for one subsequence of P symbols out of L possible symbols, it means that there is no error, and the position of the extended subset corresponds to the position of the subsequence of P symbols whose Hamming distance with the extended subset of P symbols is 0. Based on the extended subset position, it is possible to resolve the time ambiguity following the same procedure as described when the snapshot duration contains N symbols.

[0105] Figures 17a and 17b show the principle for detecting and correcting erroneously taken symbols in an extended subset of P symbols. In Figure 17a, an overlay sequence [0000100111101011] is assumed. This particular de Brown sequence is suitable for detecting and correcting erroneously taken symbols for any extended subset of P=10 symbols up to N err,max It has been shown that it is possible to detect P = 3 erroneously taken symbols. In other words, a snapshot containing P = 10 symbols taken at any position in the overlay sequence and contaminated with 1, 2 or 3 demodulation errors is detected because the extended subset of the corresponding erroneous symbols does not occur at any position in the overlay sequence. Based on this overlay sequence, assume a particular snapshot measured from a second position in the overlay sequence, this snapshot is N err= 1 erroneous symbol: [0101001111]. The erroneous symbol is located at a second position in this snapshot (the "error-free" snapshot is [0001001111]). Figure 17b then shows a table providing the Hamming distance calculated between the erroneous subset and any subsequence of 10 symbols out of the 16 possible symbols in the overlay sequence. In the example of the present disclosure, the Hamming distance varies between 1 and 10. The correct position of the extended subset in the overlay sequence corresponds to the index of the subsequence in the error-free overlay (i.e., de Brown) sequence that exhibits the smallest Hamming distance calculated with the extended subset containing 10 symbols, i.e., 1. This index is equal to 2, which is effectively confirmed from Figure 17b. It is therefore possible to correct the corresponding extended subset and to extract the relative distance of the (corrected) subset with respect to the implicit time marker to resolve the time ambiguity. The maximum number of detectable errors in the de Brown sequence of the present disclosure is N err,max = 3, so the maximum JPEG2024535752000030.jpgThis is clearly the case in this example, since 10132 symbols can be corrected.

[0106] Although the error detection and correction de Brown processing penalizes the latency in deriving synchronization by requiring a longer signal snapshot duration containing P symbols instead of N symbols, this new feature increases the reliability of the synchronization. To support such detection and correction of demodulation errors, it must be shown that a de Brown sequence selected from a large pool of existing de Brown sequences with length L still satisfies the SO(N,L) property according to the definition and thus, without detection and correction of demodulation errors, it is still possible to assume a minimum signal snapshot containing "only" N symbols to support TAI resolution. The use of a longer snapshot length to support error detection and correction is an implementation choice of the radio receiver depending on its particular application. Given the expected demodulation errors, such error detection processing of the de Brown sequence can be dimensioned to provide a measure regarding the reliability of the synchronization of the radio receiver with the transmitted overlay sequence. Length P=N+N Ext The embodiment of the present disclosure based on the use of an extended subset of Ext It should be noted that this is not limited to the case where the first (subset) part containing Q symbols is adjacent to the second (subset) part containing Q symbols, but may also be the case where both (subset) parts are spaced apart or spaced apart by Q symbols. If we assume an extended subset divided into two (subset) parts, a similar method for the selection of the de Brown sequence that guarantees error detection and correction can be followed, in which case an additional optimization parameter with the "inter-part spacing" Q is assumed for this optimal selection. However, it should be noted that having spaced snapshots penalizes the latency in time ambiguity resolution after the error detection and correction step, and also forces the radio receiver to be active for a longer duration corresponding to an even more extended snapshot covering a complete period over both (subset) parts of the extended subset. Further embodiments are disclosed below for reducing the error rate in time ambiguity resolution due to erroneously estimated positions of signal snapshots relative to implicit time markers by modulating a truncated de Brown sequence derived from the original de Brown sequence onto the carrier of a wireless signal. To illustrate this principle, the upper part of Fig. 18a shows a block diagram of length L=2 7 18a represents partial autocorrelation values ​​obtained between a snapshot sequence containing a first subset formed with the first symbol of N=7 of the original de Brown sequence of N=128 symbols (a subset padded with 121 "0"s) and any second subset of N=7 symbols in the original de Brown sequence (the second subset is also zero padded). The first symbol of this second subset starts with index 1 of the original de Brown sequence and ends with index 128, resulting in 128 possible second subsets and thus partial autocorrelation values. The total of 128 possible partial autocorrelation values, which can be called noise-free partial autocorrelation, varies between -7 and 7. When the snapshot sequence is associated with the second subset containing 7 symbols starting with index 1 and 0, the value 7 is obtained. The lower part of FIG. 18a represents the distribution of the corresponding noise-free partial autocorrelation values. It can be observed that this distribution is symmetric for positive and negative noise-free partial autocorrelation values. Furthermore, the corresponding distribution of partial autocorrelation values ​​is shown to be the same for an original de Brown sequence of a given length L, as an inherent property of the de Brown generation. Although the occurrence of noise-free partial autocorrelation values ​​as a function of offset differs for each original de Brown sequence, their distribution is the same. It can be further observed that the second largest noise-free partial autocorrelation peak value, 5, is approximately 5% over the 128 partial correlation values. The second largest noise-free partial autocorrelation value is referred to as the first side peak noise-free partial autocorrelation value.

[0107] Now, assume that the soft decoding method described above is used to retrieve the position of the snapshot relative to the implicit time marker, i.e., the snapshot signal after removal of the Doppler offset and embedding in noise is first zero-padded before being correlated with the entire original overlaid Brownian sequence. If the received signal is embedded in a large noise level, i.e., the signal processing is performed at a low (C / N0), it may happen that the noisy partial autocorrelation obtained at a signal snapshot with a noise-free partial autocorrelation of 5 exceeds the noisy autocorrelation obtained at a signal snapshot with a noise-free autocorrelation of 7. In this case, the resolution of the time ambiguity will be inaccurate due to misleading information about the actual position of the snapshot relative to the implicit time marker signal. To avoid such a situation, one solution is to truncate the original overlaid sequence by removing U symbols such that the number of large noise-free partial autocorrelation values ​​(5 and -5 in the example of this disclosure) is reduced. An example is shown in the upper part of Fig. 18b, showing 120 different noise-free partial autocorrelation values ​​obtained in the case of a truncated de Brown sequence containing 120 symbols after applying a truncation of U = 8 symbols. In this example, the truncation is obtained by removing the last 8 symbols of the original de Brown sequence. It is shown in the lower part of Fig. 18b that the occurrence of a noise-free partial autocorrelation with a value of +5 (i.e. the first side peak noise-free partial autocorrelation value) is then reduced from 5% to 4% without truncation. By using such a truncated de Brown sequence as an overlay sequence, the number of misleading or ambiguity positions of the snapshot signal in the presence of noise can be reduced. It has been shown that by further truncating the de Brown sequence, it is possible to reduce the number of side peak values, which can improve the noise robustness of the time resolution performance. The disadvantage of this method is that the effective length of the overlay sequence is reduced, which affects the time ambiguity interval.In the examples of this disclosure, the last U=8 symbols of the original de Brown sequence are truncated to derive a truncated de Brown sequence used as an overlay sequence. However, it is possible to truncate U consecutive symbols anywhere in the original de Brown sequence to generate a truncated de Brown sequence. Furthermore, U can vary between 0 (i.e., no truncation) and (LN), the maximum meaningful truncation value for a snapshot containing N symbols.

[0108] In a further alternative and advantageous embodiment of the invention, different sequences are transmitted by different transmitters, such as satellites of a global navigation satellite system, as shown in FIG. 19. This alternative scheme, also based on a combined processing of different "de Brownian" sequences, is proposed to improve time ambiguity resolution performance, such as latency, and uses similar elements to the previous embodiment. In this previous embodiment, the same overlay sequence (possibly with different lengths and modulated with different signal components) was transmitted by each of the different satellites of the GNSS. It is disclosed below that envisages advantages that can be obtained by transmitting different sequences to different satellites or groups of satellites obtained from a clustering of the constellation.

[0109] Figure 19a shows two extreme satellite positions relative to a user device. The first corresponds to the case where the satellite is at the horizon (elevation angle 0°) and the second corresponds to the case where the satellite is exactly at the zenith of the user device (elevation angle 90°). The same figure also shows the radius of the Earth, R earth (=6378.137 km) and the satellite's semi-major axis (SMA) D are also shown. The horizontal distance to the satellite is d hor =(D 2 -R 2 earth For a GPS constellation with an SMA of 26559.70 km, d hor= 25782.49 km, or 85.9 ms. For the GALILEO constellation with an SMA of 29601.3 km, d hor = 28905.99 km, or 96.3 ms. Similarly, the distance to the zenith satellite is d Zen =(DR earth For a GPS constellation with an SMA of 26559.70 km, d Zen = 20181.56 km, or 67.2 ms. For the GALILEO constellation with an SMA of 29601.3 km, d Zen = 23223.16 km, or 77.4 ms. From this Fig. 19a, the difference between the overlay sequence edges transmitted by the two satellites is about 20 ms (GPS 85.9 ms - 67.2 ms, Galileo 96.3 ms - 77.4 ms).

[0110] In Fig. 19b, it is assumed that the constellation is divided into two groups of satellites: satellites transmitting sequences of length L1 = 32 symbols, symbolized by solid lines (the conventions of Figs. 13a and 13b are also used here), and satellites transmitting sequences of length L2 = 31 symbols, symbolized by dashed lines (the conventions of Figs. 13a and 13b are also used here). The allocation of the satellites is made to ensure a relatively uniform reception of the overlay sequences transmitted by the satellites of the first and second groups. Such a subdivision can be achieved, for example, by alternately arranging the satellites belonging to each group in each orbital plane of the constellation. Here, for ease of understanding, the characteristics of the sequences used in the illustrations of Figs. 19a and 19b are applied again. We then apply a common overlay symbol duration to both sequences and assume a maximum difference in propagation times greater than 20 ms. Then, assuming an additional portion of the overlay sequence in addition to the snapshots of N (N = 5 in the example) to account for the propagation differences, it is possible to measure N symbols from the first sequence and N symbols from the second sequence. The same principles can then be applied to those presented in the previous scheme to derive the extended TAI. The mathematical derivation for the design of the parameters (L, snapshot time, ...) can also be applied in this alternative embodiment, if different de Brown sequences are transmitted by different satellites, assuming that the additional condition on the symbol duration for the maximum difference in propagation applies. The main difference is that the edges of the overlay sequences are not received synchronously, since they are transmitted from different satellite positions.

[0111] The corresponding alternative scheme can be extended here as well by considering different sequence lengths, number of truncated symbols or symbol durations, similar to the variations shown in the previous scheme applied when different overlay sequences are transmitted by the same satellite. Also, the constellation can be divided into more than two subgroups, each assigned a different sequence length. Further extensions of the scheme can also be disclosed. In this new scheme, two overlay sequences are transmitted by the constellation. One is transmitted by the first half of the constellation satellites and the other is transmitted within a few tens of milliseconds (50 ms≦T S1 Short overlay symbol duration T ≦100 ms S1 , and the other is transmitted by the second half of the constellation satellites and takes several hundred milliseconds (100 ms ≤ T S2 Longer symbol duration T (≦1000 ms) S2 For these communicating devices, only short snapshots are needed to ensure a reliable estimation of the synchronization error, as already explained in the general application of the invention. For non-communicating devices, snapshots of longer duration (e.g. 1-2 seconds) are needed to extend the time ambiguity resolution and provide absolute time by processing and combining both types of overlay sequences.

[0112] A further declension of the above embodiment considers the case where different overlay sequences of the same length are transmitted by different satellites. The advantage of this scheme is that it improves the latency of time ambiguity resolution, not by improving the time ambiguity interval, but by improving the retrieval performance of the symbols contained in the signal snapshot. The same principles as shown in the above scheme where the same satellite transmits two or more overlay sequences of the same length are applicable here. Nevertheless, an additional constraint on the symbol duration based on the maximum difference in propagation time between any two satellites needs to be considered here. A further alternative scheme is disclosed below to facilitate demodulation of overlay sequences such as de Brown for receivers of a type that have access to the relative phase transitions of the signal but not to the absolute phase. In the above description, the "de Brown" sequence is modulated with the phase of the GNSS signal. This will be explained in more detail.

[0113] For a receiver to uniquely determine any subsequence N within the L symbols, it needs to know the absolute phase of the signal. If the receiver cannot resolve this 180° phase ambiguity (i.e., determine the difference between what is interpreted as a 0 or a 1), then all subsequences N will be N * For any de Brownian sequence, there exists a reversed subsequence N of length L. * also exists, but at a different position in the overlay sequence than one of the subsequences N. To unambiguously demodulate the binary state of a single symbol from a phase-modulated GNSS signal, the receiver must resolve the phase (i.e., track the signal with a phase-locked loop (PLL)). However, PLL processing imposes several implementation constraints (e.g., closed-loop processing) that are not compatible with snapshot and low-power devices, and performance penalties (e.g., additional delay due to pull-in transition between acquisition and tracking for symbol retrieval).

[0114] Alternative GNSS signal processing, tracking, and techniques such as frequency-locked loop (FLL) have been identified as more amenable to supporting the aforementioned low-power / snapshot receiver terminals. In fact, FLL is known to have a simpler implementation, provide less sensitive tracking, and not exhibit as large a delay during pull-in as PLL. The main reason is that in a typical GNSS signal processing flow, FLL processing starts immediately after the acquisition step that only resolves the residual frequency to ensure bit synchronization and PLL loop closure to resolve the remaining phase ambiguities, followed by carrier tracking and PLL tracking for unambiguous data demodulation. Furthermore, FLL can operate in harsher environments (e.g., higher noise and interference levels) than PLL. As a major drawback, FLL can only help identify relative phase changes (i.e., detect when the signal phase has changed between two binary symbols, -1 to +1 and +1 to 1).

[0115] In summary, the PLL signal tracking mode is less robust than the FLL and requires a pull-in phase before initial loop closure, which can generate typical delays on the order of tens to hundreds of milliseconds, and as a result may not be applicable for snapshot receiver processing. Thus, the de Brown sequence is utilized on the overlay symbols to support the ability to derive the GNSS system time for a PLL tracking receiver, as well as the phase transitions to support the ability to derive the GNSS system time for a snapshot or FLL based receiver operation. Here, the relative phase change is encoded by a sequence with uniqueness. In the following, the term transition sequence refers to the de Brown sequence modulated on the phase transition (i.e., transition [1] or no transition [0]). The term integration sequence refers to the overlay symbol sequence that leads to the transition sequence after the operation of binary integration. Those skilled in the art will understand that two overlay symbols of the integration sequence are required to encode one phase transition of the transition sequence. Two consecutive identical binary symbols [0,0] or [1,1] of the integration sequence result in the [0] binary state of the “transition sequence” (i.e. no transition), and two consecutive different binary symbols [0,1] or [0,1] of the “integration sequence” result in the [1] binary state of the “transition sequence” (i.e. two consecutive overlay symbol changes equal a phase transition). The redundancy between [0,0] and [1,1] (or between [0,1] and [1,0]) to code [0] (or [1]) comes from the anti-phase relationship between the overlay symbols (i.e. 180 degree phase ambiguity) as mentioned above. The “integration sequence” then serves as an overlay sequence that is phase modulated onto the GNSS signal. As a consequence, the “integration sequence” of the de Brown “transition sequence” must itself inherit the SO(N+1,L) property to allow the modulation of the sequence by SO(N,L). Furthermore, note that two "integration sequences" exist in an anti-phase relationship and both can independently generate the same de Brownian "transition sequence".

[0116] When receiving a de Brown “transition sequence”, an increased observation period of N´=N+1 symbols is needed to demodulate the N phase states (i.e., transition or no transition). However, even when using an N+1 symbol observation period to decode the N phase states, there are also cases where no transition occurs (N+1 consecutive overlay symbols with the same binary state) and, as a result, no phase change occurs (N 0s). However, in this particular case, the reference point is prevented from resolving the synchronization error (no transitions available for “bit synchronization”) and a design mitigation must be found to avoid this situation. The first and most simple mitigation is to increase the observation period to N+2 symbols, but this has a penalty on the required minimum observation period (N+2) and on the TAI gain (L / N ratio). This penalty is especially obvious and significant for shorter sequences. An alternative mitigation consists in removing a special subsequence of “N 0s (or all 0s)” from the original de Brown sequence, resulting in a “truncated transition sequence”. To preserve the SO property, we need to apply the mechanism introduced in Fig. 20 to generate a truncated transition sequence based on the de Brownian original sequence.

[0117] [Reference 9]: "Combinatorial generation", Ruskey Frank, University of Victoria, Victoria, BC, Canada. October 1, 2003 Figure 20 describes the method used to generate an "integrated sequence" that satisfies the SO(N+1,L) property based on an "original sequence", where the square boxes correspond to sequences and the diamond boxes correspond to processes. The "original sequence" (Figure 20(a)) with all 0 / 1 states at the beginning / end of the sequence can be generated using the algorithm described in [Reference 9]. This algorithm can guarantee the following properties: 1) A sequence of N zeros (all zeros) surrounded by N ones (all ones) on one side and N-2 zeros starting from junction 1 on the other side. 2) There is no subsequence with N-1 0s and N-1 1s. The algorithm in [9] naturally inherits these properties, being based on a de Brownian graph initialization with all-zeros states. However, as introduced in [6], the same sequence can in general be found as one of 2^(2^(N-1)-N) existing de Brownian sequences for each N, L=2^N. These features ensure that a truncated sequence with the SO property can be generated (FIG. 20(b)): after removing all N zeros, there is a truncated sequence of length (2^N)-N. In this truncated sequence, the junction of the N all-one subsequences is necessarily the junction of another subsequence according to feature (a). In the truncated sequence, this results in the only violation of the SO property at that stage, since N+1 ones follow each other. This case can be corrected by purging from this subsequence a single one located to the left or right of the already purged sequence of N zeros, and the SO property of the original subsequence remains valid as guaranteed by feature (b). After this first step of truncating N+1 symbols from the original sequence to derive a truncated sequence, it is possible to apply a second step of truncating K further symbols from this truncated sequence to derive a truncated transition sequence. Note that this second truncation step of K symbols is optional. A "truncated transition sequence" is obtained with length L' = LN-1-K and the property SO(N,L'). Thus, an FLL-based receiver can resolve time ambiguities within an interval of L' symbols by observing N plus 1 symbols.

[0118] According to [Reference 9], the "original sequence" with the characteristics outlined above always has 2^(N-1) ones. By applying the inventive truncation procedure and the additional number K of truncated symbols, the number of remaining transitions in the "truncated transition sequence" can always be an odd number. The corresponding individual "truncated transition sequences" are then concatenated and integrated, resulting in successive individual "integrated sequences" that are phase modulated onto the GNSS signal. The odd number of transitions causes every other overlay "integrated sequence" to be dephased (Figure 20(c)). Thus, every other individual "integrated sequence" with L' symbols is phase shifted by 180 degrees with respect to the preceding sequence, forming a unique concatenated "overlay symbol sequence" of 2 x L' symbols. This allows the use of 2 x L' overlay symbols to code a cyclic and infinitely long "truncated transition sequence" of L' symbols (Figure 20(d)). Each of the two representations of the “integration sequence” (i.e., positive and negative phasing) has the SO(N+1,L´) property, where every subsequence of N+1 symbols of the positive “integration sequence” reflects its inverted value in the corresponding subsequence of N+1 symbols of the negative “integration sequence”, and the resulting concatenated “overlay symbol sequence” (formed by L´ positive “integration sequence” symbols and L´ negative “integration sequence” symbols) is unique and satisfies the property SO(N+1,2×L´). Thus, a receiver capable of resolving phase ambiguities (i.e., using a PLL) can double the TAI for the FLL operation by identifying any N+1 symbol subsequence in the 2×L´ long “overlay symbol sequence” (Figure 20(e)).

[0119] The above procedure used to derive a truncated transition sequence based on an original de Brown sequence is based on a particular category of original de Brown sequences generated according to the algorithm presented in [Reference 9] and satisfies properties (a) and (b). More specifically, they exhibit a subset of N "0"s followed by N "1"s or N "1"s. However, it is possible to apply a similar procedure to other original de Brown sequences in which the subsets of N "0"s and N "1"s are not adjacent. In that case, the procedure consists of purging a subset of N "0"s from the original sequence and purging a single "1" on one side of this subset of N "0"s to obtain a truncated sequence of length LN-1. Furthermore, it is possible to generate a truncated transition sequence by truncating K additional symbols, optionally. It is contemplated that some of the steps described herein as software methods may also be implemented in hardware, for example as circuitry that cooperates with a processor to perform various method steps. Portions of the present invention may also be implemented as a computer program product in which computer instructions, when processed by a computer, adapt the operation of a computer such that the methods and / or techniques of the present invention are performed or otherwise provided. Instructions for performing the methods of the present invention may be stored on fixed or removable media, transmitted via a data stream in a broadcast or other signal-bearing medium, and / or stored in a working memory in a computing device that operates according to the instructions.

[0120] While various embodiments incorporating the teachings of the present invention have been shown and described in detail herein, those skilled in the art can readily devise many other various embodiments which still incorporate these teachings. Finally, it should be noted that the embodiments of the present invention have been described above in terms of functional blocks. From the functional description of these blocks given above, it will be clear to a person skilled in the art of designing electronic devices how the embodiments of these blocks can be manufactured using well-known electronic components. Therefore, a detailed architecture of the contents of the functional blocks has not been shown. While the principles of the present invention have been described above in connection with specific apparatus, it is to be clearly understood that this description is made only by way of example and not as a limitation on the scope of the invention as defined in the appended claims.

Claims

1. 1. A method for resolving time ambiguities in a radio navigation satellite system between a radio transmitter (TX1) of a plurality of transmitters of the radio navigation satellite system having a first time scale and a radio receiver (RX1) of a plurality of radio receivers of the radio navigation satellite system having a second time scale, the radio transmitter (TX1) being coupled to the radio receiver (RX1), and the radio transmitter transmitting a radio signal to the radio receiver (RX1), the method comprising: generating, by the wireless transmitter (TX1), an overlay sequence including a set of symbols for each time ambiguity interval, the set of symbols having a predetermined length L, the overlay sequence satisfying a condition of a single occurrence of a subset of symbols within the set of symbols for the time ambiguity interval, and each of the time ambiguity intervals including an implicit time marker; transmitting, by the radio transmitter (TX1) to the radio receiver (RX1), the radio signal comprising the overlay sequence modulated with a primary code modulated onto a carrier wave of the radio signal; receiving said radio signal by said radio receiver (RX1); capturing a snapshot of the radio signal by the radio receiver (RX1), the snapshot comprising a subset of symbols comprising N symbols of the overlay sequence, making the ratio L / N as large as possible; processing the snapshot by the radio receiver (RX1) to determine a relative position of the implicit time marker of the radio signal based on a position of the subset of symbols included in the snapshot within the set of symbols of the time ambiguity interval; resolving the time ambiguity between the first time scale and the second time scale by evaluating a delay between the implicit time marker based on the processing of the snapshots expressed in the first time scale and the implicit time marker in the overlay sequence generated based on the second time scale, the overlay sequence comprising an M-ary sequence based on an M-ary de Brown sequence; A method comprising:

2. A radio transmitter (TX1) configured to resolve time ambiguities in a radio navigation satellite system between the radio transmitter (TX1) of the radio navigation satellite system having a first time scale and a radio receiver (RX1) of the radio navigation satellite system having a second time scale, the radio transmitter (TX1) configured to transmit a radio signal to the radio receiver (RX1), the radio transmitter (TX1) comprising: a sequence generation means (11) configured to generate an overlay sequence comprising a set of symbols for each time ambiguity interval, the set of symbols having a predetermined length L, the overlay sequence satisfying a condition of a single occurrence of a subset of symbols within the set of symbols for the time ambiguity interval, and each of the time ambiguity intervals comprising an implicit time marker; a transmitting means (12) configured to transmit a radio signal to the radio receiver (RX1), the radio signal comprising the overlay sequence modulated with a primary code modulated onto a carrier wave of the radio signal, the overlay sequence comprising a set of symbols for each time ambiguity interval, the set of symbols having a predetermined length L, each of the time ambiguity intervals comprising an implicit time marker, and the overlay sequence comprising an M-ary sequence based on an M-ary de Brown sequence; A radio transmitter (TX1), characterized in that it comprises:

3. 3. The radio transmitter (TX1) of claim 2, wherein the sequence generating means (11) is further configured to generate a plurality of different overlay sequences.

4. A radio receiver (RX1) is configured to resolve time ambiguities in a radio navigation satellite system between a radio transmitter (TX1) of the radio navigation satellite system having a first time scale and the radio receiver of the radio navigation satellite system having a second time scale, the radio transmitter (TX1) is configured to transmit a radio signal to the radio receiver (RX1), the radio signal being modulated with a primary code modulated onto a carrier wave of the radio signal and including an overlay sequence satisfying a condition of a single occurrence of a subset of symbols having a predetermined length (L), the overlay sequence including a set of symbols for each time ambiguity interval, each of the time ambiguity intervals including an implicit time marker, the radio receiver (RX1) receiving means (21) adapted to receive said radio signal; snapshot capture means (22) configured to take a snapshot of the radio signal, the snapshot comprising a subset of the symbols of the overlay sequence; The radio receiver (RX1) further comprises: and processing means (23) configured to determine the relative positions of the implicit time markers of the radio signal based on the positions of the subset of symbols comprising N symbols of the overlay sequence included in the snapshot, to maximize the ratio L / N; the processing means (23) is further configured to resolve the time ambiguity between the first time scale and the second time scale by evaluating a delay between the implicit time markers based on processing of the snapshots expressed in the first time scale and the implicit time markers in the overlay sequence generated based on the second time scale. A radio receiver (RX1).

5. the subset of symbols included in the snapshot is extended with an additional subset of symbols of the overlay sequence, the additional subset having a length of N symbols, the extended subset of symbols comprising P=N+N symbols; the processing means (23) is further configured to calculate a Hamming distance between the extended subset of symbols included in the snapshot and each subsequence of the overlay sequence having a length of P=N+NExt symbols; the processing means (23) is further configured to detect an error in the extended symbol subset if a minimum value over all Hamming distances calculated between the extended symbol subset included in the snapshot and each subsequence of the overlay sequence comprising P=N+NExt symbols is non-zero or is zero and occurs multiple times, the processing means (23) is further configured to determine the relative position of the implicit time marker of the radio signal based on the subset of extended symbols included in the snapshot if the minimum value over all Hamming distances calculated between the subset of extended symbols included in the snapshot and each subsequence of the overlay sequence comprising P=N+NExt symbols is 0 and occurs once. Radio receiver (RX1) according to claim 4, characterized in that it is

6. The processing means (23) determines whether the minimum over all Hamming distances calculated between the extended subset of symbols included in the snapshot and each subsequence of the overlay sequence comprising P=N+NExt symbols is equal to a further predetermined minimum value depending on the selected overlay sequence. select the subsequence of the overlay sequence containing P=N+NExt symbols that results in a minimum Hamming distance, if P=N+NExt does not exceed the maximum difference between the subsequence of the overlay sequence containing P=N+NExt symbols and the extended subset of symbols included in the snapshot and further configured to correct the error by correcting the symbols. Radio receiver (RX1) according to claim 5, characterized in that it is

7. the receiving means (21) is further configured to receive a first radio signal from a first radio transmitter and at least a second radio signal from a second radio transmitter, the first radio signal and at least the second radio signal including overlay sequences, the first overlay sequence and the at least the second overlay sequence being different; the receiving means is further configured to combine the overlay sequence of the first radio signal and the overlay sequence of at least the second radio signal into an aggregated overlay sequence; the snapshot capture means (22) is configured to obtain a snapshot of the aggregated overlay sequence of the first radio signal and at least the second radio signal, the snapshot including a subset of symbols of the aggregated overlay sequence. A radio receiver (RX1), characterized in that: the radio receiver (RX1) further comprises processing means (23) configured to determine relative positions of the implicit time markers of the radio signal based on positions of the subset of symbols of the aggregated overlay sequence included in the snapshot comprising N symbols, the processing means (23) is further configured to resolve the time ambiguity between the first time scale and the second time scale by evaluating the delay between the implicit time markers based on the processing of the snapshots expressed in the first time scale and the implicit time markers in the aggregated overlay sequence generated based on the second time scale. Radio receiver (RX1) according to any one of claims 4, 5 or 6, characterized in that it is

8. The sequence generation means (11) generating a truncated transition sequence based on an original sequence consisting of an original de Brown sequence having a length of L symbols by first removing N symbols containing "0" from said original sequence, followed by removing a single symbol containing "1" from said original sequence to result in a truncated sequence, and optionally removing an additional K symbols from said truncated sequence to result in a truncated transition sequence of length L-N-1-K; generating a first integrated sequence indicating a phase transition of the truncated transition sequence and a second integrated sequence indicating a phase of the inverted truncated transition sequence, the first integrated sequence being in antiphase with the second integrated sequence; generating a concatenated integrated sequence by concatenating the first integrated sequence and the second integrated sequence; further configured as follows: the concatenated integrated sequence is configured to be modulated with a primary code modulated onto a carrier wave of the radio signal. Radio transmitter (TX) according to claim 2 or 3, characterized in that it comprises:

9. the snapshot capture means (22) is configured to capture a snapshot of a radio signal, the snapshot comprising a subset of symbols of the overlay sequence consisting of a concatenated aggregate sequence generated by a radio transmitter (Tx) according to claim 8, the snapshot comprising N+1 symbols; The processing means (23) determining N transitions from the subset of symbols of the overlay sequence included in the snapshot; determining a position of the subset of symbols included in the snapshot relative to the implicit time marker of the wireless signal based on the N transitions from the subset of symbols included in the snapshot; and further configured as follows: Radio receiver (RX1) according to any one of claims 4, 5 or 6, characterized in that it is

10. 7. The radio receiver (RX1) of claim 4, 5 or 6, wherein the processing means (23) is further configured to determine the relative position of the implicit time marker of the radio signal based on a position of the subset of symbols included in the snapshot within the set of symbols of the time ambiguity interval by searching for the subset of symbols included in the snapshot in entries of a repository (25), wherein the repository (25) contains, for each entry, a plurality of symbols of the snapshot and relative positions of the plurality of symbols of the snapshot with respect to the time marker within the time ambiguity interval of the radio signal.

11. the radio receiver (RX1) further comprises snapshot sequence generation means (24) configured to generate, based on the extracted symbols, a snapshot sequence corresponding to the snapshot of the overlay sequence modulated onto the radio signal transmitted by the transmitter (TX1); the processing means (23) is configured to determine the relative positions of the time markers within the radio signal by partially autocorrelating the snapshot sequence with an overlay sequence corresponding to the snapshot sequence. Radio receiver (RX1) according to claim 4, 5 or 6, characterized in that it is

12. the radio receiver (RX1) further comprises snapshot sequence generation means (24) configured to generate, based on samples derived from the snapshot signal, a snapshot sequence corresponding to the snapshot of the overlay sequence modulated onto the radio signal transmitted by the transmitter (TX1); the processing means (23) is configured to determine the relative positions of the time markers within the radio signal by partially autocorrelating the snapshot sequence with an overlay sequence corresponding to the snapshot sequence. Radio receiver (RX1) according to claim 4, 5 or 6, characterized in that it is

13. Radio receiver (RX1) according to claim 4, 5 or 6, characterized in that the radio receiver (RX1) implements a phase locked loop to extract the phase of the radio signal.

14. Radio receiver (RX1) according to claim 4, 5 or 6, characterized in that the radio receiver (RX1) implements a frequency locked loop to extract phase variations of the radio signal.

15. 1. A radio navigation system for resolving time ambiguities in a radio navigation satellite system between a radio transmitter (TX1) of a plurality of radio transmitters of the radio navigation satellite system having a first time scale and a radio receiver (RX1) of a plurality of radio receivers of the radio navigation satellite system having a second time scale, wherein the transmitter is coupled to the at least one radio receiver (RX1) of the plurality of radio receivers, and the radio transmitter (TX1) is configured to transmit a radio signal to the radio receiver (RX1), The system comprises a radio transmitter (TX1) according to claim 2, The system further comprises a radio receiver (RX1) according to claim 4, 5 or 6. A radio navigation system comprising:

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