Image forming apparatus

Abstract

To generate a high voltage of the sine wave AC with good power efficiency with a more inexpensive and smaller-sized configuration.SOLUTION: The inductance L2 of a second inductor is larger than the inductance L1 of a first inductor. When a resonance frequency formed by the first inductor and a capacitor is defined as a first frequency f0, a resonance frequency formed by the capacitor, the first inductor, and the second inductor as a second frequency f1, and the drive frequency of a driving signal as a drive frequency f, a driving unit 30 performs a switching operation within a range of f0≤f≤f1. A pressure raising unit 10a is driven by the driving unit 30, and generates a high voltage of the sine wave AC which is a voltage higher than DC voltage.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] The present invention relates to an image forming apparatus. [Background technology]

[0002] Image forming apparatuses detect the amount of remaining toner to notify the user of the remaining toner amount. One method for detecting the amount of remaining toner is a capacitance detection method that uses a high AC voltage. For example, in Patent Document 1, a sine wave high voltage is generated by utilizing the bandpass filter function of a piezoelectric transformer, which is a resonator. Compared to a conventional circuit that generates high voltage by driving a high-voltage wound transformer with a sine wave, Patent Document 1 discloses a method that reduces the number of components, thereby reducing the cost, size, and energy efficiency of the circuit. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Japanese Patent Application Laid-Open No. 2009-251099 Summary of the Invention [Problem to be solved by the invention]

[0004] However, recently, there has been a problem that the cost of the piezoelectric transformers used has become relatively high compared to that of high-voltage wire-wound transformers. Meanwhile, conventional high-voltage generators generate a substantially sinusoidal AC voltage for detecting the remaining toner amount using a band-pass filter, a push-pull amplifier circuit, and a high-voltage wire-wound transformer. However, even such high-voltage generators still have the problems of high cost and poor power efficiency.

[0005] The present invention has been made under these circumstances, and has as its object to generate a high voltage sine wave AC with a smaller, less expensive configuration and with high power efficiency. [Means for solving the problem]

[0006] In order to solve the above-mentioned problems, the present invention has the following configuration.

[0007] (1) A high-voltage generator that converts a DC voltage into an AC voltage, comprising: a booster unit having a first inductor having one end connected to the DC voltage or ground potential, a second inductor having one end connected in series to the other end of the first inductor, and a capacitor connected in parallel to both ends of the first inductor; and a driver unit that is connected in series to the other end of the second inductor and performs a switching operation in response to a drive signal, wherein an inductance value L2 of the second inductor is greater than an inductance value L1 of the first inductor, and where a first frequency f0 is a frequency at which the first inductor and the capacitor resonate, a second frequency f1 is a frequency at which the capacitor and the first inductor and the second inductor resonate, and a drive frequency f is a drive frequency f, the driver unit performs the switching operation within a range of f0≦f≦f1, and the booster unit is driven by the driver unit to generate a sinusoidal AC high voltage that is higher than the DC voltage.

[0008] (2) A capacitance detection device comprising a first electrode member and a second electrode member that pairs with the first electrode member, and that detects the capacitance between the first electrode member and the second electrode member, characterized in that the capacitance detection device comprises the high-voltage generation device described in (1), and the high-voltage generation device applies the sinusoidal AC high voltage to the first electrode member.

[0009] (3) An image forming apparatus for forming an image on a recording material, comprising: an image carrier for carrying an electrostatic latent image; a developing means for developing the electrostatic latent image with toner to form a toner image; a storage chamber for storing toner supplied to the developing means; and a capacitance detection device as described in (2) attached to the storage chamber, wherein the capacitance detection device detects the capacitance of toner that has entered between the first electrode member and the second electrode member, and the image forming apparatus is characterized in that the image forming apparatus comprises an estimation means for estimating the amount of toner stored in the storage chamber based on the capacitance detected by the capacitance detection device. [Effects of the Invention]

[0010] According to the present invention, a sinusoidal AC high voltage can be generated with a smaller, less expensive configuration and with high power efficiency. [Brief explanation of the drawings]

[0011] [Figure 1] 1 is a diagram showing an image forming apparatus according to first to third embodiments; [Figure 2] 1 is a diagram showing details of the process cartridges of Examples 1 to 3; [Figure 3] FIG. 10 is a diagram showing the detection waveform of remaining toner amount detection in Examples 1 to 3. [Figure 4] Block diagram showing a high voltage generator according to a first embodiment [Figure 5] FIG. 1 is a diagram showing the circuit details of a booster unit according to a first embodiment; [Figure 6] FIG. 10 is a diagram showing frequency characteristics of the impedance of the booster unit in Examples 1 to 3. [Figure 7] 1 is a diagram showing a drive voltage waveform and a current waveform of a booster unit in Example 1. [Figure 8] FIG. 10 is a diagram showing the relationship between the inductance L2 and the step-up ratio, the current consumption, and the series resonance frequency in Examples 1 to 3. [Figure 9] Block diagram showing a high voltage generator according to a second embodiment [Figure 10] FIG. 10 is a diagram showing the relationship between the drive frequency, the current consumption, and the applied output voltage in the second embodiment. [Figure 11] FIG. 10 is a diagram showing the circuit details of the booster unit of the second embodiment. [Figure 12] FIG. 10 is a diagram showing the circuit details of the booster unit of the third embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0012] Hereinafter, embodiments will be described in detail with reference to the accompanying drawings. Note that the following embodiments do not limit the invention according to the claims. Although multiple features are described in the embodiments, not all of these multiple features are necessarily essential to the invention, and multiple features may be combined arbitrarily. Furthermore, in the accompanying drawings, the same reference numerals are used to designate the same or similar components, and redundant explanations will be omitted.

[0013] In the present invention, by combining the bandpass filter function of series resonance current and the high impedance of the power supply line due to parallel resonance, a configuration is devised in which versatile, inexpensive components and a simple circuit can be used. This reduces the number of components and uses only versatile circuit components, making it possible to reduce the cost, size, and loss of the circuit. An embodiment of the present invention will be described below. [Example]

[0014] (Image forming device) 1 is a configuration diagram of an image forming apparatus 700 according to a first embodiment. The letters Y, M, C, and K at the end of the reference numerals indicate that the colors of the toner images formed by the components indicated by the reference numerals are yellow, magenta, cyan, and black, respectively. In the following description, when it is not necessary to distinguish between colors, the reference numerals will be used without the letters Y, M, C, and K at the end.

[0015] The image forming apparatus 700 has a removable process cartridge 705. During image formation, the photosensitive drum 701 of the process cartridge 705 is driven to rotate clockwise in FIG. 1. The charging unit 702 charges the surface of the photosensitive drum 701, which is a rotating image carrier, to a uniform potential. The exposure unit 707 exposes the surface of each photosensitive drum 701 to light to form an electrostatic latent image on each photosensitive drum 701. The developing unit 750 develops the electrostatic latent image on the photosensitive drum 701 with toner, thereby forming a toner image on the photosensitive drum 701. The primary transfer roller 706 transfers the toner image on the photosensitive drum 701 to the intermediate transfer belt 720. During image formation, the intermediate transfer belt 720 is driven to rotate counterclockwise in FIG. 1. The toner images on each photosensitive drum 701 are transferred to the intermediate transfer belt 720 in a superimposed state, forming a full-color toner image on the intermediate transfer belt 720 (primary transfer). The toner image on the intermediate transfer belt 720 is transported to a position facing the secondary transfer roller 711 by the rotation of the intermediate transfer belt 720 .

[0016] Meanwhile, the recording material in cassette 713 is conveyed by conveying rollers 714, 715, and 716 along conveying path 709 to a position facing secondary transfer roller 711. Secondary transfer roller 711 transfers the color toner image on intermediate transfer belt 720 onto the recording material (secondary transfer). The recording material onto which the unfixed toner image has been transferred is conveyed to fixing unit 717. Fixing unit 717 applies heat and pressure to the recording material to fix the toner image to the recording material. The recording material is then discharged outside the apparatus by discharge rollers 721. Control unit 500 includes a microcomputer (hereinafter referred to as "micon") 501, and controls the entire image forming apparatus 700.

[0017] (Process cartridge) Next, the process cartridge 705 will be described in more detail with reference to FIG. 2. FIG. 2(a) is a cross-sectional view of the process cartridge 705, and FIG. 2(b) is a perspective view. The process cartridge 705 is formed of a photosensitive drum 701, a cleaning unit 730, and a developing unit 750. A cleaning blade 731 provided in the cleaning unit 730 removes toner (residual toner) remaining on the surface of the photosensitive drum 701 after primary transfer. The toner removed from the surface of the photosensitive drum 701 by the cleaning blade 731 is stored in a waste toner storage chamber arranged in the cleaning unit 730.

[0018] The developing unit 750 has a developing chamber 751, a toner storage chamber 756 (storage chamber), and a communication port 754 that communicates between the developing chamber 751 and the toner storage chamber 756. The developing chamber 751 is provided with a developing roller 752, a supply roller 753, and a developing blade 755 as developing means for supplying toner to the photosensitive drum 701. The developing roller 752 carries toner and conveys the toner to the photosensitive drum 701 by coming into contact with the photosensitive drum 701 while rotating during image formation. The supply roller 753 rotates while making contact with the developing roller 752. Furthermore, a developing blade 755 as a layer thickness regulating member that regulates the thickness of the toner layer formed on the developing roller 752 is disposed so as to abut against the surface of the developing roller 752.

[0019] Toner storage chamber 756 is provided with an agitator 757 for agitating toner 760 stored therein and transporting the toner to supply roller 753 through communication port 754. Agitator 757 has a rotation shaft 757a parallel to the rotation axis direction of developing roller 752 and flexible agitator sheets 757b and 757c. One end of agitator sheets 757b and 757c is attached to rotation shaft 757a and the other end is free. As rotation shaft 757a rotates, agitator sheets 757b and 757c rotate, thereby agitating toner 760. Note that developing chamber 751 is located above toner storage chamber 756, and toner 760 in toner storage chamber 756 pumped up by agitator 757 is supplied to developing chamber 751 through communication port 754.

[0020] A recess 770 is formed on the inner wall surface of the toner storage chamber 756. The recess 770 is located at a position where the toner 760 will not collect but will fall under its own weight when the toner is not being stirred. Furthermore, a first conductive member 771 serving as a first electrode member and a second conductive member 772 serving as a second electrode member that is paired with the first conductive member 771 are provided on the inner wall surface of the recess 770, and are arranged substantially parallel to the rotational axis direction of the developing roller 752. Furthermore, the shape of the recess 770 when viewed from the rotational axis direction (thrust direction) of the developing roller 752 is triangular. The first conductive member 771 and the second conductive member 772 are made of a metal sheet such as SUS or a sheet member such as a conductive resin.

[0021] (Method of detecting remaining toner) Next, a method for detecting the remaining toner amount will be described. In the first embodiment, when toner passes between the two conductive members provided in the recess 770, the detected voltage changes based on the electrostatic capacitance of the toner. Then, from the waveform of the detected voltage, the duration during which the toner enters the recess 770 is measured, and the total amount of toner in the toner storage chamber 756 is estimated.

[0022] The process cartridge 705 is provided with a first contact 781 electrically connected to the first conductive member 771 and a second contact 782 electrically connected to the second conductive member 772. An AC voltage is applied to the first contact 781 by a high-voltage application unit 51. An AC current based on the electrostatic capacitance between the first conductive member 771 and the second conductive member 772 is converted into a voltage and detected by a capacitance detection unit 101, which serves as a capacitance detection device, via the second contact 782. The high-voltage application unit 51 corresponds to a high-voltage generation device that converts a DC voltage into an AC voltage.

[0023] 3 is a diagram showing a voltage waveform detected by the capacitance detection unit 101, with the horizontal axis representing time and the vertical axis representing the detected voltage (detected voltage). Note that the high voltage applied to the first contact point 781 by the high voltage application unit 51 has a voltage amplitude of 100 to 200 V and a frequency of several tens of kHz, for example.

[0024] Because the dielectric constant of toner is higher than that of air, when toner 760 conveyed by agitator 757 enters recess 770 (when toner is present), the capacitance between first conductive member 771 and second conductive member 772 increases. On the other hand, when agitator 757 passes through recess 770 and the toner in recess 770 falls due to its own weight (when toner is absent), the toner is discharged from recess 770, and the capacitance between first conductive member 771 and second conductive member 772 decreases. In other words, the time duration t during which toner passes between first conductive member 771 and second conductive member 772 due to agitator 757 periodically changes depending on the amount of toner remaining in recess 770 (referred to as the "toner remaining amount"). This periodic change is used to estimate the amount of toner remaining in toner storage chamber 756. Microcomputer 501, which serves as an estimation unit, estimates the amount of toner contained in toner storage chamber 756 based on the capacitance detected by capacitance detection unit 101.

[0025] As shown in FIG. 3, the detected voltage V1 based on the capacitance when there is toner in the recess 770 is significantly different from the detected voltage V2 based on the capacitance when there is no toner in the recess 770. In this case, a threshold value Vc is set, and whether or not toner has entered the recess 770 is detected based on the threshold value Vc. The time when toner has entered the recess 770 and the detected voltage has reached the threshold value Vc is defined as tc. The time when the toner in the recess 770 has been discharged and the detected voltage has reached the threshold value Vc is defined as td. The time from time td (or time tc) to the next time td (or time tc) is the period of one rotation of the agitator 757.

[0026] The time period t (t = tc - td) from time tc to time td during which the output voltage is below the threshold value Vc is measured as the time during which toner has entered the recess 770. This time period t varies depending on the amount of toner remaining in the toner storage chamber 756. Therefore, the microcomputer 501 can estimate the amount of remaining toner by measuring the time period t shown in FIG. 3 from the detected voltage based on the capacitance detected by the capacitance detection unit 101 and the threshold value Vc.

[0027] (Overall circuit description) 4 is a block diagram of the remaining toner amount detection mechanism in the first embodiment, which is composed of a microcomputer 501, a high voltage application unit 51a, a capacitance detection unit 101, and the above-mentioned first contact unit 781 and second contact unit 782. The high voltage application unit 51a further comprises a constant voltage control circuit 201 and a voltage controlled oscillator 2 (hereinafter referred to as VCO2).

[0028] When the microcomputer 501 outputs an ON signal, the ON signal is input to the gate terminal of a field-effect transistor (hereinafter referred to as FET) 211 of the constant voltage control circuit 201. The source terminal of the FET 211 is connected to ground (hereinafter referred to as GND) potential, and the drain terminal is connected to a DC power supply voltage Vcc via a resistor R211. In response to the input of the ON signal, the constant voltage control circuit 201 outputs a predetermined voltage to the VCO 2, which then outputs a clock signal of a predetermined frequency to the resonant high voltage generation circuit 3. The resonant high voltage generation circuit 3 generates a sine wave AC high voltage corresponding to the input clock signal and applies it to the first contact 781. The constant voltage control circuit 201 then performs feedback control to maintain the sine wave AC high voltage at a predetermined voltage value.

[0029] (Explanation of constant voltage control circuit) The constant voltage control circuit 201 shown in FIG. 4 includes an operational amplifier IC201. The operational amplifier IC201 receives a target voltage, set by dividing resistors R211 and R212, at its non-inverting input terminal to obtain the required voltage for toner level detection. The inverting input terminal of the operational amplifier IC201 receives a detection voltage, which is the result of dividing and half-wave rectifying the generated sinusoidal AC high voltage using diode D201, resistors R201 and R202, and capacitor C202. The operational amplifier IC201 then sends a differential signal based on the target voltage and the detection voltage to the VCO2. The VCO2 then sends a clock signal with a frequency corresponding to the input voltage level to the resonant high-voltage generation circuit 3. In this way, the sinusoidal AC high voltage is constantly controlled using feedback control by the VCO2 and the operational amplifier IC201. The capacitor C201 is used for phase compensation of the operational amplifier IC201.

[0030] (Explanation of the capacitance detection unit) In the capacitance detection unit 101 shown in FIG. 4, when a high AC voltage is applied to the first contact 781, an AC current flows through the capacitance formed between the first contact 781 and the second contact 782. The magnitude of the AC current changes in proportion to the capacitance. Diodes D101 and D102 are used to rectify the AC current, and a unidirectional half-wave AC current flows through each diode. Of these, the current flowing through diode D102 is input to an integrating circuit formed by operational amplifier IC101, resistor R101, and capacitor element C101. With the potential of the non-inverting input terminal as a reference, a detection value corresponding to the flowing current is converted into a voltage, output, and detected by the AD port of the microcomputer 501.

[0031] The output voltage detected at the AD port varies due to the following factors: For example, the output voltage varies due to variations in the capacitance between the first conductive member 771 connected to the first contact 781 and the second conductive member 772 connected to the second contact 782, and variations in the resistance of the conductive members. The output voltage also varies due to changes in the dielectric constant of the toner caused by environmental changes such as temperature and humidity in the image forming apparatus 700. Therefore, the potential of the non-inverting input terminal of the operational amplifier IC101, which serves as the reference potential for the detected voltage (input to the inverting input terminal), can be variably set by the microcomputer 501. Specifically, the microcomputer 501 outputs a PWM signal, and the FET 111 performs on / off operations in response to the PWM signal. The PWM signal is input to the gate terminal of the FET 111. The source terminal of the FET 111 is connected to the GND potential, and the drain terminal is connected to the DC power supply voltage Vcc via resistor R111. A pulse signal that is the inverted logic of the PWM signal output from microcomputer 501 is generated at the drain terminal of FET111. This pulse signal is converted to a DC voltage by a low-pass filter consisting of resistor R112 and capacitor C112 and input to the non-inverting input terminal of operational amplifier IC101.

[0032] (Resonant high voltage generating circuit) Next, the resonant high-voltage generator circuit 3 will be described in detail. A single-ended drive resonant high-voltage generator circuit 3a according to the first embodiment is shown in FIG. 5. The resonant high-voltage generator circuit 3a comprises a voltage conversion switching unit 6, a driver 30, and a booster 10a. The voltage conversion switching unit 6 converts a clock signal 5, e.g., with a peak value of 3.3 V, sent from the VCO 2 described above, into a clock signal with a peak value of 24 V, e.g., which corresponds to the power supply voltage Vcc. The clock signal 5 is input to the gate terminal of an FET 6 in the voltage conversion switching unit 6. The source terminal of the FET 6 is connected to the GND potential, and the drain terminal is connected to the power supply voltage Vcc via a resistor R6.

[0033] The clock signal with a peak value of 24V sent from the voltage conversion switching unit 6 is divided by resistors R31 and R32 to a predetermined peak value and input to the gate terminal of FET 31 in the driving unit 30. The source terminal of FET 31 is connected to the GND potential, and the drain terminal is connected to a snubber circuit 31, which will be described later. FET 31 performs a switching operation in accordance with the period (in other words, the frequency (driving frequency)) of the clock signal as a driving signal input to the gate terminal, and switches and drives the boosting unit 10a.

[0034] The booster 10a is composed of an inductance element L2 as a second inductor, an inductance element L1 as a first inductor, and a capacitor element C1. The booster 10a outputs a voltage according to the impedance and frequency characteristics of each element from the application output unit 7. The inductance elements L1 and L2 are, for example, coils.

[0035] A snubber circuit 31 consisting of a diode D31, a resistor R31, and a capacitor C31 is formed at the drain terminal of the FET 31, and suppresses the flyback voltage generated at the drain terminal when the FET 31 is turned off to a predetermined voltage or less.

[0036] The inductance element L1 and the capacitor element C1 of the booster unit 10a are connected in parallel, one end of which is connected to a power supply voltage Vcc (DC voltage), and the other end of which is connected in series to one end of the inductance element L2. The other end of the series-connected inductance element L2 is connected to the drain terminal of the FET 31. That is, the driver 30 is connected in series to the other end of the inductance element L2. A feature of the booster unit 10a of the first embodiment is that the inductance element L2 is configured to have both a parallel resonant frequency and a series resonant frequency, with the relationship between the two frequencies being optimized.

[0037] (Boost section impedance characteristics) The impedance frequency characteristics of this booster unit 10a were measured using a drum-type wire-wound inductor and a film capacitor, and the results are shown in Figure 6. Figure 6 is a graph with frequency [Hz] on the horizontal axis and impedance [Ω] on the vertical axis. The constants of each element are as shown in the figure. That is, L1 (described later) = 100 μH, C1 (described later) = 0.1 μF, and L2 (described later) is increased from (a) to (f) as follows: 68 μH, 100 μH, 220 μH, 470 μH, 680 μH, and 1000 μH.

[0038] There are two resonant frequencies: a parallel resonance caused by inductance element L1 and capacitor element C1, and a series resonance caused by inductance element L2, inductance element L1, and capacitor element C1. Hereinafter, the resonant frequency of parallel resonance will be referred to as the parallel resonant frequency, and the resonant frequency of series resonance will be referred to as the series resonant frequency.

[0039] The parallel resonant frequency f0 (first frequency f0) and the series resonant frequency f1 (second frequency f1) are approximately as follows: Note that in reality, they may differ due to the presence of parasitic capacitance and parasitic inductance. f0=1 / {2π√(L1×C1)} f1=1 / (1 / (2×π))×√{(L1+L2) / (L1×L2×C1)} L1: Inductance value of inductance element L1 L2: Inductance value of inductance element L2 C1: Capacitance value of capacitor element C1 Here, in the graph of FIG. 6, L1 and C1 are fixed, and therefore the parallel resonance frequency f0 does not change in the graph of FIG.

[0040] In Figure 6(a), the impedance at the series resonance frequency f1 was about 2.5 Ω. The resonance impedance and resonance frequency measured when only the inductance element L2 and the capacitor element C1 were connected in series are shown below. 1.2Ω Series resonance frequency 49.6kHz (Condition L2: 100μH, C1: 0.1μF) 1.5Ω Series resonance frequency 33.5kHz (Condition L2: 220μH, C1: 0.1μF) 2.5Ω Series resonance frequency 22.8kHz (Condition L2: 470μH, C1: 0.1μF) 2.9Ω Series resonance frequency 19.2kHz (Condition L2: 680μH, C1: 0.1μF) 3.8Ω Series resonance frequency 15.8kHz (Condition L2: 1000μH, C1: 0.1μF) In other words, the impedance during series resonance in Figure 6(a) is very low and is not significantly affected by the parallel resonance circuit, so it can be said that series resonance exists as an almost independent phenomenon.

[0041] On the other hand, in Figure 6(f), although there is a clear series resonance frequency f1 where the frequency characteristic becomes convex downward in the figure, the impedance at series resonance frequency f1 is high at approximately 130 Ω. The impedance at series resonance frequency f1 in Figure 6(f) appears to be affected by the high impedance created by the parallel resonance circuit. Here, the resonant impedance and resonant frequency measured when inductance element L1 is fixed at 100 μH and capacitor element C1 at 0.1 μF, and inductance element L2 is changed, are shown below. 2.5Ω Series resonance frequency 78.7kHz (Condition L2: 68μH) Values ​​in Figure 6(a) 3.8Ω Series resonance frequency 70.3kHz (Condition L2: 100μH) Values ​​in Figure 6(b) 6.4Ω Series resonance frequency 64.5kHz (condition L2: 150μH) 12Ω Series resonance frequency 60.5kHz (Condition L2: 220μH) Values ​​in Figure 6(c) 22Ω Series resonance frequency 57.8kHz (Condition L2: 330μH) 41Ω Series resonance frequency 55.5kHz (Condition L2: 470μH) Values ​​in Figure 6(d) 80Ω Series resonance frequency 54.7kHz (Condition L2: 680μH) Values ​​in Figure 6(e) 130Ω Series resonance frequency 52.8kHz (Condition L2: 1000μH) Values ​​in Figure 6(f)

[0042] In other words, depending on the relative value of inductance element L2 relative to inductance element L1, it is possible to increase the impedance even with series resonance. Since the value of L2 is 150 μH, which is larger than L1, it is possible to increase the series resonance impedance above that of the inductance element L2 and capacitor element C1 alone. If the only goal was to increase the impedance, adding a resistor would result in power loss and a decrease in the step-up ratio.

[0043] (General series resonant circuits and general parallel resonant circuits) Here, we will explain the characteristics of a typical series resonant circuit in which an inductance element and a capacitor element are connected in series. In a series resonant circuit, a large current flows at the series resonant frequency, making it possible for the circuit to function as a bandpass filter using the series resonant frequency, and it is also possible for the reactance component to generate an AC voltage greater than the power supply voltage. However, because the current flows extremely quickly, power loss occurs due to the parasitic resistance components of the inductance element and capacitor element, and heat generation is likely to become an issue. In addition, because an extremely large current flows at the series resonant frequency, there is also the issue that frequency control is relatively difficult.

[0044] Furthermore, in the case of a typical parallel resonant circuit in which an inductance element and a capacitor element are connected in parallel, the current flowing through the inductance element and the capacitor element is out of phase by 180°, resulting in almost no current flowing from the power supply voltage and high impedance. Looking at it from another perspective, it can also be seen as current flowing from the inductance element to the capacitor element like a pendulum. Furthermore, when switching is performed on the connection point between the parallel-connected inductance element and the capacitor element, a large flyback voltage is generated when switching is off, and the potential is forcibly reduced to zero when switching is on. This causes the voltage waveform to become distorted and far from a sine wave.

[0045] (Boosting section of Example 1) On the other hand, as described above, in the booster unit 10a of Example 1, depending on the relative relationship of the values ​​of the inductance element L2 to the inductance element L1, it is possible to increase the impedance even with series resonance. Also, it was found that in the booster unit 10a, it is possible to achieve low loss by appropriately increasing the impedance at the series resonance frequency f1 while maintaining the bandpass filter effect due to series resonance. Among the characteristics shown in Figure 6, Figures 6(c), 6(d), and 6(e) are appropriate.

[0046] The characteristics shown in FIG. 6(d) will be used as an example for a more detailed explanation. As shown in FIG. 6(d), the inductance element L2 is 470 μH, the inductance element L1 is 100 μH, and the capacitor element C1 is 0.1 μF. In the booster 10a, the gate waveforms obtained when the FET 31 shown in FIG. 5 is single-ended driven at the series resonant frequency f1 are shown in FIG. 7(b), and the respective current waveforms are shown in FIG. 7(a). Here, the current flowing through the inductance element L2 is represented by iL2 (dotted line in FIG. 7(a)), the current flowing through the inductance element L1 by iL1 (dashed line in FIG. 7(a)), and the current flowing through the capacitor element C1 by iC1 (solid line in FIG. 7(a)). The directions of the respective currents iL2, iL1, and iC1 are as shown in FIG. 5.

[0047] The current iL2 flowing through inductance element L2, i.e., the consumption current flowing from the power supply voltage Vcc to the booster unit 10a, is small. On the other hand, the current iL1 flowing through inductance element L1 and the current iC1 flowing through capacitor element C1 are greater than the consumption current. This is because the currents flow in opposite phases to each other due to the parallel resonance between inductance element L1 and capacitor element C1. The current flowing through inductance element L2 is opposite in phase to that of inductance element L1. In terms of series resonance, this can also be seen as a series resonance current flowing from capacitor element C1 to the inductance circuit in which inductance elements L1 and L2 are connected in parallel. Furthermore, currents of the same polarity flow through capacitor element C1 and inductance element L2. On the other hand, although a current of opposite polarity flows through inductance element L1, if we reverse the direction of the arrow for iL1 in Figure 5, it can also be seen as a current of the same polarity as inductance element L2 flowing toward the power supply voltage Vcc.

[0048] As a result, a nearly sinusoidal current several times greater than the current consumption from the power supply voltage Vcc can be passed through the inductance element L1 and the capacitor element C1. It was found that a voltage several times greater than the power supply voltage Vcc can be generated at the application output unit 7 due to the reactance of the inductance element L1 and the capacitor element C1. That is, the resonant high-voltage generating circuit 3 of the first embodiment is configured, and then the values ​​of the inductance elements L1 and L2 are set to an appropriate relationship. That is, the inductance value L2 of the inductance element L2 is set to be greater than the inductance value L1 of the inductance element L1 (L2 > L1). The drive frequency f of the drive unit 30 is set within the range of the parallel resonant frequency f0 to the series resonant frequency f1 (f0 ≦ f ≦ f1), and switching operation is performed to drive the booster unit 10a. This solves both the problem of reducing loss, which is difficult with series resonant frequency driving of a series resonant circuit, and the problem of generating a sine wave, which is difficult with parallel resonant frequency driving of a parallel resonant circuit.

[0049] However, as shown in Figure 7, the resonant current flowing through inductance element L2 is slightly distorted relative to the current flowing through parallel resonant inductance element L1 and capacitor element C1, and the current phase at the zero crossing point is slightly different. On the other hand, while the resonant current flowing through inductance element L2 is flowing as the drain current of FET 31, keeping FET 31 on increases the voltage at output section 7 and is also effective in improving operational stability and noise suppression. Therefore, since it is desirable to keep FET 31 on while the resonant current flowing through inductance L2 is flowing as the drain current of FET 31, it is desirable to set the on-duty width of FET 31 in Figure 5 higher than 50%. Figure 7(b) shows the gate drive voltage, or in other words, the on-duty Ton and off-duty Toff of the drive signal (Ton > Toff).

[0050] (Optimum values ​​for inductances L1 and L2) Next, we will explain appropriate constants for inductance elements L1 and L2. Figures 8(a) to 8(c) show the results of measuring the characteristics of the booster unit 10a, which change depending on the value of inductance element L2, when inductance element L1 is 100 μH and capacitor element C1 is 0.1 μF. In Figure 8(a), the vertical axis shows the boost ratio. In Figure 8(a), the generated sinusoidal AC high voltage is expressed as a magnification factor relative to the power supply voltage Vcc. Figure 8(b) shows the current consumption characteristics, and Figure 8(c) shows the series resonance frequency f1. The horizontal axis in all Figures 8(a) to 8(c) shows the value of inductance element L2.

[0051] If the value of inductance element L2 is less than approximately 200 μH, the generated high voltage is small relative to the large current consumption, and the series resonance frequency f1 is 1.2 times or more the parallel resonance frequency, which is a range in which disturbances in the resonance current are a concern. Experimental measurements have shown that the series resonance frequency f1 should be at least 1.3 times the parallel resonance frequency f0 (f1≦f0×1.3). Furthermore, if the current consumption increases, it becomes necessary to improve the specifications of the elements in the drive unit 30, which can easily increase costs.

[0052] On the other hand, the larger the value of inductance element L2, the less current is consumed, making it possible to reduce loss. However, when it exceeds approximately 700 μH, the generated high voltage becomes small, about three times the power supply voltage, making it unsuitable for boosting purposes. Of course, if a boost ratio of about three times the target voltage is acceptable, it can be used with even lower loss.

[0053] Next, Figure 8(d) and subsequent figures show the measurement results using inductance elements and capacitor elements with different constants. Here, the following changes were made to the above example to achieve roughly the same resonance frequency as Figure 8(c). That is, the inductance value of inductance element L1 was changed to 330 μH, which is 3.3 times, and the capacitance value of capacitor element C1 was changed to 0.033 μF, which is 0.33 times, and the characteristic graphs are shown in Figures 8(d), 8(e), and 8(f).

[0054] If the value of inductance element L2 is less than approximately 650 μH, the generated high voltage is small relative to the large current consumption, and the series resonance frequency f1 is 1.2 times or more the parallel resonance frequency f0, which is a range in which disturbances in the resonance current are a concern. On the other hand, as the value of inductance element L2 increases, the current consumption decreases, making it possible to achieve low loss. However, if the value exceeds approximately 2300 μH, the generated high voltage becomes small, making it unsuitable for voltage boosting applications.

[0055] Therefore, we found that the optimal value for inductance element L2 is approximately two to seven times the value of inductance element L1 (2 × L1 ≦ L2 ≦ 7 × L1). If a higher boost ratio is more important than lower loss, a value closer to two should be used. If lower loss is more important than higher boost ratio, a value closer to seven should be used. Furthermore, even though Figures 8(a) and 8(d) have the same resonant frequency, Figure 8(d), with its larger inductance element value, achieves lower loss and a higher boost ratio. We analyzed the factors behind this using actual inductance elements and capacitors and found that as the inductor constant increases, both its reactance and parasitic resistance increase. On the other hand, as the capacitor constant increases, the parasitic resistance increases more than the increase in reactance. This resistance increases losses and suppresses boost.

[0056] However, the resistance component R cannot be generally defined due to differences in the wire diameter and core size of the inductor, as well as differences in the type of capacitor, etc. For this reason, we measured the voltage generated using actual elements: a typical drum-type wound inductor with a diameter of about 8 mm, and a film capacitor with a size of about 5 mm to 10 mm.

[0057] This shows the magnification of the generated sine wave voltage (amplitude) to the power supply voltage when the inductance element L2, inductance element L1, and capacitor element C1 have the values ​​shown below. Note that the step-up ratio is given a slight priority over low loss, so the inductance L2 is set to a constant that is nearly three times the inductance L1. Under the conditions below, the resonant frequencies are approximately the same. Boost ratio Approximately 10 times Condition 1: L2: 1000μH, L1: 330μH, C1: 0.033μF Boost ratio Approximately 7.6 times Condition 2: L2: 470μH, L1: 150μH, C1: 0.068μF Boost ratio Approximately 5.3 times Condition 3: L2: 330μH, L1: 100μH, C1: 0.10μF Boost ratio Approximately 3.2 times Condition 4: L2: 220μH, L1: 68μH, C1: 0.15μF Boost ratio Approximately 1.9 times Condition 5: L2: 100μH, L1: 33μH, C1: 0.33μF Step-up ratio: approx. 1.2x Condition 6: L2: 68μH, L1: 22μH, C1: 0.47μF

[0058] In the case of the above general components, in order to obtain a voltage boosted by more than 5 times, it was necessary to set the inductance L1 to a value greater than or equal to condition 3. The larger L1 and smaller C1 are, the higher the boost ratio becomes. When defined as √(L1 / C1), √(L1 / C1)≧√(100 / 0.10)≒30 However, in reality, there is an upper limit to the inductance value of an inductance element that can be manufactured with the same core size, and it is desirable to determine the constant while also taking into account the component size and cost.

[0059] As described above, according to the first embodiment, a sinusoidal AC high voltage can be generated with high power efficiency using a smaller and less expensive configuration. [Example]

[0060] Next, the second embodiment will be described, focusing on the differences from the first embodiment. In the second embodiment, the microcomputer 501 is responsible for the generation of the clock signal by the VCO 2 of the first embodiment and the feedback control by the constant voltage control circuit 201. Also, while the circuit of the driving unit 30 in the first embodiment is a single-ended driving system, the boosting unit 10 is driven by a circuit of a push-pull driving system.

[0061] (Overview of the overall circuit operation) 9 is a block diagram of the remaining toner amount detection mechanism in the second embodiment, which is composed of a microcomputer 501, a high voltage application unit 51b, a capacitance detection unit 101, a first contact unit 781, and a second contact unit 782. The high voltage application unit 51b further comprises an output voltage detection circuit 202.

[0062] When the microcomputer 501 outputs a clock signal of a predetermined frequency to the resonant high-voltage generation circuit 3, the resonant high-voltage generation circuit 3 generates a sine-wave AC high voltage according to the input clock signal and applies it to the first contact 781. This sine-wave AC high voltage is divided and half-wave rectified by the diode D201, resistors R201 and R202, and capacitor C202 of the output voltage detection circuit 202, and then sent to the AD terminal of the microcomputer 501. The microcomputer 501 gradually increases the frequency of the clock signal sent to the resonant high-voltage generation circuit 3 in a sweep manner so that this detected voltage becomes a target voltage stored in a preset memory (not shown) inside the microcomputer.

[0063] The resonant high-voltage generating circuit 3 outputs a voltage corresponding to the switching-driven frequency. The microcomputer 501 repeatedly increases the sweep frequency and monitors the detected voltage, thereby performing feedback control of the AC voltage generated by the software of the microcomputer 501. Note that the diode D202 prevents the voltage from the power supply voltage Vcc2 from being input to the AD port of the microcomputer 501.

[0064] If the sweep is initiated from a frequency extremely low relative to the resonant frequency, the impedance of the inductance element becomes extremely low. This can result in a transient current flowing that is larger than normal, potentially damaging the circuit. However, increasing component specifications to prevent circuit damage would increase costs. Therefore, the clock signal being swept must start at a relatively high frequency. On the other hand, if the frequency is too close to the series resonant frequency at which the output voltage is highest, overshooting is likely to occur immediately after the sweep begins. Therefore, it is desirable to initiate the sweep at a frequency approximately 0.1 to 0.8 times the series resonant frequency f1 (0.1 × f1 ≦ f ≦ 0.8 × f1). This also applies to VCO2 in Example 1.

[0065] Fig. 10 shows an example of the frequency characteristics measured when the power supply voltage is 24 V, the inductance element L2 is 3300 μH, the inductance element L1 is 1000 μH, and the capacitor element C1 is 0.01 μF. The graph in Fig. 10 shows the frequency (drive frequency) of the clock signal.

[0066] The series resonance frequency f1 is approximately 58 kHz, where the maximum voltage is generated. Figure 10(a) shows the relationship between drive frequency and current consumption, and Figure 10(b) shows the relationship between drive frequency and the amplitude of the AC voltage generated. If the target voltage is 150 Vpp, the frequency sweep begins at approximately 40 kHz, which is within the above range. As shown in Figure 10(b), this will reach 150 Vpp at approximately 57.2 Hz, and as shown in Figure 10(a), the current consumption will be approximately 25 mA.

[0067] (Push-pull drive system drive unit) Next, a resonant high-voltage generation circuit 3b of Example 2 incorporating a drive unit 20 of a push-pull drive system will be described with reference to Fig. 11(a). The resonant high-voltage generation circuit 3b is composed of a voltage conversion switching unit 6, a drive unit 20, and a boost unit 10a. The configurations of the voltage conversion switching unit 6 and the boost unit 10a are the same as those of Example 1, and therefore a description thereof will be omitted.

[0068] The driver 20 of the second embodiment includes a transistor TR20, a coupling capacitor C20, and a diode D20. The transistor TR20 has a collector terminal connected to a power supply voltage Vcc, an emitter terminal connected to the coupling capacitor C20, and a base terminal connected to the voltage conversion switching unit 6. The diode D20 is connected between the emitter and base terminals of the transistor TR20. Specifically, the diode D20 has an anode terminal connected to the emitter terminal of the transistor TR20 and the coupling capacitor C20, and a cathode terminal connected to the base terminal of the transistor TR20 and the voltage conversion switching unit 6.

[0069] When the clock signal sent from the voltage conversion switching unit 6 is at high logic, it causes a base current to flow through the transistor TR20. This causes the amplified current to flow from the emitter terminal of the transistor TR20 to the boost unit 10a via the coupling capacitor C20 as a source current, driving the boost unit 10a. When the clock signal sent from the voltage conversion switching unit 6 transitions to low logic, the base current of the transistor TR20 stops flowing, turning the transistor TR20 off. Then, the current is sunk from the boost unit 10a via the diode D20 and the coupling capacitor C20, driving the transistor TR20. Due to the presence of the coupling capacitor C20, only push-pull AC current flows.

[0070] Unlike the first embodiment, the driver 20 of the second embodiment switches with low impedance, sourcing current from the power supply voltage Vcc and sinking current to the GND potential. Therefore, no flyback voltage is generated, as in the first embodiment, eliminating the need for a snubber circuit or high-voltage FET, enabling a more inexpensive configuration. Furthermore, the push-pull drive has the advantage that the zero-cross phases of the AC currents flowing through the inductance element L2, the inductance element L1, and the capacitor element C1 do not shift from one another. To improve the accuracy of the sine wave waveform, it is desirable to drive the driver 20 with a 50% on-duty cycle.

[0071] (Modification of the booster section) Here, we will explain the resonant high-voltage generator circuit 3c in Figure 11(b), which is different from the resonant high-voltage generator circuit 3b in Figure 11(a). The push-pull drive unit 20 and voltage conversion switching unit 6 are the same as those in Figure 11(a). The booster circuit 10a in Figure 11(a) shows a case where the inductance element L2 is connected to the power supply voltage Vcc via the inductance element L1. The booster circuit 10b in Figure 11(b) shows a case where the inductance element L2 is connected to the GND potential via the inductance element L1. That is, in Figure 11(b), one end of the inductance element L1 is connected to the ground potential. There is no significant difference between the two circuits other than the difference in the DC potential that serves as the reference for operation, which is the DC potential of the coupling capacitor C20. However, the circuit in Figure 11(a) connected to the power supply voltage Vcc has the advantage of having a slightly higher operating potential and a higher voltage at the application output unit 7.

[0072] Furthermore, boost operation by switching drive is possible even without coupling capacitor C20. However, if the on-duty of the clock signal that drives switching deviates from 50% due to variations in components, etc., DC consumption current will be superimposed, resulting in a slight increase in current consumption. Furthermore, because coupling capacitor C20 only passes AC current, it also has the effect of suppressing the flow of DC overcurrent caused by a longer on-duty period during on-off transients of the power supply voltage.

[0073] Furthermore, although the above description is of a push-pull drive circuit using discrete semiconductor transistors and diodes, it is also possible to use an integrated semiconductor half-bridge gate driver IC or the like instead.

[0074] As described above, according to the second embodiment, a sinusoidal AC high voltage can be generated with high power efficiency using a smaller and less expensive configuration. [Example]

[0075] Next, the third embodiment will be described, focusing on the differences from the first embodiment. Whereas the booster unit 10 in the first and second embodiments is configured with an inductance element and a capacitor element, the third embodiment uses a high-voltage winding transformer 12 (transformer) to generate a high voltage with a higher output.

[0076] FIG. 12 shows a resonant high-voltage generator circuit 3d according to a third embodiment. The resonant high-voltage generator circuit 3d includes a voltage conversion switching unit 6, a driver 21, and a booster 11. The driver 21 according to the third embodiment includes transistors TR20 and TR21, and a coupling capacitor C20. The collector terminal of the transistor TR20 is connected to the power supply voltage Vcc, the emitter terminal is connected to the coupling capacitor C20, and the base terminal is connected to the voltage conversion switching unit 6. The transistor TR21 is connected between the emitter and base terminals of the transistor TR20. That is, the emitter terminal of the transistor TR21 is connected to the emitter terminal of the transistor TR20 and the coupling capacitor C20, the base terminal is connected to the base terminal of the transistor TR20 and the voltage conversion switching unit 6, and the collector terminal is connected to the GND potential. The configuration of the voltage conversion switching unit 6 is the same as that of the first or second embodiment, and therefore a description thereof will be omitted.

[0077] The booster 11 is composed of an inductance element L12, a high-voltage winding transformer 12, and a high-voltage capacitor element C11. Furthermore, the high-voltage winding transformer 12 comprises a primary winding L11 and a high-voltage secondary winding L13. The inductance element L12 and the primary winding L11 are switched by the driver 21, as in the second embodiment, and a sinusoidal AC high voltage with an amplitude several times the power supply voltage Vcc is generated at the end of the primary winding L11.

[0078] In the third embodiment, the high voltage generated at the primary winding L11 is further boosted in accordance with the turns ratio between the primary winding L11 and the high-voltage secondary winding L13, and a voltage is generated at the secondary winding L13. That is, in the first and second embodiments, for example, when the power supply voltage is 24 V, a sinusoidal high voltage of approximately 100 V to 200 V is generated. In contrast, the third embodiment can generate a sinusoidal high voltage with an amplitude of several kV, and the increased dynamic range makes it easier to detect even with a smaller capacitance. Furthermore, by configuring a rectifier circuit using a high-voltage diode on the secondary side, it is possible to obtain a DC high voltage from this sinusoidal AC high voltage.

[0079] The parallel resonant frequency f0 is uniquely determined by the primary winding L11, high-voltage secondary winding L13, and high-voltage capacitor element C11 of the high-voltage winding transformer 12. The series resonant frequency f1 is uniquely determined by the primary winding L11, secondary winding L13, high-voltage capacitor element C11, and inductance element L12 of the high-voltage winding transformer 12. In the third embodiment, the inductance obtained by combining the inductance element L12 and the leakage inductance of the high-voltage winding transformer 12 functions in the same way as the inductance element L2 in the first and second embodiments. The capacitance of the high-voltage capacitor element C11 can be calculated by converting it to the primary side according to the turns ratio between the primary winding L11 and the high-voltage secondary winding L13.

[0080] Next, the driver 21 of the push-pull drive system will be described. When the clock signal sent from the voltage conversion switching unit 6 is at a high logic level, it causes a base current to flow through the transistor TR20. This causes the amplified current to flow from the emitter terminal of the transistor TR20 to the source current of the booster unit 11 via the coupling capacitor C20, driving the booster unit 11. When the clock signal sent from the voltage conversion switching unit 6 transitions to a low logic level, the base current of the transistor TR20 stops flowing, turning the transistor TR20 off. Meanwhile, the transistor TR21 draws a base current, sinking the amplified current from the collector terminal of the transistor TR21 to the GND potential via the coupling capacitor C20, driving the transistor TR21. The driver 21 described in the third embodiment can reduce the current sunk by the FET 6.

[0081] As described above, according to the third embodiment, a sinusoidal AC high voltage can be generated with high power efficiency using a smaller and less expensive configuration.

[0082] <Other embodiments> The present invention can also be realized by supplying a program that realizes one or more functions of the above-described embodiments to a system or device via a network or a storage medium, and having one or more processors in the computer of the system or device read and execute the program. It can also be realized by a circuit (e.g., ASIC) that realizes one or more functions.

[0083] The disclosure of this embodiment includes the following configuration. (Configuration 1) In a high voltage generator that converts DC voltage into AC voltage, a booster unit including a first inductor having one end connected to the DC voltage or ground potential, a second inductor having one end connected in series to the other end of the first inductor, and a capacitor connected in parallel to the first inductor; a drive unit connected in series to the other end of the second inductor and performing a switching operation in response to a drive signal; an inductance value L2 of the second inductor is greater than an inductance value L1 of the first inductor; When the frequency of resonance between the first inductor and the capacitor is a first frequency f0, the frequency of resonance between the capacitor, the first inductor, and the second inductor is a second frequency f1, and the drive frequency of the drive signal is a drive frequency f, the drive unit performs the switching operation within a range of f0≦f≦f1, The high voltage generating device is characterized in that the boosting unit is driven by the driving unit and generates a sine wave AC high voltage that is higher than the DC voltage. (Configuration 2) The L2 is 2×L1≦L2≦7×L1 The high voltage generator according to configuration 1, which satisfies the above. (Configuration 3) When the capacitance of the capacitor is C, L1 and C are expressed as follows: √(L1 / C)≧30 The high-voltage generator according to the first or second aspect of the present invention satisfies the following conditions: (Configuration 4) The first frequency f0 and the second frequency f1 are f1≦f0×1.3 4. The high-voltage generator according to any one of configurations 1 to 3, wherein the following is satisfied: (Configuration 5) 5. The high-voltage generating device according to any one of configurations 1 to 4, wherein the driving unit gradually increases the driving frequency from a value within a range of 0.1 to 0.8 times the second frequency. (Configuration 6) the driving unit has a field effect transistor with a single-ended driving method, 6. The high voltage generator according to any one of configurations 1 to 5, wherein the drive signal has an on-duty width higher than 50%. (Configuration 7) the driving unit is a driving circuit using a push-pull driving method, 6. The high voltage generator according to any one of configurations 1 to 5, wherein the drive signal has an on-duty width of 50%. (Configuration 8) 8. The high-voltage generating device according to any one of configurations 1 to 7, further comprising a rectifier circuit that rectifies the sinusoidal AC high voltage generated by the booster unit to generate a high DC voltage. (Configuration 9) 9. The high-voltage generator according to any one of configurations 1 to 8, wherein the first inductor and the second inductor are coils. (Configuration 10) the first inductor is a transformer; The high voltage generator according to any one of configurations 1 to 8, wherein the second inductor is a coil. (Configuration 11) a first electrode member; a second electrode member that pairs with the first electrode member; a capacitance detection device for detecting a capacitance between the first electrode member and the second electrode member, A high-voltage generator according to any one of configurations 1 to 10 is provided, The capacitance detection device is characterized in that the high voltage generator applies the sinusoidal AC high voltage to the first electrode member. (Configuration 12) 12. The electrostatic capacitance detection device according to claim 11, wherein the electrostatic capacitance is formed by toner of an image forming device. (Configuration 13) an image carrier that carries an electrostatic latent image; a developing means for developing the electrostatic latent image with toner to form a toner image; a storage chamber for storing toner to be supplied to the developing means; the capacitance detection device according to configuration 11 or 12 attached to the accommodation chamber; An image forming apparatus for forming an image on a recording material, comprising: the capacitance detection device detects the capacitance of the toner that has entered between the first electrode member and the second electrode member; an estimation unit that estimates the amount of toner contained in the container based on the capacitance detected by the capacitance detection device; [Explanation of symbols]

[0084] 6 Switching section 10a Booster section 30 Drive unit C1 capacitor element L1, L2 inductance elements

Claims

1. In a high voltage generator that converts DC voltage into AC voltage, a booster unit including a first inductor having one end connected to the DC voltage or a ground potential, a second inductor having one end connected in series to the other end of the first inductor, and a capacitor connected in parallel to the first inductor; a drive unit connected in series to the other end of the second inductor and performing a switching operation in response to a drive signal; Equipped with an inductance value L2 of the second inductor is greater than an inductance value L1 of the first inductor; When the frequency of resonance between the first inductor and the capacitor is a first frequency f0, the frequency of resonance between the capacitor, the first inductor, and the second inductor is a second frequency f1, and the drive frequency of the drive signal is a drive frequency f, the drive unit performs the switching operation within a range of f0≦f≦f1, The high voltage generating device is characterized in that the boosting unit is driven by the driving unit and generates a sine wave AC high voltage that is higher than the DC voltage.

2. The L2 is 2×L1≦L2≦7×L1 2. The high voltage generator according to claim 1, wherein the following is satisfied:

3. When the capacitance of the capacitor is C, L1 and C are expressed as follows: √(L1 / C)≧30 2. The high voltage generator according to claim 1, wherein the following is satisfied:

4. The first frequency f0 and the second frequency f1 are f1≦f0×1.3 2. The high voltage generator according to claim 1, wherein the following is satisfied:

5. 2. The high-voltage generator according to claim 1, wherein the driving unit gradually increases the driving frequency from a value within a range of 0.1 to 0.8 times the second frequency.

6. the driving unit has a field effect transistor with a single-ended driving method, 2. The high voltage generator according to claim 1, wherein the drive signal has an on-duty width higher than 50%.

7. the driving unit is a driving circuit using a push-pull driving method, 2. The high voltage generator according to claim 1, wherein the drive signal has an on-duty width of 50%.

8. 2. The high-voltage generating device according to claim 1, further comprising a rectifier circuit that rectifies the sinusoidal AC high voltage generated by the booster to generate a high DC voltage.

9. 2. The high voltage generator according to claim 1, wherein the first inductor and the second inductor are coils.

10. the first inductor is a transformer, 2. The high voltage generator according to claim 1, wherein the second inductor is a coil.

11. a first electrode member; a second electrode member that pairs with the first electrode member; a capacitance detection device for detecting a capacitance between the first electrode member and the second electrode member, A high voltage generator according to any one of claims 1 to 10, The capacitance detection device is characterized in that the high voltage generator applies the sinusoidal AC high voltage to the first electrode member.

12. 12. The electrostatic capacitance detection device according to claim 11, wherein the electrostatic capacitance is formed by toner of an image forming device.

13. an image carrier that carries an electrostatic latent image; a developing means for developing the electrostatic latent image with toner to form a toner image; a storage chamber for storing toner to be supplied to the developing means; The capacitance detection device according to claim 11 attached to the accommodation chamber; An image forming apparatus for forming an image on a recording material, comprising: the capacitance detection device detects the capacitance of the toner that has entered between the first electrode member and the second electrode member; an estimation unit that estimates the amount of toner contained in the container based on the capacitance detected by the capacitance detection device;

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