Transmitting device for generating an OOK-modulated spread DFT-S-OFDM wake-up signal

Abstract

An embodiment of the present invention relates to a transmitting device (100) configured to transmit an OFDM signal (510) in a communication system (500). The OFDM signal (510) comprises N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit It is obtained by spreading a sequence of N bits. bit Each spreading sequence in the sequence of N spreading sequences is a linear phase sequence with a constant rotation phase angle Φ. symb The modulation symbols are N symb The N Fourier coefficients are multiplied by the discrete Fourier transform precoder to obtain the N Fourier coefficients. Finally, the N Fourier coefficients are mapped onto the K OFDM subcarriers. symb An OFDM signal (510) comprising Fourier coefficients is transmitted. Furthermore, the present invention also relates to a corresponding method and computer program.

Description

[Technical Field]

[0001] FIELD OF THE INVENTION Embodiments of the invention relate to a transmitting device for a communication system. Furthermore, embodiments of the invention also relate to a corresponding method and computer program. [Background technology]

[0002] The concept of a wake-up signal (WUS) has been introduced into several communication standards in the wireless industry, with the goal of helping devices significantly reduce their functionality, and therefore power consumption, until the reception of such a specific WUS.

[0003] In previous 3GPP releases, Long Term Evolution Machine Type Communications (LTE-M) and Narrowband Internet of Things (NB-IoT) defined the so-called Machine Type Communication (MTC) Wake-Up Signal (MWUS) and Narrowband WUS (NWUS), respectively. These are specific types of OFDM signals designed for typical 3GPP New Radio (NR) receivers: OFDM-modulated Zadoff-Chu (ZC) sequences encoding cell identity (ID). As a result, it is an OFDM-based WUS and therefore maintains orthogonality with other signals, but also requires a high-precision analog-to-digital converter (ADC) for high-level synchronization and detection provided by the primary radio of the receiving NR device. Compared to other data channels that typically require repetition for coverage extension, MWUS / NWUS have much shorter transmission durations, carrying only a small number of bits, enabling energy savings in the receiver detector. However, energy savings are still modest because the NR receiver still needs to be in deep sleep mode, which accounts for a significant portion of the user equipment's (UE's) overall energy consumption. So far, the NWUS / MWUS feature does not appear to have been rolled out in production by network operators.

[0004] Furthermore, the current 3GPP RAN1 Rel-18 standardization is devoted to low-power WUS (LP-WUS) considerations. It is envisioned that significantly more power savings can be achieved if the primary radio of an NR receiver can be completely turned off when no messages are coming. To this end, NR devices are equipped with an additional low-power detection receiver, named a low-power wake-up receiver (LP-WUR). The WUR monitors possible incoming traffic and triggers it only when necessary, while the primary radio can be completely turned off for maximum power savings. Summary of the Invention

[0005] It is an object of embodiments of the present invention to provide a solution that alleviates or overcomes the drawbacks and problems of conventional solutions.

[0006] Another object of embodiments of the present invention is to provide a low complexity On-Off Keying (OOK) signal that can be used as, for example, a WUS.

[0007] The above and further objects are solved by the subject matter of the independent claims. Further embodiments of the invention can be found in the dependent claims. [Means for solving the problem]

[0008] According to a first aspect of the present invention, the above mentioned objects and other objects are achieved by a transmitting device for a communication system, the transmitting device comprising: N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit Spread a sequence of bits and bit each of the spreading sequences in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ; N symbTo obtain the Fourier coefficients, symb multiplying the modulation symbols by a discrete Fourier transform precoder; N mapped to K OFDM subcarriers symb Transmitting an orthogonal frequency division multiplexed OFDM signal containing Fourier coefficients This is achieved by a transmitting device configured to:

[0009] The transmitting device may be part of or fully incorporated in any suitable communication device configured for communication in a communication system. Furthermore, the transmitting device may have the capability not only to transmit communication signals but also to receive communication signals in a communication system.

[0010] An advantage of the transmitting device according to the first aspect is that a multi-bit OOK signal can be provided with lower complexity compared to conventional solutions. Furthermore, a flatter ON / OFF modulation state can also be provided, thereby potentially improving robustness against quantization errors from a low-precision ADC at the receiver device. The transmitting device according to the first aspect also allows for better control of the signal spectrum compared to conventional solutions.

[0011] In an embodiment of the transmitting device according to the first aspect, N bit Spreading bits N symb To obtain a sequence of N repeated bits, bit repeating bits; N symb To obtain N modulation symbols, symb The N repeated bits are multiplied by the concatenated spreading sequence, and the concatenated spreading sequence is a linear phase sequence with a constant rotation phase angle Φ. bit is a concatenation of spreading sequences Based on.

[0012] The advantage of this embodiment is that the definition and implementation of a single concatenated spreading sequence is bitThis may be simpler than defining and implementing a sequence of individual spreading sequences.

[0013] In an embodiment of the transmitting device according to the first aspect, N bit bits are N bit 1 / 2 bits are Manchester coded bits based on a sequence of 1 / 2 bits.

[0014] An advantage of this embodiment is that Manchester encoding allows for a transmitted signal with a constant energy level, at the cost of halving the information rate, and also eliminates the need for threshold determination for detection in the receiver device.

[0015] In an embodiment of the transmitting device according to the first aspect, the spreading sequence r l [m] is given by the following formula:

number

[0016] The advantage of this embodiment is that two angles Φ and Φ are used to generate the spreading sequence. l only needs to be defined and stored on the sending device.

[0017] In an embodiment of the transmitting device according to the first aspect, the constant rotation phase angle Φ is equal to π.

[0018] An advantage of this embodiment is that such a selection of the phase angle minimizes the envelope variation of the OOK condition.

[0019] In an embodiment of the transmitting device according to the first aspect, the spreading sequence r l [m] is an alternating sequence of values ​​+1 and -1 respectively.

[0020] The advantage of this embodiment is that it has very low complexity since no calculations, ie no multiplications, are required for the sign change.

[0021] In an embodiment of the transmitting device according to the first aspect, the spreading sequence r l [m] is an alternating sequence of two binary shift keying symbols.

[0022] An advantage of this embodiment is that it reuses constellation symbols that are already defined and implemented in 3GPP systems.

[0023] In an embodiment of the transmitting device according to the first aspect, the constant rotation phase angle Φ is given by the following formula:

number

[0024] An advantage of this embodiment is that it makes it possible to null certain Fourier coefficients, i.e., set Fourier coefficients equal to zero so that, for example, the DC subcarrier can be filtered by the circuitry of the receiver device.

[0025] In an embodiment of the transmitting device according to the first aspect, the discrete Fourier transform precoder has a size N symb ≦K.

[0026] The advantage of this embodiment is that the size of the Discrete Fourier Transform (DFT) precoder N symb But that is the number of bits N bitThe advantage of this is that the DFT precoder size N is an integer factor of the number of bits, and as a result, each bit can be spread with the same spreading factor and therefore transmitted with the same energy. Also, a DFT precoder size smaller than the WUS bandwidth K has much less complexity than a typical OFDM inverse fast Fourier transform (IFFT) size. To further reduce the complexity, the DFT precoder size N symb may be chosen to be a power of two, for example.

[0027] In an embodiment of the transmitting device according to the first aspect, the transmitting device comprises: N symb Based on the periodic repetition of N Fourier coefficients, symb Expand the Fourier coefficients to K Fourier coefficients It is configured as follows.

[0028] The advantage of this embodiment is that it symb The advantage of using more subcarriers is that it allows for mapping of the Fourier coefficients to a larger number, K, of subcarriers. Using more subcarriers allows for generating an OOK signal with sharper transitions between ON and OFF states and less variation within states. Using more subcarriers may also exploit frequency diversity to improve detection at the receiver device.

[0029] In an embodiment of the transmitting device according to the first aspect, the transmitting device comprises: To obtain the frequency-shaped Fourier coefficients, symb Multiply the N Fourier coefficients or K Fourier coefficients by the frequency domain spectrum shaping window coefficients It is configured as follows.

[0030] An advantage of this embodiment is that the frequency domain spectral shaping further flattens the OOK conditions and improves robustness to detection errors.

[0031] In an embodiment of the transmitting device according to the first aspect, the frequency domain spectral shaping window coefficients are real-valued symmetric coefficients from a bell-shaped function.

[0032] An advantage of this embodiment is that such an FDSS window is known to concentrate the energy of the DFT-s-OFDM pulse well in time, which improves the shape of the OOK signal.

[0033] In an embodiment of the transmitting device according to the first aspect, the frequency domain spectral shaping window coefficients are Kaiser window coefficients with shaping parameter β=2.

[0034] An advantage of this embodiment is that it provides a good least squares approximation of the ideal OOK signal.

[0035] In an embodiment of the transmitting device according to the first aspect, the frequency domain spectral shaping window coefficients W0[k] are given by the following equation:

number

[0036] An advantage of this embodiment is that it corresponds to an optimal least-squares approximation of an ideal OOK signal.

[0037] In an embodiment of the transmitting device according to the first aspect, the transmitting device comprises: To obtain the phase-shifted Fourier coefficients, the frequency-shaped Fourier coefficients are multiplied by a frequency-domain phase shift, which is a shift parameter T shift Based on It is configured as follows.

[0038] An advantage of this embodiment is that it can improve the time position of the OOK state by maximizing the energy of the OOK state in the target time domain period.

[0039] In an embodiment of the transmitting device according to the first aspect, the shift parameter T shift The value of is the number of samples in the OFDM signal, N fft and N symb It depends on modulation symbols.

[0040] The advantage of this embodiment is that it allows the OOK signal to fft spans an OFDM signal of N samples symb Since it consists of a multiplexing of individual time domain pulses, it may be sufficient to control the time position as mentioned above.

[0041] In an embodiment of the transmitting device according to the first aspect, the shift parameter T shift The value of is given by one of the following formulas:

number

number

number

[0042] The advantage of this embodiment is that it provides near-optimal time localization, since it corresponds to half the time difference between two consecutive time-domain pulses.

[0043] In an embodiment of the transmitting device according to the first aspect, the OFDM signal is a wake-up signal.

[0044] According to a second aspect of the present invention, the above mentioned objects and other objects are achieved by a method for a transmitting device, the method comprising: N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit spreading a sequence of N bits, bit each of the spreading sequences in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ; N symb To obtain the Fourier coefficients, symb multiplying the modulation symbols with a discrete Fourier transform precoder; N mapped to K OFDM subcarriers symb transmitting an OFDM signal comprising the Fourier coefficients; This is achieved by a method comprising:

[0045] The method according to the second aspect can be extended to embodiments corresponding to the embodiments of the transmitting device according to the first aspect, and therefore embodiments of the method include features of the corresponding embodiments of the transmitting device.

[0046] The advantages of the method according to the second aspect are the same as those of the corresponding embodiment of the transmitting device according to the first aspect.

[0047]

[0010] Embodiments of the present invention also relate to a computer program product comprising a computer readable medium and the mentioned computer program, wherein the computer program is contained on the computer readable medium and may comprise one or more from the group of read only memory (ROM), programmable ROM (PROM), erasable PROM (EPROM), flash memory, electrically erasable PROM (EEPROM), hard disk drive, etc.

[0048] Further uses and advantages of embodiments of the present invention will become apparent from the following detailed description.

[0049] The accompanying drawings are intended to clarify and explain various embodiments of the present invention. [Brief explanation of the drawings]

[0050] [Figure 1] 1 illustrates a transmitting device according to an embodiment of the present invention. [Figure 2] 3 shows a flowchart of a method for a transmitting device according to an embodiment of the present invention; [Figure 3] 1 illustrates a receiver device according to an embodiment of the present invention. [Figure 4] 1 illustrates a communication system according to one embodiment of the present invention. [Figure 5] 2 shows another block diagram of a transmitting device according to an embodiment of the present invention; [Figure 6] To illustrate the OOK state flattening effect of a selected linear phase ramp with spreading sequence r[m], a two-bit sequence

[0010] is transmitted with K=72 without FDSS. [Figure 7] We demonstrate the OOK state-flattening effect of a selected linear phase ramp in the spreading sequence r[m]. An 8-bit sequence [1 0 0 1 1 0 1 0] is transmitted with K=72 using FDSS with β=4. [Figure 8] We show the difference between equations (21) and (22) and their approximations with Φ = π, where a Manchester-encoded bit string of length N = 4 is transmitted with K = 72 and FDSS with β = 4 is used. Figure 8(a) shows the envelope of [1 0 0 1 1 0 1 0], and Figure 8(b) shows the average power of the DFT coefficients. [Figure 9] Figure 9(a) shows the nulling of the DC subcarrier for Nbit0=4 and K=72, where FDSS with β=4 is used. Figure 9(b) shows the envelope of [1 0 0 1 1 0 1 0], and Figure 9(b) shows the average power of the DFT coefficients. [Figure 10] Figure 10(a) shows the BER as a function of the phase ramp angle Φ and different FDSS coefficients β for αBPF=2 and Se=128, and Figure 10(b) shows the BER for αBPF=1 and Se=64. [Figure 11] We show the BER as a function of SNR, where Se=128 in Fig. 11(a) and Se=64 in Fig. 11(b). [Figure 12] This shows the BER performance of the WUR analog circuit filtering the DC component. [Figure 13] The time shift correction of the OOK signal by the FD phase shift is shown. [Figure 14] The advantage of FD phase shifting (TD cyclic shifting) is shown in Figure 14(a) which shows the envelope of [1 0 0 1 1 0 1 0], and Figure 14(b) shows the Manchester decoding after downsampling. [Figure 15] The BER performance is shown for Nbit0=4 and αBPF=1 with and without FD phase shift. [Figure 16] Figure 16(a) shows the PAPR as a function of phase ramp angle Φ and different FDSS coefficients β for K=24, Ne=0, and NbitO=2, Figure 16(b) shows the PAPR for K=24, Ne=8, and NbitO=8, Figure 16(c) shows the PAPR for K=72, Ne=0, and NbitO=4, and Figure 16(d) shows the PAPR for K=72, Ne=8, and NbitO=8. [Figure 17] 1 shows a representation of TD pulse multiplexing. [Figure 18] 1 illustrates the benefit of spectrum spreading in producing equal spreading factors per bit. DETAILED DESCRIPTION OF THE INVENTION

[0051] To achieve very low power consumption, WUR may better use a simple noncoherent envelope detector; as a result, WUS using OOK modulation is considered well suited. OOK typically modulates bits with two amplitude values, designated ON and OFF states (see Table 1). In practice, the amplitude values ​​of the signal states vary and depend on the pulse shape. Ideally, the OFF state has a constant amplitude value of 0, and the ON state has a constant amplitude of A≠0, where A=1 is often assumed by convention.

[0052] [Table 1]

[0053] Because all current NR channels / signals use OFDM modulation, it is desirable for legacy OFDM-based NR transmitters to be able to generate a WUS even when using a different waveform, such as OOK. Furthermore, it is desirable for the WUS to be able to be orthogonally frequency multiplexed directly with other simultaneous OFDM transmissions without interfering with them. To achieve this, the WUS should be generated based on OFDM by populating some dedicated subcarriers, where a set (e.g., K) of subcarriers for the WUS are multiplexed with subcarriers carrying other data symbols. These are then multiplied by N subcarriers before the addition of a cyclic prefix (CP). fft It can be processed with a point IFFT.

[0054] Formally, the transmitted OFDM signal s[n] is a WUS s[n] generated by a single OFDM modulation. W [n] and the data signal s D It is a superposition with [n]. Sample index - N CP ≦n≦N fft One CP-OFDM symbol with −1 is calculated as follows (normalization factor omitted for simplicity):

number

number

number

number

number

number

[0055] Unless otherwise stated, CP Head N CP =144 with N fft = 2048, which are common 3GPP numerologies, but are not limited to these.

[0056] It is therefore an object of the present invention to propose a scalable OOK-OFDM WUS waveform that is compliant with 3GPP NR transmitters and reuses already legacy components of 3GPP signals. Another object is to provide a solution that has low complexity compared to conventional solutions.

[0057] Therefore, embodiments of the present invention disclose the use of bit spreading sequences to control the shape of a signal waveform and / or its spectrum. Applications include, but are not limited to, WUS transmission in 3GPP NR. Embodiments of the present invention also disclose spreading sequences that enable very flat envelopes for the ON and OFF states of the signal, thereby providing robustness against detection errors due to noise and fading when using low-precision ADC envelope detectors in receiver devices.

[0058] Accordingly, Figure 1 illustrates a transmitting device 100 according to one embodiment of the present invention. In the embodiment illustrated in Figure 1, the transmitting device 100 comprises a processor 102, a transceiver 104, and a memory 106. The processor 102 is coupled to the transceiver 104 and the memory 106 by communication means 108 known in the art. The transmitting device 100 may be configured for wireless and / or wired communication in a communication system. The wireless communication capability may comprise an antenna or antenna array 110 coupled to the transceiver 104, and the wired communication capability may comprise, for example, a wired communication interface 112 coupled to the transceiver 104.

[0059] The processor 102 may be referred to as one or more general-purpose central processing units (CPUs), one or more digital signal processors (DSPs), one or more application-specific integrated circuits (ASICs), one or more field-programmable gate arrays (FPGAs), one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, or one or more chipsets. The memory 106 may be read-only memory, random-access memory (RAM), or non-volatile RAM (NVRAM). The transceiver 304 may be a transceiver circuit, a power controller, or an interface that provides the ability to communicate with other communication modules or devices, such as network nodes and network servers. The transceiver 104, the memory 106, and / or the processor 102 may be implemented in separate chipsets or in a common chipset. The transmitting device 100 being configured to perform a particular action may be understood in this disclosure to mean that the transmitting device 100 comprises appropriate means, such as the processor 102 and the transceiver 104, configured to perform the action.

[0060] According to an embodiment of the present invention, the transmitting device 100 bitEach bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit Spread a sequence of bits and bit The transmitting device 100 is configured such that each spreading sequence in the sequence of N spreading sequences is a linear phase sequence with a constant rotation phase angle Φ. symb To obtain N Fourier coefficients, symb The transmitting device 100 is further configured to multiply the N modulation symbols by a Discrete Fourier Transform precoder. symb The system is further configured to transmit an orthogonal frequency division multiplexed OFDM signal 510 comprising the Fourier coefficients.

[0061] Furthermore, in one embodiment of the present invention, the transmitting device 100 for the communication system 500 comprises N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit Spread a sequence of bits and bit Each spreading sequence in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ, symb To obtain N Fourier coefficients, symb The transmitting device 100 comprises a processor configured to multiply the N modulation symbols by a discrete Fourier transform precoder. symb The device further comprises a transceiver configured to transmit an orthogonal frequency division multiplexed OFDM signal 510 comprising the Fourier coefficients.

[0062] Furthermore, in yet another embodiment of the present invention, a transmitting device 100 for a communication system 500 comprises a processor and a memory having stored thereon computer readable instructions, the computer readable instructions, when executed by the processor, causing the processor to: bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit Spread a sequence of N bits bit Each spreading sequence in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ, symb To obtain N Fourier coefficients, symb N modulation symbols are multiplied by a discrete Fourier transform precoder and mapped to K OFDM subcarriers. symb transmit an orthogonal frequency division multiplexed OFDM signal 510 containing Fourier coefficients.

[0063] 2 shows a flowchart of a corresponding method 200 that may be performed in a transmitting device 100, such as the transmitting device shown in FIG. bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb To obtain N modulation symbols, bit a step 202 of spreading a sequence of N bits; bit Each spreading sequence in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ. symb To obtain N Fourier coefficients, symb The method 200 includes a step 204 of multiplying the N modulation symbols with a Discrete Fourier Transform precoder. symb The method includes a step 206 of transmitting an OFDM signal 510 comprising the number of Fourier coefficients.

[0064] Figure 3 shows a receiver device 300 according to one embodiment of the present invention. In the embodiment shown in Figure 3, the receiver device 300 comprises a processor 302, a transceiver 304, and a memory 306. The processor 302 is coupled to the transceiver 304 and the memory 306 by communication means 308 known in the art. The receiver device 300 further comprises an antenna or antenna array 310 coupled to the transceiver 304, meaning that the receiver device is configured for wireless communication in a communication system.

[0065] The processor 302 may be referred to as one or more general-purpose CPUs, one or more DSPs, one or more ASICs, one or more FPGAs, one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, or one or more chipsets. The memory 306 may be read-only memory, RAM, or NVRAM. The transceiver 104 may be a transceiver circuit, a power controller, or an interface that provides the ability to communicate with other communication modules or devices. The transceiver 304, the memory 306, and / or the processor 302 may be implemented in separate chipsets or in a common chipset. The receiver device 300 being configured to perform a particular action may be understood in this disclosure to mean that the receiver device 300 comprises appropriate means, such as the processor 302 and the transceiver 304, configured to perform the action.

[0066] According to an embodiment of the present invention, the receiver device 300 is configured to receive an OFDM signal 510 transmitted by the transmitting device 100. The OFDM signal 510 is an OOK signal due to bit spreading according to an embodiment of the present invention. Therefore, the receiver device 300 decodes the bits of the OOK signal by non-coherent detection of the envelope fluctuations of the OOK signal. A typical low-power wake-up receiver architecture for OOK signal detection is to first process the received signal 510 in the analog domain by low-pass filtering for interference cancellation and noise reduction, and then perform envelope detection directly. This processed signal is then sampled and converted to the digital domain before bit detection is performed. If the detected bit sequence corresponds to a specific bit sequence implemented in the receiver device 300, the receiving unit of the receiver device 300 for WUS detection triggers the wake-up of other radio units of the receiver device 300 if the OFDM signal 510 is a WUS.

[0067] 4 illustrates a communication system 500, such as a 3GPP NR, in accordance with an embodiment of the present invention. The communication system 500 in the disclosed embodiment comprises a transmitting device 100 and a receiver device 300 configured to communicate and operate in the communication system 500. In a non-limiting example, the transmitting device 100 may be part of a network access node, such as a base station, and the receiver device 300 may be part of a client device, such as a UE. The network access node may be connected to a core network of the communication system via a communication interface.

[0068] Thus, the network access node and the client device are configured to communicate in the downlink (DL) and uplink (UL), where the network access node uses N OFDM subcarriers mapped to K OFDM subcarriers generated in accordance with an embodiment of the present invention. symbThis implies that an OFDM signal 510 including fourier coefficients can be transmitted. Further details related to embodiments of the present invention will be described in the context of 3GPP 5G NR. Therefore, 3GPP 5G terminology, definitions, expressions, and system architectures will be used. However, it may be noted that embodiments of the present invention are not limited thereto.

[0069] Generally speaking, embodiments of the present invention can be considered to be based on the inherent time-domain multiplexing properties of DFT precoded OFDM, similar to DFT-s-OFDM already standardized in NR. For WUS applications, the DFT precoder is a DFT precoder of size N symb ≤ K, i.e., less than or equal to the number of subcarriers in the WUS bandwidth allocation, although the DFT precoder may have a different size for non-WUS applications. Each bit is spread and mapped to a sequence of modulation symbols before DFT precoding.

[0070] 5 shows a block diagram of a transmitting device 100 incorporated into the processing chain of a general communication device according to an embodiment of the present invention. A serial-to-parallel (S / P) block 130 is connected to the input of the transmitting device 100. The bit stream is divided into N bit The bit string b[l](l=0,…,N bit −1) is converted into parallel bits and then input to the transmitting device 100.

[0071] The parallel bits are provided to a spreader block 132 where each bit b[l] is spread over a total of N symb =N bit N seg N modulation symbols seg to generate corresponding modulation symbols of length N seg The spreading sequence r l [n] is multiplied by n bit bits are multiplexed together. The bits are symb =N bit Nseg coefficient N to obtain a sequence of modulation symbols seg It is spread by

[0072] Each individual bit b[l] (l=0,…,N bit -1) is the spreading sequence r l [n](n=0,…,N seg −1), so that the modulation symbols are given by:

number

number

[0073] In an embodiment of the present invention, the individual spreading sequences r l [n] is the so-called concatenated spreading sequence r[m] (m=0,…,N symb -1).

number

[0074] Therefore, the modulation symbols are of length N symb =N bit N seg Repeated bit strings of

number

[0075] The modulation symbols as output from the spreader block 132 are provided to the DFT precoding block 134 of Figure 5, thus converting the modulation symbols into Fourier coefficients. The modulation symbols are DFT precoded in the DFT precoding block 134 to provide a sequence of Fourier coefficients.

number

[0076] Before mapping the Fourier coefficients onto the K subcarriers, the output of the DFT block 134 may be provided to a signal processing block 136 that extends, shapes, and shifts the phase of the Fourier coefficients.

[0077] Before mapping the Fourier coefficients to the K OFDM subcarriers, the Fourier coefficients of the DFT precoder 134 may be provided to an optional signal processing block 136, where the Fourier coefficients are expanded by spectral expansion (SE) to fill the subcarrier allocation K, and frequency-domain spectral shaping (FDSS) windows and phase shifts may also be considered to achieve further shaping effects on the OFDM signal for improved performance. Because the size of the DFT precoder is at most equal to the number of subcarriers in the WUS bandwidth allocation, which is typically much smaller than the IFFT size of OFDM, this structure has much lower implementation complexity than conventional solutions.

[0078] SE DFT precoder size N symb is the diffusion rate N seg =N symb / N bit is an integer. To achieve this, we use N symb Instead of having a discrete Fourier transform precoder of size N symb ≦K. In one example, N symb The Fourier coefficients are N symb The Fourier coefficients are expanded to K Fourier coefficients based on a periodic repetition of the four Fourier coefficients.

[0079] Given a subcarrier allocation of K subcarriers, symb ≦K, and N symb / N bit The size of the DFT precoder, N, is an integer. symb Select, for example, the largest possible, and populate K subcarriers with SE as needed, as follows: For k=0,…,K-1, D (se) [k]=D[k+L(mod N symb )](11) where L is an integer shift. Often, N symb =K can be selected, SE is not required, D (se)[k] = D[k]. Otherwise, the relevant case is N e =(KN symb ) so that it is an even number. symb and then choose L=-N, since the following shift has the advantage of creating a symmetric spectrum: e / 2(mod N symb ) which can be written as:

number

[0080] SE also allows for modifying the number of resulting DFT-s-OFDM time-multiplexed pulses that make up the OFDM signal. Fewer pulses mean wider pulses, thus providing additional degrees of freedom to modify the overall signal shaping. In conventional solutions, SE is used as a way to reduce the peak-to-average power ratio (PAPR) at the expense of destroying the orthogonality between pulses, which increases inter-pulse interference. The motivation for using SE in this solution is different: it is used to control the spreading factor. Furthermore, for the purpose of creating an OOK signal, there is no advantage to maintaining orthogonality between pulses.

[0081] FDSS After SE, FDSS can be applied to the Fourier coefficients as follows: X ’ [k]=W[k]D (se) [k](13) where {W[0],...,W[K-1]} are the FDSS window coefficients. The FDSS allows for further shaping of the OOK waveform. Thus, N symb The Fourier coefficients or K Fourier coefficients are multiplied with the FDSS window coefficients to obtain frequency-shaped Fourier coefficients.

[0082] A related embodiment of an FDSS window is a low-PAPR window, typically real and symmetric, whose coefficients are derived from a bell-shaped function. Such a window further mitigates signal envelope fluctuations and thus smooths out OOK conditions.

[0083] In an embodiment of the present invention, the FDSS window coefficients are Kaiser window coefficients with a shaping parameter β due to their convenient parameterization. Such window coefficients have also been shown to concentrate sufficient energy of DFT-s-OFDM in the time domain (TD), which is relevant for OOK signal design. The shaping parameter may be equal to 2, i.e., β=2. It may be pointed out that the case of β=0 gives a rectangular window and is therefore equivalent to having no FDSS at all.

[0084] Other types of FDSS windows are possible, such as so-called truncated root raised cosine (RRC) filters with parameters (0.5, -0.65) or (0.5, 0.1667), 2-tap filters with coefficients [1-0.28], 3-tap filters with coefficients [-0.335 1-0.335] or [-0.28 1-0.28], etc. Thus, in an embodiment, the FDSS window coefficients are instead given by the following equation:

number

[0085] FD Phase Shift A frequency domain (FD) phase shift may also be applied to the Fourier coefficients to further shape the signal. Thus, the frequency-shaped Fourier coefficients may be multiplied with an FD phase shift to obtain phase-shifted Fourier coefficients, where the FD phase shift is a shift parameter T shift Based on.

[0086] In the example, the FD phase shift is calculated by the WUS s in Eq. (6). WTo create a TD cyclic shift in [n], it can be applied to the Fourier coefficients as follows:

number

[0087] A related embodiment is to set the pulse width to half the time difference between two successive pulses, i.e., approximately

number

number

number

number

number

number

number

number

[0088] By combining equations (11), (13), and (14) together, the signal processing block 136 obtains the Fourier coefficients D[k], k=0, . . . , N given by equation (10): symb The output of the DFT block 134 is converted to the WUS Fourier coefficients as follows: For k=0,…,K,

number

[0089] Furthermore, mapper block 138 maps the WUS Fourier coefficients from signal processing block 136 to the K subcarriers allocated for the WUS. Other data formulated in equation (2), such as other WUS or other types of data for other receiving devices, may be input to mapper block 138 and frequency multiplexed together within the same OFDM symbol. The output of mapper block 138 is provided to OFDM IFFT block 140, thereby generating a time-domain OFDM symbol. Finally, CP block 142 adds a cyclic prefix to the OFDM signal before transmission in the communication system.

[0090] Manchester Coding OOK modulation is typically performed after optional Manchester encoding of the information bits. The Manchester encoding scheme is shown in Table 2.

[0091] [Table 2]

[0092] Manchester coding creates a bit stream with a constant mean of 1 / 2. The advantage of this is that the envelope of the modulated OOK signal has a constant DC component that does not carry any information. Therefore, estimating this DC component allows for finding the optimal detection threshold. Ideally, the DC level of the envelope can be estimated and subtracted so that the WUS decision boundary is zero. However, due to fading in wireless transmissions, such threshold selection typically does not perform well when ambiguous decoding states such as "0 0" or "1 1" can occur. A better approach utilizing the principles of Manchester coding is to compare the amplitude of a first signal state with the amplitude of a second signal state to obtain information bits. It may be noted that Manchester-coded OOK is a form of pulse position modulation (PPM), and the solution disclosed herein can be directly adapted to transmit any OFDM-based pulse position modulation.

[0093] Therefore, in the embodiment of the present invention, the bit string to be spread is N bit =2N bit0 length N such that bit0 is the Manchester encoded version of the original bit string of N, or similarly, bit bits are N bit The Manchester coded bits are based on a sequence of 2 bits. The number of modulation symbols per information bit (before coding) is then 2N, which serves to create both ON and OFF states for each bit. seg N symb = K, the number of modulation symbols per OOK state, i.e., the spreading factor

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[0094] As mentioned above, the spreading sequence r[m] used in this specification allows to control the signal shape and spectrum. For this purpose, a constant envelope sequence is used as the concatenated spreading sequence r[m] according to the following formula:

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[0095] Therefore, the individual spreading sequences r l [m] is the formula

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[0096] Furthermore, one may constrain the phase of the concatenated spreading sequence r[m] to follow a linear phase with a rotating phase angle Φ, i.e., m=0,…,N symb For -1, Φ m =Φm+Φ0(18) where Φ and Φ are constant angles. The constant Φ only affects the overall phase of the signal and may be irrelevant from the perspective of a receiver with a non-coherent detector, so unless otherwise stated, we consider Φ = 0. As a result, each individual spreading sequence in equation (17) has a rotating phase angle Φ.

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[0097] Alternatively, a low complexity embodiment is to choose the same spreading sequence for each bit, which is spread over a constant angle Φ l corresponds to setting to zero, resulting in a linear phase sequence r that is independent of the bit index. l [n]=e jΦn Although such a solution can often retain most of the advantages, we have confirmed from simulations that it is suboptimal, especially since it allows us to control the coherent combining of TD-multiplexed pulses within each OOK state, but not between successive OOK states.

[0098] If the modulation symbols use equation (18) with Φ 0 =0, then equation (16) becomes:

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[0099] This is equivalent to having the bits spread by an alternating sequence of +1 and -1, and explicitly, equation (14) becomes r[m]=(-1) m Therefore, the individual spreading sequences r l [m] is in this case an alternating sequence of values ​​+1 and -1 respectively. Similarly, equation (19)

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[0100] The alternating sequence of +1 and -1 can be interpreted as an alternating sequence of two binary phase shift keying (BPSK) constellation symbols. The BPSK constellation in the 3GPP standard is

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[0101] These embodiments offer very low implementation complexity and result from minimizing the phase difference between the overlapping lobes of two adjacent pulses, as described in Appendix A. The analysis in Appendix A is only approximate, in the sense that it considers only two adjacent pulses; other adjacent pulses also contribute to the fluctuations of the state envelope. This analysis gains greater validity when increasing the shaping from the FDSS window, as other side lobes then become increasingly attenuated. Figures 6 and 7 show the resulting signal shapes depending on different angles used in the linear phase ramp. Without an FDSS, several angles, such as Φ = π / 3, π / 2, or π, provide an ON state with similar fluctuations, while Φ = π results in slightly less energy leakage in the OFF state. With an FDSS, as Φ approaches π, both the ON and OFF states become apparently constant.

[0102] More precisely, in Appendix A it is shown that minimizing the phase difference between the overlapping lobes of two adjacent pulses can be achieved by choosing the phase ramp of the concatenated spreading sequence r[m] as follows:

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[0103] Numerical evaluation using Φ = π instead of equations (21) and (22) results in a barely noticeable difference in waveform shape, as shown in Figure 8, but corresponds to different values ​​of the Fourier coefficients. However, when choosing Φ = π / M, it can be seen that the Fourier coefficients often have some nulls, where M is an integer related to the number of bits per OFDM symbol and the number of symbols for modulating the ON and OFF states. If this is undesirable, it can be easily avoided by using small deviations from these angles as in equations (21) and (22) compared to Φ = π, without affecting the waveform shape, as shown in Figure 8.

[0104] Subcarrier Nulling Another interesting embodiment is to select a linear phase of the concatenated spreading sequence r[m] that allows to null certain subcarriers. For example, in the design of Wifi WUS, the direct current (DC) subcarrier was chosen to be 0 in the case filtered by the WUR circuit. In Appendix B, the index k null ∈{0,…,N symb −1}, the output of the DFT precoding in equation (10) is seg>1, it can be nulled by choosing a constant rotation phase angle equal to D[k null It has been shown that ]=0.

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[0105] In Figure 9, K=N symb =72, N bit0 = 4, and the null index k null It has been confirmed that the subcarrier nulling effect can be obtained by considering N = 36. seg = 9, the rotation phase angle

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[0106] In the disclosed solution, a set of appropriately selected parameters can generate Fourier coefficients that generate an OFDM signal that provides the smallest least-squares (LS) approximation (also referred to as the LS method) of an ideal OOK signal under a given bandwidth allocation constraint. In conventional solutions, the Fourier coefficients that provide the LS approximation are a high-complexity method because they require the introduction of a second DFT / FFT of the same size as the OFDM modulation. Even though only K FFT outputs are required, only limited complexity reduction over the full FFT can be achieved by using a so-called pruning FFT algorithm. In fact, the performance improvement from the pruning FFT generally occurs when K<<N, at the cost of significant algorithm optimization effort. fft O(N fft log2N fft ) instead of O(N fft This is a fairly modest level of log2K.

[0107] The disclosed solution allows generating the same minimum LS approximation signal, but with a much smaller implementation complexity than the traditional LS solution. fft / N bit and N seg =N symb / N bit are both integers, Φ=π,

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[0108] Compared to the Kaiser window discussed earlier, we confirmed that the expression W0[k] can be closely approximated by a Kaiser window with shaping parameter β ≈ 2. Similarly, more complex LS methods can be used with shaping parameter β LS When used with an FDSS Kaiser window having

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[0109] The advantage of the disclosed solution compared to the naive method for LS approximation is that the same OOK signal can be obtained with a much smaller implementation complexity, since the complexity of both methods depends on the size of the respective DFT precoder. This is shown in Table 3 using two numerical examples, where it can be seen that the complexity reduction can be 2-3 orders of magnitude. This is because the FFT size in OFDM modulation is typically large, while the WUS signal subcarrier allocation is typically considered small. Furthermore, here the naive LS method is used to achieve the N fft It is assumed that this is implemented with an optimized pruning FFT algorithm at the log2K level, but otherwise this is fft log2N fft Also, the FFT size considered here is N, as in LTE. fft = 2048, but the standard FFT size in the NR specification is twice as large, N fft = 4096, which doubles the complexity of the LS method alone.

[0110] [Table 3]

[0111] The disclosed solution targets good bit error rate (BER) performance for low-power WUR. BER evaluation confirms that the embodiment with a spreading sequence r[m] with a rotating phase angle Φ = π provides the best performance, but the impact of the FDSS window is less significant. In the case of a receiver with a DC blocker, the embodiment with a corresponding null DC subcarrier is shown to maintain good performance.

[0112] The BER is a function of the WUS signal-to-noise ratio (SNR), i.e., the WUS component s[n] of the transmitted signal s[n]. W It is calculated as the power of [n] divided by the total noise power. Consider a very simple, low-power receiver where 0.15 BER is claimed to be sufficient for WUS. The OFDM transmitter has N modulated subcarriers with a total of 600 subcarriers with a subcarrier spacing of 15 kHz. fft Use = 2048, N bit0 Further assume that the WUS transmits a Manchester-coded signal with K = 4. The WUS signal is assumed to be of K = 72 subcarriers transmitted in the center of the band, while the other subcarriers on either side are modulated by random BPSK symbols. The signal reaches the receiver through a multi-tap wireless channel. Assume a time-domain linear C (TDL-C) channel model with Rayleigh fading as specified in 3GPP, with a desired delay spread of 100 ns and a speed of 3 km / h.

[0113] The received analog signal is first passed through a bandpass filter (BPF) centered on the WUS signal band to remove co-channel interference, and then sent to an envelope detector consisting of an operator norm followed by a lowpass filter to smooth the signal, whose cutoff bandwidth is determined by the coefficient α BPF and α LPFAssume third-order Butterworth filters for the bandpass filter (BPF) and lowpass filter (LPF), each with a WUS bandwidth scaled according to [0.01]. The signal is then passed through the ADC, after which the bits are decoded. The ADC is considered to be a low-precision one operating at a minimum sampling rate of one sample per OOK state and 2-bit amplitude quantization. The ADC sampling is aimed at the center of the OOK state, taking into account a reference time previously obtained by synchronization with the aid of a preamble or blind synchronization, for example, based on CP redundancy. S e Symmetric sampling interval [-S e ,S e ]. When assuming a Manchester coded signal, detection is performed by direct amplitude comparison of two consecutive samples.

[0114] In Fig. 10, we show the BER for a fixed SNR as a function of the phase ramp angle Φ and different FDSS coefficients β, where the bandwidth scaling factor of the LPF is adapted to the WUS bandwidth, i.e., α LPF = 1, and select SNRs of -2 dB and 0 dB, respectively. BPF Consider two values ​​of BPF, = 2 and 1, covering two ranges of synchronization error, i.e., the entire segment of the OOK state.

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[0115] Figure 11 compares the BER as a function of SNR between the disclosed solution and the conventional solution. e For =128, β=0, S e For =64, we choose Φ=π with β=5. As can be seen in Figure 11, the proposed solution e = 64, i.e., only when the synchronization error is not too large can the LS approximation (called the LS method in the figure) be slightly better in BER by using more FDSS shaping. This improvement is small, and in general the disclosed solution provides the same BER performance as the LS method by using Φ = π and β = 2, as explained above. Nevertheless, recall that the disclosed solution is much less complex than the LS method while providing additional degrees of freedom for optimization.

[0116] For further comparison, the disclosed solution is shown to offer significant improvement over a similar but naive scheme in which bits are spread by mapping them to random symbols in a BPSK or π / 2-BPSK constellation before applying DFT-s-OFDM modulation. Note that π / 2-BPSK incorporates a linear phase ramp with an angle of π / 2 between successive symbols by construction. Using FDSS with π / 2-BPSK can further improve performance; here, we find that the best shaping is β=3. The BER for all curves can be improved by narrowing the bandwidth of the BPF or LPF. Nevertheless, for lower power consumption, a larger filter bandwidth may be desirable instead.

[0117] Finally, consider an embodiment in which the WUR blocks the DC component of the received signal at its analog front. The BER performance evaluation is performed as before with the parameter K=N symb =72, N bit0 =4, N seg = 9, index k null With a DC subcarrier at =36, the phase ramp angle is

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[0118] The FD phase shift corresponding to cyclically shifting the main lobes of the TD multiplexed pulses by half their period, i.e.

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[0119] Figure 13 shows the localized correction obtained for a 2-bit OOK signal where the ON and OFF states are expected to span each half of the OFDM symbol duration. e N = 8 was used for spectrum extension, resulting in K = 24 subcarriers symb It consists of π = 16 pulses. Linear phase ramping with Φ = π is used, and the Fourier coefficients are shaped by an FDSS window with parameter β = 4. As can be seen in Figure 13, the OFF state has its envelope reaching an amplitude close to unity.

[0120] Correcting this time offset allows for greater robustness against synchronization errors. Figure 14 shows how this FD phase shift helps reduce the decoding error in the case of large synchronization errors. Here, the signal is downsampled at a minimum rate of one sample per state, and downsampling starts at the 220th sample instead of the 128th sample in the middle of the waveform. The 8-bit sequence considered corresponds to the Manchester encoding of the bits [1 0 1 1]. Therefore, the bits can be decoded by amplitude (or energy) comparison of two consecutive samples, as shown in Figure 15(b). As can be seen, T shiftThe signal taking into account the time offset correction of will correctly decode the bits to [1 0 1 1], while the signal without time offset compensation will output two decoded bit errors in the sequence [1 0 0 0].

[0121] The difference in BER performance with and without FD phase shift is shown in Figure 15, where we assume the same system scenario as described earlier. Here, the synchronization error bound is calculated for one perfect OOK condition, i.e., S e = 128, and the OOK waveforms are generated using Φ = π and β = 2. Two cases of bandwidth allocation are considered, namely K = 24 and K = 72, and it can be seen that the performance gap is larger with smaller bandwidth allocation because the pulse lobes are larger in this case and have more energy leakage to other OOK states without proper TD shifting as achieved by the proposed FD phase rotation.

[0122] A possible alternative to FD phase shifting is to use guard symbols, in which some symbols at the input of the DFT precoder are systematically set to zero. Guard symbols can also be used between different states to avoid energy leakage between the ON and OFF states. However, using guard symbols is generally suboptimal because it reduces the width of the ON state and makes the signal more sensitive to synchronization errors. Note that guard symbols are different from guard subcarriers at the input of the OFDM modulation. Guard subcarriers can also be beneficial to the disclosed solution to reduce interference from simultaneous data transmission.

[0123] The PAPR performance of the proposed solution is considered both for standalone WUS transmission, i.e., when there is no other simultaneous data transmitted together. The maximum PAPR as a function of the rotation phase angle Φ is given by K and N bit0 Four different combinations of N are shown in Figure 16. bit0 In the case of =8, Ne A spectral spread with .gtoreq.8 is used. A good PAPR of about 4 dB is achieved primarily in the range that coincides with the angle that provides good BER performance.

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[0124] It can be seen that the special case of the Fourier coefficients in equation (58) can be implemented as DFT precoding of the bits without explicit spreading as follows:

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[0125] Like before

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[0126] As previously mentioned, the transmitting device 100 disclosed herein may be any type of suitable communications device. Non-limiting examples are network access nodes and client devices.

[0127] A network access node herein may also be referred to as a radio network access node, access network access node, access point (AP), or base station (BS), e.g., a radio base station (RBS), which in some networks may be referred to as a transmitter, "gNB," "gNodeB," "eNB," "eNodeB," "NodeB," or "B-node" depending on the standard, technology, and terminology used. Radio network access nodes may be of different classes or types, e.g., macro eNodeB, home eNodeB, or pico base station, based on transmit power and therefore cell size. A radio network access node may also be a station, which is any device that includes an IEEE 802.11-compliant medium access control (MAC) and physical layer (PHY) interface to the wireless medium (WM). The radio network access node may be configured for communication in fifth generation (5G) wireless systems such as 3GPP-related Long Term Evolution (LTE), LTE-Advanced, New Radio (NR), and their evolutions, as well as IEEE-related Wi-Fi, Worldwide Interoperability for Microwave Access (WiMAX®), and their evolutions.

[0128] A client device herein may be referred to as a user device, user equipment (UE), mobile station, Internet of Things (IoT) device, sensor device, wireless terminal, and / or mobile terminal, and is enabled to communicate wirelessly in a wireless communication system, sometimes referred to as a cellular radio system. A UE may also be referred to as a mobile phone, cellular telephone, computer tablet, or laptop with wireless capabilities. A UE in this context may be, for example, a portable, pocketable, handheld, computer-integrated, or vehicle-mounted mobile device enabled to communicate voice and / or data with another communication entity, such as another receiver or server, via a radio access network (RAN). A UE may also be a station, which is any device that includes an IEEE 802.11-compliant MAC and PHY interface to a WM. A UE may be configured for communication in 5G wireless systems, such as 3GPP-related LTE, LTE-Advanced, NR, and their evolutions, as well as IEEE-related Wi-Fi, WiMAX, and their evolutions.

[0129] Furthermore, any method according to the embodiments of the present invention may be implemented in a computer program having code means which, when executed by a processing means, causes the processing means to perform the steps of the method. The computer program is included in a computer-readable medium of a computer program product. The computer-readable medium may include essentially any memory, such as the above-mentioned ROM, PROM, EPROM, flash memory, EEPROM, or hard disk drive.

[0130] It should further be appreciated that the transmitting device 100 comprises necessary communication capabilities, e.g., in the form of functions, means, units, elements, etc., to perform or implement embodiments of the present invention. Examples of other such means, units, elements, and functions are processors, memories, buffers, control logic, encoders, decoders, rate matchers, de-rate matchers, mapping units, multipliers, decision units, selection units, switches, interleavers, de-interleavers, modulators, demodulators, inputs, outputs, antennas, amplifiers, receiver units, transmitter units, DSPs, TCM encoders, TCM decoders, power supply units, power feeders, communication interfaces, communication protocols, etc., suitably arranged together to perform the solutions.

[0131] Thus, the processor of transmitting device 1000 may comprise, for example, one or more instances of a CPU, processing unit, processing circuit, processor, ASIC, microprocessor, or other processing logic capable of interpreting and executing instructions. Thus, the term "processor" may refer to a processing circuit comprising multiple processing circuits, such as, for example, any, some, or all of those mentioned above. The processing circuit may further perform data processing functions for inputting, outputting, and processing data, including data buffering and device control functions, such as call processing control or user interface control.

[0132] Finally, it is to be understood that the present invention is not limited to the embodiments described above, but also covers any and all embodiments falling within the scope of the appended independent claims.

[0133] Appendix A Without loss of generality, we assume that the starting WUS subcarrier is K0 = 0. After inserting equations (9) to (15) into equation (6), T shift Assuming that is an integer, WUS is equal to:

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[0134] This filter is the inverse discrete Fourier transform of the FDSS window. Without FDSS windowing, W[0]=…=W[N sc -1]=1, this means that N sc This is further reduced to a DFT-s-OFDM pulse in the form of a Dirichlet kernel with modulated subcarriers.

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[0135] In a typical windowing function W[k], the pulse |g m [n]| remains essentially sinc shaped, but with more or less attenuated side lobes.

[0136] An illustration of the resulting TD pulse multiplexing effect of DFT precoding in OFDM is shown in Figure 17. Here, K = 24 subcarriers are considered, but the size N e Since SE = 8 is used, N symb = 16 pulses. The pulses are

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[0137] Now, by inserting equation (19) into equation (31), the signal becomes equal to:

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[0138] Each pulse g m [n] is the sample interval

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[0139] Based on this structure, the main design goal becomes finding the relevant pulse phase rotation that specifically guarantees coherent combining of pulses within the same segment.

[0140] The phase difference between two adjacent pulses of index m and (m+1) is:

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[0141] The function θ[n]={0 or π} corresponds to the sign difference between the real parts of the pulses and varies as a function of n. Nevertheless, in the case without FDSS and without spectral broadening, it can be verified that this constant is equal to θ[n]=0 for all samples between two adjacent pulses.

[0142] Therefore, θ[n]=0, L=0, and N e Assuming =0, we get:

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[0143] Therefore,

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[0144] In the SE case, it can be verified that θ[n]=0 also for samples where the main lobes of adjacent pulses intersect, so we can assume that θ[n]=0 here as well.

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[0145] Appendix B The modulation symbol is d[m]=b r [m]e jΦm is chosen to be, where b ris a sequence made up of segments of 1's and segments of 0's, each segment of 1's having length N seg Recall that we have N of 1. ones There are segments, and segment index is 1.

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[0146] Therefore, N seg >1 and

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[0147] In the absence of SE, D[k null ] is the subcarrier index k null ∈{0,…,K-1}. Instead, in the case of SE with a shift L, the Fourier coefficients D[k null ]teeth,

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[0148] Appendix C As mentioned earlier, the Fourier coefficients that generate the LS approximation are N fft It is obtained from a point FFT.

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[0149] For the center subcarrier, we obtain:

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[0150] In the solution disclosed herein, the bits are multiplied by the coefficient N seg =N symb / N bit and is diffused by a pulse phase ramp with angle Φ, so that k=0,…,N symb For -1 we get:

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[0151]

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[0152] Instead,

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[0153] Now, comparing equations (52) and (58), it can be seen that the Fourier coefficients differ only by an amplitude factor and an overall phase, both of which are independent of the data bits. Explicitly, in the disclosed solution,

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[0154] Appendix D Considering the subcarrier allocation K, the spreading factor N seg We show that for cases where , is not an exact integer, using SE provides better performance than using two spreading factors for the ON and OFF states when possible. The evaluation scenario is for Φ = π, β = 2, and α BPF = 1. Here, K=72 and N bit0 = 8, and therefore

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[0155] 100 sending devices 102 processors 104 Transceiver 106 memory 108 Means of communication 110 Antenna 200 ways 300 receiver devices 302 processor 304 Transceiver 306 memory 308 Means of communication 310 Antenna 500 Communication Systems 510 OFDM signal

Claims

1. A transmitting device (100) for a communication system (500), said transmitting device (100) comprising: N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb to obtain N modulation symbols. bit Spreading a sequence of N bits bit each of the spreading sequences in the sequence of spreading sequences is a linear phase sequence with a constant rotation phase angle Φ; N symb to obtain the N Fourier coefficients symb multiplying the modulation symbols by a discrete Fourier transform precoder; The N subcarriers are mapped to K OFDM subcarriers. symb transmitting an orthogonal frequency division multiplexed OFDM signal (510) comprising fourier coefficients A transmitting device (100) configured to:

2. N bit Spreading bits N symb to obtain a sequence of N repeated bits. bit repeating bits; N symb To obtain N modulation symbols, symb The N repeated bits are multiplied by a concatenated spreading sequence, and the concatenated spreading sequence is a linear phase sequence with a constant rotation phase angle Φ. bit is a concatenation of spreading sequences The transmitting device (100) of claim 1, based on

3. N bit bits are N bit 3. The transmitting device (100) according to claim 1 or 2, wherein the coded bits are Manchester coded bits based on a sequence of / 2 bits.

4. the spreading sequence r l [m] is given by the following formula: [Equation 1] where l is the bit index, m is the modulation symbol index, e is the natural exponential function, j is the imaginary unit, and Φ l The transmitting device (100) according to any one of claims 1 to 3, wherein is a constant angle that depends on the bit index l.

5. The transmitting device (100) of claim 4, wherein the constant rotational phase angle Φ is equal to π.

6. the spreading sequence r l The transmitting device (100) according to claim 4 or 5, wherein [m] is an alternating sequence of values ​​+1 and −1, respectively.

7. the spreading sequence r l The transmitting device (100) of claim 4 or 5, wherein [m] is an alternating sequence of two binary shift keying symbols.

8. The constant rotational phase angle Φ is given by the following equation: [Equation 2] where N seg is the spreading sequence r l [m], and k null 5. The transmitting device (100) of claim 4, wherein λ is an index of the nulled Fourier coefficients and λ is any non-zero integer.

9. The discrete Fourier transform precoder has size N symb The transmitting device (100) of any one of claims 1 to 8, with ≦K.

10. N symb based on periodic repetition of the N Fourier coefficients, symb Expand the Fourier coefficients to K Fourier coefficients The transmitting device (100) of any one of claims 1 to 9, configured to:

11. To obtain the frequency-shaped Fourier coefficients, symb multiplying the K Fourier coefficients or the K Fourier coefficients by a frequency domain spectrum shaping window coefficient; The transmitting device (100) according to claim 9 or 10, configured to:

12. The transmitting device (100) of claim 11, wherein the frequency domain spectral shaping window coefficients are real-valued symmetric coefficients from a bell-shaped function.

13. The transmitting device (100) of claim 12, wherein the frequency domain spectral shaping window coefficients are Kaiser window coefficients with a shaping parameter β=2.

14. The frequency domain spectrum shaping window coefficients W 0 [k] is given by the following formula: [Equation 3] where N fft The transmitting device (100) of claim 11, wherein ∑ is the number of samples of the OFDM signal (510) and sin( ) is a sine function.

15. The frequency-shaped Fourier coefficients are multiplied by a frequency-domain phase shift to obtain phase-shifted Fourier coefficients, the frequency-domain phase shift being a shift parameter T shift Based on A transmitting device (100) according to any one of claims 11 to 14, configured to:

16. The shift parameter T shift The value of is the number of samples N of the OFDM signal (510). fft and N symb The transmitting device (100) of claim 15, wherein the transmitting device (100) depends on modulation symbols.

17. The shift parameter T shift The value of is given by any one of the following formulas: [Equation 4] where N fft is the number of samples of the OFDM signal (510), [Equation 5] is the ceiling function, [Equation 6] The transmitting device (100) of claim 16, wherein: is a floor function and round[ ] is a rounding function.

18. The transmitting device (100) of any one of claims 1 to 17, wherein the OFDM signal (510) is a wake-up signal.

19. A method (200) for a transmitting device (100), said method (200) comprising: N bit Each bit in the sequence of bits and N bit N spreading sequences are multiplied by the corresponding spreading sequences in the sequence symb to obtain N modulation symbols. bit A step (202) of spreading a sequence of N bits, bit Step (202), wherein each of the spreading sequences in the sequence of spreading sequences is a linear phase sequence having a constant rotation phase angle Φ; N symb to obtain the N Fourier coefficients symb multiplying (204) modulation symbols by a Discrete Fourier Transform precoder; The N subcarriers are mapped to K OFDM subcarriers. symb transmitting (206) an OFDM signal (510) comprising the Fourier coefficients; A method (200) comprising:

20. 20. A computer program having a program code for performing the method according to claim 19, when the computer program runs on a computer.

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