Continuously Variable Active Reactance System and Method

Abstract

Various embodiments are described for controlling the resonant frequency of a resonator. The system includes at least one resonant circuit and an active variable reactance circuit that controls the resonant frequency of the at least one resonant circuit. The active variable reactance circuit includes an electrically controllable switching element and a switch controller subcircuit configured to switch the electrically controllable switching element at a frequency of a radio frequency (RF) current or voltage passing through or across the device such that the RF current flowing from a first terminal to a second terminal is substantially sinusoidal.

Description

[Technical Field]

[0001] This disclosure relates to a continuously variable active reactance system and method.

[0002] [Cross-reference of related applications] This application claims the benefits and priority of U.S. Provisional Patent Application No. 63 / 071,048, entitled "Continuously Variable Active Reactance Systems and Methods," filed on 27 August 2020, the contents of which are incorporated herein by reference in their entirety. [Background technology]

[0003] Resonant radio-power systems utilize magnetic or electrical coupling between LC resonators to transmit power. The efficiency of such systems depends on the resonator quality factor; a higher quality factor results in higher radio-power transmission efficiency. However, resonators with higher quality factors exhibit a narrower resonant response curve; that is, to fully realize the potential efficiency improvements gained from a high quality factor, the drive frequency must be brought close to the resonant frequency of the resonator. Therefore, in systems with a fixed drive frequency, and / or systems with two or more resonators, precisely controlling the resonant frequencies of all resonators is essential to optimizing system performance. [Brief explanation of the drawing]

[0004] Many aspects of this disclosure can be better understood by referring to the following drawings. The elements of the drawings are not necessarily drawn to a fixed scale, and the focus is on clearly illustrating the principles of the disclosure. Furthermore, in the drawings, the same reference number indicates a corresponding part across multiple drawings.

[0005] [Figure 1A] Figure 1A is an example of a simplified schematic diagram of an LC resonator having electrically controlled variable capacitance according to various embodiments of the present disclosure. [Figure 1B] Figure 1B is another example of a simplified schematic diagram of an LC resonator having electrically controlled variable inductance according to various embodiments of the present disclosure. [Figure 2A] Figure 2A is an example of a simplified circuit diagram for providing an actively variable capacitance according to various embodiments of the present disclosure. [Figure 2B] Figure 2B is a simulated ideal waveform diagram of the simplified circuit diagram of Figure 2A according to various embodiments of the present disclosure. [Figure 3A] Figure 3A is an example of a simplified circuit diagram for providing an actively variable capacitance according to various embodiments of the present disclosure. [Figure 3B] Figure 3B is a simulation ideal waveform diagram of a simplified circuit diagram of Figure 3A according to various embodiments of the present disclosure. [Figure 4A] Figure 4A is a graph showing the effective capacitive reactance versus duty cycle for various embodiments of the present disclosure. [Figure 4B] Figure 4B is a graph showing effective induced susceptance versus duty cycle for various embodiments of the present disclosure. [Figure 5] Figure 5 is a photograph of a printed circuit board (PCB) that implements an actively variable capacitance circuit used to collect the measurements shown in Figures 6 and 7, according to various embodiments of the present disclosure. [Figure 6] Figure 6 is a graph showing the measured waveforms of the actively variable capacitance of the circuit of Figure 5 according to various embodiments of the present disclosure. [Figure 7] Figure 7 is a graph showing the measured resonant frequency versus duty cycle of a resonator including the active variable capacitance circuit of Figure 5, according to various embodiments of the present disclosure. [Figure 8] Figure 8 is an example of a simplified diagram of an automatic tuning resonant repeater according to various embodiments of the present disclosure. [Figure 9] Figure 9 shows photographs of an automatic tuning resonant repeater and a direct drive resonant according to various embodiments of the present disclosure. [Figure 10]Figure 10 is a schematic block diagram of a system for an actively variable capacitive reactance according to various embodiments of the present disclosure. [Figure 11] Figure 11 is a schematic block diagram of a system for active variable inductive reactance according to various embodiments of the present disclosure. [Figure 12] Figure 12 is a graph showing the simulation results of the ideal voltage and current waveforms for an actively variable capacitance. [Figure 13] Figure 13 is a graph showing the effective capacitive reactance as a function of the duty cycle. [Figure 14] Figure 14 is a graph showing a simulation of the ideal voltage and current waveforms for an actively variable inductance. [Figure 15] Figure 15 is a graph showing the effective induced susceptance as a function of the duty cycle. [Figure 16] Figure 16 is a schematic block diagram of a circuit for automatically adjusting the RF current amplitude of an externally driven LC resonator using tuning control with an actively variable reactance, according to various embodiments of the present disclosure. [Figure 17] Figure 17 is a schematic block diagram of a circuit for automatic RF current regulation of an LC resonator using actively variable capacitance, according to various embodiments of the present disclosure. [Figure 18] Figure 18 shows an E-class RF generator with automatic zero-voltage switching according to various embodiments of the present disclosure. [Figure 19] Figure 19 is a chart showing measured gate and drain voltage waveforms of a Class E amplifier with automatic zero-voltage switching under various load conditions, according to various embodiments of the present disclosure. [Figure 20] Figure 20 is a chart showing measured gate and drain voltage waveforms for a Class E amplifier with automatic zero-voltage switching for various tuning conditions, according to various embodiments of the present disclosure. [Figure 21]Figure 21 is a chart showing the duty cycle of an AZVS amplifier as measured as a function of the resonant frequency of the tank circuit, according to various embodiments of the present disclosure. [Figure 22] Figure 22 is a schematic diagram of an E-class RF generator with AZVS and automatic tuning according to various embodiments of the present disclosure. [Figure 23] Figure 23 is a schematic diagram of an E-class RF generator that achieves ZVS by tuning using a variable reactance in series with an LC tank circuit, according to various embodiments of the present disclosure. [Figure 24] Figure 24 is a schematic diagram of a Class E amplifier, according to various embodiments of the present disclosure, that has been reconfigured to draw direct current (DC) power from the same two terminals that an RF generator uses to output RF power. [Figure 25] Figure 25 is a simplified schematic diagram of a resonant magnetic loop antenna driven by a distributed RF generator according to various embodiments of the present disclosure. [Figure 26] Figure 26 is a schematic block diagram of an RF generator for use as part of a distributed RF generator system with AZVS and auto-tuning, according to various embodiments of the present disclosure. [Modes for carrying out the invention]

[0006] As described above, in systems with a fixed drive frequency and / or systems with two or more resonators, it is essential to precisely control the resonant frequencies of all resonators to optimize system performance. Resonator detuning can be caused by various factors, such as fluctuations in component values ​​and interactions with the environment. Therefore, it is desirable to have a method for controlling the resonant frequencies of resonators to compensate for tuning errors.

[0007] Referring here to the drawings, Figures 1A and 1B show two exemplary embodiments for controlling the resonant frequency of an LC resonator. In both examples, the resonant LC tank circuit 10 includes an inductor L T and capacitor CT It has a variable capacitor C. Refer to Figure 1A. VAR is C T It is in series with the other component, which allows the total series capacitance of the LC resonator to be changed. The resonant frequency of the resonator in Figure 1A is given by the following equation.

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[0008] In Figure 1B, the variable inductance L VAR is L T It is in parallel with the other components, and the total parallel inductance of the LC resonator can be changed. The resonant frequency of the resonator in Figure 1B is given by the following equation.

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[0009] The variable reactance element may be electrically controllable so that the LC resonators shown in Figures 1A and 1B can respond to environmental changes without human control. Electrically controllable reactance can be created using an electromechanical system, a nonlinear device, or an array of individually switchable inductors or capacitors. However, all of these approaches have disadvantages when used for wireless power transmission. Electromechanical systems and individually switchable reactance arrays are costly and complex, while nonlinear devices require DC control voltages and currents much larger than the RF voltage and current amplitudes of the LC tank circuit 10, which is often impractical for high-voltage systems. Therefore, it is desirable to find a means of constructing an electrically controllable reactance element that does not require an electromechanical device or nonlinear reactance, and can achieve a continuous tuning range with minimal complexity using minimal switching devices.

[0010] One way to achieve electrical control is to use an electric motor to adjust the tuning of a variable capacitor or inductor. However, these electromechanical solutions are often bulky and complex. Another way to achieve electrical control is to use nonlinear reactance elements such as varactors or saturable ferrites. However, these nonlinear devices require a DC control voltage or current that is much larger than the amplitude of the RF current or voltage passing through or across the device in order to operate as variable reactance. This condition can be easily met when resonating an LC resonator with a weak signal, such as in the tank circuit of a radio receiver. However, if the LC resonator is part of a high-power radio system where the amplitude of the RF voltage and current within the resonator is already quite large, satisfying this condition is often difficult or impossible. Another method that does not require electromechanical components while maintaining linearity is to use electrically controlled switches, such as metal-oxide-semiconductor field-effect transistors (MOSFETs), bipolar junction transistors (BJTs), or relays, to switch the reactance elements inside and outside the LC resonator.

[0011] Each state of the switch provides a single tuning state. Therefore, the number of possible states is the number of switches plus two. However, this approach has various drawbacks. First, only specific individual tuning frequencies can be achieved. Tuning over a continuous range is not possible. Second, the system requires more switches, making it more complex as the number of tuning states increases. Third, the system requires digital control, which also complicates the system. Therefore, it is desirable to provide electrical control that can achieve a continuous tuning range with minimal complexity without requiring electromechanical devices or non-linear reactances and using a minimum number of switching devices.

[0012] The desirable attributes described above may be achieved with an active variable reactance. Thus, FIGS. 2A and 2B show an exemplary circuit in which an active variable reactance can be constructed using a single electrically controllable switching device S1 that is switched on and off at the frequency of a sinusoidal RF current or voltage within the LC tank circuit 10. The LC tank circuit 10 may be powered indirectly, for example, via inductive or capacitive coupling to an external RF source having a drive frequency fd.

[0013] The tuning of the resonator shown in FIGS. 2A and 3A may be adjusted by changing the duty cycle of the switch. Using just one of these electrically controllable switching devices, these circuits can generate a variable reactance without requiring the large DC voltages and currents required by complex electromechanical systems, switch arrays, or non-linear systems.

[0014] In the active variable capacitance shown in FIG. 2A, the LC tank circuit 10 ensures that the RF current I flowing from terminal T1 to terminal T2 RF is substantially sinusoidal. This RF current I RF is shown in the upper plot of FIG. 2B. The central plot shows the switch control rectangular wave. The phase of the rectangular wave is selected such that the zero crossing of the negative slope of the RF current I RF occurs at the midpoint of the low period. The lower plot is Cs >>C T Capacitor C under the assumption that s The voltage V across both ends Cs This indicates that when the switch is off, capacitor C s It is first charged, then current I RF It is discharged by I RF The phase relationship between the switch-controlled square wave and the switch ensures zero-voltage switching, and when the switch is turned on, C s Discharging ensures that the switch does not consume energy. Voltage waveform V Cs (t) contains many harmonics. However, the LC tank circuit 10 functions primarily as a filter that responds to the fundamental wave. Therefore, the effective reactance χ c This can be defined by dividing the amplitude of the fundamental wave by the amplitude of the RF current.

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[0015] In the actively variable inductance shown in Figure 3A, the LC tank circuit 10 controls the RF voltage V between terminal T1 and terminal T2. RF This ensures that the RF voltage is approximately a sine wave. This RF voltage is shown in the upper plot of Figure 3B. The middle plot shows a switch-controlled square wave. The phase of the square wave is determined by the RF voltage V. RF The zero crossing of the negative slope is selected so that it occurs at the midpoint of the long period. The plot below shows L s >>L TAssuming the inductor L s Current I flowing through Ls This indicates that when the switch is on, the voltage V RF The inductor current first rises, then falls. RF The phase relationship between the current waveform I and the switch-controlled rectangular wave ensures zero-current switching and guarantees that the switch does not consume energy. Ls (t) contains many harmonics. However, the LC tank circuit 10 functions as a filter that primarily responds to the fundamental wave. Therefore, the effective susceptance B l This can be defined by dividing the amplitude of the fundamental wave by the amplitude of the RF voltage.

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[0016] Figure 5 shows an actively variable capacitance circuit soldered to an inductive loop made from copper tape, although other materials may be used. The loop was inductively driven by an external RF source. The switch was implemented with a parallel pair of MOSFETs. To generate the gate drive signal, a sinusoidal voltage across a capacitor connected in series with the loop is sent to a comparator via a phase-shift network to generate a square wave, which is converted to a triangular wave by an RC filter and fed to a second comparator to generate a square wave with a controllable duty cycle.

[0017] Figure 6 shows the measured gate, drain, and RF current waveforms, which closely match the ideal waveform shapes shown in Figure 2B. More specifically, Figure 5 shows the measured waveforms of an actively variable capacitance circuit. Current I RF This was determined from the time derivative of the voltage across the series capacitor. A 200 MHz low-pass filter was applied in software to remove the high-frequency noise amplified by the derivative.

[0018] To test the effectiveness of Equation 6, the resonator was inductively coupled to the driven loop. The resonant frequency was measured by varying the gate drive duty cycle and finding the drive frequency at which the RF current in the resonator was maximized for each duty cycle value. The results, plotted in Figure 7, show a good agreement with the predictions of Equation 6. Specifically, Figure 7 shows the duty cycles against the measured resonant frequency of a resonator including the actively variable capacitance circuit shown in Figure 5.

[0019] The ability to electrically control the tuning of LC resonators has many applications in resonant radio power transfer. One example is the system shown in Figure 8. In this system, an actively variable capacitance is used in combination with a feedback circuit to maintain a constant RF current amplitude within a radio resonant repeater. The constant RF current resonant repeater 20 (e.g., the lower resonator in the schematic diagram of Figure 8 and the photograph in Figure 9) is driven at 6.78 MHz and inductively coupled to a directly fed resonator 25 (the upper resonator in the schematic diagram of Figure 8 and the photograph in Figure 9). Each resonator measures approximately 25 cm x 50 cm. The resonant repeater 20 maintains a constant RF current amplitude I RF The tuning is automatically adjusted to maintain the desired state. The direct resonator and resonant repeater each supply power to two tuned wireless 360mW LED loads (e.g., the LED lights shown in Figure 9). Note that the current amplitudes of inductors L1 and L2 are equal, and their circulation directions are opposite. In the direct-fed resonator, capacitors C5 and C6 are selected so that the RF current amplitudes of L3 and L4 are equal, and their circulation directions are opposite.

[0020] Capacitors Cs, C1, and C2 can be selected such that, for any duty cycle of the actively variable capacitance, the resonant frequency of the resonant repeater is always higher than the drive frequency generated by the RF generator. This means that the RF current amplitude is a monotonic function of the duty cycle. The automatic tuning circuit controls the RF current amplitude I of the resonant repeater 20. RF The rectified DC voltage across C3, which is proportional to the value of the resonant repeater, is measured. To keep this rectified DC voltage constant, a feedback loop may be provided to adjust the duty cycle of the gate-driven square wave. As a result, this feedback loop automatically changes the tuning of the resonant repeater, and the inductive coupling k13 and k24 is changed by moving the resonant repeater relative to the directly fed resonant, even if the RF current amplitude I RF The goal was to maintain a constant value. The system shown in Figure 8 was able to maintain the RF current amplitude in the resonant repeater within ±1% of 2.83A when the separation between the direct resonator and the resonant repeater varied from 2.0 cm to 5.5 cm.

[0021] The actively variable capacitance and actively variable inductance circuits shown in Figures 2A and 3A, respectively, enable the resonant frequency of an LC resonator to be electrically controlled over a continuous range at high RF power levels with low cost and complexity, using a single switching device. Electrical control of resonance has many applications in resonant radio power transmission. One example shown herein is the automatic control of RF current amplitude in a resonant repeater via a feedback loop controlling an actively variable capacitance. Other potential applications include making radio transmitters or receivers less susceptible to component variations and environmental perturbations.

[0022] Figures 10 and 11 illustrate exemplary embodiments of how an electrically controllable linear variable reactance can be configured using an electrically controllable switching element S1 that passes through the device or is switched on and off at the frequency of an RF current or voltage across the device. Figure 10 includes a switch controller 300, a DC power input 305, a reactance control input 310, an electrically controllable switching element S1, terminals T1 and T2, a capacitor C1, and a current pickup device 315. The switch controller 300 includes a DC power input 320, an RF pickup input 325, a duty cycle control input 330, and a switch control output 335. The electrically controllable switching element S1 can be any type of electrically controllable switching device, such as a MOSFET, a BJT, or a pair of MOSFETs arranged as a bidirectional switch. Figure 10 shows an exemplary embodiment of a variable capacitive reactance that can be placed in series with an LC resonator as shown in Figure 1A, and Figure 11 shows an exemplary embodiment of a variable inductive reactance that can be placed in parallel with an LC resonator as shown in Figure 1B.

[0023] In some cases, it is desirable to arrange multiple sets of parallel capacitors and switches or series inductors and switches in parallel. For example, all sets share the same two terminals, and all switches share the same switch control signal. This is useful, for example, when the actively variable reactance is part of a resonator composed of a conductor that is large compared to the physical size of the inductor, capacitor, and / or switch. Therefore, multiple inductors, capacitors, and / or switches can be used so that the current distribution through all of the parallel device matches the intrinsic current distribution in the large conductor.

[0024] Similar to Figure 10, Figure 11 includes a switch controller 400, a DC power input 405, a reactance control input 410, an electrically controllable switching element S1, terminals T1 and T2, and an inductor L1. The switch controller 400 includes a DC power input 420, a duty cycle control input 425, and a switch control output 430. The switch controller 400 may further include a positive RF pickup input 435 and a negative RF pickup input 440.

[0025] When the device shown in Figure 10 is placed in series with the LC resonator, the LC resonator receives an RF current I flowing from terminal T1 to terminal T2. RF This ensures that the RF current is approximately sinusoidal. A current pickup component generates a voltage proportional to this RF current. This pickup may consist of any device that generates a signal proportional to the RF current flowing through an LC resonator fitted with a transformer, a series resistor or reactance, or an actively variable reactance circuit. The switch controller 300 may include a switch controller subcircuit. In one example, the switch controller 300 may take the RF pickup sine wave as input and generate a square wave output with a variable duty cycle. The duty cycle may be controlled by a reactance control input 310. The phase of the square wave is I RF It is selected to move 90 degrees further. In other words, I RFThe zero crossing of the negative slope occurs at the midpoint of the period when the square wave is in its low state.

[0026] The square wave output drives the switching element S1. The switch turns on when the square wave is high and turns off when the square wave is low. When the apparatus shown in Figure 11 is placed in parallel with an LC resonator, the LC resonator uses the RF voltage V between terminals T1 and T2. RF This guarantees that the waveform is approximately a sine wave.

[0027] The switch controller uses the RF voltage V RF It receives the input and generates a square wave output with a variable duty cycle. Generally, the RF pickup does not necessarily have to come from two terminals T1 and T2, but can come from any device attached to an LC resonator that generates a signal proportional to the RF voltage across the LC resonator to which an active variable reactance circuit is attached. The duty cycle is controlled by the reactance-controlled input signal. The phase of the square wave is V RF It is selected to be delayed by 90 degrees. In other words, V RF The negative slope zero crossing occurs at the midpoint of the period when the square wave is in the high state. The square wave output drives the switching element S1. The switch turns on when the square wave is in the high state and turns off when the square wave is in the low state.

[0028] Figure 12 shows how the change in the duty cycle of a switch-controlled square wave changes the effective capacitive reactance between terminals T1 and T2 of the circuit shown in Figure 10. Sinusoidal current I RF This is shown in the plot above. The central plot shows a switch-controlled square wave. Note that I RF The negative slope zero crossing occurs at the midpoint of the low period of the switch-controlled square wave. The plot below shows the voltage V across capacitor C1. C1 This indicates that when switch S1 is on, V C1 It is zero. When the switch is off, the current I RF This charges capacitor C1 and increases its voltage. RFWhen is zero, the voltage reaches its peak. RF When V becomes negative, C1 V begins to decrease. C1 When it becomes zero, the switch turns on again. RF Due to the phase relationship between the switch-controlled square wave and V C1 It is guaranteed that the switch turns on when the voltage is zero. This zero-voltage switching condition is necessary to prevent the switch S1 from consuming energy by discharging C1 when it turns on.

[0029] During the time when the Switch is off, V C1 (t) is equal to a sine wave with a DC offset added. The amplitude of the sinusoidal component is I RF It becomes equal to the value obtained by multiplying the amplitude by the reactance of C1. Total waveform V C1 (t) can be decomposed into a Fourier series. An LC resonator acts as a filter and responds primarily to the Fourier component closest to its resonant frequency. Therefore, the operation of a tunable LC resonator can be analyzed by examining only the fundamental component of this Fourier series. The voltage amplitude of this fundamental component is I RF By dividing by the amplitude, the effective capacitive reactance X eff We find this as a function of the duty cycle δ.

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[0030] Effective capacitance C VAR It can be defined as follows:

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[0031] Figure 14 shows how a change in the duty cycle of a switch-controlled square wave changes the effective inductive reactance between terminals T1 and T2 of the circuit shown in Figure 11. (Sine wave voltage V) RF =V T1 -V T2 The above plot shows the result. The central plot shows the switch-controlled square wave. Note that V RF The negative slope zero crossing occurs at the midpoint of the high period of the switch-controlled square wave. The plot below shows the current I flowing through inductor L1. L1 This indicates that when switch S1 is off, I L1 It is zero. When the switch is on, the voltage V RF V increases the current through inductor L1. RF When V is zero, the current reaches its peak. RF When it becomes negative, L1 It begins to decrease. L1 When it becomes zero, the switch turns off again. RF Due to the phase relationship between the switch-controlled square wave, I L1 It is guaranteed that the switch will turn off when is zero. This zero-current switching condition is I L1 It is necessary to prevent energy from being consumed from inductive kickback by turning off switch S1 when the value is not zero.

[0032] While the switch is on, L1 (t) is equal to a sine wave with a DC offset added. The amplitude of the sinusoidal component is V RF The amplitude is equal to the value obtained by dividing it by the reactance of L1. Total waveform I L1(t) can be decomposed into a Fourier series. The LC resonator functions as a filter and mainly responds to the Fourier component closest to its resonance frequency. Therefore, the operation of a tunable LC resonator can be analyzed by simply examining the fundamental component of this Fourier series. By dividing the current amplitude of this fundamental component by the amplitude of V RF the effective inductive susceptance B eff is obtained as a function of the duty cycle δ. [Number] Here, X L1 is the reactance of inductor L1. [Number] Here, ω is the angular frequency of the RF voltage V RF . The relationship between the duty cycle and the effective inductive susceptance is plotted in Fig. 15. By changing the duty cycle δ, the effective susceptance can be continuously changed within the range of 0 to 1 / X L1 .

[0033] The effective inductive reactance is equal to the reciprocal of the effective inductive susceptance. [Number] The effective inductance L VAR can be defined as follows. [Number] Therefore, the effective variable inductance is as follows. [Number]

[0034] Note that the RF pickup sources shown in Figures 10 and 11 are merely examples of how the appropriate frequency and phase of the switch-controlled square wave can be generated. Typically, the RF pickup can be obtained from any RF source within the LC resonator. For example, the RF pickup may be generated from the RF voltage across the impedance in series with the LC resonator, the RF current through the admittance in parallel with the LC resonator, or from inductive or capacitive coupling to the LC resonator. Any of these methods can be used as long as an appropriate phase relationship is maintained between the switch-controlled square wave and the voltage and current of the LC resonator, i.e., the switch-controlled square wave leads the current through terminals T1 and T2 of the variable capacitive reactance circuit by 90 degrees, and the switch-controlled square wave lags the voltage between terminals T1 and T2 of the variable inductive reactance circuit by 90 degrees.

[0035] Switch control subcircuits in variable capacitive reactance and variable inductive reactance circuits may require DC power to function. If the variable active reactance device is part of a system with an external power supply, the switch control subcircuit may receive power from this external power supply. However, in some cases, it is desirable to power the LC resonator without connecting to an external DC power supply. For example, the LC resonator may be part of a resonant repeater that receives radio power from a radio power source and delivers it to a separate radio power receiver. In such cases, it is convenient for the switch control subcircuit to generate DC power from the RF power present in the LC resonator. This may be achieved in various ways. For example, it may be achieved by rectifying the RF voltage across the reactance in series with the LC resonator, by rectifying the voltage across the LC resonator, or by rectifying the voltage induced by inductive or capacitive coupling to the LC resonator.

[0036] Another example is that the actively variable reactance may be part of an actively powered LC resonator, but may be located on the resonator in a position where it is not possible to route a DC power cable without interrupting the tuning of the resonator. In such a case, the actively variable reactance may be powered by rectifying the RF present in the LC resonator. Alternatively, the LC resonator may be configured to support both RF and DC currents simultaneously. This may be achieved by placing an inductive choke in parallel with a capacitive junction that needs to carry the DC current, or by placing a bypass capacitor in series with the inductor of a junction that needs a DC potential difference. In this way, DC power may be supplied to the actively variable reactance through the structure of the LC resonator itself without requiring additional wiring.

[0037] The reactance control input allows for the electrical control of the resonant frequency of an LC resonator when the LC resonator incorporates either capacitive or inductive active variable reactance. This can be used as part of a feedback loop to adjust the RF current amplitude within the LC resonator when the LC resonator is placed in an external oscillating electric or magnetic field.

[0038] In some embodiments, this may be achieved as follows: Firstly, the tuning range must be selected such that the resonant frequency of the LC resonator with the active variable reactance device is always greater than or always less than the drive frequency for all duty cycles. That is, for a given drive frequency, a given duty cycle does not pass through the point where the resonant frequency of the LC resonator is equal to the drive frequency. Alternatively, the range of allowable duty cycles may be limited so that the duty cycle is only allowed to change over a range where the resonant frequency is always above or always below the drive frequency.

[0039] For a fixed drive amplitude and a fixed drive frequency, this ensures a monotonic relationship between the duty cycle and the amplitude of the RF current in the LC resonator. Second, the amplitude of the RF current in the LC resonator may be compared with a desired set value. This comparison may be achieved in various ways, such as rectifying the RF voltage across both ends of the LC resonator or the RF voltage across both ends of the impedance in series with the LC resonator and comparing the rectified DC voltage with a reference voltage. Third, in order to reduce the error between the detected RF current amplitude in the LC resonator and the desired RF current amplitude in the LC resonator, a feedback circuit may be constructed to increase or decrease the duty cycle of the active variable reactance, that is, to change the reactance control input.

[0040] As an example of such a system, consider the block diagram shown in FIG. 16. This system consists of a series LC circuit formed by L T and C T connected to two terminals T1 and T2 of an automatic RF current regulator circuit. The automatic RF current regulator may include an active variable capacitor C VAR in series with a second capacitor C2. The total series capacitance of the entire system is given by the following formula.

Equation

Equation

Equation

Equation

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[0041] Therefore, under the tuning condition X0 > 0, the RF current amplitude |I RF | is X eff It can be seen that it is a monotonic function of and is guaranteed to be a monotonic function of δ.

[0042] In the system shown in Figure 16, the rectifier generates a DC signal proportional to the RF current flowing through the LC resonator. This signal is compared to a setpoint by the RF amplitude comparison and reactance control circuit 500. The error between the RF amplitude signal and the internal setpoint is amplified and used to generate a reactance control signal sent to the active variable reactance circuit 505. Since the RF amplitude DC signal is a monotonic function of the RF current amplitude, and the RF current amplitude is a monotonic function of the reactance control signal, an overall gain coefficient may be selected for the entire feedback loop so that the entire feedback loop is negative. Typically, the sign of this gain coefficient depends on whether the slope of the monotonic function is positive or negative.

[0043] In addition to the RF amplitude output, the rectifier subcircuit 510 provides a second DC power output, which supplies DC power to the active variable reactance circuit 505 and the RF amplitude comparison and reactance control circuit 505. This DC power may be adjusted so that its voltage does not depend on the RF current amplitude of the LC resonator. The series capacitor C2 provides a useful RF voltage source proportional to the RF current amplitude of the LC resonator. Therefore, the voltage across capacitor C2 is used as the RF pickup input to the active variable reactance subcircuit 505.

[0044] An example of an actively variable reactance for automatic RF current regulation is shown in Figure 17. The circuit may be part of an LC resonator. An inductance and a capacitance are connected in series between terminals T1 and T2. This inductance and series capacitance, together with capacitors C1 and C2, form an LC resonator. An external RF power supply induces a voltage across the inductance. This circuit rectifies the RF voltage across capacitor C2 to provide a DC input to DC-DC converter IC1. IC1 provides 5VDC to the rest of the circuit.

[0045] The RF sine wave from capacitor C2 is filtered and fed to the input of comparator IC3, which generates a square wave output. The duty cycle of the square wave is determined by the voltage at the inverting input of IC3.

[0046] The square wave output of IC3 is sent to gate driver IC2, which drives the gate of MOSFET Q1. Resistors R1, R2, and R3 and capacitors C6, C7, and C8 shift the phase of the sine wave at the non-inverting input of IC3, so that the square wave at the gate of Q1 generates an RF current I RF The signal is selected to be led 90 degrees from terminal T1 to terminal T2. Note that the phase shift caused by R1, R2, R3, C6, C7, and C8 can compensate for the phase shift caused by propagation delay through IC3 and IC2.

[0047] MOSFET Q1 may be connected in parallel with capacitor C1. When MOSFET Q1 is always on, these capacitors are short-circuited, and the effective reactance of the capacitor in series with the LC resonator becomes zero. When MOSFET Q2 is always off, the entire reactance of C1 is placed in series with the LC resonator. When the duty cycle of MOSFET Q1 is intermediate between 0% and 100%, the effective series reactance of C1 is between zero and its maximum value. Note that diode D2 is connected from source to drain of MOSFET Q1. This diode is in parallel with the internal body diode of Q1 and has the same polarity. It can be selected to have a smaller forward voltage drop than the body diode of Q1, thus preventing conduction through the body diode of Q1. This is useful when the reverse recovery time of the body diode of Q1 is long compared to the RF period. In such cases, diode D2 can be selected to have a reverse recovery time much shorter than this period.

[0048] Diode D2 ensures that the drain-source voltage of Q1 is not negative than the forward voltage drop of D2. This ensures that no current flows through the body diode of Q1 when Q1 is off. Diode D2 ensures that a DC charge exists on capacitor C1. Note that if the reverse recovery time of Q1's internal body diode is sufficiently fast, diode D2 may not be necessary. In such cases, Q1's internal body diode performs the role of D2. The duty cycle of Q1 is set by a feedback loop including op-amp IC4, resistor R8, and capacitor C. 12 This filters the gate-driven square wave and generates a DC voltage proportional to the duty cycle. This voltage is supplied to the non-inverting input of IC4. The output of IC4 is R 12 and C 15 The signal passes through the low-pass filter formed by this process and is sent to the inverting input of IC3.

[0049] The voltage at the inverting input of IC3 sets the duty cycle of its output square wave. The feedback loop, including IC4, modulates the duty cycle of the square wave until the voltage at the inverting input of IC4 equals the voltage at the non-inverting input of IC4. Therefore, the duty cycle of the gate-driven square wave is equal to the DC voltage at the inverting input of IC4 divided by 5V.

[0050] The voltage at the inverting input of IC4 is set by a second feedback loop including op-amp IC5. The output of IC5 is connected to the inverting input of IC4 via resistor R9. The inverting input of IC4 is connected to capacitor C 13 It is connected to its own output via R 11 It is connected to ground via R 10 It is connected to +5V via this. The output voltage of IC5 is limited to 0V to +5V, so resistors R9 and R 10 , and R 11 This acts as a voltage divider at the inverting input of IC4, setting the upper and lower limits of the DC voltage. These limits set the upper and lower limits of the duty cycle of the gate-driven square wave, respectively. Capacitor C 13 R9, R 10 , and R 11 It also functions as a low-pass filter. IC5 compares the rectified voltage of C4 to a predetermined voltage setting and adjusts the output voltage to keep the rectified voltage of C4 constant. This works as follows:

[0051] As the output voltage of IC5 increases, the duty cycle of the gate-driven square wave increases. This causes Q1 to turn on for most of the cycle, and the effective reactance of capacitor C1 decreases. The LC resonator is tuned so that its resonant frequency is higher than the drive frequency. Therefore, decreasing the series capacitive reactance decreases its resonant frequency. This brings it closer to resonance with the drive frequency, and the amplitude of the induced RF current increases. This increases the RF voltage across capacitor C2 and the rectified voltage across capacitor C4. This voltage is then passed through the low-pass filter and R4, R5, C 10 , and C 11The voltage passes through a voltage divider formed by and . The filtered voltage is then passed through resistor R 13 It is sent to the inverting input of IC5 via this. Capacitor C 14 and resistor R 13 This forms an additional low-pass filter. As the DC voltage across capacitor C4 increases, the voltage at the inverting input of IC5 also increases, causing the output voltage of IC5 to decrease and completing a negative feedback loop. Therefore, IC5 must maintain a constant RF voltage amplitude across capacitor C2, which means that a constant RF current must be maintained circulating within the LC resonator.

[0052] The actively variable reactance can be coupled to the LC resonator in several ways. For example, if the LC resonator includes multiple capacitors or multiple inductors in series, the actively variable reactance may be placed in parallel with one of these capacitors or inductors. Alternatively, if the LC resonator includes multiple inductors or multiple capacitors in parallel, the actively variable reactance may be placed in series with one of these capacitors or inductors. Furthermore, the actively variable reactance may be inductively or capacitively coupled to the LC resonator.

[0053] The LC resonators shown in Figures 1A and 1B are the simplest reactance circuits incorporating active variable reactance. In general, active variable reactance elements can be placed in any circuit requiring variable reactance, as long as the circuit filters the voltage harmonics of the variable capacitance or the current harmonics of the variable inductance. While LC resonators naturally perform this filtering, other combinations of reactance elements with more complex arrangements can also perform this filtering.

[0054] For example, an active variable reactance may be incorporated into a T-filter, a π-filter, or other more complex filters. Such filters may be low-pass, high-pass, band-pass, or band-stop filters. The active variable reactance allows for the electrical control of the filter's cutoff frequency or frequency. Alternatively, the active variable reactance may be used to electrically control the filter's input or output impedance, thereby achieving electrically controlled impedance matching.

[0055] Another example is a resonator composed of multiple LC circuits, such as two adjacent LC resonator loops. Such a resonator may have multiple resonant modes. In a resonant radiopower system, the coupled resonator system is driven by one of its resonant modes. Active variable reactances can be coupled to one or more resonators to tune the desired resonant operating mode so that its natural frequency is equal to the frequency of the drive source. Alternatively, the tuning of one or more LC circuits may be continuously changed using active variable reactances to alter the natural mode structure of the entire coupled system. The system may be readjusted so that a particular natural mode resonates with the drive frequency while keeping the drive frequency constant.

[0056] In this way, the relative phase and amplitude of voltages and currents across the entire LC structure may be selected and set by one or more active variable reactances. Additionally, one or more fixed inductive or capacitive reactances may be included in series or parallel with the switch reactance elements in the active variable reactance circuit 505. These additional components may be used for blocking DC voltages, blocking DC currents, adding or removing additional reactances or susceptances, etc. The RF voltage across the additional series components, or the current flowing through the additional parallel components, may be used as a source for the RF pickup or rectified to provide DC power. For example, Figure 17 shows an example of an active variable capacitor using an additional capacitor placed in series with the switch capacitor C1. Capacitor C2 provides the RF voltage used for the RF pickup and is rectified to provide DC power.

[0057] Reactance control signal. In some cases, the active variable reactance circuit 505 does not need to share ground with the control system that generates the reactance control input signal. Nor does it need to share ground with the DC power. In such cases, it is desirable to provide a method for delivering the reactance control signal and / or DC power to the active variable reactance circuit 505 without directly connecting the grounds of the two circuits, which could cause an RF short circuit or the like.

[0058] In such cases, the reactance control input may be provided to the active variable reactance circuit 505 via an RF choke that allows the DC reactance control signal to pass through but blocks RF. This can be provided via a photocoupler that can be configured to transfer analog or digital signals. If digital signals are transferred, the active variable reactance circuit 505 may have additional circuitry to convert the digital signals into analog signals that can be used for reactance control. Alternatively, the reactance control signal may be communicated via optical fiber.

[0059] Another method for transferring the reactance control signal from the controller to the active variable reactance circuit 505 is to transmit the signal wirelessly. This wireless transmission can be achieved using analog or digital communication at a frequency different from the frequency of the RF power present in the LC resonator.

[0060] Alternatively, superposition of DC and RF signals may be used within the structure of the circuit in which the active variable reactance is installed. If an RF choke or bypass capacitor is used to isolate the DC voltage between terminals T1 and T2, or the DC current flowing through terminals T1 and T2, from the DC current flowing through the internal inductor L1, or the DC voltage across capacitor C1, then the DC voltage between T1 and T2, or the DC current through T1 and T2, can be controlled by an external source and used as a reactance control signal for the active variable reactance.

[0061] If an actively variable reactance also requires DC power, the coupling to and rectification of the RF power supply present in the LC resonator may be used as the DC power source. Alternatively, the DC voltage between T1 and T2 or the DC current passing through T1 and T2 can be used as the DC power. For a specific range of DC voltage and / or current, this externally applied DC voltage and / or current may be used simultaneously as both a DC power source and a source for the reactance control signal.

[0062] Tuning an LC resonator driven by an RF generator. In some implementations of resonant radio power, the power source is an LC resonator driven by a Class E RF generator that takes external DC power and converts it to RF power in the LC resonator. This Class E RF generator may be designed to maintain high DC-RF efficiency by operating the switching element (usually a MOSFET) in a zero-voltage switching state. This zero-voltage switching state may be maintained by varying the duty cycle of the switching element so that the switching element is reliably turned on when the voltage reaches zero. A feedback system may be incorporated into the Class E RF generator to adjust the duty cycle in order to maintain the RF generator in this state.

[0063] Such a system is called automatic zero-voltage switching (AZVS), and it allows a Class E amplifier to maintain high efficiency even if the LC resonator is detuned within a specific predefined range.

[0064] Figure 18 shows an example of an E-class RF generator with AZVS. RF drive frequency f d This is set by the frequency of the RF sawtooth wave input. The comparator converts the sawtooth wave into a square wave by comparing it with a DC voltage level. The square wave drives the gate of MOSFET Q1. The drain and source of MOSFET Q1 are connected in parallel with capacitor C1. Capacitor C0 and inductor L0 form the LC tank circuit 10 of the Class E RF generator. f0 indicates the resonant frequency of the LC tank circuit 10 and is defined as follows:

number

[0065] Inductor L0 is a magnetic loop antenna that couples RF power to a resonant radio receiver. DC power is supplied to the circuit via an RF choke connected to the drain of Q1. The system may also be characterized by a dimensionless constant K defined as follows:

number

[0066] The second comparator compares the drain voltage of Q1 with a reference voltage Vref close to 0V. The D latch latches the output of the comparator at the rising edge of the gate-driven square wave. Therefore, the output of the latch is when the drain voltage of Q1 is Vref at the moment Q1 is turned on. ref This indicates whether the value is higher or lower. The output of the D latch is filtered using a low-pass filter. If Q1 is turned on too early, the output of the D latch will be high, the DC output of the low-pass filter will rise slowly, and the rising edge of the gate-driven square wave will shift to the latter half of the cycle. If Q1 is turned on too late, the output of the D latch will be low, the DC output of the low-pass filter will fall slowly, and the rising edge of the gate-driven square wave will shift to the earlier part of the cycle. Therefore, in the AZVS feedback circuit, the drain voltage of Q1 is V at the moment Q1 is turned on. ref The duty cycle of the gate-driven square wave is controlled to be equal to V. ref When set very close to 0V, the circuit achieves zero-voltage switching.

[0067] It is desirable to maintain a constant RF current within the LC resonator under both various load and tuning conditions. This constantity allows the receiver to be kept in a constant ambient field regardless of the loop load. It is also desirable to prevent deviations in RF current amplitude beyond the designed operating point from causing the specific absorption rate of nearby human tissue to exceed regulatory limits.

[0068] In a specific tuning of f0, the Class E RF generator with AZVS maintains an RF current amplitude in the LC resonator that is substantially independent of the RF load, as long as the DC current remains much lower than the RF current amplitude in the LC resonator. This can be seen in Figure 19.

[0069] However, when the resonator is detuned, the ratio of RF current to DC power voltage changes. This can be seen in Figure 20.

[0070] The tuning error of the LC resonator may be detected using the duty cycle of the E-class RF generator with AZVS. As long as the E-class RF generator with AZVS is operating under light load conditions, the duty cycle is primarily a function of the LC resonator tuning and is largely independent of the load. The relationship between the resonant frequency of the LC tank circuit f010 and the AZVS duty cycle δ is plotted in Figure 14. Under light load conditions, the relationship between the resonant frequency f0 and the AZVS duty cycle δ is approximately as follows:

number

[0071] Equation 22 is given when K << 1 and I DC < RF This is an effective approximation under light load conditions defined as I DC and I RF These are the DC current and RF current of the Class E amplifier (see Figure 18).

[0072] Light load condition (i.e., I DC < RF In this case, the RF current amplitude is a function of the duty cycle δ of the AZVS.

number

[0073] ​​The difference between the actual duty cycle of an E-class RF generator with AZVS and a specific reference value can be used as an error signal in the feedback loop to control the tuning of the LC resonator. The tuning range of an LC resonator with actively variable reactance must be selected such that the RF current amplitude of the LC resonator driven by the E-class RF generator with AZVS is a monotonic function of the reactance control input.

[0074] Subsequently, the feedback loop can be modified to change the tuning of the LC resonator to make the actual duty cycle equal to the reference value. Once the duty cycle is fixed, the ratio of RF current to DC power voltage is fixed. This keeps the RF current amplitude substantially constant across a range of tuning and load conditions.

[0075] Figure 22 shows an example of how automatic tuning works. The Class E RF generator with AZVS is connected to an LC tank circuit 10, which consists of a capacitor C0 and an inductor L0 connected in series, via an actively variable capacitive reactance. In this example, the actively variable capacitive reactance obtains DC power from the same source as the Class E RF generator with AZVS. The resonant frequency f0 of the tank circuit is given by the following equation.

number

number

[0076] The feedback loop works as follows: Effective capacitance C of the actively variable capacitance reactance VAR The series capacitance C increases monotonically with increasing reactance control input. series is, C VAR It increases monotonically with increasing C. The resonant frequency f0 is C. seriesIt decreases monotonically as increases.

[0077] The duty cycle of the E-class RF generator with AZVS decreases monotonically as f0 decreases. The AZVS feedback voltage is proportional to 1. Therefore, the AZVS feedback voltage increases monotonically as f0 decreases. The output of the auto-tuning feedback subcircuit decreases monotonically as the AZVS feedback voltage increases. The output of the auto-tuning feedback subcircuit is sent to the reactance control input of the active variable capacitive reactance, completing the negative feedback loop.

[0078] The automatic tuning feedback subcircuit compares the AZVS feedback voltage with a fixed voltage VSET, and the high DC gain of the negative feedback loop operates so that the AZVS feedback voltage is equal to VSET in the equilibrium state. This circuit has two feedback loops: one inside the AZVS-equipped Class E RF generator, and the other a larger loop controlling the tuning of the LC tank circuit 10. The cutoff frequency of the low-pass filter in the AZVS feedback loop must be selected to be higher than the cutoff frequency of the low-pass filter in the automatic tuning feedback subcircuit. This ensures that the AZVS is always in a near-steady state with respect to the time scale of the automatic tuning feedback system.

[0079] The circuit shown in Figure 22 automatically adjusts the tuning of the LC tank circuit 10 to keep the duty cycle δ of the AZVS-equipped E-class RF generator constant.

[0080] The duty cycle setting is determined by the voltage, VSET. As shown in Figures 13 and 12, the AZVS duty cycle is primarily a function of the resonant frequency f0 and is substantially load-independent. Therefore, by fixing the AZVS duty cycle δ, the resonant frequency f0 of the tank circuit is also fixed.

[0081] Therefore, automatic tuning makes the system less susceptible to fluctuations in component values ​​affecting the inductance of L0 or environmental perturbations. Consequently, this system maintains a constant RF current amplitude in the LC tank circuit 10 under various tuning and load conditions.

[0082] Furthermore, it is possible to eliminate the internal feedback loop of the AZVS-equipped Class E RF generator so that MOSFET Q1 is driven at a constant duty cycle. Instead of changing the duty cycle to maintain the ZVS condition, an actively variable capacitance reactance may be used to change the tuning of the LC tank circuit 10 until the ZVS condition is met.

[0083] Figure 23 shows an example of how this is implemented. When Q1 is turned on, the drain voltage of Q1 is V REF The comparator and D latch detect whether it is above or below V. When Q1 is turned on, the drain voltage of Q1 is V REF If the value exceeds a certain threshold, the output of the D latch becomes high. This output passes through the inverter and the low-pass filter. When the output of the D latch is high, the output of the inverter becomes low, the output of the low-pass filter gradually decreases, and the reactance control input of the variable reactance decreases.

[0084] When the reactance control input decreases, the resonant frequency of the LC tank circuit 10 rises, and the drain voltage at the moment Q1 turns on decreases. This completes the feedback loop.

[0085] In the example in this section, a MOSFET was used as the switching device. However, any general switching device will work as long as its switching speed is faster than the drive frequency. The LC tank circuit 10 was also tuned using an actively variable capacitive reactance. However, any general actively variable reactance, or combination of actively variable reactances, such as an actively variable inductance or a combination of actively variable inductance and capacitance, may be used.

[0086] In general, the active variable capacitance reactance shown in Figure 22 and the variable reactance shown in Figure 23 may be any type of electrically controllable variable reactance.

[0087] Tuning of an LC resonator driven by a Class E RF generator with distributed AZVS. In some cases, it is desirable to drive a resonant magnetic loop antenna from multiple RF generators arranged in series around the loop. Figure 25 shows an example of a loop with three distributed RF generators. A parallel RF choke has been added to the series capacitor in the loop, modifying it so that DC current bypasses the capacitor. This allows DC current to flow into the loop in addition to the RF current, and makes it possible to draw DC power from the same two terminals from which the RF generator outputs RF power.

[0088] Figure 24 shows an example of how to modify an E-class RF generator to take DC power from the same two terminals used to output RF power. It is also important that all RF generators are phase-locked, and Figure 24 shows an example of how this can be achieved. The RF generator has a built-in current sense transformer that picks up a signal proportional to the RF current flowing through the magnetic loop antenna. The gate drive circuit generates a square wave that is phase-locked to this signal. Since all distributed RF generators share the same RF current in series, this phase-locking mechanism ensures that all RF generators are in phase with respect to each other.

[0089] Similar to a magnetic loop antenna driven by a single RF generator, it is sometimes desirable to use actively variable reactance to tune a magnetic loop antenna driven by multiple distributed RF generators. Figure 26 shows how this can be achieved by modifying the RF generator. Specifically, Figure 26 shows a block diagram of a single RF generator for use in a distributed RF generation system with automatic tuning. As before, the output terminal serves two purposes: RF output and DC input. The DC input power is sent to the RF power subcircuit and control DC power supply, providing a stable DC voltage to the circuit and other parts.

[0090] The RF power subcircuit incorporates switching elements such as MOSFETs and receives the gate drive signal from the duty cycle controller subcircuit. The RF power from the power FET drain voltage is passed through a pickup transformer, connected in series with an active variable reactance, and output to the output terminal.

[0091] The power FET drain voltage is sensed by the ZVS detector at the rising edge of the gate drive signal, detecting whether the power FET drain voltage is above or below a specific reference voltage at the moment the power FET is switched on. This can be achieved, for example, using a D latch and comparator, as shown in Figure 18. If the rising edge of the gate drive signal occurs while the power FET drain voltage is higher than the reference voltage, the duty cycle control output of the ZVS detector slowly increases over time.

[0092] The duty cycle control output of the ZVS detector slowly decreases over time when the rising edge of the gate drive signal occurs while the drain voltage of the power FET is below the reference voltage. The characteristic timescales of the rise and fall of the duty cycle control output are determined by the cutoff frequency of the low-pass filter in the ZVS detector subcircuit.

[0093] The duty cycle control output of the ZVS detector is sent to a duty cycle control subcircuit. This subcircuit takes an injection-locked square wave as input and outputs a gate-driven square wave with a duty cycle that monotonically decreases as the duty cycle control signal increases. Note that the sign of the logic or the analog control signal may be inverted without changing the operation of the entire circuit, as long as the signs of all inputs and outputs of the subcircuit to which the control signal is connected are consistently inverted.

[0094] The RF power subcircuit, ZVS detection subcircuit, and duty cycle controller subcircuit form a closed feedback loop, which automatically adjusts the duty cycle of the gate drive square wave so that the RF power subcircuit always operates in a zero-voltage switching state.

[0095] An injection-locked square wave is output from an injection-locked oscillator subcircuit, which receives a sine wave pickup input from the secondary side of the pickup transformer. The injection-locked oscillator subcircuit generates a square wave with a specific frequency tolerance added to or subtracted from the system's operating frequency. If a signal is present at the pickup input, the injection-locked oscillator locks its frequency to the pickup input frequency. There is a small phase difference between the phase of the injection-locked square wave and the phase of the pickup input, which is proportional to the frequency difference between the signal present at the pickup input and the intrinsic frequency of the injection-locked oscillator.

[0096] The primary side of the pickup transformer is in series with the magnetic loop antenna on which the RF generators are located. Therefore, the pickup signal is proportional to the RF current circulating through the magnetic loop antenna, which is shared by all the RF generators connected in series. By locking the frequency of the injection-locked square wave to the frequency of the RF current circulating through the magnetic loop antenna, all RF generators are frequency-locked relative to each other. Each RF generator has a small phase difference with respect to the average phase of all the distributed RF generators due to the difference between the natural frequency of its injection-locked oscillator and the average natural frequency of all the injection-locked oscillators.

[0097] The phase difference between RF generators can be reduced by tightening the tolerance of the intrinsic frequency of the injection-locked oscillator. When the AZVS feedback system reaches a steady state, each RF generator has a duty cycle that depends on the overall tuning of the magnetic loop antenna. Figure 21 shows an example of this relationship for a single Class E RF generator. Thus, the duty cycle control signal of each RF generator circuit can be used as a local measure of the detuning of the entire loop, and the difference between the duty cycle control signal and the desired setpoint can be used as an error signal to control the actively variable reactance that changes the tuning of the magnetic loop antenna.

[0098] A magnetic loop antenna has a tuning range Δf0 / f that is independent of the loop size and is a specific fraction. d Often tuning is required, and f0 is the range of resonant frequencies in which the loop needs to be tuned, f d This is the drive frequency of the loop. The resonant frequencies of the magnetic loop antenna are as follows:

number

number

[0099] Therefore, it is desirable to add an adjustable active variable reactance in series with the loop to correct these errors. However, the magnitude of this correction factor increases with the loop size. Consequently, retuning the loop using a single active variable reactance may be impractical when the loop size is large.

[0100] A natural solution to this problem is to add an active variable reactance to every RF generator, as shown in the block diagram in Figure 26. As the loop size increases, the number of distributed RF generators increases, and therefore the number of active variable reactances connected in series to the loop also increases. Thus, the sum of the series reactances of all active variable reactances increases with the loop size, making it possible to achieve a nearly constant fractional tuning range for loops of various sizes.

[0101] The duty cycle control signal can be used in each RF generator circuit as a local measure of the overall loop detuning, and may therefore be used to control the active variable reactance subcircuit within each RF generator. The auto-tuning control subcircuit forms a feedback loop that outputs a reactance control signal proportional to the difference between the duty cycle control signal and a predefined duty cycle setpoint. The gain of this feedback loop can be selected to exhibit negative feedback. A low-pass filter must be added to this feedback loop within the auto-tuning control subcircuit so that the AZVS feedback loop has a time to stabilize to a steady state on a much faster timescale than the auto-tuning feedback loop.

[0102] As mentioned above, there is variation among all RF generators, resulting in small phase shifts between them; therefore, the automatic tuning control subcircuit should not have infinite gain in DC. Consequently, when tuning the magnetic loop antenna so that the average duty cycle of all RF generators equals the desired duty cycle setpoint, each RF generator will have a small phase shift, and its duty cycle will differ slightly from the setpoint. If this difference is amplified with infinite gain by the automatic tuning control subcircuit, the active variable reactance subcircuit will saturate at its highest or lowest reactance. Therefore, the magnitude of the gain of the automatic tuning control subcircuit must be selected such that the product of the magnitude of the duty cycle error and the gain factor provides a reactance-controlled output signal within the range of values ​​to which the active variable reactance subcircuit responds.

[0103] Phase errors between RF generator circuits can be corrected using a third feedback loop designed to operate on a slower timescale than both the auto-tuning feedback loop and the AZVS feedback loop. When the auto-tuning feedback loop is in a steady state, the remaining difference between the duty cycle control signal and the desired setpoint generates an error signal proportional to the phase error of its RF generator, and proportional to the error of its natural frequency. This error may be corrected using a phase control signal input to an injection-locked oscillator subcircuit. This phase control signal is used to adjust the natural frequency and / or phase of the injection-locked oscillator subcircuit and can be configured to generate negative feedback such that the difference between the duty cycle control signal and the desired setpoint is driven toward zero.

[0104] This third feedback loop is optional and may not be necessary if the natural frequency error of the injection-locked oscillator is sufficiently small. However, it is useful to make the system more tolerant of the injection-locked oscillator's frequency error, which allows for the use of less expensive components with larger tolerances.

[0105] Although this disclosure has described a limited number of embodiments, those skilled in the art who are interested in this disclosure will understand that other embodiments can be devised that do not deviate from the scope of this disclosure as disclosed herein. Accordingly, the scope of this disclosure should be limited only by the appended claims.

[0106] The features, structures, or properties described above can be combined in any preferred manner in one or more embodiments, and the features discussed in the various embodiments are, where possible, interchangeable. The following description provides numerous specific details to fully understand the embodiments of the disclosure. However, those skilled in the art will understand that the technical solutions of the disclosure may be implemented without one or more of the specific details, or that other methods, components, materials, etc., may be employed. In other examples, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the disclosure.

[0107] In this specification, relative terms such as “up,” “down,” and “up,” “down” are used to describe the relative relationship between one component and another, but these terms are used only for convenience, for example, as directions in the examples shown in the drawings. It should be understood that when the device is inverted, the “up” component described above becomes the “down” component. When one structure is “on” another structure, it may be that the structure is integrally formed on the other structure, that the structure is “directly” placed on the other structure, or that the structure is “indirectly” placed on the other structure via the other structure.

[0108] In this specification, terms such as “a,” “an,” “the,” and “said” are used to indicate the presence of one or more elements or components. The terms “comprise,” “include,” “have,” “contain,” and their variations are used open-ended and, unless otherwise specified in the appended claims, mean to include additional elements, components, etc., in addition to those described.

[0109] In this specification or in the claims, terms such as “first,” “second,” “third,” etc., are used solely as labels and not as a limitation on the number of subjects unless otherwise specified. Where multiple components are shown, it is understood that, to the extent applicable, the components may be referred to in the claims as the “first” component, the “second” component, the “third” component, etc.

[0110] The embodiments described above in this disclosure are merely possible implementations set up to clearly illustrate the principles of this disclosure. Many variations and modifications can be made to the embodiments described above without substantially departing from the spirit and principles of this disclosure. All such modifications and variations are intended to be included within the scope of this disclosure and protected by the following claims.

[0111] (1) A system comprising an active variable reactance circuit configured to control the resonant frequency of at least one resonant circuit, wherein the active variable reactance circuit includes an electrically controllable switching element, a passive reactant connected to at least one terminal of the electrically controllable switching element, and a switch controller subcircuit configured to switch the electrically controllable switching element at the frequency of a radio frequency (RF) current or voltage passing through or across the device.

[0112] (2) The system according to (1), wherein the electrically controllable switching element is one of a metal-oxide-semiconductor field-effect transistor (MOSFET), a bipolar junction transistor (BJT), and a pair of MOSFETs arranged as a bidirectional switch.

[0113] (3) The system according to (1) or (2), wherein the passive reaction element provides a variable capacitive reactance in series with at least one resonant circuit, or the passive reaction element provides a variable inductive reactance in parallel with at least one resonant circuit.

[0114] (4) The system according to any one of (1) to (3), further comprising a current pickup device configured to generate a voltage proportional to the RF current and to supply the voltage to a switch controller subcircuit.

[0115] (5) The current pickup device is one of a transformer, a series resistor, and a series reactance device, as described in any one of items (1) to (4).

[0116] (6) The system according to any one of (1) to (5), wherein the switch controller subcircuit is configured to receive a sinusoidal RF pickup signal from a current pickup device as input, generate a square wave output signal with a variable duty cycle, and the square wave output signal drives an electrically controllable switching element.

[0117] (7) A system according to any one of (1) to (6), in which the variable duty cycle is controlled by a reactance control input signal supplied to a switch controller subcircuit.

[0118] (8) The phase of the square wave output signal is a 90-degree lead over the RF current, as described in any one of the systems in (1) to (7).

[0119] (9) The system according to any one of (1) to (8), wherein at least one resonant circuit is part of a resonant repeater that receives radio power from a radio power source and supplies power to a separate radio power receiver.

[0120] (10) The system according to any one of (1) to (9), further comprising an automatic RF current regulator circuit, the automatic RF current regulator circuit being configured to compare the DC signal generated by at least one resonant circuit with an internal setpoint using an RF amplitude comparison and reactance control circuit that identifies a DC signal proportional to the RF current flowing through at least one resonant circuit, and amplifies the error between the RF amplitude signal and the internal setpoint, and transmits the amplified error as a reactance control signal to an active variable reactance subcircuit.

[0121] (11) The system according to any one of (1) to (10), further comprising at least one resonant circuit.

[0122] (12) The system described in any one of (1) to (11), wherein the electrically controllable switching element is a single electrically controllable switching element of an active variable reactance circuit.

[0123] (13) A system comprising an active variable reactance circuit configured to control the resonant frequency of at least one resonant circuit, the active variable reactance circuit comprising an electrically controllable switching element in parallel with at least one capacitor, and a switch controller subcircuit configured to switch the electrically controllable switching element at the frequency of a high-frequency (RF) current or voltage passing through or across the device, such that the RF current flowing from a first terminal to a second terminal is substantially sinusoidal.

[0124] (14) A system comprising an active variable reactance circuit configured to control the resonant frequency of at least one resonant circuit, the active variable reactance circuit comprising an electrically controllable switching element connected in series with at least one inductor, and a switch controller subcircuit configured to switch the electrically controllable switching element at the frequency of a high-frequency (RF) current or voltage passing through or across the device such that the RF voltage between a first terminal and a second terminal is substantially sinusoidal.

[0125] (15) A method comprising: controlling the resonant frequency of at least one resonant circuit by an active variable reactance circuit; providing an electrically controllable switching element; providing a passive reactant connected to at least one terminal of the electrically controllable switching element; and switching the electrically controllable switching element by a switch controller subcircuit at the frequency of a radio frequency (RF) current or voltage passing through or across the device.

[0126] (16) The system according to claim 15, wherein the electrically controllable switching element is one of a metal-oxide-semiconductor field-effect transistor (MOSFET), a bipolar junction transistor (BJT), and a pair of MOSFETs arranged as a bidirectional switch.

[0127] (17) The method according to (15) or (16), wherein the switch controller provides a variable capacitive reactance arranged in series with the resonant circuit, or the switch controller provides a variable inductive reactance arranged in parallel with the resonant circuit.

[0128] (18) The method according to any one of (15) to (17), further comprising the steps of generating a voltage proportional to an RF current using a current pickup device, and providing the voltage to a switch controller subcircuit.

[0129] (19) The method according to any one of items (15) to (18), wherein the current pickup device is one of a transformer, a series resistor, and a series reactance device.

[0130] (20) The method according to any one of (15) to (19), further comprising the steps of: (20) receiving an RF pickup signal in the form of a sinusoidal wave from a current pickup device as input by a switch controller subcircuit; and (20) generating a square wave output signal having a variable duty cycle by the switch controller subcircuit, wherein the square wave output signal drives an electrically controllable switching element.

[0131] (21) The method according to any one of (15) to (20), wherein the variable duty cycle is controlled by a reactance control input signal supplied to a switch controller subcircuit.

[0132] (22) The phase of the square wave output signal is determined by leading the RF current by 90 degrees, as described in any one of (15) to (21).

[0133] (23) The method according to any one of (15) to (22), wherein at least one resonant circuit is part of a resonant repeater that receives radio power from a radio power source and powers a separate radio power receiver.

[0134] (24) The method according to any one of (15) to (23), further comprising the step of using an automatic RF current regulator circuit, the step of using an automatic RF current regulator circuit comprising: identifying a DC signal generated by at least one resonant circuit that is proportional to an RF current flowing through at least one resonant circuit; comparing the DC signal to an internal setpoint using an RF amplitude comparison and reactance control circuit; amplifying the error between the RF amplitude signal and the internal setpoint; and transmitting the amplified error as a reactance control signal to an active variable reactance subcircuit. [Prior art documents] [Patent Documents]

[0135] [Patent Document 1] U.S. Provisional Patent Application No. 63 / 071,048

Claims

1. Equipped with an active variable reactance circuit, The aforementioned active variable reactance circuit is, Electrically controllable switching element, A passive reaction element connected to at least one terminal of the electrically controllable switching element, A resonator connected to at least one terminal of the electrically controllable switching element; a high-frequency (RF) pickup component configured to generate a pickup voltage proportional to the high-frequency (RF) current or voltage passing through the resonator or across the resonator; A switch controller subcircuit that receives the pickup voltage and control voltage and generates a switch control signal having a phase based on the pickup voltage and a duty cycle based on the control voltage, which switches the electrically controllable switching element by the frequency of the high-frequency (RF) current or voltage passing through the resonator or across the resonator; Includes, Here, the switch controller subcircuit comprises a comparator and an operational amplifier feedback loop, where the comparator receives the pickup voltage as the first input and the comparison voltage as the second input, and the comparison voltage becomes the output of the operational amplifier feedback loop, and Here, the operational amplifier feedback loop receives the switch control signal as the first input and the control voltage as the second input, and outputs the comparison voltage to the comparator. system.

2. The electrically controllable switching element is one of the following: a metal-oxide-semiconductor field-effect transistor (MOSFET), a bipolar junction transistor (BJT), and a pair of MOSFETs arranged as a bidirectional switch. The system according to claim 1.

3. The passive reaction element provides a variable capacitance reactance in series with the resonator, or The passive reaction element provides a variable inductive reactance in parallel with the resonator. The system according to claim 1.

4. Here, the operational amplifier feedback loop is First passive filter network; Second passive filter network; An operational amplifier having a first input connected to the switch control voltage via the first passive filter network, and a second input connected to the control voltage and the operational amplifier output via the second passive filter network; and The system includes a third passive filter network that filters the output of the operational amplifier and generates the comparison voltage as the input to the comparator. The system according to claim 1.

5. The aforementioned high-frequency (RF) pickup component is one of the following: a transformer, a series resistor, and a series reactance device. The system according to claim 4.

6. The switch controller subcircuit is configured to receive a sinusoidal RF pickup signal from the high-frequency (RF) pickup component as input, generate a square wave output signal with a variable duty cycle, and drive the electrically controllable switching element. The system according to claim 4.

7. The variable duty cycle is controlled by a reactance control input signal supplied to the switch controller subcircuit. The system according to claim 6.

8. The phase of the aforementioned rectangular wave output signal is such that it leads the RF current by 90 degrees. The system according to claim 6 or claim 7.

9. The aforementioned resonator is part of a resonant repeater that receives wireless power from a wireless power source and supplies power to a separate wireless power receiver. The system according to claim 1.

10. It further includes an automatic RF current regulator circuit, The aforementioned automatic RF current regulator circuit is Identifying a DC signal generated by the resonator, which is proportional to the RF current flowing through the resonator. The DC signal is compared to an internal setpoint using RF amplitude comparison and reactance control circuits, and The error between the RF amplitude signal and the internal set value is amplified, and the amplified error is transmitted to the active variable reactance subcircuit as a reactance control signal. It is structured in such a way. The system according to claim 1.

11. The electrically controllable switching element is a single electrically controllable switching element of the active variable reactance circuit. The system according to claim 1.

12. The steps include controlling the resonant frequency of at least one resonant circuit using an active variable reactance circuit, The steps include providing an electrically controllable switching element, The steps include providing a passive reaction element connected to at least one terminal of the electrically controllable switching element, and a resonator connected to at least one terminal of the electrically controllable switching element, A step that generates a pickup voltage proportional to the high-frequency (RF) current or voltage passing through the resonator or across the resonator, using a high-frequency (RF) pickup component, The switch controller subcircuit includes the steps of: receiving the pickup voltage and the control voltage; and generating a switch control signal having a phase based on the pickup voltage and a duty cycle based on the control voltage, which switches the electrically controllable switching element at the frequency of the high-frequency (RF) current or voltage passing through the resonator or across the resonator; Equipped with, Here, the switch controller subcircuit includes a comparator and an operational amplifier feedback loop. Here, the comparator receives the pickup voltage as the first input and the comparison voltage as the second input, and the comparison voltage becomes the output of the operational amplifier feedback loop, and Here, the op-amp feedback loop receives the switch control signal as the first input and the control voltage as the second input, and A method for outputting the comparison voltage to the comparator via the operational amplifier feedback loop.

13. The electrically controllable switching element is one of the following: a metal-oxide-semiconductor field-effect transistor (MOSFET), a bipolar junction transistor (BJT), and a pair of MOSFETs arranged as a bidirectional switch. The method according to claim 12.

14. The switch controller subcircuit provides a variable capacitive reactance in series with the resonant circuit, or The switch controller subcircuit provides a variable inductive reactance in parallel with the resonant circuit. The method according to claim 12.

15. Here, the operational amplifier feedback loop is First passive filter network; Second passive filter network; An operational amplifier having a first input connected to the switch control voltage via the first passive filter network, and a second input connected to the control voltage and the operational amplifier output via the second passive filter network; and The system includes a third passive filter network that filters the output of the operational amplifier and generates the comparison voltage as the input to the comparator. The method according to claim 12.

16. The aforementioned high-frequency (RF) pickup component is one of the following: a transformer, a series resistor, and a series reactance device. The method according to claim 15.

17. The switch controller subcircuit receives a sinusoidal RF pickup signal from the high-frequency (RF) pickup component as input. The steps include: generating a square wave output signal having a variable duty cycle using the switch controller subcircuit; Furthermore, The square wave output signal drives the electrically controllable switching element. Furthermore, The method according to claim 15.

18. The variable duty cycle is controlled by a reactance control input signal supplied to the switch controller subcircuit. The method according to claim 17.

19. The phase of the aforementioned rectangular wave output signal is such that it leads the RF current by 90 degrees. The method according to claim 17.

20. The at least one resonant circuit is part of a resonant repeater that receives radio power from a radio power source and supplies power to a separate radio power receiver. The method according to claim 12.

21. The step further includes using an automatic RF current regulator circuit, The step of using the automatic RF current regulator circuit is: A step of identifying a DC signal generated by the at least one resonant circuit, which is proportional to the RF current flowing through the at least one resonant circuit, The steps include comparing the DC signal to an internal set value using an RF amplitude comparison and reactance control circuit, A step of amplifying the error between the RF amplitude signal and an internal setpoint, The steps include transmitting the amplified error as a reactance control signal to an active variable reactance subcircuit, including, The method according to claim 12.

22. The operational amplifier feedback loop is a first operational amplifier feedback loop; The operational amplifier in the first operational amplifier feedback loop is the first operational amplifier; and The switch controller subcircuit further includes a second operational amplifier feedback loop comprising a second operational amplifier, which receives a reference voltage as a first input and a second voltage as a second input, and outputs the control voltage, where the second voltage is a direct current (DC) voltage proportional to the high frequency (RF) current or voltage. The method according to claim 16.

23. The operational amplifier feedback loop is a first operational amplifier feedback loop; The operational amplifier in the first operational amplifier feedback loop is the first operational amplifier; and The switch controller subcircuit further includes a second operational amplifier feedback loop comprising a second operational amplifier, which receives a reference voltage as a first input and a second voltage as a second input, and outputs the control voltage, where the second voltage is a direct current (DC) voltage proportional to the high frequency (RF) current or voltage. The system according to claim 4.

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