receiver

Abstract

To provide a receiver.SOLUTION: A receiver according to the present invention includes: an I branch including an I branch mixer that down-converts an RF (radio frequency) signal and outputs an I component and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that down-converts the RF signal and outputs a Q component and a Q branch channelization filter that separates the baseband signal from the output signal of the Q branch mixer; a frequency-dependent mismatch estimator that calculates frequency-dependent mismatch between the I branch and the Q branch; and a frequency-dependent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I branch and the Q branch on the basis of the calculation result of the frequency-dependent mismatch estimator.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] This technology generally relates to receivers. [Background technology]

[0002] A direct conversion receiver uses quadrature mixing to bring a radio frequency (RF) signal down to the baseband (BB) in one step. Quadrature mixing uses two mixers to multiply the received signal by two local oscillator (LO) signals with a 90-degree phase difference. The I and Q signals, separated into baseband I and Q branches, are then processed by an ADC after passing through a low-pass filter (LPF) and an amplifier, and are converted into discrete-time signal samples. [Overview of the project] [Problems that the invention aims to solve]

[0003] An ideal receiver operates under the assumption of three conditions: a 90-degree phase difference between two LO signals, identical gain, and identical frequency characteristics for the LPF, amplifier, and DAC of the two I / Q branches. Actual semiconductor manufacturing processes introduce element-level mismatches unrelated to the design, which prevents these three conditions from being met, resulting in LO phase / gain mismatch and baseband filter mismatch.

[0004] The receiver mismatch described above has both frequency-independent and frequency-dependent characteristics. Such IQ mismatch in a receiver acts by adding a conjugate image to the original signal being received, degrading the signal-to-noise ratio (SNR) of the received signal and increasing the bit error rate (BER), similar to an interferer.

[0005] One of the problems that this technology aims to solve is to provide a technology that can compensate for the two inconsistencies mentioned above, which are shortcomings of the conventional technology. [Means for solving the problem]

[0006] This embodiment is a receiver, the receiver comprising: an I-branch including an I-branch mixer that downconverts an RF signal (radio frequency signal) to output an I component and an I-branch channeling filter that separates a baseband signal from the output signal of the I-branch mixer; a Q-branch including a Q-branch mixer that downconverts the RF signal to output a Q component and a Q-branch channeling filter that separates a baseband signal from the output signal of the Q-branch mixer; a frequency-dependency mismatch estimation unit that calculates the frequency-dependency mismatch between the I-branch and the Q-branch; and a frequency-dependency mismatch compensation unit that compensates for the frequency-dependency mismatch between the I-branch and the Q-branch based on the calculation results of the frequency-dependency mismatch estimation unit.

[0007] In one embodiment of this present invention, the frequency-dependent mismatch compensation unit further includes a compensation filter having a transfer function in the I path that corresponds to the result of dividing the transfer function of the Q branch channel filter by the transfer function of the I branch channel filter.

[0008] In one embodiment of this design, the frequency-dependent mismatch compensation unit further includes a delay line in the Q path corresponding to the delay of the compensation filter.

[0009] In one embodiment of this product, the frequency-dependent mismatch estimation unit determines the transfer function of a compensation filter that compensates for frequency-dependent mismatch from a value obtained by normalizing the time average of the tone-image correlation of the signal output from the channeling filter with the average power of the signal output from the channeling filter.

[0010] In one embodiment of this design, when the value is m(k), the frequency-dependent mismatch estimation unit determines:

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[0011] In one embodiment of this product, the frequency-dependent mismatch estimation unit obtains a value by normalizing the time average of the tone-image correlation using a pilot signal including SSB multitones with the average power of the signal output from the channelization filter.

[0012] This embodiment is a receiver, the receiver comprising: an I-branch mixer that downconverts an RF signal (radio frequency signal) to output an I component, and an I-branch channeling filter that separates a baseband signal from the output signal of the I-branch mixer; a Q-branch mixer that downconverts the RF signal to output a Q component, and a Q-branch channeling filter that separates a baseband signal from the output signal of the Q-branch mixer; a frequency-independent mismatch estimation unit that calculates the frequency-independent mismatch between the I-branch and the Q-branch; and a frequency-independent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I-branch and the Q-branch based on the calculation results of the frequency-independent mismatch estimation unit.

[0013] In one embodiment of this design, the independent mismatch estimation unit calculates the gain mismatch and phase mismatch of the I-branch mixer and the Q-branch mixer from the time average of the power of the output signals of the I-branch mixer and the Q-branch mixer, using the normalized value of the time average of the squared output signals of the I-branch mixer and the Q-branch mixer.

[0014] In one embodiment of this present invention, the dependency mismatch estimation unit is:

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[0015] In one embodiment of this design, the frequency-independent mismatch compensation unit includes a first multiplier connected to the I branch, a second multiplier connected to the Q branch, a third multiplier that amplifies the output signal of the first multiplier, and an adder that sums the outputs of the second multiplier and the third multiplier.

[0016] In one embodiment of this design, the gain of the first multiplier is gRX corresponding to, the gain of the second multiplier is cos(1 / θ RX ) corresponding to, the gain of the third multiplier is tan(θ RX ).

Advantages of the Invention

[0017] According to the present embodiment, there is provided an advantage that frequency-dependent and frequency-independent mismatches of a receiver can be compensated for.

Brief Description of the Drawings

[0018] [Figure 1] It is a diagram showing an outline of a practical receiver with mismatches. [Figure 2] It is a diagram showing an outline of a frequency-dependent mismatch compensation unit. [Figure 3] It is a diagram showing an outline of a frequency-independent mismatch compensation unit.

Modes for Carrying Out the Invention

[0019] Hereinafter, the present embodiment will be described with reference to the accompanying drawings. FIG. 1 is a diagram showing an outline of a practical receiver 10 with mismatches. Referring to FIG. 1, a receiver 10 according to the present embodiment includes an I branch 100 including an I branch mixer 110 that down-converts an RF signal (radio frequency signal, r(t)) to output an I component, and a Q branch 200 including a Q branch mixer 210 that down-converts the RF signal r(t) to output a Q component.

[0020] In one embodiment, the I branch 100 may further include an I branch channelizing filter 120 that separates a baseband signal from an output signal of the I branch mixer 110, and the Q branch 200 may further include a Q branch channelizing filter 220 that separates a baseband signal from an output signal of the Q branch mixer 210.

[0021] The receiver 10 may include a frequency-dependent mismatch estimation unit (FD est) 310 that calculates the frequency-dependent mismatch between the I branch 100 and the Q branch 200, and a frequency-dependent mismatch compensation unit 320 that compensates for the frequency-dependent mismatch between the I branch 100 and the Q branch 200 based on the calculation results of the frequency-dependent mismatch estimation unit 310.

[0022] Furthermore, in one embodiment of the receiver 10, the receiver 10 may include a frequency-independent mismatch estimation unit 410 that calculates the frequency-independent mismatch between the I branch 100 and the Q branch 200, and a frequency-independent mismatch compensation unit 420 that compensates for the frequency-independent mismatch between the I branch 100 and the Q branch 200 based on the calculation results of the frequency-independent mismatch estimation unit 410.

[0023] In the illustrated embodiment, the receiver 10 is exemplified as including a frequency-dependent mismatch estimation unit 310 and a frequency-dependent mismatch compensation unit 320, and a frequency-independent mismatch estimation unit 410 and a frequency-independent mismatch compensation unit 420. However, an embodiment of the receiver not shown may include only one of either a frequency-dependent mismatch estimation unit and a frequency-dependent mismatch compensation unit, or a frequency-independent mismatch estimation unit and a frequency-independent mismatch compensation unit.

[0024] Referring to Figure 1, the received signal r(t) is branched and input to I branch 100 and Q branch 200. The signal r(t) is downconverted to the signal cosω by the mixer 110 of I branch 100. RX It is mixed with t and down-converted. Signal r(t) is mixed with the down-converted signal and down-converted in mixer 210 of Q branch 200.

[0025] The signal x was downconverted by mixer 110 on branch I. I (t) is input to the I-branch channelization filter 120, and the I-branch channelization filter 120 processes the baseband signal y I(t) is output. The time-domain impulse response of the I-branch channeled filter 120 is h I RX (t) can be displayed. The signal x is down-converted by mixer 210 in the Q branch. Q (t) is input to the Q branch channelization filter 220. The Q branch channelization filter 220 is 、 Baseband signal y Q Output (t). The time-domain impulse response of the Q-branch channeled filter 220 is h Q RX It can be displayed as (t).

[0026] In one embodiment, the I-branch channeling filter 120 and the Q-branch channeling filter 220 may be low-pass filters (LPFs). In one embodiment, the outputs of the I-branch channeling filter 120 and the Q-branch channeling filter 220 may be supplied to an analog-to-digital converter (ADC). In one embodiment, the I-branch 100 may further include a delay line 123 that is delayed by a corresponding delay time to match the delay of the Q-branch 200. The outputs of the I-branch channeling filter 120 and the Q-branch channeling filter 220 may be supplied to an analog-to-digital converter (ADC).

[0027] Ideally, the signals supplied to each mixer by the local oscillator (not shown) for downconversion should have the same amplitude. Furthermore, ideally, the signals supplied to mixer 110 on branch I and mixer 210 on branch Q should have a 90-degree phase difference.

[0028] However, there are differences in the magnitude of the signals provided by the actual local oscillator (LO), and the phase difference of the provided signals does not exactly amount to 90 degrees. In such cases, the signal magnitude mismatch is called a gain mismatch g RX Displayed as θ, the phase mismatch is θ RXThis is displayed as follows. However, the gain and phase mismatch of the mixer during down-conversion cannot be distinguished from the LO mismatch and does not need to be displayed separately. As will be discussed later, gain mismatch and phase mismatch are independent of time (and frequency).

[0029] L is a mismatched signal provided to the mixer. RX (t) is expressed by the following formula:

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[0030] The output signal x(t), which has been down-converted to an LO signal containing mismatches, is expressed by the following formula:

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[0031] Furthermore, when equation 4 is summarized in the frequency domain, it becomes equation 5 below.

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[0032] In the ideal case where there are no effects from mismatch, g RX θ is 1, RX Since is 0, consequently the β value is 0, and the transmission frequency ω TX and receiving frequency ω RX Since they are the same, the baseband signal r BB(t) can be obtained. However, when gain mismatch and phase mismatch occur, the β value is not 0, and due to IQ mismatch, the unintended r BB (t) is the conjugate image of the signal r BB * It can be confirmed that component (t) is further generated.

[0033] The conjugate image generated in this way is r BB * The (t) component acts as an interference signal to the target signal, degrading the signal-to-noise ratio and receiving performance of the receiver. Also, as illustrated in Equation 5, from the perspective of the frequency domain, the target signal r BB The coefficients α and β of (t) and its conjugate image are constants, not functions of frequency, and in this sense, this can be called a frequency-independent IQ mismatch.

[0034] Filter h of baseband I branch 100 and Q branch 200 RX I (t), h RX Q The mismatch in (t) includes the difference in poles and zeros due to the circuit elements in the semiconductor process that implement the anti-aliased low-pass filter in the analog baseband, and the difference in propagation delay up to just before the ADC sample and hold.

[0035] When there is an IQ mismatch in the baseband filter, the input and output of the filter stage satisfy equation 6 below.

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[0036] (x i (t): Output signal of branch mixer, x Q (t): Output signal of Q branch mixer, h i RX (t): I branch channeled filter impulse response, h Q RX (t): Impulse response of Q-branch channelized filter,

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[0037] In equation 6, the target signal x(t) is not transmitted as is, and the conjugate image x * It can be seen that (t) is added. Looking at this from the frequency domain, the coefficients H of the target signal and its conjugate image are e RX (f), H o RX (f) is a function of frequency, and the interference from the image is also a function of frequency. From this perspective, this can be called a frequency-dependent IQ mismatch.

[0038] When all IQ mismatches of the LO, mixer, and baseband branch are determined, the demodulated signal y(t) can be expressed by equation 7 below. Due to the mismatch between the I branch 100 and the Q branch 200 at the receiving end 10, a complex conjugate term is generated in the output signal as shown in equation 6, and from this, an image is formed in the frequency domain.

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[0039] In equation 7, H I RX If we take (f) as a common factor, we get H D RX (f) = H Q RX (f) / H I RX (f) is expressed as the ratio of the transfer function of the Q-branch channeling filter to the transfer function of the I-branch channeling filter. Therefore, the Q-branch channeling filter 220 is a filter (h) that shows the change in the I-branch channeling filter 120 and the Q-branch channeling filter 222 relative to the I-branch channeling filter 120 in the time domain, as shown in Figure 1. D RX (t))223 can be expressed as a cascaded connection. Equation 7 above can be summarized as follows:

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[0040] H I RX (f) is common to both the I branch and the Q branch, and H is only in the Q branch. D RX (f) = H Q RX (f) / H I RX A system model can be considered that includes a frequency-independent mismatch compensation unit 410, which includes a frequency-dependent mismatch compensation filter 322 having a transfer function of (f). This can be considered in the discrete time domain h D RX (n) can be used to represent this as shown in Figure 2. D RX (t) can be summarized as shown in equation 9 below, and from this we can express the mismatch between the baseband filters of the IQ branch.

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[0041] The mismatch between the IQ branch baseband filters is due to the H of the Q branch. D RX (f) is expressed as, and to compensate for this, H is added to the I branch. D RX (f) Add (f). To achieve this, after AD sampling, h to the I branch. D RX (t) Discrete-time version of h D RX (n) is used to compensate. h D RX (n) is an IIR (Infinite Impulse Response), so for actual implementation, we apply the FIR (Finite Impulse Response) approximation. In this case, the Q branch includes a tapped-delay line 325, h D RXBalance with the integer group delay D of (n). From this, the receive frequency-dependent IQ mismatch can be compensated.

[0042] The conversion formulas of the complex envelopes of the gain mismatch and phase mismatch derived in Equations 4 and 5 can be obtained as shown in Equation 10 below, and from this, the conversion formula in the matrix form of the IQ vector can be obtained as shown in Equation 11. (z(t)=r BB (t)) [Number]

[0043] Summarize Equation 10 into a determinant and find its inverse matrix, which is as shown in Equation 11 above. From this, the gain mismatch and phase mismatch of the receive-side LO can be compensated.

[0044] FIG. 3 is a diagram showing an overview of the frequency-independent mismatch compensation unit 420. Referring to FIG. 3, the frequency-independent mismatch compensation unit 420 includes a first multiplier 421 connected to the I branch, a second multiplier 422 connected to the Q branch, a third multiplier 423 that multiplies the output signal of the first multiplier, and an adder 424 that adds the output of the second multiplier 422 and the output of the third multiplier 423.

[0045] The frequency-independent mismatch compensation unit 420 shown in FIG. 3 implements Equation 11. The gain of the first multiplier 421 included in the frequency-independent mismatch compensation unit 420 corresponds to g RX . The gain of the second multiplier 422 corresponds to cos(1 / θ RX ). As one embodiment, cos(1 / θ RX ) can be approximated to 1+(θ RX 2 ) / 2 using a Taylor series and used as the gain of the second multiplier 422.

[0046] The gain of the third multiplier corresponds to tan(θ RX ). As one embodiment, tan(θ RX) using a Taylor series θ RX +(θ RX 2 This can be approximated as ) / 3 and used as the gain of the third multiplier 423. In this way, the frequency-independent mismatch is compensated, and the received signal r with the mismatch compensated is obtained. I and r Q You can obtain it.

[0047] To compensate for frequency-independent IQ mismatch, g, which is the IQ gain mismatch between the LO and mixer circuits, is used. RX θ, which is an IQ phase mismatch RX The value must be estimated. In the above equation 4, when X(k) is the N-point normalized DFT of the discrete-time signal x(n) obtained by sampling the signal x(t), (-N / 2≦k≦N / 2-1), the k-th frequency component X(k) includes not only the k-th frequency component R(k) of the original signal, but also the complex conjugate R of the -k-th frequency component, which is the conjugate image. * (-k) is included.

[0048] X(k) = αR(k) + βR * (-k), X(-k) = αR(-k) + βR * Since (k), the time average of the product of X(k) and X(-k) is given by equation 12 below.

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[0049] In equation 12, R(k) and R(-k) are uncorrelated for thermal noise and general signals, and E[R(k)]=0, so equation 12 can be summarized as equation 13.

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[0050] The relationship between the expected values ​​of the power of the preceding and succeeding signals affected by IQ mismatch is given by equation 14 below.

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[0051] From the two equations above, we can obtain the following equation 15, which shows the signal gain mismatch and phase mismatch of the receiver LO and mixer, g. RX θ RX We can obtain an indicator to estimate this.

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[0052] In other words, g is the gain mismatch component of the mixer. RX and phase mismatch component θ RX E[|xn|] is the time average power of the mixer's output signal. 2 ] is the time average of the square of the mixer's output signal E[xn 2 This can be obtained by normalizing the signal using ]. This estimated index is obtained for any received signal r without any special conditions. BB It can operate on (t) and does not require a separate pilot signal, so it can be called a blind estimation.

[0053] If X(k) is an ergodic process, then the time average E[X(k)X(-k)] must be the same as the ensemble average. The fact that the ensemble average calculation result of X(k)X(-k) is the same as the time average, as shown in equation 16 below, indicates that X(k) is an ergodic process. Therefore, the ensemble average can be used as an estimation index for the time average.

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[0054] The estimation index mentioned above is the ensemble mean of tone image correlation. However, the ensemble mean of tone image correlation can be summarized as shown in the following formula.

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[0055] The mean of the squared signal provides an estimation index as shown in equation 18 below.

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[0056] The power of the signal before and after contamination due to IQ mismatch satisfies equation 19 below.

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[0057] From the two equations, the time-domain signal x after contamination due to IQ mismatch is obtained. n g is an index that estimates IQ gain mismatch and phase mismatch using only g. RX θ RX We can obtain an indicator to estimate this.

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[0058] As mentioned above, in order to compensate for frequency-dependent IQ mismatch, h D RX h is the discrete signal version of (t). D RX (n) must be estimated. From the signal y(t) output under the influence of the conjugate image in the channeling filter, h D RX We estimate (n). For discrete signal samples of signal y(t), let y(n) be and Y(k) be the N-point normalized DFT of y(n). Then Y(k) satisfies the relationship in Equation 21, according to the continuous-time version of the formula described above.

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[0059] When an SSB (Single-Side Band) multitone signal that satisfies half of the positive numbers in the frequency bins of an N-point DFT is input as a pilot signal, R BB (-k) = R BB * (-k)=0, and the formula can be summarized as shown in formula 22.

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[0060] The time-averaged tone image correlation Y(k)Y(-k), normalized by the average power of Y(k), is defined as m(k) as follows.

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[0061] In equation 23, h D (n) is a real filter, therefore the conjugate symmetry theorem H D (-k) = H D * Applying (k), we get H as shown in equation 24 below. D (kg RX e -jθRX It is possible to calculate this.

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[0062] Since we input a positive single-sideband (SSB) multitone, the m(k) calculation is only valid in the interval 1 ≤ k ≤ (N / 2-1) (E[|Y(k)| 2 ]=0 for -N / 2≦k≦0). However, the conjugate symmetric theorem H D (-k) = H D * (k) by H D Since the argument (arg) of (0) is 0, θ RX = -arg{Γ(0)}. However, since Γ(0) cannot be calculated directly, we estimate it by extrapolating arg{Γ(k)}.

[0063] Also, LO gain mismatch g RX This can be included in HD(k) and does not need to be calculated separately, but if it is a mismatch in the low-pass filter, |H D (0)|=1, so g RX It can be calculated as follows: =|Γ(0)|. Similarly, we can estimate |Γ(k)| by extrapolating. H in the interval -N / 2≦k≦0 D (k) can be filled in using the conjugate symmetry theorem.

[0064] Thus, for the entire interval -N / 2≦k≦N / 2-1, H D Since (k) has been estimated, we can take the N-point normalized IDFT and apply an appropriate window to obtain the compensated FIR filter hD(n), which can be expressed mathematically as shown in equation 25 below.

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[0065] In other words, by calculating the ratio of the time average of the correlation calculation between the signal and the image to the average power of the signal for each discrete frequency position, it is possible to check for mismatches and phase mismatches in the filter for each frequency position. [Explanation of Symbols]

[0066] 10 Receivers 100 I Branch 110 I Branch Mixer 120 I-branch channelization filter 123 Delay element 200 Q branch 210 Q Branch Mixer 220 Q Branch Channeling Filter 223 Difference Filter 310 Frequency-dependent mismatch estimator 320 Frequency dependent mismatch compensation section 410 Frequency-independent mismatch estimator 420 Frequency independent mismatch compensation section

Claims

1. An I-branch includes an I-branch mixer that downconverts an RF signal (radio frequency signal) to output an I component, and an I-branch channelization filter that separates a baseband signal from the output signal of the I-branch mixer; A Q branch comprising: a Q branch mixer that downconverts the RF signal to output a Q component; and a Q branch channeling filter that separates the baseband signal from the output signal of the Q branch mixer; A frequency-dependent mismatch estimation unit that calculates the frequency-dependent mismatch of the I branch and the Q branch; and A frequency-dependent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I branch and the Q branch based on the calculation results of the frequency-dependent mismatch estimation unit; The frequency-dependent mismatch estimation unit determines the transfer function of a compensation filter that compensates for frequency-dependent mismatch from the time average of the tone-image correlation of the signals output from the I-branch channelization filter and the Q-branch channelization filter, normalized by the average power of the signals output from the I-branch channelization filter and the Q-branch channelization filter. Receiver.

2. The frequency-dependent mismatch compensation unit is, The receiver according to claim 1, further comprising an I-path compensation filter having a transfer function corresponding to the result of dividing the transfer function of the Q-branch channelization filter by the transfer function of the I-branch channelization filter.

3. The receiver according to claim 2, wherein the frequency-dependent mismatch compensation unit further includes a delay line in the Q path corresponding to the delay of the compensation filter.

4. When the above value is m(k), The frequency-dependent mismatch estimation unit is, [Math 1] Satisfying the relationship, The argument and absolute value of Γ(k) are estimated by extrapolating to H Ｄ Find (k), The aforementioned H Ｄ A normalized N-point inverse DFT is performed on (k), and a window is applied to obtain the impulse response h of the compensation filter. Ｄ The receiver according to claim 1, which determines (n).

5. The frequency-dependent mismatch estimation unit is, The receiver according to claim 1, wherein the time average of the tone-image correlation is obtained using a pilot signal including SSB multitones, and the value is normalized by the average power of the signal output from the channelization filter.

6. An I-branch includes an I-branch mixer that downconverts an RF signal (radio frequency signal) to output an I component, and an I-branch channelization filter that separates a baseband signal from the output signal of the I-branch mixer; A Q branch comprising: a Q branch mixer that downconverts the RF signal to output a Q component; and a Q branch channeling filter that separates the baseband signal from the output signal of the Q branch mixer; A frequency-independent mismatch estimation unit that calculates the frequency-independent mismatch of the I branch and the Q branch; and A frequency-independent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I branch and the Q branch based on the calculation results of the frequency-independent mismatch estimation unit; The frequency-independent mismatch estimation unit calculates the gain mismatch and phase mismatch of the I-branch mixer and the Q-branch mixer from a value obtained by normalizing the time average of the power of the output signals of the I-branch mixer and the Q-branch mixer by the time average of the squares of the output signals of the I-branch mixer and the Q-branch mixer. Receiver.

7. The frequency-independent mismatch estimation unit, [Math 2] (wherein X(k): output of the mixer in the discrete-time domain, R(k): input of the mixer in the discrete-time domain) is calculated to determine the gain mismatch (g ＲＸ ), phase mismatch (θ ＲＸ The receiver according to claim 6, which estimates ).

8. The frequency-independent mismatch compensation unit is, A first multiplier connected to the aforementioned I branch, A second multiplier connected to the aforementioned Q branch, A third multiplier that amplifies the output signal of the first multiplier, The receiver according to claim 7, further comprising an adder that sums the output of the second multiplier and the output of the third multiplier.

9. The gain of the first multiplier is g ＲＸ Appropriate to, The gain of the second multiplier is cos(1 / θ). ＲＸ Appropriate to, The gain of the third multiplier is tan(θ). ＲＸ The receiver according to claim 8, which corresponds to ).

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