Power conversion equipment and air conditioning equipment

Abstract

A power conversion device (100) comprises: an inverter (3) that converts DC power into AC power and supplies the AC power to a motor (5); and a control unit (4) that generates a switching signal for the switching elements (31a to 31c, 32a to 32c) of the inverter (3). A control unit (4) performs two-phase modulation in which switching operations of switching elements (31a, 32a, 31b, 32b, or 31c, 32c) of one phase among three phases are sequentially stopped, and inserts a three-phase modulation period for performing a three-phase switching operation immediately before and / or immediately after the timing at which a phase in which a switching operation is stopped during two-phase modulation on the basis of a current flowing into and out of an inverter (3) shifts from a switching period to a switching-stopped period, and also immediately before and / or immediately after the timing at which a phase in which the switching operation is stopped shifts from a switching-stopped period to a switching period.

Description

[Technical Field]

[0001] This disclosure relates to a power conversion device and an air conditioning device equipped with an inverter that converts DC power to AC power and supplies it to a three-phase load. [Background technology]

[0002] PWM (Pulse Width Modulation) drive is commonly used as the driving method for each switching element in an inverter. There are two common modulation methods for sinusoidal modulation using PWM: "three-phase modulation" and "two-phase modulation". When using both three-phase and two-phase modulation, three-phase modulation is generally used, and switching from three-phase to two-phase modulation is often done when switching losses need to be reduced.

[0003] An inverter is equipped with a leg consisting of an upper element and a lower element (hereinafter referred to as "upper and lower elements") connected in series. The drive signal for driving the upper and lower elements of the inverter includes a period called a dead time, during which the upper and lower elements are simultaneously turned off to prevent a leg short circuit that would occur if the upper and lower elements were turned on simultaneously. On the other hand, the dead time corresponds to an external disturbance voltage from the perspective of a three-phase load. For this reason, for example, if the three-phase load is a motor, the dead time can cause ripple in the motor current and motor torque, which can adversely affect noise or vibration.

[0004] To prevent this adverse effect, Non-Patent Document 1 discloses a technique for reducing motor current and motor torque ripple by superimposing a correction voltage on the voltage command of each phase based on the DC bus voltage, motor current, and carrier frequency used to generate the drive signal. This correction is called dead time correction.

[0005] Patent Document 1 below discloses a control method for an inverter circuit in a three-phase voltage inverter that obtains three-phase AC voltage from a DC power supply. In this method, the inverter selects either a bottom-mounted two-phase modulation method or an top-mounted two-phase modulation method to reduce the maximum leakage current in the low-speed range, and a top-and-bottom mounted two-phase modulation method to ensure speed stability in the high-speed range. The bottom-mounted two-phase modulation method is a modulation method in which the voltage amplitude command of each phase is set to the minimum value every 120 degrees, which is 1 / 3 of one electrical angle cycle, and the lower element of the inverter's upper and lower elements is kept on for a period of 120 degrees. The top-mounted two-phase modulation method is a modulation method in which the sign of the bottom-mounted two-phase modulation method is reversed, that is, a modulation method in which the upper element of the inverter's upper and lower elements is kept on for a period of 120 degrees. The top-and-bottom mounted two-phase modulation method is a method in which both the bottom-mounted two-phase modulation and the top-mounted two-phase modulation are performed alternately every 60 degrees within one electrical angle cycle. [Prior art documents] [Patent Documents]

[0006] [Patent Document 1] Japanese Patent Publication No. 2006-217673 [Non-patent literature]

[0007] [Non-Patent Document 1] Hidehiko Sugimoto et al., "Theory and Practical Design of AC Servo Systems," Sogo Denshi Shuppansha, pp. 54-58. [Overview of the Initiative] [Problems that the invention aims to solve]

[0008] When performing three-phase modulation, dead time correction is performed by superimposing a correction voltage on the voltage command of each phase. On the other hand, when performing two-phase modulation, dead time correction is not performed during the switching pause period when switching operation is suspended. Therefore, when three-phase modulation and two-phase modulation are used in combination, during the three-phase modulation period immediately before and after the switching pause period in which two-phase modulation is performed, the voltage operation margin, which is the margin for voltage superposition, is small due to the influence of the minimum pulse width constraint for the protection of the switching elements or for ensuring the current detection function, making it difficult to achieve the desired dead time correction. As a result, there was a problem in that motor current and motor torque ripple could not be sufficiently suppressed.

[0009] This disclosure has been made in view of the above, and aims to provide a power converter that can sufficiently suppress motor current and motor torque ripple even when a drive system using both three-phase modulation and two-phase modulation is employed. [Means for solving the problem]

[0010] To solve the above-mentioned problems and achieve the objective, the power conversion device according to this disclosure comprises an inverter that converts DC power to AC power and supplies it to a three-phase load, and a control unit that generates switching signals for a plurality of three-phase switching elements provided in the inverter and outputs them to the inverter. The control unit performs two-phase modulation to sequentially pause the switching operation of the switching elements of one of the three phases, and, based on the current flowing into and out of the inverter, inserts a three-phase modulation period in which all three phases perform switching operation at at least one of the timings immediately before and immediately after the timing when the phase whose switching operation is paused transitions from the switching period to the switching pause period, and at least one of the timings immediately before and immediately after the timing when the phase whose switching operation has been paused transitions from the switching pause period to the switching period. [Effects of the Invention]

[0011] The power conversion device according to this disclosure has the effect of sufficiently suppressing motor current and motor torque ripple even when a drive method using both three-phase modulation and two-phase modulation is employed. [Brief explanation of the drawing]

[0012] [Figure 1] A diagram illustrating the basic configuration and basic functions of the power converter according to Embodiment 1. [Figure 2] Figure 1 shows another example configuration that incorporates the basic functions of the power converter shown in Figure 1. [Figure 3] Figure 1 shows another example configuration that incorporates the basic functions of the power converter shown in Figure 1. [Figure 4] Block diagram illustrating the basic functions related to the generation of switching signals in the control unit according to Embodiment 1. [Figure 5] Diagram illustrating the challenges of conventional technology [Figure 6] A diagram illustrating the second three-phase voltage modulation wave generated inside the control unit according to Embodiment 1. [Figure 7] This figure shows an example of a characteristic table referenced internally within the control unit according to Embodiment 1. [Figure 8] This figure shows the relationship between the u-phase Td correction value generated inside the control unit according to Embodiment 1 and the u-phase current. [Figure 9] This diagram illustrates the relationship between the first, second, and third three-phase voltage modulated waves generated inside the control unit according to Embodiment 1. [Figure 10] A diagram illustrating the method for setting the shift amount in Embodiment 1. [Figure 11] A diagram illustrating other methods for setting the shift amount in Embodiment 1. [Figure 12] Block diagram of the control unit that implements modulation scheme selection control as described with reference to Figure 10 and Figure 11. [Figure 13] This figure shows examples of phase current and q-axis current waveforms obtained by conventional two-phase modulation. [Figure 14] This figure shows examples of phase current and q-axis current waveforms when using the power conversion device according to Embodiment 1. [Figure 15] Figure 1 for explaining the three-phase modulation insertion period inserted by the control according to Embodiment 2. [Figure 16] Figure 2 illustrates the three-phase modulation insertion period inserted by the control according to Embodiment 2. [Figure 17] Figure 1 for explaining the three-phase modulation insertion period inserted by the control according to Embodiment 3. [Figure 18] Figure 2 illustrates the three-phase modulation insertion period inserted by the control according to Embodiment 3. [Figure 19] A diagram illustrating the three-phase modulation insertion period inserted by the control according to Embodiment 4. [Figure 20] A diagram showing an example configuration of an air conditioning system according to Embodiment 6. [Figure 21] This figure shows an example of a hardware configuration that realizes the functions of the control unit in Embodiments 1 to 5. [Figure 22] Figures showing other examples of hardware configurations that realize the functions of the control unit in Embodiments 1 to 5. [Modes for carrying out the invention]

[0013] The power conversion device and air conditioning device according to the embodiments of this disclosure will be described in detail below with reference to the attached drawings.

[0014] Embodiment 1. Figure 1 is a diagram illustrating the basic configuration and basic functions of a power converter 100 according to Embodiment 1. In Figure 1, the power converter 100 is connected between a commercial power supply 1 and a motor 5. The commercial power supply 1 is an example of an AC power supply. The motor 5 is a three-phase motor mounted on a three-phase load. When the power converter 100 is mounted on an air conditioning system, the compressor that compresses the refrigerant and the fan that blows air to the heat exchanger that performs heat exchange of the refrigerant are examples of the three-phase load.

[0015] The power conversion device 100 comprises a converter 2, an inverter 3, and a control unit 4. The converter 2 rectifies the power supply voltage applied from the commercial power supply 1 and outputs it to the inverter 3. If the converter 2 has a boost function, the converter 2 outputs a boosted voltage obtained by boosting the power supply voltage to the inverter 3. In other words, the converter 2 rectifies the power supply voltage applied from the commercial power supply 1 and, if necessary, boosts the said power supply voltage.

[0016] Inverter 3 is connected to the output terminal of converter 2 by electrical wiring 6a and 6b. Electrical wiring 6a and 6b are also called DC buses. Electrical wiring 6a is the DC bus on the high-potential side, and electrical wiring 6b is the DC bus on the low-potential side.

[0017] Inverter 3 has switching elements 31a, 31b, 31c, 32a, 32b, and 32c, each having a freewheeling diode connected in antiparallel. In inverter 3, switching elements 31a to 31c are the upper elements as described above, and switching elements 32a to 32c are the lower elements as described above. Inverter 3 controls the switching elements 31a to 31c and 32a to 32c to be turned on or off by the control unit 4, converting the DC power output from converter 2 into AC power with a desired amplitude and phase, and supplying it to motor 5. In Figure 1, the case where switching elements 31a to 31c and 32a to 32c are IGBTs (Insulated Gate Bipolar Transistors) is shown, but MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) may be used instead of IGBTs. In the case of MOSFETs, due to their structure, they have a built-in parasitic diode, so a configuration in which the diodes are not connected in antiparallel may be adopted.

[0018] Furthermore, the inverter 3 has shunt resistors 33a, 33b, and 33c for detecting the current flowing through each phase of the inverter 3. Shunt resistor 33a is connected between the lower element, the switching element 32a, and the low-potential electrical wiring 6b. Shunt resistors 33b and 33c are connected in the same way. The detected values ​​of shunt resistors 33a to 33c are input to the control unit 4. The control unit 4 calculates the current flowing through each phase of the inverter 3 by converting the voltages detected by the shunt resistors 33a to 33c into current. Once the current flowing through each phase of the inverter 3 is determined, the three-phase output current output from the inverter 3 to the motor 5 can also be determined.

[0019] The control unit 4 also receives the detected bus voltage Vdc from the converter 2. The bus voltage Vdc is the voltage between electrical wiring 6a and 6b, which are DC buses. The bus voltage Vdc may also be the voltage across a smoothing capacitor (not shown in Figure 1) that smooths the output voltage of the converter 2.

[0020] The control unit 4 generates switching signals for the three-phase switching elements 31a-31c and 32a-32c provided in the inverter 3 based on the current flowing through each phase of the inverter 3 and the bus voltage Vdc, and outputs them to the inverter 3. The switching elements 31a-31c and 32a-32c are controlled to turn on and off by the switching signals.

[0021] Furthermore, in the power converter 100, the configuration and arrangement of each part shown in the basic configuration of Figure 1 is just one example, and the configuration and arrangement of each part are not limited to the example shown in Figure 1. The power converter 100 according to Embodiment 1 may be configured as shown in Figure 2, for example. Figure 2 is a diagram showing another example configuration that has the basic functions of the power converter 100 shown in Figure 1.

[0022] In Figure 2, shunt resistors 33a to 33c are removed, while shunt resistor 34 is inserted into the electrical wiring 6b. Figure 1 is a configuration called a 3-shunt system, while Figure 2 is a configuration called a 1-shunt system. In the 1-shunt system, the current flowing through each phase of the inverter 3 is detected based on the timing of when the switching elements 31a to 31c and 32a to 32c are turned on or off. Note that the method of detecting the current flowing through each phase of the inverter 3 based on the detected value of the shunt resistor 34 in the 1-shunt system is well known, and further explanation is omitted here.

[0023] Furthermore, the power converter 100 according to Embodiment 1 may be configured as shown in Figure 3, for example. Figure 3 is a diagram showing yet another configuration example that has the basic functions of the power converter 100 shown in Figure 1.

[0024] In Figure 3, the shunt resistors 33a to 33c are removed, while current detectors 35a and 35b are inserted into the electrical wiring 7 connecting the inverter 3 and the motor 5. Each of the current detectors 35a and 35b detects the current of one phase of the three-phase output current, which is the output current of the inverter 3. The detected values ​​of the current detectors 35a and 35b are input to the control unit 4. The control unit 4 calculates the current of the remaining phase based on the detected values ​​of any two phases detected by the current detectors 35a and 35b.

[0025] Typical current detectors 35a and 35b include ACCTs (Alternating Current Transformers) capable of detecting only the AC component, and DCCTs (Direct Current Transformers) capable of detecting both the DC and AC components. However, any device capable of detecting the three-phase output current may be used.

[0026] FIG. 4 is a block diagram for explaining the basic functions related to the generation of switching signals in the control unit 4 according to Embodiment 1. The control unit 4 according to Embodiment 1 generates switching signals by using three-phase modulation and two-phase modulation in combination. Regarding the implementation of this control, functional blocks as shown in FIG. 4 are configured inside the control unit 4. Specifically, the control unit 4 includes a modulation method selection unit 41, a modulation wave generation unit 42, a Td correction unit 43, a PWM modulation unit 44, and a Td addition unit 45. Note that "Td" in the Td correction unit 43 and the Td addition unit 45 means dead time.

[0027] When a modulation rate command Vk, which is a positive value, and a voltage phase θ are given, the modulation wave generation unit 42 generates the first three-phase voltage modulation waves Vu1 , , * , , * , * , , * , * ,

[0030] , * , , , * ,

[0029] , * ,

[0028] , * , , * , , * , , * , Vv1 * , Vw1 * as follows.

[0028] Vu1 * = Vk × cos θ Vv1 * = Vk × cos(θ - 2 / 3π) Vw1 * = Vk × cos(θ - 4 / 3π) …(1)

[0029] [[ID=3B]]The first three-phase voltage modulation waves Vu1 * , Vv1 * , Vw1 * essentially correspond to the desired voltage to be output from the inverter 3 and are generated based on the voltage command output from the upper control system (not shown). The voltage phase θ is the phase of the three-phase output voltage, which is the output voltage of the inverter 3, and is the phase when the rotation of the motor 5 is viewed in terms of electrical angle.

[0030] The modulation method selection unit 41 selects and indicates the modulation method based on the modulation rate command Vk and the voltage phase θ. When instructed to perform three-phase modulation, the modulation wave generation unit 42 converts the first three-phase voltage modulation waves Vu1 <> * , Vv1 <> * , Vw1 <> * into the second three-phase voltage modulation waves Vu2 <> * , Vv2 <> *VW2 * It outputs as follows. Also, when two-phase modulation is instructed, the modulation wave generation unit 42 generates a second three-phase voltage modulation wave Vu2 shown in equation (2) below. * ,Vv2 * VW2 * It generates and outputs to the Td correction unit 43.

[0031] Vu2 * =Vu1 * -Vcom Vv2 * =Vv1 * -Vcom VW2 * =Vw1 * -Vcom …(2)

[0032] In equation (2) above, Vcom is a three-phase common signal. As shown in equation (2) above, the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * This is the first three-phase voltage modulated wave Vu1 * ,Vv1 * VW1 * Since it is generated by subtracting a three-phase common signal Vcom of the same value from it, the line-to-line voltage values ​​between each phase are maintained. Furthermore, when implementing the under-mounted two-phase modulation shown in Patent Document 1, the three-phase common signal Vcom can be calculated, for example, by the following equation (3).

[0033] Vcom=min(Vu1 * ,Vv1 * VW1 * ) + 1 … (3)

[0034] In equation (3) above, min(Vu1 * ,Vv1 * VW1 * ) is the first three-phase voltage modulated wave Vu1 * ,Vv1 * VW1 * This is a function that finds the minimum value among them.

[0035] The Td correction unit 43 generates a second three-phase voltage modulated wave Vu2 based on the three-phase output currents iu, iv, iw, bus voltage Vdc, and carrier frequency fc. * ,Vv2 * VW2 * The corrected third three-phase voltage modulated wave Vu3 * ,Vv3 * VW3 * The Td correction unit 43 implements a current-to-deadness characteristic when the dead time Td is viewed as an external disturbance voltage. The Td correction unit 43 refers to the current-to-deadness characteristic corresponding to the values ​​of the three-phase output currents iu, iv, and iw to generate the second three-phase voltage modulated wave Vu2. * ,Vv2 * VW2 * Correct it.

[0036] The PWM modulation unit 44 internally generates a carrier signal with the commanded carrier frequency fc, and a third three-phase voltage modulated wave Vu3 * ,Vv3 * VW3 * The system compares this with the carrier signal and generates a switching signal SW1 based on their relative magnitudes. This switching signal SW1 is the signal before the dead time Td is added. The Td addition unit 45 generates a switching signal SW by adding the dead time Td to the switching signal SW1 and outputs it to the inverter 3.

[0037] Next, we will explain the problems of the conventional technology. Figure 5 is a diagram illustrating the problems of the conventional technology. The operating waveform in Figure 5 is for two-phase modulation, and the solid line, dashed line, and thick dashed line represent the u-phase voltage modulation wave Vu2, respectively. * , v-phase voltage modulated wave Vv2 * and w-phase voltage modulated wave Vw2 * This represents the three-phase output currents iu, iv, and iw, with the thick solid line representing the u-phase current iu. The horizontal axis shows the electrical angle and phase angle, and the vertical axis shows the voltage value of each modulated wave or the current value of the u-phase current iu. The voltage values ​​on the vertical axis are normalized to ±1, and the values ​​on the vertical axis correspond to the modulation rate. The same applies to Figures 6, 8, 9, and 15-19, which will be described later.

[0038] In the example in Figure 5, the u-phase voltage modulated wave Vu2 is in the range of 120 to 240 degrees. * Since the first limiter value Limit1 has been reached, the u-phase switching operation is paused. In Figure 5, this range of electrical angular phase angle is represented as "Yu". In this paper, the range of electrical angular phase angle in which switching operation is paused during two-phase modulation is called the "switching pause period", and the range of electrical angular phase angle outside the switching pause period, i.e., in which switching operation is not paused, is called the "switching period". This switching period is the three-phase modulation period in which three-phase modulation is performed.

[0039] In Figure 5, focus on the periods Xa and Xb indicated by the thick dashed rectangular frames. Period Xa is the three-phase modulation period immediately preceding the u-phase switching pause period Yu in which two-phase modulation is performed. Period Xb is the three-phase modulation period immediately following the u-phase switching pause period Yu in which two-phase modulation is performed. As explained in the section [Problems to be Solved by the Invention], there is a constraint of minimum pulse width when generating the switching signal SW. The minimum pulse width is set to protect the switching elements 31a~31c and 32a~32c, and also to ensure the current detection function.

[0040] In Figure 5, the u-phase voltage modulated wave Vu2 * Hatching is applied to the area enclosed by the rectangular frame during periods Xa and Xb. The size of the hatched area represents the voltage adjustment margin for Td correction. A larger hatched area means a larger voltage adjustment margin. Extending periods Xa and Xb would increase the voltage adjustment margin, but this method does not solve the problem. This is because the closer we get to the u-phase switching pause period Yu, the greater the u-phase voltage modulation wave Vu2 *The difference between this value and the first limiter value Limit1 becomes smaller, and the voltage control margin decreases rapidly, making it difficult to adequately correct Td. Therefore, the periods immediately before and after the u-phase switching pause period Yu are periods in which voltage control errors exist that prevent sufficient Td correction. On the other hand, during the u-phase switching pause period Yu, there is no switching operation and no Td correction is performed, so there are no voltage control errors. Consequently, at the timing of the transition from period Xa to the u-phase switching pause period Yu, and from the u-phase switching pause period Yu to period Xb, the voltage control error changes in a step-like manner, causing current ripple and torque ripple.

[0041] Therefore, in Embodiment 1, the voltage control error near the boundary between period Xa and u-phase switching pause period Yu, and near the boundary between u-phase switching pause period Yu and period Xb is reduced, and the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * We will modify the calculation method. The specific calculation method is as follows:

[0042] First, the three-phase common signal Vcom is generated using equation (4) below, instead of equation (3) above.

[0043] Vcom=min(Vu1 * ,Vv1 * VW1 * ) +(1-Δduty)(Δduty>0) …(4)

[0044] In equation (4) above, Δduty is the amount of shift to shift the first limiter value Limit1 in the voltage direction. The second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * This is calculated by substituting the three-phase common signal Vcom obtained from equation (4) above into equation (2) above. For the sake of explanation, equation (2) above is reproduced as equation (5).

[0045] Vu2* =Vu1 * -Vcom Vv2 * =Vv1 * -Vcom VW2 * =Vw1 * -Vcom …(5)(Reprinted)

[0046] Figure 6 shows the second three-phase voltage modulation wave Vu2 generated inside the control unit 4 according to Embodiment 1. * ,Vv2 * VW2 * This is a diagram for explanation purposes. Figure 6 shows the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * The waveform for one period of the electrical angle is shown. Since the shift amount Δduty is set to Δduty>0, the lower limit value is shifted in the positive direction of the voltage by +Δduty from "-1" (=Limit1). In this paper, the limiter value shifted in the voltage direction by the shift amount Δduty is denoted as "Limit2" and referred to as the "second limiter value". There is a relationship between the first limiter value Limit1, the second limiter value Limit2, and the shift amount Δduty: "Limit2 = Limit1 + Δduty". The appropriate range for Δduty will be discussed later.

[0047] Furthermore, in Embodiment 1, the Td correction unit 43 uses the following equation (6) to generate a third three-phase voltage modulated wave Vu3 * ,Vv3 * VW3 * Generates.

[0048] Vu3 * =Vu2 * +Vtd_u Vv3 * =Vv2 * +Vtd_v VW3 * =Vw2 * PROVtd_w …(6)

[0049] In equation (6) above, Vtd_u, Vtd_v, and Vtd_w are Td correction values ​​for each phase (uvw). These u-phase Td correction value Vtd_u, v-phase Td correction value Vtd_v, and w-phase Td correction value Vtd_w can be calculated by referring to a characteristic table as shown in Figure 7. Figure 7 is a diagram showing an example of a characteristic table referenced internally by the control unit 4 according to Embodiment 1. The horizontal axis of Figure 7 shows the absolute values ​​of the instantaneous values ​​of the three-phase output currents iu, iv, and iw, and the vertical axis shows the absolute value |Vtd| of the Td correction value Vtd.

[0050] The Td correction unit 43 takes the absolute values ​​of the instantaneous values ​​of the three-phase output currents iu, iv, and iw as arguments and uses the characteristic table in Figure 7 to determine the absolute value of the Td correction value Vtd|. Furthermore, the Td correction unit 43 generates the Td correction values ​​Vtd_u, Vtd_v, and Vtd_w using the following equation (7).

[0051] Vtd_u = |Vtd| × sign(iu) Vtd_v=|Vtd|×sign(iv) Vtd_w=|Vtd|×sign(iw) …(7)

[0052] In equation (7) above, sign(iu) is a function that obtains the sign of the instantaneous value of the u-phase current iu, and takes one of the values ​​of "1", "0", or "-1". The same applies to sign(iv) and sign(iw). The three-phase output currents iu, iv, and iw can be obtained in any of the power converters 100 shown in Figures 1 to 5.

[0053] Figure 8 shows the relationship between the u-phase Td correction value Vtd_u and the u-phase current iu, which are generated internally within the control unit 4 according to Embodiment 1. In Figure 8, the u-phase Td correction value Vtd_u is shown by a solid line, and the u-phase current iu is shown by a dashed line. As shown in Figure 8, the direction of correction of the u-phase Td correction value Vtd_u is reversed depending on the sign of the instantaneous value of the u-phase current iu. The other v-phase Td correction value Vtd_v and w-phase Td correction value Vtd_w have a similar relationship.

[0054] FIG. 9 is a diagram for explaining the relationship between the first, second, and third three-phase voltage modulation waves generated inside the control unit 4 according to the first embodiment. In FIG. 9, the same waveforms and the same elements as those in FIGS. 5 and 6 are denoted by the same reference numerals.

[0055] In FIG. 9, for the sake of avoiding complexity, only the u-phase voltage modulation waves Vu1 * , Vv1 * , Vw1 * of the first three-phase voltage modulation wave and Vu3 * , Vv3 * , Vw3 * of the third three-phase voltage modulation wave are shown. * , Vu3 * are shown.

[0056] Comparing FIG. 9 with FIG. 5, in FIG. 9, periods Xa and Xb are respectively periods Xa' and Xb'. Also, in FIG. 9, the u-phase switching pause period Yu is the u-phase switching pause period Yu'.

[0057] Comparing the u-phase switching pause period Yu and the u-phase switching pause period Yu', the u-phase switching pause period Yu' is shorter than the u-phase switching pause period Yu. The reason for this is that the period Xa shown in FIG. 5 includes only the period immediately before the timing when the u-phase switching operation transitions from the switching period to the switching pause period, whereas the period Xa' shown in FIG. 9 includes the period immediately after the timing when the u-phase switching operation transitions from the switching period to the switching pause period. Similarly, the period Xb shown in FIG. 5 includes only the period immediately after the timing when the u-phase switching operation transitions from the switching pause period to the switching period, whereas the period Xb' shown in FIG. 9 includes the period immediately before the timing when the u-phase switching operation transitions from the switching pause period to the switching period. As a result, the relationship Yu' < Yu holds between the u-phase switching pause period Yu and the u-phase switching pause period Yu'.

[0058] As shown in the above equation (6), the u-phase voltage modulation wave Vu3 * is superimposed with the u-phase Td correction value Vtd_u with respect to the u-phase voltage modulation wave Vu2 * The u-phase Td correction value Vtd_u is a voltage corresponding to the absolute value of the instantaneous value of the three-phase output currents iu, iv, iw (in the case of the u-phase, the u-phase current iu) as shown in FIG. 7. Therefore, in the section where the u-phase voltage modulation wave Vu2 * is less than or equal to the second limiter value Limit2 which is the lower limit value, the waveform of the u-phase voltage modulation wave Vu3 * superimposed with the u-phase Td correction value Vtd_u becomes a waveform with a gently depressed bottom as shown in FIG. 9. By appropriately setting the magnitude of the shift amount Δduty according to the magnitude of the u-phase Td correction value Vtd_u, the u-phase voltage modulation wave Vu3 * changes up and down with a gentle slope across -1. And during the period when the u-phase voltage modulation wave Vu3 * is below the first limiter value Limit1, it is a period when the switching operation does not occur in the u-phase.

[0059] Therefore, in the periods Xa' and Xb' according to the first embodiment shown in FIG. 9, it is possible to appropriately perform the Td correction. Also, in the periods Xa' and Xb' according to the first embodiment, at the timing of the switching from the period Xa to the u-phase switching pause period Yu and from the u-phase switching pause period Yu to the period Xb, the step-like change in the voltage control error can be reduced, so that the current ripple and torque ripple can be suppressed.

[0060] Although FIG. 9 describes the u-phase, the same description can be made for the other v- and w-phases. The waveforms of the voltages and currents of the u-, v-, and w-phases are waveforms maintaining symmetry with a phase angle difference shift of 120 [degrees] from each other, and it is needless to say that the same explanation can be made. The following description also shows only the u-phase.

[0061] As is clear from the above description, according to the power converter of Embodiment 1, a three-phase modulation period in which all three phases perform switching operations is inserted immediately before and immediately after the timing when the phase whose switching operation is paused during two-phase modulation transitions from the switching period to the switching pause period. Furthermore, a three-phase modulation period in which all three phases perform switching operations is inserted immediately before and immediately after the timing when the phase whose switching operation has been paused due to the implementation of two-phase modulation transitions from the switching pause period to the switching period.

[0062] Next, we will explain an appropriate method for setting the shift amount Δduty. First, in control using both three-phase and two-phase modulation, in order to minimize the change in voltage control error at the timing when the switching period and switching pause period switch to each other, as shown in Figure 9, the u-phase voltage modulated wave Vu3 after Td correction is used. * It is preferable that the voltage gradually falls below the first limiter value Limit1. Here, as shown in equation (6) above, the u-phase voltage modulated wave Vu2 * ,Vu3 * Vu3 * =Vu2 * Because of the relationship +Vtd_u, it is desirable to determine the shift amount Δduty in relation to the Td correction value Vtd_u. Therefore, when Vtd_peak is the maximum value of the absolute value |Vtd_u| of the Td correction value Vtd_u under the operating conditions, the shift amount Δduty is set to satisfy equation (8) below.

[0063] Vtd_peak × 0.3 < Δduty <Vtd_peak×0.7 …(8)

[0064] Furthermore, as shown in Figure 7, the absolute value |Vtd_u| and the maximum value Vtd_peak of the Td correction value Vtd_u have the characteristic of fluctuating according to the magnitude of the u-phase current iu. Therefore, setting the shift amount Δduty based on equation (8) above is equivalent to setting it according to the effective value of the motor current output from inverter 3 to motor 5. For this reason, the shift amount Δduty may also be set as shown in equation (9) below.

[0065] Δduty = Ktd_I × (Effective value of motor current) …(9) However, Ktd_I: proportionality coefficient

[0066] Furthermore, if motor 5 is a motor that drives a fluid load such as a fan or compressor, the load torque is approximately proportional to the square of the rotational speed of motor 5. Therefore, the effective value of the motor current when driving these fans or compressors increases with increasing rotational speed. For this reason, when the three-phase load is a fluid load such as a fan or compressor, instead of determining the shift amount Δduty as a motor current-dependent characteristic as in equation (9) above, it is possible to determine it by converting the description to a rotational speed characteristic. A concrete concept is shown in Figure 10. Figure 10 is a diagram used to explain the method for setting the shift amount Δduty in Embodiment 1.

[0067] The upper part of Figure 10 shows the characteristics of the change in the effective value of the motor current according to the rotational speed. Therefore, as shown in the lower part of Figure 10, it is desirable to set the shift amount Δduty in accordance with the characteristics of the change in the effective value of the motor current. Generally, rotational speed is less prone to sudden changes than current, so setting the shift amount Δduty according to the rotational speed has the advantage of enabling stable Td correction.

[0068] Figure 11 is a diagram illustrating another method for setting the shift amount Δduty in Embodiment 1. The characteristic change of the motor current RMS value shown in the upper part of Figure 11 is the same as the characteristic shown in the upper part of Figure 10. In Figure 10, the shift amount Δduty was set according to the characteristic change of the motor current RMS value, but the characteristic for determining the setting value may be switched depending on the rotational speed range. In the example shown in the lower part of Figure 11, the rotational speed range is set as follows: from zero speed to the first speed is the low speed range, from the first speed to the second speed is the medium speed range, and the second speed and above, including the maximum rotational speed, is the high speed range.

[0069] The low-speed range, where the rotational speed is low, is a frequently used operating condition when starting the inverter 3, and the voltage applied to the motor 5 is small. Therefore, in this low-speed range, it is desirable to maintain constant three-phase modulation in order to eliminate as much disturbance voltage as possible associated with switching between two-phase and three-phase modulation. Accordingly, as shown in Figure 11, the shift amount Δduty is set to a relatively large value, at least up to about 0.5. By setting the shift amount Δduty to such a value, it becomes possible to set it so that no switching pause period is inserted, regardless of what Td correction value Vtd is superimposed.

[0070] Furthermore, in the high-speed range with high rotational speed, the induced voltage of the motor is high, and the influence of disturbance voltages caused by the dead time Td is small. For this reason, even with conventional two-phase modulation, the effects of current ripple and torque ripple are small. It is also desirable to reduce the three-phase modulation period to reduce switching losses. Therefore, as shown in Figure 11, the shift amount Δduty is set to zero. By setting the shift amount Δduty to zero, the control operation becomes equivalent to that of conventional two-phase modulation, and it is possible to improve the operating efficiency of inverter 3 by reducing switching losses.

[0071] Furthermore, in the medium speed range where the rotational speed is moderate, the curve characteristics are connected by an exponentially decreasing curve so that the change in shift amount Δduty between the low speed range and the high speed range is smooth. Note that Figure 11 is just one example; any curve that makes the change in shift amount Δduty smooth is acceptable, and it may even be a straight line.

[0072] Figure 12 is a block diagram of the control unit 4 that implements the modulation method selection control described with reference to Figures 10 and 11. In Figure 12, components that are the same as or equivalent to those in Figure 4 are denoted by the same reference numerals. In comparison with Figure 4, in Figure 12, the modulation method selection unit 41 is replaced by the modulation method selection unit 41A. In addition to the modulation rate command Vk and voltage phase θ, the modulation method selection unit 41A receives the motor current RMS value Irms or rotational speed Rrot from a higher-level control system (not shown). In addition to the functions of the modulation method selection unit 41, the modulation method selection unit 41A has the added function of calculating the shift amount Δduty based on the motor current RMS value Irms or rotational speed Rrot. The modulation method selection unit 41A sets the shift amount Δduty according to the change characteristics of the motor current RMS value Irms, as described with reference to Figure 10, for example, and outputs it to the modulation wave generation unit 42. Alternatively, the modulation method selection unit 41A, as explained with reference to Figure 11, for example, determines the rotational speed range from the rotational speed Rrot, sets the shift amount Δduty according to the rotational speed range, and outputs it to the modulation wave generation unit 42. The modulation wave generation unit 42 generates a second three-phase voltage modulation wave Vu2 according to the instructions of the modulation method selection unit 41A. * ,Vv2 * VW2 * This generates the following. Although not shown in Figure 12, the Td correction values ​​Vtd_u, Vtd_v, and Vtd_w for each phase calculated by the Td correction unit 43 may also be reflected in the calculation of the shift amount Δduty performed by the modulation method selection unit 41A.

[0073] Figure 13 shows an example of the waveforms of the phase current and q-axis current due to conventional two-phase modulation. Figure 14 shows an example of the waveforms of the phase current and q-axis current when using the power converter 100 according to Embodiment 1. In each figure, the phase current is shown on the upper side and the q-axis current is shown on the lower side. The phase current is the current of any one of the three-phase output currents iu, iv, and iw. The q-axis current is the current component that contributes to the motor torque when the phase current is converted to a rotating Cartesian coordinate system. The waveforms in Figures 13 and 14 are the result of driving the inverter 3 at a constant frequency of electrical angle 50 Hz while deliberately setting the response of the current control system to a low value in order to understand the effect of disturbance voltage caused by the dead time Td and the effect of switching between the switching period and the switching pause period in two-phase modulation. The larger the torque ripple generated in the motor 5, the larger the pulsation of the q-axis current.

[0074] Figure 13 shows the operating waveform when the shift amount Δduty = 0, where the phase current is distorted and the q-axis current shows pulsations at a frequency three times the electrical angle of 50 Hz. On the other hand, Figure 14 shows that the distortion of the phase current is suppressed, and the pulsations at three times the frequency are also suppressed. Therefore, current ripple can be suppressed by using the power converter 100 according to Embodiment 1. This makes it possible to suppress torque ripple generated in the motor 5, and thus also suppress noise caused by torque ripple. Furthermore, since current ripple can be suppressed by using the power converter 100 according to Embodiment 1, losses in wiring resistance and winding resistance caused by current ripple can be suppressed.

[0075] As described above, the power conversion device according to Embodiment 1 comprises an inverter that converts DC power to AC power and supplies it to a three-phase load, and a control unit that generates switching signals for a plurality of three-phase switching elements provided in the inverter and outputs them to the inverter. The control unit performs two-phase modulation, which sequentially pauses the switching operation of the switching elements of one of the three phases, and inserts a three-phase modulation period in which all three phases perform switching operation, immediately before and immediately after the timing when the phase whose switching operation is paused transitions from the switching period to the switching pause period, and immediately before and immediately after the timing when the phase whose switching operation has been paused transitions from the switching pause period to the switching period, based on the current flowing into and out of the inverter. This control makes it possible to reduce the step-like change in voltage control error at the timing of the transition from the switching period to the switching pause period, and from the switching pause period to the switching period. As a result, even when a drive method that uses both three-phase modulation and two-phase modulation is adopted, it is possible to sufficiently suppress the ripple of the motor current and motor torque.

[0076] In the power converter configured as described above, the control unit generates a first three-phase voltage modulated wave based on a voltage command output from a higher-level control system, and calculates a three-phase common signal for setting a switching pause period while maintaining the line-to-line voltage values ​​for the generated first three-phase voltage modulated wave. The control unit also calculates a second three-phase voltage modulated wave by superimposing the three-phase common signal on the first three-phase voltage modulated wave. Furthermore, the control unit generates a switching signal based on a third three-phase voltage modulated wave obtained by correcting errors caused by the dead time added to the switching signal for the second three-phase voltage modulated wave. The three-phase common signal can be calculated based on a first difference, which is the difference between a first three-phase voltage modulated wave with a value close to a predetermined first limiter value and the said first limiter value. When inserting a three-phase modulation period, the three-phase common signal can be calculated based on the second difference, which is the difference between a first three-phase voltage modulated wave with a value close to the first limiter value and a second limiter value with an absolute value smaller than the first limiter value. When the difference between the first limiter value and the second limiter value is taken as the third difference, this third difference can be determined based on at least one of the following: the effective value of the three-phase output current, which is the output current of the inverter, or the voltage-current phase difference, which is the phase difference between the three-phase output voltage, which is the output voltage of the inverter, and the three-phase output current.

[0077] When the three-phase load is a motor, the control unit may perform three-phase modulation without performing two-phase modulation under operating conditions where the motor's rotational speed is below a predetermined threshold. On the other hand, under operating conditions where the motor's rotational speed is above a predetermined threshold, the control unit may not insert a three-phase modulation period. Furthermore, under operating conditions where the motor's rotational speed is below a predetermined threshold, the control unit may determine the value of the third difference based on the value of the disturbance voltage caused by the dead time, thereby creating a three-phase modulation period in which each phase switches for the entire duration. Also, under operating conditions where the motor's rotational speed is above a predetermined threshold, the control unit may set the third difference to zero, thereby not inserting a three-phase modulation period for the entire duration.

[0078] Embodiment 2. The magnitude of the disturbance voltage caused by the dead time Td, and its correction voltage value, the Td correction value Vtd, are generated according to the instantaneous magnitudes of the three-phase output currents iu, iv, and iw, as described above. Furthermore, since the polarity of the disturbance voltage and the Td correction value Vtd depends on the polarity of the three-phase output currents iu, iv, and iw, the remaining margin of the modulation wave operation width also depends on the magnitude and polarity of the three-phase output currents iu, iv, and iw. Therefore, the second three-phase voltage modulation wave Vu2 * ,Vv2 * VW2 * Near the first limiter value Limit1, which is the lower limit, the instantaneous values ​​of the three-phase output currents iu, iv, iw are large, and the second three-phase voltage modulation wave Vu2 * ,Vv2 * VW2 * Under conditions where the polarity of the voltage and the polarity of its correction voltage value, Td correction value Vtd, are the same, the remaining margin of the modulated wave operation width decreases, and the risk of residual disturbance voltage due to insufficient Td correction increases.

[0079] On the other hand, in order to maintain the switching loss suppression effect, which is the inherent advantage of two-phase modulation, it is desirable to make the two-phase modulation period as long as possible. In light of these risks and advantages, in Embodiment 2, the two-phase modulation and three-phase modulation are switched multiple times within a period of electrical angular phase angle from 0 to 360 degrees. By doing so, a voltage operation margin for the three-phase voltage modulated wave is provided by switching from two-phase modulation to three-phase modulation during the switching period immediately before or after the switching pause period of two-phase modulation.

[0080] Specifically, the control unit 4 switches between two-phase modulation and three-phase modulation multiple times within an electrical angular phase angle of 0 to 360 degrees, based on at least one of the following: voltage-current phase difference, current values ​​of three-phase output currents iu, iv, iw, Td correction value Vtd, and modulation rate. When switching between two-phase modulation and three-phase modulation, a period of three-phase modulation is inserted immediately before or after the switching timing, according to at least one of the current-voltage phase difference, current values ​​of three-phase output currents iu, iv, iw, and Td correction value Vtd.

[0081] In Figure 5, despite the switching period, the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * In order to ensure that Td correction functions reliably even during periods Xa and Xb close to the first limiter value Limit1, which is the lower limit, a second three-phase voltage modulated wave Vu2 can be used to perform modulation rate manipulation equivalent to the Td correction value Vtd. * ,Vv2 * VW2 * To maintain the line voltage of the inverter output, the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * This applies not only to the phases close to the first limiter value Limit1, but also to the second three-phase voltage modulation wave Vu2 of all phases. * ,Vv2 * VW2 * It is necessary to shift it upward. Performing such a shift operation will result in the second three-phase voltage modulated wave Vu2 of all phases. * ,Vv2 * VW2 * None of these phases will fall below the first limiter value Limit1, and there will be no phase that is "sticking to the bottom". This is equivalent to inserting a period of three-phase modulation in which all three phases are switching. In order to maintain the switching loss suppression effect, which is the original benefit of two-phase modulation, it is desirable to shorten the period in which three-phase modulation is inserted as much as possible. For this reason, in periods Xa and Xb, the second three-phase voltage modulated wave Vu2 * ,Vv2 * VW2 * When the phase angle conditions allow for sufficient voltage control margin, it is desirable to immediately revert to two-phase modulation.

[0082] As described above, with the intention of ensuring the minimum modulation wave operation width and inserting the minimum three-phase modulation period, the three-phase modulation insertion period X3in is set based on at least one of the voltage-current phase difference, the current values ​​of the three-phase output currents iu, iv, and iw, the Td correction value Vtd, and the modulation rate, and during the said three-phase modulation insertion period X3in, the three-phase common signal Vcom is provided as shown in equation (10) below.

[0083] (Three-phase modulation insertion period) Vcom=min(Vu1 * ,Vv1 * VW1 * ) + (1 - Δduty) (Except during the three-phase modulation insertion period) Vcom=min(Vu1 * ,Vv1 * VW1 * )+1 …(10)

[0084] Figures 15 and 16 are the first and second diagrams illustrating the three-phase modulation insertion period X3in inserted by the control according to Embodiment 2. In Figures 15 and 16, the same waveforms and elements as in Figure 9 are denoted by the same reference numerals.

[0085] In Figures 15 and 16, to avoid complexity, the first three-phase voltage modulated wave Vu1 is shown. * ,Vv1 * VW1 * And for the three-phase output currents iu, iv, iw, the u-phase voltage modulated wave Vu1 * And only the u-phase current iu is shown. Figure 15 shows the u-phase current iu as the u-phase voltage modulated wave Vu1 * This is an example where the phase is leading relative to the u-phase current iu and the u-phase voltage modulated wave Vu1. * This is an example where the phase lags relative to the given phase. In Figures 15 and 16, the three-phase modulation insertion period X3in is shown by a thick solid rectangle.

[0086] In Embodiment 1, periods Xa' and Xb' were provided before and after the u-phase switching pause period Yu in conventional two-phase modulation to obtain an appropriate voltage operation margin. Although not specifically mentioned in the explanation of Figure 9, the periods Xa' and Xb' shown in Figure 9 are basically equal in phase angle width. On the other hand, in Embodiment 2, in the example of Figure 15, the periods Xa'', and Xb'' shown in the thick dashed rectangular frame are not necessarily Xa''=Xb''. As shown in Figure 15, the u-phase current iu is equal to the u-phase voltage modulated wave Vu1 *When the phase is leading relative to the current, the polarity of the u-phase Td correction value Vtd_u reverses at the zero-crossing timing of the u-phase current iu, so there is no need to provide a voltage adjustment margin after the zero-crossing. For this reason, it is possible to switch from three-phase modulation to two-phase modulation at that timing. In the example in Figure 15, the switch from three-phase modulation to two-phase modulation is performed at the zero-crossing timing, so the relationship between period Xa'' and period Xb'' is Xa''>Xb''.

[0087] On the other hand, as shown in Figure 16, the u-phase current iu is the u-phase voltage modulated wave Vu1 * If the phase lags relative to the current iu, the polarity of the u-phase Td correction value Vtd_u reverses at the zero-crossing timing of the u-phase current iu, so the u-phase voltage modulated wave Vu1 * The polarity of the current and the polarity of the u-phase Td correction value Vtd_u become the same, resulting in a smaller voltage operation margin. For this reason, it is desirable to switch from two-phase modulation to three-phase modulation at the zero-crossing timing of the u-phase current iu. In the example in Figure 16, where the modulation is switched from two-phase to three-phase modulation at the zero-crossing timing, the relationship between period Xa''' and period Xb''' is Xa''' <Xb'''となっている。

[0088] Furthermore, the widths of the three-phase modulation insertion period X3in, namely periods Xa'',Xb'' and Xa''',Xb'''', may be changed according to the Td correction value Vtd, in addition to the voltage-current phase difference mentioned above. The Td correction value Vtd depends on the magnitude of the three-phase output currents iu, iv, and iw. When the current values ​​of the three-phase output currents iu, iv, and iw are small, the operating margin of the modulation wave may be small. For this reason, the widths of the periods Xa'',Xb'',Xa''',Xb'''' can be shortened accordingly.

[0089] Furthermore, the voltage control margin increases as the modulation rate decreases. Therefore, the width of the period Xa'',Xb'',Xa''',Xb'''' can be shortened accordingly. In addition, when switching between two-phase modulation and three-phase modulation, the switching timing may be controlled based on the width of the period Xa'',Xb'',Xa''',Xb'''', i.e., the width of the three-phase modulation insertion period X3in.

[0090] As described above, according to the power converter of Embodiment 2, the control unit controls the timing and duration of the three-phase modulation period based on at least one of the following: the current value of the inverter's three-phase output current, the phase difference between the inverter's three-phase output voltage and the three-phase output current, a voltage correction value for correcting errors caused by dead time, and the modulation rate of the three-phase output voltage. This control eliminates unnecessary voltage operation margins and allows for setting the width of the truly necessary three-phase modulation insertion period. As a result, the period for inserting three-phase modulation can be shortened as much as possible, and the switching loss suppression effect, which is an inherent advantage of two-phase modulation, can be maintained.

[0091] Embodiment 3. In Embodiments 1 and 2, a three-phase modulation period was inserted in which all three phases were in switching operation immediately before and immediately after the timing when the phase whose switching operation was paused transitioned from the switching period to the switching pause period, and immediately before and immediately after the timing when the phase whose switching operation was paused transitioned from the switching pause period to the switching period, when two-phase modulation was implemented. On the other hand, for example, in operating conditions or applications with small current amplitude, the influence of disturbance voltage caused by the dead time Td is small, so even if a period with a small voltage operation margin remains, the generation of current ripple is small. For this reason, the width of the three-phase modulation insertion period X3in can be further reduced compared to Embodiments 1 and 2. Embodiment 3 will be described with reference to Figures 17 and 18.

[0092] Figures 17 and 18 are the first and second diagrams illustrating the three-phase modulation insertion period X3in inserted by the control according to Embodiment 3. In Figures 17 and 18, the same waveforms and elements as those in Figures 15 and 16 are denoted by the same reference numerals.

[0093] In Figures 17 and 18, to avoid complexity, the first three-phase voltage modulated wave Vu1 is shown. * ,Vv1 * VW1 * And for the three-phase output currents iu, iv, iw, the u-phase voltage modulated wave Vu1 *And only the u-phase current iu is shown. Figure 17 shows the u-phase current iu as the u-phase voltage modulated wave Vu1 * This is an example where the phase is leading relative to the u-phase current iu and the u-phase voltage modulated wave Vu1. * This is an example where the phase lags relative to the given phase. In Figures 17 and 18, the three-phase modulation insertion period X3in is shown by a thick solid rectangle.

[0094] In the example in Figure 17, the three-phase modulation insertion period X3in is set only immediately before the phase that pauses switching operation transitions from the switching period to the switching pause period, and only immediately before the phase that has paused switching operation transitions from the switching pause period to the switching period. In the example in Figure 18, the three-phase modulation insertion period X3in is set only immediately after the phase that pauses switching operation transitions from the switching period to the switching pause period, and only immediately after the phase that has paused switching operation transitions from the switching pause period to the switching period. In other words, in this embodiment, when the three-phase output currents iu, iv, iw are in phase lead relative to the three-phase output voltage, the three-phase modulation insertion period X3in is set only immediately before the timing of the switch between the switching period and the switching pause period, and when the three-phase output currents iu, iv, iw are in phase lag relative to the three-phase output voltage, the three-phase modulation insertion period X3in is set only immediately after the timing of the switch between the switching period and the switching pause period.

[0095] In the example in Figure 17, the three-phase modulation insertion period X3in is inserted only immediately before the timing of the switchover between the switching period and the switching pause period, but this does not prevent the three-phase modulation insertion period X3in from being set immediately after the switchover timing. Similarly, in the example in Figure 18, the three-phase modulation insertion period X3in is set only immediately after the timing of the switchover between the switching period and the switching pause period, but this does not prevent the three-phase modulation insertion period X3in from being set immediately before the switchover timing. According to the control of Embodiment 3, the insertion period of three-phase modulation can be limited, so that switching losses can be suppressed while suppressing current ripple and torque ripple. In other words, by using the power converter according to Embodiment 3, it is possible to achieve both the suppression of current ripple and torque ripple and the suppression of switching losses.

[0096] As described above, according to the power converter of Embodiment 3, the control unit performs two-phase modulation by sequentially pausing the switching operation of the switching elements of one of the three phases, and inserts a three-phase modulation period in which all three phases perform switching operation at at least one of the timings immediately before and immediately after the transition of the phase whose switching operation is paused from the switching period to the switching pause period, and at least one of the timings immediately before and immediately after the transition of the phase whose switching operation has been paused from the switching pause period to the switching period, based on the current flowing into and out of the inverter. This control limits the insertion period of three-phase modulation. This makes it possible to achieve both the suppression of current ripple and torque ripple and the suppression of switching losses.

[0097] Embodiment 4. To maintain the line voltage of the three-phase inverter output, it is necessary to shift the modulation waves of all phases in the same direction, and this shift operation causes the neutral point potential of the motor 5 to fluctuate. Since fluctuations in the neutral point potential have adverse effects such as accelerating the progression of motor shaft electrolytic corrosion, fluctuations in the neutral point potential are undesirable. Therefore, in Embodiment 4, the fluctuation of the neutral point potential is suppressed while ensuring the accuracy of Td correction. The control according to Embodiment 4 will be described below with reference to Figure 19. Figure 19 is a diagram for explaining the three-phase modulation insertion period X3in inserted by the control according to Embodiment 4.

[0098] In Figure 19, to avoid complexity, only the u-phase current iu is shown for the three-phase output current. Also in Figure 19, the three-phase modulation insertion period X3in is shown by a thick solid rectangular frame. In Figure 19, in order to give symmetry to the three-phase modulation insertion period X3in, a three-phase modulation insertion period X3in with an equal phase angle width Δθ, i.e., a phase angle width of 2Δθ, is set before and after the phase angle of 120 degrees, when the u-phase switching pause period Yu in conventional two-phase modulation begins. Similarly, a three-phase modulation insertion period X3in with a predetermined equal phase angle width Δθ, i.e., a phase angle width of 2Δθ, is set before and after the phase angle of 240 degrees, when the u-phase switching pause period Yu in conventional two-phase modulation ends. Figure 19 is an example of insertion for the u-phase switching pause period Yu, but the same insertion is applied to the switching pause periods of the v and w phases. Therefore, looking at each phase (uvw), equal three-phase modulation periods are inserted in which all three phases perform switching operations immediately before and after the timing when the phase that pauses switching operation transitions from the switching period to the switching pause period, and immediately before and after the timing when the phase that has paused switching operation transitions from the switching pause period to the switching period. According to the control of Embodiment 4, since the phase angle width of the three-phase modulation periods inserted immediately before and after the timing when the switching pause period begins, and immediately before and after the timing when the switching pause period ends, is equal for all four, it is possible to suppress fluctuations in the neutral point potential while ensuring the accuracy of Td correction.

[0099] As described above, according to the power converter of Embodiment 4, three-phase modulation periods with equal phase angle widths are inserted at four points: immediately before and immediately after the timing when the switching pause period begins, and immediately before and immediately after the timing when the switching pause period ends. This makes it possible to suppress fluctuations in the neutral point potential while ensuring the accuracy of Td correction. As a result, it is possible to suppress the progression of motor shaft electrolytic corrosion while ensuring the accuracy of Td correction.

[0100] Embodiment 5. As shown in Figures 7 and 8, the Td correction value Vtd described in Embodiment 1 is a voltage corresponding to the magnitude and polarity of the three-phase output current. Therefore, the closer the phase of the three-phase voltage modulated wave and the phase of the Td correction value Vtd (which is synonymous with the phase of the three-phase output current) are to being in phase, the better the symmetry of the three-phase voltage modulated wave is maintained. To more effectively suppress fluctuations in the neutral point potential, a voltage command is generated such that the load power factor is 1 with respect to the three-phase output current.

[0101] As a method for controlling the load power factor to 1, for example, the method described in Japanese Patent Publication No. 10-243700 can be used. Alternatively, if the three-phase load motor 5 is, for example, a surface magnet type motor, a well-known vector control method may also be used. Specifically, the three-phase output currents iu, iv, and iw detected by the method in Figure 1 are transformed into dq-axis currents Id and Iq on a Cartesian coordinate system that rotates synchronously with the rotor of the motor 5, and the d-axis current command Id * and q-axis current command Iq * It generates the following. Furthermore, the dq axis current command Id in each dq axis. * IQ * Based on the difference between the coordinate-transformed detected current values ​​Id and Iq, voltage commands for each axis are generated. At that time, the d-axis current command Id * By assigning zero to the d-axis current Id and controlling it so that it becomes zero, the load power factor can be controlled to 1 more precisely. Therefore, if the motor 5 is a surface magnet type motor, the load power factor can be suitably controlled to 1 by using this type of vector control.

[0102] As described above, according to the power converter of Embodiment 5, the control unit controls the load power factor to approach 1 in accordance with the three-phase output current, which is the output current of the inverter. By controlling the load power factor to approach 1, the phase difference between each phase in the three-phase output current and the three-phase output voltage approaches zero, so that the switching pause period due to two-phase modulation and the period in which the current of each phase becomes large coincide. Since the effect of reducing switching losses is approximately proportional to the magnitude of the three-phase output current, the effect of reducing switching losses can be enhanced in addition to the effects described in Embodiments 1 to 4. Furthermore, in the case of a surface magnet type motor, the load power factor can be strictly controlled to 1 by controlling the d-axis current command to zero, so the effect of reducing switching losses can be further enhanced.

[0103] Embodiment 6. Figure 20 shows an example of the configuration of an air conditioning system 200 according to Embodiment 6. The air conditioning system 200 according to Embodiment 6 comprises a power converter 100 as described in Embodiments 1 to 5, a compressor 50, a fan motor 5b, a fan 52 driven by the fan motor 5b, and a refrigeration cycle 110. The compressor 50 comprises a compressor motor 5a and a compression element 51 that compresses the refrigerant. The compressor motor 5a is the drive source for the compressor 50.

[0104] The power converter 100 has two inverters (not shown) that supply power to the compressor motor 5a, which is the drive source for the compressor 50, and the fan motor 5b, which is the drive source for the fan 52. At least one of the two inverters is the inverter 3 described in Embodiments 1 to 5. The converter 2 described in Embodiments 1 to 5 is intended to output a rectified voltage and may be common to both inverters 3, or it may be provided individually for each of the two inverters 3.

[0105] In the refrigeration cycle 110, the refrigerant circuit is composed of a compressor 50, a four-way valve 121, a heat source side heat exchanger 122, a load side heat exchanger 132, and an expansion device 131. The compressor 50 compresses the refrigerant, the heat source side heat exchanger 122 and the load side heat exchanger 132 perform heat exchange of the refrigerant, and the fan 52 blows air to the heat source side heat exchanger 122. Regarding the components of the refrigeration cycle 110, Figure 20 shows a configuration in which the four-way valve 121 and the heat source side heat exchanger 122 are provided on the outdoor unit 120, and the expansion device 131 and the load side heat exchanger 132 are provided on the indoor unit 130. Note that the configuration in Figure 20 is just one example, and the air conditioning system 200 according to Embodiment 6 is not limited to the configuration in Figure 20.

[0106] By applying any of the power converters 100 according to Embodiments 1 to 5 to the air conditioning system 200 according to Embodiment 6, any of the effects described in Embodiments 1 to 5 can be enjoyed. Specifically, the current ripple of at least one of the motors, the compressor motor 5a which is the driving source for the compressor 50 and the fan motor 5b which is the driving source for the fan 52, is suppressed, thereby suppressing the torque ripple emitted by the motor. As a result, noise from the air conditioning system 200 caused by torque ripple can be suppressed, and losses caused by current ripple can also be suppressed, making it possible to improve the performance of the air conditioning system 200.

[0107] Finally, the hardware configuration for realizing the functions of the control unit 4 described above will be explained with reference to Figures 21 and 22. Figure 21 is a diagram showing an example of a hardware configuration for realizing the functions of the control unit 4 in Embodiments 1 to 5. Figure 22 is a diagram showing another example of a hardware configuration for realizing the functions of the control unit 4 in Embodiments 1 to 5.

[0108] In order to implement some or all of the functions of the control unit 4 in Embodiments 1 to 5, the configuration can include a processor 300 that performs calculations, a memory 302 in which the program read by the processor 300 is stored, and an interface 304 that performs signal input and output, as shown in Figure 21.

[0109] The processor 300 is an example of a computing means. The processor 300 may be a computing means referred to as a microprocessor, microcomputer, CPU (Central Processing Unit), or DSP (Digital Signal Processor). The memory 302 may also include non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable ROM), and EEPROM (Electrically EPROM), as well as magnetic disks, flexible disks, optical disks, compact disks, minidiscs, and DVDs (Digital Versatile Discs).

[0110] Memory 302 stores a program that performs the functions of the control unit 4 in embodiments 1 to 5. The processor 300 can perform the above-described processing by exchanging necessary information via interface 304, executing the program stored in memory 302, and referring to the table stored in memory 302. The calculation results from the processor 300 can be stored in memory 302.

[0111] Furthermore, when implementing some of the functions of the control unit 4 in embodiments 1 to 5, the processing circuit 303 shown in Figure 22 can also be used. The processing circuit 303 may be a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Information input to and output from the processing circuit 303 can be exchanged via the interface 304.

[0112] Alternatively, some of the processing in the control unit 4 may be performed by the processing circuit 303, while the processing not performed by the processing circuit 303 may be performed by the processor 300 and memory 302.

[0113] The configurations shown in the above embodiments are merely examples, and it is possible to combine them with other known technologies, combine different embodiments, and omit or modify parts of the configuration without departing from the gist of the invention.

[0114] For example, in Figures 1 to 3, the commercial power supply 1, which is an AC power supply, can be single-phase or three-phase. The commercial power supply 1 and converter 2 operate as DC power sources that supply DC power to the inverter 3, but a DC power source such as a battery may be used instead of the commercial power supply 1 and converter 2. Also, in Figure 2, the modulation rate command Vk and voltage phase θ input to the control unit 4 are generated based on current feedback control including vector control or feedforward control, which are well known in higher-level control systems. Furthermore, although embodiments 1 to 5 were described assuming a bottom-sticking two-phase modulation scheme, the same can be achieved using an top-sticking two-phase modulation scheme in which the upper limit is set to "1" and stuck to the upper limit side. [Explanation of Symbols]

[0115] 1 Commercial power supply, 2 Converter, 3 Inverter, 4 Control unit, 5 Motor, 5a Compressor motor, 5b Fan motor, 6a, 6b, 7 Electrical wiring, 31a~31c, 32a~32c Switching elements, 33a~33c, 34 Shunt resistor, 35a, 35b Current detector, 41, 41A Modulation method selection unit, 42 Modulated wave generation unit, 43 Td correction unit, 44 PWM modulation unit, 45 Td addition unit, 50 Compressor, 51 Compression element, 52 Fan, 100 Power converter, 110 Refrigeration cycle, 120 Outdoor unit, 121 Four-way valve, 122 Heat source side heat exchanger, 130 Indoor unit, 131 Expansion device, 132 Load side heat exchanger, 200 Air conditioning unit, 300 Processor, 302 Memory, 303 Processing circuit, 304 Interface.

Claims

1. An inverter that converts DC power to AC power and supplies it to a three-phase load, A control unit that generates switching signals for a plurality of three-phase switching elements provided in the inverter and outputs them to the inverter, Equipped with, The control unit performs two-phase modulation, which sequentially pauses the switching operation of the switching elements of one of the three phases, and, based on the current flowing into and out of the inverter, inserts a three-phase modulation period in which all three phases perform switching operations at at least one of the timings immediately before and immediately after the transition of the phase whose switching operation is paused from the switching period to the switching pause period, and at least one of the timings immediately before and immediately after the transition of the phase whose switching operation has been paused from the switching pause period to the switching period. Power converter.

2. The control unit inserts the three-phase modulation period immediately after the timing when the phase whose switching operation is suspended transitions from the switching period to the switching suspension period, and immediately after the timing when the phase whose switching operation has been suspended transitions from the switching suspension period to the switching period, if the three-phase output current, which is the output current of the inverter, is in phase leading the three-phase output voltage, which is the output voltage of the inverter. The power conversion device according to claim 1.

3. The control unit inserts the three-phase modulation period immediately before the phase in which the switching operation is paused transitions from the switching period to the switching pause period, and immediately before the phase in which the switching operation has been paused transitions from the switching pause period to the switching period, if the three-phase output current, which is the output current of the inverter, is out of phase with respect to the three-phase output voltage, which is the output voltage of the inverter. The power conversion device according to claim 1.

4. The control unit performs two-phase modulation, which sequentially pauses the switching operation of the switching elements of one of the three phases, and, based on the current flowing into and out of the inverter, inserts a three-phase modulation period in which all three phases perform switching operation immediately before and after the timing when the phase whose switching operation is paused transitions from the switching period to the switching pause period, and immediately before and after the timing when the phase whose switching operation has been paused transitions from the switching pause period to the switching period. The power conversion device according to claim 1.

5. The control unit controls the timing and duration of the three-phase modulation period based on at least one of the following: the current value of the three-phase output current of the inverter, the phase difference between the three-phase output voltage of the inverter and the three-phase output current, a voltage correction value for correcting errors caused by dead time, and the modulation rate of the three-phase output voltage. The power conversion device according to claim 4.

6. Three-phase modulation periods with equal phase angle widths are inserted at four points: immediately before and immediately after the timing when the switching pause period begins, and immediately before and immediately after the timing when the switching pause period ends. The power conversion device according to claim 4.

7. The control unit generates a first three-phase voltage modulated wave based on a voltage command output from a higher-level control system, calculates a common three-phase signal for setting the switching pause period while maintaining the line-to-line voltage values ​​for the generated first three-phase voltage modulated wave, calculates a second three-phase voltage modulated wave by superimposing the common three-phase signal onto the first three-phase voltage modulated wave, and further generates the switching signal based on a third three-phase voltage modulated wave obtained by correcting the error caused by the dead time applied to the switching signal for the second three-phase voltage modulated wave. The power conversion device according to claim 1.

8. The control unit calculates the three-phase common signal based on a first difference, which is the difference between the first three-phase voltage modulated wave that is close to a predetermined first limiter value and the first limiter value, and when performing the process of inserting the three-phase modulation period, it calculates the three-phase common signal based on a second difference, which is the difference between the first three-phase voltage modulated wave that is close to the first limiter value and a second limiter value that has a smaller absolute value than the first limiter value. The power conversion device according to claim 7.

9. The third difference, which is the difference between the first limiter value and the second limiter value, is determined based on at least one of the following: the effective value of the three-phase output current, which is the output current of the inverter, or the phase difference between the three-phase output voltage, which is the output voltage of the inverter, and the three-phase output current. The power conversion device according to claim 8.

10. The aforementioned three-phase load is a motor, The control unit performs three-phase modulation without performing two-phase modulation under operating conditions where the motor's rotational speed falls below a predetermined threshold. The power conversion device according to claim 1.

11. The aforementioned three-phase load is a motor, Under operating conditions where the motor's rotational speed falls below a predetermined threshold, the control unit determines the value of the third difference based on the value of the disturbance voltage caused by the dead time, thereby creating a three-phase modulation period in which each phase switches for the entire duration. The power conversion device according to claim 9.

12. The aforementioned three-phase load is a motor, The control unit does not insert the three-phase modulation period under operating conditions where the motor's rotational speed exceeds a predetermined threshold. The power conversion device according to claim 1.

13. The aforementioned three-phase load is a motor, The control unit, under operating conditions where the motor's rotational speed exceeds a predetermined threshold, sets the third difference to zero, thereby preventing the insertion of a three-phase modulation period for the entire duration. The power conversion device according to claim 9.

14. The control unit controls the load power factor to approach 1 according to the output current of the inverter. The power conversion device according to claim 1.

15. The aforementioned three-phase load is a surface magnet type motor, The control unit generates a current command that results in a load power factor of 1 through vector control based on the output current of the inverter. The power conversion device according to claim 14.

16. A power conversion device according to any one of claims 1 to 15, A compressor that compresses the refrigerant, A heat exchanger that performs heat exchange of the aforementioned refrigerant, A fan that blows air to the aforementioned heat exchanger, At least one of the motors that drives the compressor and the motor that drives the fan is driven by the power converter. Air conditioning system.

Images