Power conversion devices, motor drives and refrigeration cycle application equipment

Abstract

The power conversion device (1) includes a converter (100), a capacitor (200), and an inverter (300), as well as a control device (3) that drives and controls the converter (100) and the inverter (300). The converter (100) and the inverter (300) each include at least one current detection unit (301a, 301b) for detecting a converter current flowing through the converter (100) and an inverter current flowing through the inverter (300). The control device (3) includes a capacitor current estimation unit (700) that uses detection values ​​of the converter current and the inverter current detected by the current detection units (301a, 301b) and estimates the capacitor current by switching calculation processing depending on the operating state of the converter (100) and using the capacitor current flowing through the capacitor (200) as a calculation object.

Description

[Technical field]

[0001] The present disclosure relates to a power conversion device that converts AC power into desired power, a motor drive device, and a refrigeration cycle application device. [Background technology]

[0002] Conventionally, in a power conversion device that converts an AC voltage from an AC power source into a DC voltage and further converts the DC voltage into a desired AC power, a smoothing capacitor is sometimes provided to reduce fluctuations in the ripple voltage of the DC voltage. A ripple current corresponding to the ripple voltage of the DC voltage flows through this capacitor.

[0003] The ripple current flowing through a smoothing capacitor causes the capacitor to heat up. Furthermore, if an excessive ripple current continues to flow through a smoothing capacitor, the capacitor may break down or its lifespan may deteriorate. One measure to reduce the ripple current in a capacitor is to increase the capacity of the capacitor. However, increasing the capacity of a capacitor increases the cost and weight of the device. Suppressing increases in the cost and weight of the device are common challenges that are always required of power conversion devices, regardless of the application.

[0004] Conventionally, various control techniques for reducing the ripple current of a capacitor have been proposed. For example, there is a technique for appropriately controlling the motor current output from an inverter so as to suppress the pulsation of the current flowing in and out of the capacitor. However, in any of these types of control, when the control for reducing the ripple current of the capacitor does not function normally, an excessively large ripple current continues to flow through the capacitor, making it difficult to avoid the breakdown or deterioration of the capacitor's life. Therefore, it is important to grasp the ripple current flowing through the capacitor during the operation of the power conversion device. For example, the following Patent Document 1 is a document disclosing a technique for estimating the capacitor current flowing through a capacitor. This Patent Document 1 discloses a technique for estimating the capacitor current using a detected value of a DC voltage and a capacitor capacitance value. [Prior art documents] [Patent documents]

[0005] [Patent Document 1] JP 2005-295712 A Summary of the Invention [Problem to be solved by the invention]

[0006] However, in the technology of Patent Document 1, the capacitor capacitance value used to estimate the capacitor current has errors due to individual differences and changes over time with respect to the nominal value, so there is a problem in that errors also occur in the estimated capacitor current.

[0007] The present disclosure has been made in consideration of the above, and aims to provide a power conversion device that can accurately estimate capacitor current even when the capacitor capacitance value has individual differences or changes over time from the nominal value. [Means for solving the problem]

[0008] In order to solve the above-mentioned problems and achieve the object, a power conversion device according to the present disclosure includes a converter, a capacitor, and an inverter, and a control device that drives and controls the converter and the inverter. The converter includes at least one switch element and converts a first AC voltage applied from an AC power source into a DC voltage. The capacitor smoothes the voltage output by the converter. The inverter converts the DC voltage smoothed by the capacitor into a second AC voltage, and applies the converted second AC voltage to a motor to rotate it. The converter and the inverter each include at least one current detection unit to detect a converter current flowing through the converter and an inverter current flowing through the inverter. The control device drives and controls the converter and the inverter, respectively. The control device also includes a capacitor current estimation unit that uses the detection values ​​of the converter current and the inverter current detected by the current detection unit, and performs an estimation calculation of the capacitor current by switching calculation processing depending on the operating state of the converter, with the capacitor current flowing through the capacitor as the calculation target. Effect of the Invention

[0009] According to the power conversion device according to the present disclosure, it is possible to achieve an effect of being able to accurately estimate the capacitor current even when the capacitor capacitance value has individual differences or changes over time with respect to the nominal value. [Brief description of the drawings]

[0010] [Figure 1] FIG. 1 is a diagram showing an example of the configuration of a power conversion device and a motor drive device according to a first embodiment; [Diagram 2] FIG. 2 is a diagram for explaining the operation of a converter provided in the power conversion device according to the first embodiment; [Diagram 3] FIG. 1 is a diagram for explaining the operation of a converter control unit and a converter according to the first embodiment; [Figure 4] FIG. 1 is a block diagram showing a configuration example of an inverter control unit provided in a control device according to a first embodiment. [Diagram 5] FIG. 1 is a diagram showing a configuration example of an inverter according to a first embodiment; [Figure 6] FIG. 1 is a diagram showing eight switching patterns in an inverter according to a first embodiment. [Figure 7] 1 is a time chart for explaining the operation of an inverter and an inverter control unit according to the first embodiment; [Figure 8] FIG. 1 is a block diagram showing a configuration example of a capacitor current estimation unit provided in a control device according to a first embodiment; [Figure 9] FIG. 1 is a block diagram showing a configuration example of a current effective value calculation unit provided in a capacitor current estimation unit according to a first embodiment; [Figure 10] FIG. 13 is a diagram showing a configuration example of a power conversion device and a motor drive device according to a second embodiment. [Figure 11] FIG. 11 is a diagram for explaining the operation of a converter provided in a power conversion device according to a second embodiment. [Figure 12] FIG. 13 is a diagram showing a configuration example of a power conversion device and a motor drive device according to a third embodiment. [Figure 13] FIG. 11 is a block diagram showing a configuration example of an inverter control unit and a capacitor current estimation unit provided in a control device according to a third embodiment. [Figure 14] 13 is a time chart for explaining the operation of an inverter, an inverter control unit, and a capacitor current estimating unit according to a third embodiment; [Figure 15] FIG. 13 is a diagram showing a configuration example of a power conversion device and a motor drive device according to a fourth embodiment. [Figure 16] FIG. 13 is a diagram showing a configuration example of a power conversion device and a motor drive device according to a fifth embodiment. [Figure 17] FIG. 13 is a diagram showing a configuration example of a refrigeration cycle application device according to a sixth embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] Hereinafter, a power conversion device, a motor drive device, and a refrigeration cycle applied device according to embodiments of the present disclosure will be described in detail with reference to the accompanying drawings.

[0012] Embodiment 1 1 is a diagram showing an example of the configuration of a power conversion device 1 and a motor drive device 2 according to a first embodiment. The power conversion device 1 is connected to a commercial power source 110 and a compressor 400. The power conversion device 1 converts a first AC power supplied from the commercial power source 110 into a second AC power having a desired amplitude and phase, and supplies the second AC power to the compressor 400. The commercial power source 110 may be single-phase or three-phase. In this paper, a case where the commercial power source 110 is single-phase will be described.

[0013] A compressor 400 shown in Fig. 1 is an example of a mechanical device to be driven. The compressor 400 includes a motor 401 driven by the power conversion device 1 and a mechanical load 402. The motor 401 drives the mechanical load 402, and the mechanical load 402 performs mechanical work. In this paper, the compressor 400 is described as an example, but the present invention is also applicable to mechanical devices other than the compressor 400. A motor drive device 2 according to the first embodiment is configured by the power conversion device 1 and the motor 401 provided in the compressor 400.

[0014] In the compressor 400, a periodic load torque changes when the refrigerant is sucked, compressed, and discharged. When the speed of the motor 401 is approximately constant, the load torque in the compressor 400 has a periodic pulsating waveform. Note that the mechanical structure of the compressor 400 for compressing the refrigerant is not particularly limited. The compressor 400 may be a rotary compressor, a reciprocating compressor, a scroll compressor, or a screw compressor. The compressor 400 may also be a compressor other than these.

[0015] 1, the power conversion device 1 includes a converter 100, a capacitor 200, an inverter 300, current detection units 301a and 301b, voltage detection units 201a and 201b, and a control device 3. The control device 3 controls the driving of the converter 100 and the inverter 300 based on at least one of the detection values ​​detected by the current detection units 301a and 301b and the voltage detection units 201a and 201b.

[0016] Converter 100 is connected between commercial power supply 110 and capacitor 200. Converter 100 is a power converter that includes at least one switch element and converts a first AC voltage applied from commercial power supply 110, which is an AC power supply, into a DC voltage and outputs the DC voltage. Capacitor 200, voltage detection unit 201b, and inverter 300 are connected to the output terminal of converter 100.

[0017] When obtaining DC voltage from an AC power source, it is common to use a power factor correction circuit, which has the functions of controlling the AC current so as to comply with harmonic standards and suppressing fluctuations in the output DC bus voltage.

[0018] The converter 100 shown in FIG. 1 is configured with a power factor correction circuit called a simplified switching circuit. Specifically, the converter 100 includes a reactor 120 installed on the input side of a commercial power source 110, a diode rectifier 130, and a simplified switching cell 160 in which a switch element 150 is connected to two diodes 140a and 140b. A current detection unit 301a and voltage detection units 201a and 201b can be used to detect the current and voltage in the converter 100 as shown in FIG. 1. When open loop control is applied, the current detection unit 301a and the voltage detection units 201a and 201b as shown in FIG. 1 do not need to be used. Note that, although the switch element 150 is represented by the symbol of an IGBT (Insulated Gate Bipolar Transistor), a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) may also be used, and the switch element 150 is not limited to an IGBT. 1 shows an example of a circuit configuration including a simple switching cell 160 as the converter 100 using a power factor correction circuit, but other circuit configurations may be used. Examples of other power factor correction circuits will be described in the embodiments described later.

[0019] The current detection unit 301a is installed at the lower output end of the diode rectifier 130 as shown in FIG. 1. The lower output end of the diode rectifier 130 is the side where the current due to the power rectified by the diode rectifier 130 is fed back to the commercial power source 110. The current detection unit 301a detects the converter current Iconv flowing through this portion and outputs the detected current value to the control device 3. The installation position of the current detection unit 301a is not limited to the position shown in FIG. 1, and may be installed at any position as long as the converter current Iconv can be detected or estimated. For example, instead of installing the current detection unit 301a at the lower output end of the diode rectifier 130, the current detection unit 301a may be installed at a position where the input AC current Iac flowing through the reactor 120 is detected. In this case, by using the relationship of Iconv=|Iac|, the converter current Iconv can be estimated and used for control calculation in the control device 3.

[0020] Capacitor 200 is a smoothing capacitor that smoothes the voltage output by converter 100. Examples of capacitor 200 include an electrolytic capacitor and a film capacitor. Capacitor 200 only needs to have a capacity that can smooth the power rectified by converter 100 to some extent.

[0021] The inverter 300 is connected to the capacitor 200 and the motor 401. The inverter 300 converts the DC voltage smoothed by the capacitor 200 into a second AC voltage by switching an internal switch element, and applies the converted second AC voltage to the motor 401 to drive the motor 401 to rotate. The second AC voltage is an AC voltage having a desired amplitude, frequency, and phase. The motor 401 rotates at a desired rotation speed by the second AC voltage.

[0022] The current detection unit 301b detects the inverter current Iinv fed back from the inverter 300 at least twice per control period of the inverter 300, and outputs the detected current value to the control device 3. The inverter current Iinv is a current flowing in and out of the inverter 300. When the motor 401 is a three-phase Y-connection, a technique for restoring the three-phase currents Iu, Iv, Iw flowing through the motor 401 from the inverter current Iinv is known, and this makes it possible to restore the three-phase currents Iu, Iv, Iw flowing through the motor 401. Of course, a current detector may be installed to directly detect the three-phase currents Iu, Iv, Iw output from the inverter 300, and the detection value may be output to the control device 3.

[0023] The control device 3 receives as input the detection value of the converter current Iconv detected by the current detection unit 301a, the detection value of the inverter current Iinv detected by the current detection unit 301b, and the detection value of the DC bus voltage Vdc detected by the voltage detection unit 201b, and performs various control calculations using a converter control unit 500, an inverter control unit 600, and a capacitor current estimation unit 700, which will be described later, to output each switching signal Q0, Q1 to the converter 100 and the inverter 300, and also performs an estimation calculation to estimate the capacitor current Ic.

[0024] Although not shown, the control device 3 includes a CPU (Central Processing Unit), a ROM (Read Only Memory), a RAM (Random Access Memory), an A / D (Analog Digital) converter, and other electronic circuits for various interfaces. The control device 3 is configured to read out a program stored in the ROM and expand it in the RAM, and the CPU executes various processes. In FIG. 1, one control device 3 is configured to realize the functions of the converter control unit 500, the inverter control unit 600, and the capacitor current estimation unit 700, but each function may be configured as an individual control device, that is, a control device including a CPU, a ROM, a RAM, an A / D converter, and other electronic circuits for various interfaces.

[0025] In a general power conversion device in a refrigeration cycle application device, the converter 100 and the inverter 300 each have at least one current detection unit in order to realize the functions of the converter control unit 500 and the inverter control unit 600 described later or to protect the switch elements of the converter 100 and the inverter 300 from overcurrent. Therefore, if the capacitor current Ic can be estimated using the current detection units provided in the converter 100 and the inverter 300, there is no need to additionally provide a new current detection unit in the capacitor 200. Therefore, in the power conversion device 1 according to the first embodiment, a capacitor current estimation unit 700 that estimates the capacitor current Ic flowing through the capacitor 200 based on the detection values ​​of the converter current Iconv and the inverter current Iinv detected by the current detection units 301a and 301b provided in the power conversion device 1 is configured inside the control device 3. In addition, the method of estimating the capacitor current Ic using the detection values ​​of the current detection units 301a and 301b has the advantage that there is no error due to differential calculation or error due to variation in the capacitor capacitance value compared to the conventional method of estimating it from the voltage detection value.

[0026] Next, control calculations performed in the converter control unit 500, the inverter control unit 600, and the capacitor current estimating unit 700 configured in the control device 3 according to the first embodiment will be described in detail.

[0027] The converter control unit 500 outputs a switching signal Q0 which is an ON / OFF command value for the switch element 150 of the converter 100 by referring to at least the detection value of the AC voltage Vac or its synchronous signal. The calculation method of the converter control unit 500 is, as shown in FIG. 1, a method of inputting the detection values ​​of the DC bus voltage Vdc and the converter current Iconv and determining the switching signal Q0 by feedback control so that the DC bus voltage Vdc and the converter current Iconv become desired values. A representative feedback control method is PID (Proportional Integral Differential) control. Alternatively, a control method of determining the switching signal Q0 by open loop control by referring to a switching pattern stored in advance in the control device 3 without using the voltage detection value and the current detection value is also known, and this known method may be used. In addition, the switching method of the switching signal Q0 may be a high-frequency switching method or a switching method called a simple switching method. The simple switching method is also called a partial switching method, in which switching control is performed on the switch element 150 of the converter 100 only once or a few times per half cycle of the AC power supply.

[0028] Next, the operations of the converter control unit 500 and the converter 100 will be described with reference to several drawings. Fig. 2 is a diagram illustrating the operation of the converter 100 provided in the power conversion device 1 according to the first embodiment.

[0029] 2, a path of a current flowing due to power supplied from the commercial power source 110 is switched by switching control of the switch element 150 provided in the converter 100, that is, by on / off (hereinafter referred to as "ON / OFF") control of the switch element 150. Specifically, when the switch element 150 is controlled to be OFF, the input AC current Iac flowing from the commercial power source 110 is rectified by the diode rectifier 130 via the reactor 120 as shown by the thick dashed line in the figure, and becomes a charging current flowing to the capacitor 200. At this time, no current flows through the simple switching cell 160. When the switch element 150 is controlled to be ON, a short-circuit current flows through the path passing through the simple switching cell 160 as shown by the thick solid line in the figure, and magnetic energy is stored in the reactor 120 via the diode rectifier 130. At this time, the capacitor 200 is not charged by the power of the commercial power source 110. When the switching element 150 is switched from ON to OFF by the switching control of the switching element 150, the magnetic energy stored in the reactor 120 is transferred to the capacitor 200 via the diode rectifier 130 at the same time. In the converter 100 equipped with the power factor correction circuit of the first embodiment, this series of operations makes it possible to widen the conduction angle of the input AC current Iac compared to when the switching control is not performed, i.e., when the switching element 150 is always OFF, and therefore it is possible to improve the power factor to a certain degree. Note that, regardless of the ON / OFF control of the switching element 150, a current caused by the discharge of the capacitor 200 flows to the inverter 300.

[0030] Fig. 3 is a diagram for explaining the operation of the converter control unit 500 and the converter 100 of the first embodiment. The left side of Fig. 3 shows waveforms of the AC voltage Vac of the commercial power source 110, the input AC current Iac, and the switching signal Q0 when the converter control unit 500 and the converter 100 are operated in "(a) simple switching method". The right side of Fig. 3 shows waveforms of the AC voltage Vac, the input AC current Iac, and the switching signal Q0 when the converter control unit 500 and the converter 100 are operated in "(b) high frequency switching method". The lower side of each of Figs. 3(a) and 3(b) shows an example of a switching signal Q0 whose "High" level is "ON" and whose "Low" level is "OFF".

[0031] In the simple switching method, the switch element 150 is ON / OFF controlled once or multiple times per half power cycle by open loop control. This causes an input AC current Iac to flow as shown in FIG. 3(a). Although FIG. 3 shows a case where the switching control is performed only once per half power cycle for the switch element 150, the number of times of switching control may be any number of times. The simple switching method is inferior to the high-frequency switching method in power factor improvement function, but since the number of times of switching is small, it is possible to suppress the switching loss of the switch element 150 to be low. Note that, in the simple switching method, the energy stored in the reactor 120 can be controlled by controlling the short circuit start time Tdl, the short circuit time Ton, and the number of switching times Nsw. The short circuit start time Tdl is the time from the zero cross point of the AC voltage Vac to the first time the switch element 150 is turned ON. The short circuit time Ton is the time during which the switch element 150 is kept ON. The number of switching times Nsw is the number of times per half power cycle that the switch element 150 is ON / OFF controlled. The DC bus voltage Vdc can be steplessly boosted to an arbitrary higher voltage value, as compared to a case where switching control is not performed on the switch element 150.

[0032] The high-frequency switching method is a method of feedback-controlling the DC bus voltage Vdc and the input current Iin. The input current Iin is a current flowing from the converter 100 to the capacitor 200 and the inverter 300. In the high-frequency switching method, as shown in FIG. 3(b), the switching frequency in the simple switching method is set to several kHz or more, and the ON / OFF time of the switch element 150 is controlled so that the DC bus voltage Vdc is a desired value and the input AC current Iac approaches a sine wave synchronized with the AC voltage Vac. In addition, in the high-frequency switching method, as shown in FIG. 3(b), it is possible to make the waveform of the input AC current Iac approach a sine wave synchronized with the AC voltage Vac, so that the power factor can be improved to approximately 1. In addition, the high-frequency switching method has a higher boosting capability than the simple switching method. Therefore, the DC bus voltage Vdc in the high-frequency switching method can be steplessly boosted to an arbitrary voltage value higher than when no switching control is performed on the switch element 150.

[0033] The control method in converter control unit 500 of the first embodiment is not limited to the simple switching method and high-frequency switching method described above, and various known control methods may be used.

[0034] FIG. 4 is a block diagram showing a configuration example of an inverter control unit 600 provided in the control device 3 of the first embodiment. The inverter control unit 600 is configured to include a speed control unit 601, a d-axis current command determination unit 602, a current restoration unit 603, a current control unit 604, a position estimation unit 605, and a PWM (Pulse Width Modulation) signal generator 606. The operation of the inverter control unit 600 will be described below. Note that here, a case where control is performed on a dq rotating coordinate synchronized with the rotor position of the motor 401 will be described, but the dq rotating coordinate is an example, and control may be performed in other coordinate systems. In addition, in order to perform control on the dq rotating coordinate, information on the rotor position of the motor 401 is required, but the rotor position of the motor 401 may be detected by a position sensor (not shown). Alternatively, the position estimation unit 605 may calculate a speed electromotive force using the output voltage of the inverter 300 and the current flowing through the motor 401, and estimate the rotor position using the calculated speed electromotive force.

[0035] A speed control unit 601 outputs a q-axis current command Iq* generated by performing speed control so that the speed command ω* coincides with the estimated speed ωest estimated by a position estimation unit 605. PID control is well-known as a speed control method.

[0036] The d-axis current command determination unit 602 determines a d-axis current command Id* and outputs it to a current control unit 604. Although not shown, the d-axis current command Id* is determined by referring to a three-phase voltage command Vuvw*, a q-axis current command Iq*, a speed command ω*, etc. Typical methods for determining the d-axis current command Id* include a method for determining the d-axis current command Id* so that the motor drive device 2 achieves maximum efficiency driving, and a method for determining the d-axis current command Id* by flux-weakening control so as to suppress voltage saturation of the motor 401.

[0037] The current restoration unit 603 uses the detection value of the inverter current Iinv detected at least twice per control period of the inverter 300 to calculate a three-phase current restoration value Iuvw, which is a restoration value of the three-phase currents Iu, Iv, and Iw of the motor 401. The operation of the current restoration unit 603 will be further described with reference to several drawings. In the following description, the portion between the capacitor 200 and the inverter 300 through which the inverter current Iinv flows is referred to as the "DC bus portion."

[0038] Fig. 5 is a diagram showing a configuration example of the inverter 300 according to the first embodiment. In Fig. 5, the inverter 300 includes six switch elements Sup, Sun, Svp, Svn, Swp, and Swn. In this document, the switch elements Sup, Svp, and Swp are appropriately referred to as "upper arm switch elements" or simply "upper arms," ​​and the switch elements Sun, Svn, and Swn are appropriately referred to as "lower arm switch elements" or simply "lower arms."

[0039] As described above, the current restoration unit 603 calculates the three-phase current restoration value Iuvw using the detected value of the inverter current Iinv, and in so doing, uses information about the switching states of the six switch elements Sup, Sun, Svp, Svn, Swp, and Swn of the inverter 300.

[0040] 6 is a diagram showing eight switching patterns in the inverter 300 of the first embodiment. The switching patterns are patterns of switching states in the inverter 300. The six switch elements Sup, Sun, Svp, Svn, Swp, and Swn of the inverter 300 are switched and controlled by the inverter control unit 600 with switching patterns of ON / OFF control with different combinations. These switching patterns can be classified into eight switching patterns (a) to (h) as shown in FIG. 6. In this case, the relationship between the three-phase currents Iu, Iv, and Iw and the inverter current Iinv is as shown in (a) to (h) in FIG. 6, respectively.

[0041] First, let us consider "(a) all upper arms OFF" and "(h) all upper arms ON". In either of these cases, no power is supplied from the DC bus, and the three-phase currents Iu, Iv, and Iw flowing through the motor 401 circulate through the motor 401 and the upper arms or lower arms of the inverter 300. Therefore, no current flows through the DC bus, and Iinv=0.

[0042] Next, let us look at "(b) Only the U-phase upper arm is ON." In this case, the power supplied from the DC bus is supplied to the motor 401 via the U-phase upper arm, and is returned to the DC bus via the V-phase lower arm and the W-phase lower arm. Therefore, Iinv=-Iv-Iw, and from the relationship Iu+Iv+Iw=0 according to Kirchhoff's law, Iinv=Iu.

[0043] For the other switching patterns (c) to (g), the relationship between the three-phase currents Iu, Iv, Iw and the inverter current Iinv can be derived in the same manner as in (b). The current restoration unit 603 restores the three-phase currents Iu, Iv, Iw from the inverter current Iinv by using the relationship between the three-phase currents Iu, Iv, Iw and the inverter current Iinv shown in FIG.

[0044] FIG. 6 shows that the three-phase currents Iu, Iv, and Iw can be restored from the inverter current Iinv. On the other hand, only one of the three-phase currents Iu, Iv, and Iw can be restored from one detection value of the inverter current Iinv. In order to realize the control of the inverter control unit 600, all values ​​of the three-phase currents Iu, Iv, and Iw are required for each control period. Even if one phase is calculated from Kirchhoff's law, it is necessary to detect the currents of at least the remaining two phases. That is, in order to realize the control of the inverter control unit 600, it is necessary to detect the inverter current Iinv at least twice for each control period of the inverter control unit 600. In addition, when the switching state of the inverter 300 is "(a) all upper arms are OFF" and "(h) all upper arms are ON", current detection is not possible. Therefore, current detection must be performed at least twice when the switching state is any one of (b) to (g), and the currents that can be detected at that time must be different phase currents.

[0045] Fig. 7 is a time chart for explaining the operation of the inverter 300 and the inverter control unit 600 according to the first embodiment. Fig. 7 shows phase voltage commands Vu*, Vv*, Vw* given to each phase of the inverter 300, switching signals given to the switch elements Sup, Svp, Swp of the upper arm of the inverter 300, the inverter current Iinv, the timing of detection of the inverter current Iinv, and the like.

[0046] Fig. 7 shows an example in which the magnitude relationship between the phase voltage commands Vu*, Vv*, Vw* of each phase is Vu*>Vv*>Vw*. The switching signals of the upper arm switch elements are determined by triangular wave comparison PMWs for the respective phase voltage commands Vu*, Vv*, Vw*, and are as shown in the second to fourth rows of Fig. 7. As a result, the switching pattern of the inverter 300 is as shown in the bottom row.

[0047] Black dots are shown on the waveform of the inverter current Iinv in Fig. 7, and these dots indicate the timing for detecting the inverter current Iinv. As shown in Fig. 7, by detecting the inverter current Iinv when the switching patterns during the carrier period are (b) and (c), it becomes possible to detect the U-phase current Iu and the W-phase current Iw. Furthermore, if the U-phase current Iu and the W-phase current Iw can be detected, then it becomes possible to detect the V-phase current Iv according to Kirchhoff's law.

[0048] Further focusing on Fig. 7, the detection timings in (b) and (c) of the switching patterns for detecting the inverter current Iinv are included in the period in which the carrier wave is monotonically decreasing. That is, in the example of Fig. 7, the inverter current Iinv is detected once each when the switching patterns included in the half carrier cycle during which the carrier wave is monotonically decreasing are (b) and (c). This makes it possible to restore all of the three-phase currents Iu, Iv, and Iw for each control cycle of the inverter control unit 600 based on the detection value of the inverter current Iinv.

[0049] In FIG. 7, the switching patterns (b) and (c) can be rephrased as the times before and after the switching state of the V-phase switching element Svp is switched, as can be seen from the waveform of the V-phase switching element Svp in the third stage. Therefore, the detection timing of the inverter current Iinv can be rephrased as the detection being performed once before and once after the switching state of the switching element Svp is switched. Here, FIG. 7 shows a case where the relationship between the phase voltage commands Vu*, Vv*, and Vw* of each phase is Vu*>Vv*>Vw*, and the phase voltage command Vv* is an intermediate phase that is the middle of the phase voltage commands Vu*, Vv*, and Vw* of each phase when they are arranged in order of magnitude. Therefore, it can be generalized that the detection of the inverter current Iinv is performed once before and once after the switching state of the upper arm switching element of the intermediate phase is switched.

[0050] As described above, the current restoration unit 603 detects the inverter current Iinv once each before and after the switching state of the intermediate phase is switched for each carrier down half cycle in which the carrier wave monotonically decreases, thereby restoring the current values ​​of two phases out of the three-phase currents Iu, Iv, and Iw. The current value of the remaining one phase is calculated by Kirchhoff's law to restore the three-phase currents Iu, Iv, and Iw from the detection value of the inverter current Iinv. Note that the example shown here is merely an example, and values ​​detected three or more times for each carrier cycle may be used. For example, although detection is performed twice for each carrier down half cycle in FIG. 7, detection may be performed twice for each carrier down half cycle and carrier up half cycle, and the average value of the detection values ​​in each half cycle may be used. In addition, when the switching pattern is any of (b) to (g), current detection may be performed multiple times for each, and the average value may be used to perform current restoration. In these cases, although the amount of calculation increases, the change in the phase current during one carrier cycle can be averaged to some extent to restore the current.

[0051] A current control unit 604 performs current control based on the d-axis current command Id*, the q-axis current command Iq*, the three-phase current restored value Iuvw, and the estimated position θest, and outputs a three-phase voltage command Vuvw*. As a method of current control, PID control is well known, as is the case with speed control.

[0052] The position estimation unit 605 calculates a speed electromotive force based on the three-phase voltage command Vuvw* and the three-phase current restored value Iuvw, and estimates the rotor position and speed based on the calculated speed electromotive force. Various methods have been proposed for position sensorless control of the motor 401, and for example, sensorless vector control using an adaptive observer is well known.

[0053] The PWM signal generator 606 generates a PWM signal based on the three-phase voltage command Vuvw* and the DC bus voltage Vdc. The generated PWM signal is output to the inverter 300 as a switching signal Q1 to the inverter 300. Carrier comparison modulation is a well-known method for generating a PWM signal, but other methods may also be used.

[0054] As described above, the inverter control unit 600 outputs a PWM signal as the switching signal Q1 to the inverter 300, and the inverter 300 performs switching operation according to the PWM signal. This enables the motor 401 to operate in accordance with the desired speed command ω*.

[0055] 8 is a block diagram showing an example of the configuration of a capacitor current estimating unit 700 provided in the control device 3 of embodiment 1. The capacitor current estimating unit 700 includes an estimated current calculating unit 710 and a current effective value calculating unit 720.

[0056] The estimated current calculation unit 710 calculates an estimated capacitor current Ic_est based on the detected value of the converter current Iconv, the detected value of the inverter current Iinv, and the switching signal Q0 of the converter.

[0057] In the circuit configuration of the power conversion device 1 shown in FIG. 1, the capacitor current Ic flowing through the capacitor 200 satisfies the relationship Ic=Iin-Iinv according to Kirchhoff's law. Therefore, if the input current Iin is detected in addition to the inverter current Iinv, it is possible to estimate the capacitor current Ic without installing a current detection unit at the part of the electrical wiring to which the capacitor 200 is connected. Here, the current path when the switch element 150 of the converter 100 is OFF and the current path when it is ON, as shown in FIG. 2, are referred to. When the switch element 150 of the converter 100 is ON, the input current Iin is 0. In this case, the capacitor current Ic flowing through the capacitor 200 is Ic=-Iinv. Furthermore, when the switch element 150 of the converter 100 is OFF, the input current Iin coincides with the converter current Iconv, so the capacitor current Ic flowing through the capacitor 200 is Ic=Iconv-Iinv.

[0058] Taking the above relationships into consideration, the estimated current calculation unit 710 calculates the estimated capacitor current Ic_est based on the detected value of the converter current Iconv, the detected value of the inverter current Iinv, and the switching signal Q0 of the converter 100 according to the following equation (1).

[0059]

number

[0060] At least in the power conversion device 1 assumed in embodiment 1, the method of using the detection values ​​of current detection units 301a, 301b installed in the converter 100 and the inverter 300, respectively, and switching the calculation formula depending on the switching state of the switch element 150 of the converter 100 to calculate the estimated capacitor current Ic_est, which is an estimate of the capacitor current Ic, is considered to be a new method that, as far as the inventors know, has not been mentioned in other documents.

[0061] As described above, in the new method according to the first embodiment, by calculating the estimated capacitor current Ic_est, it is possible to estimate the capacitor current Ic without installing an additional new current detection unit at the portion of the electrical wiring that connects the capacitor 200. Furthermore, in the new method according to the first embodiment, as shown in formula (1), the detection values ​​detected by the current detection units 301a and 301b are directly used, and therefore there is an advantage that there is no influence due to differential calculation and errors in the capacitor capacitance value compared to the conventional method of estimating with reference to the DC bus voltage Vdc and the capacitor capacitance value.

[0062] The current effective value calculation unit 720 receives the estimated capacitor current Ic_est and calculates the effective value of the estimated capacitor current Ic_est. Here, the effective value of the estimated capacitor current Ic_est is the current effective value normalized by a reference frequency according to the frequency characteristics of the temperature rise caused by the ripple current of the capacitor 200.

[0063] After a sufficient amount of time has passed since the start of power supply from the commercial power source 110, and in a steady state where the motor 401 is operating at a constant load and speed, the capacitor current Ic becomes an AC current with a DC component, which is the average value of the capacitor current Ic, of 0. Therefore, the magnitude of the capacitor current Ic is determined by the peak value or effective value of the capacitor current Ic.

[0064] When a current flows through the capacitor 200, the capacitor 200 generates heat due to the influence of the internal resistance of the capacitor 200. The internal resistance of the capacitor 200 has a frequency characteristic, so that the lower the frequency, the larger the internal resistance. Therefore, the lower the frequency of the capacitor current Ic, the larger the temperature rise of the capacitor 200. Here, the AC component of the capacitor current Ic includes a plurality of AC components, such as an AC component synchronized with the frequency of the commercial power supply 110 and an AC component synchronized with the fluctuation of the power consumption of the mechanical load 402. Therefore, in order to determine the magnitude of the capacitor current Ic, it is preferable to determine the effective value rather than the peak value. Furthermore, in order to determine the magnitude of the influence on the temperature rise of the capacitor 200, it is preferable to calculate the current effective value normalized by a reference frequency according to the frequency characteristic of the internal resistance of the capacitor 200.

[0065] Based on the above, the current effective value calculation unit 720 performs frequency analysis using the estimated capacitor current Ic_est as an input, and calculates the amplitude value of each frequency component contained in the estimated capacitor current Ic_est. Although the Fast Fourier Transform (FFT) is a well-known frequency analysis method, a frequency analysis method other than FFT may also be used. After performing the frequency analysis, the current effective value calculation unit 720 calculates the amplitude values ​​Ic_f0, Ic_f1, Ic_f2, ..., Ic_f N and frequency correction coefficients K(f0), K(f1), K(f2), …, K(f N ) to calculate the frequency-corrected estimated capacitor current effective value Ic_est_rms according to the following equation (2).

[0066]

number

[0067] In the above formula (2), the frequency correction coefficients K(f0), K(f1), K(f2), …, K(f N ) may be defined as a function of the frequency f. Alternatively, the frequency correction coefficients K(f0), K(f1), K(f2), ..., K(f N ) may be a process that the current effective value calculation section 720 refers to.

[0068] In the above formula (2), when calculating the effective value of the estimated capacitor current Ic_est by adding up each frequency component, the amplitude values ​​Ic_f0, Ic_f1, Ic_f2, ..., Ic_f N and frequency correction coefficients K(f0), K(f1), K(f2), …, K(f N ) to calculate the effective value, and the amplitude values ​​Ic_f0, Ic_f1, Ic_f2, …, Ic_f N In addition, in the frequency analysis described above, the amplitude values ​​of all the frequency components Ic_f0, Ic_f1, Ic_f2, ..., Ic_f N It is not necessary to calculate Ic_est_rms, but it is possible to calculate only the amplitude value of a specific frequency component. By taking these points into consideration, it is possible to calculate the estimated capacitor current effective value Ic_est_rms with a small calculation load. A well-known method for calculating the amplitude value of a specific frequency component is calculation using Fourier series expansion.

[0069] As described above, the AC component of the capacitor current Ic includes various AC components, such as an AC component synchronized with the frequency of the commercial power supply 110 and an AC component synchronized with the fluctuation of the power consumption of the compressor 400. Here, the AC voltage Vac applied to the diode rectifier 130 is full-wave rectified through the diode rectifier 130. As a result, as the AC component synchronized with the frequency of the commercial power supply 110, the current amplitude value of the component twice the frequency of the AC voltage Vac applied from the commercial power supply 110 and its harmonic components becomes large. Therefore, when the current effective value calculation unit 720 calculates the amplitude value of a specific frequency component, the estimated current calculation unit 710 calculates the amplitude value of the component that is an integer multiple of the frequency of the AC voltage Vac contained in the capacitor current Ic. In this way, the current effective value calculation unit 720 can calculate the effective value of the AC component synchronized with the frequency of the AC voltage Vac contained in the capacitor current Ic.

[0070] Next, an example of an AC component synchronized with the fluctuation of the power consumption of the compressor 400 will be described. As described above, in the compressor 400, periodic load torque pulsation occurs when the refrigerant is sucked, compressed, and discharged. This load torque pulsation is generally a load torque pulsation synchronized with the rotation speed of the compressor 400. When the torque generated by the motor 401 is a constant torque, the load torque pulsation causes speed pulsation in the rotation speed of the compressor 400.

[0071] In the prior art, in order to suppress this type of speed pulsation, pulsation suppression control is performed in which the motor torque is pulsated in synchronization with the load torque pulsation. In this pulsation suppression control, the motor torque is pulsated, so that the inverter current Iinv and the motor current, which are currents flowing through the inverter 300 and the motor 401, which are DC loads seen from the capacitor 200, fluctuate in synchronization with the load torque pulsation. In addition, it is expected that the capacitor current Ic also generates current pulsation synchronized with the load torque pulsation, and the current amplitude of the frequency component becomes large. Therefore, it is desirable for the current effective value calculation unit 720 to calculate the amplitude value of the component synchronized with the load torque pulsation, that is, the frequency component that is an integer multiple of the rotation speed of the motor 401, as the amplitude value of the specific frequency component. In this way, the current effective value calculation unit 720 can calculate the effective value of the AC component synchronized with the fluctuation of the current flowing through the inverter 300 and the motor 401, which may be included in the capacitor current Ic.

[0072] In addition, due to interference between an AC component synchronized with the frequency of the commercial power source 110 and an AC component synchronized with fluctuations in the power consumption of the compressor 400, a ripple current due to the sum and difference components of the respective frequency components may occur in the capacitor current Ic. As a specific example of this case, consider a case where the frequency of the commercial power source 110 is 50 Hz and the compressor 400 rotates at a speed of 30 revolutions per second. In this case, it is expected that a ripple current of 130 Hz (=100 Hz+30 Hz) due to the sum component of the double component of the frequency of the commercial power source 110 and the frequency of the rotation speed, and a ripple current of 70 Hz (=100 Hz-30 Hz) due to the difference component between the double component of the frequency of the commercial power source 110 and the frequency of the rotation speed are generated in the capacitor current Ic. Therefore, when the current effective value calculation unit 720 calculates the amplitude value of a specific frequency component, it is desirable to perform the calculation including these frequency components. In this way, the current effective value calculation unit 720 can calculate the effective value of the AC component synchronized with the sum and difference components of the double component of the frequency of the commercial power source 110 and the frequency component of the rotation speed, which may be included in the capacitor current Ic. Note that, although the sum and difference components of the double component of the frequency of the commercial power source 110 and the frequency component of the rotation speed are given as examples here, frequency components that are combinations of these natural number multiple components may also be targeted.

[0073] In addition to the frequency components described above, it is expected that the capacitor current Ic will also contain ripple currents of frequency components synchronized with the switching frequency of the converter 100 and frequency components synchronized with the switching frequency of the inverter 300. Therefore, when the effective current value calculator 720 calculates the amplitude value of a specific frequency component, it is desirable to include these frequency components in the calculation. If there is room for calculation processing, the calculation may further include frequency components that are combinations of natural number multiple components of these frequencies. In this way, the effective current value calculator 720 can calculate the effective values ​​of frequency components synchronized with the switching frequency of the converter 100 and the switching frequency of the inverter 300 that may be included in the capacitor current Ic.

[0074] 9 is a block diagram showing an example of the configuration of the current effective value calculation section 720 provided in the capacitor current estimating section 700 of Embodiment 1. The current effective value calculation section 720 is configured to include Fourier series expansion calculation sections 721a to 721h and a frequency component summing section 722.

[0075] The Fourier series expansion calculation units 721a to 721h input the estimated capacitor current Ic_est and calculate the amplitude value of a specific frequency component included in the estimated capacitor current Ic_est by performing a Fourier series expansion calculation. The Fourier series expansion calculation units 721a and 721b in Fig. 6 calculate and output, as the specific frequency component, an amplitude value Ic_2fac of a double component and an amplitude value Ic_4fac of a quadruple component of the frequency of the commercial power source 110. Moreover, the Fourier series expansion calculation units 721c and 721d calculate and output, as the specific frequency component, an amplitude value Ic_fω of a single component of the rotation speed of the motor 401 and an amplitude value Ic_2fω of a double component. Moreover, the Fourier series expansion calculation units 721e and 721f calculate and output, as the specific frequency components, an amplitude value Ic_2fac+fω of a frequency sum component and an amplitude value Ic_2fac-fω of a frequency difference component between a component twice the frequency of the commercial power source 110 and a component once the rotation speed of the motor 401. Furthermore, the Fourier series expansion calculation units 721g and 721h calculate and output, as the specific frequency components, an amplitude value Ic_2fconv of a component twice the switching frequency of the converter 100 and an amplitude value Ic_2finv of a component twice the switching frequency of the inverter 300.

[0076] A procedure in which the Fourier series expansion calculation unit 721a generates, as a specific frequency component, the amplitude value Ic_2fac of the component twice the frequency of the commercial power supply 110 will be representatively described below.

[0077] First, the Fourier series expansion calculation unit 721a multiplies the estimated capacitor current Ic_est by a sine wave component sin(2fac×t) and a cosine wave component cos(2fac×t) having a frequency twice that of the commercial power supply 110, respectively, in order to detect the double component of the frequency of the commercial power supply 110. The amplitudes of the sine wave component and the cosine wave component corresponding to twice the frequency of the commercial power supply 110 are twice the average values ​​of the sine wave term and the cosine wave term obtained by the multiplication in one cycle. The Fourier series expansion calculation unit 721a calculates the square root of the sum of the squares of the amplitudes of the sine wave component and the cosine wave component corresponding to twice the frequency of the commercial power supply 110, thereby calculating the amplitude value Ic_2fac of the double component of the frequency of the commercial power supply 110 included in the estimated capacitor current Ic_est. The above is the calculation of the Fourier series expansion by the Fourier series expansion calculation unit 721a. The calculations of the Fourier series expansion calculation units 721b to 721h are the same as those of the Fourier series expansion calculation unit 721a except for the frequency to be detected. Furthermore, if the estimated capacitor current Ic_est is a periodic waveform, the output signals of the Fourier series expansion calculation units 721a to 721h are approximately constant.

[0078] The frequency component summation unit 722 refers to each amplitude value calculated in the Fourier series expansion calculation units 721a to 721h and the frequency correction coefficient K(f) corresponding to each frequency, and substitutes them into the above equation (2) to calculate, thereby summing up all the frequency components and calculating the estimated capacitor current effective value Ic_est_rms.

[0079] 9, it is possible to extract only frequency components having a specific high degree of influence from among the frequency components contained in the capacitor current Ic. This makes it possible to calculate the estimated capacitor current effective value Ic_est_rms with high accuracy and with a small calculation load.

[0080] As described above, the power conversion device according to the first embodiment includes a converter and an inverter, and a control device that drives and controls the converter and the inverter. The converter and the inverter each include at least one current detection unit to detect a converter current flowing through the converter and an inverter current flowing through the inverter. The control device includes a capacitor current estimation unit that uses the detection values ​​of the converter current and the inverter current detected by the current detection unit, and performs an estimation calculation of the capacitor current by switching the calculation process depending on the operating state of the converter, with the capacitor current flowing through the capacitor as the calculation target. The power conversion device according to the first embodiment has an effect that the capacitor current can be estimated without installing a current detection unit at a part of the electric wiring to which the capacitor is connected. Furthermore, the power conversion device according to the first embodiment has an advantage that no error due to differential calculation and no error due to the capacitor capacitance value occurs, compared to the conventional technology in which the capacitor current is estimated using the detection value detected by the current detection unit. Furthermore, the power conversion device according to the first embodiment has an effect that the capacitor current can be accurately estimated even if the capacitor capacitance value has individual differences or changes over time with respect to the nominal value.

[0081] In the power conversion device according to the first embodiment, the capacitor current estimation unit provided in the control device performs frequency analysis on the estimated capacitor current to calculate the current amplitudes of a plurality of specific frequency components included in the capacitor current. The specific frequency components are frequency components having a high degree of influence among the frequency components included in the capacitor current. The capacitor current estimation unit normalizes each of the current amplitudes of the calculated plurality of specific frequency components according to the frequency characteristics of the capacitor, and calculates the effective value of the estimated capacitor current using the normalized current amplitudes of the plurality of specific frequency components. According to the power conversion device configured in this manner, when calculating the estimated capacitor current, which is an estimated value of the capacitor current, only the frequency components having a high degree of influence are extracted from the frequency components included in the capacitor current to calculate the capacitor current, so that it is possible to obtain an effect of being able to perform calculation processing with high real-time performance with a small calculation load, and also to obtain an estimated value of the capacitor current with high accuracy.

[0082] In the power conversion device according to the first embodiment, the specific frequencies analyzed by the capacitor current estimation unit may be configured to include at least one of a frequency N times the frequency of the AC voltage applied from the AC power source, a frequency M times the rotation frequency of the mechanical device to be driven, a frequency sum that is the sum of the N-fold frequency and the M-fold frequency, and a frequency difference that is the difference between the N-fold frequency and the M-fold frequency, where N and M are any natural numbers. The components of the capacitor current corresponding to these specific frequencies are frequency components with a high degree of influence. Therefore, by extracting the components of the capacitor current corresponding to these specific frequencies, an effect is obtained that an estimate value of the capacitor current with high accuracy can be obtained.

[0083] In addition, in the power conversion device according to the first embodiment, the specific frequencies analyzed in the capacitor current estimation unit may be configured to include at least one of a frequency K times the switching frequency of the converter and a frequency L times the switching frequency of the inverter, where K and L are any natural numbers. The components of the capacitor current corresponding to these specific frequencies are frequency components with a high degree of influence. Therefore, by extracting the components of the capacitor current corresponding to these specific frequencies, it is possible to obtain an accurate estimated value of the capacitor current.

[0084] Embodiment 2 In the second embodiment, a power conversion device 1a and a motor drive device 2a having a different configuration of a converter 100 from that of the first embodiment will be described. FIG. 10 is a diagram showing a configuration example of a power conversion device 1a and a motor drive device 2a according to the second embodiment. Compared with the configuration shown in FIG. 1, the converter 100 is replaced with a converter 100a in FIG. 10. Also, compared with the configuration shown in FIG. 1, the capacitor current estimation unit 700 provided in the control device 3 is replaced with a capacitor current estimation unit 700a in FIG. The motor drive device 2a according to the second embodiment is composed of a power conversion device 1a and a motor 401 provided in a compressor 400. The other configurations are the same or equivalent to those in FIG. 1, and the same reference numerals are used for the same or equivalent components, and duplicated explanations will be omitted.

[0085] The converter 100a includes a diode rectifier 130, a reactor 120a installed on the output side of the diode rectifier 130, a switch element 150a, and a diode 140c. The current and voltage detection in the converter 100a can be performed using a current detection unit 301a and voltage detection units 201a and 201b as shown in FIG. 10. When open loop control is applied, the current detection unit 301a and the voltage detection units 201a and 201b as shown in FIG. 10 do not need to be used. Although the switch element 150a is represented by the symbol of an IGBT in FIG. 10, a MOSFET may be used, and the switch element 150a is not limited to an IGBT. The ON / OFF of the switch element 150a is switched by a switching signal Q0 output from the converter control unit 500.

[0086] The basic function of the converter 100a of the second embodiment is the same as that of the converter 100 of the first embodiment. That is, the converter 100a of the second embodiment is connected between the commercial power supply 110 and the capacitor 200, converts a first AC voltage applied from the commercial power supply 110 to a DC voltage, and outputs the DC voltage to the capacitor 200 and the inverter 300. However, unlike the first embodiment, in the converter 100a, the reactor 120a is installed on the DC side, not on the AC side. The converter 100a is different from the converter 100 of the first embodiment in that a short-circuit operation caused by an ON operation of the switch element 150a is also performed on the DC side.

[0087] The control device 3 receives as input the detection value of the converter current Iconv detected by the current detection unit 301a, the detection value of the inverter current Iinv detected by the current detection unit 301b, and the detection value of the DC bus voltage Vdc detected by the voltage detection unit 201b, and performs control calculations by the converter control unit 500, the inverter control unit 600, and the capacitor current estimating unit 700, as described in embodiment 1, to output switching signals for the converter 100 and the inverter 300, and also performs an estimation calculation to estimate the capacitor current Ic.

[0088] The converter control unit 500 outputs a switching signal Q0 that is an ON / OFF command value for the switch element 150 of the converter 100. In the power conversion device 1a according to the second embodiment, the switching signal Q0 may be calculated using either the simple switching method or the high-frequency switching method described in the first embodiment, or a known switching method different from these methods may be used.

[0089] FIG. 11 is a diagram for explaining the operation of the converter 100a included in the power conversion device 1a according to the second embodiment. The path of the current flowing due to the power supplied from the commercial power source 110 is switched as shown in FIG. 11 by ON / OFF control of the switch element 150a included in the converter 100a. Specifically, when the switch element 150a is controlled to be OFF, the input AC current Iac flowing from the commercial power source 110 is rectified by the diode rectifier 130 as shown by the thick dashed line in the figure, and becomes a charging current flowing to the capacitor 200 via the reactor 120. When the switch element 150a is controlled to be ON, as shown by the thick solid line in the figure, a short-circuit current flows through the path passing through the switch element 150a, and magnetic energy is stored in the reactor 120 by the rectified current via the diode rectifier 130. At this time, the capacitor 200 is not charged by the power of the commercial power source 110. When the switching element 150a is switched from ON to OFF by the switching control of the switching element 150a, the magnetic energy stored in the reactor 120 is transferred to the capacitor 200 via the diode 140c at the same time. In the converter 100a having the power factor correction circuit of the second embodiment, the series of operations described above makes it possible to widen the conduction angle of the input AC current Iac compared to when the switching control is not performed, that is, when the switching element 150a is always OFF, and therefore it is possible to improve the power factor to a certain degree. Note that, regardless of the ON / OFF control of the switching element 150a, a current caused by the discharge of the capacitor 200 flows to the inverter 300.

[0090] Next, the operation of the capacitor current estimation unit 700 will be described. As in the first embodiment, the capacitor current Ic flowing through the capacitor 200 satisfies the relationship Ic=Iin-Iinv according to Kirchhoff's law. Therefore, if the input current Iin is detected in addition to the inverter current Iinv, it is possible to estimate the capacitor current Ic without installing a current detection unit at the portion of the electric wiring to which the capacitor 200 is connected. Here, the current path when the switch element 150a of the converter 100a is OFF and the current path when it is ON, as shown in FIG. 11, are referred to. When the switch element 150 of the converter 100 is ON, the input current Iin is 0. In this case, the capacitor current Ic flowing through the capacitor 200 is Ic=-Iinv. Furthermore, when the switch element 150 of the converter 100 is OFF, the input current Iin coincides with the converter current Iconv, so the capacitor current Ic flowing through the capacitor 200 is Ic=Iconv-Iinv.

[0091] As explained above, the capacitor current Ic according to ON / OFF of the switch element 150a of the converter 100a of the second embodiment is ultimately the same as the capacitor current Ic according to ON / OFF of the switch element 150 of the converter 100 of the first embodiment, and can be expressed by the above-mentioned formula (1). Therefore, the calculation process in the capacitor current estimator 700 may be the same as that in the first embodiment. Therefore, the same effect as that in the first embodiment can be obtained by using the power conversion device according to the second embodiment.

[0092] Converter 100a shown in FIG. 10 is an example of a converter having equivalent functions to converter 100 shown in FIG. 1. The control described in embodiment 1 can be applied to any converter having equivalent functions, even if the converter has a configuration different from that of FIG. 10.

[0093] Embodiment 3 FIG. 12 is a diagram showing an example of the configuration of a power conversion device 1b and a motor drive device 2b according to the third embodiment. Compared with the configuration shown in FIG. 1, in FIG. 12, the control device 3 is replaced with a control device 3a, and in the control device 3a, the capacitor current estimation unit 700 is replaced with a capacitor current estimation unit 700a. Also, compared with the configuration shown in FIG. 1, in FIG. 12, the switching signal Q1 output from the inverter control unit 600 to the inverter 300 is also input to the capacitor current estimation unit 700a. The motor drive device 2b according to the third embodiment is composed of the power conversion device 1b and a motor 401 provided in the compressor 400. The other configurations are the same or equivalent to those in FIG. 1, and the same reference numerals are attached to the same or equivalent components, and duplicated explanations are omitted.

[0094] Consider the waveforms in the time chart shown in FIG. 7. When the inverter 300 is not operating in overmodulation, that is, when the phase voltage commands Vu*, Vv*, and Vw* of each phase are all equal to or less than Vdc / 2, the switching pattern of the inverter 300 in one carrier period of the carrier wave changes in a total of four states, which are a combination of two of the switching patterns (a) and (h) in which the inverter current Iinv is 0, and the switching patterns (b) to (g) in which the inverter current Iinv matches the absolute value of any of the three-phase currents Iu, Iv, and Iw flowing through the motor 401. At this time, the inverter current Iinv has a pulse-like waveform according to the ON / OFF of the switching operation of each phase, as shown in FIG. 7. Therefore, the inverter current Iinv includes a ripple current consisting of a double component of the carrier frequency of the inverter 300 and its harmonic components. Furthermore, since the capacitor current Ic is expressed by the above formula (1), a ripple current similar to the ripple current of the inverter current Iinv flows in the capacitor current Ic.

[0095] To detect the influence of harmonic components of the ripple current, it is necessary to detect the inverter current Iinv at a cycle that is sufficiently fast with respect to the carrier cycle of the inverter 300 while simultaneously performing the calculation using formula (1). A guide for a cycle that is sufficiently fast is, for example, 1 / 10 or less of the carrier cycle. When performing such a process to speed up the detection cycle, the control device 3a requires an A / D converter, a CPU, and the like capable of high-speed calculations, which leads to increased costs and weight of the control device 3a and the power conversion device 1b.

[0096] There is also a method in which an integrating circuit and an AD conversion port for calculating the average value are used to detect the average value of the inverter current Iinv at every certain period, and the detected average value is used to estimate the capacitor current Ic. With this method, it is not necessary to speed up the detection period, and it is possible to detect the influence of harmonic components due to the carrier frequency. However, even with this method, it is necessary to provide an additional integrating circuit, which leads to increased costs and weight of the device. Also, variations in the constants of the resistors and capacitors that make up the integrating circuit can cause current detection errors, which is undesirable.

[0097] Therefore, the power conversion device 1b according to the third embodiment proposes the following method. First, the capacitor current estimation unit 700a calculates an average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300 by referring to a detection value of the inverter current Iinv detected at least twice for each carrier period of the inverter 300 and a switching signal Q1 to the inverter 300. Then, the capacitor current estimation unit 700a calculates an estimated capacitor current Ic_est for each carrier period of the inverter 300 using the calculated average value Iinv_m of the inverter current Iinv and the converter current Iconv. This method makes it possible to calculate the average value Iinv_m of the inverter current Iinv and estimate the capacitor current Ic without performing current detection at high speed or requiring the provision of an additional circuit for current detection.

[0098] Next, a specific description will be given of the control calculation by the capacitor current estimator 700a according to the embodiment 3. Fig. 13 is a block diagram showing an example of the configuration of the inverter control unit 600 and the capacitor current estimator 700a provided in the control device 3a according to the embodiment 3.

[0099] The configuration of the inverter control unit 600 is the same as that shown in Fig. 4 in the first embodiment, and the calculation processing of each unit is also the same, so the description will be omitted. The difference from the first embodiment is that the switching signal Q1 output from the inverter control unit 600 to the inverter 300 is also input to the capacitor current estimation unit 700a.

[0100] Capacitor current estimator 700a includes an estimated current calculator 710a, an effective current value calculator 720, and an inverter current average value calculator 730. Compared to the configuration shown in Fig. 8, capacitor current estimator 700a shown in Fig. 13 differs in that estimated current calculator 710 is replaced with estimated current calculator 710a, and inverter current average value calculator 730 is added to the front stage of estimated current calculator 710a. Also, while the calculation period of capacitor current estimator 700 is not limited in the first embodiment, capacitor current estimator 700a of the third embodiment differs in that calculation processing is performed in synchronization with the carrier period of inverter 300.

[0101] The inverter current average value calculation unit 730 calculates the average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300 by referring to the detection value of the inverter current Iinv, which is detected at least twice per carrier period of the inverter 300, and the switching signal Q1 of the inverter 300.

[0102] The operation of the inverter current average value calculation unit 730 will be further described with reference to several drawings. Fig. 14 is a time chart for explaining the operation of the inverter 300, the inverter control unit 600, and the capacitor current estimation unit 700a according to the third embodiment. Fig. 14 shows phase voltage commands Vu*, Vv*, Vw* given to each phase of the inverter 300, switching signals given to the switching elements Sup, Svp, Swp of the upper arm of the inverter 300, the inverter current Iinv, and timing for detecting the inverter current Iinv. In Fig. 14, the parts related to the operation of the inverter 300 and the inverter control unit 600 are the same as those in Fig. 7.

[0103] In the first embodiment, with reference to Fig. 6, it has been shown that depending on the switching state of the inverter 300, the inverter current Iinv may be 0 or may match the absolute value of any one of the three-phase currents Iu, Iv, and Iw of the motor 401. Also, in the first embodiment, with reference to Fig. 7, it has been shown that the inverter current Iinv matches the absolute value of any one of the three-phase currents Iu, Iv, and Iw of the motor 401 before and after the switching state of the upper arm switch element of the intermediate phase, which is the middle one when the phase voltage commands Vu*, Vv*, and Vw* of each phase are arranged in order of magnitude, is changed. This is also true in the operating state of Fig. 14. Therefore, as an operation of the inverter current average value calculation unit 730, to calculate the average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300, in FIG. 14, the switching state durations Ti, Tj before and after the switching state of the upper arm switch element of the middle phase is switched, the detection values ​​Iinv1, Iinv2 of the inverter current Iinv detected during each period, and one carrier period Tc of the carrier wave are used to perform calculation according to the following equation (3).

[0104]

number

[0105] In this paper, of the switching state durations Ti and Tj, the switching state duration Ti may be referred to as the "first switching state duration" and the switching state duration Tj may be referred to as the "second switching state duration". When the carrier frequency of the inverter 300 is not changed, one carrier period Tc is a fixed value. Therefore, the elements that vary in the formula (3) are the switching state durations Ti and Tj and the detection values ​​Iinv1 and Iinv2 of the inverter current Iinv. Therefore, it can be said that the average value Iinv_m of the inverter current Iinv can be calculated based on the product of the switching state duration Ti and the detection value Iinv1 of the inverter current Iinv, and the product of the switching state duration Tj and the detection value Iinv2 of the inverter current Iinv.

[0106] As can be seen from Fig. 14, the switching state durations Ti, Tj can be calculated using the ON times Tup, Tvp, Twp of the upper arm switch elements of each phase during one carrier period. That is, they can be calculated by referring to the switching signal Q1 calculated by the inverter control unit 600. In Fig. 14, since the V phase is the intermediate phase, Ti = (Tup - Tvp) / 2 and Tj = (Tvp - Twp) / 2. In the case of other combinations in which the intermediate phase is the U phase or W phase, the calculations can be made in a similar manner.

[0107] In addition, the detection values ​​Iinv1 and Iinv2 of the inverter current Iinv detected before and after the upper arm switch element of the middle phase used in the formula (3) is switched may be the same as the detection values ​​used for restoring the three-phase currents Iu, Iv, and Iw of the motor 401 in the current restoration unit 603 of the inverter control unit 600. Therefore, in the capacitor current estimating unit 700a, it is not necessary to perform additional current detection at different timings to estimate the capacitor current Ic, and it is sufficient to commonly use the detection values ​​detected for restoring the three-phase currents Iu, Iv, and Iw. This makes it possible to calculate the average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300 and calculate the estimated capacitor current Ic_est without increasing the calculation load of the A / D conversion.

[0108] Alternatively, in order to improve the accuracy of the current detection, the inverter current Iinv may be detected multiple times before and after the upper arm switch element of the middle phase is switched, and the average values ​​of the detected inverter currents may be set as Iinv1 and Iinv2 to calculate the average value Iinv_m of the inverter current Iinv. In FIG. 14, for the sake of simplicity, the inverter current Iinv is shown to be constant during each of the switching state durations Ti and Tj. However, in reality, a current change occurs during each of the switching state durations Ti and Tj in response to a change in the phase current of the motor 401. On the other hand, by detecting the current multiple times before and after the switching, the current change during the switching state durations Ti and Tj can be taken into account, thereby improving the accuracy of the current detection. Since the calculation load of the A / D conversion increases due to the multiple current detections, there is a trade-off between the improvement of the accuracy of the current detection and the increase in the calculation load, and it is only necessary to determine whether or not to perform the current detection multiple times depending on which one is to be prioritized.

[0109] 14 shows an example in which current detection is performed in a carrier down cycle, which is a period in which the carrier wave monotonically decreases, to calculate the average value Iinv_m of the inverter current Iinv, but the present invention is not limited to this example. Current detection may be performed in a carrier up cycle, which is a period in which the carrier wave monotonically increases, to calculate the average value Iinv_m of the inverter current Iinv. Current detection may also be performed in both the carrier down cycle and the carrier up cycle, and the average value of the current in the period before and after switching in the carrier down cycle and the average value of the current in the period before and after switching in the carrier up cycle may be further averaged to calculate the average value Iinv_m of the inverter current Iinv.

[0110] As described above, in the method of embodiment 3, the inverter current average value calculation unit 730 calculates the average value Iinv_m of the inverter current Iinv for each carrier period by referring to the detection value of the inverter current Iinv detected at least twice per carrier period of the inverter 300 and the switching signal Q1 to the inverter 300.

[0111] Estimated current calculation unit 710a calculates an estimated capacitor current Ic_est by referring to the detection value of converter current Iconv detected in synchronization with the carrier cycle of inverter 300, average value Iinv_m of inverter current Iinv calculated by inverter current average value calculation unit 730, and switching signal Q0 of converter 100. Note that the method of calculating estimated capacitor current Ic_est differs only in that inverter current Iinv in equation (1) shown in embodiment 1 is replaced with average value Iinv_m of inverter current Iinv for calculation, and is the same in that the calculation equation is switched according to switching signal Q0 of converter 100.

[0112] The current effective value calculation unit 720 receives the estimated capacitor current Ic_est as input and calculates the estimated capacitor current effective value Ic_est_rms for each carrier period of the inverter 300. The calculation contents of the current effective value calculation unit 720 are similar to those of the current effective value calculation unit 720 described in the first embodiment.

[0113] As described above, in the power conversion device according to the third embodiment, the inverter includes a plurality of switch elements whose switching is controlled by a phase voltage command generated by the control device and given to each phase of the inverter. When a phase in which the magnitude relationship of the phase voltage commands for the plurality of switch elements is intermediate is defined as an intermediate phase, a capacitor current estimation unit included in the control device calculates an average value of the inverter current for each switching period of the inverter based on a product of a first switching state duration before the switching state of the switch element of the intermediate phase is switched and a detection value of the inverter current detected during the first switching state duration, and a product of a second switching state duration after the switching state of the switch element of the intermediate phase is switched and a detection value of the inverter current detected during the second switching state duration. According to the power conversion device according to the third embodiment, it is possible to enjoy the same effect as the power conversion device according to the first embodiment. In addition, by using the method of the third embodiment, it is possible to perform accurate current detection, and an effect is obtained that it can be realized without providing a new detector and additional circuit for current detection.

[0114] Embodiment 4 Fig. 15 is a diagram showing an example of the configuration of a power conversion device 1c and a motor drive device 2c according to embodiment 4. Compared with the configuration shown in Fig. 1, in Fig. 15, the control device 3 is replaced with a control device 3b, and an alarm signal generating unit 750 is added to the control device 3b. The motor drive device 2c according to embodiment 4 is composed of the power conversion device 1c and a motor 401 provided in a compressor 400. The other configurations are the same or equivalent to those in Fig. 1, and the same reference numerals are used to denote the same or equivalent components, and duplicated explanations will be omitted.

[0115] The alarm signal generating unit 750 outputs an alarm signal ALM when a specific condition occurs based on the estimated capacitor current effective value Ic_est_rms. Here, the specific condition is, for example, when the estimated capacitor current effective value Ic_est_rms becomes larger than the capacitor current limit value Ic_lim stored in advance in the control device 3b.

[0116] Even in a general power conversion device, an allowable current is determined for each capacitor. If a capacitor is operated for a long time while a current larger than the allowable current continues to flow through it, the capacitor may break down or the life of the capacitor may rapidly deteriorate. Therefore, in the power conversion device 1c according to the fourth embodiment, a capacitor current limit value Ic_lim, which is a limit value of the capacitor current set in advance, is stored in the control device 3b. The control device 3b constantly monitors the estimated capacitor current effective value Ic_est_rms calculated during the operation of the power conversion device 1c, and when the estimated capacitor current effective value Ic_est_rms exceeds the capacitor current limit value Ic_lim, the alarm signal generating unit 750 of the control device 3b outputs an alarm signal ALM indicating that the estimated capacitor current effective value Ic_est_rms has exceeded the capacitor current limit value Ic_lim.

[0117] The alarm signal ALM notifies the user of the device or a higher-level control device (not shown) that there is a possibility of a breakdown or deterioration of the life of the capacitor 200. The power conversion device 1c may have a function of automatically stopping the switching signal Q1 to the inverter control unit 600 if the alarm signal ALM is not released for a certain period of time. In this way, various measures such as warnings and protection can be taken before the device breaks down.

[0118] As described above, the control device provided in the power conversion device according to the fourth embodiment includes an alarm signal generating unit that outputs an alarm signal when the estimated value of the capacitor current exceeds a preset limit value of the capacitor current. According to the power conversion device according to the fourth embodiment, it is possible to monitor the capacitor current during operation of the power conversion device, and to determine whether or not there is a possibility of failure or deterioration of the capacitor's life, and, if there is a possibility of failure or deterioration of the capacitor's life, it is possible to appropriately notify the user of the device, a higher-level control device, or the like of information to that effect.

[0119] Embodiment 5. Fig. 16 is a diagram showing an example of the configuration of a power conversion device 1d and a motor drive device 2d according to embodiment 5. Compared to the configuration shown in Fig. 15, in Fig. 16, the control device 3b is replaced with a control device 3c, and a speed command drooping unit 760 is added to the control device 3c. The motor drive device 2d according to embodiment 5 is composed of the power conversion device 1d and a motor 401 provided in a compressor 400. The other configurations are the same or equivalent to those in Fig. 15, and the same reference numerals are used to denote the same or equivalent components, and duplicated explanations will be omitted.

[0120] Although not limited to the power conversion device 1d according to the fifth embodiment, the greater the power consumption of the inverter 300 and the motor 401, which are located on the load side as viewed from the capacitor 200, the greater the capacitor current Ic flowing through the capacitor 200. Therefore, when the estimated capacitor current effective value Ic_est_rms exceeds the capacitor current limit value Ic_lim and an alarm signal ALM is output, the power consumption on the load side can be reduced by lowering the rotation speed of the motor 401, and the capacitor current Ic can also be reduced.

[0121] 16 , when speed command drooping unit 760 receives an alarm signal ALM, it generates a signal Δω for reducing the rotation speed of motor 401. Speed ​​command drooping unit 760 generates a corrected speed command ω** (=ω*-Δω) by subtracting signal Δω from speed command ω*, and outputs the generated corrected speed command ω** to inverter control unit 600.

[0122] The inverter control unit 600 performs the calculations shown in the first embodiment based on the corrected speed command ω** instead of the speed command ω*, and controls the speed of the motor 401 so that the rotation speed coincides with the corrected speed command ω**.

[0123] Although the output power to the motor 401 is reduced by the operation of the speed command drooping unit 760, the operation of the power conversion device 1d can be continued while preventing the breakdown or deterioration of the capacitor 200.

[0124] As described above, the control device provided in the power conversion device according to the fifth embodiment includes a speed command drooping unit that receives an alarm signal output from an alarm signal generating unit and reduces a rotation speed command of the motor. According to the power conversion device according to the fifth embodiment, when an alarm signal is received indicating that the capacitor may have a failure or a deterioration in its life, the motor rotation speed command is reduced to reduce the output power to the motor, thereby providing an effect of allowing the operation of the power conversion device to be continued while preventing the capacitor from failing or deteriorating in its life.

[0125] Embodiment 6 FIG. 17 is a diagram showing a configuration example of a refrigeration cycle-applied equipment 900 according to the sixth embodiment. The refrigeration cycle-applied equipment 900 according to the sixth embodiment includes the power conversion device 1 described in the first embodiment. Note that the power conversion device 1 may be replaced with the power conversion devices 1a to 1d described in the second to fifth embodiments. The refrigeration cycle-applied equipment 900 according to the sixth embodiment can be applied to products including a refrigeration cycle, such as air conditioners, refrigerators, freezers, and heat pump water heaters. Note that in FIG. 17, components having the same functions as those in the first embodiment are denoted by the same reference numerals as those in the first embodiment.

[0126] The refrigeration cycle device 900 includes a compressor 400 incorporating the motor 401 in the first embodiment, a four-way valve 902, an indoor heat exchanger 906, an expansion valve 908, and an outdoor heat exchanger 910, which are attached via refrigerant piping 912.

[0127] Inside the compressor 400, a compression mechanism 904 that compresses the refrigerant, and a motor 401 that operates the compression mechanism 904 are provided.

[0128] The refrigeration cycle device 900 can perform heating or cooling operation by switching the four-way valve 902. The compression mechanism 904 is driven by the motor 401 which is variably controlled in speed.

[0129] During heating operation, as indicated by the solid arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, passes through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910, and returns to the compression mechanism 904.

[0130] During cooling operation, as indicated by the dashed arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, passes through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906 and the four-way valve 902 and returns to the compression mechanism 904.

[0131] During heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 reduces the pressure of the refrigerant to expand it.

[0132] The power conversion device 1 can estimate the capacitor current Ic flowing through the capacitor 200 by using the current detection units 301a, 301b installed in the converter 100 and the inverter 300, respectively. Therefore, the capacitor current Ic can be estimated without installing an additional current detection unit at a portion of the electric wiring connecting the capacitor 200. As a result, by monitoring the capacitor current Ic during operation of the refrigeration cycle application device 900, it is possible to obtain an effect that various measures such as appropriate warning and protection can be taken before the device breaks down or the capacitor reaches the end of its life due to an overcurrent in the capacitor 200.

[0133] The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies, or the embodiments may be combined with each other. Also, parts of the configurations may be omitted or modified without departing from the spirit of the invention. [Explanation of symbols]

[0134] 1, 1a to 1d power conversion device, 2, 2a to 2d motor drive device, 3, 3a to 3c control device, 100, 100a converter, 110 commercial power source, 120, 120a reactor, 130 diode rectifier, 140a to 140c diode, 150, 150a, Sup, Sun, Svp, Svn, Swp, Swn switch element, 160 simple switching cell, 200 capacitor, 201a, 201b voltage detection unit, 300 inverter, 301a, 301b current detection unit, 400 compressor, 401 motor, 402 mechanical load, 500 converter control unit, 600 inverter control unit, 601 speed control unit, 602 d-axis current command determination unit, 603 current restoration unit, 604 current control unit, 605 position estimation unit, 606 PWM signal generator, 700, 700a capacitor current estimation unit, 710, 710a estimated current calculation unit, 720 current effective value calculation unit, 721a to 721h Fourier series expansion calculation unit, 722 frequency component summation unit, 730 inverter current average value calculation unit, 750 alarm signal generation unit, 760 speed command drooping unit, 900 refrigeration cycle applicable equipment, 902 four-way valve, 904 compression mechanism, 906 indoor heat exchanger, 908 expansion valve, 910 outdoor heat exchanger, 912 refrigerant piping.

Claims

1. a converter including at least one switch element, which converts a first AC voltage applied from an AC power supply into a DC voltage; a capacitor that smoothes the voltage output by the converter; an inverter that converts the DC voltage smoothed by the capacitor into a second AC voltage and applies the second AC voltage to a motor to rotate it; a control device that controls the drive of the converter and the inverter, the converter and the inverter each include at least one current detection unit for detecting a converter current flowing through the converter and an inverter current flowing through the inverter; the control device includes a capacitor current estimation unit that uses the detection values ​​of the converter current and the inverter current detected by the current detection unit, and estimates the capacitor current by switching calculation processing depending on the operating state of the converter, with a capacitor current flowing through the capacitor as a calculation object; The capacitor current estimation unit calculates current amplitudes of a plurality of specific frequency components included in the capacitor current by performing frequency analysis on the estimated capacitor current, normalizes each of the calculated current amplitudes of the plurality of specific frequency components in accordance with the frequency characteristics of the capacitor, and calculates an effective value of the estimated capacitor current using the normalized current amplitudes of the plurality of specific frequency components. A power conversion device characterized by:

2. the inverter includes a plurality of switch elements whose switching is controlled by a phase voltage command generated by the control device and applied to each phase of the inverter; When a phase having an intermediate relationship between the magnitudes of the phase voltage commands for the plurality of switch elements is determined as an intermediate phase, the capacitor current estimation unit An average value of the inverter current is calculated for each switching period of the inverter based on the product of a first switching state duration before the switching state of the switching element of the intermediate phase is switched and a detected value of the inverter current detected during the first switching state duration, and the product of a second switching state duration after the switching state of the switching element of the intermediate phase is switched and a detected value of the inverter current detected during the second switching state duration.

2. The power conversion device according to claim 1.

3. The plurality of specific frequencies analyzed by the capacitor current estimation unit include at least one of a frequency N times the frequency of the AC voltage applied from the AC power supply, a frequency M times the rotation frequency of the mechanical device to be driven, a frequency sum that is the sum of the N-times frequency and the M-times frequency, and a frequency difference that is the difference between the N-times frequency and the M-times frequency, where N and M are any natural numbers.

2. The power conversion device according to claim 1.

4. The plurality of specific frequencies analyzed by the capacitor current estimation unit include at least one of a frequency that is K times the switching frequency of the converter and a frequency that is L times the switching frequency of the inverter, where K and L are any natural numbers.

2. The power conversion device according to claim 1.

5. 5. The power conversion device according to claim 1, wherein the control device includes an alarm signal generating unit that outputs an alarm signal when the estimated value of the capacitor current exceeds a preset limit value of the capacitor current.

6. 6. The power conversion device according to claim 5, wherein the control device includes a speed command drooping unit that receives the alarm signal and reduces a rotation speed command for the motor.

7. A motor drive device comprising the power conversion device according to any one of claims 1 to 4.

8. A refrigeration cycle device comprising the power conversion device according to any one of claims 1 to 4.

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