Power conversion device

Abstract

The purpose of the present invention is to provide a power conversion device capable of preventing a high-frequency current from flowing through a smoothing inductor, and of suppressing a decrease in efficiency associated with a step-up operation.　A power conversion device (10) comprises: a flat smoothing inductor (Lac) to which an AC current (is) is input from an AC power supply (91); a rectifier (13) and a step-up chopper circuit (14) that are connected to the output side of the smoothing inductor (Lac); a smoothing capacitor (Cdc) that is connected to the output side of the step-up chopper circuit (14); and an inverter (16) that converts a voltage (Vdc) across the smoothing capacitor into an AC and supplies AC output power to a motor (92). A filter capacitor (Cx) is provided between the input side of the step-up inductor (Lx) of the step-up chopper circuit (14) and the output side of the smoothing inductor (Lac).

Description

Power Conversion Device

[0001] The present disclosure relates to a power conversion device.

[0002] In a power conversion device that converts AC power from an AC power source to drive a motor or the like, it is desirable to improve the power factor of the input power. Conventional methods for improving the power factor of input power include the full PAM (Pulse Amplitude Modulation) method and the electrolytic capacitor-less method. The full PAM method performs full-wave rectification using diode rectification, followed by boosting and power factor improvement using a boost chopper circuit. The electrolytic capacitor-less method achieves power factor improvement by pulsating the DC bus voltage using a small-capacity capacitor connected to the DC bus and injecting the resulting power pulsation into the motor. Both methods achieve power factor improvement without using a large smoothing inductor, and solve the problems associated with increased size of the power conversion device, but each method also has its own unique issues.

[0003] First, in the full-PAM system, the addition of a boost chopper circuit increases the number of passing elements, resulting in a decrease in overall efficiency. Furthermore, the switching elements of the boost chopper circuit operate at high frequencies, causing problems such as switching noise, which also leads to a decrease in efficiency. In the capacitor-less system, distortion of the input current waveform that occurs near the voltage zero crossing makes it difficult to comply with harmonic regulations. While applying flux-weakening control to the capacitor-less system is one way to resolve the harmonic regulation issues, the increase in motor current and inverter current that accompanies flux-weakening control still leads to a decrease in efficiency.

[0004] Therefore, there are methods in which the boost chopper circuit only performs boost operation, power factor improvement is performed using a method similar to that of a capacitor-less method, and the increase in motor current, etc., associated with flux-weakening control is suppressed by adjusting the fluctuating frequency of the output power (see, for example, Patent Document 1).

[0005] JP 2016-73203 A

[0006] However, in the technology described in Patent Document 1, a reactor connected between an AC power supply and a full-wave rectifier circuit is used as a boost inductor. Such a reactor typically functions as a smoothing inductor, through which the main current of the power conversion device flows. The main current has the AC power supply frequency as its fundamental frequency, and is primarily composed of a low-frequency current. However, since a high-frequency current flows during boost operation, the current flowing through the reactor is a current in which high-frequency components are superimposed on low-frequency components, resulting in increased copper loss and core loss. Furthermore, increased copper loss and core loss result in reduced efficiency.

[0007] The present disclosure discloses a technology for solving the above-mentioned problems, and aims to obtain a power conversion device that can prevent high-frequency current from flowing through a smoothing inductor and suppress the decrease in efficiency associated with boost operation.

[0008] The power conversion device of the present disclosure includes: a first inductor that receives AC current from an AC power supply; a second inductor connected to the output side of the first inductor; a boost chopper circuit having a switching element connected between a high-potential side DC bus and a low-potential side DC bus on the output side of the second inductor; a rectifier that has a first diode having an anode connected to the output end of the first inductor and a cathode connected to the high-potential side DC bus and is provided between the first inductor and the boost chopper circuit; a first capacitor connected between the high-potential side DC bus and the low-potential side DC bus on the output side of the boost chopper circuit; an inverter that converts a voltage across the first capacitor to AC and supplies the AC output power to a motor; and a second capacitor having one end connected to a first electric circuit that connects the output end of the first inductor and the input end of the second inductor and the other end connected to a second electric circuit that is connected in parallel with the first electric circuit.

[0009] According to the power conversion device of the present disclosure, it is possible to prevent a high frequency current from flowing through the smoothing inductor and suppress a decrease in efficiency due to the boost operation.

[0010] 3B is an enlarged view of the vicinity of the voltage zero crossing in FIG. 3B. It is a functional block diagram of a boost chopper control unit according to the first embodiment. It is a diagram showing an example of the hardware configuration of a boost chopper control unit and a motor control unit according to the first embodiment. It is a diagram showing the configuration of a power conversion device according to the second embodiment. It is a diagram showing the configuration of a power conversion device according to the third embodiment. It is a diagram showing the configuration of a power conversion device according to a modified example of the third embodiment. It is a diagram showing the configuration of a power conversion device according to the first embodiment. It is a diagram showing the configuration of a power conversion device according to the second embodiment. It is a diagram showing the configuration of a power conversion device according to the third embodiment. It is a diagram showing the configuration of a power conversion device according to a modified example of the third embodiment. It is a diagram showing the configuration of a power conversion device according to the first embodiment. It is a diagram showing the configuration of a power conversion device according to the first embodiment. It is a diagram showing the configuration of a power conversion device according to the second embodiment. It is a diagram showing the configuration of a power conversion device according to the third embodiment. It is a diagram showing the configuration of a power conversion device according to the modified example of the third embodiment. It is a diagram showing the configuration of a power conversion device according to the first ... first embodiment. It is a diagram showing the configuration of a power conversion device according to the first embodiment. It is a diagram showing the configuration of

[0011] Embodiment 1. Embodiment 1 will be described with reference to Figures 1 to 5. Figure 1 is a diagram showing the configuration of a power conversion device in embodiment 1. Power conversion device 10 converts AC power from an AC power supply 91 and supplies the converted AC power to a motor 92. Power conversion device 10 includes, from the input side, an input filter 11, a smoothing inductor unit 12, a rectifier 13, a boost chopper circuit 14, a smoothing capacitor unit 15, and an inverter 16. Power conversion device 10 also includes a boost chopper control unit 17 that controls the boost chopper circuit 14, and a motor control unit 18 that controls the motor 92 via control of the inverter 16.

[0012] The AC power supply 91 generates an AC voltage vs and supplies an AC current is to the power conversion device 10. The AC power supply 91 supplies single-phase AC power supply power ps, which is the product of the AC current is and the AC voltage vs. The motor 92 is a motor having three-phase armature windings (not shown), for example, u-phase, v-phase, and w-phase. The motor 92 is supplied with a u-phase current iu, a v-phase current iv, and a w-phase current iw from the power conversion device 10, and is driven by these three-phase currents flowing through the armature windings of the corresponding phases.

[0013] The input filter 11 includes at least one of a smoothing capacitor, a normal mode filter, and a common mode filter (none of which are shown), and has two input terminals and two output terminals. Note that in AC, the high-potential and low-potential electrical paths periodically switch positions, but for ease of explanation, the upper side in FIG. 1 is designated as the high-potential side, and the lower side is designated as the low-potential side. Therefore, the upper side of the two input terminals and the lower side of the two output terminals of the input filter 11 are designated as the high-potential side and the low-potential side, respectively. An AC current is is input to the high-potential input terminal of the input filter 11.

[0014] The smoothing inductor section 12 is provided on the output side of the input filter 11 and includes a smoothing inductor Lac, i.e., a first inductor. The input end of the smoothing inductor Lac is connected to the high-potential output terminal of the input filter 11. The output end of the smoothing inductor Lac is connected to the rectifier 13. A main current of the power conversion device 10 flows through the smoothing inductor Lac. Because this main current has a roughly sinusoidal waveform, it is preferable that the smoothing inductor Lac be made of a material with a high magnetic flux density, such as an electromagnetic steel sheet or a dust-based material.

[0015] The rectifier 13 is provided on the output side of the smoothing inductor unit 12 and includes a full-wave rectifier circuit composed of four diodes D1 to D4, each with its cathode connected to the high-potential side and its anode connected to the low-potential side. This full-wave rectifier circuit has a first series connection, consisting of diodes D1 and D2 connected in series, on the input side, and a second series connection, consisting of diodes D3 and D4 connected in series, on the output side. The cathodes of diodes D1 and D3 (corresponding to the first diode) are connected to the high-potential side DC bus DCH, i.e., the high-potential side DC bus, and the anodes of diodes D2 and D4 are connected to the low-potential side DC bus DCL, i.e., the low-potential side DC bus. The junction of diodes D1 and D2 (the junction of the anode of diode D1 and the cathode of diode D2) is connected to the output end of the smoothing inductor Lac. The connection point between diode D3 and diode D4 (the connection point between the anode of diode D3 and the cathode of diode D4) is connected to the low-potential side output terminal of the input filter 11. The connection point between diode D1 and diode D2 and the connection point between diode D3 and diode D4 each serve as an AC input terminal of the rectifier 13. The output terminal of the rectifier 13 serves as the connection point between the cathodes of diodes D1 and D3 and the high-potential side DC bus DCH. In other words, the output destination of the rectifier 13 is the high-potential side DC bus DCH.

[0016] A smoothing inductor current iac is input to the rectifier 13. The smoothing inductor current iac input to the rectifier 13 is split into two at the AC input terminal of the rectifier 13. One current flows to the high-potential side, is rectified by diodes D1 and D3, and flows through the high-potential side DC bus DCH as current irec1. The other current flows to the boost chopper circuit 14, is rectified by diodes D5 and D6, and flows as current irec2. Therefore, the sum of current irec1 and current irec2 is equal to the smoothing inductor current iac. Which of current irec1 and current irec2 mainly flows depends on whether the boost chopper circuit 14 is operating, and this will be described in detail later.

[0017] The boost chopper circuit 14 includes, from the input side, diodes D5 and D6, a filter capacitor Cx (i.e., a second capacitor), a boost inductor Lx (i.e., a second inductor), and a series connection of a switching element Sx and a diode Dx. The diodes D5 and D6 are connected in parallel, with their anodes connected to the AC input terminal of the rectifier 13. As shown in the figure, the diode D5 is connected to the high-potential side and the diode D6 is connected to the low-potential side, with the anode of the diode D5 connected to the output terminal of the smoothing inductor Lac. Therefore, the diode D5 corresponds to a second diode. The anode of the diode D6 is connected to the low-potential output terminal of the input filter 11. The cathodes of the diodes D5 and D6 are connected to the filter capacitor Cx and the boost inductor Lx.

[0018] One end of the filter capacitor Cx is connected to the cathodes of the diodes D5 and D6, and the other end is connected to the low-potential-side DC bus DCL. The input end of the boost inductor Lx is connected to one end of the filter capacitor Cx and the cathodes of the diodes D5 and D6, and the output end is connected to the collector of the switching element Sx and the anode of the diode Dx. The diodes D5 and D6 are located on an electrical path connecting the output end of the smoothing inductor Lac and the input end of the boost inductor Lx, i.e., on the first electrical path. The low-potential-side DC bus DCL is connected in parallel with the first electrical path, and the other end of the filter capacitor Cx is connected thereto, so in the first embodiment, the low-potential-side DC bus DCL corresponds to the second electrical path.

[0019] The boost inductor Lx functions only near the current zero crossing point, and therefore a current with a higher frequency and a smaller magnitude than the main current flows through it. For this reason, it is desirable to make the boost inductor Lx from a material with low core loss, such as a dust-based material or a ferrite-based material. Furthermore, the inductance of the boost inductor Lx may be dynamically changed by controlling the magnetic flux of the core of the boost inductor Lx.

[0020] The switching element Sx is configured, for example, with an IGBT (Insulated Gate Bipolar Transistor) and a diode connected in anti-parallel to the IGBT, with its collector connected to the output end of the boost inductor Lx and the anode of the diode Dx. The emitter of the switching element Sx is connected to the low-potential side DC bus DCL. The gate of the switching element Sx is connected to a control circuit (not shown). The anode of the diode Dx is connected to the output end of the boost inductor Lx and the collector of the switching element Sx, and its cathode is connected to the high-potential side DC bus DCH.

[0021] The switching element Sx may be configured with a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) and its parasitic diode, or may be configured with a MOSFET and a diode that is a separate element. The semiconductor material of the switching element, such as an IGBT or MOSFET, may be silicon (Si), or a wide bandgap semiconductor material having a bandgap wider than that of silicon, such as silicon carbide (SiC), gallium nitride (GaN), or gallium oxide (Ga2O3). That is, the switching element Sx may be a Si-MOSFET, a Si-IGBT, a SiC-MOSFET, a SiC-IGBT, a GaN power transistor, a Ga2O3 power transistor, or the like. When a MOSFET is used as the switching element Sx, the above-mentioned collector, emitter, and gate are replaced with drain, source, and base. By using a wide bandgap semiconductor material for the switching element Sx, a boost chopper circuit capable of high voltage resistance, good heat dissipation, and high-speed switching can be obtained. The switching element Sx may also be configured with multiple switching elements, and these switching elements may be connected in multiple parallel or multiple series. This can improve the current capacity and withstand voltage of the switching element Sx. When configuring the switching element Sx using multiple switching elements, multiple types of switching elements, such as the above-mentioned Si-IGBT or SiC-MOSFET, may be mixed.

[0022] Although the boost chopper circuit 14 is configured by one boost chopper circuit in the first embodiment, the boost chopper circuit 14 may be configured by a multi-phase boost chopper circuit in which a plurality of boost chopper circuits are connected in parallel. When a plurality of boost chopper circuits are used, the elements (switching elements, boost inductors, etc.) of each boost chopper circuit may have the same characteristics or different characteristics.

[0023] Diodes D5 and D6 rectify the smoothing inductor current iac and pass the rectified current irec2 to the output side, generating a full-wave rectified voltage of the AC power supply 91. Therefore, in the first embodiment, the filter capacitor voltage Vx, which is the voltage across the filter capacitor Cx, becomes the full-wave rectified voltage of the AC power supply 91. The current irec2 flows through the boost inductor Lx as the boost inductor current ix depending on the on / off state of the switching element Sx. The filter capacitor voltage Vx serves as the input voltage of the boost chopper circuit 14, and the boost chopper circuit 14 operates to generate an output voltage vx and an output current isub. The output current isub is output to the high-potential side DC bus DCH. Therefore, the current idc flowing through the high-potential side DC bus DCH on the output side of the boost chopper circuit 14 is the sum of the current irec1 rectified by the rectifier 13 and the output current isub of the boost chopper circuit 14. The operation of the boost chopper circuit 14 is controlled by a boost chopper control unit 17. The boost chopper control unit 17 will be described in detail later.

[0024] The smoothing capacitor unit 15 is provided on the output side of the rectifier 13 and the boost chopper circuit 14. One end of the smoothing capacitor unit 15 is connected to the high-side DC bus DCH and the other end is connected to the low-side DC bus DCL. This smoothing capacitor unit 15 includes a smoothing capacitor Cdc (i.e., a first capacitor). The current iinv passing through the smoothing capacitor unit 15 is input to the inverter 16 as the input current iinv. The smoothing capacitor Cdc is a small-capacity capacitor, such as a filter capacitor, that actively pulsates the DC bus voltage Vdc to inject power pulsation into the motor, thereby achieving a power factor correction without the use of a capacitor. For example, the capacitance of the smoothing capacitor Cdc may be set so that the DC bus voltage Vdc, which is the voltage across the smoothing capacitor Cdc, pulsates at twice the frequency of the AC voltage vs (the same frequency as the AC current is). In this case, the DC bus voltage Vdc pulsates with the full-wave rectified waveform of the AC voltage vs. Furthermore, the smoothing capacitor Cdc forms an LC filter in combination with the smoothing inductor Lac, and decouples the high frequency components contained in the input current iinv.

[0025] The inverter 16 is provided between the smoothing capacitor unit 15 and the motor 92, and is configured by connecting three legs in parallel, each corresponding to a u-phase, v-phase, and w-phase. The leg corresponding to the u-phase is configured by a series connection of switching elements S1 and S2. The collector of the switching element S1 is connected to the high-potential side DC bus DCH, and the emitter is connected to the collector of the switching element S2. The emitter of the switching element S2 is connected to the low-potential side DC bus DCL. The connection point between the emitter of the switching element S1 and the collector of the switching element S2 (the potential of this connection point is designated vu) is connected to the armature winding of the motor 92 corresponding to the u-phase, and a u-phase current iu flows from this connection point to the motor 92.

[0026] The leg corresponding to the v-phase is composed of a series connection of switching elements S3 and S4. The collector of switching element S3 is connected to the DC bus DCH, and the emitter is connected to the collector of switching element S4. The emitter of switching element S4 is connected to the DC bus DCL. The connection point between the emitter of switching element S3 and the collector of switching element S4 (the potential at this connection point is vv) is connected to the armature winding corresponding to the v-phase of the motor 92, and a v-phase current iv flows from this connection point to the motor 92. The leg corresponding to the w-phase is composed of a series connection of switching elements S5 and S6. The collector of switching element S5 is connected to the DC bus DCH, and the emitter is connected to the collector of switching element S6. The emitter of switching element S6 is connected to the DC bus DCL. The connection point between the emitter of switching element S5 and the collector of switching element S6 (the potential at this connection point is vw) is connected to the armature winding corresponding to the w-phase of motor 92, and w-phase current iw flows from this connection point to motor 92. Inverter 16 supplies output power to motor 92 by supplying currents of each phase as described above. Note that a multilevel inverter, a dual inverter, or the like may be used as inverter 16.

[0027] Like the switching element Sx of the boost chopper circuit 14, the switching elements S1 to S6 can be configured with, for example, an IGBT and a diode connected in anti-parallel to the IGBT. Also, like the switching element Sx, a MOSFET may be used instead of the IGBT, and a wide bandgap semiconductor material may be used as the semiconductor material of the switching elements. Furthermore, each of the switching elements S1 to S6 may be configured with multiple switching elements connected in parallel or in series. This, like the switching element Sx described above, improves the current capacity and withstand voltage of the switching elements S1 to S6. Furthermore, an inverter capable of high-speed switching can be obtained.

[0028] The inverter 16 is controlled by a motor control unit 18. The motor control unit 18 calculates a duty cycle based on a motor rotation speed command value or a torque command value, and generates a gate signal from the duty cycle. The motor control unit 18 controls the switching operations of the switching elements S1 to S6 using the gate signal to generate the desired u-phase current iu, v-phase current iv, and w-phase current iw. The motor control unit 18 controls the inverter 16 in a manner similar to that of a typical capacitor-less inverter (a capacitor-less system). The motor control unit 18 sequentially updates switching information (on / off commands) for each switching element of the inverter 16 to drive the motor 92 at a desired speed and torque. The motor control unit 18 also controls each switching element of the inverter 16 so that the AC power supply power ps and the motor power pm match. For example, a torque command value for the motor 92 is input, and the motor control unit 18 calculates an on / off signal Swi so that the torque of the motor 92 matches the torque command value. Here, the on / off signal Swi represents the on / off signals for the switching elements S1 to S6. Furthermore, when feedback control is performed, the motor control unit 18 receives as feedback the detected values ​​of the motor currents (u-phase current iu, v-phase current iv, and w-phase current iw), and controls these motor currents to match the current command values. Furthermore, the motor control unit 18 may control the motor using only the currents of two phases, rather than all three phases of the motor current. Furthermore, the inverter input current iinv, the rectified current irec1, the smoothing inductor current iac, and the smoothing inductor voltage vLac (not shown) may be used as control parameters. In other words, any parameter that has some correlation with the motor current may be used.

[0029] Furthermore, since the motor control unit 18 performs control in conjunction with the boost chopper control unit 17, it transmits, as necessary, a motor power command value Pref, which is a command value for the motor power pm, to the boost chopper control unit 17. It may also receive a status signal St, which indicates whether the boost chopper circuit 14 is operating, from the boost chopper control unit 17, and perform motor control according to the operating state of the boost chopper circuit 14. Note that the transmission and reception of data between the boost chopper control unit 17 and the motor control unit 18 is not limited to the above, and may transmit and receive, as necessary, sensor detection values ​​such as AC current is, motor current detection values ​​(including dq current), inverter output voltages (vu, vv, vw), duty command values, and the like, and use them in the respective controls.

[0030] Various motors can be used for the motor 92, such as an IM (Induction Motor), an IPMSM (Interior Permanent Magnet Synchronous Motor), an SRM (Switched Reluctance Motor), or a Synchronous Reluctance Motor (SynRM). The wiring of the armature windings of the motor 92 is not particularly limited, and open winding, Y-connection, Δ-connection, or the like can be used. While the first embodiment describes a single-phase three-phase motor 92 and inverter 16, they may also be dual-phase, or may have a multi-phase configuration other than three-phase.

[0031] The operation of the power conversion device 10 configured as described above will be briefly explained using a simplified equivalent circuit. First, the power supply mode of the power conversion device 10 differs significantly depending on whether the boost chopper circuit 14 is operating. FIG. 2A is a diagram showing a simplified equivalent circuit of the power conversion device according to the first embodiment and the flow of power supply when the boost chopper circuit is stopped, and FIG. 2B is a diagram showing the flow of power supply when the boost chopper circuit is operating. As shown in FIG. 2A, when the boost chopper circuit 14 is stopped, the power conversion device 10 has a circuit topology similar to that of a capacitor-less inverter. In this case, the power flow is as follows: AC power source 91 (AC power source power ps) → smoothing inductor Lac → diode D1 → inverter 16 (output power prec1) → motor 92 (motor power pm). When the boost chopper circuit 14 is operating, the power conversion device 10 has a circuit topology similar to that of a full-PAM system, as shown in FIG. 2B. In this case, the power flow is AC power supply 91 (AC power supply power ps) → smoothing inductor Lac → boost chopper circuit 14 → inverter 16 (output power prec2) → motor 92 (motor power pm).

[0032] Similarly to the power flow, the current flow is such that when the boost chopper circuit is stopped, the current irec1 mainly flows via the diode D1 of the rectifier 13, and when the boost chopper circuit is operating, the current irec2 mainly flows without passing through the diode D1 of the rectifier 13. The current irec2 is output as the output current isub due to the boost operation and power factor correction by the boost chopper circuit 14. Even when two types of current (and output power) are generated in motor control, it is necessary to control the motor so that the AC power supply power ps, which is the input power, and the motor power pm are equal. The AC power supply power ps is the product of the AC voltage vs and the AC current is, and since equations (1) and (2) hold for the AC voltage vs and the AC current is, respectively, the AC power supply power ps can be expressed as shown in equation (3). Here, Vs is the effective value of the AC voltage vs, and Is is the effective value of the AC current is. ωs is the angular frequency of the AC power supply 91, and t is time. In the formula, vs, is, and ps represent the values ​​of the AC voltage vs, the AC current is, and the AC power supply power ps. Similarly, hereinafter, when symbols for components are used in formulas, they represent the detected values ​​of the components unless otherwise specified.

[0033] To control the motor so that the AC power supply power p and the motor power p are equal, the motor power p can be adjusted to the right-hand side of equation (3). However, in actual motor control, distortion occurs in the waveforms of the AC voltage v and the AC current is near the voltage zero crossing of the AC power supply 91, i.e., in the region where the AC voltage v approaches zero volts. Distortion in the waveforms of the AC voltage v and the AC current is makes it difficult to comply with harmonic regulations. While flux-weakening control could be applied to motor control to comply with harmonic regulations, this increases the motor current and inverter current, leading to reduced efficiency and increased costs. For this reason, in the first embodiment, different control is performed near the voltage zero crossing and in other regions. That is, in regions other than near the voltage zero crossing, the boost chopper circuit 14 is stopped to achieve the power flow shown in FIG. 2A , while the boost chopper circuit 14 is operated near the voltage zero crossing to achieve the power flow shown in FIG. 2B . 2B, the boost chopper circuit 14 is present between the AC power supply 91 and the motor 92, so that the power factor is corrected and the DC bus voltage Vdc is boosted. In this way, in the first embodiment, the boost operation and the power factor are corrected as needed.

[0034] In the power flow shown in FIG. 2A, current irec1 has a nearly continuous current waveform due to the smoothing inductor Lac at the input. Furthermore, diodes D1 to D4 switch on and off at half the cycle of AC power supply power ps, i.e., at the same frequency as AC voltage vs, resulting in almost no recovery loss. In the power flow shown in FIG. 2B, boost inductor current ix contains harmonic components. However, a filter capacitor Cx is connected between the AC input terminal of rectifier 13 and boost inductor Lx, and the ripple component of boost inductor current ix flows primarily through filter capacitor Cx. Furthermore, a smoothing inductor Lac is connected to the input side of rectifier 13, which attenuates the ripple component. Therefore, harmonic currents are prevented from flowing through diodes D1 to D4, and diodes D1 to D4 do not require high-speed switching. Therefore, low-cost general rectifier diodes (hereinafter referred to as low-speed diodes) can be used for diodes D1 to D4.

[0035] Furthermore, since the filter capacitor Cx is connected between the diodes D5, D6 and the boost inductor Lx, and the smoothing inductor Lac is connected to the input sides (anodes) of the diodes D5, D6, the same thing can be said about the diodes D5, D6 as well, i.e., the diodes D5, D6 do not need to be switched at high speed, and low-speed diodes can be used.

[0036] Next, the operating waveforms of the power conversion device 10 will be described in comparison with a conventional capacitor-less system. FIG. 3A shows current and voltage waveforms in the conventional capacitor-less system. In FIG. 3A, the solid line indicates the AC voltage vs, and the dotted line indicates the AC current is. The dashed line indicates the DC bus voltage Vdc. In FIG. 3A, the region where the AC voltage vs is smaller than the voltage Vmin corresponds to the voltage zero crossing. As shown in FIG. 3A, near the voltage zero crossing, the DC bus voltage Vdc remains constant at voltage Vmin. As the DC bus voltage Vdc leaves the voltage zero crossing, it increases and reaches a voltage peak Vmax, becoming approximately the same value as the AC voltage vs. The DC bus voltage Vdc then begins to decrease and drops to voltage Vmin. The AC current is follows an AC curve similar to the AC voltage vs except near the voltage zero crossing, but drops rapidly to almost zero when it approaches the voltage zero crossing. Therefore, the conduction period occurs only in the region other than near the voltage zero crossing. The AC current is becomes almost zero near the voltage zero crossing because the current does not conduct through the diode in the region where the DC bus voltage Vdc is small.

[0037] FIG. 3B shows current and voltage waveforms in the first embodiment. In FIG. 3B , during one cycle of the AC voltage vs (one cycle of the AC current is), there is a boost chopper circuit stop period T1 during which the boost chopper circuit 14 is stopped, and a boost chopper circuit operation period T2 during which the boost chopper circuit 14 is operated to perform boost operation and power factor correction. During the boost chopper circuit stop period T1, the power flow is as shown in FIG. 2A , similar to the case of the electrolytic capacitor-less system. On the other hand, during the boost chopper circuit operation period T2, the power flow is as shown in FIG. 2B . In the first embodiment, the boost chopper circuit operation period T2 is set to the vicinity of the voltage zero crossing, and the DC bus voltage Vdc increases to Vmid near the voltage zero crossing. Vmid is the voltage increase due to the boost operation of the boost chopper circuit added to Vmin. As shown in FIG. 3B , in the first embodiment, the DC bus voltage Vdc is maintained at a value greater than Vmin even near the voltage zero crossing. Therefore, in the first embodiment, the AC current is is conducted even near the voltage zero crossing, and there is no distortion in the waveform of the AC current is.

[0038] Even when the motor current increases, such as during flux-weakening control, the boost chopper circuit can be operated to improve the power factor and boost the DC bus voltage Vdc, thereby equalizing the AC power supply power p and the motor power p. Constantly boosting the DC bus voltage Vdc, as in the conventional full-PAM system, can result in excessive motor current and reduced efficiency in the inverter 16 and the motor 92. In contrast, the first embodiment boosts the DC bus voltage Vdc only near the voltage zero crossing, thereby minimizing the aforementioned efficiency reduction. Furthermore, in the conventional full-PAM system, the rectified current irec2 flows through the boost inductor Lx and diodes D5 and D6, requiring a large-capacity inductor for the boost inductor Lx, which can lead to increased size and cost. In contrast, in the first embodiment, the boost chopper circuit operation period T2 is a portion of the cycle of the AC voltage vs and is shorter than the cycle of the AC voltage vs. Therefore, the generation of losses and efficiency associated with the operation of the boost chopper circuit are reduced compared to the conventional full-PAM system. Furthermore, since it is only necessary to operate near the voltage zero crossing, a small-capacity boost chopper circuit can be used as the boost chopper circuit 14. This allows for the overall device to be made smaller and less expensive.

[0039] In the example shown in FIG. 3B , there are periods during one cycle of the AC voltage vs (AC current is) during which the boost chopper circuit 14 operates and periods during which it does not operate. However, from the viewpoint of efficiency, there may be periods during one cycle of the AC voltage vs (AC current is) during which the boost chopper circuit 14 does not operate at all, or from the viewpoint of power factor improvement, the boost chopper circuit 14 may operate during all periods during one cycle of the AC voltage vs (AC current is).

[0040] When the boost chopper circuit 14 is operated, a high-frequency current flows through the boost inductor Lx. To prevent this high-frequency current from flowing into the AC power supply 91, an LC filter is required between the AC power supply 91 and the boost inductor Lx. In the first embodiment, this LC filter is configured with a smoothing inductor Lac and a filter capacitor Cx. The smoothing inductor Lac is used to cut switching current ripples of the motor 92 and is also provided in conventional power conversion devices. That is, in the first embodiment, the high-frequency current caused by the operation of the boost chopper circuit 14 is prevented from flowing into the AC power supply 91 simply by adding the filter capacitor Cx. Therefore, an increase in the volume of the power conversion device 10 due to the prevention of the flow of high-frequency current is suppressed, and an increase in cost is also suppressed.

[0041] FIG. 3C is an enlarged view of the vicinity of the voltage zero crossing in FIG. 3B, showing an enlarged waveform when the boost chopper circuit 14 is operating. As described above, the power flow during the boost chopper circuit operation period T2 is the same as the power flow shown in FIG. 2B, and the current flow is also similar. Furthermore, the power flow during periods other than the boost chopper circuit operation period T2 (the boost chopper circuit stop period T1) is the same as the power flow shown in FIG. 2A, and the current flow is also similar. Looking at FIG. 3C, it can be seen that the current irec1 is almost zero during the boost chopper circuit operation period T2, and the current irec2 is almost zero outside the boost chopper circuit operation period T2. However, as described above, the boost chopper circuit 14 operates near the voltage zero crossing, and the AC current is also small near the voltage zero crossing. Therefore, when comparing the current irec1 and current irec2 during their respective conduction periods, the current irec2 is lower than the current irec1. This means that the power capacity of the boost chopper circuit 14 can be made smaller than the rated power capacity of the power conversion device 10 .

[0042] The power modes of the power conversion device 10 include a continuous conduction mode (CCM), in which there is no zero current period; a discontinuous conduction mode (DCM), in which there is a zero current period; and a critical conduction mode (CRM), which is on the border between CCM and DCM. CCM can reduce the peak current value, but generates on-state losses in the switching elements and recovery losses in the high-speed diodes. On the other hand, CRM and DCM are characterized by increasing the peak current value, but almost no on-state losses in the switching elements or recovery losses in the diodes. Furthermore, CRM and DCM can reduce the inductance of the boost inductor Lx and the volume of the inductor compared to CCM.

[0043] As can be seen from Figures 3B and 3C, the vicinity of the voltage zero crossing is also the vicinity of the current zero crossing. In other words, the boost chopper circuit 14 operates in DCM or CRM near the current zero crossing to control the boost inductor current ix. Therefore, the power capacity of the boost chopper circuit 14 can be small. This means that the high-speed diode Dx can have a small capacity. Furthermore, since soft switching is performed during the zero-current period, noise can be reduced, efficiency can be improved, and costs can be reduced compared to conventional boost chopper circuits that perform hard switching in CCM. The average value of the boost inductor current ix is ​​smoothed by the smoothing inductor Lac and the filter capacitor Cx to become the smoothing inductor current iac. Since the smoothing inductor current iac must be controlled to track the motor power pm, during the boost chopper circuit operation period T2, the boost chopper circuit 14 controls the boost inductor current ix, thereby controlling the smoothing inductor current iac and causing the smoothing inductor current iac to track the motor power pm while also improving the power factor. During the boost chopper circuit stop period T1 when the boost inductor current ix does not flow, the power factor is controlled by the motor control by the motor control unit 18.

[0044] Next, the control of the boost chopper circuit 14 will be further described. Fig. 4 is a functional block diagram of the boost chopper control unit according to the first embodiment. The boost chopper control unit 17 includes an input unit 171, a boost chopper operation determination unit 172, a DCM calculation unit 173, a CRM calculation unit 174, a mode determination unit 175, an on / off signal generation unit 176, and an output unit 177. Note that the arrows connecting the functional units in Fig. 4 correspond to the processing flow.

[0045] The input unit 171 receives various command values ​​and detected values. The command values ​​and detected values ​​received by the input unit 171 include, for example, the AC voltage vs, the motor power command value Pref, the output current isub, and the boost inductor current ix. These detected values ​​and command values ​​are continuously or periodically sent to the input unit 171 from sensors or other control units (such as the motor control unit 18 or a higher-level controller) provided in the circuit.

[0046] The boost chopper operation determination unit 172 determines whether to operate or stop the boost chopper circuit 14 based on various command values ​​and detection values ​​received by the input unit 171. For example, in a region where the AC voltage vs is relatively high, it determines to stop the boost chopper circuit. In this case, the boost chopper operation determination unit 172 generates a status signal St indicating that the boost chopper circuit is in a stopped state. The status signal St is transmitted from the output unit 177 to the motor control unit 18. This causes the motor control unit 18 to recognize that the boost chopper circuit is in a stopped state and to perform motor control using a conventional electrolytic capacitor-less method. Note that the determination of whether the AC voltage vs is relatively high is made by comparing the current AC voltage vs with a predetermined threshold vth. That is, the boost chopper operation determination unit 172 determines to stop the boost chopper circuit if vs > vth.

[0047] When the AC voltage vs is relatively small (when vs≦vth), the boost chopper operation determination unit 172 determines to operate the boost chopper circuit of the boost chopper circuit 14. In this case, the boost chopper operation determination unit 172 generates a status signal St indicating that the boost chopper circuit is in an operating state, and outputs the status signal St to the motor control unit 18 via the output unit 177. This allows the motor control unit 18 to recognize that the boost chopper circuit is in an operating state. Furthermore, when the boost chopper circuit is to be operated, the boost chopper operation determination unit 172 controls the input power of the inverter 16. Therefore, the motor control unit 18 transmits a motor power command value Pref to the boost chopper control unit 17.

[0048] An example of the threshold value vth is the command value Vref of the DC bus voltage Vdc. Furthermore, instead of the determination based on the AC voltage vs, the determination may be based on the AC current is. In this case, the current command value Iref, which is the current command value of the input current iinv, serves as the threshold value for the determination. The current command value Iref can be calculated from the motor power command value Pref.

[0049] The boost chopper circuit 14 has two operating modes: DCM and CRM. The boost chopper control unit 17 first calculates parameters for both modes and selects the more appropriate mode based on the calculation results. Therefore, when it is determined that the boost chopper circuit is to operate, calculations are then performed by both the DCM calculation unit 173 and the CRM calculation unit 174. However, it is also possible to determine in advance which operating mode the circuit will operate, and calculate parameters only for that operating mode.

[0050] The DCM calculation unit 173 calculates various parameters for DCM. In DCM, control is performed to satisfy the following equation (4). In equation (4), iL_dcm indicates the magnitude of the instantaneous current of the boost inductor current ix in DCM, and is expressed as shown in equation (5). In equation (5), Tsw_on_dcm is the on-time in DCM with respect to the switching period Tsw of the switching element Sx, and is expressed as shown in equation (6). In equation (5), Lx is the self-inductance of the boost inductor Lx.

[0051] The CRM calculation unit 174 calculates various parameters for CRM. In CRM, control is performed to satisfy the following equation (7). In equation (7), iL_crm indicates the magnitude of the instantaneous current of the boost inductor current ix in CRM, and is expressed as in equation (8). In equation (8), Tsw_on_crm is the on-time of the switching element Sx with respect to the switching period Tsw_crm in CRM, and is expressed as in equation (9). Tsw_crm_off, which is the off-time with respect to the switching period Tsw_crm, is expressed as in equation (10). Furthermore, the switching period Tsw_crm is expressed as in equation (11).

[0052] The current command value Iref when the boost chopper circuit 14 is operating may be a command value of a sine wave with the frequency of the AC power supply 91 as the fundamental wave, and harmonic components such as third, fifth, and seventh harmonic components may be superimposed on the current command value Iref.

[0053] The mode determination unit 175 determines in which mode, DCM or CRM, the boost chopper circuit of the boost chopper circuit 14 should be operated, based on the calculation results of the DCM calculation unit 173 and the CRM calculation unit 174. The mode determination unit 175 selects the mode that results in smaller loss, for example, under the constraint that the switching frequency is equal to or less than an upper limit value.

[0054] The on / off signal generation unit 176 generates an on / off signal Swx for the switching element Sx based on the calculation results by the DCM calculation unit 173 and the CRM calculation unit 174 and the determination result by the mode determination unit 175. When the mode determination unit 175 selects DCM, the on / off signal generation unit 176 generates the on / off signal Swx using the calculation result by the DCM calculation unit 173. When the mode determination unit 175 selects CRM, the on / off signal Swx is generated using the calculation result by the CRM calculation unit 174.

[0055] The output unit 177 outputs the on / off signal generated by the on / off signal generation unit 176 to the switching element Sx of the boost chopper circuit 14, and also outputs various command values ​​and detection values ​​in the boost chopper control unit 17 to the outside (such as the motor control unit 18) as necessary.

[0056] The operation of the boost chopper control unit described above is an example, and the detected values ​​and calculation information used in the motor control unit 18 may be used to control the boost chopper circuit 14, or the boost inductor voltage vL (not shown), the smoothing inductor current iac, the smoothing inductor voltage vLac, the current iSx (not shown) flowing through the switching element Sx, the current iDx (not shown) flowing through the diode Dx, the filter capacitor voltage Vx, the DC bus voltage Vdc, etc. may be used.

[0057] 5 is a diagram showing an example of the hardware configuration of the boost chopper control unit and the motor control unit according to the first embodiment. The boost chopper control unit 17 and the motor control unit 18 are mainly composed of a processor 81, a memory 82 serving as a main storage device, and an auxiliary storage device 83. The processor 81 is composed of, for example, a microcomputer, a CPU (Central Processing Unit), an ASIC (Application Specific Integrated Circuit), an IC (Integrated Circuit), a DSP (Digital Signal Processor), an FPGA (Field Programmable Gate Array), a microcontroller, etc. The processor 81 may also include various signal processing circuits, etc. The memory 82 is a volatile storage device such as a random access memory (RAM) that updates and sequentially rewrites stored data, while the auxiliary storage device 83 is a nonvolatile storage device such as a flash memory, a read-only memory (ROM), or a hard disk. The auxiliary storage device 83 stores predetermined programs and data such as fixed value data to be executed by the processor 81. The processor 81 reads the programs and fixed value data as appropriate, executes the programs, and performs various arithmetic operations. At this time, the predetermined programs are temporarily stored in the memory 82 from the auxiliary storage device 83, and the processor 81 reads the programs from the memory 82. Arithmetic operations by the various functional units of the boost chopper control unit 17 and the motor control unit 18 are realized by the processor 81 executing the predetermined programs as described above. The results of the arithmetic operations by the processor 81 are temporarily stored in the memory 82 and then stored in the auxiliary storage device 83 according to the purpose of the executed arithmetic operations.

[0058] The boost chopper control unit 17 and the motor control unit 18 also include input / output interfaces with the outside of the device, specifically, an input circuit 84 that receives various inputs from the outside, an output circuit 85 that outputs various types of data to the outside, and a communication device 86 that transmits and receives various types of communication data.

[0059] A part or all of the hardware configuration such as the processor 81 described above may be separate for the boost chopper control unit 17 and the motor control unit 18, or the same hardware may be shared.

[0060] In the conventional full-PAM system, the boost chopper circuit operates at a high frequency, which increases the switching loss of the switching elements, leading to reduced efficiency of the power conversion device. Furthermore, when operating in CCM, recovery loss and switching noise increase, and the input filter circuit connected between the AC power supply and the diode rectifier circuit may become larger. Furthermore, the smoothing capacitor connected between the output terminal of the boost chopper circuit and the input terminal of the inverter must be selected to satisfy the current ripple, as chopped current from the boost chopper circuit and the inverter flows in. Therefore, electrolytic capacitors with high equivalent series resistance (ESR) require an increased number of parallel connections.

[0061] The power conversion device disclosed in Patent Document 1 adds two diodes and one switching element and utilizes a smoothing inductor to boost the DC bus voltage. However, unlike the full-PAM system, it does not have a power factor correction function; power factor correction is achieved by motor control, as in the capacitor-less system. However, in the power conversion device disclosed in Patent Document 1, the high-frequency switching operation for boosting the voltage also causes the rectifier diodes used in the capacitor-less system to repeatedly switch on and off at high frequency. When low-speed diodes are driven at high frequency, recovery loss occurs. To reduce recovery loss, high-speed diodes such as fast recovery diodes (FRDs) must be used, but using these diodes increases costs.

[0062] Furthermore, because Patent Document 1 uses a smoothing inductor for boost operation, an input filter with a high attenuation effect on the harmonic components of normal mode components is required to prevent high-frequency current from leaking into the AC voltage. This poses problems of cost and size. Furthermore, because the boost chopper circuit and high-speed diode also perform hard switching, there is a risk of an increase in common-mode noise, and common-mode noise countermeasures also pose problems of the input filter becoming larger and more costly.

[0063] The power conversion device 10 according to the first embodiment includes a boost chopper circuit 14. The boost chopper circuit 14 stops operating when the AC voltage vs, which is the power supply voltage of the AC power source 91, is relatively low, and operates when the AC voltage vs is relatively high, thereby correcting the power factor and boosting the DC bus voltage VDC. Operating the boost chopper circuit 14 in a low-voltage, low-current range allows the components of the boost chopper circuit 14 to be configured with small-capacity components. Furthermore, in the boost chopper circuit 14, an additional filter capacitor Cx provided between the boost inductor Lx and diodes D5 and D6 removes much of the ripple component of the boost inductor current ix. This prevents a high-frequency current from flowing through the smoothing inductor Lac. This prevents a current in which high-frequency components are superimposed on low-frequency components from flowing through the smoothing inductor Lac, thereby suppressing increases in copper loss and core loss and reducing efficiency associated with boost operation.

[0064] In addition, the configuration is such that ripples are removed using an additional filter capacitor Cx and an existing smoothing inductor Lac, so ripples are removed with a minimum number of additional components, and it is possible to apply slow diodes to the rectifying diodes D1 to D6.

[0065] Furthermore, by controlling the boost inductor current ix flowing through the boost inductor Lx by DCM or CRM, the boost inductor Lx can be made smaller, and the switching loss of the switching element Sx can be reduced. Furthermore, the recovery loss of the diode Dx, which is a high-speed diode, can be suppressed to almost zero. Furthermore, noise (common-mode noise, etc.) generated by switching in the boost chopper circuit 14 can be suppressed, and an increase in the size of the input filter 11 can be avoided.

[0066] Second Embodiment Next, a second embodiment will be described with reference to Fig. 6. Note that components that are the same as or equivalent to those in Figs. 1 to 5 are assigned the same reference numerals, and descriptions thereof will be omitted. Fig. 6 is a diagram showing the configuration of a power conversion device according to the second embodiment. A boost chopper circuit 24 in a power conversion device 20 is provided with a noise suppression inductor Lacx, i.e., a third inductor. The input end of inductor Lacx is connected between the output side of the input filter 11 and the input end of the smoothing inductor Lac, and the output side is connected to the anode of diode D5.

[0067] In addition, the cathode of a diode D7 is connected between the anode of the diode D5 and the output terminal of the inductor Lacx. The anode of the diode D7 is connected to the DC bus DCL on the low potential side. The cathode of a diode D8 is connected between the anode of the diode D6 and the AC input terminal of the rectifier 13 (the connection point between the diodes D3 and D4). The anode of the diode D8 is connected to the DC bus DCL on the low potential side.

[0068] As described above, the same effect as in the first embodiment can be obtained even if the input of the boost chopper circuit 14 is configured between the smoothing inductor Lac and the AC power supply 91. If the input filter 11 has an inductance equivalent to that of the inductor Lacx, the inductor Lacx can be omitted by using the input filter 11 instead of the inductor Lacx. In this case, the anode of the diode D5 is directly connected between the output side of the input filter 11 and the input end of the smoothing inductor Lac.

[0069] Third Embodiment Next, a third embodiment will be described with reference to FIG. 7 . Components identical to or corresponding to those in FIGS. 1 to 6 are designated by the same reference numerals, and their description will be omitted. FIG. 7 is a diagram showing the configuration of a power conversion device according to the third embodiment. The boost chopper circuit 34 of the power conversion device 30 is the boost chopper circuit 14 of the power conversion device 10 without the filter capacitor Cx. In the power conversion device 30, the filter capacitor Cx is connected between the rectifier 13 and the diodes D5 and D6. More specifically, one end of the filter capacitor Cx is connected to an electric path (corresponding to the “first electric path”) between the high-potential AC input terminal of the rectifier 13 (the connection point between the anode of the diode D1 and the cathode of the diode D2) and the anode of the diode D5. Therefore, one end of the filter capacitor Cx is connected to the output terminal of the smoothing inductor Lac, and is connected to the AC power source 91 via the smoothing inductor Lac. The other end of the filter capacitor Cx is connected to the electric path (corresponding to the "second electric path") between the AC input terminal on the low potential side of the rectifier 13 (the connection point between the anode of the diode D3 and the cathode of the diode D4) and the anode of the diode D6. Therefore, the other end of the filter capacitor Cx is connected to the AC power supply 91 without passing through the smoothing inductor Lac.

[0070] In this case, the filter capacitor Cx is located closer to the input side than the diodes D5 and D6 between the smoothing inductor Lac and the boost inductor Lx. Meanwhile, the filter capacitor Cx is located closer to the output side than the smoothing inductor Lac and the diodes D1 to D4. Thus, in the third embodiment, since there is no filter capacitor Cx between the diodes D5 and D6 and the boost inductor Lx, it is inevitable that ripples in the boost inductor current ix will flow into the diodes D5 and D6. However, the flow of ripples into the smoothing inductor Lac is suppressed in the same way as in the first embodiment. Furthermore, since the flow of ripples into the diodes D1 to D4 is also suppressed, slow diodes can be used for the diodes D1 to D4.

[0071] 8 is a diagram showing the configuration of a power conversion device according to a modification of the third embodiment. A boost chopper circuit 341 of the power conversion device 301 is configured by omitting the filter capacitor Cx from the boost chopper circuit 14 of the power conversion device 10. In the power conversion device 301, the filter capacitor Cx is connected between the smoothing inductor Lac and the rectifier 13. More specifically, one end of the filter capacitor Cx is connected to an electric path (corresponding to a "first electric path") between the output terminal of the smoothing inductor Lac and the high-potential side AC input terminal of the rectifier 13. Therefore, one end of the filter capacitor Cx is connected to the output terminal of the smoothing inductor Lac and is connected to the AC power supply 91 via the smoothing inductor Lac. Furthermore, the other end of the filter capacitor Cx is connected to an electric path (corresponding to a "second electric path") between the low-potential side output terminal of the AC power supply 91 and the low-potential side AC input terminal of the rectifier 13. Therefore, the other end of the filter capacitor Cx is connected to the AC power supply 91 without passing through the smoothing inductor Lac. In the power conversion device 301, the filter capacitor Cx is located between the smoothing inductor Lac and the boost inductor Lx and is closer to the output side than the smoothing inductor Lac. Therefore, the flow of ripples into the smoothing inductor Lac is suppressed in the same way as in the first embodiment.

[0072] Although various exemplary embodiments and examples are described in this disclosure, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless variations not illustrated are anticipated within the scope of the technology disclosed in this specification. For example, this includes cases where at least one component is modified, added, or omitted, or where at least one component is extracted and combined with components of another embodiment.

[0073] For example, the connection position of the filter capacitor Cx may be between the connection point of the cathodes of the diodes D5 and D6 and the cathode of the diode D5. That is, one end of the filter capacitor Cx may be connected to the electrical path between the connection point and the cathode of the diode D5, and the other end of the filter capacitor Cx may be connected to the low-potential side DC bus DCL.

[0074] 10, 20, 30, 301 Power conversion device, 11 Input filter, 12 Smoothing inductor section, 13 Rectifier, 14, 24, 34, 341 Boost chopper circuit, 15 Smoothing capacitor section, 16 Inverter, 17 Boost chopper control section, 18 Motor control section, 91 AC power supply, 92 Motor, 172 Boost chopper operation determination section, 173 DCM calculation section, 174 CRM calculation section, 175 Mode determination section, Cdc Smoothing capacitor, Cx Filter capacitor, D1 to D8, Dx Diode, DCH High potential side DC bus, DCL Low potential side DC bus, iinv Input current, irec1, irec2 Current, is AC current, ix Boost inductor current, Lac Smoothing inductor, Lacx Inductor, Lx Boost inductor, pm Motor power, ps AC power supply power, S1 to S6, Sx switching element, T1 boost chopper circuit stop period, T2 boost chopper circuit operation period, Vdc DC bus voltage, vs AC voltage, Vx filter capacitor voltage

Claims

1. A first inductor receives alternating current from an AC power source, A boost chopper circuit having a second inductor connected to the output side of the first inductor, and a switching element connected between a high-potential DC bus and a low-potential DC bus on the output side of the second inductor, A rectifier is provided between the first inductor and the boost chopper circuit, having a first diode whose anode is connected to the output terminal of the first inductor and whose cathode is connected to the high-potential DC bus, On the output side of the boost chopper circuit, a first capacitor is connected between the high-potential DC bus and the low-potential DC bus, An inverter that converts the voltage across the first capacitor into AC and supplies AC output power to a motor, A power conversion device characterized by comprising a second capacitor, one end of which is connected to a first circuit connecting the output terminal of the first inductor and the input terminal of the second inductor, and the other end of which is connected to a second circuit connected in parallel with the first circuit.

2. The power conversion device according to claim 1, wherein the boost chopper circuit comprises a second diode whose anode is connected to the output terminal of the first inductor and whose cathode is connected to the input terminal of the second inductor, and the second capacitor has one end connected between the second diode and the second inductor and the other end connected to the DC bus on the low potential side.

3. The power conversion device according to claim 2, wherein a third inductor is provided, the input terminal of which is connected between the input terminal of the first inductor and the AC power supply, and the output terminal of which is connected to the anode of the second diode.

4. The power conversion device according to claim 1, wherein one end of the second capacitor is connected to the output terminal of the first inductor, and the other end is connected to the AC power supply without going through the first inductor.

5. The power conversion device according to any one of claims 1 to 4, wherein the power flow path from the AC power source to the motor during the operation period of the boost chopper circuit passes through the boost chopper circuit, and the power flow path during the stop period of the boost chopper circuit passes through the first diode.

6. The power conversion device according to claim 5, wherein the boost chopper circuit has the stop period and the operating period during one cycle of the alternating current.

7. The power conversion device according to any one of claims 1 to 4, wherein the boost chopper circuit stops when the voltage command value of the terminal voltage of the first capacitor is greater than a threshold, and operates when the voltage command value is less than or equal to the threshold.

8. The power conversion device according to any one of claims 1 to 4, wherein the boost chopper circuit operates with either a DCM or a CRM, or both.

9. The first capacitor has a capacitance that pulsates at twice the frequency of the alternating current. The power conversion device according to any one of claims 1 to 4, wherein the inverter varies the output power in synchronization with the AC power of the AC power supply.

10. The power conversion device according to any one of claims 1 to 4, wherein an input filter comprising at least one of a smoothing capacitor, a normal mode filter, and a common mode filter is provided between the AC power supply and the first inductor.

11. The power converter according to any one of claims 1 to 4, wherein the power capacity of the boost chopper circuit is smaller than the rated power capacity of the power converter.