Ac / DC conversion device, rotary machine drive device, and refrigeration cycle application apparatus

Abstract

An AC / DC conversion device (2) comprises: a rectifier circuit (20) that has a switching element (215) and that rectifies the power supply voltage applied from an AC power supply (1); a capacitor (216) that is connected to DC busbars (9a, 9b) and that smooths the output voltage of the rectifier circuit (20); a reactor (212) that is disposed further to the AC power supply (1) side than the capacitor (216); and a control unit (6) that uses a carrier signal to generate a switching signal for controlling the switching element (215). A carrier frequency that is the frequency of the carrier signal does not fall below a lower limit value, and is changed within a half cycle of the power supply voltage such that the harmonic component included in the power supply current flowing between the AC power supply (1) and the rectifier circuit (20) conforms to the harmonic standard value of the power supply current.

Description

AC / DC converters, rotating machine drives, and refrigeration cycle application equipment

[0001] The present disclosure relates to an AC / DC converter that converts AC power into desired DC power, and to a rotating machine drive device and refrigeration cycle equipment that are equipped with the AC / DC converter.

[0002] A converter circuit is commonly used to obtain DC voltage from an AC power source. The converter circuit maintains a constant bus voltage and controls the power supply current to comply with harmonic specifications. A converter circuit and its control method, the "simple switching method" (also known as the "partial switching method"), performs switching at least once per half cycle of the power supply voltage, which is the voltage of the AC power supply. This method has the advantage of being able to control the bus voltage lower than the peak value of the power supply voltage. However, when the bus voltage is set lower than the peak value of the power supply voltage in the simple switching method, the operating circuit switches from a boost chopper to a capacitor-input diode rectifier, which creates the problem of distorting the power supply current.

[0003] To address this issue, the conventional technology disclosed in Patent Document 1 below achieves this by repeating the design for each load power to determine whether the combination of reactor capacity and switching timing complies with the harmonic standard values.

[0004] Japanese Patent Application Laid-Open No. 2000-125545

[0005] However, the conventional technology described in Patent Document 1 involves a method of checking whether compliance with harmonic standard values ​​can be achieved through repeated trials, which has the problem that the number of trials increases exponentially as the number of pulses increases.Furthermore, there is no clear guideline for quantitatively and uniquely designing control gains, which has the problem that it takes a long time to complete the design.

[0006] Therefore, instead of a trial-and-error approach, a control method that feeds back current and voltage has been considered. For example, a PS (Proportional Sinusoidal) controller is devised, in which a P (Proportional) controller and an S (Sinusoidal transfer function) controller that has good tracking ability for sinusoidal command are connected in parallel. This method eliminates the need for a trial-and-error approach, making it possible to shorten the time required for design work.

[0007] There is a view that the carrier frequency should be as low as possible if the power supply current can comply with harmonic specifications. This is because the lower the carrier frequency, the less switching loss there is in the converter circuit. However, in the case of a PS controller, the controller's operation follows a sinusoidal command, so a low carrier frequency can cause the controller's operation to become unstable, and in the worst case, the control operation can fail. Note that this problem is not limited to PS controllers, but can also occur in controllers such as proportional integral (PI) controllers.

[0008] The present disclosure has been made in consideration of the above, and aims to provide an AC / DC conversion device that can comply with harmonic standard values ​​while shortening the time required for design work and stabilizing control operations.

[0009] In order to solve the above-mentioned problems and achieve the object, an AC-DC converter according to the present disclosure includes a rectifier circuit, a capacitor, a reactor, and a control unit. The rectifier circuit has at least one switching element and rectifies a power supply voltage applied from an AC power supply. The capacitor is connected to a DC bus and smooths the output voltage of the rectifier circuit. The reactor is arranged closer to the AC power supply than the capacitor, and the switching element is arranged closer to the AC power supply than the capacitor. The control unit generates a switching signal for controlling the switching element using a carrier signal. The carrier frequency, which is the frequency of the carrier signal, is changed within a half cycle of the power supply voltage so that it does not fall below a lower limit and so that harmonic components contained in the power supply current flowing between the AC power supply and the rectifier circuit comply with the harmonic standard value for the power supply current.

[0010] The AC-DC converter according to the present disclosure has the effect of being able to comply with harmonic standard values ​​while shortening the time required for design work and stabilizing control operations.

[0011] 7 is a block diagram illustrating an example of the configuration of a rotating machine drive device according to embodiment 1; FIG. 8 is a diagram illustrating an example of the configuration of an AC / DC converter according to embodiment 1; FIG. 9 is a block diagram illustrating an example of the configuration of a control unit according to embodiment 1; FIG. 10 is a block diagram illustrating an example of the configuration of a voltage control unit according to embodiment 1; FIG. 11 is a block diagram illustrating an example of the configuration of a current control unit according to embodiment 1; FIG. 8 is a diagram showing a second frequency variable pattern in a simulation performed using the current control unit shown in FIG. 7; FIG. 9 is a diagram showing an example of current harmonic characteristics when a PI controller is applied to the current control unit shown in FIG. 7; FIG. 10 is a diagram showing an example of current harmonic characteristics when a PS controller is applied to the current control unit shown in FIG. 7; FIG. 11 is a diagram showing an example of a configuration of an AC / DC converter according to embodiment 2; FIG. 12 is a diagram showing an example of a configuration of an AC / DC converter according to embodiment 3; FIG. 13 is a diagram showing an example of a configuration of an AC / DC converter according to embodiment 4;

[0012] DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An AC-DC converter, a rotary machine drive device, and a refrigeration cycle application device according to embodiments of the present disclosure will be described in detail below with reference to the accompanying drawings.

[0013] Embodiment 1. Fig. 1 is a block diagram showing an example of the configuration of a rotating machine driving device 8 according to Embodiment 1. The rotating machine driving device 8 is connected to an AC power source 1 and a load 4 including a motor 41. The rotating machine driving device 8 includes an AC-DC converter 2 and a DC-AC converter 3. When the rotating machine driving device 8 is used in an air conditioner, the load 4 is a compressor or a fan, and the motor 41 is a compressor motor or a fan motor.

[0014] 2 is a diagram showing an example of the configuration of an AC-DC converter 2 according to the first embodiment. The AC-DC converter 2 according to the first embodiment mainly includes a control unit 6, a rectifier circuit 20, a reactor 212, and a capacitor 216. The AC-DC converter 2 also includes a current detection unit 211 and voltage detection units 217a and 217b as means for detecting voltage or current. In this document, when the voltage detection units 217a and 217b are distinguished from each other without reference numerals, the voltage detection unit 217b will be referred to as a "first voltage detection unit" and the voltage detection unit 217a will be referred to as a "second voltage detection unit."

[0015] The rectifier circuit 20 includes single-phase diode bridge cells 213a and 213b, each of which has four diodes bridge-connected, and a switching element 215 connected in parallel across the single-phase diode bridge cell 213b. The single-phase diode bridge cells 213a and 213b are connected in parallel to each other across the AC power supply 1. The rectifier circuit 20 shown in FIG. 2 is called a "simple switching circuit." The single-phase diode bridge cell 213b and the switching element 215 constitute a switching cell 225. The switching element 215 performs a switching operation at least once per half cycle of the power supply voltage.

[0016] The capacitor 216 is connected between the DC bus 9a and the DC bus 9b. The reactor 212 is arranged closer to the AC power supply 1 than the capacitor 216. The rectifier circuit 20 receives the power supply voltage applied from the AC power supply 1 via the reactor 212 and rectifies the received power supply voltage. The capacitor 216 smoothes the output voltage of the rectifier circuit 20.

[0017] The voltage detector 217b detects the bus voltage, which is the voltage of the DC buses 9a and 9b to which the capacitor 216 is connected. The voltage detector 217a detects the power supply voltage. The current detector 211 detects the power supply current flowing between the AC power supply 1 and the rectifier circuit 20.

[0018] The control unit 6 receives the detection values ​​of the voltage detection units 217a and 217b and the current detection unit 211. The control unit 6 generates a switching signal for controlling the on / off of the switching element 215 based on the detection values.

[0019] An example of the switching element 215 is an insulated gate bipolar transistor (IGBT) as shown in the figure, but is not limited to an IGBT. Any element capable of switching operation may be used as the switching element 215. Another example of the switching element 215 is a metal-oxide-semiconductor field-effect transistor (MOSFET).

[0020] 2 is configured as a closed loop using the detected values ​​of the voltage detectors 217a, 217b and the current detector 211, but may be configured as an open loop using a target value, an estimated value, etc. When the AC-DC converter 2 is configured as an open loop, it is also possible to control the switching element 215 without using the detected values ​​of the voltage detectors 217a, 217b and the current detector 211.

[0021] 3 is a block diagram showing an example configuration of the control unit 6 according to the first embodiment. The control unit 6 includes a voltage control unit 61, a current control unit 62, and a switching signal generation unit 63. The voltage control unit 61 generates a first current command value using a first voltage command value. The current control unit 62 generates a second voltage command value using the first current command value. The switching signal generation unit 63 generates a switching signal using the second voltage command value.

[0022] 4 is a block diagram showing an example configuration of the voltage control unit 61 according to the first embodiment. The voltage control unit 61 includes a voltage controller 611 and a subtractor 612. The voltage control unit 61 generates a first current command value using a first voltage command value that is a command value for the bus voltage. More specifically, the subtractor 612 generates a voltage deviation that is the difference between the first voltage command value and the voltage detected by the voltage detection unit 217b. The voltage controller 611 generates the first current command value using the voltage deviation output from the subtractor 612. The voltage controller 611 can be configured, for example, by a PI controller.

[0023] The transfer function G when the voltage controller 611 is configured as a PI controller ＡＶＲ（ｓ） can be expressed by the following equation (1).

[0024]

[0025] Here, the transfer function G ＡＶＲ（ｓ） "AVR" in the above is an abbreviation for "Automatic Voltage Regulator." Also, in the above formula (1), K ｐＡＶＲ is the proportional gain, and K ｉＡＶＲ is the integral gain and s is the Laplace operator. In the PI controller, the proportional gain K ｐＡＶＲ and integral gain K ｉＡＶＲ can be determined arbitrarily. ｐＡＶＲ may be set to zero and configured as an I (Integral) controller, or an integral gain K ｉＡＶＲ may be set to zero and configured as a P controller.

[0026] FIG. 5 is a block diagram showing an example configuration of the current control unit 62 according to the first embodiment. The current control unit 62 includes a current controller 621, a subtractor 622, and a multiplier 623. The current control unit 62 generates a second voltage command value using a second current command value obtained by causing the first current command value to follow a sine wave. More specifically, the multiplier 623 multiplies the first current command value by an excitation signal. The excitation signal is a sine wave synchronized with the phase of the power supply voltage. The sine wave is generated based on the voltage phase, which is the phase of the detected voltage detected by the voltage detection unit 217a. The output of the multiplier 623 is input to the subtractor 622 as a second current command value. The subtractor 622 generates a current deviation, which is the difference between the second current command value and the detected current detected by the current detection unit 211. The current controller 621 generates the second voltage command value using the current deviation output from the subtractor 622. The current controller 621 can be configured as, for example, a PS controller.

[0027] Fig. 6 is a block diagram showing an example of the configuration when the current controller 621 shown in Fig. 5 is configured as a PS controller. As shown in Fig. 6, the current controller 621 is a controller configured such that a P controller 621A and an S controller 621B are connected in parallel, and the outputs of the P controller 621A and the S controller 621B are added by an adder 621C and output. The S controller 621B outputs a signal having an angular frequency ω ｎ In the S controller 621B, the angular frequency ω ｎ The reason why the tracking performance improves for a pulsating input can be explained by the internal model principle. The internal model principle states that if the controller's denominator has the same factor as the denominator polynomial of the command value expressed in Laplace transform, the command value can be tracked without deviation. Note that the current controller 621 may be configured as a PIS (Proportional Integral Sinusoidal) controller by inserting an I controller in parallel with the PS controller.

[0028] The transfer function G when the current controller 621 is configured as a PS controller ＡＣＲ（ｓ） can be expressed by the following equation (2).

[0029]

[0030] Here, the transfer function G ＡＣＲ（ｓ） "ACR" in the above expression is an abbreviation for "Automatic Current Regulator." ｐＡＣＲ is the proportional gain, and K ｓＡＣＲ is the S control gain, and ω ｎ is the angular frequency and s is the Laplace operator. In the PS controller, the proportional gain K ｐＡＣＲ , S control gain K ｓＡＣＲ and angular frequency ω ｎ can be determined arbitrarily.

[0031] Fig. 7 is a block diagram showing a configuration example in which a target value filter 624 is introduced into the current control unit 62 shown in Fig. 5. Fig. 8 is a block diagram used to explain the transfer function of the current control system 7 in the AC-DC converter 2 including the current control unit 62 shown in Fig. 7.

[0032] In Fig. 7, a target value filter 624 is inserted in the preceding stage of the current control unit 62 shown in Fig. 5. The target value filter 624 is inserted in order to adjust the response of the transfer function of the current control unit 62 shown in Fig. 5. Specifically, the response of the transfer function of the current control unit 62 is adjusted by ensuring that the zero points of the transfer function of the current control unit 62 are canceled out by the poles of the target value filter 624.

[0033] Here, the closed loop transfer function G in the current control system 7 of FIG. ｃｌｏｓｅ（ｓ） can be expressed by the following equation (3).

[0034]

[0035] In the above formula (3), G ｃ is the transfer function of any controller 72, and G Ｐ is the transfer function of an arbitrary controlled plant 73. If the above equation (3) has a zero point, the closed-loop response cannot be determined by the poles alone. Therefore, the closed-loop transfer function G ｃｌｏｓｅ（ｓ） A target value filter 74 having the same pole as the zero point of is inserted in the front stage of the closed loop. Then, the transfer function G Ｆ（ｓ） and the closed loop transfer function G ｃｌｏｓｅ（ｓ）The desired response is realized by canceling out the influence of the zero point between the target value filter 74 and the transfer function G′. ｃｌｏｓｅ（ｓ） can be expressed by the following equation (4).

[0036]

[0037] In the above formula (4), G Ｘ is an arbitrary transfer function. Ｘ is the transfer function G' ｃｌｏｓｅ（ｓ） The transfer function G may be set to the value of the zeroth power of the Laplace operator s in the denominator polynomial of the above equation, or may have any zero point. Ｆ（ｓ） The controller response can be adjusted by applying the target value filter 74 having the following formula: Note that a PI controller may also be used as a controller that achieves the same effect as the target value filter 74.

[0038] Here, the transfer function G of the target value filter 74 Ｆ（ｓ） (=G Ｘ / (G Ｃ G Ｐ ) can be expressed, for example, by the following equation (5).

[0039]

[0040] Moreover, the above formula (2) can be expressed as the following formula (6).

[0041]

[0042] Therefore, by multiplying the above equation (5) by the above equation (6), it is clear that the zeros of equation (6) are canceled out by the poles of equation (5). In this way, pole-zero cancellation can be achieved by inserting the target value filter 74 in the front stage of the closed loop.

[0043] 9 is a block diagram showing an example configuration of switching signal generation unit 63 according to embodiment 1. Switching signal generation unit 63 includes a normalizer 630, a carrier signal generator 631, and a comparator 632. Normalizer 630 normalizes the second voltage command value input to switching signal generation unit 63. Note that if the control gain of current controller 621 that outputs the second voltage command value is designed taking normalization into consideration, normalizer 630 becomes unnecessary.

[0044] The carrier signal generator 631 generates a carrier signal used to generate a switching signal and outputs it to the comparator 632. The carrier signal is a bipolar or unipolar signal having a triangular waveform that operates at a given frequency. The comparator 632 generates a switching signal based on the output of the scaler 630 and the output of the carrier signal generator 631. More specifically, the comparator 632 compares the output value of the scaler 630 with the output value of the carrier signal generator 631, and if the output value of the scaler 630 is greater than the output value of the carrier signal generator 631, it outputs a signal to turn on the switching element 215. On the other hand, if the output value of the scaler 630 is smaller than the output value of the carrier signal generator 631, the comparator 632 outputs a signal to turn off the switching element 215. Conversely to this operation, a configuration may be adopted in which an ON operation signal is generated when the output value of the normalizer 630 is smaller than the output value of the carrier signal generator 631, and an OFF operation signal is generated when the output value of the normalizer 630 is larger than the output value of the carrier signal generator 631. Furthermore, the carrier signal generated by the carrier signal generator 631 does not have to be triangular wave-shaped, and may have any shape as long as it can generate a switching signal from the second voltage command value.

[0045] Furthermore, in the first embodiment, the carrier signal generator 631 changes the carrier frequency, which is the frequency of the carrier signal, within one cycle of the power supply voltage so that the harmonic components contained in the power supply current comply with the harmonic standard value of the power supply current. In this document, this control is referred to as "carrier frequency variable control" as appropriate. By performing the carrier frequency variable control, it is possible to disperse the harmonic components contained in the power supply current. Furthermore, by employing the carrier frequency variable control, it is possible to suppress an increase in the number of switching operations in the rectifier circuit 20. Note that the details of the carrier frequency variable control in the first embodiment will be described later.

[0046] Next, the operating characteristics when PS control is applied to the current controller 621 will be clarified by comparing them with the conditions when PI control is applied to the current controller 621.

[0047] Fig. 10 is a diagram showing example operating waveforms of the power supply voltage, bus voltage, and power supply current when PI control is applied to the current controller 621 shown in Fig. 7. The upper part of Fig. 10 shows waveforms of the absolute values ​​of the bus voltage and power supply voltage. The lower part of Fig. 10 shows waveforms of the detected power supply current, the fundamental wave component of the detected power supply current, and the power supply current command value. The detected power supply current is the detected waveform of the power supply current detected by the current detection unit 211.

[0048] The operating conditions in Fig. 10 are those in which the bus voltage is equal to or less than the peak absolute value of the power supply voltage, as shown in the upper part of Fig. 10. Under these conditions, the rectifier circuit 20 operates as a capacitor-input diode rectifier circuit rather than as a boost circuit, making it impossible to control the bus voltage. In this case, an excessive value accumulates in the PI control integrator, causing a windup phenomenon, and the detected power supply current cannot follow the power supply current command value.

[0049] In contrast, Fig. 11 is a diagram showing example operating waveforms of the power supply voltage, bus voltage, and power supply current when PS control is applied to the current controller 621 shown in Fig. 7. As with Fig. 10, the upper part of Fig. 11 shows waveforms of the absolute values ​​of the bus voltage and power supply voltage, and the middle part of Fig. 11 shows waveforms of the detected power supply current, the fundamental wave component of the detected power supply current, and the power supply current command value. The lower part of Fig. 11 shows a switching signal for controlling the switching element 215.

[0050] As shown in the middle part of Figure 11, the fundamental wave component of the detected power supply current is nearly identical to the power supply current command value. This shows that if PS control, which has high tracking performance for a sinusoidal input, is applied to the current controller 621, the fundamental wave of the detected power supply current can track the power supply current command value even under conditions in which the bus voltage is equal to or less than the peak absolute value of the power supply voltage. Note that Figure 11 shows the results for operating conditions in which the bus voltage is equal to or less than the peak absolute value of the power supply voltage, but it goes without saying that the fundamental wave of the detected power supply current will track the power supply current command value even under operating conditions in which the bus voltage exceeds the peak absolute value of the power supply voltage.

[0051] One of the features of the control technique of the first embodiment is that, in addition to applying a current controller 621 to the control unit 6, PS control, which has high tracking performance for a sine wave input, is further applied to the current controller 621. While the application of PS control to improve tracking performance for a sine wave input is considered to be a relatively common practice, the application of PS control to comply with harmonic standard values ​​is considered to be a novel technique that has never been used before.

[0052] Another feature of the control technique of the first embodiment is the adoption of variable carrier frequency control, as described above. Variable carrier frequency control in the first embodiment will be described below. In the following description, a case where the power supply frequency, which is the frequency of the power supply voltage, is 50 Hz will be exemplified. Note that the same description can be applied to a case where the power supply frequency is 60 Hz by simply replacing the numerical values.

[0053] If the carrier frequency is fixed, for example, at 2 kHz, switching occurs a maximum of 20 times during a half cycle of the power supply voltage. In this case, the dominant harmonic component contained in the power supply current is 2 kHz. This 2 kHz frequency corresponds to the 40th harmonic component of a 50 Hz power supply frequency (= 50 x 40 = 2 kHz). Therefore, if the carrier frequency is fixed at 2 kHz, it will require a great deal of effort to make the 40th harmonic component of the power supply current comply with the harmonic specification value. Note that if the carrier frequency is fixed, the same problem will occur even if the carrier frequency is changed to another frequency below the 40th harmonic.

[0054] Furthermore, when the carrier frequency is fixed, it is possible to set the carrier frequency to a frequency higher than the 40th order of the power supply frequency. However, when the carrier frequency is high, the number of times that the switching element 215 switches increases, which causes another problem of increased switching loss and switching noise.

[0055] On the other hand, when variable carrier frequency control is adopted, the carrier frequency components can be dispersed, which makes it possible to reduce switching loss and switching noise while also reducing the effort required to make the harmonic components of the power supply current comply with harmonic standard values.

[0056] Next, a specific method for generating a carrier signal that can realize carrier frequency variable control will be described with reference to Fig. 12 . Fig. 12 is a diagram illustrating the method for generating a carrier signal in the first embodiment. Fig. 12 shows how the carrier count changes when a carrier signal is generated using an up-down counter. The horizontal axis of Fig. 12 represents time. The up-down counter is one of the carrier signal generation methods used in FPGAs (Field-Programmable Gate Arrays) and the like. The carrier count is a count value set by the up-down counter.

[0057] The carrier signal generator 631 increases the count value by one for each clock up to the maximum carrier count crpk set according to the carrier frequency. After the count value reaches the maximum value crpk, the carrier signal generator 631 decreases the count value by one for each clock. In this way, the carrier signal generator 631 generates a triangular wave carrier signal that operates at a desired frequency by setting the maximum carrier count crpk and increasing or decreasing the carrier count value by one. Here, the maximum carrier count crpk is defined by the following equation (7):

[0058]

[0059] In the above formula (7), t ｃｌｋ is the clock tick time, and f ｓｗ is the desired switching frequency.

[0060] Furthermore, in order to set an arbitrary carrier frequency, the carrier signal generator 631 sets the maximum carrier count crpk in the next control cycle at the carrier top, which is the peak of the carrier count. Here, the maximum carrier count crpk corresponding to the set carrier frequency is calculated using the following equation (8).

[0061]

[0062] In the above formula (8), f ｓｅｌｅｃｔ is the set value of the carrier frequency corresponding to the desired carrier frequency. As can be seen from FIG. ｓｗ +f ｓｅｌｅｃｔ The maximum value crpk of the carrier count changes depending on the value of , and the carrier period also changes accordingly. Therefore, by changing the maximum value crpk of the carrier count using equation (8), it is possible to change the carrier frequency.

[0063] There are various possible patterns for changing the carrier frequency within one cycle of the power supply voltage. However, as explained in the section "Problems to be Solved by the Invention," if the carrier frequency is too low, the control operation of the control unit 6 becomes unstable, and in the worst case, the control operation of the control unit 6 may fail. To address this issue, the inventors performed carrier frequency variable control while setting a lower limit value for the carrier frequency, and performed simulations using several frequency variable patterns to determine whether the harmonic components contained in the power supply current complied with the harmonic standard value for the power supply current. Based on the simulation results, the inventors found that there is a preferable lower limit value for the carrier frequency. This lower limit value will be explained below.

[0064] Fig. 13 is a diagram showing a first frequency variable pattern in a simulation performed using the control unit 6 according to embodiment 1. Fig. 14 is a diagram showing a second frequency variable pattern in a simulation performed using the control unit 6 according to embodiment 1.

[0065] In both the first and second frequency variable patterns shown in FIGS. 13 and 14, the carrier frequency is varied between 0 and 0.01 [s], which is a half cycle of the power supply voltage. ｃｒ１ →Second frequency f ｃｒ２ →Third frequency f ｃｒ３ →1st frequency f ｃｒ１ The difference between the two is that the lower limit of the first frequency f ｃｒ１ 13 is set to 150 [Hz], whereas in the second frequency variable pattern shown in FIG. 14 it is set to 140 [Hz]. ｃｒ２ and the third frequency f ｃｒ３ is the same value for both.

[0066] The inventors of the present application have determined that the second frequency f ｃｒ２ and the third frequency f ｃｒ３ and the first frequency f ｃｒ１When simulations were performed by changing the carrier frequency lower limit, it was found that the operation of the control unit 6 did not become unstable if the lower limit of the carrier frequency was set to 150 Hz, as shown in Figure 13. On the other hand, when the lower limit of the carrier frequency was set to less than 150 Hz, as shown in Figure 14, some of the simulation results showed that the operation of the control unit 6 became unstable. As explained in the section [Problem to be Solved by the Invention], this result is thought to be due to the fact that in the case of a PS controller, the operation of the controller follows a sinusoidal wave command, and therefore the operation of the controller becomes unstable at a low carrier frequency.

[0067] Fig. 15 is a diagram showing an example of current harmonic characteristics when a PI controller is applied to the current control unit 62 shown in Fig. 7. Fig. 16 is a diagram showing an example of current harmonic characteristics when a PS controller is applied to the current control unit 62 shown in Fig. 7. In both figures, the horizontal axis indicates the harmonic order, and the vertical axis indicates the effective value of the current harmonic.

[0068] The current harmonic standard used in Figures 15 and 16 is IEC 61000-3-2 Class A. In Figures 15 and 16, the current harmonic standard values ​​from 2nd to 40th shown in IEC 61000-3-2 Class A are shown by dashed lines. Note that IEC 61000-3-2 Class A is an example of a current harmonic standard, and the standard is not limited to this standard.

[0069] 15 and 16, the effective values ​​of the second to fortieth order current harmonics during rated operation are shown by the solid line, the dashed-dot line, and the dashed-double-dot line. The harmonic components during rated operation represent the remaining components remaining after subtracting the fundamental wave component of the power supply current from the power supply current that flows when the AC-DC converter 2 is operated at rated power. In this paper, the second to fortieth order harmonics are referred to as "low-order harmonics."

[0070] In Fig. 15, the two-dot chain line represents the case where the lower limit of the carrier frequency is 150 [Hz], the one-dot chain line represents the case where the lower limit of the carrier frequency is 190 [Hz], and the solid line represents the case where the lower limit of the carrier frequency is 200 [Hz]. In Fig. 15, all of the low-order harmonics from the 2nd to the 40th order meet the standard values ​​only when the lower limit is 200 [Hz]. Note that, although not shown in the results of Fig. 15, when the lower limit of the carrier frequency was 150 [Hz], the control operation of the control unit 6 was unstable.

[0071] 16, the dashed-two-dot line represents the case where the lower limit of the carrier frequency is 100 Hz, the dashed-dot line represents the case where the lower limit of the carrier frequency is 140 Hz, and the solid line represents the case where the lower limit of the carrier frequency is 150 Hz. In FIG. 16, all of the low-order harmonics from the 2nd to the 40th order meet the standard values ​​only when the lower limit is 150 Hz. Although not shown in the results of FIG. 16, when the lower limit of the carrier frequency is 100 Hz, the control operation of the control unit 6 was unstable.

[0072] The results of Figures 15 and 16 can explain the following: (a1) In both PI control and PS control, if the lower limit of the carrier frequency is too low, the operation of the control unit 6 becomes unstable. (a2) In the case of PI control, if the lower limit of the carrier frequency is 190 [Hz] or higher, the operation of the control unit 6 is considered to be stable. (a3) ​​In the case of PS control, if the lower limit of the carrier frequency is 140 [Hz] or higher, the operation of the control unit 6 is considered to be stable. (a4) In both PI control and PS control, if the lower limit of the carrier frequency is too low, some low-order harmonics will not meet the harmonic specification values. (a5) In the case of PI control, if the lower limit of the carrier frequency is 200 [Hz] or higher, it is considered that all low-order harmonics will meet the harmonic specification values. (a6) In the case of PS control, if the lower limit of the carrier frequency is 150 [Hz] or higher, it is considered that all low-order harmonics will meet the harmonic specification values.

[0073] As described above, the AC-DC converter according to the first embodiment includes a rectifier circuit that rectifies a power supply voltage applied from an AC power supply, a capacitor that smooths the output voltage of the rectifier circuit, and a reactor that is located closer to the AC power supply than the capacitor. The rectifier circuit has at least one switching element that is located closer to the AC power supply than the capacitor, and a control unit generates a switching signal for controlling the switching element using a carrier signal. The carrier frequency of the carrier signal is changed within a half cycle of the power supply voltage so that harmonic components contained in the power supply current flowing between the AC power supply and the rectifier circuit comply with the harmonic standard value of the power supply current. This spreads the harmonic components contained in the power supply current across a frequency band of low-order harmonics from the second to the fortieth order, thereby shortening the time required for design work. Furthermore, the AC-DC converter according to the first embodiment changes the carrier frequency so that it does not fall below a predetermined lower limit, thereby stabilizing the operation of the control unit. Therefore, the AC-DC converter according to the first embodiment enables compliance with the harmonic standard value while stabilizing control operation.

[0074] In order to realize the above functions, the control unit according to the first embodiment is configured to include a voltage control unit that generates a first current command value using a first voltage command value that is a command value for the bus voltage, a current control unit that generates a second voltage command value using a second current command value that causes the first current command value to follow a sine wave, and a switching signal generation unit that generates a switching signal using the second voltage command value.

[0075] Furthermore, the control unit according to the first embodiment sets a lower limit for the carrier frequency that is changed within a half cycle of the power supply voltage, and changes the carrier frequency so that it does not fall below the set lower limit. The lower limit is preferably set to 200 Hz or greater. If the lower limit is 200 Hz or greater, it is possible to obtain an AC / DC converter that complies with harmonic specifications while shortening the time required for design work and stabilizing control operation, regardless of whether a PI controller or a PS controller is used in the current control unit.

[0076] Furthermore, in the control unit according to embodiment 1, if the controller applied to the current control unit is a PS controller, the lower limit may be set to 150 Hz or more. In the case of a PS controller, the operation of the controller follows a sinusoidal wave command, so the lower limit can be set even smaller than in the case of a PI controller.

[0077] Embodiment 2 In embodiment 2, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0078] Fig. 17 is a diagram showing a configuration example of an AC-DC converter 2 according to embodiment 2. In the AC-DC converter 2 shown in Fig. 17, a rectifier circuit 20 is configured with a single-phase H-bridge cell including four switching elements 220a, 220b, 220c, and 220d. Note that the configuration and operation of the rectifier circuit 20 shown in Fig. 17 are publicly known, and further description thereof will be omitted here.

[0079] The control unit 6 generates switching signals to drive the four switching elements 220a, 220b, 220c, and 220d using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in FIG. 17 can also achieve the same effects as those of embodiment 1.

[0080] 17, the switching elements 220a, 220b, 220c, and 220d are shown as IGBTs, but any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 17 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detected values ​​of the voltage detectors 217a and 217b and the current detector 211 do not need to be used.

[0081] Embodiment 3 In embodiment 3, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0082] FIG. 18 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a third embodiment. In the AC-DC converter 2 of FIG. 18, a rectifier circuit 20 is configured with a single-phase H-bridge cell including two diodes 218a and 218b and two switching elements 220c and 220d. In the rectifier circuit 20 illustrated in FIG. 18, one leg is configured with a series circuit of the diodes 218a and 218b, and the other leg is configured with a series circuit of the switching elements 220c and 220d. The configuration and operation of the rectifier circuit 20 illustrated in FIG. 18 are publicly known, and therefore further description thereof will be omitted here.

[0083] The control unit 6 generates switching signals to drive the two switching elements 220c and 220d using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 18 can also achieve the same effects as those of embodiment 1.

[0084] 18, the switching elements 220c and 220d are shown as IGBTs, but any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 18 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 217a and 217b and the current detector 211 do not need to be used.

[0085] Embodiment 4 In embodiment 4, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0086] FIG. 19 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a fourth embodiment. In the AC-DC converter 2 of FIG. 19, a rectifier circuit 20 is configured with a single-phase H-bridge cell including two diodes 218a and 218c and two switching elements 220b and 220d. In the rectifier circuit 20 illustrated in FIG. 19, the diodes 218a and 218c are arranged in the upper arms of two legs, and the switching elements 220b and 220d are arranged in the lower arms of two legs. Note that the configuration and operation of the rectifier circuit 20 illustrated in FIG. 19 are publicly known, and therefore further description thereof will be omitted here.

[0087] The control unit 6 generates switching signals to drive the two switching elements 220b and 220d using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 19 can also achieve the same effects as those of embodiment 1.

[0088] 19, the switching elements 220b and 220d are shown as IGBTs, but any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 19 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 217a and 217b and the current detector 211 do not need to be used.

[0089] Embodiment 5 In embodiment 5, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0090] 20 is a diagram showing a configuration example of an AC-DC converter 2 according to embodiment 5. In the AC-DC converter 2 shown in FIG. 20, the rectifier circuit 20 is configured with a single-phase H-bridge cell including two diodes 218a and 218b, four switching elements 220a, 220b, 220c, and 220d, a capacitor 216b, and a voltage detector 217c. The voltage detector 217c may be provided outside the rectifier circuit 20.

[0091] In the rectifier circuit 20 shown in FIG. 20 , one leg is configured as a series circuit of diodes 218a and 218b, and the other leg is configured as a series circuit of switching elements 220a, 220b, 220c, and 220d. Capacitor 216b is connected between the connection point of switching elements 220a and 220b and the connection point of switching elements 220c and 220d. Voltage detection unit 217c detects the voltage of capacitor 216b and outputs the detected value to control unit 6. Control unit 6 generates switching signals for controlling switching elements 220a, 220b, 220c, and 220d based on the detected values ​​of voltage detection units 217a, 217b, and 217c and current detection unit 211. Note that the configuration and operation of rectifier circuit 20 shown in FIG. 20 are known, and further description thereof will be omitted here.

[0092] The control unit 6 generates switching signals to drive the four switching elements 220a, 220b, 220c, and 220d using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 20 can also achieve the same effects as those of embodiment 1.

[0093] 20, the switching elements 220a, 220b, 220c, and 220d are shown as IGBTs, but any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 20 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 217a, 217b, and 217c and the current detector 211 do not need to be used.

[0094] Sixth Embodiment In a sixth embodiment, a different example of the AC-DC converter 2 including the control unit 6 described in the first embodiment will be described. Components having the same or equivalent functions as those of the AC-DC converter 2 described in the first embodiment will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0095] 21 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a sixth embodiment. In the AC-DC converter 2 of FIG. 21 , a rectifier circuit 20 includes a single-phase H-bridge cell 221 and a switching cell 222. The single-phase H-bridge cell 221 includes two diodes 218a and 218c and two switching elements 220b and 220d. The switching cell 222 includes four switching elements 220e, 220f, 220g, and 220h, a capacitor 216c, and a voltage detection unit 217c. The voltage detection unit 217c may be provided outside the switching cell 222.

[0096] In a single-phase H-bridge cell 221 shown in Fig. 21, diodes 218a and 218c are arranged in the upper arms of two legs, and switching elements 220b and 220d are arranged in the lower arms of two legs. In a switching cell 222 shown in Fig. 21, four switching elements 220e, 220f, 220g, and 220h are bridge-connected. A capacitor 216c is connected in parallel to a first leg consisting of switching elements 220e and 220f and a second leg consisting of switching elements 220g and 220h.

[0097] The voltage detection unit 217c detects the voltage of the capacitor 216c and outputs the detected value to the control unit 6. The control unit 6 generates switching signals for controlling the switching elements 220b, 220d, 220e, 220f, 220g, and 220h based on the detected values ​​of the voltage detection units 217a, 217b, and 217c and the current detection unit 211. The configuration and operation of the rectifier circuit 20 shown in FIG. 21 are known, and further description thereof will be omitted here.

[0098] The control unit 6 generates switching signals to drive the six switching elements 220b, 220d, 220e, 220f, 220g, and 220h using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in FIG. 21 can also achieve the same effects as those of embodiment 1.

[0099] 21, the switching elements 220b, 220d, 220e, 220f, 220g, and 220h are shown as IGBTs, but any elements capable of switching may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 21 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detected values ​​of the voltage detectors 217a, 217b, and 217c and the current detector 211 do not need to be used.

[0100] Embodiment 7 In embodiment 7, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0101] FIG. 22 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a seventh embodiment. In the AC-DC converter 2 of FIG. 22, the rectifier circuit 20 is composed of a single-phase diode bridge cell 213a and a switching cell 225. The switching cell 225 includes a single-phase diode bridge cell 213b and a series circuit formed by two switching elements 220a and 220b. The series circuit is connected in parallel to the single-phase diode bridge cell 213b. In FIG. 22, the capacitor 216 in FIG. 2 is replaced with two capacitors 216a and 216b connected in series. The series-connected capacitors 216a and 216b are connected between the DC buses 9a and 9b.

[0102] The control unit 6 generates switching signals for controlling the switching elements 220a and 220b based on the detection values ​​of the voltage detection units 217a and 217b and the current detection unit 211. The configuration and operation of the rectifier circuit 20 shown in Fig. 22 are publicly known, and further description thereof will be omitted here.

[0103] The control unit 6 generates switching signals to drive the two switching elements 220a and 220b using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 22 can also achieve the same effects as those of embodiment 1.

[0104] 22, the switching elements 220a and 220b are shown as IGBTs, but any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 22 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 217a and 217b and the current detector 211 do not need to be used.

[0105] Embodiment 8 In embodiment 8, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0106] Fig. 23 is a diagram showing a configuration example of an AC-DC converter 2 according to an eighth embodiment. While the AC-DC converter 2 in Fig. 17 shows a single-phase AC power supply 1, Fig. 23 shows this changed to a three-phase AC power supply 5. As a result, in the AC-DC converter 2 in Fig. 23, the rectifier circuit 20 is configured with a three-phase full-bridge cell 226 including six switching elements 220a, 220b, 220c, 220d, 220e, and 220f. Reactors 212a, 212b, and 212c are inserted in each phase between the three-phase AC power supply 5 and the rectifier circuit 20, and current detectors 211a and 211b are arranged in any two of the three phases.

[0107] The voltage detection unit 227 detects the voltage of each phase of the three-phase AC power supply 5 and outputs the detected value to the control unit 6. The current detection units 211a and 211b detect the currents flowing in any two of the three phases and output the detected values ​​to the control unit 6. The current in the remaining phase can be determined by calculation within the control unit 6, taking advantage of the fact that the currents in the phases are three-phase balanced.

[0108] The control unit 6 generates switching signals for controlling the switching elements 220a, 220b, 220c, 220d, 220e, and 220f based on the detection values ​​of the voltage detection units 217b and 227 and the current detection units 211a and 211b. Note that the configuration and operation of the rectifier circuit 20 shown in Fig. 23 are publicly known, and further description thereof will be omitted here.

[0109] The control unit 6 generates switching signals to drive the six switching elements 220a, 220b, 220c, 220d, 220e, and 220f using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in FIG. 23 can also achieve the same effects as those of embodiment 1.

[0110] Although the switching elements 220a, 220b, 220c, 220d, 220e, and 220f are shown as IGBTs in FIG. 23 , any elements capable of switching operation may be used. Although the AC-DC converter 2 shown in FIG. 23 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217b and the current detectors 211a and 211b do not need to be used. Furthermore, the control according to the eighth embodiment may be performed on αβ coordinates or on three-phase coordinates.

[0111] Embodiment 9 In embodiment 9, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiments 1 and 8 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0112] FIG. 24 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a ninth embodiment. In the AC-DC converter 2 illustrated in FIG. 24 , the rectifier circuit 20 includes a three-phase diode bridge cell 228 and a three-phase simplified PAM cell 229. The three-phase diode bridge cell 228 includes six diodes connected in a full bridge configuration. The three-phase simplified PAM cell 229 includes single-phase diode bridge cells 213a, 213b, and 213c, and switching elements 215a, 215b, and 215c connected in parallel to the single-phase diode bridge cells 213a, 213b, and 213c, respectively. The single-phase diode bridge cells 213a, 213b, and 213c are connected between the three-phase lines and the reactors 212a, 212b, and 212c and the three-phase diode bridge cell 228. The three-phase simple PAM cell 229 also includes a capacitor 216d, one end of which is connected to the single-phase diode bridge cells 213a, 213b, and 213c, and the other end of which is connected to the DC bus 9b. The capacitor 216d may be provided outside the three-phase simple PAM cell 229.

[0113] The control unit 6 generates switching signals for controlling the switching elements 215a, 215b, and 215c based on the detection values ​​of the voltage detection units 217b and 227 and the current detection units 211a and 211b. Note that the configuration and operation of the rectifier circuit 20 shown in Fig. 24 are publicly known, and further description thereof will be omitted here.

[0114] The control unit 6 generates switching signals to drive the three switching elements 215a, 215b, and 215c using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 24 can also achieve the same effects as those of embodiment 1.

[0115] Although the switching elements 215a, 215b, and 215c are shown as IGBTs in FIG. 24 , any elements capable of switching operation may be used. Although the AC-DC converter 2 shown in FIG. 24 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217b and the current detectors 211a and 211b do not need to be used. Furthermore, the control according to the ninth embodiment may be performed on αβ coordinates or on three-phase coordinates.

[0116] Embodiment 10 In embodiment 10, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Note that components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiments 1, 8, and 9 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0117] Fig. 25 is a diagram showing a configuration example of an AC-DC converter 2 according to embodiment 10. In the AC-DC converter 2 of Fig. 25, the capacitor 216d is removed from the configuration of the AC-DC converter 2 of Fig. 24. Other configurations are the same as or equivalent to those of Fig. 24.

[0118] The control unit 6 generates switching signals for controlling the switching elements 215a, 215b, and 215c based on the detection values ​​of the voltage detection units 227 and 217b and the current detection units 211a and 211b. Note that the configuration and operation of the rectifier circuit 20 shown in Fig. 25 are publicly known, and further description thereof will be omitted here.

[0119] The control unit 6 generates switching signals to drive the three switching elements 215a, 215b, and 215c using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 25 can also achieve the same effects as those of embodiment 1.

[0120] Although the switching elements 215a, 215b, and 215c are shown as IGBTs in FIG. 25 , any elements capable of switching operation may be used. Although the AC-DC converter 2 shown in FIG. 25 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217b and the current detectors 211a and 211b do not need to be used. Furthermore, the control according to the tenth embodiment may be performed on αβ coordinates or on three-phase coordinates.

[0121] Embodiment 11 In embodiment 11, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Note that components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiments 1 and 8 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0122] 26 is a diagram showing a configuration example of an AC-DC converter 2 according to embodiment 11. In the AC-DC converter 2 shown in FIG. 26, the rectifier circuit 20 has a configuration called a full PAM circuit. The rectifier circuit 20 includes a three-phase diode bridge cell 228, a switching element 215, and a diode 218.

[0123] The control unit 6 generates a switching signal for controlling the switching element 215 based on the detection values ​​of the voltage detection units 227, 217b and the current detection unit 211. The configuration and operation of the rectifier circuit 20 shown in Fig. 26 are well known, and further description thereof will be omitted here.

[0124] The control unit 6 generates a switching signal for driving the switching element 215 using the control method described in the first embodiment. As a result, the AC / DC converter 2 shown in Fig. 26 can also achieve the same effects as those of the first embodiment.

[0125] Although the switching element 215 is shown as an IGBT in FIG. 26 , any element capable of switching operation may be used. Although the AC-DC converter 2 shown in FIG. 26 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217 b and the current detector 211 do not need to be used. Furthermore, the control according to the eleventh embodiment may be performed on an αβ coordinate system or a three-phase coordinate system.

[0126] Embodiment 12 In embodiment 12, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Note that components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiments 1, 8, and 11 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0127] Fig. 27 is a diagram showing a configuration example of an AC-DC converter 2 according to embodiment 12. In the AC-DC converter 2 of Fig. 27, the reactors 212a, 212b, and 212c that were arranged between the three-phase AC power supply 5 and the three-phase diode bridge cell 228 in the configuration of the AC-DC converter 2 of Fig. 26 are replaced with a reactor 212. In Fig. 27, the reactor 212 is arranged between the three-phase diode bridge cell 228 and the diode 218. The rest is the same as or equivalent to Fig. 26.

[0128] The control unit 6 generates a switching signal for controlling the switching element 215 based on the detection values ​​of the voltage detection units 227, 217b and the current detection unit 211. The configuration and operation of the rectifier circuit 20 shown in Fig. 27 are well known, and further description thereof will be omitted here.

[0129] The control unit 6 generates a switching signal for driving the switching element 215 using the control method described in the first embodiment. As a result, the AC / DC converter 2 shown in Fig. 27 can also achieve the same effects as those of the first embodiment.

[0130] Although the switching element 215 is shown as an IGBT in FIG. 27 , any element capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 27 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217 b and the current detector 211 do not need to be used. Furthermore, the control according to the twelfth embodiment may be performed on an αβ coordinate system or a three-phase coordinate system.

[0131] Embodiment 13 In embodiment 13, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Note that components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiments 1 and 8 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0132] FIG. 28 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a thirteenth embodiment. In the AC-DC converter 2 of FIG. 28, the rectifier circuit 20 is configured with a three-phase diode bridge cell 228 and a three-phase bidirectional switching cell 231. The three-phase bidirectional switching cell 231 includes six switching elements 231a, 231b, 231c, 231d, 231e, and 231f. Also, in FIG. 28, the capacitor 216 of FIG. 23 is replaced with two capacitors 216a and 216b connected in series. The series-connected capacitors 216a and 216b are connected between the DC buses 9a and 9b. In the three-phase bidirectional switching cell 231, the switching elements 231a and 231b, the switching elements 231c and 231d, and the switching elements 231e and 231f are connected in series in pairs. Each series-connected pair is arranged for each phase between the three-phase diode bridge cell 228 and the connection point of the capacitors 216 a, 216 b. Note that the configuration and operation of the rectifier circuit 20 shown in FIG. 28 are well known, and further description thereof will be omitted here.

[0133] The control unit 6 generates switching signals to drive the six switching elements 231a, 231b, 231c, 231d, 231e, and 231f using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 28 can also achieve the same effects as those of embodiment 1.

[0134] Although the switching elements 231a, 231b, 231c, 231d, 231e, and 231f are shown as IGBTs in FIG. 28 , any elements capable of switching operation may be used. Although the AC-DC converter 2 shown in FIG. 28 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 227 and 217b and the current detectors 211a and 211b do not need to be used. Furthermore, the control according to the thirteenth embodiment may be performed on αβ coordinates or on three-phase coordinates.

[0135] Embodiment 14 In embodiment 14, a different example of the AC-DC converter 2 including the control unit 6 described in embodiment 1 will be described. Components having the same or equivalent functions as components of the AC-DC converter 2 described in embodiment 1 will be denoted by the same reference numerals, and overlapping descriptions will be omitted.

[0136] FIG. 29 is a diagram illustrating a configuration example of an AC-DC converter 2 according to a fourteenth embodiment. In the AC-DC converter 2 of FIG. 29, a rectifier circuit 20 is configured with a single-phase diode bridge cell 213a and an interleaved cell 219. The interleaved cell 219 is configured by combining two sets of a reactor 212, a switching element 215, and a diode 218 in the full-PAM circuit configuration described in FIG. 12. Specifically, the interleaved cell 219 includes reactors 2191a and 2191b, diodes 2192a and 2192b, and switching elements 2193a and 2193b. The configuration and operation of the rectifier circuit 20 illustrated in FIG. 29 are publicly known, and further description thereof will be omitted here.

[0137] The control unit 6 generates switching signals to drive the two switching elements 2193a and 2193b using the control method described in embodiment 1. As a result, the AC-DC converter 2 shown in Fig. 29 can also achieve the same effects as those of embodiment 1.

[0138] Although the switching elements 2193a and 2193b are shown as IGBTs in FIG. 29 , any elements capable of switching operation may be used. Furthermore, although the AC-DC converter 2 shown in FIG. 29 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, the detection values ​​of the voltage detectors 217a and 217b and the current detector 211 do not need to be used. Furthermore, although FIG. 29 shows an example in which the interleaved cells 219 are configured in two stages, the interleaved cells 219 may also be configured in three or more stages. Furthermore, the rectifier circuits 20 shown in the first to fourteenth embodiments may also be configured in an interleaved configuration.

[0139] Fifteenth Embodiment Fig. 30 is a diagram showing a configuration example of a refrigeration cycle-applied device 900 according to a fifteenth embodiment. The refrigeration cycle-applied device 900 according to the fifteenth embodiment includes the rotating machine drive device 8 described in the first embodiment. The refrigeration cycle-applied device 900 according to the fifteenth embodiment can be applied to products equipped with a refrigeration cycle, such as air conditioners, refrigerators, freezers, and heat pump water heaters.

[0140] The refrigeration cycle device 900 includes a compressor 42 incorporating the motor 41 of the first embodiment, a four-way valve 902, an indoor heat exchanger 906, an expansion valve 908, and an outdoor heat exchanger 910, all of which are attached via refrigerant piping 912. Inside the compressor 42, there are provided a compression mechanism 904 that compresses the refrigerant, and a motor 41 that operates the compression mechanism 904. The refrigeration cycle device 900 can perform heating or cooling operation by switching the four-way valve 902.

[0141] The compression mechanism 904 is driven by a variable-speed controlled motor 41. During heating operation, as indicated by the solid arrows, the refrigerant is pressurized by the compression mechanism 904 and discharged, passes through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910, and the four-way valve 902, and returns to the compression mechanism 904. During cooling operation, as indicated by the dashed arrows, the refrigerant is pressurized by the compression mechanism 904 and discharged, passes through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906, and the four-way valve 902, and returns to the compression mechanism 904. During heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 reduces the pressure of the refrigerant and causes it to expand.

[0142] The refrigeration cycle applied device 900 according to the fifteenth embodiment has been described as including the rotating machine driving device 8 described in the first embodiment, but is not limited to this. The refrigeration cycle applied device 900 may include the rotating machine driving device 8 including the rectifier circuit 20 described in any of the second to fourteenth embodiments. Furthermore, the refrigeration cycle applied device 900 may include a rectifier circuit other than the rectifier circuit 20 described in any of the first to fourteenth embodiments, as long as the control method of the first embodiment can be applied.

[0143] The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies, or the embodiments may be combined with each other, or part of the configuration may be omitted or modified without departing from the spirit of the invention. For example, the above-described control method may also be applied to a DC-AC converter.

[0144] 1 AC power supply, 2 AC / DC converter, 3 DC / AC converter, 4 Load, 5 Three-phase AC power supply, 6 Control unit, 7 Current control system, 8 Rotating machine drive unit, 9a, 9b DC bus, 20 Rectifier circuit, 41 Motor, 42 Compressor, 61 Voltage control unit, 62 Current control unit, 63 Switching signal generation unit, 72 Controller, 73 Control target plant, 74, 624 Target value filter, 211, 211a, 211b Current detection unit, 212, 212a, 212b, 212c, 2191a, 2191b Reactor, 213a, 213b, 213c Single-phase diode bridge cells, 215, 215a, 215b, 215c, 220a, 220b, 220c, 220d, 220e, 220f, 220g, 220h, 231a, 231b, 231c, 231d, 231e, 231f, 2193a, 2193b Switching elements, 216, 216a, 216b, 216c, 216d Capacitors, 217a, 217b, 217c, 227 Voltage detection units, 218, 218a, 218b, 218c, 2192a, 2192b Diodes, 219 Interleaved cells, 221 Single-phase H-bridge cells, 222, 225 Switching cells, 226 Three-phase full-bridge cells, 228 Three-phase diode bridge cells, 229 Three-phase simple PAM cell, 231 Three-phase bidirectional switching cell, 611 Voltage controller, 612, 622 Subtractor, 621 Current controller, 621A P controller, 621B S controller, 621C Adder, 623 Multiplier, 630 Standardizer, 631 Carrier signal generator, 632 Comparator, 900 Refrigeration cycle application equipment, 902 Four-way valve, 904 Compression mechanism, 906 Indoor heat exchanger, 908 Expansion valve, 910 Outdoor heat exchanger, 912 Refrigerant piping.

Claims

1. An AC-DC conversion device comprising: a rectifier circuit having at least one switching element for rectifying a power supply voltage applied from an AC power supply; a capacitor connected to a DC bus for smoothing the output voltage of the rectifier circuit; a reactor disposed on the AC power supply side of the capacitor; and a control unit for generating a switching signal for controlling the switching element using a carrier signal, wherein the switching element is disposed on the AC power supply side of the capacitor, and a carrier frequency, which is the frequency of the carrier signal, is changed within a half cycle of the power supply voltage so as not to fall below a lower limit value and so that a harmonic component included in a power supply current flowing between the AC power supply and the rectifier circuit complies with a harmonic standard value of the power supply current.

2. The AC-DC conversion device according to claim 1, wherein the control unit includes: a voltage control unit for generating a first current command value using a first voltage command value, which is a command value of a bus voltage that is the voltage of the DC bus; a current control unit for generating a second voltage command value using a second current command value obtained by making the first current command value follow a sine wave; and a switching signal generation unit for generating the switching signal using the second voltage command value and the carrier signal.

3. The AC-DC conversion device according to claim 2, wherein the lower limit value is set to 200 [Hz] or more.

4. The AC-DC conversion device according to claim 3, wherein when the current control unit includes a proportional controller and a sine wave transfer function controller, the lower limit value is set to 150 [Hz] or more.

5. A rotary machine drive device comprising the AC-DC conversion device according to any one of claims 1 to 4.

6. A refrigeration cycle application device comprising the AC-DC conversion device according to any one of claims 1 to 4.

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