Oscillator based sensors

Abstract

This application relates to methods and apparatus for sensing using controlled oscillators. A sensor apparatus has a first oscillator (200) which generates a first oscillator signal (Sosc). The first oscillator includes a sensor component (205) with an electrical property that varies with a measurand of interest to vary an oscillation frequency of the first oscillator signal. In some examples, the first oscillator may be a ring oscillator with a plurality of delay elements (201) and at least one delay element includes the sensor component which varies with a measurand of interest to vary a propagation delay of the sensing delay element.

Description

[0001] OSCILLATOR BASED SENSORS

[0002] The field of representative embodiments of this disclosure relates to methods, apparatus and / or implementations concerning or relating to sensing, e.g. electrical, physical or chemical sensing, and, in particular, to sensing using oscillators.

[0003] Sensing is required in a variety of different applications. In general, an electrical sensing circuit may comprise circuitry that generates an analog current or voltage related to a measurand of interest and an analog-to-digital converter (ADC) for converting the analog current or voltage in a digital output.

[0004] One example, where sensing is typically required, is for driving output transducers, e.g. driving loudspeakers in audio applications. In such applications it may be desirable to monitor one or more of temperature, current and voltage of the driver circuit for implementing appropriate control and / or protection of the driver circuit.

[0005] Driver circuits for driving output transducers have been implemented as an integrated circuit formed in a silicon substrate. For some applications, however, the use of compound semiconductor materials is increasingly being proposed for at least the output power stage of a circuit, such as a DC / DC converter or the output stage of a switching driver or class-D type amplifier may be implemented as an integrated circuit formed in Gallium Nitride (GaN) substrate. The properties of GaN can make it advantageous for use in some applications, for instance for high power and / or high voltage applications. However, the properties of GaN can pose challenges for integrating other circuitry, such as sensing circuitry, into the GaN device. For instance, as will be understood by one skilled in the art, whilst n-channel devices can be readily formed in a GaN substrate, the commercially available fabrication techniques are generally not able to form effective p-channel devices and thus complementary integrated circuits designs are generally not available for use in GaN circuits.

[0006] Embodiments of the present disclosure relate to methods and apparatus for sensing that at least mitigate at least some of the above-mentioned issues. In particular, at least some embodiments may be suitable for use in integrated circuits formed in compoundsemiconductors materials such as GaN, but it will be understood that embodiments may also be implemented in integrated circuits formed in silicon.

[0007] According to an aspect of the disclosure there is provided a sensor apparatus comprising: a first oscillator comprising at least one capacitance which is charged and discharged to generate a first oscillator signal and a sensor component with an electrical property that varies with a measurand of interest to vary at least one of a rate of charging and discharging of said at least one capacitance and hence an oscillation frequency of the first oscillator signal.

[0008] In some implementations, the first oscillator may comprise a first ring-oscillator comprising a plurality of delay elements and at least one of the delay elements may comprise a sensing delay element which comprises said sensor component with an electrical property that varies with a measurand of interest to vary a propagation delay of the sensing delay element

[0009] In some implementations, the sensing delay element may comprise the sensor component connected in series with a switching device. A control terminal of the switching device may be configured to be driven by an input to the sensing delay element. An output of the sensing delay element may be taken from a node between the series connected sensor component and switching device. The switching device may comprise an n-channel transistor or an enhancement high electron mobility transistor.

[0010] In some implementations, each of the plurality of delay elements of the first ring oscillator may comprise a sensing delay element.

[0011] In some implementations, the first ring oscillator may be configured so that a number of active delay elements in the ring can be selectively controlled.

[0012] In some implementations, the first oscillator may be one of a first resistive-capacitive resonance oscillator and a first resistive-capacitive relaxation oscillator.In some implementations, the sensor component may comprise a resistance having a resistance value which varies with the measurand of interest.

[0013] In some examples the measurand of interest may be temperature and the resistance has a first temperature coefficient of resistance. In some examples the measurand of interest may be strain and the variable resistance may comprise a resistive strain sensor. In some examples the resistance may comprises a two-dimensional electron gas resistive element. In some example, the sensor apparatus may be configured for sensing for one or more target chemical analytes of interest, and the sensor apparatus may comprise a functional layer overlying the two-dimensional electron gas resistive element, where the functional layer is configured to interact with the one or more target chemical analytes of interest to vary a charge or electric field distribution over the two-dimensional electron gas and hence the resistance of the two-dimensional electron gas resistive element.

[0014] In some implementations, the sensor component may comprise a variable current component that provides a current which varies with the measurand of interest.

[0015] In some examples the measurand of interest may be a magnetic field strength and the variable current component may comprise a Hall sensor configured to provide current steering between first and second output electrodes. In some implementations, the sensor apparatus may further comprise a second ring oscillator with at least one sensing delay element and the Hall sensor may be configured as a shared sensing component for a sensing delay element of each of the first and second ring oscillators. The Hall sensor may be connected as part of the sensing delay element of the first ring oscillator via the first output electrode and be connected as part of the sensing delay element of the second ring oscillator via the second output electrode. In some implementations, the sensor apparatus may comprise a plurality of Hall sensors configured in a stacked arrangement such that an output current from one of the first and second output electrodes of a first one of the Hall sensors provides an input current for a second one of the Hall sensors.

[0016] The sensor apparatus may further comprise a second oscillator configured to generate a second oscillator signal. The first and second oscillators may be configured such that any variation in frequency of the second oscillator signal with the measurand of interestis different to the variation in frequency of the first oscillator signal. In some implementations, the second oscillator may be configured such that the frequency of the second oscillator signal does not vary with the measurand of interest. The sensor apparatus may further comprise a counter configured to count a number of oscillations of one of the first and second oscillator signals in a count period. In some examples, a frequency divider configured to apply frequency division to the other of the first and second oscillator signals to define the count period. In some examples the count period may be defined by a reference clock. In some examples, a processing module may be configured to implement an exclusive-or function on the first and second oscillator signals.

[0017] The sensor apparatus may be implemented as a gallium nitride based integrated circuit.

[0018] In another aspect of the disclosure there is provided a sensor apparatus comprising: a first ring oscillator comprising a plurality of delay elements configured to generate a first oscillator signal. At least one of the delay elements comprises a sensing delay element which comprises a sensor component with an electrical property that varies with a measurand of interest to vary a propagation delay of the sensing delay element and hence an oscillation frequency of the first oscillator signal.

[0019] In another aspect, there is provided a sensor apparatus comprising: a first ring oscillator comprising a plurality of delay elements configured to generate a first oscillator signal; wherein at least one delay element comprises a sensor component connected between a first voltage and a delay element output node and a switching device connected between the delay element output node and a second voltage; and wherein the sensor component provides at least one of a variable resistance and a variable current that varies with a measurand of interest.

[0020] In another aspect, there is provided a sensor apparatus comprising: a first ring oscillator comprising a plurality of delay elements configured to generate a first oscillator signal, wherein at least one of the delay elements is configured to provide a propagation delay which varies with a measurand of interest so as to vary a frequency of the first oscillator signal; and decode circuitry configured to receive and decode the first oscillator signal so as to provide a measurement signal indicative of the measurand of interest.The sensor apparatus may comprise a second ring oscillator comprising a plurality of delay elements configured to generate a second oscillator signal and wherein the second oscillator signal is used to define a count period for counting a number of oscillations of the first oscillator signal.

[0021] It should be noted that, unless expressly indicated to the contrary herein or otherwise clearly incompatible, then any feature described herein may be implemented in combination with any one or more other described features.For a better understanding of examples of the present disclosure, and to show more clearly how the examples may be carried into effect, reference will now be made, by way of example only, to the following drawings in which:

[0022] Figure 1 illustrates a conventional controlled ring oscillator;

[0023] Figures 2a and 2b illustrate examples of delay-based oscillators with sensing functionality according to an embodiment;

[0024] Figures 3a illustrates an example of a resonance-type oscillator with sensing functionality and figure 3b illustrates an example of a relaxation-type oscillator with sensing functionality;

[0025] Figures 4a - 4c illustrates different examples of sensor arrangements using two sensing oscillators;

[0026] Figure 5 illustrates one example of a delay element with a variable load capacitance;

[0027] Figure 6 illustrates an example of a sensing component for chemical sensing;

[0028] Figure 7 illustrates an example of a sensing delay element with a Hall effect sensor for magnetic sensing;

[0029] Figure 8 illustrates sharing of Hall effect sensors between two ring oscillators; and

[0030] Figure 9 illustrates stacking of Hall sensors to provide increased sensitivity.

[0031] The description below sets forth example embodiments according to this disclosure. Further example embodiments and implementations will be apparent to those having ordinary skill in the art. Further, those having ordinary skill in the art will recognize that various equivalent techniques may be applied in lieu of, or in conjunction with, the embodiments discussed below, and all such equivalents should be deemed as being encompassed by the present disclosure.Embodiments of the disclosure relate to oscillator-based sensors, that is to sensors implemented with an oscillator that has an oscillation period, and hence frequency, that varies with a measurand of interest.

[0032] Controlled oscillators, such as voltage-controlled oscillators (VCOs), are known and may be implemented in various ways. For instance, controlled oscillators may generally be implemented as a resonance oscillator, a relaxation oscillator or a ring oscillator.

[0033] Figure 1 illustrates one example of a basic form of ring oscillator 100 as may be implemented by CMOS device on a silicon substrate. The ring oscillator 100 in this example comprises an odd plurality of inverters 101, each of which comprises a PMOS transistor MP in series with an NMOS transistor MN between high-side and low-side voltages, e.g. a supply voltage VDD and ground. The output of each inverter 101, which is taken from the midpoint of the series connected transistors MP and MN, provides the gate drive input for the next inverter. The PMOS and NMOS transistors act in a complementary manner to drive the output of the inverter 101 high or low depending on the input to the inverter 101. An output is tapped from an appropriate part of the ring arrangement to provide the output signal Sosc from the ring oscillator 100.

[0034] The output frequency of the ring oscillator 100, i.e. the frequency of the output signal Sosc, depends on the number of inverters 101 in the ring arrangement and the individual propagation delay associated with each inverter. The propagation delay of each inverter 101, i.e. the delay between a change in the input and a corresponding change in the output of the inverter 101, depends on the properties of the transistors and the drive strength of the inverters, i.e. how quickly the output of one inverter can charge or discharge the gate capacitances of the PMOS and NMOS device of the next inverter to drive the output voltage high or low as appropriate.

[0035] To provide a controllably variable output frequency, the drive strength of the inverters 101 may be controllably varied in use so as to vary the propagation delay of the inverters. Figure 1 illustrates an example where a received control signal Scon, for instance a control voltage for a VCO implementation, is used to control a current source arrangement 102 that defines the strength of the drive signal which is output from eachinverter, but it will be understood that other arrangements are possible. In general, though, the received control signal Scon varies the propagation delay of the inverters 101, and consequently the output frequency of the VCO 100.

[0036] Ring oscillators such as described with reference to figure 1 may be used for a range of different applications, such as generation of clock signals. VCOs have also been used as part of an analogue-to-digital converter (ADC) arrangement, and some VCO based ADCs can have the advantage of being small in circuit area, although non-linearity can be a significant issue.

[0037] Embodiments of the present disclosure provide a sensing functionality within an oscillator, such that the oscillation period and hence output frequency of the oscillator varies with the measurand of interest.

[0038] Figure 2a and 2b illustrates two examples of oscillators 200 with sensing functionality according to an embodiment. In each case, the oscillator 200 comprises an odd number of inverting delay elements arranged in a ring arrangement in a similar manner as the oscillator of figure 1. However, in the examples of figures 2a and 2b at least one of the delay elements includes sensing functionality.

[0039] Figure 2a illustrates an embodiment where one sensing delay element 201 , implemented with sensing functionality, is arranged as part of the ring oscillator with a plurality of nonsensing delay elements 202.

[0040] The non-sensing delay elements 202, in this example, are coupled between high-side and low-side voltages, e.g. a supply voltage VDD and ground respectively. Each of the non-sensing delay elements 202 comprises a pull-up arrangement and a pull-down arrangement for driving the output high or low depending on the state of the input. In the example of figure 2a, the pull-down arrangement comprises a transistor 203 implemented as a switching device connected between the output for the non-sensing delay element 201 and the low-side voltage, e.g. ground, with its control terminal, e.g. gate, driven by the input for the non-sensing delay element 201. The transistor 203 may be an n-channel transistor or a transistor such as an enhancement mode high electron mobility transistor (HEMT). In the example of figure 2a, the pull-up arrangement of thenon-sensing delay element 201 comprises a resistance 204. In use, when the transistor 203 is on, i.e. conducting, the output of the non-sensing delay element 201 is pulled down to the low side voltage, whereas when the transistor 203 is off, i.e. non-conducting, the output of the non-sensing delay element 201 is pulled high. The pull-up provided by the resistance 204 is thus an inherent or passive pull-up that occurs when the transistor 203 is non conducting.

[0041] The use of a resistance 204 as the pull-up element for the non-sensing delay elements 201 means that the non-sensing delay elements 201 can be implemented without the need for any p-channel transistors, which means that such delay elements can be readily implemented in compound semiconductor materials such as Gallium Nitride (GaN). However, embodiments may be implemented in any suitable material system, and in applications implemented on a silicon substrate, the non-sensing delay elements 201 could be implemented with a resistance 204 (as illustrated in figure 2a) or could be implemented using conventional inverters 101 such as described with reference to figure 1. Note, in the example of figure 2a, the resistance 204 of a non-sensing delay element 202 could be implemented by a passive resistor of a suitable material, or a combination of such resistors, or could be implemented by one or more active devices, e.g. by a depletion mode n-channel transistor having a connected gate and source.

[0042] The propagation delay of each non-sensing delay element 202 depends on the drive strength of the non-sensing delay element 202. The drive strength of the output from each non-sensing delay element 202 depends on the supply voltages, e.g. on VDD, and the properties of the n-channel transistor 203 and resistance 204.

[0043] The propagation delay for the non-sensing delay element 202 can be represented in terms of the rise time for of the output of the non-sensing delay element 202 to go high when its input is low or the fall time for the output of the non-sensing delay element 202 to go low when its input is high as appropriate.

[0044] The fall time, tf, can be represented as:

[0045]

[0046] where VDD is the value of supply voltage, Vt is the threshold voltage for the transistor 203, CL is the capacitance of the load at the output of the delay element, e.g. the gatecapacitance of the transistor 203 of the next delay element in the ring, and It is the drainsource current of the transistor 203 when on (assuming saturation) which is given by:

[0047] It = k(VDD - Vt)2Eqn. (2)

[0048] The rise time can be modelled by considering the time taken for the output voltage Vo to go from 10% of VDD to 90% of VDD. The output voltage Vo is given by:

[0049] Vo = VDD(1 - et / R CV) Eqn. (3) where R is value of the resistance 204. The rise time from Vo = 0.1 VDD to Vo = 0.9VDD can thus be determined as:

[0050] tr = RCL\n (9) Eqn. (4)

[0051] It can thus be seen that the rise time of the non-sensing delay element 202 has a dependence on the value of the resistance 204.

[0052] The sensing delay element 201 may comprise essentially the same pull-down arrangement as a non-sensing delay element 202, i.e. a transistor 203 (which may be an enhancement HEMT or n-channel transistor) connected between the output of the sensing delay element 201 and the low-side voltage, e.g. ground, with its gate controlled by the input to the sensing delay element 201. However, the pull-up arrangement for the sensing delay element 201 comprises a sensor component 205 which provides a propagation delay for the sensing delay element 201 which varies with a measurand of interest. The sensor component 205 is configured to have an electrical property that varies with the measurand of interest to provide the variable propagation delay, e.g. a variable rise time. For instance, in some implementations the sensor component 205 may comprises a variable resistance, with a resistance that varies with the measurand of interest.

[0053] For such a sensing delay element 201, assuming the transistor 203 has the same characteristics as that of a non-sensing delay element 202, the fall time tf for the output of the sensing delay element 201 will also be given by equations 1 and 2 above. The rise time for the sensing delay element 201 may also be determined by equation 4 above, but the value of the resistance of the sensor component 205 may vary with the measurand of interest, e.g. as R+ AR where AR is the variation with the measurand of interest. The rise time for the sensing delay element 201 will thus vary with themeasurand of interest, with a consequent variation in the overall cycle period of the controlled oscillator 201 and hence the output frequency. A change AR in the resistance of the sensor component 205 will result in a change Atr in the rise time for that sensing delay element which is equal to ARCdn(9) and hence a consequential change in the cycle period for the oscillator 200. It will be noted that the change in oscillation period may thus be generally linear with a change in resistance for the sensing delay element 201. This relationship between a change in resistance of the sensor component 205 and a change in output period for embodiments of the present disclosure can provide a greater linearity than between a change in voltage in a conventional VCO such as discussed with reference to figure 1.

[0054] Whilst figure 2a illustrates that just one of the delay elements of the oscillator 200 is implemented as a sensing delay element 201, in some embodiments more than one of the delay elements of the oscillator 200 may be implemented as a sensing delay element.

[0055] Figure 2b illustrates an embodiment where each of the delay elements of the oscillator 200 is implemented as a sensing delay element 201. In this case the rise time of each of the sensing delay elements 201 may vary with the value of the resistance of the relevant sensor component 205. Assuming all the sensing delay elements 201 respond in substantially the same way to the relevant measurand, this can provide a greater change in oscillator period toscfor a given change in measurand and thus increase sensitivity. In this case, the change in oscillation period is effectively an indication of the average change in measurand indicated by all of the sensing delay elements 201.

[0056] As a complete cycle of oscillation of the oscillator 200 involves the output of each delay element undergoing both a high-to-low transition and a low-to-high transition, the overall oscillation period is thus a sum of the rise and fall times of each of the delay elements and thus a variation in the rise time of the sensing delay element 201 results in a variation in the period of oscillation.

[0057] The overall period of oscillation tosccan be given by:

[0058]

[0059] = 1 + kR(VDD — Vt)\n (9) Eqn. (5)where N is the number of delay elements in the ring and R, in this case, is the average resistance of the pull-up arrangements of the delay elements, i.e. the average resistance of the sensor component 205 of the sensing delay element(s) 201 and the resistances 204 of any non-sensing delay element.

[0060] The embodiments of figures 2a and 2b thus include a sensor component 205, such as a variable resistance, within a ring oscillator, so as to provide an oscillation period, or equivalently an oscillation frequency, which varies with the measurand of interest. A ring oscillator such as described with reference to figures 2a and 2b is a delay-based oscillator. This can be seen as a form of RC (resistive-capacitive) oscillator, in that the resistance of the pull-up element of each delay element, i.e. the sensor component 205 or resistance 204 together the gate capacitance of the next delay element defines, at least partly, the oscillation period as discussed above. A ring oscillator can be advantageous in some applications, as it can be implemented as a relatively small and simple circuit, but embodiments may also be implemented with other types of RC oscillators and, in general, embodiments relate to a sensing component being implemented as part of, i.e. within, an RC oscillator. Including the sensing component within the RC oscillator can improve linearity. In general, for an RC oscillator, the oscillation period may have a dependence on R*C and if the relevant resistance is implemented as a variable resistance, the oscillation period will thus vary linearly with the resistance of the sensing element.

[0061] The oscillator could, for example, be an RC resonance type oscillator which generates a sine-wave type output via a frequency selective RC network, for instance a lead-lag oscillator such as a Wien-bridge. Figure 3a illustrates an example of a Wien-bridge type RC resonance oscillator 300a. An op-amp 301 is configured with feedback to its noninverting input by an RC network which comprises a series combination of a resistor and capacitor with values Rs and Rc and a parallel combination of a resistor and capacitor with the same values Rs and Rc. This provides a phase shift that depends on frequency. Feedback to the non-inverting input of op-amp 301 via the resistive divider formed by resistors R1 and R2 defines the gain. The frequency of oscillation is equal to 1 / (2TTRSC) and thus the oscillation period is proportional to RsC. In the example of figure 3a the resistances Rs are implemented as variable resistance sensor components 205 to provide an oscillation period that varies with the measurand. In practice, some amplitudestabilization, such as automatic gain control or the like, may be applied to reduce distortion but such a lead-lag RC resonance oscillator can provide an oscillator with a clean and tuneable output frequency, which can readily be controlled to be within a useful frequency band for audio related applications.

[0062] It will be understood that this is just one type of suitable RC resonance oscillator, where the output may be a sine wave or similar via a frequency selective RC network and in which the relevant resistance(s) could be implemented as variable resistances that vary with the measurand of interest so as to provide a sensing functionality. Other suitable RC resonance oscillators could be, for example, be based on ladder phase-shift networks, say with three RC sections, which provides a simple and cheap oscillator, or a notch filter oscillator, such as a Twin-T or Bridged-T oscillator or the like, which uses RC networks configured in a T-topology to provide feedback at the notch frequency. Other RC resonance oscillators could include active-filter loops, e.g. a biquad loop oscillator based on two-integrator loops or the like.

[0063] In some embodiments, the resonance oscillator could instead be implemented as an LR (inductive-resistance) or LCR (inductive-capacitive-resistance) resonance oscillator, but the use of inductors can have some practical implications in terms of cost and area, and thus LC resonance oscillators will be preferred for some applications.

[0064] In some implementations, the oscillator could be a relaxation type oscillator, i.e. an oscillator that generates a non-sinusoidal output such as triangle or square wave, in general by charging and discharging a capacitor via a resistance (or with some other controlled current). Again, for an RC implementations, the resistance could be a variable resistance, with a resistance that varies with the measurand of interest. Figure 3b illustrates one example of a relaxation oscillator 300b, in which the output from comparator 302 charges capacitance C via resistance Rs and the capacitor voltage is fed-back to the comparator where it is compared with a reference voltage Vref. The comparator 302 can be configured to apply some hysteresis to the comparison, and thus the period of oscillation depends on the rate of charging / discharging of the capacitor C and the relevant amount of hysteresis, and the period thus has a dependence on RsC. The resistance Rs can be implemented as a sensing component 205 with a variableresistance based on the measurand of interest so as to provide an oscillator with a sensing functionality.

[0065] Other types of relaxation oscillator can be implemented, however, such as Schmitt trigger oscillator or ASDM (asynchronous sigma-delta modulator) or some timed discharge controller such as an astable multivibrator or a 555 astable timer or the like.

[0066] In general, a resonance type oscillator such as illustrated in figures 3a may provide a sine wave type output with lower distortion than the other oscillator types, but may be more complex to implement. Relaxation type oscillators, such as illustrated in figure 3b provides non-sinusoidal outputs such as square or triangle waves which can have a wider tuning range and be relatively easy to implement. Delay based oscillators such as illustrated in figures 2a and 2b may be the simplest to implement on an integrated circuit and can be very small in circuit area, but may suffer from the greatest amount of jitter.

[0067] Embodiments of the present disclosure may be implemented to provide sensing for a range of different possible measurands, provided that the sensor component 205 has a property that varies with the measurand of interest in a way that can lead to a change in oscillation period. As mentioned above, this may, in some examples, be achieved by implementing the sensor component 205 to provide a variable resistance.

[0068] In one example, the oscillator 200 with sensing functionality may be implemented as a temperature sensor, in which case each sensor component 205 may have a temperature coefficient of resistance such that the resistor exhibits a relatively significant variation in resistance with temperature.

[0069] Such a sensor component 205 could be implemented by any suitable arrangement that provides a resistance that varies with temperature. For example, one or more resistors of a suitable resistive material could be used to provide the variable resistance. In some examples, the resistance could be provided by an active load, e.g. a n-channel depletion transistor connected to provide a resistance, e.g. with its gate and source terminals connected. In some material systems, the variable resistance may be implemented by a two-dimensional electron gas (2DEG). As will be understood by one skilled in the art, 2DEGs can be formed by suitable heterojunctions, e.g. in GaN a 2DEG can be formedat a GaN / AIGaN heterojunction. The 2DEG has a charge carrier mobility, and hence conductance / resistance, that can vary with temperature. A 2DEG can thus be utilised as a resistance which varies with temperature and at least some 2DEGs can exhibit an effective temperature coefficient of resistance which is relatively high.

[0070] An oscillator such as illustrated in any of figures 2a, 2b, 3a or 3b could thus be implemented with temperature variable resistances for the sensor components 205 to provide an oscillation frequency that varies with temperature.

[0071] For a delay based oscillator 200, such as illustrated in figures 2a or 2b, the output signal from such an oscillator 200 would thus be indicative of temperature changes affecting the sensing delay elements 201 of the oscillator 200. In at least some applications, this oscillator output signal may be demodulated or decoded to provide a measurement signal.

[0072] It would be possible to demodulate the output signal from the oscillator 200 by determining the frequency of the oscillation signal, e.g. by counting the number of oscillations of the output signal in a count period defined by a stable reference clock and converting the frequency into an indication of temperature, however it will be noted from equation 5 that the oscillation period is proportional to resistance and thus the oscillation frequency is inversely proportional to the frequency of the oscillation signal. A decode in time, rather than frequency may, therefore, be advantageous. The oscillation period could be determined by counting the number of pulses of a fast reference clock in a count period defined by the oscillation signal, but this may require a fast reference clock. Also, both these approaches would require a stable reference clock and ideally require a well-regulated power supply for the controlled oscillator 200, as the oscillation period also has a dependence on the supply voltage. Similar considerations apply to the other types of oscillator with sensing functionality.

[0073] In some embodiments therefore, a sensor system may be implemented with two oscillators, where one oscillator is used to provide demodulation of the other controlled oscillator.Figure 4a illustrates one example of such a sensor system 400a comprising first and second oscillators, labelled as 401a and 401b. In the example of figure 4 each of the first and second oscillators 401a and 401b comprises an oscillator with sensing functionality such as described with reference to figures 2a or 2b or 3a or 3b but the oscillators 401a and 401b are implemented such that they respond differently to the measurand of interest. For instance, for temperature sensing, the temperature coefficient of resistance of the sensor component 205 of the first controlled oscillator 401a is different to that of the second controlled oscillator 401b. For some applications, only one of the first and second oscillators 401a and 401b may be implemented with sensing functionality and the other oscillator may be implemented by an oscillator without sensing functionality, such a delay oscillator formed purely from a ring arrangement comprising non-sensing delay elements 202 such as discussed with reference to figure 2a.

[0074] The output signal Sosca of the first oscillator 401a is supplied to a counter 402 which counts the number of oscillations of the output signal Sosca in a defined count period to provide a count value y. The output signal Soscb of the second oscillator 401b is supplied to a frequency divider 403 which divides the frequency by a factor M and supplies the frequency divided signal to the counter 402 to define the count period. Note, for correct decoding the

[0075] For an example where the controlled oscillator is a delay oscillator such as described with reference to figures 2a or 2b, the count value y can, from equation 5 above, be seen as:

[0076]

[0077] where Ra is the average resistance average resistance of the pull-up arrangements of the delay elements for the first oscillator 401a and Rb is the average resistance average resistance of the pull-up arrangements of the delay elements for the second oscillator 401b.

[0078] This can be approximated as:

[0079] Y ~ M— Eqn. (7)If Rais expressed as Ra = RaO (1 + aaT) and Rb = RbO (1 + cibT), where T is temperature RaO and RbO are the respective resistance values at a temperature of zero and aaand Ob are the respective temperature coefficients of resistance for the first and second oscillators 401a and 401b respectively, then:

[0080]

[0081] It can be seen that the count value y will thus vary in a generally linearly manner with temperature. Note that for a correct decode, it should be ensured that the count period is greater than twice the oscillation period of the output signal Sosca of the first oscillator 401a, i.e. the frequency of the output signal Sosca of the first oscillator 401a should be greater than M / 2 times the frequency of the output signal Soscb of the second oscillator 401b.

[0082] In this example, for temperature sensing, the sensitivity, in terms of variation in count value with temperature, depends on the difference between the temperature coefficients of resistance aaand Ob for the two controlled oscillators, and thus it is generally beneficial to implement the first and second oscillators 401a and 401b with a relatively large difference in the temperature coefficient of resistance. For instance, one of the first and second oscillators 401a and 401b could be implemented with a 2DEG (with a relatively high temperature coefficient of resistance) as the variable resistance of the sensor component 205, whilst the other of the first and second oscillators 401 a and 401 b could be implemented with resistors formed from a material such as silicon carbide, silicon chromium, or silicon carbide chrome (with a relatively low temperature coefficient of resistance).

[0083] The count value y could, in some applications, be used to provide a measurement signal SM representing the measurand of interest, e.g. in this example an indication of absolute temperature or an indication of temperature change. For instance, some downstream processing (not illustrated) may be applied to translate the count value y to the desired measurement signal, for instance using a suitable look-up table (LUT) of calibrated and / or modelled count values against temperature or applying some defined transfer function.The sensor system 400a thus provides a means of sensing a measurand of interest, such as temperature in the example discussed, using oscillators. It will be noted that using the output of the second oscillator 401 b to define the count period for counting the number of oscillations of the first oscillator 401a means that any variation in the respect count period due to variations in supply voltage VDD substantially cancel. This arrangement also avoids the need for any accurate reference clock.

[0084] The sensor system 400a may advantageously be implemented in an integrated circuit with only a relatively small footprint, i.e. which only requires a relatively small circuit area. The oscillators 401a and 401b can themselves be implemented as small components and the counter 402 and frequency divider 403 can also be small in area. The sensor system 400a may also advantageously be relatively low power. As the sensing functionality is provided by the sensor component 205 of the oscillators, the power for the sensor component 205 also powers at least part of the oscillator, i.e. there is no need to separately power both a sensor and an oscillator.

[0085] As noted above, in at least some implementations, the first and second oscillators 401a and 401b can be implemented without requiring any p-channel devices, meaning that they can be readily implemented in material systems such as GaN. In some applications, the frequency divider 403 and / or the counter 402 may formed as part of the same integrated circuit (IC) with the firstand second oscillators 401a and 401b and in materials systems such as GaN, the frequency divider 402 and / or the counter 402 may thus be implemented without using any p-channel devices. One skilled in the art will understand there are various ways a frequency divider and a counter could be implemented with requiring the use of a p-channel device. In some implementations, however, the first and second oscillators 401a and 401b could be implemented as part of a first IC 404 with at least one of the frequency divider 403 and counter 402 being external, i.e. off chip, to that first IC 404, e.g. formed as part of a different IC. This may allow the first and second oscillators to be implemented as part of the first IC 404 where the sensing is required, which may be formed in a first material system, such as GaN, but the decode circuitry, i.e. the frequency divider 403 and counter 402 could be implemented as part of a second, different, IC which could be implemented in a different material system, such as silicon. This arrangement has the advantage that the output signals Sosca and Soscb, which will be output from the first IC, are essentially digital time-encoded signals. Thetransmission of such digital signals between different ICs may be more robust and / or less likely to introduce noise compared to the transmission of analogue signals, which may be the case with some other sensing arrangements where a sensor is arranged as part of the first IC and ADC is arranged to provide a decode on a different IC.

[0086] This advantage of transmitting a digital time-encoded signal indicative of the measurand also apply in the case where one or more oscillator with sensing functionality were implemented and decoded using a stable reference clock.

[0087] In some examples, it may be preferable to use such a stable reference clock. For example, for microcontroller-based decoding of two sensor controlled oscillators, deriving both measurements from a single refence clock may provide better performance than using the output of one oscillator as the timing base for the other, especially if the refence clock has high stability. Using a common reference clock places both edge counts in the same clock domain, eliminating clock-domain crossings and metastability, avoiding a ±1 -cycle gating uncertainty that arises when one aperiodic / jittery oscillator signal opens and closes the other’s measurement window, and preventing correlated jitter / beat-note aliasing between the two unknowns.

[0088] Figure 4b illustrates one example of another sensor system 400b with first and second sensing oscillators 401a and 401b, which may be formed together as part of an IC 404. In this example the first and second oscillation signals Sosca and Soscb are supplied to respective counters 402a and 402b which are clocked by a clock signal CLK at a stable reference frequency Fref. With the clock signal CLK defining the same count period for each of the counters 401a and 401b, the counters 402a and 402b can count the number of edges in the respective oscillation signals Sosca and Soscb and provide respective count values a and b. The frequency ratio between the two oscillation signals fosca / fooscb can then be given by a / b. Alternatively, average periods may be determined via input capture to provide values a and b that can be used to determine a ratio of the oscillation periods Tosca / Toscb. In either case, the individual values a and b may be provided to a processing module 405 which can be implemented to refine the estimate a / b without requiring some hardware divide operation. For instance, the processing module 405 may fit a rational approximation via continued fractions to obtain small integers that bound a / b, perform a sliding window least squares fit on pairs of aand b values to reduce variance and then run a sigma-delta ratio estimator to provide a bitstream whose long-term average converges to a / b whilst noise-shaping quantization error to high frequencies, for effective averaging.

[0089] In some implementations, rather than compare a ratio of the different frequencies or periods of the two oscillator signals, it may be of interest to consider the difference between the two oscillator signals. As noted above, an advantage of using a sensing oscillator, in particular a delay based or relaxation type oscillate, is that the outputs are digital-like signals. Embedding the sensing functionality within such an oscillator effectively digitizes at source. In this case, processing can be used to process the difference between the two oscillation signals.

[0090] Figure 4c illustrates another example of sensor system 400c with first and second sensing oscillators 401a and 401b, which may be formed together as part of an IC 404. In the example of figure 4c, the respective oscillator signals Sosca and Soscb are input to a processing moule 406 which, in some examples, may implement an XOR (exclusive-or) function.

[0091] In general, if two ideal 50% duty-cycle square wave signals S1(t) and S2(t) with fundamental frequencies f1 and f2 were input to an XOR function, the output X(t) would be given by:

[0092] X(t) = S1(t) + S2(t) -2s1(t).s2(t) Eqn. (9)

[0093] Consequently, the transition rate of X(t) is approximately the sum of the input transition rates, yielding prominent high-frequency content near f1+f2. In this sense, the XOR behaves like a nonlinear mixer, which produces a strong carrier like frequency at f1 + f2 and which has an envelope value, i.e. a time averaged component, with a frequency equal to | f1 - f21 Low-pass filtering could thus be applied to the XOR output to effectively remove the high-frequency f1 + f2 component and to leave essentially a triangle waveform with a frequency equal to the absolute value of the frequency difference | f1 - f21 . Thus, an XOR processing module 406 followed by a low-pass filter could provide a simple indication of the absolute value of the difference in frequency between the two oscillator signals Sosca and Soscb. In some applications an absolutevalue of the difference may be sufficient, but in some applications the sign of the difference may be important. In which case instead of the processing module 406 implementing an XOR function, it could be configured as a PFD (phase and frequency detector) as will be understood by one skilled in the art. PFD may, for example, be conventionally used as part of an FLL (frequency-locked loop) to provide UP or DOWN pulses for increasing or decreasing the loop frequency as required, but the output of the PFD could instead be used to be provide a difference value with an associated polarity.

[0094] The sensor systems described with respect to figures 4a-4c thus make use of controlled oscillators with sensing functionality to provide an oscillator signal whose oscillation frequency / period varies with a measurand of interest.

[0095] In some applications, it may be advantageous to be able to vary at least one of the power consumption and sensitivity of the oscillator. For instance, it may be desirable to be able to operate the oscillator in a low-power mode, which may have a reduced sensitivity for routing monitoring, but then change to a higher sensitivity, but higher power, mode for events of interest, or simply to allow a device manufacturer or user to apply a desired trade-off between power consumption and sensitivity. More generally it may be beneficial to be able to tune the oscillator.

[0096] In general, the sensitivity of the oscillator depends on the extent of the change in oscillation period for a given change in measurand. For the example of a delay-based oscillator 200 such as described with reference to figure 2a and where the sensing component 205 provides a variable resistance R, it will be clear from equation (5) above that a change in the resistance R will result in a change in the rise time of each sensing delay element (but not the fall time) and there will be a change in oscillation period that depends on the change on resistance but also on the load capacitance CL and the number N of sensing delay elements.

[0097] The sensitivity of the sensing oscillator could therefore be adjusted by changing one or both of the load capacitance and the number of (active) sensing delay elements of the oscillator. Note varying the load capacitance CL would vary both the rise time and the fall time of the relevant delay element and if the load capacitance of each of the delay elements was varied in the same way, the resulting variation in oscillation period wouldbe linear with the variation in load capacitance. The overall oscillation period also depends on the number of delay elements and thus the period also varies linearly with the number of delay elements.

[0098] The load capacitance (for at least one delay element) could be selected varied in various ways, for example figure 5 illustrates that at least tuning capacitance Ctun could be selective connected in parallel with the gate capacitance of the delay stage to vary the overall load capacitance CL.

[0099] Varying the load capacitance can tune the oscillation period, and hence oscillation frequency, of the oscillator with a resulting impact on sensitivity, but varying the load capacitance in this way does not, itself, provide any significant power saving.

[0100] As will be understood by one skilled in the art, the power consumption P of each delay element, in terms of switching losses, is given by

[0101] P = f.CL.VDD2Eqn. (10)

[0102] where f is the oscillation frequency. The power consumption thus has a dependence on the load capacitance CL and VDD. However, the power consumption also has a dependence on frequency, and as noted, the frequency also varies with CL and VDD.

[0103] From equation 5 it can be seen that the oscillation period is linear with load capacitance CL and thus reducing the load capacitance would reduce the oscillation period, with a consequent reduction in sensitivity. This would also reduce the switching loss associate with each switching operation (as CL is halved), but, as noted, the reduction in oscillation period results in an increased switching frequency and thus there is a consequent increase in the number of switching operations and the power consumption due to switching losses essentially remains the same. It will thus be understood that varying the load capacitance provides a way to tune the oscillation period / frequency of the oscillator without substantially varying the switching losses.

[0104] Varying the number N of sensing delay elements which are enabled would also vary the sensitivity of the oscillator and reducing the number of active delay elements wouldreduce sensitivity, but again this would increase the frequency, and each delay element would thus switch more often and the overall number of switching operations would stay the same, so again without any significant change in switching losses.

[0105] However, varying the number of enabled delay element, i.e. enabling / disabling some delay elements, can vary some conduction losses associated with operation of the oscillator. It will be appreciated that during operation, each delay element will swap between the high output state, where the transistor 203 is off, and a low output state in which the transistor 203 is on. In the low output state, when the transistor 203 is on, current will flow from the VDD supply to ground via the sensing component 205 (with a value based on VDD and the resistance of the sensing component), with associated power loss. In operation, about half of the delay elements will be in the low output state at any time and thus these conduction losses scale according to the number of enabled delay elements in the oscillator.

[0106] Thus, by controllably varying the number of sensing delay elements which are active as part of the ring oscillator, a trade of between power and sensitivity can be achieved -although it will be understood that for the design of oscillator such as illustrated in figure 2a, there should always be a plurality of active delay elements and the number of active delay elements should be odd, so the minimum number of active delay elements in use will be three. As an example, consider that a ring oscillator such as illustrated in figure 2b is implanted with seven sensing delay elements 201. In use, in a high-power, high sensitivity mode, all the delay elements could be activated as part of the oscillator loop. In a low-power, low sensitivity mode only three of the delay elements 201, say the first three delay elements, may be active, with the other delay elements disabled, and the output of the third delay element is thus fed-back as the input for the first delay element. In a mid-power, mid sensitivity mode, five of the delay elements may be active.

[0107] Selectively changing the number of active delay elements can thus provide a trade-off between power consumption and sensitivity, with a consequent variation in oscillation period and hence oscillation frequency. As noted above, varying the load capacitance can also be used to tune the oscillation period and hence frequency, and, in some embodiments, some tuning of the load capacitance may be applied together with a variation in the number of active delay elements. For instance, if the number of activedelay elements is reduced to save power, the load capacitance could be varied to offset at least some of the change in oscillation frequency that would otherwise result.

[0108] Referring back to equation (10) it can be seen the power consumption due to switching losses has a dependence on the supply voltage VDD, and thus another way to tune the power consumption would be to vary the supply voltage. However, the rise time as defined in equation (2) also has a dependence on the supply voltage VDD and thus the oscillation period also varies with VDD in a non-linear manner. It can be shown that reducing the supply voltage can provide an overall reduction in power consumption, but as the voltage VDD must remain high enough to allow the oscillator to function correctly, there is a limit to the power savings that can be achieved.

[0109] The resistance R of the pull up element, (i.e. the resistance 204 or the resistance of the sensing element 205 for a given value of measurand) has an impact on the rise time as defined at equation (4), but no impact on the fall time. This means that the oscillation frequency will have a dependence on R, but the dependence will be non-linear. If the resistance were increased, this would increase the oscillation period and hence reduce the oscillation frequency, with a consequent saving in power consumption.

[0110] Varying the resistance R could thus, additionally or alternatively, be a way to vary the power consumption of the oscillator. For a CMOS implementation, the resistance R could be tuned by the use of active PMOS devices forming at least part of the pull up element for the delay stages. However, as discussed previously, in some material systems, such as GaN, the lack of p-channel devices may make this difficult. In such GaN systems, tuning of the capacitance would thus be preferred for tuning of the oscillation frequency and selectively varying the number of active delay stages may be preferred for tuning the power consumption.

[0111] The examples discussed above have focussed on temperature sensing, but as noted above embodiments may be implemented to provide sensing of a range of different properties.

[0112] For instance, strain sensing based on strain gauges that provide a change of resistance with applied strain are known. A strain sensor could be implemented using such aresistive strain gauge as the sensor component 205 of any of the examples of figure 2a, 2b, 3a or 3b.

[0113] Such as strain sensing oscillator could, for example, be used for one of the first or second oscillators 401a or 401b as part a sensor system 400 for strain sensing. In this case, the other one of the first or second oscillators 401a or 401b may be implemented with no strain sensing functionality, e.g. the oscillator may simply comprise a ring arrangement of non-sensing delay elements. In some implementations, however, each of the first and second oscillators 401a and 401b may be configured as strain sensing oscillators, with the first and second oscillators 401a and 401b being configured to provide different changes in resistance for a given applied strain. For instance, the resistive strain gauge(s) used for the variable resistance(s) of the first oscillator 401a may be arranged to provide a change in resistance to a given applied strain which is of the opposite polarity to that of the resistive strain gauge(s) used for the variable resistance(s) of the second oscillator 401b. In other words, the sensor component 205 of the first oscillator 401a to a given applied strain may result in a resistance change of +ARa, whereas the same strain applied to the sensor component 205 of the second oscillator 401b may lead to a resistance change of -ARb.

[0114] As described above the sensor system 400 of figure 4a may advantageously result in variations due to power supply fluctuations substantially cancelling. In addition, if temperature is not the measurand of interest and the first and second oscillators are implemented with resistances with the same or similar temperature coefficients of resistance, variations due to temperature may also substantially cancel.

[0115] In some embodiments, the sensor component 205 could be implemented as a component that exhibits magnetoresistance, i.e. a resistance that varies with an applied magnetic field. Such a sensor component could be implemented as part of a magnetic sensor.

[0116] In one possible implementation, instead of having a continuous resistor, e.g. a continuous 2DEG resistor, the resistor may be segmented into several low-aspect-ratio elements with metal connections between them. The low-aspect-ratio can act to reduce the magneto resista nee of the 2DEG and hence the difference in resistances betweenthe low-aspect-ratio and the high-aspect-ratio resistors scales with the square of the magnetic field applied to the resistors. Conversely, it may be desirable to use segmented 2DEG resistors to eliminate magnetoresistive effects in other sensor modalities, e.g. for sensing for temperature.

[0117] In some applications, the sensor component 205 may be arranged to provide chemical sensing. The sensor component 205 may, for example, comprise a functional material that reacts with one or more target chemicals in a way that results in a change of resistance of the sensor component. In some implementations, a 2DEG may be used as a chemical sensor due to its relatively high sensitivity to surface effects.

[0118] Figure 6 illustrates one example of a suitable sensor component 205 based on a 2DEG that could be used to provide chemical sensing functionality. The sensor component 205 comprises a first and second semiconductor materials 601 and 602 arranged to form a heterojunction that can give rise to a 2DEG. For instance, the first semiconductor material 401 may be GaN and the second semiconductor material 602 may be AIGaN. In such a heterojunction, a 2DEG 603 can form at the junction interface due to polarization effects, creating a highly conductive layer. Electrodes 604a and 604b are configured to allow a current to flow between the electrodes via the 2DEG 603.

[0119] In use, the charge or electric field distribution overlying the 2DEG can impact the carrier density and conductivity of the 2DEG and hence the resistance of 2DEG. This can be used to provide chemical sensing, e.g. for sensing of target chemical analytes. If the relevant analyte adsorbs onto, or otherwise interacts or reacts with, a sensing surface overlying the 2DEG in a manner that results in a change in charge or electric field distribution, this can result in the change in resistance of the 2DEG. In the example of figure 6, an optional functional layer or catalyst, e.g. platinum, 605, is disposed over at least part of the area of the 2DEG, where the functional layer 605 comprises a layer which is tailored to interact with the target chemical analytes of interest to provide a variation in charge or electric field. Various suitable functional layers are known and used in other electrochemical sensing methods and such functional layers could be used in embodiments of the present disclosure.The sensor component 205 illustrated in figure 6 could thus be used for the sensing delay element(s) 201 of the oscillator 200 of the examples of figures 2a or 2b, or as the variable resistance 205 of the examples of figures 3a and 3b, such that in the presence of the target chemical analyte(s), the output oscillation frequency of the oscillator varies.

[0120] Depending on the application, the oscillation frequency could simply be monitored for a change above a threshold amount which indicates that a relevant analyte is present and interacting with the functional layer. In which case the output of a single such oscillator could be monitored for a change in count value above a threshold, although it may be beneficial to provide a second oscillator constructed using similar 2DEGs but without the functional layer to provide calibration for power supply and / or temperature variations. For instance, the sensor system of any of figure 4a could be implemented with one of the first and second oscillators 401a and 401b having the chemical sensing functionality and the other having the non-chemical sensing arrangement. In some implementations, the amount of change in resistance of the 2DEG may vary depending on the relative concentration of the target analyte(s) and thus the sensor component of figure 4a may be used to provide a count value y indicative of the concentration. Alternatively the sensor systems of figures 4b or 4c could be used.

[0121] It will thus be clear that any sensor component which provides a resistance which varies with a measurand of interest could be used as the sensor component 205 of an oscillator of an embodiment of the present disclosure. Such an oscillator with sensing functionality could then be used as part of a sensor apparatus such as illustrated in any of figures 4a-4c, other it will be understood that other arrangements could be used.

[0122] The examples discussed above have focused on the sensor component 205 providing a variable resistance, for example so as to provide a variable propagation delay for a sensing delay element 201 in a delay-based oscillator. However, the propagation delay of a sensing delay element 201 could also be varied in other ways, for instance by varying the current that flows to / from the output of the sensing delay element 201 via the sensor component 205.Figure 7 illustrates one example of a sensor component 205 implemented as part of a sensing delay element 201 which provides a current which varies with the measurand of interest. In example of figure 7, the sensor component comprises a Hall sensor 701.

[0123] As will be understood by one skilled in the art, a Hall sensor is a known sensor for sensing a magnetic field based on the Hall effect. Typical Hall sensors comprise a planar material, with a first pair of electrodes 702a and 702b on opposite sides of the planar material to one another on a first axis and a second pair of electrodes 702c and 702d on opposite sides of the planar material to one another on a second axis, which may typically be substantially orthogonal to the first axis. For conventional Hall sensing, the Hall sensor is used in a voltage mode, in which a known voltage may be applied across one of the first and second pairs of electrodes to generate a DC bias current and potential difference between the other of the first and second pairs of electrodes is measured to provide a measurement signal of the applied magnetic field strength.

[0124] In the sensor component 205 illustrated in figure 7, however, the Hall sensor 701 is operated in a current mode in which the supply voltage VDD is applied to one of the electrodes of each of the first and second pairs, e.g. to electrodes 702a and 702c as illustrated such that, in use, current flows from these electrodes to the other electrodes 702b and 702d and the division of the current between the electrodes 702b and 702d varies with the applied magnetic field strength. The Hall sensor in this configuration thus provides current steering between the electrodes 702b and 702d, where the extent of the current steering varies with the strength of the of the applied magnetic field, that is the proportion of current steered to the electrode 702b rather than to electrode 702d varies with the applied field strength.

[0125] In the example of figure 7, the electrode 702b is connected to the output node of the sensing delay element 201, which is also connected to the n-channel transistor 203 of the sensing delay element. In this example the electrode 702d of the Hall sensor is connected to a suitable termination 703. In use, the current that flows to the output node of the sensing delay element 201 via the Hall sensor 701 thus had a dependence on the magnetic field strength acting on the Hall sensor, and hence the rise time of the sensing delay element varies with field strength so as to provide a variable propagation delay for the sensing delay element and hence the oscillator 200.An oscillator where the sensor component(s) 205 of the sensing delay element(s) 201 is implemented by a Hall sensor in this way, can thus provide an output oscillation signal with an oscillation period / frequency which varies with magnetic field strength. An oscillator with magnetic sensing functionality could be used as part of a current sense, for instance to sense a magnetic field generated by a current flowing in a conductive path.

[0126] The output of an oscillator with magnetic sensing functionality could be decoded using a stable reference clock such as described above with reference to figure 4b, or using another oscillator such as described with reference to figure 4a. In which case, either separate hall sensors could be used configured so that one oscillator receives more current and the other less current for the same magnetic field, or the same Hall sensor may be shared by sensing delay elements of the two oscillators 200a and 200b. In the example discussed with reference to figure 7, the output node of the sensing delay element 201 is connected to electrode 702b of the Hall sensor so the rise time of the sensing delay element 201 varies with the current steered to this electrode 702b. In this example the current steered to electrode 702d is not used. However, it will be understood that any increase in current steered to electrode 702b will result in a consequent reduction in current steered to electrode 702d and vice versa. Thus, any change in the current through electrode 702b will be matched by an opposite change in the current steered to electrode 702d. Rather than simply waste the current which is steered to electrode 702d, this electrode could be connected to the output node of a sensing delay element of the second oscillator to provide an equal and opposite change in oscillation frequency. This arrangement avoids power wastage can provide greater sensitivity. Figure 8 illustrates an example of a sensor component where a first oscillator arrangement 801a comprises a plurality of transistors 203 each coupled to a first output electrode of a respective Hall sensor 701 and a second oscillator arrangement 801b comprises a plurality of transistors 203 each coupled to a respective second output electrode of the Hall sensor 701s. The Hall sensors 701 are configured such that a given change in field strength will lead to generally the same variation in current steering between the first and second output electrodes. The frequency the first and second oscillator arrangements will thus both vary with magnetic field strength in a substantially opposite fashion to one another. The output of the first and second oscillatorarrangements 801a and 801b may be decoded in a similar manner as illustrated in figure 4a or 4b.

[0127] In some embodiments, multiple Hall sensors could be arranged together in a stacked configuration to provide a greater change in current for a given change in magnetic field strength. Figure 9 illustrates one example of two-layer stack of Hall sensors 701. Each of the Hall sensors has two input electrode and first and second output electrodes and is arranged to steer current received at the input electrodes between first and second output electrodes. The Halls sensors are configured so that a given magnetic field will lead to similar steering of current between the first and second output electrodes for each of the Hall sensors. A first tier 901 of Hall sensors comprises one or more Hall sensors and, in this example, comprises two Hall sensors 701-1a and 701-1b. The second tier 902 of Hall sensors comprises two Hall sensors 701-2a and 701-2b. The first output electrode of each of the Hall sensors of the first tier is configured to provide current to the input electrodes of one of the Hall sensor of the second tier, in this example Hall sensor 701 -2a, and the second output electrode of each of the Hall sensors of the first tier is configured to provide current to the input electrodes of the other Hall sensor of the second tier, in this example 701-2a. The first electrode of the Hall sensor 701-2a provides current to the oscillator arrangement 801a and the second electrode of the Hall sensor 701-2b provides current to the other oscillator arrangement 801b.

[0128] In use, in the first tier, the Hall sensors operate to steer current between the first and second output electrodes such that a fraction Pa of the current through the Hall sensor is steered to the first output electrode and a fraction Pb (Pb = 1 - Pa) is steered to the second output electrode. Assuming the input current lin for each of the Hall sensors 701-1a and 701-1b is the same, the current supplied to the Hall sensor 701-2a of the second tier is thus equal to 2lin*Pa and the current supplied to the Hall sensor 701-2a of the second tier is thus equal to 2lin*Pa.

[0129] The Hall sensors of the second tier steer current in the same fashion and thus Pa of the input current supplied to the Hall sensor 701-2a is steered to its first output electrode. The current supplied to the first oscillator arrangement 801a is thus Pa(lin*Pa) = lin*Pa2. Similarly the current steered to the second electrode of the Hall sensor 501 -2b of the second tier, and hence to the second oscillator arrangement 801b is thus Pb(lin*Pb) =lin*Pb2. In this example the current at the second output electrode of the Hall sensor 701-2a and the first output electrode of Hall sensor 701-2b is ignored and these electrodes may be connected to a suitable termination 703.

[0130] It can thus be seen that stacking of the Hall sensors in this way effectively amplifies the current steering due to applied magnetic field.

[0131] It should be noted that Hall sensors are just one example of a sensor component that can provide a variable current for a sensing delay element of an oscillator with sensing functionality and other type of sensor component where an output current varies with a measurand of interest could be implemented.

[0132] The use of such current steering sensor have been described in the context of a delaybased oscillator to provide a variable drive current for charging / discharging the gate capacitance of the next delay element. It should be understood that a similar arrangement could be applied for a relaxation oscillator such as described with reference to figure 3b to vary the strength of a current for charging / discharging the capacitor C of the relaxation oscillator.

[0133] Embodiments of the present disclosure thus provide for oscillators with a sensor functionality, i.e. where an oscillation frequency / period of the oscillator varies with a measurement of interest. Embodiments may be implemented without requiring p-channel devices and are suitable for use in material system such as GaN, although the principles could be applied making use of p-channel devices. The examples have been described with a switching device such as an n-channel or HEMT transistor as a controlled pull-down element and the sensing functionality implemented as part of a pull-up arrangement but it will be understood that a delay element could be implemented in some material systems with a controlled pull-up arrangement and a sensor component, such as a variable resistance or current steering element, as part of a pull-down arrangement. It will also be understood that the oscillators may comprise other components, for instance series resistances may be implemented between the delay elements in some applications. The examples have been described in respect of single-ended oscillators but differential oscillators could be used in some embodiments, at the expense of an increased number of components.Embodiments may be implemented in a host device, especially a portable and / or battery powered host device such as a mobile computing device for example a laptop, notebook or tablet computer, or a mobile communication device such as a mobile telephone, for example a smartphone. The device could be a wearable device such as a smartwatch. The host device could be a games console, a remote-control device, a home automation controller or a domestic appliance, a toy, a machine such as a robot, an audio player, a video player. It will be understood that embodiments may be implemented as part of a system provided in a home appliance or in a vehicle or interactive display. There is further provided a host device incorporating the above-described embodiments.

[0134] It should be noted that the above-mentioned embodiments illustrate rather than limit the invention, and that those skilled in the art will be able to design many alternative embodiments without departing from the scope of the appended claims. The word “comprising” does not exclude the presence of elements or steps other than those listed in a claim, “a” or “an” does not exclude a plurality, and a single feature or other unit may fulfil the functions of several units recited in the claims. Any reference numerals or labels in the claims shall not be construed so as to limit their scope

Claims

CLAIMS1. A sensor apparatus comprising:a first oscillator comprising at least one capacitance which is charged and discharged to generate a first oscillator signal; and a sensor component with an electrical property that varies with a measurand of interest to vary at least one of a rate of charging and discharging of said at least one capacitance and hence an oscillation frequency of the first oscillator signal.

2. The sensor apparatus of claim 1 where the first oscillator comprises a first ringoscillator comprising a plurality of delay elements and wherein at least one of the delay elements comprises a sensing delay element which comprises said sensor component with an electrical property that varies with a measurand of interest to vary a propagation delay of the sensing delay element.

3. The sensor apparatus of claim 2 wherein the sensing delay element comprises the sensor component connected in series with a switching device, wherein a control terminal of the switching device is configured to be driven by an input to the sensing delay element and an output of the sensing delay element is taken from a node between the series connected sensor component and switching device.

4. The sensor apparatus of claim 3 where the switching device comprises is an n- channel transistor or an enhancement high electron mobility transistor.

5. The sensor apparatus of claim 4 further comprising a variable capacitance connected to a gate terminal of the n-channel transistor or an enhancement high electron mobility transistor.

6. The sensor apparatus of any of claims 2 to 5 wherein each of the plurality of delay elements of the first ring oscillator comprises a sensing delay element.

7. The sensor apparatus of any of claims 2 to 6 wherein the first ring oscillator is configured so that a number of active delay elements in the ring can be selectively controlled.

8. The sensor apparatus of claim 1 wherein the first oscillator is one of a first resistive-capacitive resonance oscillator and a first resistive-capacitive relaxation oscillator.

9. The sensor apparatus of any of claims 1 to 8 wherein the sensor component comprises a resistance having a resistance value which varies with the measurand of interest.

10. The sensor apparatus of claim 9 wherein the measurand of interest is temperature and the resistance has a first temperature coefficient of resistance.

11. The sensor apparatus of claim 9 wherein the measurand of interest is strain and the variable resistance comprises a resistive strain sensor.

12. The sensor apparatus of claim 9 wherein the resistance comprises a two- dimensional electron gas resistive element.

13. The sensor apparatus of claim 12 configured for sensing for one or more target chemical analytes of interest, wherein the sensor apparatus comprises a functional layer overlying the two-dimensional electron gas resistive element, wherein the functional layer is configured to interact with the one or more target chemical analytes of interest to vary a charge or electric field distribution over the two-dimensional electron gas and hence the resistance of the two-dimensional electron gas resistive element.

14. The sensor apparatus of any of claims 1 to 8 wherein the sensor component comprises a variable current component that provides a current which varies with the measurand of interest.

15. The sensor apparatus of claim 14 wherein the measurand of interest is a magnetic field strength and the variable current component comprises a Hall sensor configured to provide current steering between first and second output electrodes.

16. The sensor apparatus of claim 15, when dependent on claim 2, further comprising a second ring oscillator with at least one sensing delay element wherein the Hall sensor is configured as a shared sensing component for a sensing delay element of each of the first and second ring oscillators, wherein the Hall sensor is connected as part of the sensing delay element of the first ring oscillator via the first output electrode and is connected as part of the sensing delay element of the second ring oscillator via the second output electrode.

17. The sensor apparatus of claim 16 wherein the variable current component comprises a plurality of Hall sensors configured in a stacked arrangement such that an output current from one of the first and second output electrodes of a first one of the Hall sensors provides an input current for a second one of the Hall sensors.

18. The sensor apparatus of any of claims 1 to 15 further comprising a second oscillator configured to generate a second oscillator signal, wherein the first and second oscillators are configured such that any variation in frequency of the second oscillator signal with the measurand of interest is different to the variation in frequency of the first oscillator signal.

19. The sensor apparatus of claim 18 wherein the second oscillator is configured such that the frequency of the second oscillator signal does not vary with the measurand of interest.

20. The sensor apparatus of claim 18 or claim 19 further comprising a counter configured to count a number of oscillations of one of the first and second oscillator signals in a count period, and a frequency divider configured to apply frequency division to the other of the first and second oscillator signals to define the count period.

20. The sensor apparatus of claim 18 or claim 19 further comprising a counter configured to generate respective counts of a number of oscillations of each of the first and second oscillator signals in a count period defined by a reference clock.

21. The sensor apparatus of claim 18 or claim 19 comprising a processing module configured to implement an exclusive-or function on the first and second oscillator signals.

22. The sensor apparatus of claim 1 implemented as a gallium nitride based integrated circuit.

23. A sensor apparatus comprising:a first ring oscillator comprising a plurality of delay elements configured to generate a first oscillator signal;wherein at least one delay element comprises a sensor component connected between a first voltage and a delay element output node and a switching device connected between the delay element output node and a second voltage;wherein the sensor component provides at least one of a variable resistance and a variable current that varies with a measurand of interest.

24. A sensor apparatus comprising:a first ring oscillator comprising a plurality of delay elements configured to generate a first oscillator signal, wherein at least one of the delay elements is configured to provide a propagation delay which varies with a measurand of interest so as to vary a frequency of the first oscillator signal; anddecode circuitry configured to receive and decode the first oscillator signal so as to provide a measurement signal indicative of the measurand of interest.

25. The sensor apparatus of claim 24 wherein the sensor apparatus comprises a second ring oscillator comprising a plurality of delay elements configured to generate a second oscillator signal and wherein the second oscillator signal is used to define a count period for counting a number of oscillations of the first oscillator signal.

Images