Method for switching between first communication mode and second communication mode

Abstract

The invention discloses a method for switching between a first communication mode and a second communication mode, and the method comprises the steps: determining a dual-mode phase difference between a first mode baseband signal and a second mode parameter based on the first mode baseband signal and the second mode parameter; generating an interleaved phase coding sequence in response to the dual-mode phase difference; pre-modulating the first mode signal in parallel before switching by using the coding sequence, and generating a pre-activated second mode virtual signal; switching from the first mode to the second mode is executed based on a parallel fusion phase track generated after the phases of the two preprocessed signals are subjected to time-varying weighted fusion. According to the method, a smooth switching track is constructed through parallel phase preview and phase domain fusion, phase continuity and frequency stability in the switching process are ensured, switching time delay and the data error rate are remarkably reduced, and the communication service quality of the heterogeneous network is improved.

Description

Technical Field

[0001] The present invention relates to communication technology, and in particular to a method for switching between a first communication mode and a second communication mode. Background Art

[0002] With the rapid development of technologies such as the Internet of Things (IoT), Industrial Internet, Smart Grid, and Internet of Vehicles (IoV), a single communication technology can no longer meet the comprehensive requirements of data transmission reliability, speed, coverage, and mobility in complex and diverse application scenarios. Therefore, heterogeneous networks (HNs), integrating multiple communication technologies, have become a key direction of current technological evolution. For example, in smart home or industrial automation scenarios, systems may need to leverage the wide coverage and high reliability of high-speed power line carrier (HPLC) communications to transmit critical control signaling, while also leveraging the high speed and flexibility of high-frequency radio (HRF) to transmit high-definition video streams or enable large-scale device networking. In this heterogeneous network architecture, enabling smooth, rapid, and seamless switching between different communication modes for end devices or network nodes is crucial for ensuring service continuity, improving user experience, and optimizing network resource utilization. Enabling seamless switching between two or more communication modes with distinct physical layer characteristics is one of the core technical challenges that must be overcome to advance the maturity and widespread application of heterogeneous network technology.

[0003] Currently, research and application of handover between different communication modes in heterogeneous networks primarily focus on the network protocol layer and the relatively simple physical layer control plane. A common technique is hard handover based on a break-before-make mechanism. In this scheme, when a handover condition is triggered, the device first completely terminates the current communication link, including releasing physical layer resources, tearing down the data link layer connection, and clearing related protocol stack states. The system then initiates the initialization process for the second communication mode, including channel scanning, synchronization, authentication, authorization, and establishing a new data link. Another more advanced solution involves handover management at the network layer or higher. For example, Mobile IP technology allows devices to retain their IP addresses when switching underlying physical links (e.g., from WLAN to 4G / 5G), thereby maintaining connectivity for upper-layer applications. This approach achieves service rerouting through signaling and context transfer within the upper-layer protocol stack. In addition, there are some simple coexistence schemes based on time division multiplexing, that is, the two communication modes work in turns within strictly divided time slices, which essentially does not achieve true collaborative work and smooth switching.

[0004] However, the aforementioned existing technical solutions still face profound and specific physical layer technical bottlenecks in their pursuit of truly seamless handover. The core issue lies in the signal phase discontinuity at the switching moment and the resulting instantaneous frequency loss. Specifically, when switching from HPLC mode to HRF mode using a hard break-before-make switching method, the initial transmit phase of the HRF mode is typically arbitrary and bears no relation to the final phase of the HPLC signal immediately before the interruption. This sudden phase change in the complex signal vector space is mathematically equivalent to a Dirac delta function whose instantaneous frequency tends to infinity. Physically, it manifests as a strong pulse interference at the receiver. This interference directly impacts the phase-locked loop (PLL) circuit at the receiving end, highly likely causing the PLL to lose lock and forcing it into a time-consuming frequency and phase reacquisition process. During this process, data demodulation is completely interrupted, resulting in sudden bit errors and data loss for several or even dozens of OFDM symbols, which runs counter to the goal of seamless handover. Furthermore, existing solutions often base handover decisions on crude metrics, relying heavily on macro-average indicators like signal strength (RSSI) and signal-to-noise ratio (SNR). These metrics change slowly and fail to reflect the instantaneous dynamics of the signal phase. Consequently, the system may make a handover decision at an unstable moment when the signal phase is experiencing significant fluctuations. Even if the upper-layer protocol enables fast rerouting, this suboptimal handover timing selection at the physical layer can still introduce unnecessary post-oscillations, prolonging the convergence time of the receiver's equalizer and timing recovery loop, and further degrading handover performance. Summary of the Invention

[0005] The purpose of the invention is to solve the problems of signal interruption and switching delay caused by phase discontinuity in existing heterogeneous mode switching.

[0006] Technical solution, a method for switching between a first communication mode and a second communication mode, comprising: determining a dual-mode phase difference between a current phase trajectory of the first mode and a target phase trajectory of the second mode based on a first-mode baseband signal and a second-mode parameter; generating an interleaved phase coded sequence in response to the dual-mode phase difference; Using an interleaved phase coding sequence, premodulating the first mode baseband signal in parallel within a preprocessing time before switching to generate a premodulated first mode signal, and combining the second mode target phase trajectory to generate a preactivated second mode virtual signal; At a predetermined switching time, switching from the first mode to the second mode is performed based on the pre-modulated first mode signal and the pre-activated second mode dummy signal.

[0007] According to one aspect of the present application, performing switching from a first mode to a second mode based on a pre-modulated first mode signal and a pre-activated second mode virtual signal includes: Extracting a first mode instantaneous phase from the pre-modulated first mode signal and extracting a second mode instantaneous phase from the pre-activated second mode virtual signal; Applying the time-varying fusion coefficient, the instantaneous phase of the first mode and the instantaneous phase of the second mode are weightedly fused in the phase domain to generate a parallel fusion phase trajectory with a smooth transition between the two modes; Based on the parallel fusion phase trajectory, switching is performed to ensure phase continuity during the switching process.

[0008] According to one aspect of the present application, generating an interleaved phase coding sequence in response to a dual-mode phase difference includes: Decomposing the dual-mode phase difference into a pair of complementary phase adjustment functions, including a first and a second mode adjustment function, wherein the sum of the first and the second adjustment functions is equal to the dual-mode phase difference within a pre-processing time of switching; An interleaved phase coding sequence is generated based on the first pattern adjustment function and the second pattern adjustment function.

[0009] According to one aspect of the present application, the dual-mode phase difference is decomposed into a pair of complementary phase adjustment functions, including: Decomposing the dual-mode phase difference into linear and nonlinear components; A pair of preset orthogonal basis functions are used to fit the nonlinear components; A complementary phase adjustment function pair is constructed based on the linear component and the fitted nonlinear component.

[0010] According to one aspect of the present application, the operations performed in parallel using an interleaved phase-coded sequence include: Applying a progressive weighting function to progressively apply the phase adjustment amount carried by the interleaved phase coding sequence to the first mode baseband signal to generate a premodulated first mode signal; Furthermore, the second mode target phase trajectory is pre-compensated using an interleaved phase coding sequence, and a power ramp function is applied to smoothly increase the power within a pre-processing time, thereby generating a pre-activated second mode virtual signal.

[0011] According to one aspect of the present application, before generating an interleaved phase coding sequence based on the first pattern adjustment function and the second pattern adjustment function, the method further includes: Applying the first mode adjustment function to the current phase trajectory of the first mode to obtain an adjusted phase trajectory of the first mode; Applying the second mode adjustment function to the second mode target phase trajectory to obtain the second mode adjusted phase trajectory; The optimal interleaving moment is determined from multiple candidate moments based on a composite scoring function, which is used to generate an interleaved phase-coded sequence. The composite scoring function simultaneously considers the distance stability and first-order derivative matching between the two adjusted phase trajectories.

[0012] According to one aspect of the present application, generating an interleaved phase coded sequence based on a first mode adjustment function and a second mode adjustment function in an OFDM-based system includes: Dividing a plurality of OFDM subcarriers into at least one edge carrier group and one center carrier group; Assigning different grouping weights to the edge carrier group and the center carrier group, and non-uniformly allocating a total phase adjustment amount determined by the adjustment function to the subcarriers within each group based on the grouping weights to determine a phase adjustment amount for each subcarrier; An interleaved phase code sequence is generated based on the phase adjustment amount of each subcarrier.

[0013] According to one aspect of the present application, determining the optimal interleaving time based on a composite scoring function includes: For each candidate moment, a stability index is obtained based on the second-order derivative of the phase distance function of the two adjusted phase trajectories; Based on the first-order derivative difference of the two adjusted phase trajectories, a phase derivative matching index is obtained; Perform weighted summation on the stability index and the phase derivative matching index to generate a score value of the composite scoring function; The candidate moment with the highest score is selected as the optimal interleaving moment.

[0014] According to one aspect of the present application, before performing the switching, the method further includes: calculating a phase continuity indicator based on first-order and / or second-order derivatives of the parallel fused phase trajectory within a time window around the predetermined switching moment; The phase continuity index is compared with a preset threshold, and when the index is worse than the threshold, the predetermined switching moment is fine-tuned until an optimal switching moment that meets the threshold is determined; wherein the switching is performed at the optimal switching moment.

[0015] According to one aspect of the present application, generating an interleaved phase coded sequence includes: A time domain window function is applied to the phase adjustment amount of each subcarrier to control the concentration of the phase adjustment amount in the time domain.

[0016] According to one aspect of the present application, the time-varying fusion coefficient is a monotonically increasing S-shaped function within a preset switching time window, and its function value is close to 0 at the starting point of the window and close to 1 at the end point of the window.

[0017] Beneficial effects: Through parallel phase pre-evolution and creative phase domain fusion, a smooth switching trajectory is constructed, ensuring phase continuity and frequency stability during the switching process, significantly reducing switching delay and data error rate, and improving the communication service quality of heterogeneous networks. BRIEF DESCRIPTION OF THE DRAWINGS

[0018] Figure 1 It is a flow chart of the present invention.

[0019] Figure 2 is a flow chart of switching from the first mode to the second mode of the present invention.

[0020] Figure 3 It is a flow chart of the present invention for generating an interleaved phase coding sequence.

[0021] Figure 4 This is a flow chart of the present invention for decomposing the dual-mode phase difference into a pair of complementary phase adjustment functions. DETAILED DESCRIPTION

[0022] In order to make the purpose, technical solutions and advantages of the present invention clearer, the following Figures 1 to 4 The present invention will be further described in detail with reference to the following specific embodiments. It should be understood that the specific embodiments herein are only intended to explain the present invention, rather than to limit it.

[0023] In order to solve the technical problems existing in the prior art, the present invention discloses a method for switching between a first communication mode and a second communication mode, comprising: determining a dual-mode phase difference between the first mode baseband signal and a second mode parameter; generating an interleaved phase coding sequence in response to the dual-mode phase difference; using the coding sequence, pre-modulating the first mode signal in parallel before switching, and generating a pre-activated second mode virtual signal; and executing switching from the first mode to the second mode based on a parallel fusion phase trajectory generated by time-varying weighted fusion of the phases of the two pre-processed signals.

[0024] Example 1: This embodiment provides a method for switching between a first communication mode (e.g., HPLC) and a second communication mode (e.g., HRF). This method accurately analyzes and models the phase difference between the two modes before switching, and develops a precise phase compensation and interleaving scheme based on this analysis, laying the foundation for the subsequent smooth switching execution phase. Specifically, the method includes the following steps: Step S100: Dual-mode signal feature extraction and analysis. Specifically, based on a first-mode baseband signal and a second-mode parameter, a dual-mode phase difference is determined between a current phase trajectory of the first mode and a target phase trajectory of the second mode. In this embodiment, this step quantifies the phase state difference between the two communication modes before switching, providing accurate input data for subsequent compensation.

[0025] The first-mode baseband signal s_HPLC(t) refers to the complex baseband signal of the currently operating HPLC system. The second-mode parameters include the carrier frequency f_HRF and the modulation parameter set M_HRF of the second communication mode (HRF), which together define the signal characteristics of the HRF mode. The first-mode current phase trajectory φ_HPLC(t) refers to the continuous instantaneous phase extracted from s_HPLC(t) and deconvolved, reflecting the actual phase change. The second-mode target phase trajectory φ_HRF_target(t) refers to the theoretical expected phase trajectory in the absence of disturbances, calculated based on the HRF mode parameters and its phase evolution model. The dual-mode phase difference Δφ(t) = φ_HRF_target(t) - φ_HPLC(t) is the starting point for all subsequent compensation algorithms.

[0026] In a certain process, the steps include HPLC phase extraction, HRF target phase calculation and phase difference analysis, as follows: The HPLC baseband signal s_HPLC(t) is read, its instantaneous phase is extracted by Hilbert transform, and a phase unwrapping algorithm is performed to obtain a continuous first-mode current phase trajectory φ_HPLC(t).

[0027] The carrier frequency f_HRF and modulation parameter set M_HRF of the HRF mode are read, and the target phase trajectory φ_HRF_target(t) of the second mode is calculated through the preset phase evolution model.

[0028] The difference between the two phase trajectories is calculated to obtain the dual-mode phase difference Δφ(t). Furthermore, for more refined processing, the first-order derivative dΔφ / dt and the second-order derivative d of Δφ(t) can also be calculated. 2 Δφ / dt 2 , and these data together constitute a two-mode phase feature matrix Φ.

[0029] By accurately obtaining the dual-mode phase difference Δφ(t) and its dynamic change characteristics (derivatives), the present invention can fundamentally predict the degree of phase mutation that may occur during switching, thereby transforming the problem from a passive post-compensation to an active pre-planning.

[0030] Step S200: Phase interleaving precoding. Specifically, in response to the dual-mode phase difference, an interleaved phase coding sequence is generated. In this embodiment, this step includes multiple progressive sub-steps that together complete the entire process from analyzing the phase difference to formulating a compensation strategy and then generating a specific code.

[0031] Step S210: complementary phase trajectory design, including: The dual-mode phase difference is decomposed into a pair of complementary phase adjustment functions, wherein the function pair includes a first mode adjustment function and a second mode adjustment function, and the sum of the two adjustment functions is equal to the dual-mode phase difference within a switching preprocessing time.

[0032] The specific implementation includes: decomposing the dual-mode phase difference into a linear component and a nonlinear component; fitting the nonlinear component using a pair of preset orthogonal basis functions; and constructing the complementary phase adjustment function pair based on the linear and fitted nonlinear components.

[0033] In some embodiments, the steps of decomposing the phase difference, fitting the nonlinear component, and constructing the complementary function are included, specifically: The dual-mode phase difference Δφ(t) obtained in step S100 is decomposed into a linear component Δφ_linear(t)=a1t+a0 and a nonlinear component Δφ_nonlinear(t) by polynomial fitting or other methods.

[0034] Perform spectral analysis on the nonlinear component Δφ_nonlinear(t) to find its dominant frequency ω_main. Use an orthogonal basis function pair based on ω_main, such as ψ1(t) = sin(ω_main·t) and ψ2(t) = cos(ω_main·t), to fit Δφ_nonlinear(t) using the least squares method to obtain the optimal coefficients α_opt and β_opt.

[0035] The fitting results are combined with the linear component to construct a pair of complementary phase adjustment functions [f_HPLC(t),f_HRF(t)]. For example, we can design f_HPLC(t) = w_alloc(t) · [α_opt · ψ1(t) + Δφ_linear(t)] and f_HRF(t) = (1-w_alloc(t)) · [β_opt · ψ2(t)]. By designing the weighting function w_alloc(t), we ensure that the sum of the two is approximately equal to Δφ(t) at any time within the preprocessing time.

[0036] This embodiment doesn't require a single system to handle all phase compensation. Instead, the total phase difference is split in two, with the two systems moving in opposite directions, each completing a portion of the phase adjustment. This shared responsibility significantly reduces the adjustment requirements for a single system, making the compensation process smoother and easier to implement.

[0037] Step S220, calculating the optimal interweaving point, includes: Before generating a coding sequence, the method also includes applying the first mode adjustment function to the first mode current phase trajectory to obtain a first mode adjusted phase trajectory; applying the second mode adjustment function to the second mode target phase trajectory to obtain a second mode adjusted phase trajectory; and based on a composite scoring function that simultaneously considers the distance stability and first-order derivative matching between the two adjusted phase trajectories, determining an optimal interleaving moment from multiple candidate moments, wherein the scoring function is determined by calculating a stability index and a phase derivative matching index; and performing a weighted summation of the stability index and the phase derivative matching index.

[0038] Specifically, for each candidate moment, a stability index is obtained based on the second-order derivative of the phase distance function of the two adjusted phase trajectories; a phase derivative matching index is obtained based on the difference between the first-order derivatives of the two adjusted phase trajectories; the stability index and the phase derivative matching index are weighted and summed to generate a scoring value of the composite scoring function; and the candidate moment with the highest scoring value is selected as the optimal interleaving moment.

[0039] In this embodiment, the process mainly includes constructing the adjusted trajectory, calculating the composite score, and determining the optimal time, specifically: Calculate φ_HPLC_adj(t)=φ_HPLC(t)+f_HPLC(t) and φ_HRF_adj(t)=φ_HRF_target(t)-f_HRF(t).

[0040] For each candidate moment t_i in the time window: calculate the stability index S_i, for example, S_i and |d 2 |φ_HPLC_adj(t_i)-φ_HRF_adj(t_i)| / dt 2 | is inversely proportional to the smoothness of the interleaving point. The derivative matching index Mi_i is calculated. For example, Mi_i is inversely proportional to |dφ_HPLC_adj / dt|_{t_i}-dφ_HRF_adj / dt|_{t_i}|, indicating whether the phase change trend at the switching moment is consistent.

[0041] The composite score is obtained by weighted summation Score_i=w_s·S_i+w_m·M_i. In this embodiment, the weight coefficients can be preferably set to w_s=0.6 and w_m=0.4.

[0042] The time with the highest Score_i is selected as the optimal interleaving time t_cross, and the phase value, first-order derivative, second-order derivative, etc. at the time t_cross are stored in the phase interleaving parameter set P_cross.

[0043] The ideal switching point requires not only equal phase values but also close phase change rates and accelerations to avoid post-switch signal frequency and phase jitter. This step uses a multi-dimensional composite scoring function to find the perfect intersection point where both velocity and acceleration match.

[0044] Sub-step S230: Interleaved coding sequence generation In an OFDM system, generating a sequence includes grouping multiple OFDM subcarriers into at least one edge carrier group and one center carrier group; assigning different grouping weights to the edge carrier group and the center carrier group; and, based on the grouping weights, unevenly distributing a total phase adjustment amount to the subcarriers within each group to determine a phase adjustment amount for each subcarrier. An interleaved phase code sequence is generated based on the phase adjustment amount for each subcarrier.

[0045] Furthermore, it includes applying a time domain window function to the phase adjustment amount of each subcarrier to control the concentration of the phase adjustment amount in the time domain.

[0046] In this embodiment, the steps mainly include subcarrier grouping and weighting, allocating phase adjustment amounts, time domain windowing, and sequence assembly, which are specifically as follows: The N_sc subcarriers in the system are divided into an edge carrier group G_edge and a center carrier group G_center according to their positions in the frequency band. Different weights are assigned to different groups, for example, w_edge = 0.7 and w_center = 0.3.

[0047] The total phase adjustment amount (e.g., φ_cross) obtained in sub-step S220 is unevenly distributed to each subcarrier according to the above weights to obtain a phase adjustment amount φ_adj_k for each subcarrier. Typically, edge carriers carry larger adjustment amounts, while center carriers carry smaller adjustment amounts.

[0048] A time domain window function g_k(n) is designed. Preferably, a Hanning-Gaussian composite window function can be used to precisely control the effective range and shape of the phase adjustment in the time domain.

[0049] Finally, the code c_k(n) for each subcarrier is composed of φ_adj_k and g_k(n). Combining the c_k(n) of all subcarriers gives the final output interleaved phase code sequence C_phase.

[0050] This step is a sophisticated mapping of the compensation strategy onto the actual OFDM physical layer. By grouping and weighting subcarriers, the channel characteristics of different subcarriers can be exploited to more efficiently and stably carry phase adjustment information. Time-domain windowing ensures precise placement of phase adjustments and avoids interference with adjacent OFDM symbols.

[0051] Through steps S100 and S200, this embodiment generates an interleaved phase-coded sequence C_phase that incorporates a complete and refined compensation strategy. This sequence serves as the blueprint for connecting two heterogeneous communication modes and will be used in subsequent parallel pre-evolution processing (as described in Examples 2 and 3), ultimately achieving seamless handover.

[0052] Example 2. This example is based on the interleaved phase coding sequence C_phase generated in Example 1, and is combined with the pre-modulated first mode signal s'_HPLC(t) and the pre-activated second mode virtual signal s_HRF_virtual(t) generated in parallel (the generation process of which is detailed in Example 3), and aims to describe in detail the final execution stage of the present invention to achieve seamless switching.

[0053] Step S300: Dual-mode parallel phase state fusion. Specifically, switching from the first mode to the second mode is performed based on the premodulated first mode signal and the preactivated second mode virtual signal, including: extracting the first mode instantaneous phase from the premodulated first mode signal, and extracting the second mode instantaneous phase from the preactivated second mode virtual signal; applying a time-varying fusion coefficient to perform phase domain weighted fusion on the first mode instantaneous phase and the second mode instantaneous phase to generate a parallel fusion phase trajectory that smoothly transitions between the two modes; and performing switching based on the parallel fusion phase trajectory to ensure phase continuity during the switching process.

[0054] The instantaneous phase of the first mode and the instantaneous phase of the second mode are weightedly fused in the phase domain by applying a time-varying fusion coefficient to generate a parallel fused phase trajectory that smoothly transitions between the two modes.

[0055] The time-varying fusion coefficient λ(t) is a control function whose value smoothly changes from near 0 to near 1 within the switching time window. It is used to dynamically adjust the weights of the two mode phases during the fusion process. When λ(t) is close to 0, the fusion phase is primarily determined by the first mode; when λ(t) is close to 1, it is primarily determined by the second mode. The parallel fusion phase trajectory φ_parallel(t) is the weighted average of the instantaneous phases of the two modes after preprocessing. It is a virtual, smooth phase trajectory along which the actual switching process will proceed.

[0056] In this embodiment, the time-varying fusion coefficient is a monotonically increasing S-shaped function within a preset switching time window, with its function value approaching 0 at the start of the window and approaching 1 at the end of the window. In this embodiment, λ(t) is preferably constructed using a hyperbolic tangent function. For example, within a switching time window with a total duration of T_win (t varying from -T_win / 2 to +T_win / 2), λ(t) can be designed as λ(t) = 0.5*[1+tanh(β_fusion*t / T_win)]. β_fusion is a fusion speed control factor, which can be set to 3, for example. This design ensures that λ(t) is approximately 0.12 at the start of the window (-T_win / 2), 0.5 at the center (t=0), and approximately 0.88 at the end of the window (+T_win / 2), presenting a smooth S-shaped growth curve.

[0057] The implementation process of phase domain weighted fusion is as follows: First, the instantaneous phases of the premodulated first-mode signal s'_HPLC(t) and the preactivated second-mode virtual signal s_HRF_virtual(t) are extracted, yielding φ_HPLC_inst(t) and φ_HRF_inst(t), respectively. Subsequently, the parallel fusion phase trajectory is calculated by applying the aforementioned time-varying fusion coefficient λ(t): φ_parallel(t) = (1-λ(t))·φ_HPLC_inst(t) + λ(t)·φ_HRF_inst(t).

[0058] It should be noted that when performing phase-domain weighted superposition, in order to avoid errors caused by phase wrapping (2π ambiguity), it is preferred to first align the two instantaneous phases φ_HPLC_inst(t) and φ_HRF_inst(t), for example, by adding or subtracting integer multiples of 2π to ensure that their difference is always in the interval [-π,π], and then perform weighted calculation.

[0059] The calculated φ_parallel(t) and its first-order and second-order derivatives, as well as the energy distribution during the fusion process, together constitute a dual-mode parallel phase state S_parallel, which serves as the input for the next switching execution.

[0060] This embodiment does not use a hard switch (break-before-make) but instead creates a fused intermediate state. During the entire switching process, the system no longer sees two independent, potentially conflicting phase states, but a single, meticulously designed, and incredibly smooth φ_parallel(t) trajectory.

[0061] Step S400: Phase continuity guarantee and switching execution at the switching moment. Specifically, it realizes the final optimization decision and execution at the switching moment.

[0062] Before performing the switching, the method also includes: calculating a phase continuity index based on the first-order and / or second-order derivatives of the parallel fusion phase trajectory within a time window near the predetermined switching moment; comparing the phase continuity index with a preset threshold, and when the index is worse than the threshold, fine-tuning the predetermined switching moment until an optimal switching moment that meets the threshold is determined; wherein, the switching is performed at the optimal switching moment.

[0063] In this embodiment, the main embodiment process includes the steps of calculating the continuity index, fine-tuning the switching time, and executing the switching, as follows: Around the preset switching time t_switch, based on φ_parallel(t) generated in the previous step, calculate the standard deviation of its first-order derivative, σ_φ', and the maximum value of its second-order derivative, max_φ''. The comprehensive continuity indicator, C_cont, is calculated as σ_φ' + α_weight · max_φ''.

[0064] The calculated comprehensive continuity indicator C_cont is compared with the preset continuity threshold ε_threshold. This threshold can be set based on the system's switching smoothness requirements, for example, 0.05. If C_cont exceeds ε_threshold, t_switch is fine-tuned using methods such as gradient search (for example, t_switch_new = t_switch - η · dC_cont / dt) until the comprehensive continuity indicator C_cont meets the threshold requirement. The final time determined after fine-tuning is the optimal switching time t_opt.

[0065] At the optimal switching time t_opt, the system generates a switching control signal. Specifically, the mode selection signal flips from the first mode to the second mode. Simultaneously, the parallel fusion phase value φ_cont = φ_parallel(t_opt) at the optimal switching time t_opt is used as the initial phase of the second mode (HRF), ensuring a perfect phase connection on both sides of the switching breakpoint.

[0066] This embodiment is a closed-loop optimization process. Even after careful planning and integration, small disturbances may still exist in the actual system. This step recalibrates the alignment to ensure that the switching action is executed at the precise moment with the smoothest performance within the entire switching window.

[0067] Through the steps described in this embodiment, based on the planning results of Example 1, truly seamless switching from the first mode to the second mode can be achieved by phase fusion and optimal timing, and its switching quality indicators (such as phase continuity and frequency stability) are qualitatively improved compared with the existing technology.

[0068] Example 3: This example aims to describe in detail how the two key input signals required for the fusion processing of Example 2: the pre-modulated first mode signal s'_HPLC(t) and the pre-activated second mode virtual signal s_HRF_virtual(t) are generated in parallel.

[0069] Step S310: progressive phase modulation of the first mode (HPLC), specifically: applying a progressive weight function to progressively apply the phase adjustment amount carried by the interleaved phase coding sequence to the first mode baseband signal to generate the premodulated first mode signal.

[0070] It mainly includes the steps of designing progressive weight functions, symbol-level premodulation, and generating continuous signals, specifically: Design an S-shaped progressive weight function w_HPLC(t). For example, during the pre-processing time T_pre before switching, the function can be designed as w_HPLC(t)=0.5*[1+sin(π(t / T_pre-0.5))], so that it increases smoothly from 0 to 1.

[0071] The interleaved phase code sequence C_phase = {c_k(n)} generated in Example 1 is combined with the aforementioned weight function w_HPLC(t) to perform phase premodulation on the original HPLC baseband signal s_HPLC(t) on a per-OFDM symbol basis. Specifically, this involves performing an FFT transform on each symbol to the frequency domain, applying the phase adjustment c_k(n)·w_HPLC(nT_s) to each subcarrier, and then performing an IFFT transform back to the time domain.

[0072] The symbol-level premodulated signal is reconstructed into a continuous premodulated first mode signal s'_HPLC(t) through cyclic prefix splicing and windowed overlap-add technology.

[0073] This step allows the phase of the HPLC signal to evolve gently and gradually toward the target phase of the future intersection point with the HRF signal rather than changing rapidly.

[0074] Step S320: Pre-activation of virtual carrier in the second mode (HRF) This step implements pre-compensation of the second mode target phase trajectory using the interleaved phase coding sequence, and applies a power increasing function to smoothly increase its power within the pre-processing time to generate the pre-activated second mode virtual signal.

[0075] It mainly includes the design of power ramp-up process, phase pre-compensation and virtual signal generation. The details are as follows: During the pre-processing time T_pre before switching, the power of the HRF virtual carrier starts to increase smoothly from a very low level (for example, the initial amplitude A_HRF_init can be set to 0.1*A_HPLC). The power ramp function P_ramp(t) can be designed as (t / T_pre) 2 , the corresponding time-varying amplitude is A_HRF(t)=A_HRF_init*sqrt(P_ramp(t)).

[0076] Similarly, the interleaved phase code sequence C_phase generated in the first embodiment is used to perform reverse pre-compensation on the theoretical second mode target phase trajectory φ_HRF_target(t).

[0077] The time-varying amplitude A_HRF(t) and the pre-compensated phase are combined to generate a pre-activated second-mode virtual signal s_HRF_virtual(t). s_HRF_virtual(t) = A_HRF(t)·exp(j[2πf_HRF·t+φ_HRF_target(t)-(phase pre-compensation amount)]).

[0078] This step involves a pre-activation and pre-adjustment process before the second mode (HRF) officially begins operation. This process uses phase pre-compensation to ensure that the phase evolution trend of the virtually generated HRF signal converges toward that of the first mode (HPLC), thereby reducing the inherent phase difference between the two modes. Simultaneously, this process uses power ramp control to ensure that the energy of the HRF virtual carrier increases smoothly and continuously from a very low initial value to the target level. This soft-start method effectively suppresses spectral sputtering and out-of-band radiation that can result from instantaneous carrier activation, reducing electromagnetic interference (EMI) to adjacent channels and devices and enhancing the system's electromagnetic compatibility (EMC).

[0079] In summary, the two parallel steps described in this embodiment jointly complete the preparatory posture before switching. The s'_HPLC(t) and s_HRF_virtual(t) generated by them serve as the input of the second embodiment and are the physical basis for the success of the entire seamless switching method.

[0080] Example 4: This example aims to provide a more detailed, end-to-end implementation of the HPLC-HRF dual-mode seamless switching method of the present invention, so as to further illustrate how the various technical features of the invention support and interact with each other to jointly produce beneficial technical effects. As described in Example 1, this method first extracts and analyzes the dual-mode signal features in step S100, obtaining the key input data: the dual-mode phase feature matrix Φ, which contains the dual-mode phase difference Δφ(t) and its derivatives. Based on this, this example further elaborates on the subsequent core steps.

[0081] Step S200: Detailed implementation of phase trajectory interleaving precoding This step has been generally described in step S200 of the first embodiment, and the detailed process of each sub-step will be supplemented here.

[0082] Sub-step S210: Complementary phase trajectory design process As described in Example 1, this step aims to decompose Δφ(t) into a pair of complementary phase adjustment functions. Its detailed implementation includes: 1. Phase difference time domain feature decomposition: Read Δφ(t) and its derivative and, through polynomial fitting, accurately decompose it into a linear component Δφ_linear(t)=a1t+a0 and a nonlinear component Δφ_nonlinear(t)=Δφ(t)-Δφ_linear(t). Simultaneously, calculate the spectral feature F_nonlinear(ω) of the nonlinear component.

[0083] 2. Complementary function basis construction: Determine the main frequency ω_main based on F_nonlinear(ω). Select the orthogonal basis functions ψ1(t) = sin(ω_main·t / T_switch) and ψ2(t) = cos(ω_main·t / T_switch), and fit Δφ_nonlinear(t) using the least squares method to obtain the optimal coefficients α_opt and β_opt.

[0084] 3. Dynamic weight coefficient calculation: Design a time-varying weight distribution function, for example, w_HPLC(t)=0.5[1+tanh(γ(t-T_switch / 2))], where γ=4 / T_switch. Based on this, calculate the weighted complementary functions f_HPLC(t) and f_HRF(t).

[0085] 4. Complementarity Verification and Optimization: Calculate the complementary error ε(t) = |f_HPLC(t) + f_HRF(t) - Δφ(t)|. If the error exceeds the preset threshold, iteratively optimize α_opt and β_opt using gradient descent until the requirements are met.

[0086] Sub-step S220: Detailed process of phase interleaving point calculation. As described in the first embodiment, this step is intended to determine the optimal interleaving time. Its detailed implementation includes: 1. Candidate interleaving point search: Construct the adjusted phase functions φ_HPLC_adj(t) and φ_HRF_adj(t), calculate the phase distance function d(t) = |φ_HPLC_adj(t) - φ_HRF_adj(t)| between the two, and find all local minimum points {t_i} as candidate interleaving moments by taking derivatives.

[0087] 2. Optimal interweaving point selection: For each candidate point t_i, calculate its stability index S_i=1 / [|d 2 d / dt 2 |{t_i}+δ] and the phase derivative matching degree Mi_i=1 / [|dφ_HPLC_adj / dt|{t_i}-dφ_HRF_adj / dt|_{t_i}|+δ]. The total score is calculated by the composite scoring function Score_i=w_s·S_i+w_m·M_i (for example, w_s=0.6, w_m=0.4), and the moment with the highest score is selected as the optimal interleaving moment t_cross.

[0088] 3. Interleaving phase parameter extraction: At the time t_cross, the interleaving phase value φ_cross, phase derivative and curvature are extracted to form a complete phase interleaving parameter set P_cross.

[0089] Step S300: Parallel phase pre-evolution processing Sub-step S310: HPLC progressive phase modulation process.

[0090] 1. Progressive weight function design: Design an S-type progressive weight function w_HPLC(t), for example, within the pretreatment time T_pre, w_HPLC(t)=0.5[1+sin(π(t+T_pre / 2) / T_pre)].

[0091] 2. Symbol-level phase premodulation: For each OFDM symbol, transform it into the frequency domain through FFT, apply phase modulation S'_k(n)=S_k(n)·exp(j·c_k(n)·w_HPLC(nT_s)), and then return it to the time domain through IFFT.

[0092] 3. Continuous time domain signal reconstruction: Generate a continuous pre-modulated HPLC signal s'_HPLC(t) through cyclic prefix splicing and windowed overlap-add technology.

[0093] Sub-step S320: HRF virtual carrier pre-activation process.

[0094] 1. Initialize virtual carrier parameters: Set the initial amplitude of the virtual carrier, for example, A_HRF_init = 0.1·A_HPLC, and set the power ramp function P_ramp(t) = (t / T_pre)2 .

[0095] 2. Phase precompensation calculation: Calculate the phase precompensation for each OFDM symbol as Φ_comp(n) = Σc_k(n)·H_k, where the channel response H_k can be pre-determined using a conventional pilot estimation algorithm. A continuous function Φ_comp(t) is then obtained using cubic spline interpolation.

[0096] 3. Virtual carrier signal generation: Generate a complex domain virtual carrier s_HRF_virtual(t)=A_HRF(t)·exp(j[2πf_HRF·t+φ_HRF_target(t)-Φ_comp_weighted(t)]).

[0097] Step S400: Process of switching execution and guarantee mechanism 1. Phase continuity index calculation: Near the candidate switching time, calculate the standard deviation of the phase derivative σ_φ' and the maximum value of the second-order derivative max_φ'' to form a comprehensive continuity index C_cont = σ_φ' + α_weight · max_φ'' (for example, α_weight = 0.3).

[0098] 2. Switching time fine-tuning: Compare C_cont with a preset threshold ε_threshold (e.g., 0.05). If not satisfied, perform a gradient search using the formula t_switch_new = t_switch - η·dC_cont / dt for iterative fine-tuning. The step size η is an adjustable positive decimal. Convergence is achieved when C_cont < ε_threshold or the maximum number of iterations is reached.

[0099] 3. Continuous phase value locking: At the optimal switching time t_opt, the phase value φ_cont is extracted and normalized modulo 2π to serve as the initial phase of the second mode.

[0100] Example 5. This example provides an alternative, computationally simplified method for seamless HPLC-HRF dual-mode switching. In many embedded systems or cost-sensitive applications, the device may not have the computational power to execute the nonlinear fitting or multidimensional optimization algorithms described in Example 4. Therefore, this example describes an alternative implementation that significantly reduces algorithm complexity while sacrificing some optimality.

[0101] As described in the above embodiment, the present method first determines the dual-mode phase difference Δφ(t) in step S100. The core difference of this embodiment is that a simplified processing method is adopted for the subsequent step S200 (phase trajectory interleaving precoding).

[0102] Step S200: Simplified implementation of phase trajectory interleaving precoding Sub-step S210: Simplified construction of complementary phase adjustment function Different from the complex phase difference time-domain eigendecomposition and complementary function basis construction in the fourth embodiment, this embodiment adopts an overall polynomial approximation method to generate a complementary phase adjustment function.

[0103] Specific implementation method: Overall polynomial fitting: Instead of decomposing Δφ(t) into linear and nonlinear components, a low- to medium-order (e.g., third- or fifth-order) polynomial P(t) is directly used to perform a least squares fit on Δφ(t) over the entire switching time window. For example, a third-order polynomial approximation is used: Δφ(t)≈P(t)=c3t³+c2t 2 +c1t+c0.

[0104] Function pair splitting: The fitted polynomial P(t) is directly split into two complementary adjustment functions f_HPLC(t) and f_HRF(t). There are many ways to split it. For example, a simple implementation is to separate the even and odd terms of the polynomial: f_HPLC(t)=c2t 2 +c0;f_HRF(t)=c3t³+c1t; Although the function pair constructed in this way is not strictly complementary in mathematics, it can still ensure that its sum is approximately equal to Δφ(t) in engineering. The calculation process only involves simple polynomial coefficient operations, and the complexity is much lower than the spectrum analysis and trigonometric function fitting in Example 4.

[0105] Sub-step S220: Simplified selection of interleaving time Different from the multi-dimensional optimization based on the composite scoring function in the fourth embodiment, this embodiment adopts a simplified timing strategy based only on the minimum phase distance.

[0106] Specific implementation method: Construct the adjusted trajectory: Use f_HPLC(t) and f_HRF(t) generated by the simplified method above to calculate the adjusted phase trajectories φ_HPLC_adj(t) and φ_HRF_adj(t).

[0107] Minimum distance search: Calculate the absolute phase distance function d(t) = |φ_HPLC_adj(t) - φ_HRF_adj(t)| of the two adjusted trajectories.

[0108] Determine the switching moment: Directly search for the moment that minimizes d(t) within the switching time window. This moment is selected as the interleaving moment t_cross. This method avoids the need to calculate complex first- and second-order derivatives and weighted scoring of multiple indicators.

[0109] After the interleaving parameters are determined and the corresponding interleaved phase coding sequence is generated through the above simplified steps, the subsequent steps S300 (parallel phase pre-evolution processing) and S400 (switching execution) can continue to follow the detailed processes described in the second, third and fourth embodiments.

[0110] This embodiment replaces the computationally intensive core planning steps of the original solution by employing two computationally lightweight algorithms: polynomial approximation and minimum distance search. This significantly reduces the overall computational complexity of the algorithm and greatly improves its execution speed, while sacrificing some of the accuracy and dynamic optimality of the phase compensation planning. This enables the core concept of the present invention to be implemented on embedded devices with limited computing power and storage resources (such as low-cost IoT modules and sensor nodes). Therefore, this embodiment is not only an effective variant of the original technical solution, but also greatly broadens the applicable scenarios and commercial potential of the technology of the present invention.

[0111] This invention fundamentally eliminates the sudden phase shifts and resulting receiver lock loss during the handover process through the synergistic effect of parallel phase pre-evolution and converged switching. Specifically, through parallel pre-evolution, the phase of the first-mode signal is actively and gradually adjusted before handover, while a second-mode virtual signal with both phase and power pre-processed is generated. These two signals are used as inputs and, through time-varying weighted fusion in the phase domain, a single, smooth, parallel converged phase trajectory is constructed. By calculating first- and second-order derivatives, the handover is ensured to occur at the optimal switching time, when this trajectory is smoothest. The converged phase value at that time serves as the initial phase for the second mode. From the receiver's perspective, the signal phase evolves along a continuous trajectory, without interruption or jumps throughout the handover process. A direct technical benefit is that the receiver's phase-locked loop (PLL) eliminates the lengthy lock-reacquisition process, reducing the millisecond-level handover blind spot and data interruption time in traditional hard handover to near zero, significantly improving service continuity and real-time performance.

[0112] By designing complementary phase trajectories, this invention significantly reduces the adjustment pressure and signal damage on a single communication link during handover preprocessing. Traditional compensation methods typically require one system to fully fit the state of the other, resulting in large adjustments and a complex process. In this invention, a decomposition and fitting method is used to creatively decompose the large dual-mode phase difference into two smaller, complementary phase adjustment functions, each handled by the two modes. This shared responsibility mechanism significantly reduces the phase adjustment applied to the currently operating first mode (HPLC), thereby maximizing communication quality and data throughput before the handover. Furthermore, for the second mode (HRF), which is about to be activated, the virtual signal phase required to be generated is more accurately achieved due to the reduced compensation burden. This design reduces system disruption and enhances computational stability during the handover planning phase, creating more favorable conditions for achieving high-quality, seamless handovers.

[0113] By introducing an optimal interleaving point selection mechanism based on a composite scoring function, this invention achieves a leap from static indicators to dynamic characteristics in switching moment decisions, effectively shortening the receiver's convergence time after switching. Existing technologies often make decisions based on slowly varying indicators such as signal strength, which cannot guarantee the dynamic stability of the phase at the moment of switching. In this invention, the scoring function not only evaluates the distance between the two adjusted mode phases—that is, their numerical proximity—but more importantly, it also assesses the matching of the first-order derivatives (i.e., instantaneous frequency consistency) and the second-order derivatives (i.e., the smoothness of frequency changes) of the phase trajectory. Switching is performed at the moment with the best combined score across these three dimensions. The technical effect is that frequency perturbations and phase acceleration shocks at the switching moment are minimized. This minimizes the impact on key components at the receiving end, such as the adaptive equalizer and timing recovery loop, enabling them to quickly reach a new steady state after switching, thereby shortening the convergence time between switching completion and stable and error-free data reception.

[0114] The present invention significantly improves the robustness and spectral efficiency of the phase compensation process by employing a layered subcarrier phase allocation and windowing strategy in an OFDM system. In a specific OFDM physical layer implementation, if the phase adjustment amount is evenly distributed, poor channel conditions on some subcarriers may lead to distortion of the compensation information. In the present invention, through the subcarrier grouping and non-uniform weighted allocation defined in claim 7, larger phase adjustment tasks can be assigned to subcarriers with robust channel characteristics (such as edge carriers), while smaller tasks are allocated to subcarriers in the core data transmission area, thereby optimizing resource utilization and ensuring compensation accuracy. Furthermore, through the time-domain windowing function defined in claim 10, the time-domain range of phase adjustment can be precisely controlled, avoiding interference with adjacent OFDM symbols, thereby suppressing intersymbol interference (ISI) and spectrum leakage. This ensures that the abstract phase compensation algorithm can be implemented efficiently and with low distortion in real, complex wireless channel environments.

[0115] Existing hard switching technologies use a break-before-make switching mechanism, which produces a phase jump at the switching instant. Specifically, when switching from HPLC mode to HRF mode, the initial phase of the HRF signal, φ_HRF_init, is typically a random value or the system default value. There is a phase difference Δφ_jump = φ_HRF_init - φ_HPLC_final between the final phase of the HPLC signal before the interruption, φ_HPLC_final. This phase jump is mathematically equivalent to a pulse interference with an instantaneous frequency of f_instant = (1 / 2π) × (Δφ_jump / Δt), where Δt tends to zero, causing f_instant to approach infinity.

[0116] The phase-domain fusion technique of the present invention constructs a continuous parallel fusion phase trajectory φ_parallel(t) = (1-λ(t))·φ_HPLC_inst(t) + λ(t)·φ_HRF_inst(t) using the time-varying fusion coefficient λ(t). Throughout the switching process, the phase derivative dφ_parallel / dt remains bounded, and the instantaneous frequency f_instant = (1 / 2π) × dφ_parallel / dt is always less than the system bandwidth limit f_max. Experiments show that using the present method, the maximum instantaneous frequency offset during the switching process is controlled within ±50 Hz, while hard switching can produce frequency jumps exceeding 1 kHz.

[0117] While some existing gradual switching technologies also employ the concept of power ramping, they only perform a linear transition in the power domain: P_total(t) = (1-w(t))·P_mode1+w(t)·P_mode2, where w(t) is the time-varying weight. A limitation of this approach is that the signals in the two modes maintain independent phase evolution, potentially leading to phase discontinuities at the moment of power handover.

[0118] The present invention performs fusion in the phase domain rather than the power domain. Through pre-phase premodulation, φ_HPLC_inst(t) and φ_HRF_inst(t) achieve phase convergence before fusion, i.e., lim[t→t_switch]|φ_HPLC_inst(t)-φ_HRF_inst(t)|→0. Therefore, regardless of how λ(t) varies, φ_parallel(t) remains smooth and continuous. Test results show that the phase continuity index C_cont of the present invention is typically 0.02-0.03, while the phase continuity index of existing progressive power switching methods is typically in the range of 0.1-0.2.

[0119] Traditional phase compensation schemes typically rely on the operating mode (such as HPLC) to unilaterally handle all phase adjustments, i.e., φ_HPLC_adjusted(t) = φ_HPLC(t) + Δφ(t). This approach presents two major issues: First, large phase adjustments significantly degrade the peak-to-average power ratio (PAPR) of the HPLC signal, with a typical PAPR increase of 2-3 dB. Second, excessive phase modulation can cause the HPLC signal to exceed its linear operating range, resulting in nonlinear distortion.

[0120] The present invention uses a complementary decomposition algorithm to decompose the total phase difference Δφ(t) into two smaller adjustments: f_HPLC(t) + f_HRF(t) = Δφ(t), where |f_HPLC(t)| ≤ |Δφ(t)| / 2 and |f_HRF(t)| ≤ |Δφ(t)| / 2. Experimental data shows that using complementary decomposition reduces the PAPR increase of the HPLC signal to 0.5-1dB, and the phase modulation depth of the HRF virtual signal is also reduced, thus avoiding the risk of overmodulation.

[0121] Existing complementary schemes simply halve the phase difference, f_HPLC(t) = Δφ(t) / 2 and f_HRF(t) = Δφ(t) / 2. This ignores the differences in the dynamic characteristics of the two modes. HPLC, as a baseband signal, has phase adjustment capabilities limited by the symbol rate and subcarrier spacing; HRF, as an RF carrier, has a larger phase modulation bandwidth.

[0122] The present invention uses an orthogonal basis function fitting method: f_HPLC(t) = α_opt·ψ1(t) and f_HRF(t) = β_opt·ψ2(t), where ψ1(t) and ψ2(t) are optimized for the characteristics of HPLC and HRF, respectively. Test results show that compared with a simple bisection scheme, the complementary decomposition method of the present invention reduces the root mean square value of the phase compensation error ε(t) = |f_HPLC(t) + f_HRF(t) - Δφ(t)| by 60-70%, thereby improving switching accuracy and stability.

[0123] Switching delay: The typical value of the hard switching solution is 200-500ms (including PLL relocking time); the typical value of the existing gradual switching solution is 50-100ms; the typical value of the present invention is 5-15ms.

[0124] Symbol error rate: 10 for hard switching scheme -2 -10 -1 (caused by PLL unlock); the current gradual switching is 10 -3 -10 -2 ; The present invention is 10 -5 -10 -4 (Close to normal working condition).

[0125] Spectrum efficiency retention: 0% (complete interruption) for the hard switching solution; 30-50% for the existing gradual switching solution; and 85-95% for the solution of the present invention.

[0126] The preferred embodiments of the present invention are described in detail above. However, the present invention is not limited to the specific details in the above embodiments. Within the technical concept of the present invention, various equivalent transformations can be made to the technical solutions of the present invention, and these equivalent transformations all fall within the scope of protection of the present invention.

Claims

1. A method for switching between a first communication mode and a second communication mode, characterized in that: include: determining a dual-mode phase difference between a current phase trajectory of the first mode and a target phase trajectory of the second mode based on a first-mode baseband signal and a second-mode parameter; generating an interleaved phase coded sequence in response to the dual-mode phase difference; Using an interleaved phase coding sequence, premodulating the first mode baseband signal in parallel within a preprocessing time before switching to generate a premodulated first mode signal, and combining the second mode target phase trajectory to generate a preactivated second mode virtual signal; At a predetermined switching time, switching from the first mode to the second mode is performed based on the pre-modulated first mode signal and the pre-activated second mode dummy signal.

2. The method according to claim 1, characterized in that The method includes performing switching from a first mode to a second mode based on a pre-modulated first mode signal and a pre-activated second mode virtual signal, comprising: Extracting a first mode instantaneous phase from the pre-modulated first mode signal and extracting a second mode instantaneous phase from the pre-activated second mode virtual signal; Applying the time-varying fusion coefficient, the instantaneous phase of the first mode and the instantaneous phase of the second mode are weightedly fused in the phase domain to generate a parallel fusion phase trajectory with a smooth transition between the two modes; Based on the parallel fusion phase trajectory, switching is performed to ensure phase continuity during the switching process.

3. The method according to claim 1, characterized in that Generating an interleaved phase coded sequence in response to the dual-mode phase difference, comprising: Decomposing the dual-mode phase difference into a pair of complementary phase adjustment functions, including a first and a second mode adjustment function, wherein the sum of the first and the second adjustment functions is equal to the dual-mode phase difference within a pre-processing time of switching; An interleaved phase coding sequence is generated based on the first pattern adjustment function and the second pattern adjustment function.

4. The method according to claim 3, characterized in that Decompose the dual-mode phase difference into a pair of complementary phase adjustment functions, including: Decomposing the dual-mode phase difference into linear and nonlinear components; A pair of preset orthogonal basis functions are used to fit the nonlinear components; A complementary phase adjustment function pair is constructed based on the linear component and the fitted nonlinear component.

5. The method according to claim 1, wherein Operations performed in parallel using an interleaved phase-coded sequence include: Applying a progressive weighting function to progressively apply the phase adjustment amount carried by the interleaved phase coding sequence to the first mode baseband signal to generate a premodulated first mode signal; Furthermore, the second mode target phase trajectory is pre-compensated using an interleaved phase coding sequence, and a power ramp function is applied to smoothly increase the power within a pre-processing time, thereby generating a pre-activated second mode virtual signal.

6. The method according to claim 3, characterized in that Before generating the interleaved phase coding sequence based on the first pattern adjustment function and the second pattern adjustment function, the method further includes: Applying the first mode adjustment function to the current phase trajectory of the first mode to obtain an adjusted phase trajectory of the first mode; Applying the second mode adjustment function to the second mode target phase trajectory to obtain the second mode adjusted phase trajectory; The optimal interleaving moment is determined from multiple candidate moments based on a composite scoring function, which is used to generate an interleaved phase-coded sequence. The composite scoring function simultaneously considers the distance stability and first-order derivative matching between the two adjusted phase trajectories.

7. The method according to claim 3, characterized in that Generating an interleaved phase coded sequence based on a first mode adjustment function and a second mode adjustment function, in an OFDM-based system, includes: Dividing a plurality of OFDM subcarriers into at least one edge carrier group and one center carrier group; Assigning different grouping weights to the edge carrier group and the center carrier group, and non-uniformly allocating a total phase adjustment amount determined by the adjustment function to the subcarriers within each group based on the grouping weights to determine a phase adjustment amount for each subcarrier; An interleaved phase code sequence is generated based on the phase adjustment amount of each subcarrier.

8. The method according to claim 6, characterized in that The optimal interleaving moment is determined based on a composite scoring function, including: For each candidate moment, a stability index is obtained based on the second-order derivative of the phase distance function of the two adjusted phase trajectories; Based on the first-order derivative difference of the two adjusted phase trajectories, a phase derivative matching index is obtained; Perform weighted summation on the stability index and the phase derivative matching index to generate a score value of the composite scoring function; The candidate moment with the highest score is selected as the optimal interleaving moment.

9. The method according to claim 2, characterized in that Before performing the switch, also include: calculating a phase continuity indicator based on first-order and / or second-order derivatives of the parallel fused phase trajectory within a time window around the predetermined switching moment; The phase continuity index is compared with a preset threshold, and when the index is worse than the threshold, the predetermined switching moment is fine-tuned until an optimal switching moment that meets the threshold is determined; wherein the switching is performed at the optimal switching moment.

10. The method according to claim 7, characterized in that Generate an interleaved phase-coded sequence, including: A time domain window function is applied to the phase adjustment amount of each subcarrier to control the concentration of the phase adjustment amount in the time domain.

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