Variable resistor power switch feedback

Abstract

The present disclosure provides "variable resistor power switch feedback". A vehicle powertrain system has a power inverter, the power inverter includes a load switch with a main emitter and a current mirror emitter, a variable resistor coupled between the current mirror emitter and the main emitter, and a controller. The controller can be configured to adjust a gate voltage based on a voltage across the variable resistor, and in response to the gate voltage exceeding a Miller plateau gate voltage, increase a variable resistor resistance so that feedback increases as the load switch saturates.

Description

Technical Field

[0001] The present application generally relates to a gate driver for a solid-state switch of a vehicle power module, wherein a variable resistor is coupled between a mirror switch emitter and a load switch emitter. Background Art

[0002] Electrified vehicles, including hybrid-electric vehicles (HEV), plug-in hybrid electric vehicles (PHEV) and battery electric vehicles (BEV), rely on traction batteries to provide power to traction motors for propulsion, and rely on power inverters in between to convert direct current (DC) power to alternating current (AC) power. A typical AC traction motor is a 3-phase motor that can be powered by 3 sinusoidal signals, each driven with 120 degrees of phase separation. The traction battery is configured to operate within a specific voltage range and provide maximum current. During vehicle operation, high-voltage components in the electric vehicle powertrain may be subjected to stress including overcurrent or short-circuit current conditions.

[0003] In addition, many electrified vehicles include a direct current-to-direct current (DC-DC) converter (also known as a variable voltage converter (VVC)) for converting the voltage of the traction battery to the operating voltage level of the electric machine. The electric machine, which may include a traction motor, may require high voltage and high current. Due to the voltage, current and switching requirements, solid-state switches such as insulated gate bipolar junction transistors (IGBTs) are often used to generate signals in power inverters and VVCs. Similarly, the components of the DC-DC converter may be subject to stresses including overcurrent or short circuit current conditions. Summary of the invention

[0004] A vehicle powertrain system has a power inverter including a load switch with a main emitter and a current mirror emitter, a variable resistor coupled between the current mirror emitter and the main emitter, and a controller. The controller can be configured to adjust a gate voltage based on a voltage across the variable resistor and, in response to the gate voltage exceeding a Miller plateau gate voltage, increase a resistance of the variable resistor such that feedback increases as the load switch saturates.

[0005] A vehicle powertrain system includes a load switch monolithically integrated with a mirror switch, a variable resistor coupled between an emitter of the mirror switch and a reference point, and a controller. The controller can be configured to adjust a gate voltage based on a voltage across the variable resistor, and in response to the gate voltage exceeding a Miller plateau gate voltage, increase a resistance of the variable resistor such that feedback increases as the load switch saturates.

[0006] A method of controlling a power switch in a vehicle powertrain includes: setting a resistance of a variable resistor; applying a charge to the gate to increase the gate voltage; and changing the resistance of the variable resistor in response to the gate voltage exceeding a Miller plateau gate voltage. The variable resistor may be coupled between a current mirror emitter and a reference point while the gate voltage of the power switch is below a turn-on threshold. BRIEF DESCRIPTION OF THE DRAWINGS

[0007] Figure 1 is a schematic diagram of a hybrid vehicle showing a typical powertrain and energy storage components with a variable voltage converter and a power inverter therebetween.

[0008] Figure 2 is a schematic diagram of a vehicle's variable voltage converter.

[0009] Figure 3 is a schematic diagram of a vehicle motor inverter.

[0010] Figure 4 is a schematic diagram of a gate driver for a power switch with a current mirror and fixed resistor feedback circuit.

[0011] Figure 5 Is has Figure 4 Plot of the characteristics of a current mirror power switch versus time.

[0012] Figure 6 is a schematic diagram of a gate driver for a power switch having a current mirror and a variable resistor feedback circuit.

[0013] Figure 7 Is has Figure 6 Plot of the characteristics of a current mirror power switch versus time.

[0014] Figure 8 is a schematic diagram of a gate driver for a power switch with a current mirror and optional resistive feedback circuit.

[0015] Fig. 9 Is has Figure 6 Plot of the characteristics of a current mirror power switch versus time.

[0016] Fig.10is a schematic diagram of a gate driver for a power switch with a current mirror and optional resistive feedback circuit. DETAILED DESCRIPTION

[0017] Embodiments of the present disclosure are described herein. However, it should be understood that the disclosed embodiments are merely examples and that other embodiments may take various forms and alternative forms. The drawings are not necessarily drawn to scale; some features may be enlarged or minimized in order to show the details of specific components. Therefore, the specific structural details and functional details disclosed herein should not be interpreted as being restrictive, but merely as a representative basis for teaching those skilled in the art to adopt the present invention in different ways. As will be understood by those of ordinary skill in the art, the various features shown and described with reference to any of the figures may be combined with the features shown in one or more other figures to produce embodiments that are not explicitly shown or described. The combination of features shown provides representative embodiments for typical applications. However, various combinations and modifications of the features consistent with the teachings of the present disclosure may be desired for specific applications or implementations.

[0018] Generally speaking, solid-state devices (SSDs) such as insulated gate bipolar junction transistors (IGBTs), metal oxide semiconductor field effect transistors (MOSFETs), or bipolar junction transistors (BJTs) are widely used in a variety of automotive and industrial applications such as electric motor drives, power inverters, direct current-to-direct current (DC-DC) converters, and power modules. The operation of IGBTs and MOSFETs is voltage controlled, where the operation is based on the voltage applied to the gate of the IGBT or MOSFET, while the operation of BJTs is current controlled, where the operation is based on the current applied to the base of the BJT. Here, the use of IGBTs will be discussed, but the structure and method can be applied to other SSDs, for example, insulated gate SSDs including IGBTs and MOSFETs. The operation of the IGBT is controlled by a gate voltage provided by a gate driver.

[0019] Here, an IGBT current mirror feedback system and method is disclosed, which can be implemented for use during the turn-on of a power device. The main principle is that a variable resistance resistor is coupled to the current mirror sense pin of the IGBT to adjust the feedback when the device is turned on.

[0020] Figure 1Depict an electrified vehicle 112 that may be referred to as a plug-in hybrid electric vehicle (PHEV). The plug-in hybrid electric vehicle 112 may include one or more motors 114 mechanically coupled to a hybrid transmission 116. The motor 114 may be capable of operating as a motor or a generator. In addition, the hybrid transmission 116 is mechanically coupled to an engine 118. The hybrid transmission 116 is also mechanically coupled to a drive shaft 120, which is mechanically coupled to wheels 122. When the engine 118 is turned on or off, the motor 114 can provide propulsion and deceleration performance. The motor 114 can also be used as a generator and can provide fuel economy benefits by recovering energy that would normally be lost as heat in a friction braking system. The motor 114 can also reduce vehicle emissions by allowing the engine 118 to operate at a more efficient speed and allowing the hybrid electric vehicle 112 to operate in an electric mode when the engine 118 is turned off in certain circumstances. The electrified vehicle 112 may also be a battery electric vehicle (BEV). In a BEV configuration, the engine 118 may not be present. In other configurations, the electrified vehicle 112 may be a full hybrid-electric vehicle (FHEV) without plug-in capability.

[0021] The traction battery or battery pack 124 stores energy that can be used by the motor 114. The vehicle battery pack 124 can provide a high voltage direct current (DC) output. The traction battery 124 can be electrically coupled to one or more power electronics modules 126. One or more contactors 142 can isolate the traction battery 124 from other components when open, and connect the traction battery 124 to other components when closed. The power electronics module 126 is also electrically coupled to the motor 114 and provides the ability to transfer energy bidirectionally between the traction battery 124 and the motor 114. For example, the traction battery 124 can provide a DC voltage, and the motor 114 can operate with three-phase alternating current (AC) to function. The power electronics module 126 can convert the DC voltage into a three-phase AC current to operate the motor 114. In the regenerative mode, the power electronics module 126 can convert the three-phase AC current from the motor 114 acting as a generator into a DC voltage compatible with the traction battery 124.

[0022] The vehicle 112 may include a variable-voltage converter (VVC) 152 electrically coupled between the traction battery 124 and the power electronics module 126. The VVC 152 may be a DC / DC boost converter configured to increase or boost the voltage provided by the traction battery 124. By increasing the voltage, the current demand may be reduced, resulting in a reduced wiring size for the power electronics module 126 and the motor 114. Additionally, the motor 114 may operate with better efficiency and lower losses.

[0023] In addition to providing energy for propulsion, the traction battery 124 can also provide energy for other vehicle electrical systems. The vehicle 112 may include a DC / DC converter module 128 that converts the high-voltage DC output of the traction battery 124 into a low-voltage DC power source compatible with the low-voltage vehicle load. The output of the DC / DC converter module 128 can be electrically coupled to an auxiliary battery 130 (e.g., a 12V battery) to charge the auxiliary battery 130. The low-voltage system can be electrically coupled to the auxiliary battery 130. One or more electrical loads 146 can be coupled to the high-voltage bus. The electrical load 146 may have an associated controller that operates and controls the electrical load 146 when appropriate. Examples of the electrical load 146 may be a fan, an electric heating element, and / or an air conditioning compressor.

[0024] The electrified vehicle 112 may be configured to recharge the traction battery 124 via an external power source 136. The external power source 136 may be a connection to an electrical outlet. The external power source 136 may be electrically coupled to a charger or electric vehicle supply equipment (EVSE) 138. The external power source 136 may be an electricity distribution network or transmission grid provided by an electric utility company. The EVSE 138 may provide circuitry and control to regulate and manage energy transfer between the power source 136 and the vehicle 112. The external power source 136 may provide DC or AC power to the EVSE 138. The EVSE 138 may have a charging connector 140 for plugging into a charging port 134 of the vehicle 112. The charging port 134 may be any type of port configured to transfer power from the EVSE 138 to the vehicle 112. The charging port 134 may be electrically coupled to a charger or an onboard power conversion module 132. The power conversion module 132 may condition the power provided from the EVSE 138 to provide appropriate voltage levels and current levels to the traction battery 124. The power conversion module 132 may interact with the EVSE 138 to coordinate the delivery of power to the vehicle 112. The EVSE connector 140 may have pins that mate with corresponding recesses of the charging port 134. Alternatively, the various components described as electrically coupled or connected may transfer power using wireless inductive coupling.

[0025] One or more wheel brakes 144 may be provided for decelerating the vehicle 112 and preventing the vehicle 112 from moving. The wheel brakes 144 may be hydraulically actuated, electrically actuated, or some combination thereof. The wheel brakes 144 may be part of a braking system 150. The braking system 150 may include other components for operating the wheel brakes 144. For simplicity, the accompanying drawings depict a single connection between the braking system 150 and one of the wheel brakes 144. Connections between the braking system 150 and other wheel brakes 144 are implied. The braking system 150 may include a controller for monitoring and coordinating the braking system 150. The braking system 150 may monitor the braking components and control the wheel brakes 144 for vehicle deceleration. The braking system 150 may respond to driver commands and may also operate autonomously to implement features such as stability control. The controller of the braking system 150 may implement a method for applying the requested braking force when another controller or sub-function requests it.

[0026] The electronic modules in the vehicle 112 can communicate via one or more vehicle networks. The vehicle network may include multiple channels for communication. One channel of the vehicle network may be a serial bus, such as a Controller Area Network (CAN). One channel of the vehicle network may include an Ethernet network defined by the Institute of Electrical and Electronics Engineer (IEEE) 802 series of standards. Additional channels of the vehicle network may include discrete connections between modules and may include power signals from the auxiliary battery 130. Different signals may be transmitted on different channels of the vehicle network. For example, a video signal may be transmitted on a high-speed channel (e.g., Ethernet), while a control signal may be transmitted on the CAN or via discrete signals. The vehicle network may include any hardware components and software components that facilitate the transmission of signals and data between modules. The vehicle network is not Figure 1 1, but may imply that the vehicle network may be connected to any electronic module present in the vehicle 112. A vehicle system controller (VSC) 148 may be present to coordinate the operation of the various components.

[0027] Figure 2A diagram depicting a VVC 152 configured as a boost converter. The VVC 152 may include input terminals that may be coupled to terminals of a traction battery 124 via contactors 142. The VVC 152 may include output terminals coupled to terminals of a power electronics module 126. The VVC 152 may be operated to cause a voltage at the output terminals to be greater than a voltage at the input terminals. The vehicle 112 may include a VVC controller 200 that monitors and controls electrical parameters (e.g., voltage and current) at various locations within the VVC 152. In some configurations, the VVC controller 200 may be included as part of the VVC 152. The VVC controller 200 may determine a reference output voltage The VVC controller 200 can be based on the electrical parameters and the reference voltage To determine a control signal sufficient to cause the VVC 152 to achieve the desired output voltage. In some configurations, the control signal may be implemented as a pulse-width modulated (PWM) signal, wherein the duty cycle of the PWM signal varies. The control signal may operate at a predetermined switching frequency. The VVC controller 200 may use the control signal to command the VVC 152 to provide the desired output voltage. The specific control signal for operating the VVC 152 may be positively correlated with the amount of boost to be provided by the VVC 152.

[0028] The output voltage of the VVC 152 can be controlled to achieve a desired reference voltage. In some configurations, the VVC 152 can be a boost converter. In a boost converter configuration where the VVC controller 200 controls the duty cycle, the input voltage With output voltage and duty cycle The ideal relationship between can be described using the following equation:

[0029] 1)

[0030] The desired duty cycle D can be determined by measuring the input voltage (e.g., traction battery voltage) and setting the output voltage to a reference voltage. The VVC 152 can be a buck converter that steps down the voltage from input to output. In a buck configuration, different expressions relating the input voltage and output voltage to the duty cycle can be derived. In some configurations, the VVC 152 can be a buck-boost converter that can increase or decrease the input voltage. The control strategies described herein are not limited to a specific variable voltage converter topology.

[0031] refer to Figure 2, the VVC 152 can increase or "step up" the voltage potential of the power provided by the traction battery 124. The traction battery 124 can provide high voltage (HV) DC power. In some configurations, the traction battery 124 can provide a voltage between 150 volts and 400 volts. The contactor 142 can be electrically coupled in series between the traction battery 124 and the VVC 152. When the contactor 142 is closed, the HV DC power can be transferred from the traction battery 124 to the VVC 152. The input capacitor 202 can be electrically coupled in parallel with the traction battery 124. The input capacitor 202 can stabilize the bus voltage and reduce any voltage and current ripple. The VVC 152 can receive the HV DC power and increase or "step up" the voltage potential of the input voltage according to the duty cycle.

[0032] The output capacitor 204 may be electrically coupled between the output terminals of the VVC 152. The output capacitor 204 may stabilize the bus voltage and reduce voltage and current ripple at the output of the VVC 152.

[0033] Further references Figure 2 , the VVC 152 may include a first switching device 206 and a second switching device 208 for stepping up an input voltage to provide a stepped-up output voltage. The switching devices 206, 208 may be configured to selectively cause current to flow to an electrical load (e.g., the power electronics module 126 and the motor 114). Each switching device 206, 208 may be individually controlled by a gate drive circuit (not shown) of the VVC controller 200 and may include any type of controllable switch (e.g., an insulated gate bipolar transistor (IGBT) or a field-effect transistor (FET)). The gate drive circuit may provide an electrical signal to each of the switching devices 206, 208 based on a control signal (e.g., a duty cycle of a PWM control signal). A diode may be coupled to each of the switching devices 206, 208. The switching devices 206, 208 may each have associated switching losses. Switching losses are those power losses that occur during state changes (e.g., on / off and off / on transitions) of the switching device. Switching losses can be quantified by the current flowing through the switching devices 206, 208 and the voltage across the switching devices 206, 208 during a transition. A switching device may also have associated conduction losses that occur when the device is turned on.

[0034] The vehicle system may include sensors for measuring electrical parameters of the VVC 152. The first voltage sensor 210 may be configured to measure an input voltage (eg, the voltage of the battery 124) and convert a corresponding input signal ( ) is provided to the VVC controller 200. In one or more embodiments, the first voltage sensor 210 can measure the voltage on the input capacitor 202, which corresponds to the battery voltage. The second voltage sensor 212 can measure the output voltage of the VVC 152 and convert the corresponding input signal ( ) is provided to the VVC controller 200. In one or more embodiments, the second voltage sensor 212 can measure the voltage on the output capacitor 204, which corresponds to the DC bus voltage. The first voltage sensor 210 and the second voltage sensor 212 may include circuitry for scaling the voltage to a level suitable for the VVC controller 200. The VVC controller 200 may include circuitry for filtering and digitizing the signals from the first voltage sensor 210 and the second voltage sensor 212.

[0035] An input inductor 214, often referred to as a boost inductor, may be electrically coupled in series between the traction battery 124 and the switch devices 206, 208. The input inductor 214 may alternate between storing and releasing energy in the VVC 152 to enable a variable voltage and current to be provided as the VVC 152 output and to enable a desired voltage boost. A current sensor 216 may measure the input current through the input inductor 214 and generate a corresponding current signal ( ) is provided to the VVC controller 200. The input current through the input inductor 214 may be a voltage difference between the input voltage and the output voltage of the VVC 152, the conduction time of the switching devices 206, 208, and the inductance of the input inductor 214. The VVC controller 200 may include circuitry for scaling, filtering, and digitizing the signal from the current sensor 216 .

[0036] The VVC controller 200 may be programmed to control the output voltage of the VVC 152. The VVC controller 200 may receive input from the VVC 152 and other controllers via the vehicle network and determine control signals. The VVC controller 200 may monitor the input signals To determine the control signal. For example, the VVC controller 200 may provide a control signal corresponding to the duty cycle command to the gate drive circuit. The gate drive circuit may then control each switching device 206, 208 based on the duty cycle command.

[0037] The control signal to the VVC 152 can be configured to drive the switching devices 206, 208 at a specific switching frequency. Within each cycle of the switching frequency, the switching devices 206, 208 can operate at a specified duty cycle. The duty cycle defines the amount of time that the switching devices 206, 208 are in the on state and the off state. For example, a duty cycle of 100% can cause the switching devices 206, 208 to operate in a continuous on state without being turned off. A duty cycle of 0% can cause the switching devices 206, 208 to operate in a continuous off state without being turned on. A duty cycle of 50% can cause the switching devices 206, 208 to operate in the on state for half a cycle and in the off state for half a cycle. The control signals of the two switches 206, 208 can be complementary. That is, the control signal sent to one of the switching devices (e.g., 206) can be the opposite version of the control signal sent to the other switching device (e.g., 208). Complementary control of switch devices 206, 208 is desirable to avoid a shoot-through condition where current flows directly through high-side switch device 206 and low-side switch device 208. High-side switch device 206 is also referred to as pass device 206 and low-side switch device 208 is also referred to as charge device 208.

[0038] The current controlled by the switching devices 206, 208 may include a ripple component having a certain amplitude, which varies with the amplitude of the current and the duty cycle and switching frequency of the switching devices 206, 208. The worst case ripple current amplitude occurs during relatively high input current conditions relative to the input current. When the duty cycle is fixed, an increase in the inductor current causes the amplitude of the ripple current to increase. The amplitude of the ripple current is also related to the duty cycle. The highest amplitude ripple current occurs when the duty cycle is equal to 50%. The general relationship between the inductor ripple current amplitude and the duty cycle can be as shown Figure 5 Based on these facts, it may be beneficial to implement measures to reduce the ripple current amplitude under high current and intermediate duty cycle conditions.

[0039] When designing the VVC 152, the switching frequency and inductance value of the inductor 214 may be selected to meet the maximum allowable ripple current amplitude. The ripple component may be a periodic change that appears on a DC signal. The ripple component may be defined by a ripple component amplitude and a ripple component frequency. The ripple component may have harmonics in the audible frequency range, which may add to the noise signature of the vehicle. In addition, the ripple component may cause difficulties in accurately controlling the devices fed by the power supply. During switching transients, the switching devices 206, 208 may be turned off at the maximum inductor current (DC current plus ripple current), which may result in a large voltage surge on the switching devices 206, 208. Due to size and cost constraints, the inductance value may be selected based on the conduction current. Typically, as the current increases, the inductance may decrease due to saturation.

[0040] The switching frequency may be selected to limit the magnitude of the ripple current component in a worst-case scenario (e.g., a condition where the highest input current and / or duty cycle is close to 50%). The switching frequency of the switching devices 206, 208 may be selected to be a frequency (e.g., 10 kHz) greater than the switching frequency (e.g., 5 kHz) of the motor / generator inverter coupled to the output of the VVC 152. In some applications, the switching frequency of the VVC 152 may be selected to be a predetermined fixed frequency. The predetermined fixed frequency is typically selected to meet noise and ripple current specifications. However, the selection of a predetermined fixed frequency may not provide optimal performance over all operating ranges of the VVC 152. The predetermined fixed frequency may provide optimal results under a specific set of operating conditions, but may be a compromise result under other operating conditions.

[0041] Increasing the switching frequency can reduce the ripple current amplitude and reduce the voltage stress on the switching devices 206, 208, but may result in higher switching losses. Although the switching frequency can be selected for the worst-case ripple condition, the VVC 152 can only operate under the worst-case ripple condition for a small proportion of the total operating time. This may result in unnecessarily high switching losses, which may reduce fuel economy. In addition, a fixed switching frequency can concentrate the noise spectrum into a very narrow range. The increased noise density in this narrow range may cause significant noise, vibration and harshness (NVH) issues.

[0042] The VVC controller 200 may be programmed to vary the switching frequency of the switching devices 206, 208 based on the duty cycle and input current. Variation of the switching frequency may improve fuel economy by reducing switching losses and reducing NVH issues while maintaining a ripple current target under worst case operating conditions.

[0043] During relatively high current conditions, the switching devices 206, 208 may experience increased voltage stress. At the maximum operating current of the VVC 152, it may be desirable to select a relatively high switching frequency that reduces the ripple component amplitude at a reasonable level of switching losses. The switching frequency may be selected based on the input current amplitude so that as the input current amplitude increases, the switching frequency increases. The switching frequency may be increased up to a predetermined maximum switching frequency. The predetermined maximum switching frequency may be a level that provides a compromise between a lower ripple component amplitude and higher switching losses. The switching frequency may be changed in discrete steps or in a continuous manner within the operating current range.

[0044] The VVC controller 200 may be programmed to reduce the switching frequency in response to the current input being less than a predetermined maximum current. The predetermined maximum current may be the maximum operating current of the VVC 152. The change in switching frequency may be based on the magnitude of the current input to the switching devices 206, 208. When the current is greater than the predetermined maximum current, the switching frequency may be set to a predetermined maximum switching frequency. As the current decreases, the magnitude of the ripple component decreases. By operating at a lower switching frequency when the current decreases, switching losses are reduced. The switching frequency may be changed based on the power input to the switching device. Since the input power is a function of the input current and the battery voltage, the input power and the input current may be used in a similar manner.

[0045] Since the ripple current is also affected by the duty cycle, the switching frequency can be changed based on the duty cycle. The duty cycle can be determined based on the ratio of the input voltage to the output voltage. In this way, the switching frequency can also be changed based on the ratio between the input voltage and the output voltage. When the duty cycle is close to 50%, the predicted ripple current amplitude is a maximum value, and the switching frequency can be set to a predetermined maximum frequency. The predetermined maximum frequency can be the maximum switching frequency value selected to minimize the ripple current amplitude. Within the duty cycle range, the switching frequency can be changed in discrete steps or in a continuous manner.

[0046] The VVC controller 200 may be programmed to reduce the switching frequency from a predetermined maximum frequency in response to the magnitude of the difference between the duty cycle and the predicted duty cycle value when the ripple component amplitude is maximum (e.g., 50%). When the magnitude of the difference is less than a threshold, the switching frequency may be set to a predetermined frequency. When the magnitude of the difference decreases, the switching frequency may increase toward a predetermined maximum frequency to reduce the ripple component amplitude. When the magnitude of the difference is less than a threshold, the switching frequency may be set to a predetermined maximum frequency.

[0047] The switching frequency may be limited to between a predetermined maximum frequency and a predetermined minimum frequency. The predetermined minimum frequency may be a frequency level greater than a predetermined switching frequency of the power electronics module 126 coupled to the output of the variable voltage converter 152. The switching frequency may also be based on parasitic inductance associated with the gate of the IGBT.

[0048] refer to Figure 3 , a system 300 for controlling a power electronics module (PEM) 126 is provided. Figure 3The PEM 126 is shown to include a plurality of switches 302 (e.g., IGBTs) configured to operate together as an inverter having a first phase bridge 316, a second phase bridge 318, and a third phase bridge 320. Although the inverter is shown as a three-phase converter, the inverter may include additional phase bridges. For example, the inverter may be a four-phase converter, a five-phase converter, a six-phase converter, etc. In addition, the PEM 126 may include a plurality of converters, wherein each inverter in the PEM 126 includes three or more phase bridges. For example, the system 300 may control two or more inverters in the PEM 126. The PEM 126 may also include a DC-to-DC converter having a high power switch (e.g., IGBT) to convert a power electronics module input voltage to a power electronics module output voltage by stepping up, stepping down, or a combination thereof.

[0049] like Figure 3 As shown, the inverter can be a DC to AC converter. In operation, the DC to AC converter receives DC power from the DC power link 306 via the DC bus 304 and converts the DC power to AC power. The AC power is transmitted via phase currents ia, ib, and ic to drive an AC motor (also referred to as motor 114), such as Figure 3 The depicted three-phase permanent-magnet synchronous motor (PMSM). In such an example, the DC power link 306 may include a DC battery for providing DC power to the DC bus 304. In another example, the inverter may operate as an AC to DC converter that converts AC power from the AC motor 114 (e.g., a generator) to DC power that the DC bus 304 may provide to the DC power link 306. In addition, the system 300 may control the PEM 126 in other power electronic topologies.

[0050] Continue to refer Figure 3 Each phase bridge 316, 318, 320 in the inverter includes a power switch 302, which can be implemented by various types of controllable switches. In one embodiment, each power switch 302 can include a diode and a transistor (eg, an IGBT). Figure 3 The diode is marked as D a1 , D a2 , D b1 , D b2 , D c1 and D c2 ,and Figure 3 The IGBTs are marked as S a1 , S a2 , S b1 , S b2 , S c1and S c2 . Power switch S a1 , S a2 , D a1 and D a2 It is part of the phase bridge A of the three-phase converter. Figure 3 In FIG. 3 , the first phase bridge A 316 is marked. Similarly, the power switch S b1 , S b2 , D b1 and D b2 is part of the phase bridge B 318, and the power switch S c1 , S c2 , D c1 and D c2 It is part of the phase bridge C 320 of the three-phase converter. Depending on the specific configuration of the inverter, the inverter may include any number of power switches 302 or circuit elements. The diode (D xx ) and IGBT(S XX ) are connected in parallel, however, for correct operation, the polarity is reversed and this configuration is often referred to as an anti-parallel connection. The diodes in this anti-parallel configuration are also known as freewheeling diodes.

[0051] like Figure 3 As shown, the current sensor CS a , CS b and CS c is provided to sense the current flow in the respective phase bridge 316 , 318 , 320 . Figure 3 The current sensor CS is shown separated from the PEM 126 a , CS b and CS c However, depending on the configuration of the PEM 126, the current sensor CS a , CS b and CS c May be integrated as part of the PEM 126 . Figure 3 The current sensor CS a , CS b and CS c Can be used with phase bridges A, B and C (i.e. Figure 3 Each of the phase bridges 316, 318, 320 in the system 300 (also in Figure 3 ) provides a corresponding feedback signal i as 、i bs and i cs . Feedback signal i as 、i bs and i csThe current signal may be a raw current signal processed by a logic device (LD) 310, or may be embedded or encoded with data or information about the current flow through the corresponding phase bridge 316, 318, 320. In addition, the power switch 302 (e.g., IGBT) may include current sensing capabilities. The current sensing capabilities may include a current mirror output configured to provide a current representative of i as 、i bs and i cs The data / signal may indicate the direction of current flow, the magnitude of current flow, or the direction and magnitude of current flow through the corresponding phase bridges A, B and C.

[0052] See again Figure 3 , the system 300 includes a logic device (LD) or controller 310. The controller or LD 310 may be implemented by various types or combinations of electronic devices and / or microprocessor-based computers or controllers. To implement the method of controlling the PEM 126, the controller 310 may execute a computer program or algorithm embedded or encoded with the method and stored in volatile and / or permanent memory 312. Alternatively, the logic may be encoded in discrete logic elements, microprocessors, microcontrollers, or logic or gate arrays stored on one or more integrated circuit chips. Figure 3 As shown in the embodiment, the controller 310 receives and processes the feedback signal i as 、i bs and i cs , to control the phase current i a 、i b and i c , so that the phase current i a 、i b and i c Flow through the phase bridges 316, 318, 320 and into the corresponding windings of the motor 114 according to various current or voltage patterns. For example, the current pattern may include the phase current i a 、i b and i c The flow pattern into and out of the DC bus 304 or the DC bus capacitor 308 is shown separated from the PEM 126. Figure 3 However, the DC bus capacitor 308 may be integrated as part of the PEM 126.

[0053] like Figure 3As shown, a storage medium 312 (hereinafter referred to as "memory") (such as a computer readable memory) can store a computer program or algorithm embedded or encoded with the method. In addition, the memory 312 can store data or information about various operating conditions or components in the PEM 126. For example, the memory 312 can store data or information about the current flow through the corresponding phase bridges 316, 318, and 320. The memory 312 can be part of the controller 310, such as Figure 3 However, memory 312 may be located in any suitable location accessible to controller 310 .

[0054] like Figure 3 As shown, the controller 310 sends at least one control signal 236 to the power converter system 126. The power converter system 126 receives the control signal 236 to control the switch configuration of the inverter, and thus controls the current flow through the corresponding phase bridges 316, 318, and 320. The switch configuration is a set of switch states of the power switch 302 in the inverter. In general, the switch configuration of the inverter determines how the inverter converts power between the DC power link 306 and the motor 114.

[0055] To control the switching configuration of the inverter, the inverter changes the switching state of each power switch 302 in the inverter to an on state or an off state based on the control signal 236. In the illustrated embodiment, to switch the power switch 302 to an on state or an off state, the controller / LD 310 provides a gate voltage (Vg) to each power switch 302 and thus drives the switching state of each power switch 302. The gate voltage Vg a1 , Vg a2 , Vg b1 , Vg b2 , Vg c1 and Vg c2 (like Figure 3 ) controls the switching state and characteristics of the corresponding power switch 302. Figure 3 302. The inverter 302 is shown as a voltage driven device in FIG. 30, but the inverter may be a current driven device or controlled by other strategies for switching the power switch 302 between an on state and an off state. The controller 310 may change the gate drive of each IGBT based on the rotational speed of the motor 114, a current mirror, or the temperature of the IGBT switch. The change in gate drive may be selected from a plurality of gate drive currents, wherein the change in gate drive current is proportional to the change in IGBT switching speed.

[0056] Also like Figure 3As shown, each phase bridge 316, 318 and 320 includes two switches 302. However, only one switch in each of the phase bridges 316, 318, 320 can be in the on state without short-circuiting the DC power link 306. Therefore, in each phase bridge, the switching state of the lower switch is generally opposite to the switching state of the corresponding upper switch. The top switch is generally referred to as the high-side switch (i.e., 302A, 302B, 302C), and the lower switch is generally referred to as the low-side switch (i.e., 302D, 302E, 302F). Therefore, the high state of the phase bridge refers to the upper switch in the phase bridge being in the on state, wherein the lower switch is in the off state. Similarly, the low state of the phase bridge refers to the upper switch in the phase bridge being in the off state, wherein the lower switch is in the on state. Therefore, the IGBT with current mirror performance can be used in all IGBTs, a subset of IGBTs (e.g., S a1 , S b1 , S c1 ) or on a single IGBT.

[0057] exist Figure 3 Two situations may occur during the enabled state of the illustrated three-phase converter example: (1) two phase bridges are in a high state and the third phase bridge is in a low state, or (2) one phase bridge is in a high state and the other two phase bridges are in a low state. Thus, one phase bridge of the three-phase converter (which may be defined as the "reference" phase for a particular enabled state of the inverter) is in an opposite state to the other two phase bridges (or "non-reference" phases) having the same state. Thus, the non-reference phases are either both in a high state or both in a low state during the enabled state of the inverter.

[0058] Figure 4 is a schematic diagram of a gate driver 400 for a power switch having a current mirror and a fixed resistor feedback circuit. Figure 4 4 is an IGBT gate drive circuit 400 with feedback, which has current sensing and open circuit (OC) / short circuit (SC) protection by monitoring the current mirror of the IGBT. The power device 402 is a power switch with a current mirror output driven by a gate controller 404 (i.e., a load IGBT with a mirror IGBT integrated on a single chip so that the gates of the load switch and the mirror switch are connected). The load current flows from the collector to the main emitter, while the current mirror flows from the collector to the mirror emitter. The current flowing out of the mirror emitter flows through the sense resistor 406, so that the voltage on the resistor 406 is equal to the current flowing through the resistor 406 multiplied by the resistance of the resistor 406. When the main IGBT (i.e., the load IGBT) current I CE A small portion of the current (i.e., the current mirror) I flows through the IGBT collector (C) and emitter (E) terminals. METhe current will flow out of the mirror IGBT, and the voltage across resistor R 406 is fed back to the gate driver integrated circuit (IC) for over-current (OC) or / and short-circuit (SC) protection. ME I CE The relationship (e.g., coefficient) is relatively stable when the IGBT is in the on state, but Figure 5 As shown in Figure 2, during the IGBT conduction period, this coefficient may change significantly, where the current mirror value I′ during the conduction process is ME It may be value I ME multiples of (may be greater than 4, 5 or 10 times). Figure 5 The current mirror spike during IGBT turn-on is shown in FIG. 1 . If a fixed resistor R 406 is used for current mirror sensing, the feedback voltage value on the sense resistor R 406 during the turn-on process will be ME = I ME R, when the IGBT is in full conduction state, I′ ME It may be value I ME is many times higher than , and thus limits the value of resistor R 406 that can be used.

[0059] If a fixed resistor R 406 is used for IGBT current mirror measurement during IGBT on-time, the gate drive circuit 404 will receive a voltage based on the IGBT current mirror that is significantly greater than a voltage based on the value of the IGBT current when in the fully on state. To avoid false triggering of the IGBT OC / SC protection, V ME The protection threshold is set higher than V′ ME . This represents an OC / SC protection threshold during turn-on, which is higher than the IGBT (e.g., >5x, 10x, or 20x) when the device is on and the channel is saturated. Based on this, the fault window is typically selected to be larger than the turn-on time (e.g., a 1 μs turn-on time may have a 5 μs fault window). Filtering can be used to mitigate problems caused by current mirror spikes in the feedback signal. However, with the help of a low-pass filter, there is a trade-off between current spike suppression and protection response. Taking advantage of the dynamic characteristics of the IGBT current mirror during turn-on, it may be necessary to increase the OC / SC protection threshold and the fault determination window time. The increase may result in an increased current protection level, thereby reducing the response time for both the dynamic process (i.e., when the IGBT is turned on) and the steady-state process (i.e., the IGBT is fully turned on steady state).

[0060] Figure 5 Is has Figure 4500 of the characteristics of a power switch of a current mirror with respect to time. The gate to emitter voltage 502, the collector to main emitter current 504, the voltage 506 on the collector and main emitter, the current mirror emitter current 508, and the mirror voltage 510 (such as the voltage on a resistor (e.g., resistor R 406) coupled between the main emitter and the current mirror emitter) are shown relative to time 520. At the firing time point 512, the gate voltage 502 increases to turn on the gate controlled switch (e.g., IGBT). As the channel in the power switch strengthens, current begins to flow between the collector and the main emitter 504 and the mirror emitter 508 (also referred to as the current mirror emitter 508). At a second time point 514 at which the gate voltage 504 is equal to the Miller plateau gate voltage, the gate voltage 504 remains generally constant as additional charge is applied to the gate. During this period, the voltage on the collector and main emitter 506 begins to decrease as the device approaches saturation. At a third time point 516, the device enters saturation and V CE 506 remains flat, while the current mirror drops to an operating level at time point 518 .

[0061] Figure 6 6 is a schematic diagram of a gate driver 600 for a power switch with a current mirror and a variable resistor feedback circuit. In this illustration, the power switch is an IGBT with a gate drive circuit 600 with feedback, which has current sensing and open circuit (OC) / short circuit (SC) protection by monitoring the current mirror of the IGBT. The power device 602 is a load IGBT with a mirror IGBT integrated on a single chip, so that the gates of the load switch and the mirror switch are connected and driven by a gate controller 604. The load current flows from the collector to the main emitter, and the current mirror flows from the collector to the mirror emitter. The current flowing out of the mirror emitter flows through a sense resistor 606, so that the voltage on the resistor 606 is equal to the current flowing through the resistor 606 multiplied by the resistance of the resistor 606. Here, the resistor 606 is a variable resistor, and the gate drive controller 604 is coupled to the variable resistor 606 so that the controller can change the resistance of the variable resistor 606. When the main IGBT (i.e., the load IGBT) current I CE A small portion of the current (i.e., the current mirror) I flows through the IGBT collector (C) and emitter (E) terminals. ME The current will flow out of the mirror IGBT, and the voltage across the variable resistor R 606 is fed back to the gate driver integrated circuit (IC) for over-current (OC) or / and short-circuit (SC) protection. ME I CE The relationship (e.g., coefficient) is relatively stable when the IGBT is in the on-state (static or non-transient operation). Figure 7As shown, this coefficient may change significantly during the IGBT turn-on (dynamic or transient operation), where the current mirror value I′ during the turn-on process is ME It may be value I ME The variable resistor 606 may be controlled to compensate for the Figure 7 Current mirror spike during IGBT turn-on as shown. For example, variable resistor R606 may initially be at a low value during turn-on until the gate voltage exceeds the Miller plateau gate voltage, and at that time, the variable resistance may increase such that the voltage across variable resistor 606 generally remains small with a small resistance multiplier R606, and after exceeding the Miller plateau, the resistance increases such that the current I ME A change in results in a larger change in the voltage across resistor 606.

[0062] For example, the controller sets the resistance of R 606 to a very low value during the on-time, and the controller then increases the resistance of R 606 back to the nominal value after the IGBT is saturated (i.e., fully turned on). The on-resistance may be 1 / 10, 1 / 50, 1 / 100, or 1 / 1000 of the value during operation (nominal operation). Figure 7 The corresponding voltage and current waveforms are shown in . Since the turn-on process is closely related to the gate voltage, the resistance of R 606 can be gate voltage dependent.

[0063] Figure 7 yes Figure 6 700 of the characteristics of a power switch and a mirror switch with respect to time. A gate-to-emitter voltage 702, a collector-to-main emitter current 704, a voltage 706 on the collector and the main emitter, a current mirror emitter current 708, and a mirror voltage 710 (such as the voltage on a resistor (e.g., resistor R 406) coupled between the main emitter and the mirror emitter) are shown with respect to time 720. When the power device receives a signal to turn on, the controller can set the resistance of the resistor 606 to a very low level. Then, at a firing time point 712, the gate voltage 702 increases to turn on the gate controlled switch (e.g., an IGBT). As the channel in the power switch strengthens, current begins to flow between the collector and the main emitter 704 and the mirror emitter 708. At a second time point 714 at which the gate voltage 704 is equal to the Miller plateau gate voltage, the gate voltage 704 remains generally constant as additional charge is applied to the gate. During this period, as the device approaches saturation, the voltage on the collector and the main emitter 706 begins to decrease. The controller may then change the resistance of the resistor 606 at a third time point 716 so that as the device enters saturation, the feedback voltage V ME can be increased based on the increased resistance of R 606 until VCE 706 remains flat and the current mirror drops to an operating level at time point 718 .

[0064] Figure 8 800 is a schematic diagram of a gate driver 800 for a power switch with a current mirror and optional resistor feedback circuit. In this illustration, the power switch is an IGBT with a gate drive circuit 800 with feedback, which has current sensing and open circuit (OC) / short circuit (SC) protection by monitoring the current mirror of the IGBT. The power device 802 is a load IGBT with a mirror IGBT integrated monolithically, so that the gates of the load switch and the mirror switch are connected and driven by a gate controller 804. The load current flows from the collector to the main emitter, while the current mirror flows from the collector to the mirror emitter. The current flowing out of the mirror emitter flows through the sense resistor R1 806 and flows through the on resistor R2 808 when the switch 810 is turned on, so that V ME The voltage across 808 is equal to the current flowing through the parallel combination of resistor R1 806 and resistor R2 808 multiplied by the resistance of the parallel combination of resistor R1 806 and resistor R2 808. The resistance of resistor 808 may be lower than the resistance of resistor R1 806.

[0065] Therefore, before t1 906, V GE The gate voltage of switch S2 is at a low level (eg, substantially zero volts) and IGBTS1 802 is in the off state, and the control unit of switch S2 detects that the gate voltage V GE And output signal V GE2 To turn on S2 810, at this point in time, the current mirror senses the voltage V ME will be substantially zero. Then, from time t1 906 to t2 908, IGBT 802 begins to conduct, and V GE2 Still remains high to keep S2 810 on. When S2 810 is on, the equivalent sense resistance is equal to R1 806 / R2 808, which is less than R2 808. R1 806 is selected to keep V ME The voltage spike is minimized. For example, the value of R1 may be close to zero. After t2 908, the control unit of switch S2 detects that the voltage is above the preset threshold V GE-pre (e.g., Miller plateau gate voltage) GE , which indicates that the IGBT 802 is in a fully on state (eg, saturated). After t2 908, the current mirror returns to the standard operating value and the corresponding IGBT current I CE , and the controller outputs a low level (e.g., zero volt) control gate voltage V GE2 , to turn off S2 810.

[0066] Fig. 9yes Figure 6 900 of the control and characteristics of the power switch and mirror switch of FIG. 902, the gate to emitter voltage 902, the variable resistance control signal 904 are shown relative to time 910. When the power switch is first turned on at time t1 906, the power switch control signal is high, enabling the second resistor to be coupled in parallel to reduce the sense resistance (e.g., the power switch 810 is turned on and R2 808 is coupled in parallel with R1 806, thereby reducing the effective resistance). When the gate voltage exceeds the Miller plateau gate voltage at point t2 908, the controller disables the parallel resistor R2 808, so that the effective resistance is then reduced to the resistance of R1 806.

[0067] Fig.10 1 is a schematic diagram of a gate driver 1000 for a power switch with a current mirror and optional resistor feedback circuit. In this illustration, the power switch is an IGBT with a gate drive circuit 1000 with feedback, which has current sensing and open circuit (OC) / short circuit (SC) protection by monitoring the current mirror of the IGBT. The power device 1002 is a load IGBT with a mirror IGBT integrated monolithically, so that the gates of the load switch and the mirror switch are connected and driven by a gate controller 1004. The load current flows from the collector to the main emitter, while the current mirror flows from the collector to the mirror emitter. During the on-time, the controller outputs a signal R C To select the on-resistance value. Signal R C This can be a single signal or multiple signals forming a bus. For example, R C It can be a multiplexed signal, such as a Serial Peripheral Interface (SPI) bus, a Universal Asynchronous Receiver / Transmitter (UART) bus, an I 2 C bus, 1-wire or other formats, or R C can be parallel buses that are each coupled to a switch or encoded to select a switch of the solid-state variable resistor 1010. In addition, along with the selection of the resistor, the signal R C An enable signal may be included to engage / disengage the selected switch. The on-resistance value is equal to the equivalent resistance of the selected resistance of the solid-state variable resistor 1010, which is connected in series with the on resistor R2 1008, both of which are connected in parallel with the operating resistor R1 1006. C When all switches are turned on, the equivalent resistance is just the resistance of R1 1006, and the current flowing out of the mirror emitter flows through the sense resistor R1 1006, making V METhe voltage across V is equal to the current flowing through resistor R1 806 multiplied by the resistance of resistor R1 806. However, during the on period, the controller may enable the resistance of solid-state variable resistor 1010. For example, if the controller outputs a signal to close the first switch, where the resistance of solid-state variable resistor 1010 is substantially zero, then V ME The voltage across will be equal to the current flowing through the parallel combination of resistor R1 1006 and resistor R2 1008 multiplied by the resistance of the parallel combination of resistor R1 1006 and resistor R2 1008. In this embodiment, the controller may vary the selection of resistance based on criteria including ambient temperature, temperature of power device 1002, power demand, or other operational aspects.

[0068] In conventional current sensing systems with OC / SC protection, the feedback resistor value is selected to be much higher than the value required for the IGBT conduction current level for normal operation (e.g. >5 times), and the fault window time must be greater than the IGBT turn-on time (e.g. 1 μs). This results in an increased current protection level and an increased protection response time. The performance of the OC / SC protection method is limited / affected by the IGBT dynamic process (IGBT turn-on). Here, a variable resistor is used to reduce the current mirror spike in the feedback signal during IGBT turn-on, lowering the OC / SC protection threshold (e.g. 2 times the normal current level) and reducing the response window time (e.g. 0.5 μs) without limiting the IGBT turn-on.

[0069] The control logic or functions performed by the controller may be represented by a flowchart or similar diagram in one or more of the accompanying drawings. These drawings provide representative control strategies and / or logic that can be implemented using one or more processing strategies (such as event-driven, interrupt-driven, multi-tasking, multi-threading, etc.). In this way, the various steps or functions shown may be performed in the order shown, in parallel, or omitted in some cases. Although not always explicitly shown, it will be recognized by those of ordinary skill in the art that one or more of the steps or functions shown may be repeatedly performed according to the specific processing strategy used. Similarly, the processing order is not necessarily required to achieve the features and advantages described herein, but is provided for ease of illustration and description. The control logic may be implemented primarily in software executed by a microprocessor-based vehicle, engine and / or powertrain controller (such as a controller). Of course, depending on the specific application, the control logic may be implemented in software, hardware, or a combination of software and hardware in one or more controllers. When implemented in software, the control logic may be provided in one or more computer-readable storage devices or media, which have stored data representing codes or instructions executed by a computer to control a vehicle or its subsystems. The computer-readable storage device or medium may include one or more of a number of known physical devices that utilize electrical, magnetic, and / or optical storage to retain executable instructions and associated calibration information, operating variables, and the like.

[0070] The process, method or algorithm disclosed herein may be delivered to or implemented by a processing device, a controller or a computer, which may include any existing programmable electronic control unit or a dedicated electronic control unit. Similarly, a process, method or algorithm may be stored as data and instructions that can be executed by a controller or a computer in many forms, including but not limited to information permanently stored on a non-writable storage medium such as a read-only memory (ROM) device and information stored in a changeable manner on a writable storage medium, such as a floppy disk, a tape, a compact disc (CD), a random access memory (RAM) device and other magnetic and optical media. A process, method or algorithm may also be implemented in a software executable object. Alternatively, a suitable hardware component (such as an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a state machine, a controller or other hardware component or device) or a combination of hardware, software and firmware components may be used to fully or partially embody a process, method or algorithm.

[0071] Although exemplary embodiments are described above, it is not intended that these embodiments describe all possible forms covered by the claims. The words used in the specification are descriptive rather than limiting, and it should be understood that various changes can be made without departing from the spirit and scope of the present disclosure. As mentioned above, the features of the various embodiments can be combined to form other embodiments of the present invention that may not be explicitly described or illustrated. Although various embodiments can be described as providing advantages or advantages over other embodiments or prior art implementations relative to one or more desired characteristics, those of ordinary skill in the art recognize that one or more features or characteristics can be deducted to achieve the desired overall system properties, depending on the specific application and implementation. These properties may include, but are not limited to: cost, strength, durability, life cycle cost, marketability, appearance, packaging, size, serviceability, weight, manufacturability, ease of assembly, etc. In this way, embodiments described as less desirable than other embodiments or prior art implementations relative to one or more characteristics are not outside the scope of the present disclosure and may be desirable for specific applications.

[0072] According to the present invention, a vehicle powertrain system is provided, which has: a power inverter, the power inverter including a load switch with a main emitter and a current mirror emitter and a variable resistor coupled between the main emitter and the current mirror emitter; and a controller, the controller being configured to adjust a gate voltage based on a voltage across the variable resistor, and in response to the gate voltage exceeding a Miller platform gate voltage, increase a resistance of the variable resistor so that feedback increases as the load switch saturates.

[0073] According to one embodiment, the variable resistor includes a first resistor coupled in parallel with a second resistor, the second resistor being selectable by a solid-state switch, wherein the first resistor has a greater resistance than the second resistor.

[0074] According to one embodiment, the variable resistor is a solid state variable resistor coupled in parallel with the sensing resistor.

[0075] According to one embodiment, the controller is further configured to select the resistance of the solid state variable resistor based on ambient temperature.

[0076] According to one embodiment, the controller is further configured to select a resistance of the solid-state variable resistor based on a temperature of the load switch.

[0077] According to one embodiment, the load switch is an insulated gate bipolar junction transistor (IGBT).

[0078] According to the present invention, a vehicle powertrain system is provided having: a load switch monolithically integrated with a mirror switch; and a variable resistor coupled between an emitter of the mirror switch and a reference point.

[0079] According to one embodiment, the reference point is the chassis or main emitter, and the load switch and the mirror switch are insulated gate bipolar junction transistors (IGBTs).

[0080] According to one embodiment, the variable resistor includes a first resistor coupled in parallel with a second resistor, the second resistor being selectable by a solid-state switch, wherein the first resistor has a greater resistance than the second resistor.

[0081] According to one embodiment, the load switch is configured to cause a current to flow into an inductor of the variable voltage converter.

[0082] According to one embodiment, the load switch is located in a power inverter and is configured to allow current to flow into a phase winding of the motor.

[0083] According to one embodiment, the above invention is further characterized by a controller configured to adjust the gate voltage based on the voltage across the variable resistor, and in response to the gate voltage exceeding the Miller plateau gate voltage, increase the variable resistor resistance so that feedback increases as the load switch saturates.

[0084] According to one embodiment, the variable resistor is a solid state variable resistor coupled in parallel with the sensing resistor.

[0085] According to one embodiment, the controller is further configured to select the resistance of the solid state variable resistor based on ambient temperature.

[0086] According to one embodiment, the controller is further configured to select a resistance of the solid-state variable resistor based on a temperature of the load switch.

[0087] According to the present invention, a method for controlling a power switch in a vehicle powertrain system includes: setting the resistance of a variable resistor coupled between an emitter of a current mirror and a reference point to a first value while a gate voltage of the power switch is below a turn-on threshold; applying charge to the gate to increase the gate voltage; and changing the resistance of the variable resistor in response to the gate voltage exceeding a Miller plateau gate voltage.

[0088] According to one embodiment, changing the resistance of the variable resistor is increasing the resistance of the variable resistor.

[0089] According to one embodiment, the variable resistor includes a first resistor coupled in parallel with a second resistor selectable by a solid-state switch, and changing the resistance of the variable resistor disconnects the second resistor.

[0090] According to one embodiment, the resistance is set based on the temperature of the power switch.

[0091] According to one embodiment, the resistance is set based on the power demand of the vehicle driveline.

Claims

1. A vehicle power transmission system, comprising: A power inverter, the power inverter comprising load switch with main emitter and current mirror emitter, and a variable resistor coupled between the main emitter and the current mirror emitter; as well as A controller is configured to adjust a gate voltage based on a voltage across the variable resistor and, in response to the gate voltage exceeding a Miller plateau gate voltage, increase a resistance of the variable resistor such that a feedback voltage increases as the load switch saturates.

2. The vehicle powertrain system of claim 1, wherein: The variable resistor includes a first resistor coupled in parallel with a second resistor selectable by a solid-state switch, wherein the first resistor has a greater resistance than the second resistor.

3. The vehicle powertrain system of claim 1, wherein: The variable resistor is a solid state variable resistor coupled in parallel with the sensing resistor.

4. The vehicle powertrain system of claim 3, wherein: The controller is further configured to select a resistance of the solid state variable resistor based on ambient temperature.

5. The vehicle powertrain system of claim 3, wherein: The controller is further configured to select a resistance of the solid-state variable resistor based on a temperature of the load switch.

6. The vehicle powertrain system of claim 1, wherein: The load switch is an insulated gate bipolar junction transistor.

7. A vehicle power transmission system, comprising: A load switch monolithically integrated with a mirror switch; a variable resistor coupled between an emitter of the mirror switch and a reference point; as well as A controller is configured to adjust a gate voltage based on a voltage across the variable resistor and, in response to the gate voltage exceeding a Miller plateau gate voltage, increase a resistance of the variable resistor such that a feedback voltage increases as the load switch saturates.

8. The vehicle powertrain system of claim 7, wherein: The reference point is the chassis or main emitter, and the load switch and the mirror switch are insulated gate bipolar junction transistors.

9. The vehicle powertrain system of claim 7, wherein: The variable resistor includes a first resistor coupled in parallel with a second resistor selectable by a solid-state switch, wherein the first resistor has a greater resistance than the second resistor.

10. The vehicle powertrain system of claim 7, wherein: The load switch is configured to cause current to flow into an inductor of the variable voltage converter.

11. The vehicle powertrain system of claim 7, wherein: The load switch is located in the power inverter and is configured to allow current to flow into a phase winding of the motor.

12. The vehicle powertrain system of claim 7, wherein: The variable resistor includes a first resistor coupled in parallel with a second resistor selectable by a solid-state switch, and changing the resistance of the variable resistor disconnects the second resistor.

13. The vehicle powertrain system of claim 7, wherein: The variable resistor is a solid state variable resistor coupled in parallel with the sensing resistor.

14. The vehicle powertrain system of claim 13, wherein: The controller is further configured to select a resistance of the solid state variable resistor based on ambient temperature.

15. The vehicle powertrain system of claim 13, wherein: The controller is further configured to select a resistance of the solid-state variable resistor based on a temperature of the load switch.

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