Radar waveform generation circuit, method, radar and computer storage medium

Abstract

The present application provides a radar waveform generation circuit, method, radar and computer storage medium. In the radar waveform generation circuit, a frequency synthesizer sends a waveform completion signal when a set waveform is completed according to a first enable signal sent by a second counter; when a first counter finishes counting the number of waveform completion signals, it sends a strobe conversion signal to a strobe; the strobe is used to replace a first time interval with a second time interval as time interval data according to the strobe conversion signal and send it to the second counter; the second counter is used to perform time counting according to the waveform completion signal and the time interval data, and send a second enable signal to the frequency synthesizer when the time counting is completed. By setting different time intervals between two sets of set waveforms, different Doppler resolutions can be set between the two sets of waveforms. Through further processing of the echo, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

Description

Technical Field

[0001] The present application relates to the technical field of radar detection, and in particular, to a radar waveform generation circuit, method, radar, and computer storage medium. Background Art

[0002] In in-vehicle millimeter-wave radar products, FMCW (Frequency Modulated Continuous Wave) radar is one of the most widely used. This radar generates a frequency-domain ramp signal (Chirp) with continuously changing frequency through a waveform generator and a voltage-controlled oscillator (VCO), transmits it to the external detection space through an antenna, and mixes and demodulates the obstacle echo signal with a slight delay (proportional to the distance) with the transmitted signal to obtain a baseband signal with a constant frequency.

[0003] FMCW radar generally uses FFT (Fast Fourier Transform) to perform a primary analysis on the time-domain baseband signal. Each FFT peak obtained from the analysis represents one or more obstacle targets.

[0004] FMCW radar generally uses FFT to form a virtual sampling sequence according to the spectral lines where the targets of the above first range Fourier transform (RFFT) are located at the moment of each Chirp and perform FFT. Each FFT spectral line peak obtained from the analysis represents one or more obstacle targets with the same distance but different Doppler velocities.

[0005] In the above generally used radar signal processing methods, the dynamic measurement range in the range dimension (maximum measurement range divided by range resolution) is determined by the number of spectral lines (bins) of the range FFT, and the dynamic measurement range in the Doppler velocity dimension (maximum measured Doppler velocity ÷ Doppler velocity resolution) is determined by the number of spectral lines of the Doppler FFT. Currently, due to the significant increase in the system complexity of autonomous driving and assisted driving, the market demand for the dynamic measurement range of in-vehicle radar products is also increasing day by day, and thus the demand for the scale of the number of spectral lines of FFT in each dimension is also increasing day by day. In engineering implementation, a larger number of points of FFT basically means higher memory (RAM) occupancy and also means higher hardware costs. At the same time, a larger number of points of FFT also requires more time for ADC sampling on the baseband signal, so a higher duty cycle and higher average transmit power are required. Generally speaking, due to the limitations of engineering implementation, it is difficult for in-vehicle radar products to improve the dynamic measurement range of products by infinitely expanding the number of points of each level of FFT, and it is difficult to improve the maximum measurement value and minimum measurement resolution. Summary of the Invention

[0006] The object of the present application is to provide a radar waveform generation circuit, method, radar and computer storage medium, so as to solve the problem that it is difficult for vehicle-mounted millimeter-wave radars to improve the maximum measurement value and the minimum measurement resolution.

[0007] In one aspect, the present application provides a radar waveform generation circuit. Specifically, the radar waveform generation circuit includes a strobe, a first memory storing a first time interval, a second memory storing a second time interval, a first counter, a second counter and a frequency synthesizer, wherein:

[0008] The frequency synthesizer is connected to the first counter and the second counter. The frequency synthesizer generates a set waveform of a frequency-modulated continuous wave according to a first enable signal sent by the second counter, and sends a waveform completion signal to the first counter and the second counter when the set waveform is completed;

[0009] The first counter is used to count the number of the waveform completion signals, and send a strobe conversion signal to the strobe when the number counting is completed;

[0010] The strobe is used to replace the selected first time interval with the second time interval according to the strobe conversion signal, so as to send the second time interval as time interval data to the second counter;

[0011] The second counter is used to read the time interval data according to the waveform completion signal, perform time counting according to the time interval data, and send a second enable signal to the frequency synthesizer when the time counting is completed.

[0012] Optionally, the strobe includes a first group of data input pins, a second group of data input pins, a selection pin and a data output pin. The first group of data input pins is connected to the first memory, the second group of data input pins is connected to the second memory, the selection pin is connected to the first counter, and the data output pin is connected to the second counter.

[0013] Optionally, the first counter includes a first counting chip, a second counting chip and a first NOT gate. The clock input terminals of the first counting chip and the second counting chip are respectively connected to the frequency synthesizer. The carry output pin of the first counting chip is connected to the counting control pin of the second counting chip. The lowest bit data output pin of the second counting chip is connected to the strobe. The first NOT gate is connected in series between the second lowest bit data output pin and the clear pin of the second counting chip.

[0014] Optionally, the frequency synthesizer includes a second AND gate, a clock source, and a frequency synthesis chip. The first input terminal of the second AND gate is connected to the clock source, the second input terminal of the second AND gate is connected to the second counter to receive the enable signal, and the output terminal of the second AND gate is connected to the frequency synthesis chip.

[0015] Optionally, the second counter includes a NOT gate array, a first AND gate, a third counting chip, a second NOT gate, and a third NOT gate. Among them, the NOT gate array is connected between the data input pin of the third counting chip and the strobe, the first AND gate is connected between the data output pin of the third counting chip and the input terminal of the second NOT gate, the output terminal of the first AND gate is connected to the frequency synthesizer to output the enable signal, the output terminal of the second NOT gate is connected to the counting control pin of the third counting chip, the output terminal of the third NOT gate is connected to the preset control pin of the third counting chip, and the input terminal of the third NOT gate is connected to the frequency synthesizer to receive the waveform completion signal.

[0016] Optionally, the second counter further includes a first NAND gate. The first input terminal of the first NAND gate is connected to the output terminal of the second NOT gate, the second input terminal of the first NAND gate is connected to the output terminal of the third NOT gate, and the output terminal of the first NAND gate is connected to the frequency synthesizer to output the enable signal.

[0017] On the other hand, the present application also provides a radar. Specifically, the radar includes an antenna and the above-mentioned radar waveform generation circuit connected to each other.

[0018] Optionally, the radar further includes a main controller for controlling the frequency modulated continuous wave, and the main controller is connected to the frequency synthesizer in the radar waveform generation circuit.

[0019] On the other hand, the present application also provides a radar waveform generation method. Specifically, the radar waveform generation method includes:

[0020] Obtaining a first enable signal;

[0021] Generating a set waveform of the frequency modulated continuous wave according to the first enable signal, and generating a waveform completion signal when the set waveform is completed;

[0022] Obtaining time gap data according to the waveform completion signal;

[0023] Performing time counting according to the time gap data, and generating a second enable signal when the time counting is completed;

[0024] The step of obtaining time gap data according to the waveform completion signal includes:

[0025] Count the quantity of the waveform completion signal, and generate a strobe conversion signal when the quantity counting is completed, so as to replace the strobe of the first time interval with the second time interval as the time interval data.

[0026] On the other hand, the present application also provides a computer storage medium. Specifically, a computer program is stored on the computer storage medium, and when the computer program is executed by a processor, the radar waveform generation method as described above can be implemented.

[0027] The radar waveform generation circuit, method, radar and computer storage medium provided by the present application can set different Doppler resolutions between two sets of waveforms by different time intervals between two sets of set waveforms. By further processing the echo, the measurement range can be effectively increased and the minimum measurement resolution can be further refined. BRIEF DESCRIPTION OF THE DRAWINGS

[0028] Figure 1 It is a flowchart of a radar waveform design method according to an embodiment of the present application.

[0029] Figure 2 It is a block diagram of a radar waveform design device according to an embodiment of the present application.

[0030] Figure 3 It is a schematic diagram of a radar wave according to an embodiment of the present application.

[0031] Figure 4 It is a flowchart of a radar wave processing method according to an embodiment of the present application.

[0032] Figure 5 It is a block diagram of a radar wave processing device according to an embodiment of the present application.

[0033] Figure 6 It is a flowchart of a sub - Doppler velocity processing method according to an embodiment of the present application.

[0034] Figure 7 It is a schematic diagram of the sampling of the pitch observation vector sequence according to an embodiment of the present application.

[0035] Figure 8 It is a block diagram of a radar waveform generation circuit according to an embodiment of the present application.

[0036] Figure 9 It is a circuit connection diagram of a strobe according to an embodiment of the present application.

[0037] Figure 10 It is a circuit connection diagram of a first counter according to an embodiment of the present application.

[0038] Figure 11 It is a circuit connection diagram of a frequency synthesizer according to an embodiment of the present application.

[0039] Figure 12 Block diagram of a radar according to an embodiment of the present application.

[0040] Figure 13 Flowchart of a radar waveform generation method according to an embodiment of the present application. Detailed implementation manners

[0041] The following will further describe in detail the specific implementation manners of the present application in conjunction with the accompanying drawings and embodiments. The following embodiments are used to illustrate the present application, but are not used to limit the scope of the present application.

[0042] In one aspect of the present application, a radar waveform design method is provided. Figure 1 Flowchart of a radar waveform design method according to an embodiment of the present application.

[0043] As Figure 1 shown, in one embodiment, the radar waveform design method includes:

[0044] S110: Set a first center wavelength, a first waveform duration, a first number of waveforms, a first waveform gap, a second center wavelength, a second waveform duration, a second number of waveforms, and a second waveform gap.

[0045] By preset waveform parameters, a variety of specific waveforms with different Doppler resolutions are achieved. Waveforms with different Doppler resolutions can be achieved by setting different center wavelengths, different waveform durations, different numbers of waveforms, and different waveform gaps. For example, different Doppler resolutions can be achieved by setting different waveform slopes or different RF bandwidths to obtain different total observation durations.

[0046] S120: According to the first center wavelength and the first waveform duration, output the frequency-modulated continuous wave of the first tone in the radar beam of the first number of waveforms, and there is a first waveform gap between adjacent frequency-modulated continuous wave waveforms of the first tone.

[0047] Output the first tone of the radar beam through the parameters of the first type of waveform.

[0048] S130: According to the second center wavelength and the second waveform duration, output the frequency-modulated continuous wave of the second tone in the radar beam of the second number of waveforms, and there is a second waveform gap between adjacent frequency-modulated continuous wave waveforms of the second tone.

[0049] Output the second tone of the radar beam through the parameters of the second type of waveform. Among them, under different waveform parameters, the second tone can have a different Doppler resolution from the first tone.

[0050] Multiple tones with different Doppler resolutions are set in the radar beam, which is convenient for processing the echo, not only to extend the measurement range of the Doppler velocity upward, but also to subdivide the measurement resolution of the Doppler velocity downward.

[0051] In one embodiment, in the step of executing S110: setting the first central wavelength, the first waveform duration, the first number of waveforms, the first waveform gap, the second central wavelength, the second waveform duration, the second number of waveforms, and the second waveform gap, it includes:

[0052] Set according to the following formula:

[0053] [λ1*N2*(t21 + t22)] / [λ2*N1*(t11 + t12)] ≠ A

[0054] Where λ1 is the first central wavelength, N1 is the first number of waveforms, t11 is the first waveform duration, t12 is the first waveform gap, λ2 is the second central wavelength, N2 is the second number of waveforms, t21 is the second waveform duration, t22 is the second waveform gap, and A is any positive integer.

[0055] In order to extend the measurement range of the Doppler velocity upward, when setting the waveform parameters, control the ratio of the Doppler resolution of the first tone to the Doppler resolution of the second tone not to be an integer.

[0056] In one embodiment, in the step of executing S110: setting the first central wavelength, the first waveform duration, the first number of waveforms, the first waveform gap, the second central wavelength, the second waveform duration, the second number of waveforms, and the second waveform gap, it includes:

[0057] Set according to the following formula:

[0058] [λ1*N2*(t21 + t22)] / [λ2*N1*(t11 + t12)] = (M + 1) / M

[0059] Where λ1 is the first central wavelength, N1 is the first number of waveforms, t11 is the first waveform duration, t12 is the first waveform gap, λ2 is the second central wavelength, N2 is the second number of waveforms, t21 is the second waveform duration, t22 is the second waveform gap, and M is a positive integer greater than 1.

[0060] When setting the waveform parameters, control the ratio of the Doppler resolution of the first tone to the Doppler resolution of the second tone to be the ratio of two adjacent integers, which is beneficial to reducing the calculation amount of the radar system.

[0061] In one embodiment, before performing step S130: outputting the frequency-modulated continuous wave of the second tone portion of the radar beam of the second waveform number according to the second center wavelength and the second waveform duration, with a second waveform gap between adjacent frequency-modulated continuous wave waveforms of the second tone portion, the following steps are included:

[0062] In one beam, the output of the second tone portion starts immediately after the output of the first tone portion is completed.

[0063] The two tone portions are as close as possible to each other in time, which can avoid blank areas in the measurement range, especially reducing the analysis blind area of the second Doppler Fourier transform.

[0064] On the other hand, the present application also provides a radar waveform design device. Figure 2 It is a block diagram of the radar waveform design device according to an embodiment of the present application.

[0065] As Figure 2 shown, in one embodiment, the radar waveform design device includes a memory 11, a processing module 12, and a frequency synthesis module 13 that are connected in sequence, where:

[0066] The memory 11 is used to store and send the first center wavelength, the first waveform duration, the first waveform number, the first waveform gap, the second center wavelength, the second waveform duration, the second waveform number, and the second waveform gap to the processing module 12;

[0067] The processing module 12 is used to first control the frequency synthesis module 13 to output the frequency-modulated continuous wave of the first tone portion of the radar beam of the first waveform number according to the first center wavelength and the first waveform duration, where there is a first waveform gap between adjacent frequency-modulated continuous wave waveforms of the first tone portion; the processing module 12 then controls the frequency synthesis module 13 to output the frequency-modulated continuous wave of the second tone portion of the radar beam of the second waveform number according to the second center wavelength and the second waveform duration, with a second waveform gap between adjacent frequency-modulated continuous wave waveforms of the second tone portion.

[0068] By presetting the parameters of the waveform, the first and second tone portions of the radar beam are output. Among them, the second tone portion has a different Doppler resolution from the first tone portion.

[0069] Setting multiple tone portions with different Doppler resolutions in the radar beam facilitates not only expanding the measurement range of the Doppler velocity upward but also subdividing the measurement resolution of the Doppler velocity downward in the processing of the echo.

[0070] In one embodiment, the memory 11 is set and stored according to the following formula:

[0071] [λ1 * N2 * (t21 + t22)] / [λ2 * N1 * (t11 + t12)] ≠ A

[0072] Wherein, λ1 is the first central wavelength, N1 is the number of the first waveforms, t11 is the duration of the first waveform, t12 is the gap between the first waveforms, λ2 is the second central wavelength, N2 is the number of the second waveforms, t21 is the duration of the second waveform, t22 is the gap between the second waveforms, and A is any positive integer.

[0073] To achieve upward expansion of the measurement range of the Doppler velocity, when setting the waveform parameters, the ratio of the Doppler resolution of the first tone part to the Doppler resolution of the second tone part is controlled not to be an integer.

[0074] In one embodiment, the number of the first waveforms output by the frequency synthesis module 13 is the same as the number of the second waveforms.

[0075] The same number of waveforms in the two tone parts is beneficial to reducing the computational load of the radar system.

[0076] In one embodiment, the waveform of the frequency-modulated continuous wave output by the frequency synthesis module 13 is selected from at least one of a triangular wave, a sawtooth wave, a stepped wave, and a sine wave.

[0077] In the practice of radar applications, according to the actual application environment, multiple different waveforms can be selected and used to meet the needs of radar ranging and velocity measurement.

[0078] In one embodiment, the frequency synthesis module 13 outputs multiple scan frames within each radar scan cycle, and multiple radar beams in different directions are output within each scan frame.

[0079] By transmitting beams in multiple different directions, echo signals in multiple different directions can be obtained, and thus the detection target can be identified more accurately.

[0080] Figure 3 This is a schematic diagram of radar waves according to an embodiment of the present application.

[0081] Please refer to Figure 3 , in one embodiment, a waveform transmitted by the radar is named "Duo", which is a multi-frame and multi-beam multi-level complex frequency-modulated continuous wave (FMCW) waveform, and its basic characteristics are as follows:

[0082] Each radar detection cycle (Cycle) is equally divided into multiple scan frames (Frame#1, Frame#2...) in time.

[0083] Multiple beams (Duo Beam#1, Duo Beam#2, Duo Beam#3...) in different directions, with different range resolutions and different Doppler resolutions, are transmitted within each scan frame.

[0084] Each beam is a kind of Duo Chirp group, which at least includes two Chirp Groups with high resolution (H part) and low resolution (L part). The Doppler resolutions of these two Chirp Groups are slightly different, always appear adjacent to each other, and are combined into the final extended result at an appropriate moment, which are called two "parts" of the Duo in this application.

[0085] The H and L parts are as close as possible in time (referring to the time-frequency diagram of the attached figure), that is, immediately after the H part ends, the L part starts to be transmitted. The number of chirps in the two parts is the same, each being half of the total number of chirps in the beam where they are located.

[0086] Setting multiple parts with different Doppler resolutions in the radar beam is convenient for processing the echo, which can not only extend the measurement range of the Doppler velocity upward, but also subdivide the measurement resolution of the Doppler velocity downward.

[0087] On the other hand, this application also provides a computer storage medium.

[0088] A computer program is stored on the computer storage medium. When the computer program is executed by a processor, it can implement the radar waveform design method as described above.

[0089] The radar waveform design method, device and computer storage medium provided by this application can effectively improve the dynamic measurement range of the product, and improve the maximum measurement value and the minimum measurement resolution by setting multiple parts with different Doppler resolutions in the radar beam.

[0090] On the other hand, this application also provides a radar wave processing method. Figure 4 It is a flowchart of the radar wave processing method according to an embodiment of this application.

[0091] As Figure 4 shown, the radar wave processing method includes:

[0092] S210: Perform Fourier transform on the baseband signals of the echoes of multiple parts in the radar beam respectively to obtain corresponding multiple frequency domain data.

[0093] In engineering practice, fast Fourier transform and discrete Fourier transform can be performed on the echo of the time domain signal to obtain the spectral data in the frequency domain, and then the detection target can be identified. The Fourier transform can include range Fourier transform and Doppler Fourier transform.

[0094] S220: Obtain corresponding multiple original signal-to-noise ratio matrices respectively according to the multiple frequency domain data.

[0095] The frequency domain data of multiple parts can respectively obtain the range-Doppler matrix data of multiple parts as the original signal-to-noise ratio matrix.

[0096] S230: Combine and accumulate multiple original signal-to-noise ratio matrices into an extended signal-to-noise ratio matrix.

[0097] For multiple original signal-to-noise ratio matrices formed by echoes of multiple voice parts with different Doppler resolutions, it is necessary to further integrate and combine them into an extended signal-to-noise ratio matrix to accumulate signal-to-noise ratio data.

[0098] S240: Screen target points according to the extended signal-to-noise ratio matrix.

[0099] Based on the extended signal-to-noise ratio matrix formed by the accumulated signal-to-noise ratio data, target points can be screened within a larger Doppler measurement range.

[0100] By processing the echoes including multiple voice parts, on the basis of the original signal-to-noise ratio matrix, further through the mapping relationship, combined and accumulated into the extended signal-to-noise ratio matrix, it can extend the Doppler measurement interval upward and effectively expand the measurement range.

[0101] In one embodiment, in the step of performing S210: Perform Fourier transform on the baseband signals of the echoes of multiple voice parts in the radar beam to obtain corresponding multiple frequency domain data, it includes:

[0102] Perform Doppler Fourier transform and / or range Fourier transform on the baseband signals of the echoes of multiple voice parts respectively.

[0103] In one embodiment, in the step of performing S220: Obtain corresponding multiple original signal-to-noise ratio matrices according to multiple frequency domain data respectively, it includes:

[0104] Obtain the logarithmic signal intensity at all coordinate positions in the original signal-to-noise ratio matrix to generate a screening signal-to-noise ratio threshold;

[0105] Perform distribution statistics on the logarithmic signal intensity of the surrounding coordinate positions of the first coordinate position to obtain the logarithmic noise intensity of the first coordinate position according to the preset noise algorithm;

[0106] Use the difference between the logarithmic signal intensity and the logarithmic noise intensity of the first coordinate position, minus the value of the screening signal-to-noise ratio threshold, as the original signal-to-noise ratio of the first coordinate position.

[0107] In one embodiment, in the step of performing S220: Obtain corresponding multiple original signal-to-noise ratio matrices according to multiple frequency domain data respectively, it includes:

[0108] Perform constant false alarm calculation on the frequency domain data of the first voice part echo to obtain the first original signal-to-noise ratio matrix;

[0109] Perform constant false alarm calculation on the frequency domain data of the second voice part echo to obtain the second original signal-to-noise ratio matrix.

[0110] In one embodiment, in the step of executing S230: combining and accumulating multiple original signal-to-noise ratio matrices into an extended signal-to-noise ratio matrix, it includes:

[0111] Mapping the coordinates of the extended signal-to-noise ratio matrix to multiple original signal-to-noise ratio matrices respectively;

[0112] Interpolating the mapped coordinate positions in multiple original signal-to-noise ratio matrices to obtain an interpolation result;

[0113] Selecting the minimum original signal-to-noise ratio among the interpolation results of the mapped coordinate positions in multiple original signal-to-noise ratio matrices;

[0114] Taking the sum of the signal-to-noise ratio cumulative value at the corresponding coordinate position in the extended signal-to-noise ratio matrix and the minimum original signal-to-noise ratio as the combined signal-to-noise ratio cumulative value.

[0115] In one embodiment, in the step of executing mapping the coordinates of the extended signal-to-noise ratio matrix to multiple original signal-to-noise ratio matrices respectively, it includes:

[0116] Obtaining the serial number of the scan frame where the original signal-to-noise ratio matrix is located in the measurement period, and obtaining the mapped coordinate positions to multiple original signal-to-noise ratio matrices according to the following formula:

[0117] [i,j]=[(Ei*ERres + k*Tf*(Ej–Edz)*EDres) / Rres,((Ej–Edz)*EDres) / Dres)%NDFFT]

[0118] Where, i is the horizontal axis value of the coordinate of the mapped original signal-to-noise ratio matrix; j is the vertical axis value of the coordinate of the mapped original signal-to-noise ratio matrix, Ei is the horizontal axis value of the coordinate of the extended signal-to-noise ratio matrix, Ej is the vertical axis value of the coordinate of the extended signal-to-noise ratio matrix, ERres is the range resolution of the extended signal-to-noise ratio matrix, EDres is the Doppler resolution of the extended signal-to-noise ratio matrix, EDz is the Doppler zero point of the extended signal-to-noise ratio matrix, k is the serial number of the scan frame where the original signal-to-noise ratio matrix is located in the measurement period, Tf is the period duration of the scan frame, Rres is the range resolution of the part where the original signal-to-noise ratio matrix is located, Dres is the Doppler resolution of the part where the original signal-to-noise ratio matrix is located, and NDFFT is the number of Doppler layers of the part where the original signal-to-noise ratio matrix is located.

[0119] In one embodiment, in the step of executing interpolating the mapped coordinate positions in multiple original signal-to-noise ratio matrices to obtain an interpolation result, it includes;

[0120] Obtaining the adjacent integer coordinates [il, jb] where the horizontal axis coordinate in the mapped coordinate [i,j] decreases by no more than one coordinate position and the vertical axis coordinate decreases by no more than one coordinate position, and interpolating the original signal-to-noise ratio at the position of the mapped coordinate [i,j] according to the following formula:

[0121] So[i, j] = So[il, jb] * (il + 1 - i) * (jb + 1 - j) +

[0122] So[il, jb + 1] * (il + 1 - i) * (j - jb) +

[0123] So[il + 1, jb] * (i - il) * (jb + 1 - j) +

[0124] So[il + 1, jb + 1] * (i - il) * (j – jb)

[0125] Among them, So is the original signal-to-noise ratio value at the corresponding coordinate position in the original signal-to-noise ratio matrix.

[0126] On the other hand, the present application also provides a radar wave processing device. Figure 5 It is a block diagram of the radar wave processing device according to an embodiment of the present application.

[0127] As Figure 5 shown, in one embodiment, the radar wave processing device includes a radar receiver 21 and a processor 22 that are connected to each other.

[0128] The radar receiver 21 is configured to receive and send radar echoes to the processor 22.

[0129] The processor 22 is configured to execute the radar wave processing method as described above.

[0130] For example, in one embodiment, the radar general baseband signal processing unit provided by the processor 22 includes a plurality of sub-units such as a range fast Fourier transform (FFT) unit, a Doppler fast Fourier transform unit, and a constant false alarm rate detection (CFAR) unit. The baseband signal processing unit composed of the above sub-units should be executed once for each of the H and L tone parts within each beam, and a raw constant false alarm rate detection signal-to-noise ratio matrix is output through the constant false alarm rate detection sub-unit. Preferably, each of the above units can be implemented using pure hardware, and the entire processing process does not require software intervention. Preferably, although the baseband signal processing device processes the H and L tone parts separately, resources should be saved as much as possible, and time-division multiplexing of the same device instance is considered instead of parallel execution of two duplicate devices.

[0131] The processor 22 sets up an extended range-Doppler (RD) signal-to-noise ratio storage unit, which is a two-dimensional matrix. Each cell in the matrix stores the signal-to-noise ratio corresponding to the range-Doppler coordinates (range value, Doppler value) of that cell. The signal-to-noise ratio reflects the likelihood of a real reflector target at that range-Doppler coordinate. The number of Doppler layers stored in the signal-to-noise ratio storage unit is W times the number of Doppler layers in the original constant false alarm detection signal-to-noise ratio matrix in the baseband signal processing device (usually the same as the number of points in the Doppler fast Fourier transform), where W is a positive number not less than 1, so as to achieve an upward expansion (higher measurement value) of the measurement range of Doppler by W times.

[0132] The processor 22 sets up an extended range-Doppler signal-to-noise ratio calculation unit, which increases or decreases the cumulative signal-to-noise ratio value in the extended range-Doppler signal-to-noise ratio storage unit according to the baseband signal processing results from each part of the duet (here mainly referring to the original constant false alarm detection signal-to-noise ratio). The specific calculation method is described in detail in the signal processing method feature description below. The calculation operation should be executed once when the baseband signal of each part of each beam in each cycle of the three-layer waveform of this application is processed.

[0133] The processor 22 sets up an extended range-Doppler signal-to-noise ratio filtering unit, that is, an extended constant false alarm detection unit. At the beginning of each radar detection cycle, through the constant false alarm detection method, the range-Doppler coordinates corresponding to the extended range-Doppler cells with a cumulative signal-to-noise ratio higher than the target determination signal-to-noise ratio threshold are selected from the extended range-Doppler signal-to-noise ratio matrix as the radar target point cloud of the current detection cycle.

[0134] Finally, the processor 22 screens the radar target point cloud to identify the detected target points.

[0135] In one embodiment, each measurement cycle is equally divided into multiple measurement frames. A complete baseband signal processing is performed for each measurement frame, and each measurement frame needs to update the global extended range-Doppler signal-to-noise ratio matrix according to its baseband signal processing results.

[0136] Before the start of the baseband signal processing of the first measurement frame in each cycle, an extended constant false alarm detection will be performed to filter out the range-Doppler coordinates with a signal-to-noise ratio higher than the set threshold from the currently accumulated extended range-Doppler signal-to-noise ratio matrix. These range-Doppler coordinates will be output as the target points of this measurement cycle and further perform subsequent operations, such as secondary Doppler discrete Fourier transform (DFT) and azimuth angle solution.

[0137] In the last measurement frame of each period, there is still one chance to further obtain a more refined Doppler measurement result with higher Doppler resolution based on a set of pitch observation vector sequences generated for each pitch of all frames in this measurement period. This measurement result will replace the Doppler value of the target extended range-Doppler coordinate obtained from the extended constant false alarm rate (CFAR) detection in the first measurement frame of this period, and the target will be split into multiple ones according to different values of the refined Doppler, thereby achieving a finer Doppler resolution.

[0138] In one embodiment, a preposed window function, such as a Hanning window, can be added to the above-mentioned fast Fourier transform operations at all levels.

[0139] In one embodiment, the above-mentioned CFAR detection algorithm can be added with a preposed digital beamforming (DBF) algorithm, so as to virtually focus the beam in a certain direction, thereby improving the signal-to-noise ratio (SNR) of the signal in the focused direction.

[0140] In one embodiment, in order to obtain a range-Doppler SNR matrix with multiple extended intervals to achieve a higher Doppler measurement value, after the CFAR detection of the H and L pitches in each measurement frame is completed, the original CFAR detection SNR matrices output by the two CFAR detections need to be "merged and accumulated" into each cell of the extended range-Doppler SNR matrix.

[0141] Based on the values of the above-mentioned extended range-Doppler SNR matrix, before the baseband signal processing of the first measurement frame in each period starts, an "extended CFAR detection" will be performed once on the currently accumulated extended range-Doppler SNR matrix, that is, the range-Doppler coordinates with SNR higher than the set extended CFAR detection SNR threshold will be screened out, and these range-Doppler coordinates will be output as the target point cloud of this measurement period.

[0142] In one embodiment, the "merging and accumulating" process for the extended range-Doppler SNR matrix usually takes the lower value of the original CFAR detection SNR interpolation results at the "corresponding positions" (detailed below) of the H and L pitches and accumulates it to the accumulated value of the previous frame of the extended range-Doppler SNR:

[0143] The current frame cumulative value of the extended range-Doppler SNR [Ei, Ej] = the cumulative value at the same position in the previous frame +

[0144] MIN{

[0145] The original CFAR detection SNR [H mapping (Ei, Ej)],

[0146] The original CFAR detection SNR [L mapping (Ei, Ej)]

[0147] }

[0148] For example, when processing the constant false alarm detection result of the j-th frame in the i-th measurement period, the original constant false alarm detection signal-to-noise ratio interpolation result of the H part corresponding to a certain extended range-Doppler cell is 123, and the original constant false alarm detection signal-to-noise ratio interpolation result of the L part is -456. At this time, the accumulated extended range-Doppler signal-to-noise ratio of this cell is 789. Then, after this update, the new accumulated signal-to-noise ratio value of this cell is: 789 + min(123, -456) = 333

[0149] In one embodiment, the calculation method of the original constant false alarm detection signal-to-noise ratio matrix is as follows: Assume that during the execution of the constant false alarm detection algorithm, the logarithmic signal intensity of a certain cell is S, the logarithmic noise intensity statistically obtained according to the intensity distribution of the surrounding cells is No, and the signal-to-noise ratio threshold used for screening and generating the target by the constant false alarm detection is T. Then, the original constant false alarm detection signal-to-noise ratio at this position is denoted as:

[0150] Original constant false alarm detection signal-to-noise ratio = S – No - T

[0151] That is to say, for a cell that can just be screened as the original constant false alarm detection target, the obtained original signal-to-noise ratio value is exactly 0; the signal-to-noise ratio of a higher-quality target is higher than 0; and the signal-to-noise ratio of a cell that is not sufficient to be an original target is lower than 0.

[0152] Regarding the noise calculation method for each cell in the constant false alarm detection, it belongs to the well-known field, and this application will not elaborate. In one embodiment, in order to improve the operation efficiency of engineering implementation, the constant false alarm detection noise calculation method supported by the radar hardware platform (such as an SoC chip) should be considered, such as simple CFAR-CA (including CFAR-CASO or CFAR-CAGO), or better-performing CFAR-OS, etc. Generally speaking, in the implementation of most constant false alarm detection algorithms, the noise at a certain cell is statistically obtained (average value, cumulative probability value, etc.) from the signal values of several neighboring cells around it, which usually represents the background signal intensity within the neighborhood range of this cell, that is, the noise intensity.

[0153] Due to the difference in Doppler resolution between H and L and the influence brought by the target moving speed, the "corresponding position" of the extended range-Doppler coordinate (Ei, Ej) in the coordinate mapping method is often a non-integer original range-Doppler coordinate. Therefore, the "original constant false alarm detection signal-to-noise ratio at the corresponding position" is usually interpolated from the original constant false alarm detection signal-to-noise ratio matrix according to the non-integer original range-Doppler coordinate.

[0154] In one embodiment, the signal-to-noise ratio interpolation method is as follows:

[0155] Assume that the cell coordinates of an extended range-Doppler SNR matrix are (Ei, Ej). After mapping this coordinate to the original constant false alarm detection SNR matrix corresponding to the H pitch, the H coordinate is (ih, jh). There are a total of 4 integer range-Doppler coordinates that are closely adjacent to this non-integer coordinate. Denote the one with the lowest bottom left (both R and D are the smallest) among these 4 coordinates as [ihl, jhb]. Then, based on the original SNRs of these 4 neighboring cells, interpolation can be performed to obtain the original constant false alarm detection SNR at (ih, jh) in the H pitch as follows:

[0156] Original constant false alarm detection SNR [ih, jh] =

[0157] Original constant false alarm detection SNR [ihl, jhb] * (ihl + 1 - ih) * (jhb + 1 - jh) +

[0158] Original constant false alarm detection SNR [ihl, jhb + 1] * (ihl + 1 - ih) * (jh - jhb) +

[0159] Original constant false alarm detection SNR [ihl + 1, jhb] * (ih - ihl) * (jhb + 1 - jh) +

[0160] Original constant false alarm detection SNR [ihl + 1, jhb + 1] * (ih - ihl) * (jh - jhb)

[0161] Similarly, the coordinate mapping method for the L pitch is the same as that for the H pitch, and will not be elaborated here.

[0162] For example, for a cell (23, 4) in an extended range-Doppler SNR matrix, after mapping to the L pitch, the coordinate is (23.1, 3.6). Then, its original constant false alarm detection SNR in the L pitch can be obtained through the above interpolation method:

[0163] L pitch original constant false alarm detection SNR [23.1, 3.6] =

[0164] L pitch original constant false alarm detection SNR [23, 3] * 0.9 * 0.4 +

[0165] L pitch original constant false alarm detection SNR [23, 4] * 0.9 * 0.6 +

[0166] L pitch original constant false alarm detection SNR [24, 3] * 0.1 * 0.4 +

[0167] L pitch original constant false alarm detection SNR [24, 4] * 0.1 * 0.6

[0168] In one embodiment, the mapping strategy from the "extended range-Doppler SNR matrix" to the "original constant false alarm detection SNR matrix" is as follows:

[0169] It is necessary to perform multiple consecutive non - overlapping expansions on the Doppler interval represented by constant false alarm rate (CFAR) detection. This is also one of the core operations of this application, aiming to expand the measurement range of Doppler. For example, if the original Doppler interval represented by CFAR detection has 32 layers (corresponding to 32 output spectral lines of Doppler fast Fourier transform), after an 8 - fold expansion, the expanded range - Doppler matrix can reach 256 layers. Considering the aliasing characteristics of Doppler fast Fourier transform, the Doppler spectral line coordinate j in the expanded range - Doppler signal - to - noise ratio matrix after expansion needs to be periodically circularly mapped to the original effective expression range of CFAR detection. In the above example, multiple expanded Doppler coordinates such as expanded Doppler coordinate Ej = 2, expanded Doppler coordinate Ej = 34, expanded Doppler coordinate Ej = 66, and expanded Doppler coordinate Ej = 98 are all mapped to the same original Doppler coordinate j = 2 of CFAR detection.

[0170] Considering that the target is moving, at the start of multiple different frames in each measurement period, the mapping relationship should be synchronized with the moving speed of the target. Only in this way can the original CFAR detection signal - to - noise ratios between multiple different frames be reasonably superimposed at the same range position. Otherwise, the misaligned superposition cannot accumulate the signal - to - noise ratio at the real target to a relatively high value.

[0171] Based on the above strategy, the following coordinate mapping method can be obtained:

[0172] Suppose in the 0th frame of a certain measurement period, the cell coordinate of the expanded range - Doppler signal - to - noise ratio matrix corresponding to a certain real target is [Ei, Ej], the range resolution of the entire expanded range - Doppler signal - to - noise ratio matrix is ERres, the Doppler resolution is EDres, and the Doppler zero point (the Doppler coordinate of the expanded range - Doppler cell representing zero speed) is EDz.

[0173] At the same time, suppose the number of Doppler layers (the number of spectral lines of Doppler fast Fourier transform) of the instantaneous CFAR detection of a certain tone (H or L) is NDFFT, the range resolution is Rres, the Doppler resolution is Dres, and the time interval (frame duration) between two adjacent measurement frames is Tf. Then:

[0174] In the 0th frame of each measurement period, the physical coordinates corresponding to the expanded coordinates [Ei, Ej] are: [Ei * ERres, (Ej – Edz) * EDres]

[0175] In the kth frame of each measurement period, due to target movement (it can be assumed to be uniform radial movement in a short time between frames), the range value in the physical coordinates changes to:

[0176] Ei * ERres + k * Tf * (Ej – Edz) * EDres

[0177] At the k-th frame, according to the range resolution and Doppler resolution of the original constant false alarm detection signal-to-noise ratio matrix, the original cell coordinates [i, j] are as follows:

[0178] [i, j] =

[0179] (Ei * ERres + k * Tf * (Ej – Edz) * EDres) / Rres,

[0180] ((Ej – Edz) * EDres) / Dres) % NDFFT

[0182] In the above formula, if the original range coordinate after mapping exceeds the effective range measurement range, it means that the target has moved outside the horizon at the k-th frame and should be regarded as an invalid mapping. The corresponding instantaneous signal-to-noise ratio should be set to an extremely low value to ensure that such targets will not obtain sufficient values in the final extended range-Doppler signal-to-noise ratio matrix to generate false targets.

[0183] The operation of taking the remainder of the mapped original Doppler coordinate by the number of spectral lines in the above formula is designed according to the aliasing characteristics of the Doppler fast Fourier transform. Its aliasing characteristics determine that the k-th original spectral line beyond the measurement range will be aliased to the position of the k % NDFFT-th original spectral line. For example, when NDFFT = 16, the 18th, 34th, 66th, and 98th original spectral lines that cannot be directly measured will all be aliased to the 2nd original spectral line, and the 2nd original spectral line is measurable.

[0184] The Doppler resolutions of each voice part are not in an integer multiple relationship. The key to upwardly expanding the Doppler measurement range in this application lies in achieving two slightly different Doppler aliasing ranges through the slightly different Doppler resolutions of the H and L voice parts. Furthermore, when mapping the same extended Doppler coordinate Ej to H and L, two different original constant false alarm detection measurable Doppler coordinates will be obtained: if the target truly exists, sufficient high measurement signal-to-noise ratios should be shown at both of these two original constant false alarm detection Doppler coordinates; otherwise, at least one of the measurement signal-to-noise ratios corresponding to the constant false alarm detection will be relatively low. By taking the lower of the two original constant false alarm detection signal-to-noise ratios to participate in the accumulation of the extended range-Doppler signal-to-noise ratio matrix, such non-existent targets can be excluded at the stage of finally performing extended constant false alarm detection (screening targets according to the extended range-Doppler signal-to-noise ratio).

[0185] In the above embodiment, through the multi-layer extended range-Doppler signal-to-noise ratio matrix, in cooperation with the H + L duet waveform to resist Doppler aliasing, the Doppler measurement range is expanded upward.

[0186] ​It should be noted that multiple tones within a beam are transmitted closely adjacent to each other, and there is no limitation on whether the highest (finest) resolution tone is in the front or the back.

[0187] In other embodiments, the design of upgrading the two tones H and L to multiple tones can improve the extended signal-to-noise ratio at the cost of increased time overhead and transmission power consumption. In essence, it is similar to the method of performing Doppler expansion with two tones listed in this application to illustrate the design principle, and it is still within the protection scope of this application.

[0188] In one embodiment, this application also provides a computer storage medium.

[0189] A computer program is stored on the computer storage medium. When the computer program is executed by a processor, it can implement the radar wave processing method as described above.

[0190] The radar wave processing method, device, and computer storage medium provided by this application can effectively improve the dynamic measurement range of the product and effectively increase the maximum measurement value by setting multiple tones with different Doppler resolutions in the radar beam.

[0191] On the other hand, this application also provides a method for processing fine Doppler velocity. Figure 6 It is a flowchart of the method for processing fine Doppler velocity according to an embodiment of this application.

[0192] As Figure 6 shown, in one embodiment, the method for processing fine Doppler velocity includes:

[0193] S310: Perform Fourier transform of the baseband signals on the echoes of multiple tones in the radar beam including the first beam to screen for target points. The Fourier transform includes the first Doppler Fourier transform.

[0194] Perform frequency domain conversion on the echoes of multiple tones of the radar, perform the first Doppler Fourier transform, and screen out the cell coordinates that may contain multiple targets with similar velocities.

[0195] S320: Obtain multiple tone observation vectors of the target points according to the first Doppler Fourier transform.

[0196] Calculate and store the instantaneous observation vectors of each tone at the coordinates of each target point.

[0197] S330: Extract the multiple tone observation vectors in each scan frame within at least one measurement period of the first beam, and form a tone observation vector sequence in the time domain order.

[0198] A beam contains multiple voice parts, and each beam is distributed within each scan frame. By extracting the multiple voice part observation vectors of the same beam within each scan frame, a virtual sampling sequence in the time domain can be formed in sequence.

[0199] S340: Perform a second Doppler Fourier transform on the voice part observation vector sequence to obtain amplitude distribution data.

[0200] Perform a second Doppler Fourier transform on the virtual sampling sequence to obtain the amplitude distribution in the frequency domain.

[0201] S350: Screen the sub-targets in the target points according to the amplitude distribution data.

[0202] Through an appropriate screening algorithm, sub-targets with similar speeds in the same target point can be screened out.

[0203] By calculating the second Doppler Fourier transform, a sub-Doppler spectrum can be obtained, which can distinguish multiple targets with close speeds that cannot be distinguished by the first Fourier transform and are located within the same output spectral line of the first Doppler Fourier transform.

[0204] In one embodiment, when performing S310: Perform a Fourier transform of the baseband signal on the echoes of multiple voice parts in the radar beam including the first beam respectively to screen the target points, the steps of the Fourier transform including the first Doppler Fourier transform include:

[0205] Obtain an extended signal-to-noise ratio matrix according to the echoes of multiple voice parts;

[0206] Before the first scan frame of the measurement period, perform a constant false alarm calculation on the extended signal-to-noise ratio matrix to screen out the target points.

[0207] As in the above embodiment, screening the target points through the extended signal-to-noise ratio matrix can expand the measurement range of the Doppler speed and screen out the target points for the sub-second Doppler Fourier transform.

[0208] In one embodiment, when performing S320: According to the first Doppler Fourier transform, obtain the steps of the multiple voice part observation vectors of the target points include:

[0209] Map the target point to the original signal-to-noise ratio matrix to obtain the original Doppler coordinates of the current voice part; obtain the original complex vector of the first Doppler Fourier transform of the original Doppler coordinates; obtain the estimated azimuth angle of the target point; according to the estimated azimuth angle, perform antenna vector merging with phase retention on the original complex vector to obtain the voice part observation vector of the current voice part.

[0210] In one embodiment, when performing the step of obtaining the original complex vector of the first Doppler Fourier transform at the original Doppler coordinates, it includes:

[0211] When the original Doppler coordinate is a non-integer coordinate, obtain the first complex vector of the smaller integer coordinate and the second complex vector of the larger integer coordinate among the two nearest non-integer coordinates;

[0212] Interpolate and combine the first complex vector and the second complex vector to obtain the original complex vector of the non-integer coordinate.

[0213] For example, in one embodiment, for a target from a cell of an extended range-Doppler signal-to-noise ratio matrix (Ei, Ej), if a second Doppler discrete Fourier transform is to be performed to subdivide the Doppler value of the target and generate multiple subdivided Doppler targets (if any), first, a sequence of "phoneme observation vectors" needs to be constructed, and the "phoneme observation vectors" of the first Doppler fast Fourier transform results from each phoneme in the sequence can be generated by the following method:

[0214] First, after the first Doppler fast Fourier transform of each phoneme is completed, map the (Ei, Ej) extended range-Doppler coordinates to the original coordinates of the original complex vector matrix of the first Doppler fast Fourier transform of the phoneme according to the velocity of the extended range-Doppler target and the Doppler resolution of the current phoneme, as described above:

[0215] {H|L map (Ei), ((Ej – Edz) * EDres) / Dres) % NDFFT}

[0216] In engineering implementation, to simplify the calculation, in many cases, the range resolution of the extended range-Doppler signal-to-noise ratio matrix is the same as the range resolution of each phoneme. Therefore, the original range coordinate i after mapping is exactly the same as Ei before mapping. If the two resolutions are different, generally, a floating-point conversion (round) operation can be used to approximate to the integer coordinate.

[0217] For the Doppler dimension coordinate, as described above, the original Doppler coordinate mapped to the current phoneme is:

[0218] Dj = ((Ej – Edz) * EDres) / Dres) % NDFFT

[0219] Since the above coordinates are generally not integer values, and the original complex vector matrix of the first Doppler fast Fourier transform can only take values at integer cells, the two nearest integer coordinates of this non-integer coordinate can be taken (i.e., floor(Dj) and floor(Dj) + 1). Then, at these two nearest integer coordinates, two groups of complex vectors composed of receiving antennas corresponding roughly to this non-integer coordinate are taken out from the original complex vector matrix. The group with Doppler slightly higher than Dj is denoted as RXt, and the group with Doppler slightly lower than Dj is denoted as RXb. For a radar with NRX receiving antennas, there is:

[0220] RXt = First Doppler Fast Fourier Transform[i, floor(Dj)+1] = {RXt(1), RXt(2), …, RXt(NRX)}

[0221] RXb = First Doppler Fast Fourier Transform[i, floor(Dj)] = {RXb(1), RXb(2), …, RXb(NRX)}

[0222] Then, according to the actual non-integer value of j, the original complex vectors of the corresponding receiving antennas of RXt and RXb are interpolated and combined to obtain the original complex vector RX of the First Doppler Fast Fourier Transform that exactly matches j:

[0223] RX = {interpolation(RXt(1), RXb(1)), interpolation(RXt(2), RXb(2)), …, interpolation(RXt(N), RXb(NRX))}

[0224] Finally, according to the estimated azimuth angle (DoA) of the target obtained in other units of the radar baseband signal processing device, perform coherent digital beamforming, that is, the antenna vector merging operation that preserves the phase, and merge the RX vector into a single total vector, and this total vector will be used as the "bin observation vector" V of the current bin:

[0225] V = Σ(RX(g) * Twiddle(g))

[0226] Where the rotation vector (Twiddle) of the digital beamforming is calculated according to the positions of the receiving antennas of the radar device. For the receiving antenna g with an absolute position of p(g) (in units of λ), when performing beamforming in the DoA direction, its rotation vector is:

[0227] Twiddle(DoA, g) = exp(sin(DoA) * 2 * Π * p(g))

[0228] In one embodiment, in the step of performing interpolation and combination of the first complex vector and the second complex vector to obtain the original complex vector at non-integer coordinates, the exact Fast Fourier Transform output vector at j can be obtained by interpolating according to the Fast Fourier Transform output vectors of the two closest cells of RXt and RXb.

[0229] The original complex vector RX(g) at the non-integer spectral line position j obtained by interpolating the first complex vector RXb(g) and the second complex vector RXt(g) of the receiving antenna g is:

[0230] β = |RXb(g)| / |RXt(g)|

[0231] α = 1 / (1 + β)

[0232] A = |RXb(g)| * Π * α / sin(Π * α)

[0233] P = ∠RXb(g) - Π * α

[0234] After obtaining A and P, calculate RX(g) according to the following trigonometric function formula:

[0235] RX(g) = A * cos(P) + A * sin(P) * J

[0236] Where g is the serial number of the receiving antenna, and the value is a positive integer greater than 0, RX(g) is the original complex vector, RXb(g) is the first complex vector, RXt(g) is the second complex vector, and J is the unit complex vector.

[0237] Adopting a mature engineering algorithm can have sufficient selectivity during the signal processing process, improve the efficiency of radar design and maintenance, and significantly reduce costs.

[0238] In one embodiment, in the step of performing antenna vector merging with phase retention on the original complex vector according to the estimated azimuth angle to obtain the pitch observation vector of the current pitch, it includes:

[0239] Calculate the pitch observation vector according to the following formula:

[0240] V = Σ(RX(g) * exp(sin(DoA) * 2 * Π * p(g)))

[0241] Where V is the pitch observation vector, g is the positive integer serial number of the receiving antenna, and the value is a positive integer greater than 1, RX(g) is the original complex vector, DoA is the estimated azimuth angle, Π is the pi, and p(g) is the absolute position of the receiving antenna.

[0242] In one embodiment, in the step of performing S330: extracting multiple pitch observation vectors in each scan frame of the first beam in at least one measurement period and forming a pitch observation vector sequence in the time domain, it includes:

[0243] Obtain the pitch observation vector of the first pitch and the pitch observation vector of the second pitch in each scan frame of at least one measurement period;

[0244] According to the pitch observation vector of the first pitch and the pitch observation vector of the second pitch, form a pitch observation vector sequence in the following formula order in the time domain:

[0245] V(Z) = [VH(1), VL(1), VH(2), VL(2), ……, VH(Z), VL(Z)]

[0246] Wherein, Z is a positive integer greater than 1, V(Z) is a sequence of pitch observation vectors, VH(Z) is the pitch observation vector of the first pitch, and VL(Z) is the pitch observation vector of the second pitch.

[0247] Whether performing the first Doppler fast Fourier transform or the second Doppler discrete Fourier transform, effective sampling points are required to construct a virtual "time domain" sampling value sequence, and then the Fourier transform is performed to complete the mapping from the "time domain" to the "frequency domain". The obtained virtual "frequency domain" data is the Doppler distribution information. From the information in the public domain, for the first Doppler fast Fourier transform, its sampling point sequence is a sequence composed of complex vectors of each ramp in the fast Fourier transform on the same range spectrum line. This sequence represents the initial phase of the target on this range spectrum line at the start time of each ramp.

[0248] Figure 7 It is a schematic diagram of sampling the sequence of pitch observation vectors according to an embodiment of the present application.

[0249] Please refer to Figure 7 , the present application proposes a similar method for constructing a virtual sampling sequence of the second Doppler discrete Fourier transform, that is, after the first Doppler fast Fourier transform of each pitch in the duet is completed, digital beamforming is performed according to the estimated azimuth angle where the target point is located to obtain the total pitch observation complex vector that combines all receiving antennas. This complex vector represents the initial phase of the target at the start time of the current pitch. By storing the "pitch observation vectors" of each pitch of the target within the measurement period, a virtual sampling sequence with an unequal time distribution can be constructed for the second Doppler discrete Fourier transform.

[0250] In one embodiment, in the step of performing S340: performing a second Doppler Fourier transform on the sequence of pitch observation vectors to obtain amplitude distribution data, it includes:

[0251] According to the sequence of pitch observation vectors, obtain the observation time corresponding to each value in the sequence of pitch observation vectors;

[0252] According to the observation time, obtain the rotation vector of the discrete Fourier transform corresponding to each sample in each spectrum line in the second Doppler Fourier transform;

[0253] According to the rotation vector of the discrete Fourier transform and the sequence of pitch observation vectors, obtain the amplitude distribution data of the second Doppler Fourier transform.

[0254] For example, please refer to Figure 7 and Figure 3 simultaneously. Each radar detection cycle (Cycle) is equally divided into multiple scan frames (Frame#1, Frame#2...) in time.

[0255] Multiple beams (Duo Beam#1, Duo Beam#2, Duo Beam#3, ……) with different directions, different range resolutions, and different Doppler resolutions are emitted within each scan frame.

[0256] Each beam is a kind of duo chirp group, which at least includes two chirp groups with high resolution (H part) and low resolution (L part).

[0257] Since the above "part observation vector" sequence is non-equidistant sampling, the fast Fourier transform algorithm cannot be used to implement it, and the discrete Fourier transform needs to be used. Generally, it is considered that the fast Fourier transform with a time complexity of 2NLog2(N) is faster than the discrete Fourier transform with a time complexity of N*N. However, in engineering implementation, the level NDDFT2 of the quadratic Doppler discrete Fourier transform is usually not too many, so the execution time is within an acceptable range. On the contrary, the discrete Fourier transform has certain advantages compared with the fast Fourier transform. The discrete Fourier transform is pipeline-operable and can therefore not be truncated by a measurement window of a fixed length. With this feature, the quadratic Doppler discrete Fourier transform of this application can perform target tracking across measurement periods, calculate the energy distribution of each sub-Doppler level infinitely in a longer time dimension, and thus obtain a result with higher resolution and greater stability. In the following, in order to elaborate on the method of this application, the calculation process of the quadratic Doppler discrete Fourier transform is still constrained within a measurement period:

[0258] Assume that a measurement period of duration Tc contains Z measurement frames. Among them, the part observation vectors of the H and L parts of a certain beam in the i-th (1≤i≤Z) measurement frame are VHi and VLi. The measurement duration of each H part is Th, and the L part follows immediately. Then, for the following part observation vector sequence:

[0259] V(Z) = [VH(1), VL(1), VH(2), VL(2), ……, VH(Z), VL(Z)]

[0260] The observation time corresponding to the i-th value among them (starting from the start time of the measurement period) is:

[0261] t(i) = Tc / Z * (i / 2) + Th * (i%2)

[0262] That is to say, the minimum time interval between part observation vectors is Th (because the L part follows the H part within the same frame), and the periodic time interval that the entire observation vector sequence can cover is Tc (i.e., NDDFT * frame duration Tf). Then, according to the definition of radar Doppler resolution, the Doppler measurement ability in this case can be known:

[0263] The finest resolution Vres = λ / (2*Tc)

[0264] The maximum measured value Vmax = λ / (2*Th)

[0265] Taking the finest resolution Vres as the output spectral line graduation value of the second - order Doppler discrete Fourier transform, the total number of output spectral lines can reach:

[0266] NDDFT = Vmax / Vres = Tc / Th

[0267] Among all the spectral lines and the output spectral lines of the second - order discrete Fourier transform, the discrete Fourier transform rotation vector corresponding to the i - th sample of the k - th spectral line is:

[0268] Twiddle(k,i) = exp(-1j*k*t(i)*(2*Π / Tc))

[0269] Then, according to the mathematical meaning of the discrete Fourier transform, the calculation method of the amplitude distribution of the second - order Doppler discrete Fourier transform corresponding to this application can be obtained:

[0270] DFT(k) = |Σ(Twiddle(k,i)*V(i))|

[0271] Among the above - mentioned output spectral lines of the second - order Doppler discrete Fourier transform, each local maximum may be a value result of a sub - divided Doppler. Assuming that the current voice observation vector sequence is interpolated from the j - th spectral line of the first - order Doppler fast Fourier transform, if the amplitude of the k - th output spectral line of the second - order Doppler discrete Fourier transform is a local maximum, then the final sub - divided Doppler velocity corresponding to this local maximum is:

[0272] Vd = λ / (2*Tc)*j + λ / (2*Th)*k

[0273] Where λ is the central wavelength. In any case, the Doppler resolution is λ / (2 * total observation duration). The reason why the resolution of the first Doppler fast Fourier transform is insufficient is that the continuous observation duration is too short (one frame duration Th or Tl). The second Doppler extends this observation duration to one period Tc. Based on the method proposed in this application, after the second Doppler discrete Fourier transform, the Doppler resolution can be successfully refined from the resolution of the first Doppler fast Fourier transform (the smaller of λ / (2*Th) and λ / (2*Tl)) to λ / (2*Tc). Moreover, a special feature of this application is that since there is a seamless connection between the H and L voices, the upper limit of the Doppler measurement of the second Doppler discrete Fourier transform is also seamlessly connected to the lower limit (i.e., the resolution) of the Doppler measurement of the first Doppler fast Fourier transform. The Doppler value at the connection is λ / (2*Tc)*j, where j is the output spectral line number of the first Doppler fast Fourier transform. In this way, the downward expansion of the Doppler measurement range is achieved completely and without continuous blind spots through the method proposed in this application.

[0274] In the above embodiment, after obtaining the initial phase of the signal through the first Doppler measurement for each voice of multiple frames of the same beam within the measurement period, the initial phase is used as a virtual time-domain sampling for the second Doppler Fourier transform. Different from the conventional second Doppler method where there is only one virtual sampling per period, here due to the special design of the duet waveform, there is continuity between H and L. Therefore, the maximum measurement speed of the second Doppler discrete Fourier transform is exactly equal to the resolution of the first Doppler measurement. That is to say, there is a seamless connection between the measurable range of the first Doppler and the measurable range of the second Doppler, and there is basically no measurement blind spot.

[0275] It is precisely because the time intervals between virtual samplings are not exactly the same that the discrete Fourier transform is used in this application instead of the fast Fourier transform. Compared with the fast Fourier transform, the discrete Fourier transform is pipelined and can therefore not be truncated by a measurement window of a fixed length. It can span the measurement period and measure in a longer time dimension to obtain more stable and reliable results.

[0276] In one embodiment, after performing step S350: screening the sub-targets in the target points according to the amplitude distribution data, it includes:

[0277] Obtain the maximum output spectral line number of the second Doppler according to the minimum measurement duration among all voices and the total duration of at least one measurement period;

[0278] Select multiple local maxima formed by sidelobe interference of the same real target from the output amplitudes of the sub-targets according to the maximum output spectral line number and the period duration of the scanning frame;

[0279] Select the spectral line position corresponding to the highest amplitude among multiple local maxima as the true target.

[0280] Since the observation vector sequence used in the quadratic Doppler discrete Fourier transform is non-equidistant sampling, it is necessary to suppress sidelobe interference when necessary, which can avoid the generation of false local maxima by these sidelobes and prevent the radar from generating false targets with incorrect Doppler velocities.

[0281] For example, in one embodiment, when the duration of the H note Th / the duration of the measurement frame Tf = 1 / m, there are m aliased intervals in the output range of each quadratic Doppler discrete Fourier transform. That is to say, a target with a velocity of Vd located at the k-th output spectral line of the quadratic Doppler discrete Fourier transform will cause a total of m local maxima in the entire output range of the quadratic Doppler discrete Fourier transform. The spectral line positions of these local maxima are respectively:

[0282] [k, k + NDDFT / m, k + 2 * NDDFT / m, k + 3 * NDDFT / m, …, k + floor(m) * NDDFT / m]

[0283] Or rather, a target with a velocity of Vd located at any one of the output spectral lines in [k, k + NDDFT / m, k + 2 * NDDFT / m, k + 3 * NDDFT / m, …, k + floor(m) * NDDFT / m] of the quadratic Doppler DFT will cause a total of m local maxima in the entire output range of the quadratic Doppler discrete Fourier transform. The spectral line positions of these local maxima are respectively:

[0284] [k, k + NDDFT / m, k + 2 * NDDFT / m, k + 3 * NDDFT / m, …, k + floor(m) * NDDFT / m]

[0285] While proposing the non-equidistant sampling quadratic Doppler discrete Fourier transform method, this application also proposes a corresponding method to avoid sidelobe interference to achieve the anti-aliasing effect of the quadratic Doppler discrete Fourier transform. Theoretical analysis can prove that among these multiple local maxima caused by the aliasing effect, the amplitude of the k-th spectral line where the true target is located is the highest, and the amplitudes at other positions are lower than that of the k-th spectral line. Therefore, this application proposes a method of only selecting the one with the highest amplitude from the above multiple mutually aliased local maxima as the sole representative of these mutually aliased local maxima for output, so as to avoid the generation of false targets due to aliasing.

[0286] It should be noted that this method of suppressing aliasing by selecting one from multiple options will not significantly affect the fine Doppler resolution between measurable multiple targets in engineering implementation. As long as multiple fine targets do not exactly fall on one of the m aliased spectral lines of each other, they can be output as independent "local maximum" true targets in the amplitude-frequency distribution of the second Doppler discrete Fourier transform without being affected at all. For example, when the radar waveform duty cycle is about 67%, if a duet waveform is used, at this time Th and Tl each account for about 33% of the duration Tf of a single measurement frame, that is, Th / Tf = 1 / 3. If each measurement period contains NF = 32 measurement frames, the total number of output spectral lines of the second Doppler discrete Fourier transform of this method is 96. For each output true target, there are only Th / Tf = 3 (including itself) positions where it is aliased, and the blind zone spectral lines are only (3 - 1) / 96, which is only about 2%, much lower than the traditional method of connecting multiple equally timed sampled measurement frames together for fast Fourier transform (in the equally timed sampling method, the blind zone is at least the waveform idle time ratio, that is, about 33%).

[0287] On the other hand, the present application also provides a device for processing fine Doppler velocity.

[0288] Please refer to Figure 5 , in an embodiment, the device for processing fine Doppler velocity includes a radar receiver 21 and a processor 22 connected to each other, where:

[0289] The radar receiver 21 is configured to receive and send radar echoes to the processor 22; the processor 22 is configured to execute the method for processing fine Doppler velocity as described above.

[0290] It should be noted that based on some calculation formulas of the present application, using some common optimization methods for transformation, such as fixed-point operation, increasing or decreasing the order of an approximate polynomial, polynomial combination based on multiplication and addition operations, etc., are all within the protection scope of the present application.

[0291] The method and device for processing fine Doppler velocity provided by the present application can refine the minimum measurement resolution by setting multiple voices with different Doppler resolutions in the radar beam and performing the first Doppler Fourier transform and the second Doppler Fourier transform.

[0292] In order to implement the foregoing radar waveform, in one aspect of the present application, a radar waveform generation circuit is also provided. Figure 8 It is a block diagram of the radar waveform generation circuit according to an embodiment of the present application.

[0293] As Figure 8 shown, in an embodiment, the radar waveform generation circuit includes a gate 30, a first memory 10, a second memory 20, a first counter 40, a second counter 50, and a frequency synthesizer 60.

[0294] The strobe 30 is respectively connected to the first memory 10 and the second memory 20, wherein the first memory 10 stores data of a first time interval, and the second memory 20 stores data of a second time interval. The strobe 30 is also connected to the first counter 40 and the second counter 50, and under the signal control of the first counter 40, the selected memory data is sent to the second counter 50.

[0295] The second counter 60 is connected to the first counter 40 and the second counter 50. The second counter 60 generates a set waveform of a frequency modulated continuous wave according to the first enable signal sent by the second counter 50, and sends a waveform completion signal to the first counter 40 and the second counter 50 when the set waveform is completed.

[0296] The first counter 40 is used to count the number of waveform completion signals, and sends a strobe conversion signal to the strobe 30 when the number counting is completed.

[0297] The strobe 30 is used to change the strobe from the first time interval to the second time interval according to the strobe conversion signal, so as to send the second time interval as time interval data to the second counter 50.

[0298] The second counter 50 is used to read the time interval data according to the waveform completion signal, perform time counting according to the time interval data, and send a second enable signal to the second counter 60 when the time counting is completed.

[0299] By setting different time intervals between two sets of set waveforms, different Doppler resolutions can be achieved between the two sets of waveforms. Through further processing of the echo, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

[0300] Figure 9 It is a circuit connection diagram of the strobe in an embodiment of the present application.

[0301] As Figure 9 shown, in an embodiment, the strobe 30 uses the chip 74LS157D, and the first memory 10 and the second memory 20 respectively store four-bit time interval data. The strobe 30 includes a first group of data input pins AI, a second group of data input pins BI, a selection pin A / B, and a data output pin Y.

[0302] Please refer to Figure 9 , the first group of data input pins AI is connected to the first memory 10, and the second group of data input pins BI is connected to the second memory 20. The selection pin A / B is connected to the first counter 40, and the data output pin YO is connected to the second counter 50.

[0303] Under the strobe control of the first counter 40, the strobe 30 strobes four-bit time gap data and sends it to the second counter 50 for time counting, thereby controlling the required time gap between two waveforms. In other embodiments, time gap data of other bit numbers can also be selected as needed, for example, it can be eight-bit or sixteen-bit, etc. Correspondingly, the strobe 30 can also use chips of other specifications.

[0304] Figure 10 It is the circuit connection diagram of the first counter in an embodiment of the present application.

[0305] As Figure 10 shown, in an embodiment, the first counter 40 includes a first counting chip 41, a second counting chip 42, and a first NOT gate M1. Among them, the counter chip uses 74LS161D.

[0306] The clock input terminals CP of the first counting chip 41 and the second counting chip 42 are respectively connected to the frequency synthesizer 50. The carry output pin RCO of the first counting chip 41 is connected to the counting control pin EP / ET of the second counting chip 42, and the counting control pin EP / ET and the clear pin CR of the first counting chip 41 are connected to the high-level terminal VCC. The lowest-bit data output pin Q0 of the second counting chip 42 is connected to the strobe 30, and the first NOT gate M1 is connected in series between the second-lowest-bit data output pin Q1 of the second counting chip 42 and the clear pin CR. In this embodiment, the preset control pin LD of the first counting chip 41 is connected to the clear pin CR, the preset control pin LD of the second counting chip 42 is connected to the clear pin CR, and the data input pins D0, D1, D2, D3 of the first counting chip 41 and the data input pins D0, D1, D2, D3 of the second counting chip 42 are all grounded.

[0307] In this embodiment, the first counting chip 41 is configured as a counter of a first specific number system (such as hexadecimal). Whenever it receives the first specific number of waveform completion signals sent by the frequency synthesizer 60, it generates a carry signal and sends it to the second counting chip 42 for enabling. The second counting chip 42 is configured as a second specific number system counter. When enabled, every time it receives the second specific number (such as 1 time) of waveform completion signals sent by the frequency synthesizer 60, it issues a strobe conversion signal to strobe the specified memory. It should be noted that in order to concisely explain the waveform implementation principle of the present application, a simplified circuit diagram is given in this embodiment. Based on the principle described in the present application, further conventional transformations of the circuit are within the protection scope of the present application. For example, in other embodiments, counters of other digit systems can be set according to the needs of different numbers of waveforms.

[0308] Figure 11 It is the circuit connection diagram of the frequency synthesizer in an embodiment of the present application.

[0309] As shown Figure 11 in FIG. 1, in one embodiment, the frequency synthesizer 60 includes a second AND gate M2, a clock source T1, and a frequency synthesis chip 61, where the frequency synthesis chip 61 uses ADF4169.

[0310] Please refer to Figure 11 . The first input terminal of the second AND gate M2 is connected to the clock source T1, the second input terminal of the second AND gate M2 is connected to the second counter 50 to receive an enable signal, and the output terminal of the second AND gate M2 is connected to the step signal input terminal STEP of the frequency synthesis chip 61.

[0311] Through the control of the enable signal, the clock signal sent by the clock source T1 can be sent to the frequency synthesis chip 61 within a set time, so as to achieve the purpose of setting a specific time interval.

[0312] Please continue to refer to Figure 11 . In one embodiment, the second counter 50 includes a NOT gate array T2, a first AND gate M3, a third counting chip 51, a second NOT gate M4, and a third NOT gate M5.

[0313] Among them, the NOT gate array T2 is connected between the data input pins D0, D1, D2, D3 of the third counting chip 51 and the selector 30, the first AND gate M3 is connected between the data output pins Q0, Q1, Q2, Q3 of the third counting chip 51 and the input terminal of the second NOT gate M4, and the output terminal of the first AND gate M3 is also connected to the frequency synthesizer 60 to output an enable signal. The output terminal of the second NOT gate M4 is connected to the counting control pin EP / ET of the third counting chip 51.

[0314] The output terminal of the third NOT gate M5 is connected to the preset control pin LD of the third counting chip 51, the clear pin CR of the third counting chip 51 is connected to the high-level terminal VCC. The input terminal of the third NOT gate M5 is connected to the waveform completion signal output terminal COMP of the frequency synthesis chip 61 to receive the waveform completion signal.

[0315] The third counting chip 51 receives a clock signal through the clock input terminal CP. In this embodiment, the clock input terminal CP of the third counting chip 51 is connected to the clock source T1. When the second counter 50 receives the waveform completion signal of the frequency synthesizer 60, through the preset control of the third counting chip 51, the time interval data sent from the selector 30 is read, inverted by the NOT gate array T2 and then counted, which is equivalent to counting down time according to the clock signal. It should be noted that in other embodiments, through appropriate circuit settings, the second counter 50 can also perform forward counting on the read time interval data.

[0316] When the reciprocal of the time interval is completed, all the data output pins Q0, Q1, Q2, and Q3 of the third counter chip 51 output high levels. Through the processing of the first AND gate M3 and the second NOT gate M4, a low level is output to the counting control pin EP / ET of the third counter chip 51, causing the third counter chip 51 to stop counting. At the same time, the high-level counting completion signal is sent to the frequency synthesizer 60 as an enable signal for outputting the next set waveform.

[0317] When a set waveform is completed by the frequency synthesizer 60, a high-level waveform completion signal is issued. After passing through the third NOT gate M5, a low level is sent to the preset control pin LD of the third counter chip 51, causing the third counter chip 51 to re-read the time interval data and start time counting again. It should be noted that during the time interval between two set waveforms, the frequency synthesizer 60 can stop outputting or output at a specific frequency during the time counting process of the second counter 50.

[0318] In the settings of some chips, within the last clock cycle of the set waveform, the frequency synthesis chip 61 outputs a high-level waveform completion signal. The high-level waveform completion signal will pull down the output signal of the first AND gate M3 to a low level. Please continue to refer to Figure 11 , in one embodiment, the second counter 50 further includes a first NAND gate M6. The first input terminal of the first NAND gate M6 is connected to the output terminal of the second NOT gate M4, the second input terminal of the first NAND gate M6 is connected to the output terminal of the third NOT gate M5, and the output terminal of the first NAND gate M6 is connected to the frequency synthesizer 60 to output an enable signal.

[0319] By connecting two NOT gates and one NAND gate, it is equivalent to forming an OR gate circuit. That is to say, as long as any one of the output signal of the first AND gate M3 or the waveform completion signal of the frequency synthesizer 60 is high level, the effect of the enable signal can be achieved, so that the frequency synthesis chip 61 can receive the last clock signal again within the last clock cycle of the set waveform, thereby finally ending the output of the current set waveform.

[0320] It should be noted that the set waveform of the frequency-modulated continuous wave output by the frequency synthesizer 60 can be selected from at least one of a triangular wave, a sawtooth wave, a stepped wave, and a sine wave.

[0321] On the other hand, the present application also provides a radar. Figure 12 It is a block diagram of the radar according to an embodiment of the present application.

[0322] Please refer to Figure 12 , the radar includes an antenna 1 and the radar waveform generation circuit 2 as described above that are connected to each other.

[0323] When the radar waveforms with different Doppler resolutions generated by the radar waveform generation circuit 2 are sent to the external space through the antenna 1 to detect the detection target. By further processing the echoes with different Doppler resolutions, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

[0324] Please continue to refer to Figure 12 , in one embodiment, the radar further includes a main controller 3 for controlling the frequency modulated continuous wave. The main controller 3 is connected to the frequency synthesizer 60 in the radar waveform generation circuit 1.

[0325] The waveform parameters of the frequency synthesizer 60 can be flexibly set through the main controller 3, providing a flexible means for further increasing the measurement range and further refining the minimum measurement resolution.

[0326] On the other hand, the present application also provides a method for generating a radar waveform.

[0327] In one embodiment, the method for generating a radar waveform includes:

[0328] Obtain the first enable signal; generate a set waveform of the frequency modulated continuous wave according to the first enable signal, and generate a waveform completion signal when the set waveform is completed; obtain time gap data according to the waveform completion signal; perform time counting according to the time gap data, and generate a second enable signal when the time counting is completed.

[0329] The step of obtaining time gap data according to the waveform completion signal includes:

[0330] Perform a quantity count on the waveform completion signal, and generate a strobe conversion signal when the quantity count is completed to replace the strobe of the first time gap to the second time gap as the time gap data.

[0331] By setting the difference in the time gap between two sets of set waveforms, different Doppler resolutions can be achieved between the two sets of waveforms. Through further processing of the echoes, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

[0332] Figure 13 It is a flowchart of the method for generating a radar waveform according to an embodiment of the present application.

[0333] The sawtooth wave is a commonly used frequency modulated continuous wave in radar detection. As Figure 13 shown, in one embodiment, the steps of the method for generating a radar waveform include:

[0334] S410: Obtain the first enable signal; enter step S20;

[0335] S420: Generate a sawtooth waveform based on the first enabling signal, and generate a waveform completion signal when a sawtooth waveform is completed; proceed to step S30;

[0336] S430: Determine whether the number of times the waveform completion signal has reached a set number. If not, proceed to step S431; if so, proceed to step S432;

[0337] S431: Obtain the stored first time interval as time interval data; proceed to step S440;

[0338] S432: Obtain the stored second time interval as time interval data; proceed to step S440;

[0339] S440: Perform time counting according to the time interval data, and generate a second enabling signal when the time counting is completed.

[0340] One pitch is composed of a set number of sawtooth waves (e.g., 16). Among the echoes of at least two pitches with different Doppler resolutions, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

[0341] On the other hand, the present application also provides a computer storage medium.

[0342] In one embodiment, a computer program is stored on the computer storage medium. When the computer program is executed by a processor, the radar waveform generation method as described above can be implemented.

[0343] When the processor executes the computer program, for the implementation of the radar waveform generation method, please refer to the above embodiments and will not be elaborated here.

[0344] It should be noted that in order to make the waveforms of adjacent multiple pitches required by the present application have different Doppler resolutions, in the above embodiments of radar waveform generation, the waveforms of two pitches are used with different waveform intervals to achieve this. For the sake of simplicity in introduction, the waveform durations, the number of waveforms, and the central wavelengths of the waveforms of the two pitches are set to be the same. In fact, in other embodiments, it can be achieved by setting the waveforms of adjacent multiple pitches to have different central wavelengths, different waveform durations, different numbers of waveforms, and different waveform intervals. For example, different total observation durations can be obtained by setting different waveform slopes or different radio frequency bandwidths to achieve different Doppler resolutions for each pitch.

[0345] The radar waveform generation circuit, method, radar, and computer storage medium provided by the present application can set different Doppler resolutions between two sets of waveforms by the difference in the time interval between two sets of set waveforms. Through further processing of the echoes, the measurement range can be effectively increased and the minimum measurement resolution can be further refined.

[0346] It should be noted that for the radar waveform generation circuit and method provided in this application, in order to illustrate the core technical features of this application, simple chips and gate circuits are used to implement frequency-modulated continuous waves with two time intervals to obtain two tone parts with different Doppler resolutions. In other cases, other software and hardware methods can be used. By simply transforming and expanding the core technical features of this application, for example, changing to using logic circuits to implement counting, adopting at least three time interval data to obtain more than three tone parts with different Doppler resolutions, etc., all fall within the protection scope of this application.

[0347] In this text, unless otherwise clearly specified and limited, the terms "installed", "connected", and "coupled" shall be understood in a broad sense. For example, it can be a fixed connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be directly connected or indirectly connected through an intermediate medium, and it can be the communication inside two components. For those of ordinary skill in the art, the specific meanings of the above terms can be understood according to specific situations.

[0348] In this text, the serial adjectives "first", "second", etc. used to describe components are only for distinguishing components with similar attributes, and do not mean that the components described in this way must follow a given order, or time, space, rank or other restrictions.

[0349] In this text, unless otherwise specified, the meanings of "a plurality of" and "several" are two or more.

[0350] Those of ordinary skill in the art can understand that all or part of the steps to implement the above method embodiments can be completed by hardware related to program instructions. The foregoing program can be stored in a computer-readable storage medium. When the program is executed, it executes the steps including the above method embodiments. The foregoing storage media include: various media such as ROM, RAM, magnetic disks or optical discs that can store program codes.

[0351] The technical features of the above-described embodiments can be combined arbitrarily. For the sake of brevity of description, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, it should be considered as the scope recorded in this specification.

[0352] In this text, the term "comprising", "including" or any other variant thereof is intended to cover non-exclusive inclusion. In addition to including the listed elements, it may also include other elements not specifically listed.

[0353] As described above, it is only the specific implementation manner of the present application, but the protection scope of the present application is not limited thereto. Any person skilled in the art within the technical scope disclosed by the present application can easily think of changes or substitutions, which should all be covered within the protection scope of the present application. Therefore, the protection scope of the present application shall be subject to the protection scope of the claims described above.

Claims

1. A radar waveform generation circuit, characterized in that, It includes a strobe, a first memory for storing a first time interval, a second memory for storing a second time interval, a first counter, a second counter, and a frequency synthesizer, where: The frequency synthesizer is connected to the first counter and the second counter. The frequency synthesizer generates a set waveform of a frequency modulated continuous wave according to a first enable signal sent by the second counter, and sends a waveform completion signal to the first counter and the second counter when the set waveform is completed; The first counter is used to count the number of the waveform completion signals, and sends a strobe conversion signal to the strobe when the number counting is completed; The strobe is used to send the first time interval as time interval data to the second counter when the number counting is not completed, and replaces the strobe of the first time interval with the second time interval according to the strobe conversion signal, so as to send the second time interval as time interval data to the second counter; The second counter is used to read the time interval data according to the waveform completion signal, perform time counting according to the time interval data, and send a second enable signal to the frequency synthesizer when the time counting is completed.

2. The radar waveform generation circuit according to claim 1, wherein The strobe includes a first group of data input pins, a second group of data input pins, a selection pin, and a data output pin. The first group of data input pins is connected to the first memory, the second group of data input pins is connected to the second memory, the selection pin is connected to the first counter, and the data output pin is connected to the second counter.

3. The radar waveform generation circuit according to claim 1, characterized in that, The first counter includes a first counting chip, a second counting chip, and a first NOT gate. The clock input terminals of the first counting chip and the second counting chip are respectively connected to the frequency synthesizer. The carry output pin of the first counting chip is connected to the counting control pin of the second counting chip. The lowest bit data output pin of the second counting chip is connected to the strobe. The first NOT gate is connected in series between the second lowest bit data output pin and the clear pin of the second counting chip.

4. The radar waveform generation circuit according to claim 1, wherein, The frequency synthesizer includes a second AND gate, a clock source, and a frequency synthesizer chip. The first input terminal of the second AND gate is connected to the clock source. The second input terminal of the second AND gate is connected to the second counter to receive the enable signal. The output terminal of the second AND gate is connected to the frequency synthesizer chip.

5. The radar waveform generation circuit according to claim 1, wherein The second counter includes a NOT gate array, a first AND gate, a third counting chip, a second NOT gate, and a third NOT gate. Among them, the NOT gate array is connected between the data input pin of the third counting chip and the strobe. The first AND gate is connected between the data output pin of the third counting chip and the input terminal of the second NOT gate. The output terminal of the first AND gate is connected to the input terminal of the second NOT gate. The output terminal of the second NOT gate is connected to the counting control pin of the third counting chip. The output terminal of the third NOT gate is connected to the preset control pin of the third counting chip. The input terminal of the third NOT gate is connected to the frequency synthesizer to receive the waveform completion signal; Among them, the second counter further includes a first NAND gate, a first input end of the first NAND gate is connected to an output end of the second NOT gate, a second input end of the first NAND gate is connected to an output end of the third NOT gate, and an output end of the first NAND gate is connected to the frequency synthesizer to output the enable signal.

6. A radar, characterized in that, It includes an antenna and a radar waveform generation circuit as described in any one of claims 1-5, which are interconnected.

7. The radar according to claim 6, characterized in that, The radar further includes a main controller for controlling the frequency modulated continuous wave, and the main controller is connected to the frequency synthesizer in the radar waveform generation circuit.

8. A radar waveform generation method, characterized in that, Applied to the radar waveform generation circuit as described in any one of claims 1-5, the radar waveform generation method includes: Obtaining a first enable signal; Generating a set waveform of the frequency modulated continuous wave according to the first enable signal, and generating a waveform completion signal when the set waveform is completed; Obtaining time gap data according to the waveform completion signal; Performing time counting according to the time gap data, and generating a second enable signal when the time counting is completed; The step of obtaining time gap data according to the waveform completion signal includes: Performing quantity counting on the waveform completion signal, using the first time gap as the time gap data when the quantity counting is not completed, and generating a strobe conversion signal when the quantity counting is completed to replace the strobe of the first time gap with the second time gap as the time gap data.

9. A computer storage medium, characterized in that, A computer program is stored on the computer storage medium, and when the computer program is executed by a processor, the radar waveform generation method as described in claim 8 can be implemented.

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