Permanent magnet synchronous motor control method and system based on speed feedback variable frequency tracking

Abstract

A permanent magnet synchronous motor control method and system based on speed feedback variable frequency tracking, relates to the field of permanent magnet synchronous motor control. The present invention is aimed at the problem of low estimation accuracy in the permanent magnet synchronous motor control without position sensor in the prior art. The present invention obtains estimated sine curves and cosine curves according to the motor's phase voltage, phase current and feedback angular velocity estimation value; obtains the rotor's angular velocity estimation value and position angle estimation value according to the estimated sine curve and cosine curve; obtains a driving signal according to the angular velocity estimation value and position angle estimation value; the present invention reduces the current error, improves the estimation accuracy, and realizes the stable operation of the permanent magnet synchronous motor.

Description

Technical Field

[0001] The present invention relates to the field of permanent magnet synchronous motor control, and in particular to a control method and a control system for a permanent magnet synchronous motor based on speed feedback variable frequency tracking. Background Art

[0002] PMSM has the advantages of high power output, low torque pulsation, high efficiency and strong fault tolerance. The drive system based on PMSM has broad application prospects in electric vehicles, ships, aerospace and other application fields that require high power and high reliability. PMSM vector control system usually needs to install position sensors to detect rotor position and speed information. However, the use of position sensors increases the cost and volume, and also reduces the reliability and safety of the control system. Therefore, in order to improve the reliability of the system, position sensorless detection technology is a feasible technical solution with important research significance. PMSM position sensorless detection technology is generally divided into two categories: one is to estimate the rotor position and speed by using the salient pole effect of the motor rotor, which is suitable for low speed and zero speed high-frequency injection method. Its estimation accuracy is independent of speed and is insensitive to parameter changes, but it has certain salient pole requirements for PMSM. In addition, the amplitude of the high-frequency interference signal must be correctly selected, otherwise electromagnetic noise will be generated. At the same time, its operating range and dynamic characteristics are also limited by the bus voltage. The other type is a method based on the mathematical model of the motor suitable for medium and high-speed areas, such as the extended Kalman filter method, the model reference method, and the sliding mode observer method. The estimation of the rotor position and speed depends on the motor back electromotive force, which will affect the estimation accuracy and the stability of the permanent magnet synchronous motor. Summary of the invention

[0003] In order to solve the above problems, the present invention provides a control method or control system of a permanent magnet synchronous motor based on speed feedback variable frequency tracking, which reduces current error, improves estimation accuracy, and realizes stable operation of the permanent magnet synchronous motor.

[0004] The present invention provides a control method for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, comprising the following steps:

[0005] S1, according to the phase voltage of the motor , Phase current and the estimated angular velocity of the feedback Get the estimated sine curve and cosine curve ;

[0006] S2, based on the estimated sine curve and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate ;

[0007] S3, according to the angular velocity estimation value , position angle estimate Get the driving signal.

[0008] Furthermore, the sinusoidal curve and cosine curve The method to obtain is:

[0009] S11, obtaining an angle estimation error parameter f;

[0010] S12. Obtain current error estimate , ;

[0011] S13, the angle estimation error f and the feedback angular velocity estimation value and current error estimate , Input into the flux observer to get the flux estimation value and , the flux observer is:

[0012] ;

[0013] in, and k θ are all positive gain parameters, and f is the angle error parameter generated by the actual angle and the estimated angle;

[0014] S14, obtaining an estimated sinusoidal curve according to the current observer with frequency tracking and the flux observer and cosine curve .

[0015] Furthermore, the current observer with frequency tracking is:

[0016] ;

[0017] in, is the FVT function.

[0018] Furthermore, the and The methods for obtaining include:

[0019] S121, according to the estimated value of stator current Get the current change , current change The angular frequency is obtained by PI regulation , and then get the stator current frequency ;

[0020] S122, the current error estimate , Enter the FVT function, the FVT function is:

[0021] ;

[0022] in, , .

[0023] Furthermore, the method for obtaining the angle error parameter f includes:

[0024] .

[0025] Further, step S2 includes:

[0026] S21, establish an extended state observer to obtain the estimated value of the rotor position angle ;

[0027] The state equation of the extended state observer is:

[0028] ;

[0029] in, is the estimated value of the motor rotor angular velocity, J is the moment of inertia, P is the number of pole pairs, T L is the load torque, Q is the total disturbance of the system;

[0030] S22, the rotor position angle estimation value As feedback, it is used to update the rotor position angle error .

[0031] The present invention provides a control system for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, comprising:

[0032] Flux observer is used to monitor the phase voltage of the motor , Phase current and the angular velocity estimate fed back Get the estimated sine curve and cosine curve ;

[0033] Extend the state observer to estimate the sinusoidal and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate ;

[0034] The current loop module is used to estimate the value according to the angular velocity , position angle estimate Get the driving signal.

[0035] Furthermore, the permanent magnet synchronous motor is a dual three-phase permanent magnet synchronous motor.

[0036] As described above, the control method or control system of the permanent magnet synchronous motor based on speed feedback variable frequency tracking provided by the present invention has the following effects:

[0037] 1. The control method of the present invention is applied to the field of motor control in multi-dimensional space, and realizes the stable operation of a dual three-phase permanent magnet synchronous motor based on position sensorless control.

[0038] 2. The present invention can obtain sine and cosine signals with speed and angle information through a new flux observer, and introduces an estimated angle error parameter in the flux observer, improves the estimation accuracy through estimated speed feedback, and uses FVT to track current frequency to reduce current error.

[0039] 3. The present invention processes the sine and cosine signals obtained by the new flux observer through a new third-order extended observer, and then estimates the position and speed of the rotor. The third-order extended observer of the present invention has the effect of resisting disturbances and improves the estimation accuracy.

[0040] 4. The present invention is suitable for applications such as aerospace, electric vehicles, etc. that have high requirements on the reliability and dynamic performance of motors. BRIEF DESCRIPTION OF THE DRAWINGS

[0041] Figure 1 is a stator current frequency detection block diagram of a specific embodiment of the present invention;

[0042] Figure 2 It is a block diagram of the FVT function structure principle of a specific embodiment of the present invention;

[0043] Figure 3 is a schematic diagram of rotor position information estimation based on a third-order ESO according to a specific embodiment of the present invention;

[0044] Figure 4 is a schematic diagram of a third-order ESO according to a specific embodiment of the present invention;

[0045] Figure 5 It is a structural block diagram of a position sensorless control system with speed feedback according to a specific embodiment of the present invention;

[0046] Figure 6 It is a main flow chart of the system of a specific embodiment of the present invention;

[0047] Figure 7 It is a main program flow chart of a specific embodiment of the present invention;

[0048] Figure 8 is a flow chart of an interruption program according to a specific embodiment of the present invention;

[0049] Fig. 9 is a DSP power supply circuit diagram of a specific embodiment of the present invention;

[0050] Fig.10 is a voltage sampling circuit diagram of a specific embodiment of the present invention;

[0051] Fig.11 is an AC current sampling circuit diagram of a specific embodiment of the present invention;

[0052] Fig.12 is a DC bias circuit diagram of a specific embodiment of the present invention;

[0053] Fig.13 is an overcurrent protection circuit diagram of a specific embodiment of the present invention;

[0054] Fig.14 2SD315AI is a driving circuit diagram of a specific embodiment of the present invention;

[0055] Fig.15 The sine and cosine curves observed by the flux observer at 1000 r / min in a specific embodiment of the present invention are ;

[0056] Fig.16 It is a 1000r / min speed waveform diagram of a specific embodiment of the present invention;

[0057] Fig.17 This is a 1000r / min angle waveform diagram of a specific embodiment of the present invention;

[0058] Fig.18 It is a 1000r / min electromagnetic torque waveform diagram of a specific embodiment of the present invention;

[0059] Fig.19 It is a waveform diagram of the dynamic change speed from 800r / min to 1000r / min of a specific embodiment of the present invention;

[0060] Fig. 20 This is a waveform diagram of the dynamic change angle from 800r / min to 1000r / min in a specific embodiment of the present invention;

[0061] Fig.21 This is a waveform diagram of a speed that changes dynamically from 50r / min to 100r / min according to a specific embodiment of the present invention;

[0062] Fig. 22 This is a dynamic angle waveform diagram of a specific embodiment of the present invention from 50r / min to 100r / min;

[0063] Fig.23is a sine-cosine curve observed by a 20r / min flux observer in a specific embodiment of the present invention;

[0064] Fig.24 It is a 20r / min speed waveform diagram of a specific embodiment of the present invention;

[0065] Fig.25 is a 20r / min angle waveform diagram of a specific embodiment of the present invention;

[0066] Fig.26 It is a 20r / min electromagnetic torque waveform diagram of a specific embodiment of the present invention. DETAILED DESCRIPTION

[0067] The following describes the embodiments of the present invention by specific examples, and those skilled in the art can easily understand other advantages and effects of the present invention from the contents disclosed in this specification. The present invention can also be implemented or applied through other different specific embodiments, and the details in this specification can also be modified or changed in various ways based on different viewpoints and applications without departing from the spirit of the present invention. It should be noted that the following embodiments and features in the embodiments can be combined with each other without conflict.

[0068] It should be noted that the illustrations provided in the following embodiments are only used to illustrate the basic concept of the present invention in a schematic manner, and thus the illustrations only show components related to the present invention rather than being drawn according to the number, shape and size of components in actual implementation. In actual implementation, the type, quantity and proportion of each component may be changed arbitrarily, and the component layout may also be more complicated.

[0069] The present invention provides a control method for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, wherein the permanent magnet synchronous motor is a dual three-phase permanent magnet synchronous motor, comprising the following steps:

[0070] S1, according to the phase voltage of the motor , Phase current and the angular velocity estimate fed back Get the estimated sine curve and cosine curve , specifically including the following steps:

[0071] S11, obtaining an angle estimation error parameter f;

[0072] The method for obtaining the angle error parameter f includes:

[0073] (1)

[0074] in, and is the sine and cosine curve of the actual angle, , p is the number of motor pole pairs, is the estimated sine and cosine curve, is the rotor permanent magnet flux.

[0075] S12. Obtain current error estimate , ;

[0076] The traditional current observer is:

[0077] (2)

[0078] In the above formula, R is the stator resistance, L is S is the stator inductance, is the motor mechanical angular velocity.

[0079] In order to reduce the current error, this embodiment uses the FVT function as the current error gain to track the current frequency, and obtains the current observer with frequency tracking as shown in formula (3):

[0080] (3)

[0081] in, is the FVT function.

[0082] Said and The methods for obtaining include:

[0083] S121, according to the estimated value of stator current Get the current change , current change The angular frequency is obtained by PI regulation , and then get the stator current frequency ;

[0084] In order to track the frequency of changes, the following Figure 1 The stator current frequency detection method based on a phase-locked loop is shown to detect the stator current frequency, specifically including:

[0085] According to the stator current estimate Get the current change :

[0086] (4)

[0087] In the above formula, This is the estimated angle obtained by the PLL (Phase Locked Loop) system.

[0088] Current change The angular frequency is obtained by PI regulation :

[0089] (5)

[0090] Then the stator current frequency is obtained for:

[0091] (6)

[0092] S122, the current error estimate , Enter the FVT function, the FVT function is:

[0093] (7)

[0094] in, , , K p , K r are the proportional gain and the peak gain at the resonant frequency, is the cut-off frequency, which can affect the resonance gain and bandwidth, To track the stator current frequency. The principle of parameter adjustment is to adjust K r To eliminate the steady-state error, adjust ω c To suppress frequency fluctuations. In the experiment, each parameter needs to be adjusted online. In order to facilitate the calculation and combine the parameters of the motor and the observer, ω c The frequency range is 2000-4000, K p and K r The parameters can be adjusted according to the real-time response and frequency characteristics of the observer, and the appropriate parameters can be determined based on the expected experimental results.

[0095] In order to adapt to the frequency change and computer digital realization, the transfer function in the time domain must be converted to the Z domain:

[0096] (8)

[0097] In the above formula, is the electrical angular velocity;

[0098] Which then translates into:

[0099] (9)

[0100] Define the state variables:

[0101] (10)

[0102] The derivative of (10) is derived to obtain the following state space equation:

[0103] (11)

[0104] The FVT function is obtained by formula (11):

[0105] (12)

[0106] Based on the above analysis, the structure diagram of the FVT function is as follows Figure 2 shown.

[0107] S13, the angle estimation error f and the feedback angular velocity estimation value and the current error estimate , Input into the flux observer to get the flux estimation value and .

[0108] The flux equation of the dual three-phase PMSM is:

[0109] (13)

[0110] And according to the voltage equation:

[0111] (14)

[0112] Combining equation (13) and equation (14) we get the estimated sine and cosine curves:

[0113] (15)

[0114] Estimate the error parameter f and the estimated speed feedback value based on the angle The flux observer is established as:

[0115] (16)

[0116] in, and k θ are all positive gain parameters, and f is the angle error parameter generated by the actual angle and the estimated angle;

[0117] S14. The estimated sine curve is obtained according to the following formula: and cosine curve :

[0118] (17)

[0119] S2, based on the estimated sine curve and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate , specifically including:

[0120] S21, establish an extended state observer to obtain the estimated value of the rotor position angle ;

[0121] like Figure 4 As shown, this embodiment adopts a third-order extended state observer (ESO), and its state equation is:

[0122] (18)

[0123] in, is the motor rotor speed, J is the moment of inertia, P is the number of pole pairs, T L is the load torque, Q is the total disturbance of the system;

[0124] S22, the rotor position angle estimation value As feedback, it is used to update the rotor position angle error ;

[0125] set up in and is the inductance of the d-axis and q-axis, when hour, is considered valid, such as Figure 3 As shown, the rotor position angle error is:

[0126] (19)

[0127] S3, according to the angular velocity estimation value , position angle estimate Get the driving signal.

[0128] like Figure 5 As shown, according to the angular velocity estimation value Get the estimated speed value n, and combine the estimated speed value n and the estimated position angle Sent to the current loop, the speed estimate n and the speed given value Make a difference and get the current through PI regulation , current , The phase voltages are respectively obtained through PI regulation, and the driving signals of the driving circuit are obtained through the SVOWM modulation algorithm. The SVPWM modulation is realized by controlling the on and off of the power devices in the inverter, thereby controlling the operation of the PMSM.

[0129] The present invention provides a control system for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, such as Figure 5 As shown, it includes a current inner loop and a speed outer loop, and the speed outer loop includes:

[0130] Flux observer is used to monitor the phase voltage of the motor , Phase current and the angular velocity estimate fed back Get the estimated sine curve and cosine curve ;

[0131] Extend the state observer to estimate the sinusoidal and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate ;

[0132] The current inner loop is used to estimate the angular velocity according to the , position angle estimate Get the driving signal.

[0133] Each module in the above-mentioned current inner loop and speed outer loop can be in the form of hardware circuit, or in the form of software program, or in the form of a combination of hardware circuit and software program. In this embodiment,

[0134] The structural principle diagram of the control system of this embodiment is as follows Figure 6 As shown, the hardware circuit of the control system is mainly composed of a control circuit, a voltage and current sampling circuit, an overcurrent protection circuit, an IGBT drive circuit, and a PMSM. The power side topology of the PMSM control system is an AC-DC-AC topology, that is, the 220V industrial frequency AC power is first passed through a rectifier bridge to obtain a DC bus voltage, and then the DC bus voltage output by the rectifier bridge is filtered and stabilized by a non-polarity filter capacitor and sent to the voltage type inverter. This embodiment uses a voltage type inverter with an anti-parallel diode structure to achieve inverter processing. The phase voltage and phase current of the motor are processed by the voltage and current sampling circuit and transmitted to the DSP control circuit. The voltage and current analog signals obtained by the sampling circuit are processed by the ADC conversion unit of the DSP for digital signal processing, and then the motor rotor position information is estimated by the flux observer on the DSP chip to achieve a double closed loop of speed and current. The current closed loop is output to the SVPWM modulation algorithm to obtain the drive signal of the drive circuit; SVPWM modulation is achieved by controlling the on and off of the power devices in the inverter, thereby controlling the operation of the PMSM.

[0135] The main program of the control circuit mainly completes system initialization and interruption, and the interruption program includes AD sampling, fault diagnosis, flux observer and ESO calculation, speed loop and current loop, etc. Figure 7In the main program flow chart of the system shown in FIG, when the system starts running, all interrupts are turned off and the system is initialized, completing the initial settings of each unit used in the program. After the initialization is completed, the interrupt is turned on, the timer is started, and the interrupt is waited.

[0136] The flowchart of the interrupt subroutine is as follows Figure 8 As shown, it is used to complete the sampling of phase voltage and current, estimate the rotor position and speed, speed loop PI regulation, current loop PI regulation, coordinate transformation, and output the control signal to the power module through the SVPWM algorithm to control the motor operation.

[0137] The control circuit in this embodiment adopts a DSP architecture, and the DSP chip is TMS320F28335 of TI. During the working process, the DSP chip is mainly responsible for the operation of the processed sampled signal, command current extraction, current tracking control algorithm, and sensorless control algorithm, and processes the phase voltage and phase current of the motor obtained through the voltage and current sampling circuit, and transmits them to the DSP control circuit. The voltage and current analog signals obtained by the sampling circuit are digitized through the ADC conversion unit of the DSP, and then the motor rotor position information is estimated on the DSP chip through the flux observer to realize the dual closed loop of speed and current, and the received modulation signal data is compared with the modulation wave and the carrier to obtain a PWM signal with a dead zone. The PWM is amplified by the drive circuit to drive the power switch tubes in each phase inverter to work.

[0138] like Fig. 9 The figure shows the DSP power supply circuit, which uses the TPS767D301 chip to power the DSP and outputs two stable DC voltages, 1.9V to supply the DSP core and 3.3V DC to supply the I / O port.

[0139] like Fig.10 The figure shows a voltage sampling circuit. The phase voltage and DC bus voltage are collected through an isolation amplifier. The principle is to use a voltage divider resistor on the high-voltage side to obtain a small voltage, and isolate the voltage in a differential manner through an isolation amplifier. The output side of the isolation amplifier is also output in a differential manner, and then passes through an operational amplifier to convert the differential signal into a single-ended signal and send it to the DSP.

[0140] like Fig.11 The figure shows the AC current sampling circuit, which collects the AC current of the motor through the current Hall sensor. The model of the current Hall sensor is HA2020 produced by YHDC. The maximum sampling current value is 100A, the power supply is ±15V, and the transformation ratio is 2000:1.

[0141] like Fig.12The figure shows a DC bias circuit, which limits the voltage amplitude of the current sampling signal to between 0-3V through the bias circuit. The bias voltage of 1.65V is generated by the voltage divider of resistors R37 and R39. Resistor R39 and capacitor C10 form a first-order RC filter circuit. The Schottky diode D2 constitutes a clamping circuit for the sampling voltage to prevent the voltage entering the DSP from being too large, which may damage the chip.

[0142] like Fig.13 The figure shows an overcurrent protection circuit, which is mainly used to prevent the phase current of the motor from exceeding the rated value of the IGBT switch tube, causing the IGBT switch tube to burn out. The circuit is composed of a comparator, which compares the sampled current signal with the limit value through the bias circuit. If the voltage value of the biased current signal is higher than 4.5V or lower than 0.6V, the control system blocks the PWM output. The selection of the limit value is related to the motor rating and the gain of the sampling circuit.

[0143] The driving circuit is used to amplify the low-level and low-power control signal output by the DSP so that it can drive the power switch tube. Fig.14 As shown, the driving circuit of this embodiment uses the driving module model 2SD315AI launched by the Swiss CONCEPT company, which has two working modes: direct mode and half-bridge mode. The 8-pin MOD of the driver is short-circuited with VDD, and it works in direct mode. At this time, channels A and B have no relationship, and the two channels work independently. RC1 and RC2 are short-circuited with GND, and the state output SO1 / SO2 also works independently. The 8-pin MOD of the driver is short-circuited with GND, and it works in half-bridge mode. A dead time is generated between the two channels. The dead time is adjusted by the RC network between pins 5 and 7. At this time, INB is connected to a high level to enable, and INA is the total input terminal of the two signals.

[0144] In order to further illustrate the present invention, a simulation experiment is carried out based on the above control system. In the simulation model, the DC bus voltage is set to 537 V, the PWM switching frequency is selected to 10 kHz, the speed is set to 1000 r / min, the initial load torque is 20 N∙m, and the load torque suddenly increases to 40 N∙m at 0.2 s. Figure 6 The sine and cosine curves observed by the flux observer with speed feedback The curve amplitude in the figure is plus or minus 1, and the phase difference is 90°. Compared with the back electromotive force, the influence of magnetic flux and speed is removed, and the speed feedback is introduced to make the estimation more accurate. Fig.16 and Fig.17 The speed and angle waveforms are 1000r / min respectively. Fig.16 It can be seen that the speed rises rapidly during the startup phase, and returns to a steady state after a certain overshoot when reaching the given speed. At the same time, the estimated speed can always follow the actual speed, and the closed-loop effect is good in the steady state. Fig.17 It can also be seen that the actual angle is highly consistent with the estimated angle. The estimated angle can always follow the actual angle with almost no phase delay. Fig.18 The electromagnetic torque waveform at 1000r / min. It can be seen from the figure that the starting torque is large. After the speed stabilizes, the torque recovers to the given value. At 0.2s, the load suddenly changes to 40N∙m. The output electromagnetic torque can quickly reach a steady state and follow the load changes, with good dynamic response.

[0145] Next, the dynamic speed change simulation is carried out. At medium and high speeds, the given speed is stepped from 800r / min to 1000r / min, and at low speeds, the speed is stepped from 50r / min to 100r / min. Fig.19 and Fig. 20 The speed and angle waveforms are from 800r / min to 1000r / min at medium and high speed. Fig.19 It can be seen that the speed rises quickly during the startup phase, and after a certain overshoot when reaching the given speed, it returns to the given speed and maintains a steady state. At the same time, the estimated speed can always follow the actual speed, and the closed-loop effect is good in the steady state. At 0.2s, the speed setting step is increased to 1000r / min, the motor can respond quickly and the speed can change with the given value. At the same time, the estimated speed is always consistent with the actual speed, which reflects the good dynamic effect of the flux observer. Fig. 20 It can also be seen that the actual angle is synchronized with the estimated angle, and the estimated angle can always follow the actual angle with almost no phase deviation. Fig.21 and Fig. 22 They are the dynamic change waveforms of speed and angle from 50r / min to 100r / min at low speed. Fig.21 It can be seen that the speed rises quickly during the startup phase, reaches a steady state within a period of time with a small overshoot, and there is basically no deviation between the estimated speed and the actual speed. The closed-loop effect is good at low-speed steady state. When the speed is given by a step change at 0.2s, the motor can respond quickly and follow the given value. At the same time, the estimated speed also follows the actual speed, which reflects the good dynamic effect of the flux observer at low speed. Fig. 22 It can also be seen that at low speeds, the actual angle and the estimated angle remain highly consistent, the estimated angle can always follow the actual angle, and there is almost no phase delay.

[0146] Static loading simulation of the motor at low speed, the motor starts with load, the given speed is 20r / min, the load torque is 20N∙m, Fig.23 The sine and cosine curves observed by the flux observer with speed feedback at 20r / min. The curve amplitude in the figure is plus or minus 1, and the phase difference is 90°. Compared with the back electromotive force, the influence of flux and speed at low speed is removed, and the introduction of speed feedback makes the estimation more accurate and more sinusoidal. Fig.15 and Fig.16 They are the speed and angle waveforms of 20r / min at low speed steady state. Fig.24 It can be seen that the speed rises steadily and slowly during the startup phase, reaches a steady state within a period of time with almost no overshoot, and the estimated speed can always follow the actual speed with basically no error, and the closed-loop effect is good in the steady state. Fig.25 The angle waveform can also verify this: the estimated speed always follows the actual speed, and the steady-state effect is good at low speeds. Fig.26 The electromagnetic torque waveform at 20r / min. The output electromagnetic torque can follow the given value and remain stable, which also shows good low-speed load-carrying capacity.

[0147] The above embodiments are merely illustrative of the principles and effects of the present invention, and are not intended to limit the present invention. Anyone familiar with the art may modify or alter the above embodiments without departing from the spirit and scope of the present invention. Therefore, all equivalent modifications or alterations made by a person of ordinary skill in the art without departing from the spirit and technical concept disclosed by the present invention shall still be covered by the claims of the present invention.

Claims

1. A control method for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, characterized in that: The steps include: S1, according to the phase voltage of the motor , Phase current and the estimated angular velocity of the feedback Get the estimated sine curve and cosine curve The sine curve and cosine curve The method to obtain is: S11, obtain the angle estimation error parameter f: ； S12. Obtaining a current error estimate based on a current observer with frequency tracking , , the current observer with frequency tracking is: ； in, is the FVT function; S13, the angle estimation error parameter f, the feedback angular velocity estimation value and current error estimate , Input into the flux observer to get the flux estimation value and , the flux observer is: ； in, and k θ are all positive gain parameters, and f is the angle error parameter generated by the actual angle and the estimated angle; S14, obtaining an estimated sinusoidal curve according to the current observer with frequency tracking and the flux observer and cosine curve ; S2, based on the estimated sine curve and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate ; S3, according to the angular velocity estimation value , position angle estimate Get the driving signal.

2. The control method of the permanent magnet synchronous motor based on speed feedback variable frequency tracking according to claim 1 is characterized in that: Said and The methods for obtaining include: S121, according to the estimated value of stator current Get the current change , current change The angular frequency is obtained by PI regulation , and then get the stator current frequency ; S122, the current error estimate , Enter the FVT function, the FVT function is: ； in, , .

3. The control method of a permanent magnet synchronous motor based on speed feedback variable frequency tracking according to claim 1 is characterized in that: Step S2 includes: S21, establish an extended state observer to obtain the estimated value of the rotor position angle ; The state equation of the extended state observation is: ； in, is the estimated value of the motor rotor angular velocity, J is the moment of inertia, P is the number of pole pairs, T L is the load torque, Q is the total disturbance of the system; S22, the rotor position angle estimation value As feedback, it is used to update the rotor position angle error .

4. A control system for a permanent magnet synchronous motor based on speed feedback variable frequency tracking, characterized in that: include: Flux observer is used to monitor the phase voltage of the motor , Phase current and the estimated angular velocity of the feedback Get the estimated sine curve and cosine curve The sine curve and cosine curve The method to obtain is: The angle estimation error parameter f is obtained as: ； Obtaining a current error estimate based on a current observer with frequency tracking , , the current observer with frequency tracking is: ； in, is the FVT function; The angle estimation error f and the feedback angular velocity estimation value are and the current error estimate , Input into the flux observer to get the flux estimation value and , the flux observer is: ； in, and k θ are all positive gain parameters, and f is the angle error parameter generated by the actual angle and the estimated angle; The estimated sinusoidal curve is obtained based on the current observer with frequency tracking and the flux observer and cosine curve ; Extend the state observer to estimate the sinusoidal and cosine curve Get the estimated value of the rotor angular velocity and position angle estimate ; The current loop module is used to estimate the value according to the angular velocity , position angle estimate Get the driving signal.

5. The control system of the permanent magnet synchronous motor based on speed feedback variable frequency tracking according to claim 4 is characterized in that: The permanent magnet synchronous motor is a dual three-phase permanent magnet synchronous motor.

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