Design Method and Structure of a Multi-Band Doherty Power Amplifier

Abstract

The present invention relates to the technical field of power amplifiers, and discloses a design method and structure of a multi-band Doherty power amplifier. The method includes: Step S1, setting the Doherty power amplifier to operate in a multi-band mode; Step S2, obtaining the phase difference between the current phase of the main path power amplifier and the current phase of the peak path power amplifier at the synthesis point of the Doherty power amplifier; Step S3, representing the first reflection coefficient of the current source end face of the main path power amplifier based on the phase difference, and representing the back-off impedance of the main path power amplifier based on the first reflection coefficient and the optimal impedance when the power amplifier operates in class B; establishing an impedance requirement for the multiple relationship between the back-off impedance of the main path power amplifier and the optimal impedance, and obtaining the phase difference that meets the impedance requirement. The method of the present invention can achieve the characteristic of dual-band impedance matching of a simple multi-order transmission line based on the obtained phase difference.

Description

Technical Field

[0001] The present invention relates to the technical field of power amplifiers, and particularly to a design method and structure of a multi-band Doherty power amplifier. Background Art

[0002] Due to the discontinuity of spectrum resources in wireless communication, the required power amplifier for operation should have the characteristics of broadband or multi-band operation. Most traditional multi-band Doherty power amplifiers utilize multi-order series-parallel transmission lines to form the characteristics of dual-band or multi-band impedance matching to achieve the design of multi-band Doherty frequencies. However, to achieve dual-band or multi-band impedance matching, not only a specific transmission line topology is required, but also a large number of impedance value calculations, which is very unfriendly to the design. For traditional Doherty power amplifiers, the currents at the combining point are in phase, which greatly limits the impedance matching space of the circuit and it is very difficult to achieve a Doherty power amplifier topology that meets specific requirements. Therefore, it is urgent to realize the design of dual-band or multi-band Doherty power amplifiers from other directions. Summary of the Invention

[0003] To solve the problem of difficulty in finding the topology of Doherty power amplifiers, this patent proposes a design method for a multi-band Doherty power amplifier. This method utilizes the phase difference between the currents of the main path and the peak path at the combining point to achieve the characteristics of impedance matching of multi-order transmission lines in two or more frequency bands.

[0004] The technical problem to be solved by the present invention is: to solve the problem of difficulty in finding the topology of transmission lines, the present invention provides a design method and structure of a multi-band Doherty power amplifier.

[0005] To achieve the above object, the technical solution adopted by the present invention is as follows: A design method for a multi-band Doherty power amplifier, characterized by comprising: Step S1, setting the Doherty power amplifier to a multi-band operation mode; Step S2, obtaining the phase difference between the current phase of the main path power amplifier and the current phase of the peak path power amplifier at the combining point of the Doherty power amplifier; Step S3, representing the first reflection coefficient of the current source end face of the main path power amplifier based on the phase difference, representing the back-off impedance of the main path power amplifier based on the first reflection coefficient and the optimal impedance when the power amplifier operates in class B; establishing an impedance requirement for the multiple relationship between the back-off impedance of the main path power amplifier and the optimal impedance, and obtaining the phase difference that meets the impedance requirement.

[0006] Further, in the step S3, the first reflection coefficient Γ of the current source end face of the main path power amplifier C is:

[0007]

[0008] Among them, e is the natural constant, j is the imaginary part symbol, Δθ is the phase difference, and the electrical length θ of the transmission line of the impedance inverter in the main path C The expression is:

[0009]

[0010] f is the operating frequency, and f1 is the center frequency.

[0011] Furthermore, in the step S3, the back-off impedance Z of the main path power amplifier CB is:

[0012]

[0013] where R opt is the optimal impedance when the power amplifier operates in class B.

[0014] Furthermore, design a dual-band Doherty power amplifier, and the impedance requirement is that the back-off impedance of the main path power amplifier is equal to twice the optimal impedance.

[0015] Furthermore, the step S3 specifically includes the following process:

[0016] Step S31, solve the saturated impedance Z at the combining point C1S is:

[0017] Z C1S = R L *(1 + e -jΔθ )

[0018] Solve the back-off impedance Z at the combining point C1B is:

[0019] Z C1B = R L ;

[0020] where R L is the resistance of the post-matching network at the combining point, j is the complex number symbol, Δθ = θ C1 - θ P1 , Δθ is the phase difference, θ C1 is the current phase of the main path power amplifier at the combining point, and θ P1 is the current phase of the peak path power amplifier at the combining point;

[0021] Step S32, when operating in the saturated state, solve the scattering matrix S of the impedance inverter as:

[0022]

[0023] Solve for the first reflection coefficient Γ of the current source end face of the main path power amplifier C The first expression is:

[0024]

[0025] Solve for the second reflection coefficient Γ of the main path power amplifier at the synthesis point end face C1 The second expression is:

[0026]

[0027] The first reflection coefficient Γ C and the second reflection coefficient Γ C1 The first relationship is:

[0028]

[0029] where θ C is the electrical length of the transmission line of the impedance inverter of the main path, R opt is the optimal impedance when the power amplifier operates in class B, Z C is the impedance of the main path power amplifier, Z C1 is the impedance at the synthesis point, Z0 is the reference impedance of the main path power amplifier from the synthesis point end face, is the conjugate impedance of Z0;

[0030] Step S33, when operating in the saturation state, the third expression for solving the relationship between the reference impedance Z0 and the phase difference Δθ of the main path power amplifier at the synthesis point end face is:

[0031]

[0032] And solve for the conjugate impedance of the reference impedance and the fourth expression for the relationship with the phase difference Δθ is:

[0033]

[0034] where, is the conjugate impedance of Z C1S ;

[0035] Step S34, when operating in the back-off state, the fifth expression for solving the second reflection coefficient and the phase difference of the main path power amplifier at the synthesis point end face is:

[0036]

[0037] Step S35, when operating in the back-off state, solve for the first reflection coefficient Γ of the current source end face of the main path power amplifier C and the sixth expression for the phase difference Δθ is:

[0038]

[0039] Among them, the electrical length θ of the transmission line of the impedance inverter on the main path C The expression is:

[0040]

[0041] Among them, f is the operating frequency, and f1 is the center frequency;

[0042] Step S36, when operating in the back-off state, the impedance Z of the main path power amplifier C is equal to the back-off impedance of the main path power amplifier (Z CB ), according to the first expression, the back-off impedance Z of the main path power amplifier and the seventh expression of the first reflection coefficient Γ CB are: C The seventh expression is:

[0043]

[0044] Step S37, establish an impedance requirement that the back-off impedance of the main path power amplifier is equal to twice the optimal impedance, and obtain the phase difference that meets the impedance requirement at the operating frequency according to the sixth expression and the seventh expression.

[0045] Furthermore, in step S33, when operating in the saturation state, it satisfies the condition Z C = R opt , according to the first expression, the first reflection coefficient Γ C = 0, according to the first relationship, the second reflection coefficient Γ C1 = 0; according to the second expression, the impedance Z at the synthesis point when operating in the saturation state (Z C1 in the saturation state is Z C1 = Z C1S ) is equal to the conjugate impedance of the reference impedance of the main path power amplifier at the end face of the synthesis point

[0046] Furthermore, in step S34, when operating in the back-off state, the impedance Z at the synthesis point C1 = R L , substitute the third expression and the fourth expression into the second expression, and the fifth expression is obtained.

[0047] The present invention also discloses a design structure of a multi-band Doherty power amplifier, which includes a power splitter, a first phase compensation line, a first input matching network, a main path power amplifier, a first output impedance matching network, a dual-band phase compensation line, a second input matching network, a peak path power amplifier, a second output impedance matching network, a second phase compensation line, a post-matching network and a resistor. The electromagnetic frequency is connected to the input end of the power splitter. The first output end of the power splitter is sequentially connected to the first phase compensation line, the first input matching network, the main path power amplifier, the first output impedance matching network, and the dual-band phase compensation line. The first output impedance matching network and the dual-band phase compensation line form an impedance inverter. The second output end of the power splitter is sequentially connected to the second input matching network, the peak path power amplifier, the second output impedance matching network, and the second phase compensation line. The dual-band phase compensation line and the second phase compensation line are connected together at the synthesis point. The synthesis point is connected to the post-matching network and then grounded through the resistor. The phase difference of the current phase at the synthesis point is determined by any of the above methods in the claims.

[0048] Compared with the prior art, the present invention has the following beneficial effects: In the technical solution of the present invention, there is a phase difference in the current at the synthesis point of the main path and the peak path. By using the phase difference of the currents of the main path and the peak path at the synthesis point, the requirements of saturated impedance and back-off impedance matching are effectively met in two or more frequency bands. BRIEF DESCRIPTION OF THE DRAWINGS

[0049] Figure 1 It is a schematic diagram of the principle of a dual-band Doherty power amplifier.

[0050] Figure 2 It is a schematic diagram of the output circuit principle of the main path.

[0051] Figure 3 It is a schematic diagram of the design method flow of the multi-band Doherty power amplifier of the present invention.

[0052] Figure 4 It is the relationship diagram of Γ CB and the phase difference at the operating frequencies of 0.8 and 2.6 GHz.

[0053] Figure 5 It is the circuit layout of the dual-band Doherty power amplifier obtained in the embodiment of the present invention.

[0054] Figure 6 It is a schematic diagram of the relationship between the phase difference and the output power at 1.8 - 2.6 GHz and 2.6 GHz in the embodiment of the present invention.

[0055] Figure 7(a) shows the relationship between the drain efficiency and the gain and the output power in the simulation of 1.70 - 1.95 GHz in the embodiment of the present invention.

[0056] Figure 7(b) shows the relationship between the drain efficiency, gain and output power in the 2.40 - 2.60 GHz simulation in the embodiment of the present invention.

[0057] Figure 8(a) shows the relationship between the drain efficiency, gain and output power in the 1.70 - 1.95 GHz simulation in the embodiment of the present invention.

[0058] Figure 8(b) shows the relationship between the measured drain efficiency, gain and output power in the 2.40 - 2.60 GHz in the embodiment of the present invention. Detailed implementation manners

[0059] In order to make the objectives, technical solutions and advantages of the present invention clearer, the present invention will be further described in detail below with reference to the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all of the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0060] In the description of the present invention, it should be noted that the terms "first" and "second" are only used for descriptive purposes and cannot be construed as indicating or implying relative importance.

[0061] As Figure 1 shown, a dual - band Doherty power amplifier, an electromagnetic frequency input signal is input to a power splitter. The first output terminal of the power splitter is sequentially connected to a first phase compensation line, a first input matching network, a main - path power amplifier, a first output impedance matching network, and a dual - band phase compensation line. The first output impedance matching network and the dual - band phase compensation line form an impedance inverter. The second output terminal of the power splitter is sequentially connected to a second input matching network, a peak - path power amplifier, a second output impedance matching network, and a second phase compensation line. The dual - band phase compensation line and the second phase compensation line are connected together at the synthesis point. After being connected, the synthesis point is connected to a matching network and then grounded through a resistor.

[0062] As Figure 2 shown in the output circuit of the main path, where Z C is the impedance of the main - path power amplifier, R opt is the optimal impedance when the power amplifier operates in class B, Γ C is the first reflection coefficient of the current - source end face of the main - path power amplifier, Γ C1 is the second reflection coefficient of the main - path power amplifier at the synthesis - point end face, and Z C1 is the impedance at the synthesis point. The above parameters meet the following requirements:

[0063] The impedance of the main - path power amplifier

[0064] The scattering matrix of the impedance inverter

[0065] Back-off impedance at the combining point

[0066] Saturation is the saturation state, and OBO is the back-off state.

[0067] In the design of a Doherty power amplifier, the key to achieving a broadband or multi-band symmetric Doherty power amplifier is that within the operating frequency band, the ratio of the real impedance at the back-off point on the current source end face to the real impedance at the saturation point is greater than 2. In the present invention, a dual-band Doherty power amplifier is taken as an example for illustration, which is a symmetric Doherty power amplifier operating at 1.70–1.95 GHz to 2.40–2.60 GHz with a 6 dB back-off. The impedance of the main path power amplifier Z C The impedance requirements at the saturation point and the back-off point are as shown in formula (1-1), and the impedance of the peak path power amplifier Z P The impedance requirements at the saturation point and the back-off point are as shown in formula (1-2).

[0068]

[0069]

[0070] Assume that the operating frequency points of the dual-band Doherty power amplifier are f1 and f2 respectively. In order to satisfy formulas (1-1) and (1-2) in both frequency bands. In the present invention, a phase difference (Δθ) is introduced between the currents of the main path power amplifier and the peak path power amplifier at the combining point, so that the impedance in more frequency bands can satisfy formula (1-1). In order to verify the proposed phase angle difference theory, a dual-band Doherty power amplifier is designed in the present invention, with Δθ being 0° at the f1 frequency point; and Δθ not being 0° at the f2 frequency point.

[0071] The design objective of the present invention is to find the phase difference that satisfies formula (1-1) for any operating frequency point f.

[0072] According to the design objective, the saturation impedance of the exact main path transmission line network (i.e., the saturation impedance Z CS ) and the back-off impedance (i.e., the back-off impedance Z CB ) of the main path power amplifier at the current source end face need to be obtained as a function of Δθ. For the convenience of analysis, it is assumed that the transmission line is passive, lossless, and reciprocal. In order to analyze the back-off point impedance of the main path (i.e., the back-off impedance Z CB ) of the main path power amplifier at the current source end face, the reflection coefficient analysis method is used to obtain the relationship between the back-off point impedance and the phase difference.

[0073] A design method for a multi-band Doherty power amplifier in this embodiment includes: Step S1, setting the Doherty power amplifier to operate in a multi-band mode; in the multi-band operating mode, some frequency bands are out of phase and some are in phase, so there is a phase difference between the current phase of the main path power amplifier and the current phase of the peak path power amplifier at the power amplifier synthesis point. Step S2, obtaining the phase difference between the current phase of the main path power amplifier and the current phase of the peak path power amplifier at the power amplifier synthesis point. Step S3, representing the first reflection coefficient of the current source end face of the main path power amplifier based on the phase difference, and representing the back-off impedance of the main path power amplifier based on the first reflection coefficient and the optimal impedance when the power amplifier operates in class B; establishing an impedance requirement for the relationship between the back-off impedance of the main path power amplifier and the multiple of the optimal impedance, and obtaining the phase difference that meets the impedance requirement. Based on the obtained phase difference, this embodiment can achieve the characteristic of dual-band impedance matching for a simple multi-order transmission line.

[0074] In some embodiments, in step S3, the first reflection coefficient Γ of the current source end face of the main path power amplifier C is:

[0075]

[0076] where e is the natural constant, j is the complex number symbol, Δθ is the phase difference, and the electrical length θ of the transmission line of the impedance inverter in the main path C The expression is:

[0077]

[0078] f is the operating frequency, and f1 is the center frequency. This embodiment can obtain the relationship between the first reflection coefficient Γ C and the phase difference Δθ.

[0079] In some embodiments, in step S3, the back-off impedance Z of the main path power amplifier CB is:

[0080]

[0081] where R opt is the optimal impedance when the power amplifier operates in class B. In this embodiment, substituting the relationship between the first reflection coefficient Γ C and the phase difference Δθ, when the impedance requirement for the relationship between the back-off impedance Z of the main path power amplifier CB and the multiple R of the optimal impedance opt is known, the phase difference Δθ can be obtained.

[0082] In some embodiments, a dual-band Doherty power amplifier is designed. According to the expression in the back-off state in formula (1-1), the back-off impedance of the main path power amplifier is equal to twice the optimal impedance.

[0083] In some embodiments, step S3 specifically includes the following process:

[0084] Step S31, according to the current-voltage relationship, obtain the current phase θ of the main path power amplifier at the synthesis point C1 and the current phase θ of the peak path power amplifier at the synthesis point P1 of the phase difference Δθ and the relationship formula of the impedance at the synthesis point.

[0085] The phase difference is:

[0086] Δθ = θ C1 -θ P1 (1-3)

[0087] Solve for the saturation impedance Z at the synthesis point C1S as:

[0088] Z C1S = R L *(1 + e -jΔθ ) (1-4)

[0089] Solve for the back-off impedance Z at the synthesis point C1B as:

[0090] Z C1B = R L (1-5)

[0091] where R L is the resistance of the post-matching network at the synthesis point, and j is the complex number symbol;

[0092] Step S32, when operating in the saturation state, the impedance inverter of the main path can achieve matching the saturation impedance Z at the synthesis point C1S to R opt (which means that when the Doherty power amplifier is in the saturation state, the saturation impedance Z at the current source end face of the main path power amplifier CS remains R throughout the operating frequency band opt , and R opt is the optimal impedance when the power amplifier operates in class B), and if the impedance inverter is passive, lossless, and reciprocal, the scattering matrix S of the impedance inverter can be obtained as:

[0093]

[0094] Solve for the first reflection coefficient Γ C of the current source end face of the main path power amplifier, and the first expression is:

[0095]

[0096] Solve for the second reflection coefficient Γ of the main path power amplifier at the end face of the synthesis point C1 The second expression is:

[0097]

[0098] The first reflection coefficient Γ C and the second reflection coefficient Γ C1 The first relationship is:

[0099]

[0100] where θ C is the electrical length of the transmission line of the impedance inverter of the main path, and R opt is the optimal impedance when the power amplifier operates in class B, Z C is the impedance of the main path power amplifier, Z C1 is the impedance at the synthesis point, and Z0 is the reference impedance of the main path power amplifier from the end face of the synthesis point, is the conjugate impedance of Z0;

[0101] Step S33, when operating in the saturation state, the third expression for solving the relationship between the reference impedance Z0 and the phase difference Δθ of the main path power amplifier at the end face of the synthesis point is:

[0102]

[0103] And solve for the conjugate impedance of the reference impedance and the fourth expression for the relationship between the phase difference Δθ is:

[0104]

[0105] where is the conjugate impedance of Z C1S ;

[0106] Step S34, when operating in the back-off state, the fifth expression for solving the second reflection coefficient and the phase difference of the main path power amplifier at the end face of the synthesis point is:

[0107]

[0108] Step S35, when operating in the back-off state, the second reflection coefficient Γ of the main path power amplifier at the end face of the synthesis point has been obtained in step S34 C1 The relationship with Δθ, then from formula (1-9), it can be known that the sixth expression for solving the first reflection coefficient Γ C and the phase difference Δθ at the current source end face of the main path power amplifier is:

[0109]

[0110] Formula (1-13) means that the first reflection coefficient (Γ C ) is determined by the magnitude and phase angle of Δθ and the impedance of the main line and the transmission line electrical length of the inverter (θ C ) is determined by θ C It is linearly related to the frequency (f), which means that when the operating frequency f is determined, the electrical length of the transmission line (θ C ) will also be determined. Among them, the impedance of the main circuit and the transmission line electrical length of the inverter θ C The expression is:

[0111]

[0112] Among them, f is the operating frequency, f1 is the center frequency;

[0113] Step S36, transform formula (1-7), when working in the fallback state, the main power amplifier impedance Z C Equal to the back-off impedance Z of the main power amplifier CB , that is, Z C =Z CB According to the first expression (1-7), the main power amplifier back-off impedance Z is obtained CB and the first reflection coefficient Γ C The seventh expression of is:

[0114]

[0115] Step S37, in order to make the real part of the main power amplifier back-off impedance real (Z CB ) satisfies the back-off impedance Z in formula (1-1) C Equal to 2R opt The analysis process is as follows: Determine the operating frequency of the Doherty power amplifier as f and the center frequency as f1, then the electrical length θ of the center frequency f1 C1 =90°, electrical length θ at operating frequency f C Determined by formula (1-14). Using formulas (1-13) and (1-15), we can get the condition that real(Z CB )=2R opt The phase difference Δθ.

[0116] In some embodiments, in step S33, when working in saturation state, the main power amplifier impedance Z is satisfied. C =R optUnder the condition, the first reflection coefficient Γ in formula (1-7) is obtained according to the first expression C = 0. According to the first relational expression (1-9), it can be known that Γ C and Γ C1 relationship, then the second reflection coefficient Γ C1 = 0; According to the second expression (1-8), the impedance Z at the synthesis point when working in the saturation state is obtained C1 (Z C1 = Z C1S ) at saturation is equal to the conjugate impedance of the reference impedance of the main path power amplifier at the end face of the synthesis point Therefore, the reference impedance of the main path power amplifier at the end face of the synthesis point can be obtained Conjugate impedance of the reference impedance

[0117] In some embodiments, in step S34, when working in the back-off state, the impedance Z of the main path at the synthesis point C1 = R L . Substitute the third expression (1-10) and the fourth expression (1-11) into the second expression (1-8) to obtain the fifth expression (1-12).

[0118] Embodiment

[0119] Use a commercial CGH40010 power amplifier as the power amplifier for the main path and the peak path. Its optimal impedance (R opt ) is 32 ohms. The operating frequency points of the dual-band Doherty power amplifier are selected at 1.8 and 2.6 GHz. Among them, θ C = 90° at 1.8 GHz. Therefore, θ C = 130° at 2.6 GHz can be obtained using formula (1-14). From formula (1-13), the relationship between the reflection coefficient Γ CB of the main path power amplifier at the end face of the synthesis point and the phase difference Δθ at 1.8 and 2.6 GHz can be obtained, as shown in Figure 4 . For the solid line, Δθ = 90°, θ C = [0°, 360°]. The dotted line represents the relationship between Γ CB and the phase difference Δθ at 1.8 GHz, θ C = 90°, Δθ = [0°, 360°]. It can be seen that Figure 4 The filled square points and the blank square points in (the square points represent the relationship between the reflection coefficient of the traditional Doherty power amplifier and the electrical length θ C ), Figure 4 The circles in represent the relationship between the reflection coefficient of the traditional Doherty power amplifier and the electrical length θ C ) coincide with the filled circular points, which means that when θC When it is 90°, in order to satisfy real(Z CB ), the phase difference Δθ at the combining point should be selected as 0°. The dashed line represents the relationship between Γ CB and Δθ at 2.6 GHz, θ C = 130°, Δθ = [0°, 360°]. It can be seen that Figure 4 The blank square points (Z CB ) deviate from real(Z CB ) = 64 Ω. In order to make real(Z CB ) satisfy real(Z CB ) = 64 Ω, the magnitude of real(Z CB ) can be controlled by changing the phase difference Δθ. When Δθ = 45°, real(Z CB ) = 64 Ω, as shown by the blank dots on the dotted line in Figure 4 .

[0120] After obtaining that the phase differences Δθ at the operating frequency points of the dual-band Doherty power amplifier at 1.8 and 2.6 GHz are 0° and 45° respectively, based on the phase difference theory, a dual-band Doherty power amplifier can be designed. The circuit layout of the dual-band Doherty power amplifier is as shown in Figure 5 ; its structural schematic diagram is as shown in Figure 1 . The phase difference at the combining point is determined by the design method of the multi-band Doherty power amplifier in any embodiment of the present invention. At 1.8 GHz and 2.6 GHz, when the Doherty power amplifier operates in the saturation state, the phase differences Δθ at the combining point are 5° and 36° respectively, as shown in Figure 6 . The simulated efficiency and gain of the designed dual-band Doherty power amplifier at 1.70 - 1.95 GHz and 2.40 - 2.60 GHz are shown in Figures 7(a) and 7(b). The measured efficiency and gain at 1.7 - 1.95 GHz and 2.40 - 2.60 GHz are shown in Figures 8(a) and 8(b).

[0121] Finally, it should be noted that the above embodiments are only preferred embodiments of the present invention to illustrate the technical solutions of the present invention, rather than limiting it, and certainly not limiting the patent scope of the present invention; although the present invention has been described in detail with reference to the foregoing embodiments, those of ordinary skill in the art should understand that they can still modify the technical solutions described in the foregoing embodiments, or perform equivalent replacements on some or all of the technical features; and these modifications or replacements do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the various embodiments of the present invention; that is to say, any meaningless changes or polishings made in the main design concept and spirit of the present invention, as long as the technical problems solved are still the same as those of the present invention, should be included in the protection scope of the present invention; in addition, directly or indirectly applying the technical solutions of the present invention to other related technical fields shall likewise be included in the patent protection scope of the present invention.

Claims

1. A design method of a multi-band Doherty power amplifier, characterized in that, Including: Step S1, setting the Doherty power amplifier to a multi-band operating mode; Step S2: Obtain the phase difference between the current phase of the main path power amplifier and the current phase of the peak path power amplifier at the synthesis point of the Doherty power amplifier; Step S3: Represent the first reflection coefficient of the current source end face of the main path power amplifier based on the phase difference, and based on the first reflection coefficient and the optimal impedance R when the power amplifier operates in class B opt represent the back-off impedance of the main path power amplifier; establish an impedance requirement for the multiple relationship between the back-off impedance of the main path power amplifier and the optimal impedance, and obtain the phase difference that meets the impedance requirement; In the step S3, the first reflection coefficient Γ of the current source end face of the main road power amplifier C is as follows: where e is the natural constant, j is the imaginary part symbol, Δθ is the phase difference, and θ is the electrical length of the transmission line of the impedance inverter in the main path C The expression is: where f is the operating frequency and f1 is the center frequency.

2. The design method of a multi-band Doherty power amplifier according to claim 1, characterized in that, In the step S3, the back-off impedance Z of the main path power amplifier CB is as follows: Among them, R opt is the optimal impedance when the power amplifier operates in Class B.

3. The design method of a multi-band Doherty power amplifier according to claim 1, characterized in that Design a dual-band Doherty power amplifier, and the impedance requirement is that the back-off impedance of the main path power amplifier is equal to twice the optimal impedance.

4. The design method of a multi-band Doherty power amplifier according to claim 1, characterized in that, The specific content of step S3 includes the following processes: Step S31, solve for the saturation impedance Z at the synthesis point C1S It is: Z C1S = R L *(1 + e -jΔθ ); Solve for the fallback impedance Z at the synthesis point C1B It is: Z C1B = R L ; where R L is the impedance that the matching network matches 50Ω to R after use, j is the symbol of the imaginary part, Δθ = θ L - θ C1 , Δθ is the phase difference, θ P1 is the current phase of the main path power amplifier at the synthesis point, θ C1 is the current phase of the peak path power amplifier at the synthesis point; P1 ​ Step S32, when operating in the saturation state, solving the scattering matrix S of the impedance inverter as: Solve for the first reflection coefficient Γ of the current source end face of the main path power amplifier C The first expression for Solve for the second reflection coefficient Γ of the main path power amplifier at the end face of the synthesis point C1 The second expression for is as follows: The first reflection coefficient Γ C and the second reflection coefficient Γ C1 The first relational expression is as follows: Among them, θ C is the electrical length of the transmission line of the impedance inverter on the main path, and R opt is the optimal impedance when the power amplifier operates in class B, and Z C is the impedance of the main path power amplifier, and Z C1 is the impedance at the synthesis point, and Z0 is the reference impedance of the main path power amplifier from the end face of the synthesis point, is the conjugate impedance of Z0; Step S33, when operating in the saturation state, solving the third expression of the relationship between the reference impedance Z0 and the phase difference Δθ at the end face of the main path power amplifier at the synthesis point as: And solve for the conjugate impedance of the reference impedance The fourth expression for the relationship with the phase difference Δθ is as follows: Among them, is the conjugate impedance of Z C1S ; Step S34, when operating in the back-off state, solving the fifth expression of the second reflection coefficient and the phase difference at the end face of the main path power amplifier at the synthesis point as: Step S35, when working in the fallback state, solve for the first reflection coefficient Γ of the current source end face of the main path power amplifier C and the sixth expression for the phase difference Δθ is as follows: Among them, the electrical length θ of the transmission line of the impedance inverter on the main road C The expression is: where f is the operating frequency and f1 is the center frequency; Step S36, when working in the back-off state, according to the first expression, obtain the back-off impedance Z of the main path power amplifier CB and the first reflection coefficient Γ at the back-off point C The seventh expression of is: Step S37, establishing an impedance requirement that the back-off impedance of the main path power amplifier is equal to twice the optimal impedance, and obtaining the phase difference that satisfies the impedance requirement at the operating frequency according to the sixth expression and the seventh expression.

5. The design method of a multi-band Doherty power amplifier according to claim 4, characterized in that, In the step S33, when working in the saturation state, the impedance Z of the main path power amplifier satisfies C = R opt Under the condition, the first reflection coefficient Γ C = 0 is obtained according to the first expression, and the second reflection coefficient Γ C1 = 0 is obtained according to the first relational expression; according to the second expression, the impedance at the synthesis point when working in the saturation state is equal to the conjugate impedance of the reference impedance of the main path power amplifier at the end face of the synthesis point.

6. The design method of a multi-band Doherty power amplifier according to claim 4, characterized in that, In step S34, when working in the fallback state, the impedance Z at the synthesis point C1 = R L . Substitute the third and fourth expressions into the second expression, and the fifth expression is obtained.

7. A design structure of a multi-band Doherty power amplifier, characterized in that, Including a power splitter, a first phase compensation line, a first input matching network, a main path power amplifier, a first output impedance matching network, a dual-band phase compensation line, a second input matching network, a peak path power amplifier, a second output impedance matching network, a second phase compensation line, a post-matching network, and a resistor. The electromagnetic frequency is connected to the input end of the power splitter. The first output end of the power splitter is sequentially connected to the first phase compensation line, the first input matching network, the main path power amplifier, the first output impedance matching network, and the dual-band phase compensation line. The first output impedance matching network and the dual-band phase compensation line form an impedance inverter. The second output end of the power splitter is sequentially connected to the second input matching network, the peak path power amplifier, the second output impedance matching network, and the second phase compensation line. The dual-band phase compensation line and the second phase compensation line are connected together at the synthesis point. The synthesis point is connected to the post-matching network and then connected to the resistor and grounded; The phase difference of the current phase at the synthesis point is determined by the method according to any one of claims 1-6.

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