A variable integrated inductor energy receiving coil and a wireless charging car

Abstract

This invention provides a variable integrated inductor energy receiving coil and a wireless charging vehicle. The energy receiving coil incorporates a variable integrated inductor, which includes an AC inductor coil for a secondary compensation network and two DC control windings for altering the equivalent self-inductance of the AC inductor coil. The two DC control windings are arranged in opposite directions on the inner and outer sides of the AC inductor coil. The advantages are: by integrating a variable inductor into the energy receiving coil, the inductance parameters and mutual inductance can be changed according to different needs. Therefore, in a dynamic wireless charging system for electric vehicles, this further improves lateral offset resistance while suppressing power fluctuations during rail switching, and compensates for the drop in system output power during lateral offset using a smaller DC current.

Description

Technical Field

[0001] This invention relates to wireless power transmission technology, specifically to a variable integrated inductor power receiving coil and a wireless charging vehicle. Background Technology

[0002] Wireless power transfer (WPT) achieves the transmission of electrical energy without electrical contact through media such as magnetic fields, electric fields, lasers, and microwaves. This technology can effectively solve the problems of limited equipment flexibility and safety hazards caused by traditional wired power supply methods. Currently, in application fields such as electric vehicles, consumer electronics, and home appliances, experts and scholars at home and abroad have conducted research on this technology and obtained considerable theoretical results.

[0003] In existing wireless power transmission systems, the circuit components at the energy receiver end are often predetermined according to fixed design requirements, with component parameters remaining constant. However, for applications like electric vehicles, to achieve continuous wireless power supply during operation, dynamic wireless charging systems for electric vehicles have been proposed. These systems utilize wireless energy generating coils in the form of multiple short guide rails to achieve dynamic wireless power transmission. However, during dynamic operation, the receiver end inevitably experiences offset, resulting in power fluctuations, especially lateral offsets. Adjustments to circuit parameters are often necessary to accommodate these power fluctuations. Summary of the Invention

[0004] To address the power drop caused by lateral offset of the receiving coil in a wireless charging system, this invention first proposes a variable integrated inductor energy receiving coil. By changing the parameters of the integrated inductor, it adapts to different offset conditions, thereby suppressing power fluctuations.

[0005] To achieve the above objectives, the specific technical solution adopted by the present invention is as follows:

[0006] A variable integrated inductor energy receiving coil is characterized by having a variable integrated inductor on the energy receiving coil. The variable integrated inductor includes an AC inductor coil for a secondary compensation network and two DC control windings for changing the equivalent self-inductance of the AC inductor coil. The two DC control windings are respectively arranged on the inner and outer sides of the AC inductor coil with opposite winding directions.

[0007] Optionally, a DC power supply is applied to the two DC control windings, wherein the magnetic field generated by the current flowing through the inner DC control winding and the outer DC control winding is in opposite directions.

[0008] Optionally, the inner DC control winding has the same number of turns as the outer DC control winding.

[0009] Optionally, the energy receiving coil is wound horizontally in the form of a rectangular coil, and the AC inductor coil and the two DC control windings in the variable integrated inductor are wound in the same plane inside the energy receiving coil.

[0010] Optionally, the top surfaces of the energy receiving coil and the variable integrated inductor are provided with a magnetic core plate and a shielding plate.

[0011] Optionally, the two DC control windings have the same number of turns.

[0012] Based on the aforementioned variable integrated inductor energy receiving coil, this invention also proposes a wireless charging vehicle, which mainly solves the problem of charging power fluctuation caused by vehicle body displacement during dynamic wireless charging of electric vehicles. The specific technical solution is to use the aforementioned variable integrated inductor energy receiving coil in the wireless charging energy receiving circuit.

[0013] Optionally, the wireless charging energy receiving circuit further includes a secondary resonant capacitor, a secondary compensation capacitor, and a rectifier filter circuit. The secondary resonant capacitor, the secondary compensation capacitor, and the variable integrated inductor form an LCC compensation network connected to the energy receiving coil. The output of the LCC compensation network is connected to the electrical load via the rectifier filter circuit.

[0014] The effects of this invention are:

[0015] This invention proposes a variable integrated inductor energy receiving coil and a wireless charging vehicle. The variable inductor is integrated on the energy receiving coil. The inductance parameters and mutual inductance effect of the integrated inductor can be changed according to different needs. Thus, in the dynamic wireless charging system of electric vehicles, the lateral anti-offset performance can be further improved on the basis of suppressing power fluctuations when switching rails. The system output power drop during lateral offset is compensated using a small DC current. Attached Figure Description

[0016] To more clearly illustrate the specific embodiments of the present invention or the technical solutions in the prior art, the accompanying drawings used in the description of the specific embodiments or the prior art will be briefly introduced below.

[0017] Figure 1 The curves showing the variation of equivalent mutual inductance under different lateral offsets in the prior art;

[0018] Figure 2 This is a schematic diagram of the structure of a dynamic wireless charging system for electric vehicles provided in a specific embodiment;

[0019] Figure 3 This is a circuit schematic diagram of a dynamic wireless charging system for electric vehicles provided in a specific embodiment;

[0020] Figure 4 This is the equivalent decoupling circuit for the secondary-side compensation network;

[0021] Figure 5 To consider the equivalent circuit of inductive coupling on the same side;

[0022] Figure 6 This is a schematic diagram of the structure of a variable integrated inductor coil;

[0023] Figure 7 This is a schematic diagram of the windings and current direction of a variable integrated inductor coil.

[0024] Figure 8 Typical magnetization curves for magnetic materials;

[0025] Figure 9 It is a hysteresis loop magnetized by a polarized DC bias magnetic field;

[0026] Figure 10 This is a design flowchart of the variable inductor magnetic integrated coupling mechanism in a specific embodiment;

[0027] Figure 11 The main structural view and dimensions of the magnetic integrated coupling mechanism;

[0028] Figure 12 A top view and dimensions of the magnetic integrated coupling mechanism;

[0029] Figure 13 This is a graph showing the relationship between the coupling coefficient k and the gap distance g.

[0030] Figure 14 For the number of turns n dc1 and number of turns n dc2 The effect of the mutual inductance difference between the secondary coil and the control coil;

[0031] Figure 15 Design region for composite mutual inductance in integrated control coil;

[0032] Figure 16 To integrate the self-inductance of the compensation inductor coil and its mutual inductance with the receiving coil under different combinations of turns;

[0033] Figure 17 The self-inductance of the integrated compensation inductor coil and its mutual inductance with the receiving coil are calculated when the number of turns on the inner and outer sides are the same.

[0034] Figure 18 To integrate the influence of different DC currents in the control coil on the flux density modulus;

[0035] Figure 19 The inverter output waveforms of the system under different receiving coil positions;

[0036] Figure 20The simulated output power and efficiency are calculated when the lower edge of the transmitting coil moves with a lateral offset of 10cm.

[0037] Reference numerals: 1-Energy transmitting coil, 11-Primary side compensation coil, 2-Energy receiving coil, 21-AC inductor coil, 22-DC control winding. Detailed Implementation

[0038] The embodiments of the technical solution of the present invention will now be described in detail with reference to the accompanying drawings. These embodiments are merely illustrative of the technical solution of the present invention and are therefore intended to limit the scope of protection of the present invention.

[0039] It should be noted that, unless otherwise stated, the technical or scientific terms used in this application should have the ordinary meaning as understood by those skilled in the art to which this invention pertains.

[0040] In existing wireless charging vehicles, whether during dynamic or static charging, the transmitting and receiving coils are prone to misalignment. Figure 1 Taking the lateral offset data shown as an example, the equivalent mutual inductance is prone to change under different lateral offsets, which leads to fluctuations in power transmission.

[0041] This embodiment provides a wireless charging vehicle, through which a dynamic wireless charging system for electric vehicles is constructed, such as... Figure 2 As shown, it includes a power transmitting end and a power receiving end. The power transmitting end adopts multiple energy transmitting rails arranged side by side. Each energy transmitting rail includes an energy transmitting coil 1 wound along a first direction and a primary side compensation coil 11 wound in series with the energy transmitting coil 1 and in the opposite direction to the first direction. The power receiving end adopts a variable integrated inductor energy receiving coil, including an energy receiving coil 2 and a variable integrated inductor. The variable integrated inductor includes an AC inductor coil 21 for the secondary side compensation network and two DC control windings 22 for changing the equivalent self-inductance of the AC inductor coil 21. The two DC control windings 22 are respectively arranged on the inner and outer sides of the AC inductor coil 21 with opposite winding directions.

[0042] Taking a dynamic wireless charging system for electric vehicles as an example, in specific implementation, a horizontal distance is set between two adjacent energy transmitting rails according to a first preset value. The energy transmitting coil 1 is wound horizontally as a rectangular coil. The primary side compensation coil 11 is wound inside the energy transmitting coil 1 in the same plane. The energy receiving coil 2 is wound horizontally as a rectangular coil, and the width of its outer contour is the same as the width of the energy transmitting coil 1. The AC inductor coil 21 and the two DC control windings 22 in the variable integrated inductor are wound inside the energy receiving coil 2 in the same plane. Typically, a primary magnetic core plate is provided on the bottom surface of the energy transmitting coil 1 and the primary side compensation coil 11, and a secondary magnetic core plate and a shielding plate are provided on the top surface of the energy receiving coil 2 and the variable integrated inductor.

[0043] In this embodiment, the power transmitting end is provided with a primary-side LCC resonant compensation network, and a primary-side series resonant capacitor is also provided on the energy transmitting coil. The wireless charging energy receiving circuit of the power receiving end also includes a secondary-side resonant capacitor, a secondary-side compensation capacitor, and a rectifier and filter circuit. The secondary-side resonant capacitor, the secondary-side compensation capacitor, and the variable integrated inductor form an LCC compensation network connected to the energy receiving coil. The output of the LCC compensation network is connected to the electrical load through the rectifier and filter circuit.

[0044] Based on the above description, the system circuit structure is as follows: Figure 3 As shown in the figure, U in It is the AC output voltage of the inverter, R L It is the load resistance, and the coil L A1 and L A2 The reverse series connection serves as the first transmitter, corresponding to the energy transmission coil 1 and the primary-side compensation coil 11 in the first energy transmission rail; coil L B1 and L B2 The reverse series connection serves as the second transmitter, corresponding to the energy transmission coil 1 and the primary-side compensation coil 11 in the second energy transmission rail; coil L s and L f3 The integrated coil acts as a receiver, corresponding to energy receiving coil 2 and a variable integrated compensation inductor. All integrated coils and the main coupling coil share the same ferrite core and shielding plate. L1 and L2 are compensation inductors, and C1, C2, and C... f1 C f2 C p1 C p2 C s C f3 It is a compensation capacitor, C o It's a filter capacitor. From Figure 3 It can be seen that there are 15 important coupled mutual inductances in the system, M A1B2 M A1B1 M A2B2 MA2B1 M A1A2 M B1B2 M A1f3 M A2f3 M B1f3 M B2f3 M A1s M A2s M B1s M B2s and M sf3 Among them, M A1B2 M A1B1 M A2B2 M A2B1 M A1A2 M B1B2 M represents the mutual inductance between the two sets of primary windings. A1f3 M A2f3 M B1f3 M B2f3 M A1s M A2s M B1s M B2s M represents the mutual inductance between the secondary and primary windings. sf3 This refers to the mutual inductance between the energy receiving coil 2 and the AC inductor coil 21 in the variable integrated compensating inductor. A and I B It is the current in the transmitting coil, I s I is the current in the receiving coil. I1 and I2 are the inverter output currents of the two compensation networks, respectively. I3 is the load output current, and ω is the system operating frequency.

[0045] For ease of understanding and description, the mutual inductance involved in the system is defined as follows:

[0046]

[0047] M A1A2 and M B1B2 These are the internal mutual inductances directly connected in series in a circuit, and are usually fixed. Therefore, they can be directly equivalent to the following components in the circuit:

[0048]

[0049] At the power receiving end, the internal mutual inductance between the energy receiving coil and the integrated inductor coil is decoupled through an equivalent T-network, such as... Figure 4 As shown, M sf3 The resulting T-shaped network affects the secondary-side circuit impedance, which also provides a way to increase output power.

[0050] L f1 and L f2 The equivalent inductance of the series branches is expressed as follows:

[0051]

[0052] The resonant network is configured as follows:

[0053]

[0054] Furthermore, a simplified circuit considering coupling between coils on the same side, such as... Figure 5 As shown.

[0055] The KVL equations are obtained as follows:

[0056]

[0057] By solving the equations, the values ​​of the transmitting coil current and the output current are calculated as follows:

[0058]

[0059] According to equation (6), the expressions for the energy transmitting coil current and the output current do not include the load R. e Therefore, the constant current output characteristics of the primary and secondary circuits are maintained.

[0060] Considering the symmetry of the system parameters, the corresponding parameters are set as follows:

[0061] L f1 =L f2 =L f (7)

[0062] To make the expression more concise, cross-inductance is represented as:

[0063]

[0064] Output power is expressed as:

[0065]

[0066] Based on the system design and optimization scheme mentioned in our team's previously submitted Chinese invention patent: 202111524549.7, stable M-mode performance can be achieved during the driving of electric vehicles. ABs Output power fluctuations during system rail switching without lateral offset are effectively suppressed.

[0067] Combination Figure 1 The analysis results show that during the movement of the electric vehicle in a lateral offset state, M ABs It remains relatively stable at lower values. According to equation (9), M can be adjusted... sf3 This improves power transmission capability, thereby suppressing the overall output power drop due to lateral offset.

[0068] However, based on the analysis of LCC-LCC wireless power transfer systems with magnetically integrated coupling mechanisms, M sf3 The adjustment will inevitably affect the resonance state. The total output current of the inverter calculated according to equation (5) is expressed as:

[0069]

[0070] Among them, L ses =L se +M sf3 ,L sed =L se -L f3 The total input impedance of the system is expressed as:

[0071]

[0072] The phase angle between the inverter output voltage and current, denoted as γ, can be used to reflect the system's resonant state. Positive and negative values ​​of γ represent inductive and capacitive offsets, respectively. Furthermore, its absolute value indicates the degree to which the system deviates from its resonant point. The tangent of γ is expressed as:

[0073]

[0074] M sf3 and L f3 These parameters are introduced by the integrated inductor coil, and their impact on output power and ZVS conditions requires further in-depth analysis. sf3 The change is defined as ΔM sf3 L f3 The change is defined as ΔL f3 Equation (9) can be rewritten as:

[0075]

[0076] According to equation (13), the output power is affected by ΔM sf3 It can be used as a control variable, as it is unaffected by ΔL. Meanwhile, the output power is unaffected by ΔL. f3 This is consistent with the constant output current characteristics on the secondary side.

[0077] Equation (12) can be rewritten as:

[0078]

[0079] in,

[0080] In equation (14), T1 is the original ZVS configuration part, and T2 is determined by ΔM. sf3 and ΔL f3The resulting additional components, together, constitute the tangent of the system's input impedance angle, determining the resonant state and ZVS condition. To achieve a relatively stable and optimized ZVS operating state for the electric vehicle while driving on the road, T2 is expected to be sufficiently small relative to the configured T1 to be negligible. Therefore, through careful and appropriate design, the output power under lateral offset can be improved, enabling the system to operate under reasonable ZVS conditions.

[0081] Regarding variable integrated inductors, in specific implementations, such as Figure 6 , Figure 7 As shown, the integrated inductor winding N ac Two DC control windings N are wound in the middle position. dc1 and N dc2 The two DC control windings are wound on the inner and outer sides respectively, and a DC power supply is applied to them. The magnetic field generated by the current flowing through the inner DC control winding is in the opposite direction to that of the outer DC control winding.

[0082] Because the magnetic flux generated by the AC winding flows through the DC winding, undesirable voltages are generated in both DC windings. Ideal decoupling between the inductor and control winding is difficult due to nonlinear BH behavior. However, the AC voltage induced in the DC winding can be mitigated by the following measures.

[0083] (1) The two DC windings are respectively close to the inner and outer edges of the AC winding, so that they have a tight coupling characteristic with the AC winding. Therefore, the control of DC power will be more effective.

[0084] (2) The two DC windings are connected in reverse series. Considering the direction of the current, the induced voltage in the intermediate winding is in the opposite direction, which makes the voltages on the DC coils cancel each other out.

[0085] A magnetic core can increase the inductance of a coil; the permeability of the core affects the inductance value. For integrated inductor coils, the magnetic field density is denoted by B, and the magnetic field strength by H. When alternating current passes through the coil, the magnetization process of the magnetic core can be represented by the relationship curve between H and B, as shown in the figure. Figure 8 As shown, the process includes an unsaturated region, a transition region, and a saturated region. Initially, both H and B are zero. In the unsaturated region, H and B exhibit a strong positive correlation and a nonlinear relationship. In the transition region, the increase in B is relatively slow. In the saturated region, B tends to stabilize. The slope of the curve relating H and B reflects the rate of magnetization of the magnetic core, corresponding to the equivalent relative permeability μ of the core material.

[0086] The general definition of inductance is the ratio of total magnetic flux linkage to the current flowing through the winding, expressed as:

[0087]

[0088] In the expression, Ψ(t) is the total magnetic flux linkage. Φ is the magnetic flux, which is positively correlated with the relative permeability of the core material. N is the number of turns of the conductor, and i is the current in the winding. Figure 9 According to Equation (15), the self-inductance of the variable integrated inductor is positively correlated with the equivalent relative permeability of the core material.

[0089] During the operation of a wireless power transfer system, the relative permeability of the magnetic core is typically constant. However, the polarization bias generated by the direct current affects the effective relative permeability of the core. The hysteresis loop magnetized by the polarized DC bias magnetic field is shown in the figure. Figure 9 As shown. When the magnetic core is magnetized by a polarized DC bias magnetic field, AC magnetization produces a small scissor-shaped hysteresis loop. As the AC H-field decreases, the hysteresis loop shrinks to a straight line. The area of ​​this loop is usually small, and its average slope can reflect the incremental relative permeability Δμ:

[0090]

[0091] Based on the definition of inductance and the effect of direct current on the magnetization process of the magnetic core, it is feasible to adjust the inductance by utilizing the permeability of the inductor core.

[0092] Based on the above description, this embodiment also provides the design flow of the magnetically integrated coupling mechanism. During implementation, to reduce the impact of the outer control coil on the self-inductance of the receiving coil and to amplify the effect of the DC magnetic field, the distance between the outer control coil and the energy receiving coil should be sufficiently small. Furthermore, to minimize the total induced voltage on the control coil, the size of the inner control coil should be as close as possible to that of the outer control coil; therefore, the distance between the inner control coil and the receiving coil should also be sufficiently small. Considering all these factors, the design flow of the proposed magnetic coupling mechanism is as follows: Figure 10 As shown.

[0093] First, design the energy receiving coil according to the application scenario requirements and relevant standards, and design the energy transmitting coil by comprehensively considering the coupling coefficient k and the actual application situation.

[0094] Secondly, given the main coupling coil, a primary-side integrated compensation coil is designed to achieve stable M at different receiver positions. ABs .

[0095] Third, based on the energy transmitting coil and the primary-side compensation coil, a secondary-side compensation inductor coil was designed to ensure M at different receiver positions. ABf ≤εM ABs .

[0096] Fourth, the relationship between the number of turns inside and outside the control coil was reasonably set to make the total induced voltage small enough, and a series of turn selection areas for the inner and outer control coils were obtained by traversal method.

[0097] Fifth, based on the expressions for the decrease in mutual inductance and output power during lateral offset, the mutual inductance M was determined. sf3 The range of variation was determined, and the number of turns of the control coil was designed accordingly.

[0098] Sixth, according to L f3 and M sf3 The relationship between the changes is evaluated according to Equation (14) for the ZVS condition.

[0099] Finally, based on the simulation design of the main coupling coil and the integrated coil, a magnetic coupling mechanism model was established in Maxwell to obtain the system parameters.

[0100] The entire system architecture modeled using Maxwell is as follows: Figure 11 and Figure 12 As shown in Table 1, the parameters describing the dimensions of the coupling mechanism are defined. A typical magnetic coupling mechanism with a power rating of 4.5 kW is then constructed to validate the design method.

[0101] Table 1. Parameter Definitions of the Variable Inductor Magnetic Integrated Coupling Mechanism

[0102]

[0103] Regarding the design of the initial magnetic coupling mechanism, the receiver structure is designed according to the SAE J2954 WPT3Z3 VA standard. The dimensions of the energy receiving coil are designed to be "400mm*400mm*5mm", with n turns. w3 The design is 10. Ferrite plates and aluminum shielding plates are used for magnetic field enhancement and electromagnetic shielding, respectively. The secondary-side ferrite plate has dimensions of "400mm*400mm*10mm", and the aluminum shielding plate has dimensions of "420mm*420mm*5mm". The transmission distance is d=150mm.

[0104] Coupling coefficient and cost are typically two key performance indicators (K) for designing the main magnetic coupling mechanism. In this system, the coupling coefficient K is closely related to power transfer performance. For roads of a certain length, shorter transmission coils require more resonant compensation networks, magnetic cores, and more expensive Litz wire. Taking the transmitting coil as an example, the initial magnetic coupling mechanism was designed considering the length, cost, leakage flux, and coupling coefficient of the electric vehicle. In this case, the transmitting coil is designed with dimensions of 900mm * 450mm * 5mm and the number of turns n. w1 The design is 6.

[0105] In this system, cross-coupling between two adjacent transmitting coils will cause the activated coil to be affected by nearby inactive coils. Furthermore, excessive cross-coupling makes it difficult to configure the series resonant compensation capacitor in practical applications. The larger the gap distance, the greater the decrease in mutual inductance and the greater the output power fluctuation. To avoid excessive cross-coupling, the gap distance g between two adjacent transmitting coils should be optimized. The relationship between the coupling coefficient k and the gap distance g is as follows: Figure 13 As shown in the figure. The results indicate that after the gap distance reaches 30 mm, the value of k can be ignored and decreases slowly. Therefore, the gap distance g is set to 30 mm.

[0106] The integrated reverse coil is designed to be 650mm*190mm*5mm in size, with n turns. w2 The design is 3. Results show that output power fluctuations remain within ±4% throughout the entire movement range. Therefore, power fluctuations during the transmit coil transition can be effectively suppressed without lateral offset. Furthermore, a secondary-side integrated compensation inductor was designed to optimize ZVS conditions; the integrated compensation inductor has dimensions of 170mm*170mm*5mm and a number of turns n. w4 The design is 6.

[0107] For the integrated control coil, its addition inevitably introduces coupling with other unipolar coils in the coupling mechanism. Theoretical analysis suggests that these couplings should be reduced. Since the mutual inductance between the integrated control coil and the primary coil is sufficiently small, only the mutual inductance between the integrated control coil and the secondary coil is considered in the design.

[0108] According to Newman's formula, the mutual inductance between the transmitting coil and the receiving coil can be expressed as:

[0109]

[0110] Where N a N b , , where represents the number of coil turns, and represents the micro-element of the transmitting coil and the receiving coil, respectively. ab yes and The distance between them. μ0 is the permeability of vacuum.

[0111] According to equation (17), for a fixed-position integrated control coil, the number of turns is the decisive factor affecting the mutual inductance. Therefore, by iterating through the number of turns of the inner and outer control coils, the mutual inductance between the integrated control coil and the secondary coil under different combinations of turn counts is obtained. The mutual inductance between the integrated compensation inductor coil and the integrated control coil are M... d1f3 and M d2f3 The mutual inductance between the receiving coil and the integrated control coil is M. d1and M d2 The equivalent mutual inductance difference between the integrated compensation inductor and the integrated control inductor is expressed as ΔM. df3 .

[0112] The equivalent mutual inductance difference between the receiving coil and the integrated control coil is expressed as ΔM. ds .

[0113]

[0114] Combination Figure 14 It can be seen that, Figure 14 (a) and Figure 14 (b) shows the number of turns n. dc1 and number of turns n dc2 The mutual inductance difference between the secondary coil and the control coil is affected. Due to I s There is a 90-degree phase difference between I3 and the receiving coil. The total equivalent induced voltage generated by the receiving coil and the integrated inductor coil can be considered as the maximum value of the two. In the prototype design of the 4.5kW power class, in order to ensure a small induced voltage, the maximum value of the composite mutual inductance difference (ΔM) is... df3 ΔM ds The value can be set below 6μH. To determine the selection range of the number of turns of the inner and outer integrated control coils, the composite mutual inductance in the integrated control coils is obtained by superposition and plotted on... Figure 15 In. Figure 15 In the design range, the region with less than 6 μH is marked as the number of turns.

[0115] In theory, the number of turns in a variable integrated inductor can be adjusted within the design range, only the required DC current range varies. To achieve output power control with a smaller control current, the number of turns configuration was further optimized through simulation. According to... Figure 15 Within the design range, when the number of external turns n d1 When n = 3, the selectable number of internal turns is n d2 The range of n is relatively large. Therefore, we first analyze n. d1 The system characteristics of the inner and outer turns when the number of turns is 3, and the self-inductance of the integrated compensation inductor and its mutual inductance with the receiving coil under different combinations of turns are plotted on the diagram. Figure 16 In the diagram, the upper curve represents L. f3 The change, the curve below represents M sf3 The changes revealed two characteristics. First, within the turn configuration of the design range, L f3 and M sf3 It changes effectively with increasing DC current. Second, relative to n d1 n d2 The smaller the value, the corresponding L f3 and M sf3The effect of direct current is more pronounced. Considering the design objectives, as the direct current increases, L is expected to... f3 The change was small, M sf3 The changes are significant. Therefore, in order to achieve a trade-off between self-inductance and mutual inductance, n should be limited within the design scope. d1 and n d2 Set them to be equal.

[0116] When the number of turns on the inner and outer sides is the same, the self-inductance of the integrated compensating inductor and its mutual inductance with the receiving coil are as follows: Figure 17 As shown. The results indicate that different turns configurations correspond to different M values. sf3 Adjustment range. In this example of a 4.5kW power stage prototype design, according to... Figure 16 Mutual inductance drop when the lateral offset y = 10cm is selected, n d1 =5 and n d2 =5.

[0117] To further verify the effect of the polarization bias generated by the DC current on the equivalent relative permeability of the magnetic core, finite element simulation analysis was performed on the flux density modulus distribution of the integrated control coils with different DC current magnitudes. Based on the system parameters designed above, the simulated flux density modulus is as follows: Figure 18 As shown.

[0118] according to Figure 18 Simulation results show that the DC current in the integrated control coil can effectively change the flux density modulus of the integrated inductor coil, altering the relative permeability of the core material and thus changing the self-inductance of the integrated compensation inductor coil. Therefore, the output power fluctuation caused by rail switching under lateral offset conditions in the dynamic wireless charging system for electric vehicles can be effectively suppressed.

[0119] To further verify the correctness of the theoretical analysis, a 4.5kW power-level EVDWC system prototype was constructed. The feasibility of the proposed method was verified through simulation and experiments.

[0120] A system circuit simulation model was established based on the simulation parameters in Table 2. The inverter output voltage and load output voltage were set to 400V. The lateral offset was set to y = 10cm, and a DC current of 10A was passed through the integrated control coil. The inverter output waveforms at different receiving coil positions are shown below. Figure 19 As shown, where v inv It is the inverter output voltage, i invA and i invB This corresponds to the inverter output current of the two transmitters, i inv is i invA and i invBThe total current is superimposed. The inverter's output voltage phase slightly leads the total current, which limits the reactive power circulating in the resonant circuit, thereby reducing power loss, while also ensuring ZVS conditions under normal parameter configuration.

[0121] Table 2 Simulation parameters and calculated resonance parameters of the coupling mechanism

[0122]

[0123] Simulation results of output power and efficiency at different receiving coil positions under a 10cm lateral offset are as follows: Figure 20 As shown in the figure, the efficiency remains high at different receiving positions. Furthermore, the overall output power is improved during the movement of the receiving coil, and power fluctuations due to lateral offset are suppressed. The output power fluctuation is suppressed to within ±5% relative to the average power.

[0124] In summary, based on the above theoretical analysis and simulation verification, this embodiment can also propose a method for suppressing lateral offset power fluctuations in a dynamic wireless charging system for electric vehicles, including the following steps:

[0125] S1: Detects the lateral offset of the electric vehicle;

[0126] S2: Determine the drop in the first mutual inductance coefficient based on the lateral offset. The first mutual inductance coefficient is the mutual inductance coefficient between two adjacent energy transmitting coils and energy receiving coils. In specific implementation, based on the system parameter configuration, a mapping relationship between the lateral offset and the first mutual inductance coefficient is pre-established through simulation.

[0127] S3: The second mutual inductance coefficient is changed by adjusting the drive current of the two DC control windings, so that the system output power is maintained within a predetermined range. The second mutual inductance coefficient is the mutual inductance coefficient between the energy receiving coil and the variable integrated compensation inductor. In specific implementation, the mapping relationship between the drive current of the two DC control windings and the second mutual inductance coefficient is established in advance through simulation.

[0128] In step S3, it can be done according to To determine the relationship between the changes in the first mutual inductance coefficient and the second mutual inductance coefficient, when the lateral displacement of the electric vehicle causes the first mutual inductance coefficient M to change... ABs When changes occur, the second mutual inductance coefficient M is changed by adjusting the drive current of the two DC control windings. sf3 So that the output power P out Maintain within the predetermined range.

[0129] By using the above method, compensation for the drop in system output power can be achieved using a smaller DC current.

[0130] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, and not to limit them. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some or all of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of the present invention, and such transformations should be covered within the scope of the claims and specification of the present invention.

Claims

1. A variable integrated inductance energy receiving coil, characterized in that, A variable integrated inductor is provided on the energy receiving coil. The variable integrated inductor includes an AC inductor coil for the secondary compensation network and two DC control windings for changing the equivalent self-inductance of the AC inductor coil. The two DC control windings are respectively arranged on the inner and outer sides of the AC inductor coil in opposite winding directions; A DC power supply is loaded on the two DC control windings, and the magnetic field directions generated by the currents flowing through the inner DC control winding and the outer DC control winding are opposite; The energy receiving coil is horizontally wound in a rectangular coil, and the AC inductor coil and the two DC control windings in the variable integrated inductor are wound on the inside of the energy receiving coil in the same plane; The variable integrated inductance energy receiving coil is used in the energy receiving circuit of an electric vehicle dynamic wireless charging system. When the lateral offset of the electric vehicle causes the first mutual inductance coefficient to change, the drive current of the two DC control windings is adjusted to change the second mutual inductance coefficient , so that the output power is maintained within a predetermined range; The first mutual inductance coefficient is the mutual inductance coefficient between adjacent two segments of the energy transmitting coil and the energy receiving coil; the second mutual inductance coefficient is the mutual inductance coefficient between the energy receiving coil and the variable integrated inductor.

2. The variable integrated inductor energy receiving coil according to claim 1, wherein: The number of turns of the inner DC control winding is the same as that of the outer DC control winding.

3. The variable integrated inductance energy receiving coil according to claim 1 or 2, characterized in that: A magnetic core plate and a shielding plate are provided on the top surfaces of the energy receiving coil and the variable integrated inductor.

4. The variable integrated inductance energy receiving coil according to claim 1, wherein: The number of turns of the two DC control windings is the same.

5. A wireless charging vehicle, characterized in that: The variable integrated inductor energy receiving coil according to any one of claims 1-4 is adopted in the wireless charging energy receiving circuit.

6. A wireless charging vehicle according to claim 5, wherein: The wireless charging energy receiving circuit further includes a secondary resonant capacitor, a secondary compensation capacitor and a rectifying and filtering circuit. The secondary resonant capacitor and the secondary compensation capacitor form an LCC compensation network with the variable integrated inductor and are connected to the energy receiving coil. The output end of the LCC compensation network is connected to an electrical load through the rectifying and filtering circuit.

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