Vehicle lamp, driving circuit, and driving-integrated light source

Abstract

A lighting circuit (200) lights a plurality of semiconductor light sources (102). A plurality of current sources (2101) to (210N) are connected in series with corresponding semiconductor light sources (102), respectively. A switching converter (220) supplies a drive voltage (VOUT) between both ends of each of a plurality of series connection circuits formed by the plurality of semiconductor light sources (102) and the plurality of current sources (210). A converter controller (230) controls a switching transistor (M1) of the switching converter (220) based on a relationship between a voltage (VLED) between both ends of any one of the plurality of current sources (210) and a reference voltage having a positive correlation with a temperature (Tj).

Description

Technical Field

[0001] This invention relates to lighting circuits. Background Technology

[0002] Vehicle lights typically switch between low beam and high beam. Low beam illuminates the area around the vehicle at a predetermined illuminance, with light distribution specifications designed to avoid glare for oncoming or following vehicles; it is primarily used for driving in urban areas. High beam, on the other hand, illuminates a wider area ahead and further ahead at a higher illuminance; it is mainly used for high-speed driving on roads with less oncoming or following traffic. Therefore, high beam provides better visibility for the driver compared to low beam, but it also presents the problem of glare for drivers of vehicles or pedestrians in front of the vehicle.

[0003] In recent years, an adaptive high beam (ADB) technology has been proposed, which dynamically and adaptively controls the high beam pattern based on the surrounding conditions of the vehicle. ADB technology detects whether there are vehicles, oncoming vehicles, or pedestrians in front of the vehicle and reduces or turns off the light in the area corresponding to the vehicle or pedestrian, thereby reducing glare to vehicles or pedestrians.

[0004] Figure 1 This is a block diagram of a lighting system 1001 with ADB functionality. The lighting system 1001 includes a battery 1002, a switch 1004, a switch converter 1006, multiple light-emitting units 1008_1 to 1008_N and multiple current sources 1010_1 to 1010_N, a converter controller 1012, and a light distribution controller 1014.

[0005] Multiple light-emitting units 1008_1 to 1008_N are semiconductor light sources such as LEDs (light-emitting diodes) or LDs (laser diodes), and are corresponding to multiple different areas on a virtual vertical screen in front of the vehicle. Multiple current sources 1010_1 to 1010_N are connected in series with the corresponding multiple light-emitting units 1008_1 to 1008_N. The driving current ILEDi generated by the current source 1010_i flows through the i-th (1≤i≤N) light-emitting unit 1008i.

[0006] Multiple current sources 1010_1 to 1010_N are configured to be independently turned on and off (or have different current flows). The light distribution controller 1014 controls the on / off (or current flows) of the multiple current sources 1010_1 to 1010_N to obtain the desired light distribution pattern.

[0007] A constant-voltage output switching converter 1006 generates a drive voltage VOUT, which is sufficient to cause multiple light-emitting units 1008_1 to 1008_N to emit light at the desired brightness. Next, considering the i-th channel, the voltage drop (forward voltage) of the light-emitting unit 1008_i when a certain drive current ILEDi flows is denoted as VFi. Furthermore, regarding the current source 1010_i, in order to generate the drive current ILEDi, the voltage across its terminals should be greater than a certain voltage (hereinafter referred to as the saturation voltage VSATi). Thus, the following inequality should hold for the i-th channel.

[0008] VOUT>VFi+VSATi…(1)

[0009] This relationship needs to be established in all channels.

[0010] [Existing technical documents]

[0011] [Patent Literature]

[0012] Patent Document 1: Japanese Patent Application Publication No. 2009-012669

[0013] Patent Document 2: Japanese Patent Application Publication No. 2015-138763 Summary of the Invention

[0014] [The problem the invention aims to solve]

[0015] Question 1.

[0016] In order to make inequality (1) hold under any circumstances, the output voltage VOUT is used as the control object for feedback. As shown in equation (2), the target value VOUT(REF) of the output voltage VOUT is set higher with a margin, and feedback control can be applied so that the output voltage VOUT of the switching converter 1006 is consistent with the target value VOUT(REF).

[0017] VOUT(REF)=VF(MERGIN)+VSAT(MERGIN)…(2)

[0018] VF(TYP) is the maximum (or typical) value of VF with an added margin. VSAT(MERGIN) is the saturation voltage VSAT with an added margin.

[0019] When this control is performed, the difference between the saturation voltage VSAT(MERGIN) and the actual saturation voltage VSAT is applied to the current source 1010, resulting in unnecessary power loss. Furthermore, when the actual forward voltage VF is lower than VF(MERGIN), their difference is included in the voltage drop of the current source 1010, resulting in unnecessary power loss.

[0020] In vehicle lighting, a very large current needs to flow to the light-emitting unit, and compared with other devices, it is difficult to take heat dissipation measures. Therefore, it is necessary to minimize the heat generated in the current source.

[0021] Question 2.

[0022] The spatial resolution of the light distribution pattern formed by the luminaire system 1001 is defined by the number N of the light-emitting units 1008. In high-resolution luminaire systems with N exceeding several hundred, the increase in the number of wires between connecting components or circuits becomes problematic. For example, when the light-emitting units are configured in a matrix of 30x30, it is impractical to connect the light distribution controller 1014 and the multiple light-emitting units 1008 with N = 900 signal lines.

[0023] Figure 22 This diagram illustrates another configuration example of the lighting system 1001. Multiple current sources 1010_1 to 1010_N are integrated into a single semiconductor chip (driver IC) 1020. This driver IC 1020 includes an interface / decoder circuit 1022, which reduces the number of wires between the driver IC 1020 and the light distribution controller 1014 by transmitting control signals for the N current sources 1010_1 to 1010_N in a time-division multiplexing manner. When using high-speed serial communication, the number of wires is reduced to a few.

[0024] The inventors of this invention are targeting Figure 22 The lighting system 1001 was studied, and the following problems were identified as a result.

[0025] exist Figure 1 In a lighting system, when an abnormality occurs in the wiring connecting the light distribution controller 1014 and a current source 1010, the light-emitting unit 1008 corresponding to that wiring becomes uncontrollable. However, in reality, the probability of all N wirings experiencing abnormalities is low, so the system is unlikely to become immediately unusable.

[0026] On the other hand, Figure 22 In the lighting system, all light-emitting units 1008 are controlled by several wires 1030. Therefore, if even one wire 1030 malfunctions, the light-emitting units 1008 will become uncontrollable. Consequently, situations may occur that dazzle oncoming vehicles or vehicles in front, or fail to illuminate necessary areas.

[0027] The same problem may occur with other light fixtures. Figure 23 This is a diagram illustrating another configuration example of a vehicle lighting fixture. The vehicle lighting fixture 2001 includes a lighting ECU 2002, an interface circuit 2010, a local controller 2020, a light distribution variable device 2030, a light source 2040, and a lighting circuit 2050.

[0028] The illumination circuit 2050 illuminates the light source 2040. The variable light distribution device 2030 includes multiple independently controllable elements and forms a light distribution pattern based on the states of these elements. For example, the variable light distribution device 2030 is a DMD (Digital Mirror Device) that reflects the emitted light from the light source 2040, and can control the on and off of reflection on a pixel-by-pixel basis.

[0029] Alternatively, in the bypass-type vehicle lighting fixture 2001, the light source 2040 includes multiple LEDs connected in series. The variable light distribution device 2030 includes multiple bypass switches connected in parallel with the multiple LEDs. A light distribution pattern is formed according to the on / off state of the multiple bypass switches.

[0030] When the light distribution variable device 2030 and the local controller 2020 are mounted on a different substrate 2006 than the luminaire ECU 2002, the luminaire ECU 2002 and the local controller 2020 are not connected by wiring on the printed circuit board, but by communication line 2004. On the substrate 2006, an interface circuit 2010 is provided to receive the control signal S1 from the luminaire ECU 2002.

[0031] The local controller 2020 converts the control signal S1 received by the interface circuit 2010 into a separate control signal S2, which indicates the status of the multiple controllable elements included in the photoluminescence variable device 2030.

[0032] In this situation, if the communication line 2004, connector, or lamp ECU 2002, or the communication line 5 between the upstream vehicle ECU 4 and lamp ECU 2002 malfunctions, the variable light distribution device 2030 will become uncontrollable. This means it may cause glare to oncoming vehicles or vehicles in front, or fail to illuminate necessary areas.

[0033] One aspect of the present invention is addressed in view of problem 1, and one of its exemplary objectives is to provide a lighting circuit that reduces power consumption. Another aspect of the present invention is addressed in view of problem 2, and one of its exemplary objectives is to provide a vehicle lighting fixture that improves safety.

[0034] [Technical solutions used to address technical problems]

[0035] 1. One aspect of the present invention relates to a lighting circuit for illuminating multiple semiconductor light sources. The lighting circuit includes: multiple current sources for series connection with corresponding terminals of the multiple semiconductor light sources; a switching converter for supplying a driving voltage between the terminals of the multiple series-connected circuits formed by the multiple semiconductor light sources and the multiple current sources; and a converter controller for controlling the switching converter based on a relationship between the voltage between the terminals of any one of the multiple current sources and a reference voltage that is positively correlated with temperature.

[0036] 2. One aspect of the present invention relates to a driving circuit for supplying driving current to multiple semiconductor light sources. The driving circuit includes: multiple current sources for being connected in series with corresponding ones of the multiple semiconductor light sources; an interface circuit connected to a processor for receiving control signals indicating the on / off states of the multiple semiconductor light sources, generating multiple individual control signals based on the control signals, and setting the on / off states of the multiple current sources; and a protection circuit for monitoring the communication between the processor and the interface circuit, and forcibly setting the multiple current sources to a predetermined state when an abnormality is detected.

[0037] Another aspect of the present invention relates to a vehicle lighting fixture. The vehicle lighting fixture includes: a host controller; an interface circuit that receives control signals from the controller; a variable light distribution device; a local controller that controls the variable light distribution device based on the control signals received by the interface circuit; and a protection circuit that monitors the communication between the processor and the interface circuit, and when an anomaly is detected, forces a predetermined pattern onto the variable light distribution device.

[0038] Furthermore, any combination of the above-mentioned constituent elements, or the result of interchanging the constituent elements or expressions of the present invention among methods, apparatuses, systems, etc., is also an effective solution of the present invention.

[0039] [Invention Effects]

[0040] According to one aspect of the present invention, power consumption can be reduced. Furthermore, according to one aspect of the present invention, functional safety can be ensured in the event of an anomaly within the vehicle lighting fixture. Attached Figure Description

[0041] Figure 1 This is a block diagram of a lighting system with ADB functionality.

[0042] Figure 2 This is a block diagram of a lighting system including the vehicle lighting fixtures of embodiment 1.

[0043] Figure 3 yes Figure 2 Waveform diagram of the operation of vehicle lights.

[0044] Figure 4(a) is a waveform diagram of the cathode voltage VLED in Embodiment 1. Figure 4 (b) is a waveform of the cathode voltage VLED in the comparison technique.

[0045] Figure 5 This is a circuit diagram illustrating an example of a configuration with multiple current sources.

[0046] Figure 6 (a) is a graph representing the I-V characteristics of a MOS transistor. Figure 6 (b) is a graph showing the relationship between temperature and overdrive voltage.

[0047] Figure 7 This is a circuit diagram of the converter controller in Example 1-1.

[0048] Figure 8 This is a circuit diagram illustrating an example of a voltage generating circuit that generates the lower limit voltage VBOTTOM.

[0049] Figure 9 This is a circuit diagram representing another example of a voltage generating circuit that generates the lower limit voltage VBOTTOM.

[0050] Figure 10 This is a circuit diagram of the converter controller in Examples 1-2.

[0051] Figure 11 This is a circuit diagram of the converter controller in Examples 1-3.

[0052] Figure 12 This is a circuit diagram of the converter controller in Examples 1-4.

[0053] Figure 13 This is a circuit diagram of the converter controller in Examples 1-5.

[0054] Figure 14 This is a circuit diagram of the converter controller in Examples 1-6.

[0055] Figure 15 yes Figure 14 The specific circuit diagram of the converter controller.

[0056] Figure 16 This is a circuit diagram of a modified example of a conduction signal generation circuit.

[0057] Figure 17 (a) to (c) are circuit diagrams representing examples of current source configurations.

[0058] Figure 18 This is a circuit diagram of the current driver IC and its peripheral circuits in Implementation Method 2.

[0059] Figure 19This is a waveform diagram of the operation of a current driver IC.

[0060] Figure 20 It is a plan view and a cross-sectional view of the integrated light source driver.

[0061] Figure 21 This is the circuit diagram of the vehicle lighting fixtures in variations 1-3.

[0062] Figure 22 This is a diagram showing another example of the configuration of a lighting system.

[0063] Figure 23 This is a diagram illustrating another example of the components of a vehicle lighting system.

[0064] Figure 24 This is a block diagram of a lighting system including the vehicle lighting fixtures of embodiment 3.

[0065] Figure 25 This is a circuit diagram illustrating an example of a configuration with multiple current sources.

[0066] Figure 26 This is a plan view of an LED chip.

[0067] Figure 27 This is a block diagram illustrating an example of the configuration of an interface circuit and a protection circuit.

[0068] Figure 28 of (a), Figure 28 (b) is a circuit diagram showing an example of the configuration of a data substitution circuit.

[0069] Figure 29 It is a plan view and a cross-sectional view of the integrated light source driver.

[0070] Figure 30 This is the circuit diagram of the vehicle lighting fixture in variation 3.1.

[0071] Figure 31 This is a block diagram of the vehicle lighting fixture according to embodiment 4.

[0072] Figure 32 This is a block diagram of a vehicle lighting fixture as shown in Variation 4.1.

[0073] Figure 33 This is a block diagram of a vehicle lighting fixture as shown in Variation 4.2. Detailed Implementation

[0074] (Summary of the implementation method)

[0075] 1. One embodiment disclosed in this specification relates to a lighting circuit configured to illuminate multiple semiconductor light sources. The lighting circuit includes: multiple current sources, each connected in series with one of the multiple semiconductor light sources; a switching converter that supplies a drive voltage between the two ends of the multiple series-connected circuits formed by the multiple semiconductor light sources and the multiple current sources; and a converter controller that controls the switching converter based on a relationship between the voltage between the two ends of any one of the multiple current sources and a reference voltage that is positively correlated with temperature.

[0076] By specifying the reference voltage as the minimum level that ensures the current source will generate a predetermined amount of drive current, the power loss of the current source can be reduced. When the reference voltage is set to a constant value independent of temperature, (i) when the reference voltage is set higher, the predetermined amount of drive current can be maintained even at high temperatures, but the losses increase at low temperatures; conversely, (ii) when the reference voltage is set lower, the losses decrease at low temperatures, but the predetermined amount of drive current cannot be maintained at high temperatures, and the brightness of the semiconductor light source decreases. In contrast, by making the reference voltage change in a positive correlation with temperature, both low power consumption at low temperatures and maintenance of drive current at high temperatures can be achieved.

[0077] Alternatively, the converter controller may turn on the switching transistor of the switching converter in response to the voltage across any one of the multiple current sources dropping to the reference voltage.

[0078] Alternatively, the current source can be a current mirror circuit. In this case, power consumption can be reduced by making the reference voltage have the temperature dependence of the pinch-off voltage (overdrive voltage) of the MOS transistor constituting the current mirror circuit or the collector saturation voltage VCE(sat) of the bipolar transistor.

[0079] Alternatively, the converter controller may include a constant voltage source that generates a constant voltage and a correction current source that generates a correction current that is positively correlated with temperature, with the reference voltage being the voltage obtained by adding the offset voltage, which is proportional to the correction current, to the constant voltage.

[0080] Alternatively, the correction current can be generated by utilizing the temperature dependence of the forward voltage of the PN junction.

[0081] Alternatively, the correction current source may include: at least one diode and a resistor connected in series; and a current mirror circuit that replicates the current flowing into the diode and generates a correction current.

[0082] Alternatively, multiple semiconductor light sources can be integrated into a first semiconductor chip, and multiple current sources can be integrated into a second semiconductor chip. The surfaces of the first and second semiconductor chips are bonded together and modularized in a package.

[0083] In one implementation, the lighting circuit can be installed in vehicle lights.

[0084] 2. One embodiment disclosed in this specification relates to a drive circuit for supplying drive current to multiple semiconductor light sources. The drive circuit includes: multiple current sources, each connected in series with a corresponding one of the multiple semiconductor light sources; an interface circuit connected to a processor, receiving control signals indicating the on / off state of each of the multiple semiconductor light sources, generating multiple individual control signals based on the control signals, and setting the on / off state of the multiple current sources; and a protection circuit that monitors the communication between the processor and the interface circuit, and when an abnormality is detected, forcibly sets the multiple current sources to a predetermined state.

[0085] According to this embodiment, a predetermined light distribution pattern can be formed by setting multiple current sources to a predetermined state when the control signal cannot be transmitted normally, and functional safety can be ensured in the event of an anomaly within the lamp. More specifically, it can suppress glare from oncoming or preceding vehicles while ensuring the light distribution required for the vehicle's operation, thereby achieving a balance between high resolution and high safety.

[0086] Alternatively, the predetermined state can correspond to the light distribution of the low beam.

[0087] Alternatively, the protection circuit may include: an anomaly detection circuit that enables an anomaly detection signal when an anomaly is detected; and a data replacement circuit that directly outputs multiple individual control signals when the anomaly detection signal is invalid, and outputs a combination of predetermined values ​​when the anomaly detection signal is enabled.

[0088] Alternatively, the data replacement circuit may include: an inverter that inverts the anomaly detection signal to generate an inverted anomaly detection signal; multiple first logic gates corresponding to multiple current sources that should be turned on in an abnormal state; and multiple second logic gates corresponding to multiple current sources that should be turned off in an abnormal state. Alternatively, each first logic gate may receive a separate control signal corresponding to one of the anomaly detection signal and the inverted anomaly detection signal, and supply its output to the corresponding current source; each second logic gate may receive a separate control signal corresponding to the other of the anomaly detection signal and the inverted anomaly detection signal, and supply its output to the corresponding current source.

[0089] Alternatively, the protection circuit can determine an anomaly if the control signal does not undergo a level transition within a certain period. This allows for the detection of communication anomalies using a simple circuit.

[0090] One embodiment disclosed in this specification relates to a vehicle lighting fixture. The vehicle lighting fixture includes: a host controller; an interface circuit that receives control signals from the host controller; a variable light distribution device; a local controller that controls the variable light distribution device based on the control signals received by the interface circuit; and an anomaly detection unit that monitors the communication between the host controller and the interface circuit to detect anomalies. The vehicle lighting fixture is configured such that, when an anomaly is detected, a predetermined pattern is forcibly set on the variable light distribution device.

[0091] According to this embodiment, a desired light distribution pattern corresponding to the pattern provided on the variable light distribution device can be formed even when the control signal cannot be transmitted normally. Therefore, while ensuring the light distribution required for the vehicle's operation, glare from oncoming or preceding vehicles can be suppressed, thereby improving safety.

[0092] Alternatively, the anomaly detection unit can also monitor the output of the local controller to detect anomalies. This ensures functional safety in the event of anomalies in the local controller or in the event of communication failures between the local controller and the variable light distribution device.

[0093] (Implementation Method)

[0094] Hereinafter, the present invention will be described with reference to the accompanying drawings and based on preferred embodiments. The same or equivalent constituent elements, components, and processes shown in the various drawings are labeled with the same reference numerals, and repeated descriptions are omitted where appropriate. Furthermore, the embodiments are not intended to limit the invention, but are merely illustrative; not all features or combinations thereof described in the embodiments are substantive aspects of the invention.

[0095] In this specification, the term "the state of connection between component A and component B" includes not only the case where component A and component B are physically directly connected, but also the case where component A and component B are indirectly connected via other components, without substantially affecting their electrical connection state or impairing the function or effect achieved through their connection.

[0096] Similarly, the term "the state in which component C is positioned between component A and component B" includes not only the case where component A and component C, or component B and component C are directly connected, but also the case where they are indirectly connected via other components, without substantially affecting their electrical connection state, or without impairing the function or effect achieved through their combination.

[0097] Furthermore, in this specification, the reference numerals on electrical signals such as voltage signals and current signals, or circuit elements such as resistors and capacitors, are used to indicate their respective voltage, current, resistance, or capacitance values ​​as needed.

[0098] <Implementation Method 1>

[0099] Figure 2 This is a block diagram of a lighting system 1 including the vehicle lighting fixture 100 according to embodiment 1. The lighting system 1 includes a battery 2, a vehicle ECU (Electronic Control Unit) 4, and the vehicle lighting fixture 100. The vehicle lighting fixture 100 is a variable beam headlight with ADB function, forming a beam pattern corresponding to the control signal from the vehicle ECU 4.

[0100] The vehicle lighting fixture 100 includes multiple (N≥2) semiconductor light sources 102_1 to 102_N, a lighting ECU 110, and a lighting circuit 200. For the semiconductor light sources 102, LEDs are preferred, but other light-emitting elements such as LDs or organic ELs can also be used. Each semiconductor light source 102 may also contain multiple light-emitting elements connected in series and / or in parallel. Furthermore, the number of channels N is not limited to a complex number and can also be 1.

[0101] The lighting ECU 110 includes a switch 112 and a microcomputer 114. The microcomputer (processor) 114 is connected to the vehicle ECU 4 via a bus such as CAN (Controller Area Network) or LIN (Local Interconnect Network), and can receive on / off indications or other information. In response to an on / off indication from the vehicle ECU 4, the microcomputer 114 turns on the switch 112. This allows the power supply voltage (battery voltage VBAT) from the battery 2 to be supplied to the lighting circuit 200.

[0102] In addition, the microcomputer 114 receives a control signal from the vehicle ECU 4 indicating the light distribution pattern and controls the lighting circuit 200. Alternatively, the microcomputer 114 may receive information from the vehicle ECU 4 indicating the situation in front of the vehicle and generate its own light distribution pattern based on that information.

[0103] The lighting circuit 200 supplies driving currents ILED1 to ILEDN to multiple semiconductor light sources 102_1 to 102_N to obtain the desired light distribution pattern.

[0104] The lighting circuit 200 includes multiple current sources 210_1 to 210_N, a switching converter 220, and a converter controller 230. The current sources 210_i (i = 1, 2, ... N) are connected in series with the corresponding semiconductor light source 102_i to stabilize the driving current ILEDi flowing through the semiconductor light source 102_i into a constant current driver with a predetermined current amount.

[0105] Multiple current sources 210_1 to 210_N are configured to be individually controlled to turn on and off according to the PWM signals SPWM1 to SPWMN generated by the light distribution controller 116. When the PWM signal SPWMi is at the on level (e.g., "high"), the drive current ILEDi flows, and the semiconductor light source 102_i is lit. When the PWM signal SPWMi is at the off level (e.g., "low"), the drive current ILEDi becomes zero, and the semiconductor light source 102_i is turned off. By changing the duty cycle of the PWM signal SPWMi, the effective brightness of the semiconductor light source 102_i is changed (PWM dimming).

[0106] The switching converter 220 supplies a drive voltage VOUT between the two ends of the series connection circuit between the semiconductor light source 102 and the current source 210. The switching converter 220 is a buck converter and includes a switching transistor M1, a rectifier diode D1, an inductor L1, and an output capacitor C1.

[0107] To maintain the drive current ILED at a predetermined amount, the voltage VCS across current source 210 should be greater than a certain threshold (referred to as the saturation voltage VSAT). Converter controller 230 monitors temperature Tj and controls switching converter 220 based on the relationship between the voltage VCS across any one of the multiple current sources 210 and a reference voltage (hereinafter referred to as the lower limit voltage VBOTTOM) that has a positive correlation with temperature Tj. The lower limit voltage VBOTTOM is set to be substantially the same as or slightly larger than the saturation voltage VSAT.

[0108] The converter controller 230 uses ripple control and, in response to the voltage VCS across any one of the multiple current sources 210 dropping to the lower limit voltage VBOTTOM, turns on the switching transistor M1 of the switching converter 220. Figure 2 In this circuit, one end of the current source 210 is grounded, so the voltage VLED of the connection node between the current source 210 and its corresponding semiconductor light source 102 becomes the voltage VCS between the two ends of the current source 210.

[0109] Furthermore, when a predetermined cutoff condition is met, the converter controller 230 will transfer the control pulse S1 to the cutoff level (high level), causing the switching transistor M1 to turn off. Alternatively, the cutoff condition can be that the output voltage VOUT of the switching converter 220 reaches a predetermined upper limit voltage VUPPER.

[0110] The above describes the composition of the vehicle lighting fixture 100. Next, its operation will be explained.

[0111] Figure 3 yes Figure 2 The waveform diagram of the operation of the vehicle lighting fixture 100 is shown below. For ease of understanding, N = 3 is assumed. Furthermore, it is assumed that component deviations of the multiple current sources 210_1 to 210_N can be ignored. Furthermore, it is assumed that VF1 > VF2 > VF3 holds true due to component deviations of the semiconductor light source 102. Furthermore, for ease of understanding, PWM dimming is not performed.

[0112] During the off period of switching transistor M1 (represented by "low" in the diagram), the output capacitor C1 of the switching converter 220 discharges through the sum of the drive currents ILED1 to ILED3, i.e., the load current IOUT, and the output voltage VOUT decreases over time. In reality, the output capacitor C1 charges or discharges based on the difference between the coil current IL flowing through inductor L1 and the load current; therefore, the increase or decrease of the output voltage VOUT may not coincide with the on / off state of switching transistor M1 on the time axis.

[0113] The voltage between the two ends of the current source 210, that is, the voltage (cathode voltage) of the connection node between the current source 210 and the semiconductor light source 102, is expressed by the following formula.

[0114] VLED1 = VOUT - VF1

[0115] VLED2 = VOUT - VF2

[0116] VLED3 = VOUT - VF3

[0117] Therefore, VLED1 to VLED3 are varied while maintaining a constant potential difference with the output voltage VOUT. Because the forward voltage VF1 of the first channel is the largest, the cathode voltage VLED1 of the first channel will become the lowest.

[0118] In the first channel, when the cathode voltage VLED1 drops to the lower limit voltage VBOTTOM, the switching transistor M1 will turn on.

[0119] When the switching transistor M1 turns on, the coil current IL flowing through the inductor L1 increases, and the output voltage VOUT rises. When the output voltage VOUT reaches the upper limit voltage VUPPER, the switching transistor M1 turns off. The lighting circuit 200 repeats this operation.

[0120] The above describes the operation of the lighting circuit 200. According to this lighting circuit 200, the voltage across the current source 210 can be maintained near the minimum level required to generate the predetermined drive current ILED, thereby reducing power consumption.

[0121] As another method (comparative technique), feedback control is considered, which uses an error amplifier to make the lowest voltage of the cathode voltages VLED1 to VLEDN close to the predetermined target voltage VREF.

[0122] Figure 4 (a) is a waveform diagram of the cathode voltage VLED in Embodiment 1. Figure 4 (b) is a waveform diagram of the cathode voltage VLED in the comparison technique. The cathode voltage VLED shown here is the lowest voltage VMIN among multiple cathode voltages.

[0123] In the comparison technique, based on the response characteristics of the phase compensation filter set in the feedback loop, the average of the lowest voltage VMIN of the cathode voltages VLED1 to VLEDN will approach the target voltage VREF. That is, the lower limit level VMIN_BOTTOM of the lowest voltage VMIN is lower than the target voltage VREF. The difference between this lower limit level VMIN_BOTTOM and the target voltage VREF will vary depending on the conditions and is uncertain. For stable circuit operation, in Figure 4 In (b), as shown by the solid line, a large differential ΔV is envisioned, thus requiring a higher VREF specification. However, as shown by the dashed line, when the differential ΔV' is small, the cathode voltage VLED will be higher than the lower limit voltage VBOTTOM, consuming unnecessary power in the current source. According to embodiment 1, as... Figure 4 As shown in (a), the lower limit level of the cathode voltage VLED can be made close to the lower limit voltage VBOTTOM, thus further reducing power consumption compared to the comparison technique.

[0124] The current source 210 includes a transistor connected in series with the semiconductor light source 102. By adjusting the voltage and current at the control terminal (gate or base) of the transistor, the drive current ILED is maintained at a certain amount. Figure 5This is a circuit diagram illustrating an example of multiple current sources 210_1 to 210_N. Current sources 210_# (# = 1, 2, ..., N) are constructed using a current mirror circuit 216 containing transistors M31 and M32. A reference current source 218 is connected to transistor M31 on the input side, amplifying the reference current IEF by a reflection factor (K) to obtain a drive current ILED that flows through transistor M32 on the output side. Transistor M33 is configured to control the on / off state of current sources 210 and is positioned between the gate, drain, and ground of transistor M31. When the SPWM# signal is "high," transistor M33 is on, current source 210 is off, and the drive current ILED becomes zero. Conversely, when the SPWM# signal is "low," transistor M33 is off, current source 210 is on, and the drive current ILED flows.

[0125] Figure 6 (a) is a graph representing the I-V characteristics of a MOS transistor. Figure 6 (b) is a graph showing the relationship between temperature and overdrive voltage. The drain current ID flowing through the MOS transistor is constant in the region where the drain-source voltage VDS is higher than the overdrive voltage VOD (or pinch-off voltage, or simply saturation voltage VDS(ast)). The overdrive voltage VOD is the difference between the gate-source voltage VGS and the threshold voltage VGS(th).

[0126] VOD = VGS - VGS(th)

[0127] Because the threshold voltage VGS(th) is negatively correlated with temperature, the overdrive voltage VOD is related to... Figure 6 The temperatures shown in (b) are positively correlated. If a bipolar transistor is used instead of a MOS transistor, the overdrive voltage VOD can be replaced with the collector-emitter saturation voltage VCE(sat).

[0128] Depend on Figure 5 The voltage VCS between the two terminals of the operating point of the current source 210, which is formed by the current mirror circuit, needs to be determined in a range higher than the overdrive voltage VOD. In other words, the overdrive voltage VOD will become... Figure 5 The saturation voltage VSAT of the current source 210.

[0129] In this embodiment 1, the lower limit voltage VBOTTOM is varied in a manner that is positively correlated with the temperature Tj. Therefore, the lower limit voltage VBOTTOM can follow the change in the saturation voltage VSAT (i.e., the overdrive voltage VOD) accompanying temperature variations.

[0130] As a comparative technique, an investigation was conducted on making the lower limit voltage VBOTTOM a constant value independent of temperature. For example, the maximum rated junction temperature Tj of the semiconductor light source 102 is 150°C. When the semiconductor light source 102 is close to the current source 210 or is mounted on a common heat sink, the maximum rated junction temperature Tj of the semiconductor light source 102 is also 150°C. In this case, (i) when the lower limit voltage VBOTTOM is set to the overdrive voltage V150 at Tj = 150°C, a predetermined amount of drive current can be maintained even at high temperatures up to 150°C. However, the junction temperature Tj of the current source 210 during actual operation is lower than 150°C. Therefore, when the difference between the overdrive voltage VOP corresponding to the junction temperature during actual operation and the overdrive voltage V150 at 150°C is denoted as ΔVLOSS, an additional power loss of ILED × ΔVLOSS will occur in the current source 210, and the heat generation will increase. If left unattended, the permissible rated ambient temperature will decrease, reducing the marketability of the vehicle lighting fixture 100. Adding larger heat sinks or fans to address this issue would increase costs.

[0131] Conversely, when the lower limit voltage VBOTTOM is determined based on, for example, the junction temperature TOP during normal operation, the losses in Tj=TOP can be reduced, but even at high temperatures (Tj≈150°C), the predetermined drive current ILED cannot be maintained, and the brightness of the semiconductor light source 102 will decrease.

[0132] Compared to the comparative technique, according to Embodiment 1, the optimal lower limit voltage VBOTTOM can be set according to temperature, thereby reducing losses at low temperatures and suppressing the decrease in brightness at high temperatures.

[0133] This invention can be used as Figure 2 The invention is understood in terms of block diagrams or circuit diagrams, or relates to devices, circuits, and methods derived from the above description, and is not limited to a specific configuration. The following description is not intended to narrow the scope of the invention, but rather to aid in understanding the nature of the invention or the operation of the circuits and to clarify them, by providing more specific configuration examples or variations.

[0134] (Example 1-1)

[0135] Figure 7This is a circuit diagram of the converter controller 230F in Example 1-1. The turn-on signal generation circuit 240F includes multiple comparators 252_1 to 252_N and a logic gate 254. Comparator 252_i compares the corresponding cathode voltage VLEDi with the lower limit voltage VBOTTOM. When VLEDi < VBOTTOM, a comparison signal is generated that is set to active (e.g., "high"). Logic gate 254 performs logical operations on the outputs (comparison signals) SCMP1 to SCMPN of the multiple comparators 252_1 to 252_N. When at least one comparison signal is set to active, the turn-on signal SON is set to active. In this example, logic gate 254 is an OR gate.

[0136] The cutoff signal generation circuit 260F generates a cutoff signal SOFF, which specifies the timing for turning off the switching transistor M1. The voltage divider circuit 261 divides the output voltage VOUT, scaling it to an appropriate voltage level. The comparator 262 compares the divided output voltage VOUT' with a threshold voltage VUPPER' obtained by scaling the upper limit voltage VUPPER. When VOUT > VUPPER is detected, the cutoff signal SOFF is set to active (e.g., "high").

[0137] Logic circuit 234 is, for example, an SR flip-flop. In response to the activation of the on signal SON, it switches its output Q to the on level (e.g., "high"), and in response to the activation of the off signal SOFF, it switches its output Q to the off level (e.g., "low"). Furthermore, when the on signal SON and the off signal SOFF are activated simultaneously, in order to place the switch-converter in a safer state (i.e., the off state of the switching transistor M1), logic circuit 234 is preferably configured as a reset-priority flip-flop.

[0138] Driver 232 drives switching transistor M1 according to the output Q of logic circuit 234. For example... Figure 2 As shown, when the switching transistor M1 is a P-channel MOSFET, the output of the driver 232, i.e. the control pulse S1, is a low voltage (VBAT-VG) when the output Q is on and a high voltage (VBAT) when the output Q is off.

[0139] Figure 8This is a circuit diagram illustrating an example of a voltage generating circuit 280 that generates a lower limit voltage VBOTTOM. The voltage generating circuit 280 includes a constant voltage source 282, a correction current source 284, and a resistor R51. The constant voltage source 282 generates a constant voltage VCONST. The correction current source 284 generates a correction current ICOMMP that is positively correlated with the junction temperature Tj. The lower limit voltage VBOTTOM is the voltage obtained by summing a bias voltage VOFS, which is proportional to the correction current ICOMMP, with the constant voltage VCONST. For example, it is also possible to connect one end of resistor R51 to the output of the constant voltage source 282, allowing the correction current ICOMMP to flow in, thereby creating a voltage drop ICOMMP×R51 across resistor R51, and then extracting the voltage across the other end of resistor R51.

[0140] VBOTTOM = VCONST + ICOM P × R51

[0141] Alternatively, in this example, the current-pull correction current source 284 can be connected to the high potential side of resistor R51, while the current-sink correction current source 284 can be connected to the low potential side of resistor R51.

[0142] The configuration of the correction current source 284 is not limited, but for example, it could utilize the temperature dependence of the forward voltage Vf of the PN junction to generate the correction current ICOMMP. Figure 8 In the calibration current source 284, multiple diodes D51-D53 and a resistor R52 are connected in series; and a current mirror circuit 286 replicates the current It flowing through the multiple diodes D51-D53. When the forward voltage of the diode is denoted as Vf,

[0143] It=(VCC-VBE-3×Vf) / R51.

[0144] When the approximation is Vf≈VBE

[0145] It = (VCC - 4 × VBE) / R51.

[0146] Because VBE is negatively correlated with temperature, the current It is positively correlated with temperature.

[0147] Figure 9This is a circuit diagram illustrating another example of a voltage generating circuit 280 that generates the lower limit voltage VBOTTOM. The voltage generating circuit 280 includes a current source 288, a MOS transistor M61, an emitter follower circuit 290, and a resistor network 292. The MOS transistor M61 is positioned in the path of the current IC generated by the current source 288. The gate of the MOS transistor M61 is biased, causing it to operate in the linear region. For example, at the gate, a power supply voltage VCC is applied via resistor R61. The drain voltage VD is then proportional to the on-resistance RON.

[0148] VD = RON × IC

[0149] The on-resistance RON of a MOS transistor is positively correlated with temperature, therefore the drain voltage VD is also positively correlated with temperature. The drain voltage VD is input to the emitter follower circuit 290. The emitter follower circuit 290 is a two-stage circuit consisting of PNP and NPN types. The base-emitter voltage VBE is canceled out, and the output voltage of the emitter follower circuit 290 becomes equal to the drain voltage VD of transistor M61.

[0150] Resistor network 292 performs a weighted sum (average) of the drain voltage VD, which has a positive temperature coefficient, and the constant voltage VCONST to generate the lower limit voltage VBOTTOM. As a result, the lower limit voltage VBOTTOM has a positive temperature coefficient.

[0151] (Examples 1-2)

[0152] Figure 10 This is a circuit diagram of the converter controller 230G of Embodiments 1-2. The turn-on signal generation circuit 240G includes a minimum value circuit 256 and a comparator 258. The minimum value circuit 256 outputs a voltage VMIN corresponding to the smallest of a plurality of cathode voltages VLED1 to VLEDN. The minimum value circuit 256 can use known techniques. The comparator 258 compares the voltage VMIN with a threshold VBOTTOM' corresponding to the lower limit voltage VBOTTOM. When VMIN < VBOTTOM', the turn-on signal SON is set to active (e.g., "high"). The lower limit voltage VBOTTOM' is positively correlated with temperature. VBOTTOM' = f(VBOTTOM) holds when the relationship between VMIN and the smallest Va among VLED1 to VLEDN is expressed as a function VMIN = f(Va).

[0153] In Example 1-1, with a large number of channels, the circuit area of ​​the comparator group is large, resulting in a larger chip size. In contrast, according to Example 1-2, since only one comparator is needed, the circuit area can be reduced.

[0154] (Examples 1-3)

[0155] Figure 11 This is a circuit diagram of the converter controller 230H in Examples 1-3. In this example, feedback control is applied to the upper limit voltage VUPPER to keep the switching frequency of the switching transistor M1 constant.

[0156] In addition to comparator 262, cutoff signal generation circuit 260H also includes frequency detection circuit 264 and error amplifier 266. Frequency detection circuit 264 monitors the output Q or control pulse S1 of logic circuit 234 and generates a frequency detection signal VFREQ representing the switching frequency. Error amplifier 266 amplifies the error between the frequency detection signal VFREQ and the reference voltage VFREQ(REF) of the target value of the specified switching frequency, generating an upper limit voltage VUPPER corresponding to the error.

[0157] According to Examples 1-3, the switching frequency can be stabilized at the target value, thus making noise countermeasures easier.

[0158] (Examples 1-4)

[0159] Figure 12 This is a circuit diagram of the converter controller 230I in Examples 1-4. Alternatively, after turning on the switching transistor M1, the converter controller 230I turns the switching transistor M1 off after a conduction time TON. That is, the turn-off condition can also be the conduction time TON from the time the switching transistor M1 turns off.

[0160] The cutoff signal generation circuit 260I includes a timer circuit 268. The timer circuit 268 responds to the turn-on signal SON, begins measuring a predetermined on-time TON, and after the on-time TON has elapsed, sets the cutoff signal SOFF to active (e.g., "high"). The timer circuit 268 can be constructed, for example, from a monostable multivibrator (single-trigger pulse generator), a digital counter, or an analog timer. Alternatively, to detect the timing of the switching transistor M1's transition, instead of the turn-on signal SON, the output Q of the logic circuit 234 or a control pulse S1 can be input to the timer circuit 268.

[0161] (Examples 1-5)

[0162] Figure 13This is a circuit diagram of the converter controller 230J in Examples 1-5. Similar to Examples 1-4, the converter controller 230J turns on the switching transistor M1 and then turns it off after a conduction time TON. The OR gate 241 acts as a conduction signal generator, generating a conduction signal SON. The timer circuit 268, such as a monostable multivibrator, generates a pulse signal SP that is high for a predetermined conduction time TON from when the conduction signal SON is active, and supplies it to the driver 232. Furthermore, during startup, considering that VG1 to VGN do not exceed the threshold of the OR gate 241, an OR gate 231 is added, and the logic sum SP' of the conduction signal SON and the output SP of the timer circuit 268 is supplied to the driver 232.

[0163] (Examples 1-6)

[0164] Figure 14 This is a circuit diagram of the converter controller 230K in Examples 1-6. The cutoff signal generation circuit 260K provides feedback control over the on-time TON, ensuring a constant switching frequency. The variable timer circuit 270 is a monostable multivibrator that generates a pulse signal SP that is high during the on-time TON, starting from when the on-signal SON is activated. The on-time TON is configured to vary according to the control voltage VCTRL.

[0165] For example, the variable timer circuit 270 may include: a capacitor; a current source that charges the capacitor; and a comparator that compares the voltage of the capacitor with a threshold. In the variable timer circuit 270, at least one of the current generated by the current source or the threshold is configured to vary according to the control voltage VCTRL.

[0166] The frequency detection circuit 272 monitors the output Q or control pulse S1 of the logic circuit 234 and generates a frequency detection signal VFREQ representing the switching frequency. The error amplifier 274 amplifies the error between the frequency detection signal VFREQ and the reference voltage VFREQ(REF) of the target value of the specified switching frequency and generates a control voltage VCTRL corresponding to the error.

[0167] According to Examples 1-6, the switching frequency can be stabilized at the target value, thus making noise countermeasures easier.

[0168] Figure 15 yes Figure 14The specific circuit diagram of the converter controller 230K is shown below. Next, the operation of the frequency detection circuit 272 will be explained. Capacitor C11 and resistor R11 form a high-pass filter, which can be regarded as a differentiating circuit that differentiates the output of OR gate 231 (or control pulse S1), or as an edge detection circuit that detects the edge of the pulse signal SP'. Transistor Tr11 turns on when the output of the high-pass filter exceeds the threshold, i.e., when the rising edge of the pulse signal SP' is generated, causing capacitor C12 to discharge. During the period when transistor Tr11 is off, capacitor C12 is charged through resistor R12. The voltage VC12 of capacitor C12 becomes a ramp synchronized with the pulse signal SP', and the duration and height of the ramp portion vary according to the period of the pulse signal SP'.

[0169] Transistors Tr12 and Tr13, resistors R13 and R14, and capacitor C13 form a peak hold circuit to maintain the peak value of the voltage VC12 across capacitor C12. The output VFREQ of the peak hold circuit is related to the period of the pulse signal SP', in other words, it is frequency-dependent.

[0170] Comparator COMP1 compares the frequency detection signal VFREQ with the reference signal VFREQ(REF) representing the target frequency. Resistor R15 and capacitor C14 form a low-pass filter, smoothing the output of comparator COMP1 and generating the control voltage VCTRL. The control signal VCTRL is output via buffer BUF1.

[0171] Next, the variable timer circuit 270 will be described. The on signal SON is inverted by the inverter 273. When the inverted on signal #SON is below the threshold VTH1, in other words, when the on signal SON is "high", the output of comparator COMP2 is "high", the trigger SRFF is set, and the pulse signal SP is "high".

[0172] During the period when the pulse signal SP is "high", transistor M21 is off. During the off period of transistor M21, current source 271 generates a variable current IVAR corresponding to the control voltage VCTRL, charging capacitor C12. When the voltage VC15 of capacitor C15 reaches the threshold VTH2, the output of comparator COMP3 is "high", the flip-flop SRFF is reset, and the pulse signal SP changes to "low". As a result, transistor M21 turns on, and the voltage VC15 of capacitor C15 is initialized.

[0173] Figure 16 This is a circuit diagram of a modified example of the conduction signal generation circuit 240. For example... Figure 7 As shown, when using comparator 252, high-precision voltage comparison can be performed; however, the circuit area will increase or the cost will increase. Therefore, as... Figure 16 As shown, a voltage comparator unit easily constructed with transistors can be used. The voltage comparator unit 253 includes: an emitter follower circuit 255 containing a PNP bipolar transistor Tr21; and a comparator circuit 257. The output (VLED+VBE) of the front-stage emitter follower circuit 255 is divided by resistors R21 and R22 and input to the base of transistor Tr22. When the monitored voltage VLED decreases, the base voltage of transistor Tr22 decreases. When it falls below the turn-on voltage of the bipolar transistor, the current to transistor Tr22 is cut off, and the output of the voltage comparator unit 253 becomes "high".

[0174] exist Figure 16 In this configuration, the outputs of multiple voltage comparator units 253 are input to OR gate 254, but this is not a limitation. Alternatively, OR gate 254 can be omitted, and the collectors of the transistors Tr22 of the multiple voltage comparator units 253 can be connected in a common ground, with a common resistor placed between the common collector and the power supply line VCC.

[0175] Figure 17 (a) to (c) are circuit diagrams showing examples of the configuration of current source 210. Figure 17 The current source 210 in (a) includes a series transistor M2, a sensing resistor RS, and an error amplifier 212. The series transistor M2 and the sensing resistor RS are connected in series in the path of the drive current ILEDi. The error amplifier 212 adjusts the voltage VG of the control electrode (in this example, the gate) of the series transistor M2 so that the voltage drop VCS of the sensing resistor RS approaches the target voltage VADIM. In this embodiment, the series transistor M2 is an N-type (N-channel) MOS transistor. The reference voltage VADIM is input to one input (non-inverting input terminal) of the error amplifier 212, and the voltage VCS (voltage drop of the sensing resistor RS) at the connection node of the series transistor M2 and the sensing resistor RS is input to the other input (inverting input terminal). By applying feedback using the error amplifier 212, the VCS is brought close to VADIM, and the drive current ILED is stabilized with a target value of ILED(REF) = VADIM / RS.

[0176] The current source 210 also includes a switch (dimming switch) 214 for PWM dimming. The dimming switch 214 is controlled by the PWM signal SPWM generated by the light distribution controller 116. When the dimming switch 214 is off, the drive current ILED flows through the current source 210. When the dimming switch 214 is on, the series transistor M2 is off, and the drive current ILED is cut off. By switching the dimming switch 214 at a high speed at a PWM frequency of 60Hz or higher, its duty cycle is adjusted, thereby performing PWM dimming on the semiconductor light source 102.

[0177] Figure 17 (b) The current source 210 uses a P-channel MOSFET as a series transistor. The polarity of the input to the error amplifier 212 is... Figure 17 (a) is the opposite.

[0178] In order to Figure 17 In the case of current source 210 in (a) or (b), the lower limit voltage VBOTTOM can be specified as follows. ΔV is an appropriate margin.

[0179] VBOTTOM = RS × ILED + VSAT + ΔV

[0180] Figure 17 (c) current source 210 and Figure 5 Similarly, it includes a current mirror circuit 216 and a reference current source 218. The current mirror circuit 216 amplifies the reference current IREF generated by the reference current source 218 by a predetermined factor determined by the reflection coefficient, thereby generating a drive current ILED. Figure 17 In the case of current source 210 in (c), the lower limit voltage VBOTTOM can be specified as follows.

[0181] VBOTTOM = VSAT + ΔV

[0182] VSAT is the saturation voltage of the current mirror circuit, and ΔV is an appropriate margin.

[0183] <Implementation Method 2>

[0184] Next, we will explain the integrated driver light source. Multiple current sources 210 can be integrated into a single semiconductor chip. Hereinafter, this will be referred to as a current driver IC (Integrated Circuit). Figure 18 This is a circuit diagram of the current driver IC 300 and its peripheral circuits according to Embodiment 2. In addition to multiple current sources 310_1 to 310_N, the current driver IC 300 also includes an interface circuit 320 and a dimming pulse generator 330.

[0185] Multiple current sources 310_1 to 310_N are configured to be independently turned on and off according to PWM signals SPWM1 to SPWMN. Current sources 310_1 to 310_N are connected in series with corresponding semiconductor light sources 102_1 to 102N through cathode pins LED1 to LEDN.

[0186] The interface circuit 320 receives multiple control data D1 to DN from an external microcomputer (processor) 114. The type of interface is not particularly limited; for example, an SPI (Serial Peripheral Interface) or I2C interface can be used. The multiple control data D1 to DN indicate the duty cycle of the multiple current sources 310_1 to 310_N and are updated at a first time interval T1. For example, the first time interval T1 is approximately 20ms to 200ms, such as 100ms.

[0187] The dimming pulse generator 330 generates multiple PWM signals SPWM1 to SPWMN corresponding to multiple current sources 310_1 to 310_N based on multiple control data D1 to DN. In embodiment 1 ( Figure 2 In implementation 2, multiple PWM signals SPWM1 to SPWMN are generated in the microcomputer 114, while in implementation 2... Figure 18 In the current driver IC300, the generation function of multiple PWM signals SPWM1 to SPWMN is built into the current driver IC300.

[0188] The duty cycle of the i-th PWM signal SPWMi gradually changes from the value before the update of the corresponding control data Di to the value after the update in a second time interval T2 that is shorter than the first time interval T1 (this is called the gradual change mode). The second time interval T2 is about 1ms to 10ms, for example, 5ms.

[0189] In addition to the gradual dimming mode, the dimming pulse generator 330 also supports a non-gradual dimming mode. In the non-gradual dimming mode, the duty cycle of the i-th PWM signal SPWMi can change instantaneously from the value before the update of the corresponding control data Di to the updated value.

[0190] The preferred non-gradient mode and the gradient mode can be dynamically changed based on settings from the microcomputer 114. Preferably, the non-gradient mode and the gradient mode can be specified individually by channel (by dimming pulse), and the setting data for the specified mode can also be attached to the control data Di.

[0191] Alternatively, part or all of the conduction signal generation circuit 240 can be integrated into the current driver IC 300. Which part is integrated depends on the circuit configuration of the conduction signal generation circuit 240, and can also be determined by reducing the number of wires between the converter controller 230 and the current driver IC 300. For example... Figure 18 As shown, when the entire conduction signal generation circuit 240 is integrated into the current driver IC 300, the wiring between the converter controller 230 and the current driver IC 300 becomes the wire that propagates the conduction signal SON. Alternatively, when using... Figure 10In the case of the conduction signal generation circuit 240G, when the minimum value circuit 256 is integrated into the current driver IC 300, the wiring between the converter controller 230 and the current driver IC 300 will become the one that propagates the minimum voltage VMIN.

[0192] Next, the operation of the current driver IC300 will be explained. Figure 19 This is the operating waveform diagram of the current driver IC300. Here, the duty cycle of the PWM signal changes linearly. For example, when T1 = 100ms and T2 = 5ms, the duty cycle can be changed in 20 steps. When the difference between the control data value before and after the update is X%, the duty cycle of the PWM signal changes by ΔY = (ΔX / 20)% every T2.

[0193] The above describes the operation of the current driver IC 300. The advantages of this current driver IC 300 will become clear through comparison with other technologies. When the duty cycle gradation function is not implemented in the current driver IC 300, the microcomputer 114 must update the control data D1 to DN indicating the duty cycle every second time interval T2. When the number of channels N in the semiconductor light source 102 exceeds tens to 100, a microcomputer 114 with high processing power is required, thus incurring a high cost. Furthermore, high-speed communication is required between the microcomputer 114 and the current driver IC 300, which can lead to noise problems.

[0194] In contrast, according to the current driver IC 300 of Embodiment 2, the processing power required by the microcomputer 114 can be reduced because the speed at which the microcomputer 114 updates the control data D1 to DN is reduced. Furthermore, since the communication speed between the microcomputer 114 and the current driver IC 300 can also be reduced, the noise problem can also be solved.

[0195] The first time interval T1 is preferably set to be variable. When the duty cycle changes little, the data communication volume can be reduced by making the first time interval T1 longer, thereby suppressing power consumption and noise.

[0196] exist Figure 19 In this method, the duty cycle can be changed linearly, but it can also be changed according to a curve such as a quadratic function or an exponential function. By using a quadratic function, it is possible to achieve a more natural dimming effect with less disharmony.

[0197] like Figure 18 As shown, multiple semiconductor light sources 102_1 to 102_N can also be integrated into a single semiconductor chip (die) 402. Furthermore, the semiconductor chip 402 and the current driver IC 300 can be housed in a single package, making it modular.

[0198] Figure 20 This is a plan view and a cross-sectional view of the integrated light source 400. Multiple semiconductor light sources 102 are formed in a matrix on the surface of the semiconductor chip 402. On the back side of the semiconductor chip 402, back electrodes A and K are provided, corresponding to the anode and cathode electrodes of each of the multiple semiconductor light sources 102. Here, only the connection relationship of one semiconductor light source 102_1 is shown.

[0199] Semiconductor chip 402 is mechanically bonded to current driver IC 300 and electrically connected. Surface electrodes 410 are provided on the surface of current driver IC 300. Figure 18 The current driver IC 300 includes LEDs 1 to N, and a surface electrode 412. The surface electrode 410 is connected to the cathode electrode K of each of the plurality of semiconductor light sources 102, and the surface electrode 412 is connected to the anode electrode A of each of the plurality of semiconductor light sources 102. The surface electrode 412 is connected to a bump (or pad) 414, which is disposed on the packaging substrate on the back side of the current driver IC 300. Alternatively, an insert (not shown) may be inserted between the semiconductor chip 402 and the current driver IC 300.

[0200] The type of package for the integrated driver light source 400 is not limited, and can be BGA (Ball Grid Array), PGA (Pin Grid Array), LGA (Land Grid Array), QFP (Quad Flat Package), etc.

[0201] When the semiconductor light source 102 and the current driver IC 300 are different modules, measures such as installing heat dissipation structures in each module can be implemented. On the other hand, in Figure 20 In such a driver-integrated light source 400, heat dissipation is required for the light source 102, which generates heat from the current source 210. Therefore, a very large heat dissipation structure may be required. By employing the lighting circuit 200 of Embodiment 2, the heat generation of the current source 210 can be suppressed, thus reducing the size of the heat dissipation structure that should be installed in the driver-integrated light source 400.

[0202] Next, variations associated with embodiments 1 and 2 will be described.

[0203] (Variation 1-1)

[0204] Alternatively, any transistor, such as the series transistor M2, can be used to construct a bipolar transistor. In this case, simply replace the gate with the base, the source with the emitter, and the drain with the collector.

[0205] (Variations 1-2)

[0206] In this implementation, the switching transistor M1 is configured as a P-channel MOSFET, but an N-channel MOSFET can also be used. In this case, a bootstrap circuit can also be added. Alternatively, instead of a MOSFET, an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor can be used.

[0207] (Variations 1-3)

[0208] In one embodiment, the current source 210 is configured as a current sinker and connected to the cathode of the semiconductor light source 102, but it is not limited thereto. Figure 21 This is a circuit diagram of the vehicle lamp 100 according to Variations 1-3. In this variation, the cathode of the semiconductor light source 102 is connected in a common ground, and a current source 210 of the current-carrying type is connected to the anode side of the semiconductor light source 102. The current source 210 can be configured to... Figure 5 or Figure 17 The current source is configured by inverting its high and low phases. The converter controller 230 controls the switching converter 220 based on the relationship between the voltage VCS across the current source 210 and the lower limit voltage VBOTTOM.

[0209] <Implementation Method 3>

[0210] Figure 24 This is a block diagram of a lighting system 1 including the vehicle lighting fixture 100 according to embodiment 3. The lighting system 1 includes a battery 2, a vehicle ECU (Electronic Control Unit) 4, and the vehicle lighting fixture 100. The vehicle lighting fixture 100 is a variable beam headlight with ADB function, forming a beam distribution corresponding to the control signal from the vehicle ECU 4.

[0211] The vehicle lighting fixture 100 includes multiple (N≥2) semiconductor light sources 102_1 to 102_N, a lighting ECU 110, a lighting circuit 200, and a current driver IC 300. LEDs are preferably used for the semiconductor light sources 102, but other light-emitting elements such as LDs or organic ELs can also be used. Alternatively, each semiconductor light source 102 may contain multiple light-emitting elements connected in series and / or in parallel. The number N of semiconductor light sources 102 is not particularly limited, but is at least tens, more preferably hundreds to thousands, or even on the order of tens of thousands. Multiple semiconductor light sources 102_1 to 102_N are integrated into a single semiconductor chip (LED chip 103).

[0212] The lighting ECU 110 includes a switch 112 and a microcomputer 114. The microcomputer (processor) 114 is connected to the vehicle ECU 4 via a bus such as CAN (Controller Area Network) or LIN (Local Interconnect Network) and can receive on / off indications or other information. In response to an on / off indication from the vehicle ECU 4, the microcomputer 114 turns on the switch 112. This allows the power supply voltage (battery voltage VBAT) from the battery 2 to be supplied to the lighting circuit 200.

[0213] In addition, the microcomputer 114 receives a control signal from the vehicle ECU 4 indicating the light distribution pattern and controls the lighting circuit 200. Alternatively, the microcomputer 114 may receive information from the vehicle ECU 4 indicating the situation in front of the vehicle and generate its own light distribution pattern based on that information.

[0214] The current driver IC 300 supplies drive currents ILED1 to ILEDN to multiple semiconductor light sources 102_1 to 102_N. The current driver IC 300 is configured to independently turn on and off the drive current ILED for each semiconductor light source 102.

[0215] The lighting circuit 200 supplies a drive voltage VOUT to the LED chip 103 and the current driver IC 300. The lighting circuit 200 may also include a switching converter 220 and a converter controller 230. In this example, a buck converter is shown, but the topology of the switching converter 220 is not particularly limited.

[0216] The current driver IC 300 includes multiple current sources 310_1 to 310_N, an interface circuit 320, a protection circuit 340, and a feedback circuit 360.

[0217] Current source 310_# (# = 1~N) is connected in series with one corresponding 102_# of multiple semiconductor light sources 102_1~102_N. Current source 310_# can be switched on and off according to control input Sp_#. In the on state, it generates drive current ILED#, and in the off state, it cuts off drive current ILED#. For example, when control input Sp_# is "high", current source 310_# is set to be on, and when control input Sp_# is "low", current source 310_# is set to be off.

[0218] Figure 25This is a circuit diagram illustrating an example configuration of multiple current sources 310. Each current source 310 includes a current mirror circuit 312, a reference current source 314, and a dimming switch 316. The reference current source 314 generates a reference current IREF. The current mirror circuit 312 replicates the reference current IREF to generate a drive current ILED. The dimming switch 316 is positioned between the gate and source of the current mirror circuit 312. When the dimming switch 316 is turned on, the operation of the current mirror circuit 312 stops, and the drive current ILED is cut off. The inverter 318 inverts the control input Sp and inputs it to the gate of the dimming switch 316. Alternatively, in a configuration with a negative logic system, the inverter 318 can be omitted, and the semiconductor light source 102 can be illuminated when the control input Sp is "low".

[0219] Next, return to Figure 24 The feedback circuit 360 outputs a feedback signal SFB corresponding to the voltages VLED1 to VLEDN between the terminals of the multiple current sources 310_1 to 310_N to the converter controller 230. For example, the feedback circuit 360 is designed according to the control method of the converter controller 230.

[0220] The converter controller 230 can also perform ripple control. In this case, the feedback circuit 360 may activate the feedback signal SFB when the lowest of the voltages VLED1 to VLEDN between the two terminals drops to a predetermined voltage (lower limit voltage VMIN). Alternatively, the converter controller 230 may activate the switching transistor M1 in response to the activation of the feedback signal SFB.

[0221] The converter controller 230 can also perform PWM (Pulse Width Modulation) control. Alternatively, the feedback circuit 360 can amplify the error between the lowest voltage among the voltages VLED1 to VLEDN between its terminals and a predetermined reference voltage VREF to generate an analog feedback signal SFB. Alternatively, the converter controller 230 can generate a PWM signal with a duty cycle corresponding to the voltage level of the feedback signal SFB and drive the switching transistor M1.

[0222] By integrating the feedback circuit 360 into the current driver IC 300, the feedback path can be consolidated into a single path, reducing the number of wires required.

[0223] Interface circuit 320 is connected to microcomputer (processor) 114 and receives control signals S1 indicating the on / off state of each of the plurality of semiconductor light sources 102_1 to 102_N. Interface circuit 320 has the function of a decoder that converts the received control signal S1 into multiple individual control signals S2_1 to S2_N. The multiple individual control signals S2_1 to S2_N define the on / off state of each of the multiple current sources 310_1 to 310_N.

[0224] Protection circuit 340 monitors the communication between microcomputer 114 and interface circuit 320. Under normal conditions, individual control signals S2_1 to S2_N based on control signal S1 are directly supplied to current sources 310_1 to 310_N. When protection circuit 340 detects an abnormality, it forcibly sets the multiple current sources 310_1 to 310_N to predetermined states. Specifically, it replaces the control inputs Sp_1 to Sp_N supplied to current sources 310_1 to 310_N with a pre-defined set of values, independent of control signal S1 (individual control signals S2_1 to S2_N).

[0225] For example, the predetermined state can also correspond to the light distribution of low beam. Figure 26 This is a plan view of LED chip 103. LED chip 103 is an array of multiple semiconductor light sources 102 configured in a matrix shape, with each semiconductor light source 102 corresponding to a pixel. The area RGNON that should be illuminated in the low beam distribution is surrounded by a thick line. When an anomaly is detected, the multiple semiconductor light sources 102 contained in this area RGNON are forcibly turned on. The multiple semiconductor light sources 102 contained in the area that should be turned off in the low beam distribution are forcibly turned off when an anomaly is detected.

[0226] Figure 27 This is a block diagram illustrating an example configuration of the interface circuit 320 and the protection circuit 340. For example, the interface circuit 320 may include a shift register 322 with an output latch. In this case, the interface circuit 320 is connected to the light distribution controller 116 via three lines: the SDATA line, the SCK line, and the RCK line.

[0227] The shift register 322 with output latch includes an N-bit shift register at the input stage and an N-bit storage register at the output stage. Serial data SDATA is fetched into the shift register at the input stage synchronously with the SCK signal. In response to the rising edge of the RCK signal, the value of the shift register is copied into the storage register, and the value is updated. The value of the storage register is retrieved as separate control signals S2_1 to S2_N.

[0228] The protection circuit 340 includes an anomaly detection circuit 342 and a data replacement circuit 344. The anomaly detection circuit 342 monitors at least one of the input signals of the interface circuit 320 (in this example, SDATA, SCK, and RCK) to detect communication anomalies. For example, the interface circuit 320 may determine an anomaly if the monitored signal does not change for a certain period of time. When an anomaly is determined, the anomaly detection signal S3 is activated.

[0229] When the anomaly detection signal S3 is invalid, the data replacement circuit 344 directly outputs individual control signals S2_1 to S2_N. When the anomaly detection signal S3 is valid, the data replacement circuit 344 replaces the corresponding control inputs Sp_1 to Sp_N of the multiple current sources 310_1 to 310_N with predetermined values. Specifically, for the semiconductor light source 102 that should be lit in case of an anomaly, Sp = H; for the semiconductor light source 102 that should be turned off in case of an anomaly, Sp = L.

[0230] Figure 28 (a) and (b) are circuit diagrams showing examples of the configuration of the data substitution circuit 344. Figure 28 The data replacement circuit 344 of (a) includes an inverter 346, a plurality of first logic gates 348, and a plurality of second logic gates 350.

[0231] Inverter 346 inverts the fault detection signal S3, generating an inverted fault detection signal S3b. Multiple first logic gates 348 correspond to multiple current sources 310 that should be turned on in the faulty state. Each first logic gate 348_i and one of the fault detection signal S3 and the inverted fault detection signal S3b (in this example, S3) receives a corresponding individual control signal S2_i and supplies its output to the corresponding current source 310_i. In this example, the first logic gate 348 is an OR gate.

[0232] Multiple second logic gates 350 correspond to multiple current sources 310 that should be turned off under abnormal conditions. Each second logic gate 350_j and another of the abnormal detection signal S3 and the inverted abnormal detection signal S3b (in this example, S3b) receives a corresponding individual control signal S2_j and supplies its output to the corresponding current source 310_j. In this example, the second logic gate 350 is an AND gate.

[0233] exist Figure 28In (b), another configuration example of the data replacement circuit 344 is shown. The data replacement circuit 344 includes multiple selectors 352_1 to 352_N corresponding to multiple current sources 310_1 to 310_N. A corresponding individual control signal S2_# and a predetermined value a# are input to each selector 352_#. For current source 310_i that should be turned on in an abnormal state, the predetermined value a# is 1, and for current source 310_j that should be turned off in an abnormal state, the predetermined value a# is 0. When the abnormal detection signal S3 is invalid (0), selector 352_# selects the individual control signal S2_#, and when the abnormal detection signal S3 is set to valid (1), selects the predetermined value a#. Figure 28 In (b), a memory is needed to hold predetermined values ​​a1 to a#, but it has the advantage of being able to change the light distribution pattern under abnormal conditions based on the values ​​in the memory. In contrast, Figure 28 In (a), the light distribution pattern in the abnormal state cannot be changed, but memory is necessary, and it has the advantage of reducing the circuit area.

[0234] Figure 29 This is a plan view and a cross-sectional view of the integrated light source 400. In the semiconductor chip 402 ( Figure 24 On the surface of the LED chip 402, multiple semiconductor light sources 102 are formed in a matrix. On the back side of the semiconductor chip 402, back electrodes A and K are provided, corresponding to the anode and cathode electrodes of each of the multiple semiconductor light sources 102. Here, only the connection relationship of one semiconductor light source 102_1 is shown.

[0235] Semiconductor chip 402 is mechanically bonded to and electrically connected to current driver IC 300. Surface electrodes 410 and 412 are provided on the surface of current driver IC 300. Surface electrode 410 is connected to the cathode electrode K of each of the plurality of semiconductor light sources 102, and surface electrode 412 is connected to the anode electrode A of each of the plurality of semiconductor light sources 102. Surface electrode 412 is connected to bumps (or pads) 414, which are disposed on the packaging substrate on the back side of current driver IC 300. Alternatively, an insert (not shown) may be inserted between semiconductor chip 402 and current driver IC 300.

[0236] The type of package for the integrated driver light source 400 is not limited, and can be BGA (Ball Grid Array), PGA (Pin Grid Array), LGA (Land Grid Array), QFP (Quad Flat Package), etc.

[0237] Finally, a variation example will be explained.

[0238] (Variation 3.1)

[0239] In embodiment 3, the current source 310 is configured as a current sinker and connected to the cathode of the semiconductor light source 102, but it is not limited to this. Figure 30 This is a circuit diagram of the vehicle lamp 100 according to Variation 3.1. In this variation, the cathodes of the semiconductor light source 102 are connected in a common circuit, and a current source 310 of the current-carrying type is connected to the anode side of the semiconductor light source 102. In this configuration, the difference between VOUT and VLED# is the voltage between the two ends of the current source 310_#. Therefore, it is also possible that the feedback circuit 360 of the ripple control mode activates the feedback signal SFB when the highest voltage among the anode voltages VLED1 to VLEDN reaches the threshold voltage VTH (=VOUT-VMIN). VMIN is the minimum value of the voltage between the two ends of the current source 310 that can operate. Alternatively, the converter controller 230 responds to the activation of the feedback signal SFB by turning on the switching transistor M1.

[0240] (Variation 3.2)

[0241] In embodiment 3, the switching transistor M1 is configured as a P-channel MOSFET, but an N-channel MOSFET can also be used. In this case, a bootstrap circuit can also be added. Alternatively, an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor can be used instead of a MOSFET.

[0242] (Variation 3.3)

[0243] In implementation method 3, the logic circuit is constructed with a positive logic system, but it is not limited to this. Alternatively, a portion or all of it may be constructed as a negative logic system.

[0244] (Variation 3.4)

[0245] The configuration of the current source 310 is not limited to using a current mirror circuit; various known configurations can be used.

[0246] <Implementation Method 4>

[0247] Figure 31 This is a block diagram of the vehicle lamp 500 according to Embodiment 4. The lighting circuit 570 illuminates the light source 560. The variable light distribution device 530 includes multiple independently controllable elements and forms a light distribution pattern based on the states of the multiple controllable elements. For example, the variable light distribution device 530 is a DMD (Digital Mirror Device) that reflects the emitted light from the light source 560 or multiple bypass switches.

[0248] The lighting ECU510 is a host controller that controls the light distribution pattern based on control signals or various information received from the vehicle ECU4 via the communication line 5.

[0249] A local controller 540 and a photodistribution variable device 530 are mounted on a substrate 504. An interface circuit 520 is connected to a luminaire ECU 510 via a communication line 502 and receives a control signal S1 indicating a photodistribution pattern. The local controller 540 controls the photodistribution variable device 530 based on the control signal S1 received by the interface circuit 520. Alternatively, the local controller 540 may have a decoder function that converts the control signal S1 from the luminaire ECU 510 into a control signal S4 suitable for the photodistribution variable device 530. For example, the control signal S4 may include multiple individual control signals that indicate the state of each of the multiple controllable elements forming the photodistribution variable device 530.

[0250] The anomaly detection unit 550 monitors the communication between the lamp ECU 510 and the interface circuit 520, and detects anomalies based on the monitoring results. The vehicle lamp 500 is configured such that when an anomaly is detected by the anomaly detection unit 550, a predetermined pattern is forcibly set on the light distribution variable device 530.

[0251] When an anomaly is detected, the anomaly detection unit 550 activates the anomaly detection signal S5. For example, the anomaly detection unit 550 may determine an anomaly when the control signal S1 of the communication line 502 does not change within a certain period of time. In response to the activation of the anomaly detection signal S5, the local controller 540 replaces multiple individual control signals S2 with a predetermined combination of values.

[0252] The interface circuit 520, the local controller 540, and the anomaly detection unit 550 can be installed in a single microcomputer or as a combination of multiple hardware components.

[0253] Implementation method 3 can be understood as a solution to implementation method 4, and they can correspond in the following way. Specifically, Figure 24 The interface circuit 320 is connected to Figure 31 The interface circuit 520 and the local controller 540 correspond to this. Furthermore, Figure 24 Multiple current sources 310_1 to 310_N (or multiple dimming switches 316 contained therein) are connected to Figure 31 The 530 is compatible with the variable light distribution device. Furthermore... Figure 27 The anomaly detection circuit 342 is connected with Figure 31 The anomaly detection unit 550 corresponds to this. Figure 27 The data substitution circuit 344 can be understood as Figure 31 It is part of the local controller 540.

[0254] (Variation 4.1)

[0255] Figure 32 This is a block diagram of vehicle lighting fixture 500A, as shown in Variation 4.1. Vehicle lighting fixture 500A, besides... Figure 31 In addition to the vehicle lighting fixture 500, the system also includes a selector 552 and a fixed light distribution indicator 554. The fixed light distribution indicator 554 generates a control signal S6 that specifies the light distribution pattern to be used in case of an anomaly. When the anomaly detection signal S5 is invalid, the selector 552 selects the control signal S4 generated by the local controller 540; when the anomaly detection signal S5 is valid, it selects the control signal S6 generated by the fixed light distribution indicator 554.

[0256] (Variation 4.2)

[0257] Figure 33 This is a block diagram of vehicle lighting fixture 500B, a variation of Example 4.2. Vehicle lighting fixture 500B, besides... Figure 32 In addition to the vehicle lighting fixture 500, a second anomaly detection unit 556 is also included. The second anomaly detection unit 556 monitors the output S4 of the local controller 540, and when an anomaly is detected, it activates the anomaly detection signal S7.

[0258] The local controller 540 responds to the activation of the abnormality detection signal S5 by replacing the control signal S4 with the signal that should be used in the abnormal state.

[0259] On the other hand, if the local controller 540 does not operate normally when the abnormality detection signal S7 is active, or if the communication line between the local controller 540 and the photodistribution variable device 530 malfunctions, then the control signal S4, which should be used in the abnormal state, cannot be correctly provided to the photodistribution variable device 530. Therefore, when the abnormality detection signal S7 is active, the appropriate control signal can be provided to the photodistribution variable device 530 by having the selector 552 select the control signal S6 generated by the fixed photodistribution indicator 554.

[0260] Although specific statements have been used to describe the present invention based on the implementation methods, the implementation methods only represent the principles and applications of the present invention. Many variations or configuration changes are allowed for the implementation methods without departing from the spirit of the present invention as defined in the claims.

[0261] [Industrial Availability]

[0262] This invention relates to lighting circuits.

[0263] [Explanation of reference numerals in the attached figures]

[0264] 1…Lighting system, 2…Battery, 4…Vehicle ECU, 100…Vehicle lighting, 102…Semiconductor light source, 103…LED chip, 110…Lighting ECU, 112…Switch, 114…Microcomputer, 116…Light distribution controller, 200…Lighting circuit, 210…Current source, M2…Series transistor, RS…Sensing resistor, 212…Error amplifier, 214…Dimmer switch, 216…Current mirror circuit, 218…Reference current source, 220…Switch converter, M1… Switching transistor, 230…converter controller, 232…driver, 234…logic circuit, 240…on signal generation circuit, 252…comparator, 254…logic gate, 256…minimum value circuit, 258…comparator, 260…off signal generation circuit, 262…comparator, 264…frequency detection circuit, 266…error amplifier, 268…timer circuit, 270…variable timer circuit, 272…frequency detection circuit, 274…error amplifier, 280… Voltage generating circuit, 282…constant voltage source, 284…corrected current source, 286…current mirror circuit, 288…current source, 290…emitter follower circuit, 300…current driver IC, 310…current source, 312…current mirror circuit, 314…reference current source, 316…dimmer switch, 320…interface circuit, 322…output latch with shift register, 330…dimmer pulse generator, 340…protection circuit, 342…abnormal detection circuit, 344…data replacement Circuit, 346…Inverter, 348…First logic gate, 350…Second logic gate, 360…Feedback circuit, 400…Drive integrated light source, 402…Semiconductor chip, 500…Vehicle lamp, 510…Lamp ECU, 520…Interface circuit, 530…Variable light distribution device, 540…Local controller, 550…Abnormal detection unit, 560…Light source, 570…Lighting circuit, 552…Selector, 554…Fixed light distribution indicator, 556…Second abnormal detection unit.

Claims

1. A drive circuit for supplying drive current to a plurality of semiconductor light sources used as vehicle lamps, characterized in that, include: Multiple current sources are connected in series with corresponding semiconductor light sources, and can be controlled to turn on and off based on binary control inputs. An interface circuit that can communicate with the processor of the lighting ECU, receives control signals from the processor instructing the activation and deactivation of the plurality of semiconductor light sources, generates a plurality of individual binary control signals based on the control signals, and outputs the plurality of individual binary control signals through multiple wirings. The protection circuit has multiple input terminals that receive multiple binary individual control signals through the multiple wiring and multiple output terminals connected to the multiple current sources. It monitors the communication between the processor and the interface circuit. When the communication is normal, it outputs the multiple individual control signals directly as multiple control inputs from the multiple output terminals. When an abnormality is detected in the communication, it outputs a combination of predetermined values ​​as multiple control inputs from the multiple output terminals.

2. The driving circuit as described in claim 1, characterized in that, The combination of the aforementioned predetermined values ​​is set to form a light distribution state corresponding to the light distribution of the near beam.

3. The driving circuit as described in claim 1, characterized in that, The above protection circuit includes: The anomaly detection circuit, upon detecting the aforementioned anomaly, activates the anomaly detection signal and... The data replacement circuit directly outputs the multiple individual control signals from the multiple output terminals when the above-mentioned abnormality detection signal is invalid, and outputs a combination of predetermined values ​​from the multiple output terminals when the above-mentioned abnormality detection signal is valid.

4. The driving circuit as described in claim 3, characterized in that, The above data replacement circuit includes: The inverter inverts the aforementioned anomaly detection signal, generating an inverted anomaly detection signal. Multiple first logic gates, corresponding to multiple current sources among the aforementioned multiple current sources that should be turned on when the aforementioned anomaly is detected, and Multiple second logic gates, which correspond to multiple current sources among the aforementioned multiple current sources that should be turned off when the aforementioned anomaly is detected; Each first logic gate receives a corresponding individual control signal from one of the above-mentioned abnormal detection signal and the above-mentioned inverted abnormal detection signal, and outputs its signal to the corresponding current source. Each second logic gate receives a corresponding individual control signal from the above-mentioned abnormality detection signal and the above-mentioned inverted abnormality detection signal, and supplies its output to the corresponding current source.

5. The driving circuit according to any one of claims 1 to 4, characterized in that, If the above protection circuit does not perform a level transition within a certain period of the control signal, it is determined to be abnormal.

6. A driving integrated light source, characterized in that, include: Multiple semiconductor light sources are integrated into a first semiconductor chip, and The driving circuit as described in any one of claims 1 to 5 is coupled to the second semiconductor chip.

7. A vehicle lamp, characterized in that, Includes the drive circuit as described in any one of claims 1 to 5.

8. A vehicle lamp, characterized in that, It is constituted to include: Upper controller, The interface circuit receives control signals from the aforementioned host controller. Variable light distribution equipment The local controller controls the variable light distribution device based on the control signals received from the interface circuit, and outputs multiple binary control signals via multiple wirings to indicate the on or off state of multiple current sources in the variable light distribution device. The first anomaly detection unit monitors the communication between the host controller and the interface circuit and detects anomalies. When the above-mentioned abnormality is detected, a predetermined light distribution pattern is formed by replacing the above-mentioned multiple control signals with a predetermined combination of values.

9. The vehicle lighting fixture as described in claim 8, characterized in that, It also includes a second anomaly detection unit, which monitors the output of the aforementioned local controller and detects anomalies.

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