A High-Quality Resonant Deadbeat Predictive Current Control Method for SynRM

Abstract

The present invention discloses a high-quality resonant deadbeat predictive current control method for SynRM, comprising the following steps: Step 1, establish a super-local model of a synchronous reluctance motor in a two-phase rotating coordinate system; Step 2, design a deadbeat predictive current controller for the synchronous reluctance motor according to the super-local model obtained in Step 1. The present invention can compensate for the disturbances in the predictive current control system by using a sliding-mode lumped disturbance observer under the super-local model. The reaching law adopted by the sliding-mode lumped disturbance observer can effectively reduce the high-frequency chattering caused by the sliding-mode switching function and can also effectively improve its convergence speed. In order to suppress the 6th harmonic in the torque, a quasi-resonant disturbance estimator is implanted in the disturbance estimation loop of the sliding-mode lumped disturbance observer, which can effectively compensate for the AC disturbance of the 6th harmonic. While suppressing external disturbances, it can also effectively improve the ability to suppress current harmonic disturbances, thereby efficiently achieving the effect of suppressing torque ripple.

Description

Technical Field

[0001] The present invention relates to the technical field of synchronous reluctance motor control, and particularly to a SynRM high-quality resonant deadbeat predictive current control method. Background Technique

[0002] The rotor of a synchronous reluctance motor (SynRM) has no permanent magnet and field winding, so it does not rely on rare earth materials and has advantages such as high efficiency, strong reliability, and low cost. As the synchronous reluctance motor is more and more widely used in the industrial field, its dynamic and steady-state performance becomes very important. Therefore, deadbeat predictive current control with high dynamic performance has become a research hotspot at present. However, deadbeat predictive current control is very dependent on the accuracy of motor parameters. In fact, the motor parameters of a synchronous reluctance motor change with the operating point and environment. For example, the stator resistance changes with temperature, and the stator inductance changes with the stator current;

[0003] In order to solve the dependence of deadbeat predictive current control on motor parameters, an extended state observer is usually used to estimate the disturbances caused by motor parameter changes, etc. However, the extended state observer cannot accurately estimate fast-changing disturbances. When the synchronous reluctance motor runs at high speed, the disturbances caused by inductance parameter changes are fast-changing disturbances, and due to the asymmetry of the magnetic circuit of the synchronous reluctance motor, the synchronous reluctance motor will generate large torque ripples and harmonic currents, and the torque ripples and harmonic currents are also fast-changing disturbances. These fast-changing disturbances cannot be accurately estimated, resulting in the feedback current of deadbeat predictive current control being unable to accurately track the given current, the current deviation becoming larger, and even causing the controller to be unstable;

[0004] Therefore, it is necessary to provide a SynRM high-quality resonant deadbeat predictive current control method to solve the above technical problems. Summary of the Invention

[0005] The purpose of the present invention is to provide a SynRM high-quality resonant deadbeat predictive current control method to solve the problems raised in the above background technique. The technical solution of the present invention provides a solution significantly different from the prior art for the technical problem that the prior art solution is too single.

[0006] To achieve the above object, the present invention provides the following technical solution: A SynRM high-quality resonant deadbeat predictive current control method, comprising the following steps:

[0007] Step 1: Establish a super-local model of a synchronous reluctance motor in a two-phase rotating coordinate system;

[0008] Step 2: Design a deadbeat predictive current controller for the synchronous reluctance motor from the super-local model obtained in Step 1;

[0009] Step 3: Design a sliding mode lumped disturbance observer to obtain the system disturbance estimation value in Step 2;

[0010] Step 4: Design a resonant sliding mode lumped disturbance observer based on Step 3 to suppress current harmonics and torque harmonics;

[0011] Step 5: Calculate the command values of the direct and quadrature axis voltage controls.

[0012] In a further embodiment, Step 1 is specifically as follows:

[0013] The voltage equation of the SynRM in the rotating coordinate axes of the d-q axes is shown as the following formula (1):

[0014]

[0015] Wherein, u d , u q are the d-axis and q-axis components of the stator voltage, R s is the stator resistance, i d , i q are the d-axis and q-axis components of the stator current, L d , L q are the d-axis and q-axis stator inductances respectively, ω e is the electrical angular velocity of the rotor, ψ d , ψ q are the d-axis and q-axis components of the stator flux linkage;

[0016] According to the input and output of the synchronous reluctance motor system, the nonlinear, complex and variable system can be replaced by a superlocal model. The first-order superlocal model of a single-input single-output system is shown as the following formula (2):

[0017]

[0018] Wherein, u and y are the control and output variables respectively; α represents a non-physical scaling factor selected by the designer; F represents the known part and unmodeled dynamics of the system;

[0019] According to formula (1) and formula (2), the superlocal models of the d-axis and q-axis current loops are shown as the following formula (3):

[0020]

[0021] Wherein, p is a differential operator, ω m is the mechanical angular velocity, α d is the d-axis designed voltage parameter, α q is the q-axis designed voltage parameter, F d is the d-axis designed system disturbance, F qDesign system disturbances for the q-axis, and

[0022] In a further embodiment, step 2 is specifically as follows:

[0023] Obtain the voltage u according to formula (3) d 、u q as the following formula (4):

[0024] [[ID=l5]]

[0025] Discretize formula (4) according to the forward Euler formula as shown in the following formula (5):

[0026]

[0027] where u d (k) is the d-axis voltage at time k, u q (k) is the q-axis voltage at time k, i d (k + 1) is the d-axis current at time k + 1, i q (k + 1) is the q-axis current at time k + 1, i d (k) is the d-axis current at time k, i q (k) is the q-axis current at time k, F d (k) is the d-axis disturbance at time k, F q (k) is the q-axis disturbance at time k, T s is the sampling time;

[0028] When calculating the desired voltage, use the current reference value as the current value at time k + 1 as shown in the following formula (6):

[0029]

[0030] where is the d-axis reference current at time k + 1, is the q-axis reference current at time k + 1;

[0031] Combine formula (5) and formula (6) to obtain the predictive current controller as the following formula (7):

[0032]

[0033] where are the AC and DC axis voltage control commands obtained through predictive calculation in the current cycle respectively. This voltage command will act on the motor through the inverter in the next cycle through SVPWM, so that the stator current of the motor at the next moment tracks the given value without error, represents i d [[ID=HO]](k), i q(k) estimated value, Denote F d (k), F q (k) estimated value;

[0034] In an actual motor drive system adopting a digital controller, the current and voltage sampling of the system, algorithm real-time calculation, and PWM duty cycle update cannot occur at the same time, which makes the deadbeat predictive current control have a delay control problem during implementation. Therefore, when designing the controller, the stator current and disturbance at the k+1 moment are predicted one beat forward, and we get The following formula (8):

[0035]

[0036] Wherein, Denote i d (k+1), i q (k+1) estimated value, Denote F d (k+1), F q (k+1) estimated value;

[0037] In a further embodiment, step 3 is specifically as follows:

[0038] In order to improve the accuracy, rapidity of parameter F d and F q estimation and the anti-interference ability of parameter estimation, the design of the sliding mode lumped disturbance observer based on the super-local model is shown in the following formulas (9) and (10):

[0039]

[0040]

[0041] Wherein, Are respectively the derivatives of the estimated values of the d-axis and q-axis currents, Are respectively the estimated values of the super-local model parameters of the d-axis and q-axis, Are respectively the derivatives of the estimated values of the super-local model parameters of the d-axis and q-axis, ξ is the gain coefficient of the control law, σ d , σ q Are respectively the d-axis and q-axis sliding mode reaching laws;

[0042] The selection of the sliding mode surface is shown in the following formula (11):

[0043]

[0044] Design a reaching law function to effectively improve the convergence speed of the sliding mode observer and reduce chattering, and its expression is shown in the following formula (12):

[0045]

[0046]

[0047] Among them, λ, η, and α are parameters in f(s), obtained from formula (13). When the system state is far from the sliding surface (|s| increases), f(s) will converge to β << 1 + λ is a very small value; when the system state is close to the sliding surface (|s| decreases), f(s) will converge to |s| α <1, -qs causes the system chattering to decay exponentially, thus ensuring the stability of the system. This adaptive reaching law can both reduce the system chattering and accelerate the convergence speed;

[0048] The d - axis and q - axis sliding - mode reaching laws are obtained from formula (12) as the following formula (14):

[0049]

[0050] According to formulas (9), (10), and (14), the estimated values of the d - axis and q - axis currents are as the following formulas (15) and (16):

[0051]

[0052]

[0053] After discretizing formulas (15) and (16) through the forward Euler formula, they are as shown in the following formulas (17) and (18):

[0054]

[0055]

[0056] In a further embodiment, step 4 is specifically:

[0057] The 5th - order and 7th - order phase - current harmonics in the abc three - phase stationary coordinate system are converted to the 6th - order harmonics in the d - q axis. The currents in the d - q axis are as shown in the following formula (19):

[0058]

[0059] Among them, I1, I5, and I7 are the amplitudes of the fundamental, 5th - order, and 7th - order current harmonics respectively; θ1, θ5, and θ7 are the phases of the fundamental, 5th - order, and 7th - order current harmonics respectively;

[0060] The transfer function of the designed quasi - resonant controller is as shown in the following formula (20):

[0061]

[0062] Among them, ω c is the cut-off frequency of the quasi-resonant controller, ω0 is the resonant frequency, and ω0 = 6ω is taken according to formula (19) e , K i is the integral gain;

[0063] The bilinear transformation is used to discretize formula (20) as shown in the following formula (21):

[0064]

[0065] Among them, b1 = 0,

[0066]

[0067] After arranging formula (21), the following formula (22) is obtained:

[0068] Y(z)+a1z -1 Y(z)+a2Y(z) = b0U(z)+b1z -1 U(z)+b2z -2 U(z) (22)

[0069] The conversion relationship is as shown in the following formula (23):

[0070]

[0071] According to formulas (22) and (23), the output difference equation is as shown in the following formula (24):

[0072] y(k) = b0u(k)+b2u(k - 2)-a1y(k - 1)-a2y(k - 2) (24)

[0073] The input of the difference equation in the system is e d and e q , so formula (24) is rewritten as:

[0074]

[0075] Based on the basic sliding mode lumped disturbance observer, a quasi-resonant disturbance estimator is added to the loop of observing the disturbance to compensate for the AC disturbance as shown in the following formula (26):

[0076]

[0077] In the formula, y d 、y qis the continuous output of the d-axis and q-axis resonant disturbance estimators;

[0078] The predicted current of the final output obtained from equations (17), (18) and (26) is as shown in equations (27) and (28) below:

[0079]

[0080]

[0081] where y d (k), y q (k) are the discrete outputs of the d-axis and q-axis resonant disturbance estimators.

[0082] In a further embodiment, step 5 is specifically:

[0083] Based on steps 2, 3 and 4, the calculation of the AC and DC axis voltage control command values can be re-obtained from equation (8), as shown in equation (29) below:

[0084]

[0085] where, is the d-axis predicted current at time k + 1 after adding the quasi-resonant disturbance estimator, is the q-axis predicted current at time k + 1 after adding the quasi-resonant disturbance estimator, is the estimated value of the d-axis hyper-local model parameter at time k + 1 after adding the quasi-resonant disturbance estimator, is the estimated value of the q-axis hyper-local model parameter at time k + 1 after adding the quasi-resonant disturbance estimator.

[0086] Compared with the prior art, the beneficial effects of the present invention are:

[0087] 1. In the present invention, the designed resonant controller can achieve current harmonic suppression under periodic disturbances by utilizing the infinite gain characteristic at the resonant frequency. Compared with the existing resonant controllers, the quasi-resonant controller increases the bandwidth at the resonant frequency, reducing the frequency sensitivity of the system and improving the stability. It has a better suppression effect on the 5th and 7th phase current harmonics. Therefore, it is more suitable for motor systems with torque fluctuations. Suppressing the phase current can make the resonant deadbeat predictive current control more accurate, with better stability and strong robustness. Part of the reason for torque fluctuations is caused by the harmonics of the phase current. Therefore, suppressing the phase current harmonics can reduce torque fluctuations and is more suitable for motor systems with torque fluctuations;

[0088] 2. The present invention uses a sliding mode lumped disturbance observer under an ultra-local model to observe the current in real time, which can compensate for the disturbances in the predictive current control system. The reaching law adopted by the sliding mode lumped disturbance observer can effectively reduce the high-frequency chattering caused by the sliding mode switching function and can also effectively improve its convergence speed. In addition, in order to suppress the 6th harmonic in the torque, a quasi-resonant disturbance estimator is implanted in the disturbance estimation loop of the sliding mode lumped disturbance observer, which can effectively compensate for the AC disturbance of the 6th harmonic. Compared with the existing technologies, while effectively suppressing external disturbances, it can also effectively improve the ability to suppress current harmonic disturbances, thus efficiently achieving the effect of suppressing torque ripple. Description of the Drawings

[0089] Figure 1 is the system block diagram of a SynRM high-quality resonant deadbeat predictive current control method of the present invention;

[0090] Figure 2 is the structural block diagram of the resonant sliding mode lumped disturbance observer adopted by a SynRM high-quality resonant deadbeat predictive current control method of the present invention (taking the d-axis as an example);

[0091] Figure 3 is the simulation result before and after torque ripple suppression of a SynRM high-quality resonant deadbeat predictive current control method of the present invention, Figure 3 (a) is the simulation waveform without using the torque ripple suppression method proposed by the present invention; Figure 3 (b) is the simulation waveform using the torque ripple suppression method proposed by the present invention. Detailed Embodiment

[0092] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0093] Please refer to Figure 1-2 , an embodiment provided by the present invention: a SynRM high-quality resonant deadbeat predictive current control method, including the following steps:

[0094] Step 1: Establish an ultra-local model of a synchronous reluctance motor in a two-phase rotating coordinate system;

[0095] Step 2: Design a deadbeat predictive current controller for the synchronous reluctance motor from the ultra-local model obtained in Step 1;

[0096] Step 3: Design a sliding mode lumped disturbance observer to obtain the system disturbance estimation value in Step 2;

[0097] Step 4: Design a resonant sliding mode lumped disturbance observer based on Step 3 to suppress current harmonics and torque harmonics;

[0098] Step 5: Calculate the control command values of the direct and quadrature axis voltages based on Steps 2, 3, and 4.

[0099] Specifically, Step 1 is as follows:

[0100] The voltage equation of SynRM in the rotating coordinate axes of the d-q axes is shown in the following formula (1):

[0101]

[0102] where, u d , u q are the d-axis and q-axis components of the stator voltage, R s is the stator resistance, i d , i q are the d-axis and q-axis components of the stator current, L d , L q are the d-axis and q-axis stator inductances respectively, ω e is the electrical angular velocity of the rotor, ψ d , ψ q are the d-axis and q-axis components of the stator flux linkage;

[0103] According to the input and output of the synchronous reluctance motor system, some nonlinear, complex, and variable systems can be replaced by a superlocal model. The first-order superlocal model of a single-input single-output system is shown in the following formula (2):

[0104]

[0105] where, u and y are the control and output variables respectively; α represents the non-physical scaling factor selected by the designer; F represents the known part and the unmodeled dynamics of the system.

[0106] Based on formula (1) and formula (2), the superlocal models of the d-axis and q-axis current loops are shown in the following formula (3):

[0107]

[0108] where, p is the differential operator, ω m is the mechanical angular velocity, α d is the d-axis designed voltage parameter, α q is the q-axis designed voltage parameter, F d is the d-axis designed system disturbance, F q is the q-axis designed system disturbance, and

[0109] Step 2 specifically is as follows:

[0110] Obtain the voltage u according to formula (3) d and u q as the following formula (4):

[0111]

[0112] Discretize formula (4) according to the forward Euler formula as shown in the following formula (5):

[0113]

[0114] where u d (k) is the d-axis voltage at time k, u q (k) is the q-axis voltage at time k, i d (k + 1) is the d-axis current at time k + 1, i q (k + 1) is the q-axis current at time k + 1, i d (k) is the d-axis current at time k, i q (k) is the q-axis current at time k, F d (k) is the d-axis disturbance at time k, F q (k) is the q-axis disturbance at time k, T s is the sampling time;

[0115] When calculating the desired voltage, the current reference value needs to be used as the current value at time k + 1 as shown in the following formula (6):

[0116]

[0117] where is the d-axis reference current at time k + 1, is the q-axis reference current at time k + 1.

[0118] Combining formula (5) and formula (6) gives the predictive current controller as the following formula (7):

[0119]

[0120] where are the AC and DC axis voltage control commands obtained through predictive calculation in the current cycle respectively. This voltage command will act on the motor through the inverter in the next cycle, enabling the stator current of the motor to track the given value without deadbeat at the next moment, represents the estimated values of i d (k) and i q (k), represents F d (k) and F q(k) estimated value;

[0121] In an actual motor drive system using a digital controller, the current and voltage sampling of the system, the real-time calculation of the algorithm, and the update of the PWM duty cycle cannot occur at the same time. This makes the deadbeat predictive current control have a delay control problem during implementation. Therefore, when designing the controller, the stator current and disturbance at the k+1 moment are predicted one beat forward, and we get The following formula (8):

[0122]

[0123] Among them, represents the estimated values of i d (k + 1) and i q (k + 1), represents the estimated values of F d (k + 1) and F q (k + 1).

[0124] Step 3 is specifically as follows:

[0125] In order to improve the accuracy, rapidity of the parameter F d and F q estimation and the anti-interference ability of the parameter estimation, the design of the sliding mode lumped disturbance observer based on the super-local model is shown in the following formulas (9) and (10):

[0126]

[0127] [[ID=...]]

[0128] Among them, are respectively the derivatives of the estimated values of the d-axis and q-axis currents, are respectively the estimated values of the d-axis and q-axis super-local model parameters, are respectively the derivatives of the estimated values of the d-axis and q-axis super-local model parameters, ξ is the gain coefficient of the control law, σ d and σ q are respectively the d-axis and q-axis sliding mode reaching laws;

[0129] The selection of the sliding mode surface is shown in the following formula (11):

[0130]

[0131] Design a reaching law function to effectively improve the convergence speed of the sliding mode observer and reduce chattering. Its expression is shown in the following formula (12):

[0132]

[0133] Note: Some of the tags like etc. seem to be incomplete in the original. I've translated as much as possible while keeping the tags intact. If there are any specific instructions or corrections regarding those incomplete tags, please let me know.

[0134] Among them, λ, η, and α are parameters in f(s), obtained from formula (13). When the system state is far from the sliding surface (|s| increases), f(s) will converge to β << 1 + λ is a very small value; when the system state is close to the sliding surface (|s| decreases), f(s) will converge to |s| α <1, -qs causes the chattering of the system to decay exponentially, thus ensuring the stability of the system. This adaptive reaching law can both reduce the chattering of the system and accelerate the convergence speed;

[0135] The d - axis and q - axis sliding - mode reaching laws are obtained from formula (12) as follows in formula (14):

[0136]

[0137] According to formulas (9), (10), and (14), the estimated values of the d - axis and q - axis currents are as follows in formulas (15) and (16):

[0138]

[0139]

[0140] By using the forward - Euler formula, formulas (15) and (16) are discretized as shown in formulas (17) and (18) below:

[0141]

[0142]

[0143] Step 4 is specifically as follows:

[0144] The 5th - order and 7th - order phase - current harmonics in the abc three - phase stationary coordinate system are converted to the 6th - order harmonics in the d - q axis. The currents in the d - q axis are as shown in formula (19) below:

[0145]

[0146] Among them, I1, I5, and I7 are the amplitudes of the fundamental - wave, 5th - order, and 7th - order harmonics of the current respectively; θ1, θ5, and θ7 are the phases of the fundamental - wave, 5th - order, and 7th - order harmonics respectively;

[0147] The transfer function of the designed quasi - resonant controller is as shown in formula (20) below:

[0148]

[0149] Among them, ω cis the cut-off frequency of the quasi-resonant controller, ω0 is the resonant frequency, and ω0 = 6ω is taken according to formula (19). e , K i is the integral gain;

[0150] The bilinear transformation is used to discretize formula (20) as shown in the following formula (21):

[0151]

[0152] Among them,

[0153]

[0154] After arranging formula (21), the following formula (22) is obtained:

[0155] Y(z)+a1z -1 Y(z)+a2Y(z) = b0U(z)+b1z -1 U(z)+b2z -2 U(z) (22)

[0156] The conversion relationship is shown in the following formula (23):

[0157]

[0158] According to formulas (22) and (23), the output difference equation is shown in the following formula (24):

[0159] y(k) = b0u(k)+b2u(k - 2)-a1y(k - 1)-a2y(k - 2) (24)

[0160] The input of the difference equation in the system is e d and e q , so formula (24) is rewritten as:

[0161]

[0162] Based on the basic sliding mode lumped disturbance observer, a quasi-resonant disturbance estimator is added to the loop for observing the disturbance to compensate for the AC disturbance as shown in the following formula (26):

[0163]

[0164] In the formula, y d , y q are the continuous output quantities of the d-axis and q-axis resonant disturbance estimators;

[0165] The predicted current of the final output is obtained from formulas (17), (18), and (26) as shown in the following formulas (27) and (28):

[0166]

[0167]

[0168] where y d (k) and y q (k) are the discrete output quantities of the resonant disturbance estimators under the d-axis and q-axis.

[0169] Step 5 is specifically as follows:

[0170] Based on Steps 2, 3, and 4, the calculation of the AC and DC axis voltage control command values can be re-obtained from formula (8) as shown in the following formula (29):

[0171]

[0172] where is the d-axis predicted current at time k + 1 after adding the quasi-resonant disturbance estimator, is the q-axis predicted current at time k + 1 after adding the quasi-resonant disturbance estimator, is the estimated value of the d-axis hyperlocal model parameter at time k + 1 after adding the quasi-resonant disturbance estimator, is the estimated value of the q-axis hyperlocal model parameter at time k + 1 after adding the quasi-resonant disturbance estimator.

[0173] Among them, the block diagram of the SynRM deadbeat predictive current control system with a resonant sliding mode lumped disturbance observer based on the hyperlocal model is as shown in Figure 1 shown. A synchronous reluctance motor vector control system is adopted, including a signal main circuit, a detection circuit, and a control circuit. The main circuit includes a three-phase inverter and a synchronous reluctance motor. The detection circuit is a current detection circuit and a sensor module. The current detection circuit is used to detect the current signal of the synchronous reluctance motor. The sensor module uses a position sensor to obtain the position and speed of the rotor of the synchronous reluctance motor. The control circuit includes a Clark transformation module, a Park transformation module, a resonant sliding mode lumped disturbance observer module, a maximum torque per ampere (MTPA) module, a deadbeat predictive current module under the hyperlocal model, a Park inverse transformation module, and an SVPWM modulation module, which are mainly used to process the signals obtained by the signal detection circuit to obtain the control signals for controlling the main circuit;

[0174] Among them, the working process of the SynRM deadbeat predictive current control system based on the resonant sliding mode lumped disturbance observer under the hyperlocal model is as follows: The current signal detection circuit detects the three-phase input currents \(i\) a 、\(i\) b 、\(i\) c of the synchronous reluctance motor in the three-phase stationary coordinate system through Hall sensors. The three-phase input currents are converted into the currents \(i\) α 、\(i\) β in the stationary two-phase coordinate system through Clark transformation (3s / 2s); The position sensor can obtain the position angle \(\theta\) e and the electrical angular velocity \(\omega\) e of the rotor of the synchronous reluctance motor; The sliding mode lumped disturbance observer under the hyperlocal model uses the voltage signals \(u\) d \((k)\), \(u\) q \((k)\) and the current signals \(i\) d \((k)\), \(i\) q \((k)\) of the motor to estimate the currents \(i\) d \((k + 1)\), \(i\) q \((k + 1)\) and the disturbances \(F\) d \((k + 1)\), \(F\) q \((k + 1)\) of the synchronous reluctance motor in the next beat. The difference between the speed reference value and the speed feedback value \(\omega\) r of the speed outer loop is taken to obtain the speed error, which passes through a PI regulator to obtain the electromagnetic torque reference value and passes through the MTPA module to obtain the dq-axis stator current reference values and the predicted currents \(i\) d \((k + 1)\), \(i\) q \((k + 1)\) and the disturbances \(F\) d \((k + 1)\), \(F\) q \((k + 1)\) obtained by the observer. After DPCC regulation, the dq-axis voltage reference values are converted into two-phase voltages in the stationary two-phase coordinate system after inverse Park transformation (2r / 2s) inverse transformation The two-phase voltages generate PWM waves through the SVPWM generation module and then drive the synchronous reluctance motor (SynRM) to work after passing through the three-phase inverter;

[0175] Among them, the parameters of the synchronous reluctance motor used in the simulation are shown in Table 1. The speed of the motor is set to 1500 r / min in the simulation, Figure 3 the rated torque of the motor is loaded to 35 N·m at 1 s in the simulation results, and the rising slope is set to 200π N·m / s, Figure 3 (a) is the simulation waveform without using the torque ripple suppression method proposed in the present invention; Figure 3(b) is the simulation waveform using the torque ripple suppression method proposed in the present invention. By comparing Figure 3 (a) and Figure 3 (b), it can be found that the method proposed in the present invention can suppress torque ripple.

[0176] Table 1 Parameters of the synchronous reluctance motor

[0177] parameter value parameter value d-axis stator inductance / mH 96.97 rated power / Kw 5.5 q-axis stator inductance / mH 17.4 rated frequency / Hz 50 stator resistance / Ω 0.5545 rated speed / (r / min) 1500 <![CDATA[Moment of inertia / (kg·m 2 )]]> 0.02 rated current / A 16 friction coefficient / (Nms / rad) 0.01 rated voltage / V 380 number of pole pairs 2 rated torque / N·m 35

[0178] The content not described in detail in this specification belongs to the prior art well-known to those skilled in the art. In the description of the present invention, unless otherwise specified, the meaning of "a plurality of" is two or more; the terms "upper", "lower", "left", "right", "inner", "outer", "front end", "rear end", "head", "tail", etc. indicate the orientation or positional relationship based on the orientation or positional relationship shown in the drawings, and are only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation, and thus cannot be construed as a limitation of the present invention. In addition, the terms "first", "second", "third", etc. are only used for descriptive purposes and cannot be construed as indicating or implying relative importance. In the description of the present invention, it should be noted that unless otherwise clearly specified and defined, the terms "connected" and "connected" should be understood in a broad sense. For example, it can be a fixed connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be directly connected or indirectly connected through an intermediate medium. For those of ordinary skill in the art, the specific meanings of the above terms in the present invention can be understood in specific cases.

[0179] For those skilled in the art, it is obvious that the present invention is not limited to the details of the above exemplary embodiments, and can be implemented in other specific forms without departing from the spirit or basic characteristics of the present invention. Therefore, from any point of view, the embodiments should be regarded as exemplary and non-limiting. The scope of the present invention is defined by the appended claims rather than the above description. Therefore, it is intended to embrace all changes falling within the meaning and scope of the equivalent elements of the claims in the present invention. Any reference signs in the claims should not be regarded as limiting the claims involved.

Claims

1. A SynRM high-quality resonant deadbeat predictive current control method, comprising the following steps: Step 1: Establish a synchronous reluctance motor hyperlocal model in a two-phase rotating coordinate system; Specifically: The voltage equations of SynRM under the rotating coordinate axes of the d-q axes are shown in the following formula (1): where, u d and u q are the d-axis and q-axis components of the stator voltage, R s is the stator resistance, i d and i q are the d-axis and q-axis components of the stator current, L d and L q are the d-axis and q-axis stator inductances respectively, ω e is the electrical angular velocity of the rotor, ψ d and ψ q are the d-axis and q-axis components of the stator flux linkage; According to the input and output of the synchronous reluctance motor control system, the non-linear, complex and variable system is replaced by a hyperlocal model. The first-order hyperlocal model of a single-input single-output system is shown in the following formula (2): Wherein, u and y are the control and output variables respectively; α represents a non-physical scaling factor selected by the designer; F represents the known part and unmodeled dynamics of the system; According to formula (1) and formula (2), the hyperlocal models of the d-axis and q-axis current loops are shown in the following formula (3): where p is the differential operator, ω m is the mechanical angular velocity, α d is the d-axis design voltage parameter, α q is the q-axis design voltage parameter, F d is the d-axis design system perturbation, F q is the q-axis design system perturbation, and Step 2: Design a deadbeat predictive current controller for the synchronous reluctance motor from the hyperlocal model obtained in Step 1; Step 3: Design a sliding mode lumped disturbance observer to obtain the system disturbance estimation value in Step 2; Step 4: Design a resonant sliding mode lumped disturbance observer on the basis of Step 3 to suppress current harmonics and torque harmonics; Step 5: Calculate the control command values of the AC and DC axis voltages.

2. The SynRM high-quality resonant deadbeat predictive current control method according to claim 1, characterized in that Wherein Step 2 is specifically: The voltage u is obtained according to Equation (3). d and u q are as shown in Equation (4) below: Discretize formula (4) according to the forward Euler formula as shown in the following formula (5): where, u d (k) is the d-axis voltage at time k, u q (k) is the q-axis voltage at time k, i d (k + 1) is the d-axis current at time k + 1, i q (k + 1) is the q-axis current at time k + 1, i d (k) is the d-axis current at time k, i q (k) is the q-axis current at time k, F d (k) is the d-axis disturbance at time k, F q (k) is the q-axis disturbance at time k, T s is the sampling time; When calculating the desired voltage, take the current reference value as the current value at the k+1 moment, as shown in the following formula (6): Among them, is the d-axis reference current at the (k + 1)-th moment, is the q-axis reference current at the (k + 1)-th moment; Combine formula (5) and formula (6) to obtain the predictive current controller as shown in the following formula (7): Among them, are the d-axis and q-axis voltage control commands obtained through predictive calculation for the current cycle. These voltage commands will act on the motor through the inverter in the next cycle after SVPWM, enabling the stator current of the motor to track the given value without deadbeat at the next moment. represents i d (k), the estimated value of i q (k). represents F d (k), the estimated value of F q (k). When designing the controller, the stator current and disturbance at the (k + 1)-th moment are predicted one beat forward to obtain the following formula (8): Among them, represents i d (k + 1), i q the estimated value of (k + 1), represents F d (k + 1), F q the estimated value of (k + 1).

3. A SynRM high-quality resonant deadbeat predictive current control method according to claim 2, characterized in that, Wherein Step 3 is specifically: To improve parameter F d and F q For the purpose of improving the accuracy and rapidity of estimation and the anti-interference ability of parameter estimation, the design of a sliding-mode lumped disturbance observer based on a super-local model is shown in the following formulas (9) and (10): wherein, are the derivatives of the estimated values of the d-axis and q-axis currents respectively, are the estimated values of the d-axis and q-axis hyper-local model parameters respectively, are the derivatives of the estimated values of the d-axis and q-axis hyper-local model parameters respectively, ξ is the gain coefficient of the control law, σ d and σ q are the d-axis and q-axis sliding mode reaching laws respectively; The selection of the sliding mode surface is shown in the following formula (11): Design an approach law function to effectively improve the convergence speed of the sliding mode observer and reduce chattering. Its expression is shown in the following formula (12): Among them, λ, η, and α are parameters in f(s), obtained from formula (13). When the system state is far from the sliding surface, that is, when |s| increases, f(s) will converge to β << 1 + λ is a very small value; when the system state is close to the sliding surface, that is, when |s| decreases, f(s) will converge to |s| α <1, -qs causes the chattering of the system to decay exponentially, thus ensuring the stability of the system. This adaptive reaching law can both reduce the chattering of the system and accelerate the convergence speed; Obtain the d-axis and q-axis sliding mode approach laws from formula (12) as shown in the following formula (14): According to formula (9), (10) and (14), obtain the estimated values of the d-axis and q-axis currents as shown in the following formula (15) and (16): Discretize formula (15) and formula (16) through the forward Euler formula as shown in the following formula (17) and (18):

4. A SynRM high-quality resonant deadbeat predictive current control method according to claim 3, characterized in that Wherein Step 4 is specifically: The 5th and 7th phase current harmonics in the abc three-phase stationary coordinate system are converted to the 6th harmonic under the d-q axes. The currents under the d-q axes are shown in the following formula (19): Wherein, I1, I5 and I7 are the amplitudes of the fundamental wave, 5th and 7th harmonics of the current respectively; θ1, θ5 and θ7 are the phases of the fundamental wave, 5th and 7th harmonics respectively; The transfer function of the designed quasi-resonant controller is shown in the following formula (20): Among them, ω c is the cut-off frequency of the quasi-resonant controller, ω0 is the resonant frequency, and ω0 = 6ω is taken according to formula (19). e , K i is the integral gain; Discretize formula (20) by bilinear transformation as shown in the following formula (21): Among them, Arrange formula (21) to obtain the following formula (22): Y(z)+a1z -1 Y(z)+a2Y(z) = b0U(z)+b1z -1 U(z)+b2z -2 U(z) (22) The conversion relationship is shown in the following formula (23): According to formula (22) and (23), obtain the output difference equation as shown in the following formula (24): y(k) = b0u(k) + b2u(k - 2) - a1y(k - 1) - a2y(k - 2) (24) The input of the difference equation in the system is e d and e q , so formula (24) is rewritten as: Based on the basic sliding mode lumped disturbance observer, a quasi-resonant disturbance estimator is added to the loop for observing the disturbance to compensate for the AC disturbance as shown in the following formula (26): where y d and y q are the continuous output quantities of the d-axis and q-axis resonant disturbance estimators; Obtain the finally output predictive current from formula (17), (18) and (26) as shown in the following formula (27) and (28): where y d (k), y q (k) are the discrete output quantities of the resonant disturbance estimators under the d-axis and q-axis.

5. A SynRM high-quality resonant deadbeat predictive current control method according to claim 4, characterized in that Wherein Step 5 is specifically: Based on Steps 2, 3, and 4, the calculation of the command values for the armature and field voltages in Formula (8) can be obtained again as shown in the following Formula (29): Among them, is the predicted d-axis current at the (k + 1)-th moment after adding the quasi-resonant disturbance estimator, is the predicted q-axis current at the (k + 1)-th moment after adding the quasi-resonant disturbance estimator, is the estimated value of the d-axis hyper-local model parameter at the (k + 1)-th moment after adding the quasi-resonant disturbance estimator, is the estimated value of the q-axis hyper-local model parameter at the (k + 1)-th moment after adding the quasi-resonant disturbance estimator.

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