Power conversion system

Abstract

A power conversion system in which a converter (10) and an inverter (20) are connected through a DC link (30) having an inductive component, the switching frequency of the PWM-controlled converter (10) and inverter (20) is set to be equal, and the switching frequency is set to be higher than the resonance frequency of a resonance circuit of the DC link (30) including a first capacitor (C c ), a second capacitor (C i ), and a cable, etc., in a manner such that the phase of a prescribed component of the voltage pulsation of the capacitors (C c , C i ) resulting from the switching action of the converter (10) and inverter (20) becomes substantially in phase, and the switching action of at least one of the converter (10) and inverter (20) is controlled.

Description

Technical Field

[0001] This invention relates to a power conversion system for driving an AC motor by AC / DC / AC conversion through a converter and inverter controlled by PWM (Pulse Width Modulation), and more specifically, to a technique for reducing the pulsation of current flowing through the DC circuit between the converter and the inverter. Background Technology

[0002] Various power conversion systems of this kind have been provided in the past. However, if the carrier frequencies used for PWM control of the converter and inverter are different, the current ripple in the capacitors (in voltage-source systems) and reactors (in current-source systems) flowing through the DC circuit between the converter and inverter increases, resulting in unstable control. In this case, the required capacitance of the capacitors and reactors increases.

[0003] Therefore, for example, in Patent Document 1, the carrier wave on the converter side and the carrier wave on the inverter side are set to the same waveform or an inverted waveform, or to waveforms with a specified phase difference at the same frequency, thereby reducing the current pulsation of the DC circuit.

[0004] In addition, Patent Document 2 describes a method for suppressing the resonant current flowing through the DC circuit between the converter and the inverter, synchronizing the frequency and phase of the carrier waves of the converter and the inverter, and adding a DC capacitor in the DC circuit such that the resonant current decreases when the phase difference between the two carrier waves is zero.

[0005] It should be noted that Patent Document 3 discloses a DC power transmission system in which capacitors for voltage smoothing are respectively included in the DC voltage sections of the converter and inverter connected via DC power supply lines. Here, the aforementioned capacitors suppress DC voltage surges caused by the switching of the converter and inverter, thereby preventing overvoltage damage to components, and also serve to stabilize the DC voltage when the AC power supply and load power change.

[0006] <Prior art documents>

[0007] <Patent Documents>

[0008] Patent Document 1: Japanese Patent Application Publication No. 4-121065 (Page 5, top left column, line 20 to Page 6, top left column, line 1, Figure 1) Figure 8 wait)

[0009] Patent Document 2: Japanese Patent Application Publication No. 2017-204976 (

[0017] ,

[0018] , Figures 1-) Figure 3 wait)

[0010] Patent Document 3: Japanese Patent No. 4373040 (

[0013] , Figure 1, etc.) Summary of the Invention

[0011] <Problem to be solved by the invention>

[0012] Patent Document 1 does not disclose a system in which the converter and inverter, which include capacitors in the DC voltage section, are connected via a DC circuit, as in Patent Document 3.

[0013] Furthermore, in the prior art of Patent Document 2, the DC circuit is constructed by simulating the reduction of the resonant current, which is determined by the impedance of the DC circuit, the conditions of the two power systems, and the voltage and current of the converter and inverter. However, the principle of suppressing the resonant current of the DC circuit is not specifically shown, nor is the connection configuration and function of the capacitors, reactors, etc. connected in the DC circuit clearly disclosed.

[0014] Moreover, the prior art of Patent Document 3 is an invention that suppresses low-order harmonics flowing into the AC power system by equipping the control circuits of the converter and inverter with active filter functions, and it does not address the issue of reducing the current pulsation of the DC circuit.

[0015] The problem solved by the present invention is to provide a power conversion system capable of reducing the pulsation of the current flowing through a DC circuit. The power conversion system has a resonant circuit, which consists of a DC circuit between the converter and the inverter (an example of a DC coupling unit) and capacitors connected to the DC voltage sections of the converter and the inverter, respectively.

[0016] <Methods for solving problems>

[0017] This invention relates to the following aspects. A first aspect is a power conversion system, comprising:

[0018] AC power supply;

[0019] The converter converts the AC power from the aforementioned AC power source into DC power through PWM control.

[0020] The inverter converts the DC power output from the converter into AC power through PWM control and supplies it to the AC motor.

[0021] The first capacitor is connected to the DC voltage section of the aforementioned converter;

[0022] The second capacitor is connected to the DC voltage section of the inverter; and

[0023] A DC coupling section connects the first capacitor and the second capacitor, and has an inductive component.

[0024] The switching frequencies of the converter and the inverter are set to be equal, and the switching frequency is set to be higher than the resonant frequency of the resonant circuit including the first capacitor, the second capacitor, and the DC coupling section.

[0025] The switching operation of at least one of the converter and the inverter is controlled in such a way that the phase of a predetermined component of the voltage ripple of the first capacitor and the second capacitor generated by the switching operation of the converter and the inverter is approximately in phase.

[0026] In the second method, in the power conversion system described in the first method, PWM pulses are generated by comparing the voltage command value and the carrier wave, respectively, and the carrier waves on the converter side and the inverter side are set to the same frequency, and the two carrier waves have a specified phase relationship.

[0027] In the third method, in the power conversion system described in the second method, the number of phases of the AC power supply and the AC motor are set to be equal, and at least one of the converter and the inverter is controlled in such a way that the fundamental frequency of the voltage of one phase of the AC power supply and the AC motor is the same and they are approximately in phase, and the phases of the carrier waves on the converter side and the inverter side are reversed.

[0028] In the fourth method, in the power conversion system described in the second method, the number of phases of the AC power supply and the AC motor are set to be equal, and at least one of the converter and the inverter is controlled in such a way that the fundamental frequency of the voltage of one phase of the AC power supply and the AC motor is the same and is substantially out of phase, and the phases of the carrier waves on the converter side and the inverter side are set to be in phase.

[0029] In the fifth approach, in the power conversion system described in any of the first to fourth approaches, at least one of the converter and the inverter has multiple power conversion units connected in parallel with DC voltage units.

[0030] In the sixth method, in the power conversion system described in the fifth method, the switching frequencies of the plurality of power conversion units are set to be the same, and the timing of the pulse generation of the DC bus current of the plurality of power conversion units constituting the converter or the inverter is staggered.

[0031] In the seventh method, in the power conversion system described in the sixth method, the timing of the pulse generation of the DC bus current of the aforementioned multiple power conversion units is distributed approximately equally.

[0032] In the eighth method, in the power conversion system described in the sixth method, the switching frequency component in the voltage pulsation of the first capacitor or the second capacitor is set to be approximately in phase in the converter and the inverter. The higher harmonic components of the switching frequency are canceled out by staggering the timing of the pulses generated by the DC bus currents of the plurality of power conversion units.

[0033] In the ninth method, in the power conversion system described in the sixth method, the timing of the pulses generated by the DC bus currents of the plurality of power conversion units is staggered for the switching frequency component in the voltage pulsation of the first capacitor or the second capacitor, and the higher harmonic components of the switching frequency are offset by setting them to be approximately in phase in the converter and the inverter.

[0034] In the tenth method, in the power conversion system described in any of the sixth to ninth methods, PWM pulses are generated by comparing the voltage command value and the carrier wave to be given to the plurality of power conversion units respectively, the carrier waves are set to the same frequency and the carrier waves have a predetermined phase relationship, and the carrier waves used in the converter and the inverter are set to the same frequency and the carrier waves have a predetermined phase relationship.

[0035] In the eleventh method, in the power conversion system described in the tenth method, the number of phases of the AC power supply and the AC motor are set to be equal, and the frequency of the fundamental wave of the AC side voltage is set to be the same. In the converter and the inverter, the amplitude of the fundamental wave of the AC side voltage and the AC side current of the plurality of power conversion units is set to be approximately equal.

[0036] In the twelfth method, in the power conversion system described in any of the fifth to eleventh methods, when a portion of the plurality of power conversion units is stopped, control is performed such that the phase of a predetermined component of the voltage pulsation of the first or second capacitor caused by the switch is substantially in phase in the converter and the inverter.

[0037] In the thirteenth method, in the power conversion system described in any of the first to twelfth methods, the AC power source is an AC generator driven by an external force. The current of the AC generator is controlled by the converter in such a way that the average DC voltage of the converter or the inverter becomes a predetermined value. The frequency of the AC generator current is given to the inverter as a quantity equivalent to the frequency command of the AC motor.

[0038] In the fourteenth embodiment, in the power conversion system described in any of the first to twelfth embodiments, the AC power source is an AC generator driven by an external force. A control device that controls the current of the AC generator via the converter, such that the average DC voltage of the converter or inverter is a predetermined value, and applies the frequency of the AC motor current as a quantity equivalent to the frequency command of the AC generator to the external force.

[0039] In the fifteenth method, in the power conversion system described in the thirteenth or fourteenth method, the amount equivalent to the output power of the inverter is added to the amount equivalent to the input power command value of the converter.

[0040] In the sixteenth embodiment, in the power conversion system described in any of the first to fifteenth embodiments, at least one of the following is adjusted to reduce the current pulsation flowing through the DC coupling section: the phase relationship between the carrier waves used for PWM control of the converter and the inverter, or the phase relationship between the fundamental AC voltages of the converter and the inverter, or the phase relationship between the fundamental AC currents.

[0041] In the seventeenth embodiment, the power conversion system described in any of the first to sixteenth embodiments is a system that drives the aforementioned AC motor of the electric fan that propels the aircraft.

[0042] <The Effects of the Invention>

[0043] According to one aspect of the present invention, without adding new components or parts, the pulsation of the current flowing through the DC coupling section between the converter and the inverter can be reduced, thereby reducing losses caused by pulsating current and stabilizing control. Furthermore, by reducing the responsibility of the capacitors in the DC voltage section for reducing pulsating current, the electrostatic capacitance of the capacitors is reduced, making miniaturization and cost reduction of the device possible. Attached Figure Description

[0044] Figure 1A This is a main circuit configuration diagram of a power conversion system according to one embodiment of the present invention.

[0045] Figure 1B This is a block diagram of the control circuit constituting the main circuit of a power conversion system according to one embodiment of the present invention.

[0046] Figure 2 It is Figure 1A A simplified circuit diagram.

[0047] Figure 3 It is shown Figure 2 A graph showing the frequency response of the absolute value of the impedance.

[0048] Figure 4A It is a waveform diagram showing the voltage of the capacitors on the converter side and the inverter side, and the current of the DC coupling section.

[0049] Figure 4B It is a waveform diagram showing the voltage of the capacitors on the converter side and the inverter side, and the current of the DC coupling section.

[0050] Figure 5 It is shown Figure 2 The diagram shows an example of the operating waveforms on the converter side of the power conversion system.

[0051] Figure 6A This is a waveform diagram showing the AC side current, AC side voltage fundamental, carrier, capacitor voltage, and DC coupling current when the carriers on the converter side and inverter side are set to be in phase.

[0052] Figure 6B This is a waveform diagram showing the AC side current, AC side voltage fundamental, carrier, capacitor voltage, and DC coupling current when the carrier on the converter side and inverter side are set to reverse phase.

[0053] Figure 7A It shows the waveforms of AC side current, AC side voltage fundamental frequency, carrier frequency, cutoff current, capacitor voltage, and DC coupling current under the condition that the power factor changes on the converter side and the inverter side.

[0054] Figure 7B It shows the waveforms of AC side current, AC side voltage fundamental frequency, carrier frequency, cutoff current, capacitor voltage, and DC coupling current under the condition that the power factor changes on the converter side and the inverter side.

[0055] Figure 8 This is a main circuit configuration diagram of a power conversion system according to another embodiment of the present invention.

[0056] Figure 9A This is a waveform diagram showing the AC side current, AC side voltage fundamental frequency, carrier frequency, cutoff current, capacitor voltage, and DC coupling current when both the converter and inverter are composed of a single power conversion unit.

[0057] Figure 9B This is a waveform diagram showing the AC side current, AC side voltage fundamental frequency, carrier frequency, cutoff current, capacitor voltage, and DC coupling current when both the converter and inverter are composed of multiple power conversion units connected in parallel.

[0058] Figure 10This is a waveform diagram showing the AC side current, AC side voltage fundamental frequency, carrier frequency, cutoff current, capacitor voltage, and DC coupling current when both the converter and inverter are composed of multiple power conversion units connected in parallel.

[0059] Figure 11 It is shown Figure 1B A block diagram of a modified example of the control circuit.

[0060] Figure 12 This is a configuration diagram of applying the present invention to an aircraft propulsion system. Detailed Implementation

[0061] Hereinafter, embodiments of the present invention will be described with reference to the figures.

[0062] Figure 1A This is a main circuit diagram illustrating one embodiment of the power conversion system of the present invention. Figure 1B This is a block diagram illustrating an example of its control circuit.

[0063] exist Figure 1A In this configuration, the AC power supply G includes a three-phase AC generator that obtains rotational force from an external force, such as an engine. The AC power supply G is connected to a three-phase voltage-source converter (hereinafter simply referred to as converter) 10, and the DC side of converter 10 is connected to a three-phase voltage-source inverter (hereinafter simply referred to as inverter) 20 via a DC coupling section 30, such as a cable. The DC coupling section 30 is an example of a DC coupling unit. The inverter 20 converts the DC power input through the DC coupling section 30 into AC power and outputs it to an AC motor M, such as a three-phase synchronous motor or asynchronous motor. The AC motor M generates a predetermined torque to drive a load (not shown).

[0064] The converter 10 includes a power conversion section 11 consisting of semiconductor switching elements (hereinafter referred to as switching elements) S1 to S6 such as IGBTs and FETs connected by a three-phase bridge, and a voltage smoothing capacitor C connected between its DC output terminals as a first capacitor. c The inverter 20 includes a capacitor C connected via a DC coupling section 30. c The capacitor C connected as a second capacitor for voltage smoothing i and including the switching element S connected at both ends by a three-phase bridge. 21 ~S 26 Power conversion unit 21.

[0065] In the above configuration, the converter 10 converts AC voltage to DC voltage through the switching action of switching elements S1 to S6, and the DC voltage is then passed through capacitor C. cThis smooths the flow and supplies it to the inverter 20 via the DC coupling section 30. In the inverter 20, through capacitor C... i Smooth the input DC voltage and through the switching element S 21 ~S 26 The switching action converts it into AC voltage, which is then supplied to the AC motor M.

[0066] Hereinafter, the attached figures will be labeled C. c C i It is used in the sense of capacitor and capacitance. Additionally, through L... ci This indicates the inductance of the DC coupling section 30. It should be noted that this inductance L... ci This includes the inductance of the cable itself, which serves as the DC coupling part 30, and the inductance of the DC reactor and the cable when a DC reactor, which is a component, is connected in the middle of the cable.

[0067] (1) Implementation of the first method

[0068] like Figure 1A As shown, this embodiment is characterized in that it has a capacitor C in the DC circuit between the converter 10 and the inverter 20. c C i and inductor L ci A power conversion system using a resonant circuit (hereinafter also referred to as a CLC resonant circuit). Furthermore, this embodiment is characterized in that the switching frequencies of the converter 10 and the inverter 20 are set to be equal and higher than the resonant frequency of the CLC resonant circuit. Moreover, this embodiment is characterized in that the capacitor C generated by the switching of the converter 10 and the inverter 20... c C i The switching operation of at least one of the converter 10 and inverter 20 is controlled in such a way that the phase of the specified components of the voltage ripple is approximately in phase. Consequently, the current I of the DC coupling section 30, which is caused by the difference in the aforementioned voltage ripple, is... ci The pulsation decreases.

[0069] First, the relationship between the resonant frequency of the CLC resonant circuit and the switching frequencies of the converter 10 and the inverter 20 is studied.

[0070] If the switching frequency or its higher harmonics frequency coincides with the resonant frequency, then C c C i L ci The vibration current in the circuit is not restricted and increases, thus affecting the capacitor C. c C iOvervoltage or overcurrent in the aforementioned circuit could damage the device. Even if such a situation is avoided, there may be a case where the switching frequency is lower than the resonant frequency. In this case, observing from the power conversion section 11 of the converter 10 or the power conversion section 21 of the inverter 20, the capacitor C located in its DC voltage section... c Capacitor C i Compared to, inductor L ci The switching frequency has a low impedance. Therefore, most of the pulsating component of the current generated by the switching action is not absorbed by the capacitor C. c C i Absorbed and flowing through inductor L ci Capacitor C c C i Its original function as a voltage smoothing element has been compromised.

[0071] As can be seen from the above, setting the switching frequency of the converter 10 and the inverter 20 (power conversion units 11 and 21) to be higher than the resonant frequency of the CLC resonant circuit is a condition for stabilizing the system operation.

[0072] use Figure 2 The diagram further illustrates the points mentioned above.

[0073] Figure 2 It is Figure 1A A simplified diagram. In this... Figure 2 In the circuit, the frequency response of the absolute value |Z| of the impedance Z observed from the converter 10 side becomes, for example... Figure 3 Therefore, if we formulate it, we can obtain Equation 1. It should be noted that the impedance observed from the inverter 20 side is also the same; the C value in the numerator of the second fractional term on the right side of Equation 1 is... i Become C c .

[0074] (Mathematical Formula 1)

[0075]

[0076] According to mathematical formula 1, the impedance Z has two singularities. The frequency of each singularity is the frequency f at which |Z| is zero. r1 (ω=ω r1 =2πf r1 ), and |Z| is an infinitely large frequency f. r2 (ω=ω r2 =2πf r2 ), which are obtained from mathematical expression 2 and mathematical expression 3 respectively.

[0077] (Mathematical Formula 2)

[0078]

[0079] (Mathematical Formula 3)

[0080]

[0081] Here, in Figure 3 Although |Z| is finite at singular points, it takes into account the resistive component in actual circuits, which is not the essential issue. Furthermore, since this resistive component generally has less influence on L... ci C c C i Its impact is relatively small, so it is ignored in the formula.

[0082] According to mathematical expressions 2 and 3, since f must exist... r1 <f r2 Therefore, the switching frequencies of converter 10 and inverter 20 need to be set to a value higher than that of inverter 20. Figure 3 f r2 High. In other words, if the switching frequency is related to f... r1 or f r2 When they are in sync, they become a resonance state, which in turn leads to overvoltage and overcurrent. If the ratio is greater than f... r2 The low current is due to most of the current component of the switching action flowing through the inductor L. ci It should be noted that, in order to ensure stable system operation, if the switching frequencies of converter 10 and inverter 20 are related to f... r2 When the frequency approaches a certain point, it becomes a near-resonance state. Therefore, in practical applications, it is preferable to set the switching frequency to f. r2 It is roughly twice as much as that.

[0083] Next, according to Figure 2 It is clearly known that, due to the capacitor C based on the switching of converter 10 and inverter 20 c C i The voltage fluctuation difference is applied to reactance L ci Therefore, by suppressing this difference, the current-carrying reactance L can be reduced. ci The pulsation of the current.

[0084] In the above Figure 2 In this circuit, the power conversion sections 11 and 21 of the converter 10 and inverter 20 respectively apply PWM-controlled voltages to the AC power supply G and the AC motor M to directly or indirectly control the current. The AC power supply G and the AC motor M typically have fundamental and higher harmonic voltages. A rectangular wave voltage based on PWM control is superimposed on these voltages. Thus, the voltage difference between the two is applied to a reactor or similar device connected between the AC power supply G and the AC motor M, thereby smoothing the current. However, since the aforementioned reactor or similar device is mostly replaced by the inductive component present in the AC power supply G and the AC motor M, therefore... Figure 2In this context, descriptions of reactors and the like are omitted; only the inductance L of the DC coupling unit 30j is described. ci .

[0085] As described above, the current flowing from the AC power source G through the converter 10 and the current flowing from the AC motor M through the inverter 20 are continuous. These continuous currents are interrupted by the switches of the power conversion units 11 and 21, resulting in pulsed currents (hereinafter referred to as interrupted currents) flowing to the DC circuit side, becoming the current supplied by capacitor C. c C i and inductor L ci The input to the CLC resonant circuit is formed. Since the waveform of the cut-off current is actually determined by the current of the AC power supply G and the switch of the power conversion unit 11, and the current of the AC motor M and the switch of the power conversion unit 21, it acts as a current source for the CLC resonant circuit.

[0086] Furthermore, the switching frequencies of the power conversion units 11 and 21 are set to a value higher than the resonant frequency f of the CLC resonant circuit observed from each unit. r2 Under high conditions, most of the interrupted current flows from the power conversion unit 11 into the capacitor C. c And the power flows from the power conversion unit 21 into the capacitor C i As a result, in capacitor C c C i In this process, a pulsating voltage is superimposed on a specified DC voltage. Furthermore, because the DC component of the interrupting current does not pass through capacitor C... c C i Therefore, it is through the inductor L of the DC coupling section 30 ci The current flows between the converter 10 and the inverter 20. Thus, the power required by the AC motor M is supplied by the AC power source G, and conversely, through regenerative operation, power is supplied from the AC motor M to the AC power source G.

[0087] For capacitor C generated by the switching action of power conversion units 11 and 21 c C i The voltage pulsation will be further explained.

[0088] From the above explanation, it is clear that, as a capacitor C... c C i The voltage ripple difference is applied to inductor L ci As a result, current pulsation occurs. Therefore, if capacitor C can be... c C i If the voltage ripple difference is suppressed to a small value, then the inductor L can be made smaller. ci The current pulsation in the medium decreases.

[0089] The capacitor C caused by the switch of the power conversion section 11, 21 c C i The voltage ripple has a component related to the switching frequency. Therefore, in order to make capacitor C c C i The voltage ripple difference is suppressed to a small extent. For the "specified component" in this component, as long as it is within capacitor C... c and capacitor C i The switching operations of power conversion units 11 and 21 can be controlled in a manner that ensures they are approximately in phase. In this case, "specified component" refers to capacitor C. c C i The voltage ripple has a "major frequency component", a "frequency component that is particularly desired to be reduced", or a "time component" in the time waveform of the voltage ripple during the period of large voltage amplitude.

[0090] The diagram that specifically illustrates the points mentioned above is... Figure 4A as well as Figure 4B .

[0091] Figure 4A as well as Figure 4B The capacitor C is shown on the converter 10 side and the inverter 20 side. c C i voltage E c E i and the current I of the DC coupling section 30 ci waveform, Figure 4A It is E c (shown by solid line), E i (Shown by dashed lines) The case is roughly the opposite phase. Figure 4B It is E c E i In roughly the same phase. Voltage E in any case. c E i The difference between these voltage ripples is applied to the inductor L of the DC coupling section 30. ci Thus, current I flows through it. ci .

[0092] exist Figure 4A In the middle, for roughly out-of-phase E c E i The voltage pulsation, since the larger the amplitude of each voltage, the larger the voltage difference, therefore, during this period, the current flowing through inductor L... ci I ci It contains large current ripples. On the other hand, in Figure 4B In the middle, due to E c E i The voltage ripples are approximately in phase, therefore the voltage applied to inductor L is thus... ciThe ripple of the differential voltage decreases, I ci The current pulsation also decreases.

[0093] Therefore, if Figure 4B That way with E c E i By controlling the switching operations of converter 10 and inverter 20 in a manner that makes the voltage ripples approximately in phase, the current I flowing through DC coupling section 30 can be suppressed. ci The pulse.

[0094] (2) Implementation of the second method

[0095] Figure 5 yes Figure 2 An example of the operating waveforms of the power conversion system is shown above, depicting the carrier wave (triangular wave), voltage command value, cutoff current, and capacitor C on the converter 10 side. c voltage E c and the current I of the DC coupling section 30 ci .

[0096] for Figure 5 The cutoff current is used to illustrate the timing when the carrier wave reaches its maximum value, indicated by auxiliary lines. From this, it can be seen that the capacitor C based on the cutoff current... c The frequency of the voltage pulsations is dominated by the frequency of the carrier wave. Furthermore, detailed observation reveals that capacitor C... c The voltage ripple increases during pulses containing cutoff current and decreases during pulses where cutoff current is absent (when the cutoff current is zero). This can be understood as the cutoff current flowing into capacitor C. c It should be noted that on the inverter 20 side (not shown), capacitor C... i voltage E i It decreases during periods when a pulse with interrupted current is present and increases during periods when no pulse is present.

[0097] In PWM converters and PWM inverters, it is common practice to generate PWM pulses by comparing a carrier wave, represented as a triangular wave, with a voltage command value. In this case, it is well known that the current between the capacitors in the power conversion section and the DC voltage section of the converter / inverter is a pulsed cutoff current, the main frequency component of which is an integer multiple of the carrier wave frequency. Consequently, the frequency components of the capacitor voltage pulsation include the fundamental frequency component of the carrier wave, its higher harmonic components, and multiple components from its sideband waves.

[0098] Therefore, as described above, through capacitor C c C i The difference in voltage ripple determines the current ripple in the DC coupling section 30. To reduce this current ripple, the two capacitors C...c C i Setting the voltage pulsation to be approximately in phase is effective. Accordingly, when generating PWM pulses for converter 10 and inverter 20 by comparing the voltage command value and the carrier wave, it is preferable to set the carrier wave frequencies used in converter 10 and inverter 20 to be equal and relative to the frequencies of the two capacitors C. c C i The voltage pulsation is adjusted to make the phase relationship of the two carriers approximately in phase.

[0099] Assuming the frequencies of the two carrier waves are not equal, capacitor C c C i The voltage ripples are out of phase with time, making it impossible to suppress the current ripples in the DC coupling section 30. Furthermore, it is also impossible to suppress the current ripples in the DC coupling section 30 when the specified phase relationship cannot be maintained between the two carrier waves.

[0100] (3) Examples of the third and fourth methods

[0101] Next, as embodiments of the third and fourth methods, a specific method for reducing the current ripple of the DC coupling section 30 will be described.

[0102] Generally, in PWM converters and PWM inverters, the capacitor C in the steady state... c C i The voltage pulsation amplitude varies periodically at twice the number of phases of the fundamental wave on the AC side. Therefore, in order to make capacitor C c C i The timing for the voltage ripple to increase is consistent on both the converter 10 side and the inverter 20 side. First, the number of phases of the converter 10 and the inverter 20, in other words, the number of phases of the AC power supply G and the AC motor M, need to be set to be equal.

[0103] The following explanation focuses on the most representative three-phase scenario.

[0104] First, regarding capacitor C c C i The relationship between the amplitude of the voltage pulsation and the phase of the fundamental voltage on the AC side of converter 10 and inverter 20 is explained.

[0105] The above Figure 5The voltage command value shown corresponds to the fundamental voltage of one phase on the AC side. This voltage command value is compared with the carrier wave to generate PWM pulses that cause the voltage output to the AC side of converter 10 and inverter 20. As a result, the cutoff current becomes a pulse-shaped waveform that is zero during the period when the switching elements of the upper or lower bridge arms of all three phases are turned on (referred to as the zero-phase period), and equal to the current of any phase on the AC side during the other periods.

[0106] according to Figure 5 In the cutoff current, during the period when the amplitude of the voltage command value (fundamental wave) is smaller than the amplitude of the carrier wave, there are two pulses of cutoff current per cycle of the carrier wave. Here, focusing on the time interval of the cutoff current pulses, when the voltage command value is roughly equal to the amplitude (i.e., near its maximum or minimum), it is observed that adjacent pulses of the cutoff current repeatedly approach and disperse. It can be seen that this period of approaching and dispersing (period A) repeats several times within half a cycle of the voltage command value, that is, three times in this case. Between one period A and the next period A, the period (period B) in which adjacent pulses of the cutoff current are roughly equally distributed exists the same number of times.

[0107] The reasons are explained briefly below.

[0108] When the voltage command value (fundamental wave) is a three-phase balanced sine wave, the voltage command value of the other two phases is at the moment when the amplitude of one phase is the largest (in Figure 5 The amplitude (not shown in the figure) is half the amplitude of the aforementioned phase, and the reference numerals are reversed. When the voltage command values ​​of the three phases at this time point are compared with the common carrier wave, the zero-phase period becomes shorter near the time point when the carrier wave reaches its peak and longer near the time point when the carrier wave reaches its trough. That is, the adjacent pulses that cut off the current are close near the time point when the carrier wave reaches its peak, and the adjacent pulses that cut off the current are discrete near the time point when the carrier wave reaches its trough.

[0109] This phenomenon repeats each time the three-phase voltage command value (voltage fundamental wave) alternates between positive and negative values, becoming a maximum and a minimum value.

[0110] In addition, the position of the current-cutting pulse directly affects the capacitor C. c C i The magnitude of the voltage ripple. That is, if adjacent pulses that interrupt the current in converter 10 are close together, then the current flows into capacitor C at a high frequency. c Therefore, capacitor C c The voltage increases significantly. On the other hand, if the adjacent pulses of the cutoff current are discrete, then since current does not flow into capacitor C during the zero-phase period... c Therefore, due to the current flowing out to the inverter 20 side, capacitor C... cThe voltage decreases significantly. On the opposite side of inverter 20, capacitor C... i The voltage decreases significantly if the adjacent pulses of the interrupting current are close, and increases significantly if they are discrete.

[0111] The above content can be accessed through Figure 5 Confirmed. That is, although some parts overlap with what has already been explained,

[0112] • The pulses of the cut-off current repeat near and discretely occur three times within half a cycle of the three-phase voltage command value (voltage fundamental wave) during period A. Between periods A, there is a period B in which the pulses of the cut-off current are roughly equally distributed.

[0113] During period A, capacitor C c The amplitude of the voltage pulsation increases.

[0114] • Capacitor C on the 10 side of the converter c The voltage increases significantly during the period when the current-cutting pulse approaches, and decreases significantly during the period when the pulse is discrete.

[0115] As described above, in the converter 10 and the inverter 20, in capacitor C c C i Setting the voltage ripple of both capacitors to be the same during periods of large voltage ripple is effective in reducing the ripple current in the DC coupling section 30. Therefore, firstly, in order to make capacitor C c C i The period of large voltage ripple is consistent between converter 10 and inverter 20. For both converter 10 and inverter 20, the voltage command value (i.e., the frequency of the fundamental voltage wave) of any phase on the AC side is set to be the same and in phase. This is because capacitor C c C i The magnitude of the voltage ripple is directly affected by the proximity and dispersion of the cutoff current pulse, and the proximity and dispersion of the cutoff current pulse are determined by the phase of the voltage command value in carrier comparison-based PWM control.

[0116] Furthermore, in order to use capacitor C c C i During periods of large voltage ripple, the voltage ripples of both converters are set to be in phase, and the carrier frequencies used by converter 10 and inverter 20 are set to be the same but out of phase. That is, as described above, due to the proximity of the pulses relative to the cutoff current and the discrete capacitor C... c C i The voltage increases and decreases are opposite in converter 10 and inverter 20, therefore, in order to make capacitor C c C iThe voltage increases or decreases in a consistent manner, so that the pulses that interrupt the current are brought closer together and then reversed.

[0117] The diagram shown is Figure 6. Figure 6A This is the case where the carrier waves of converter 10 and inverter 20 are in phase. Figure 6B In the case of reverse phase, the AC side current, AC side voltage fundamental frequency, carrier wave, and capacitor C of converter 10 and inverter 20 are shown from above in each figure. c C i voltage E c E i and the current I of the DC coupling section 30 ci It should be noted that the scale of the chart is... Figure 6A as well as Figure 6B The common ground is in the middle.

[0118] exist Figure 6A as well as Figure 6B In the diagram, since the fundamental AC voltage is synchronized in both converter 10 and inverter 20, the waveforms overlap. Furthermore, an example is shown where the AC current of inverter 20 is in phase with the fundamental voltage, meaning the power factor is 1. These figures clearly show that the periods of larger amplitude voltage ripples in the capacitor occur simultaneously in both converter 10 and inverter 20. However, in… Figure 6A The voltage E shown is in phase with the carrier wave. c E i Becoming roughly the opposite, in contrast, in Figure 6B The voltage E is shown when the carrier wave is inverted. c E i They become roughly in phase. Therefore, in Figure 6B In the case shown, it is clearly known that... Figure 6A Compared to the situation shown, it can suppress current I. ci The pulse.

[0119] It should be noted that, by analogy to the above description, the same effect can be obtained when the fundamental voltage of any phase on the AC side of converter 10 and inverter 20 is set to the same frequency and out of phase, and the carrier waves of converter 10 and inverter 20 are set to the same frequency and in phase.

[0120] here, Figure 7A This is a waveform diagram showing the AC side current, AC side voltage fundamental, carrier, cut-off current, capacitor voltage, and DC coupling current when the power factor angle φ on the converter 10 side is set to approximately 0° (power factor is approximately 1) and the power factor angle φ on the inverter 20 side is set to -10° (current lag phase). Figure 7BThis diagram shows the waveforms of AC side current, AC side fundamental voltage, carrier wave, cutoff current, capacitor voltage, and DC-coupled current when the power factor angle φ on the converter 10 side is approximately set to 0° (power factor approximately 1) and the power factor angle φ on the inverter 20 side is set to -30° (current lag phase). It should be noted that the fundamental voltages of the converter 10 and inverter 20 are in phase, while the carrier waves are out of phase. Furthermore, in the cutoff current waveform, the solid line represents the current on the converter 10 side, and the dashed line represents the current on the inverter 20 side.

[0121] Thus, even when the current phases of converter 10 and inverter 20 are different, as described above, the approach and dispersion of the current-cutting pulse are determined solely by the phase of the fundamental voltage wave, independent of the power factor. Therefore, it can be known that capacitor C c C i The magnitude of the voltage pulsation is roughly the same in both the converter 10 and the inverter 20, and the same effect can be achieved.

[0122] (4) Implementation of the fifth method

[0123] This invention can also be applied to situations where at least one of the converter and inverter is composed of multiple power conversion units in which DC voltage sections are connected in parallel. One example of this is provided as another embodiment of the invention. Figure 8 As shown in the image.

[0124] Should Figure 8 This configuration consists of converter 10A, which connects two power conversion units 11 and 12 in parallel, and inverter 20A, which connects two power conversion units 21 and 22 in parallel. Converter 10A is connected to one AC power supply G, and inverter 20A is connected to one AC motor M. It should be noted that the switching frequencies of power conversion units 11 and 12, as well as power conversion units 21 and 22, are all the same.

[0125] In this embodiment, as in the embodiments described above, capacitor C is used. c C i The voltage ripple is controlled in such a way that the components are roughly in phase, and the power conversion units 11, 12 and 21, 22 are controlled respectively to suppress the current ripple of the DC coupling unit 30.

[0126] (5) Examples of the sixth and seventh methods

[0127] exist Figure 8 In the configuration, the capacitor C of the 10A converter c And the 20A inverter capacitor C iThe voltage fluctuations are generated by the interaction of multiple power conversion units 11 and 12 connected in parallel, as well as the interaction of power conversion units 21 and 22.

[0128] In contrast, the timing of the pulses generating the DC bus current (cutoff current) is staggered by interpolating the PWM pulses supplied to the power conversion units 11 and 12 in the converter 10A, and similarly, the timing of the pulses generating the DC bus current is staggered by interpolating the PWM pulses supplied to the power conversion units 21 and 22 in the inverter 20A. This allows for the suppression of the pulses generating the DC bus current in capacitor C. c Capacitor C i Voltage pulsation.

[0129] For example, if the PWM pulses of the power conversion sections 11 and 12 within the converter 10A are staggered in a manner that makes them approximately equal, the timing of the pulse generation of the DC bus current flowing through each power conversion section 11 and 12 can be staggered approximately equally. As a result, the timing of the pulses flowing into the capacitor C can be... c The sum of currents is smoothed out, thus effectively suppressing capacitor C. c Voltage pulsation. When there are two power converters connected in parallel, it is sufficient to set them so that the PWM pulses alternate approximately. The method for staggering the PWM pulses of the power converters is the same for power converters 21 and 22 of inverter 20A.

[0130] It should be noted that, according to this embodiment, it is undoubtedly the same as the embodiments described above, which helps to reduce the current ripple of the DC coupling section 30.

[0131] (6) Implementation of the eighth method

[0132] exist Figure 8 In the configuration, it is preferable to use capacitor C c C i The switching frequency components in the voltage pulsation are roughly synchronized in the converter 10A and the inverter 20A. In addition, the timing of the pulses generated by the DC bus currents of the multiple power conversion units 11, 12 and 21, 22 is staggered and canceled out for the higher harmonic components of the switching frequency.

[0133] If we consider the case where the converter 10A and inverter 20A are physically separated to a certain extent, then the management of the phase of these carriers will naturally have errors, and the impact will be greater as the frequency of the components increases. Therefore, the lowest frequency switching frequency component in the current ripple of the DC coupling section 30 caused by the switching operations of the converter 10A and inverter 20A is suppressed by managing the phase of the carriers of the converter 10A and inverter 20A, thereby minimizing the impact of management errors.

[0134] In contrast, since it is assumed that the multiple power conversion units 11 and 12 connected in parallel are physically close to each other (e.g., within the same housing), the phase of the carrier wave can be managed with high precision compared to that of the converter 10A and the inverter 20A. Therefore, high-order harmonic components of the switching frequency can be suppressed simply by the interaction of the multiple power conversion units connected in parallel.

[0135] Here, the cables constituting the DC coupling section 30 are considered to play a role in the distribution constant. This effect is more significant, for example, the longer the cable length or the closer the positive and negative conductors are.

[0136] The effect of the distributed constant is generally stronger at higher frequencies. Therefore, when high-frequency voltage fluctuations are applied to both ends of a cable, even if it is intended to cancel them out by managing the phase at both ends of the cable, the distributed constant of the cable can be taken into account. Specifically, high-frequency current flows through the capacitance between the positive and negative conductors. Therefore, the higher harmonic components of the switching frequency at both ends of the cable are canceled out by the interaction of multiple power conversion units 11, 12, 21, and 22, and the relatively lower frequency switching frequency components are suppressed by managing the phase at both ends of the cable. Thus, it is difficult to be affected by the distributed constant of the cable, and current ripples can be effectively suppressed.

[0137] (7) Implementation of the Ninth Method

[0138] Contrary to the eighth embodiment described above, for the capacitor C of converter 10A and inverter 20A... c C i The switching frequency component in the voltage pulsation can be made to stagger and cancel out the timing of the pulses generated by the DC bus currents of the multiple power conversion units 11, 12 and 21, 22. On the other hand, the higher harmonic components of the switching frequency can be set to be approximately in phase in the converter 10A and the inverter 20A.

[0139] That is, for the switching frequency component, which typically has the highest proportion of current ripples generated in the DC coupling section 30 of cables, etc., the ripple component is reduced at its source by staggering the timing of the pulse generation of the DC bus current of power conversion sections 11 and 12 and the timing of the pulse generation of the DC bus current of power conversion sections 21 and 22 at both ends of the DC coupling section 30. The two ends of the DC coupling section 30 refer to, for example, the interiors of converter 10A and inverter 20A respectively. Furthermore, for the residual high-order harmonic components of the switching frequency in the current ripples generated in the DC coupling section 30, the interaction between converter 10A and inverter 20A, that is, by reducing the timing of the pulse generation of the DC bus current of power conversion sections 21 and 22, the ripple component is reduced at its source. c Ci The voltage ripples are set to be in phase and thus cancel each other out.

[0140] (8) Implementation of the tenth method

[0141] This embodiment further specifies the case where the converter 10A and the inverter 20A are composed of multiple power conversion units connected in parallel.

[0142] Similar to the case of a single power conversion unit, when generating PWM pulses by comparing voltage command values ​​and carrier waves, for multiple power conversion units connected in parallel, the carrier waves for each power conversion unit are set to the same frequency and a predetermined phase difference is maintained between each carrier wave. This allows for easy timing-staggered generation of pulses for the cut-off current output from each power conversion unit. This is achieved by using... Figure 5 The explanations and other details are easy to understand.

[0143] As an example, when there are two power converters connected in parallel, it is possible to reverse the carrier waves supplied to each power converter. Alternatively, when there are three power converters, the phase difference of the carrier waves can be set to 120° and evenly staggered. Since the timing of the pulses generating the interrupting current from the multiple power converters is evenly staggered in either case, it is possible to make capacitor C... c C i The voltage ripple decreases, which in turn reduces the current ripple in the DC coupling section 30. Alternatively, shifting the phase of the carrier wave by canceling out the high-order harmonics of the switching frequency is also effective.

[0144] Furthermore, similar to the case where there is an odd number of power conversion units, when there are multiple power conversion units, by setting the carrier frequencies of converter 10A and inverter 20A to be the same, and adjusting the phase relationship between the carrier on the converter 10A side and the carrier on the inverter 20A side, it is possible to reduce the capacitor C c C i The main components of the voltage ripple are set to be approximately in phase to suppress the current ripple of the DC coupling section 30.

[0145] (9) Implementation of the eleventh method

[0146] like Figure 8 As shown, when the converter 10A and inverter 20A have multiple power conversion units connected in parallel, if the fundamental frequency of the AC side voltage of the converter 10A and inverter 20A is set to be the same, then capacitor C can be made to... c C iThe timing of the voltage pulsation amplitude variation is consistent. Furthermore, by setting the fundamental amplitudes of the AC side voltage and AC side current of the multiple power conversion units connected in parallel to be approximately equal, the cut-off current generated by each power conversion unit also becomes a similar waveform. By setting a phase difference in these cut-off currents, the pulsating current can be effectively canceled out.

[0147] Figure 9A as well as Figure 9B This is a waveform diagram showing the simulation results of this embodiment.

[0148] Figure 9A For comparison, waveforms of AC side current, AC side fundamental voltage, carrier voltage, cutoff current, capacitor voltage, and DC coupling current are shown when both converter 10 and inverter 20 have a single power conversion unit. Figure 9A In this process, the carrier waves on the converter 10 side and the inverter 20 side become out of phase, equivalent to the above. Figure 6B Examples.

[0149] on the other hand, Figure 9B Is it like this? Figure 8 The waveform diagrams of the various parts of the converter 10A and inverter 20A, which are respectively connected in parallel, are shown. Figure 9B This illustrates the case where the fundamental AC voltages of converter 10A and inverter 20A are synchronized, and for each group of power conversion units 11, 21, 12, and 22, the amplitudes of the fundamental AC voltages and the AC side current are set to be approximately equal. Furthermore, Figure 9B This diagram illustrates a scenario where the carrier frequencies of the four power conversion units 11, 12, 21, and 22 are set to the same value, and their respective phases are sequentially set to 0°, 90°, 180°, and 270°. Specifically, in converter 10A, the phases of the carriers relative to power conversion units 11 and 12 are offset by 90°, and in inverter 20A, the phases of the carriers relative to power conversion units 21 and 22 are offset by 90°. Furthermore, the phases of the carriers between converter 10A and inverter 20A are offset by 180° as a whole. Figure 9B The diagram only shows the carrier waves for the power conversion unit 11 in the converter 10A and the power conversion unit 21 in the inverter 20A (with a phase difference of 180° between them), omitting the illustrations of the carrier waves for the other power conversion units 12 and 22.

[0150] In addition, for Figure 9B The cutoff current is shown in the diagram. The solid line indicates the power conversion section 11 side within the converter 10A, and the dashed line indicates the power conversion section 21 side within the inverter 20A. For capacitor C... c C iThe voltage of capacitor C is shown by the solid line. c voltage E c The dashed line shows the capacitor C. i voltage E i .

[0151] It should be noted that, in Figure 9A as well as Figure 9B In the middle, the proportions of the corresponding waveforms are the same.

[0152] According to this embodiment ( Figure 9B Between converter 10A and inverter 20A, the main component of the current ripple in the DC coupling section 30, namely the frequency component of the carrier wave, becomes in phase and cancels each other out. Simultaneously, between power conversion sections 11 and 12 in converter 10A and between power conversion sections 21 and 22 in inverter 20A, the twice-frequency component of the carrier wave becomes in phase and cancels each other out. As a result, it can be seen that the current ripple in the DC coupling section 30 and... Figure 9A Compared to further reductions.

[0153] and, Figure 10 and Figure 9B Conversely, it is a waveform diagram showing the configuration in which the carrier frequency component is canceled between the converter 10A and the inverter 20A, and between the power conversion units 11 and 12 in the converter 10A and between the power conversion units 21 and 22 in the inverter 20A.

[0154] In this example, the carrier frequencies of the four power conversion units 11, 12, 21, and 22 are set to the same frequency, and their phases are set sequentially to 0°, 180°, 90°, and 270°. Figure 10 The diagram only shows the carrier waves for the power conversion section 11 within the converter 10A and the power conversion section 21 within the inverter 20A (with a 90° phase difference between them); illustrations of the carrier waves for the other power conversion sections 12 and 22 are omitted. Furthermore, for the interrupted current, solid lines show the power conversion section 11 side within the converter 10A, and dashed lines show the power conversion section 21 side within the inverter 20A. For capacitor C... c C i The voltage of capacitor C is shown by the solid line. c voltage E c The dashed line shows the capacitor C. i voltage E i .

[0155] In Figure 10 In, and Figure 9B Similarly, it can be seen that the current ripple of the DC coupling section 30 is significantly reduced.

[0156] It should be noted that if... Figure 9B and Figure 10 By comparison, it can be seen that capacitor C c C i The frequency components of the voltage ripples are different. This is because, in order to address the issue of capacitor C on the 10A side of the converter... c and capacitor C on the 20A side of the inverter i The respective voltage ripples are generally set to be in phase to cancel each other out, thus reducing the current ripple of the DC coupling section 30. This is the objective of the present invention. The frequency components of the capacitor voltages to be canceled out are different (in...). Figure 9B The frequency component of the carrier wave is in the middle; in contrast, in... Figure 10 (The middle part is its twice frequency component).

[0157] (10) Embodiment of the twelfth method

[0158] When multiple power conversion units are connected in parallel to form the converter 10A and the inverter 20A, even if the system stops due to a single power conversion unit failure, the remaining power conversion units can continue to operate the system.

[0159] In this case, if the remaining power conversion unit operates in the same way as all the power conversion units, the balance that suppresses the current ripple of the DC coupling unit 30 through the interaction of each power conversion unit is disrupted, resulting in a problem of increased current ripple.

[0160] Therefore, the key point of this embodiment is to alleviate the above-mentioned problems by correcting the operating state of the remaining power conversion unit.

[0161] For example, such as Figure 8 As shown, in the configuration of converter 10A and inverter 20A having two power conversion units 11 and 12 and power conversion units 21 and 22 connected in parallel, when one power conversion unit on the converter 10A side stops, the following correction of the operating state can be considered.

[0162] (a) The operation of one of the two power conversion units 21 and 22 on the inverter 20A side is stopped, and the state is set so that both the converter 10A and the inverter 20A are operated by one power conversion unit. The current pulsation of the DC coupling unit 30 is suppressed by the operation described above.

[0163] (b) Make the two power conversion units 21 and 22 on the inverter 20A side perform the same operation as one power conversion unit, that is, make the AC side voltage, AC side current and carrier common to suppress the current pulsation of the DC coupling unit 30.

[0164] (11) Examples of methods thirteen through fifteen

[0165] This embodiment relates to a control circuit for a power conversion system, and pertains to the above-mentioned... Figure 1B The structure and function of the control circuit. The following, although for control... Figure 1A The situation of converter 10 and inverter 20 will be explained, but in the control Figure 8 In the case of a converter 10A and an inverter 20A connected in parallel as shown, the same control circuit can be applied.

[0166] It should be noted that, as described below, the configuration of the control circuit of the present invention is not limited to any particular type. Figure 1B Examples.

[0167] exist Figure 1B In the inverter 20, the power obtained by multiplying the torque of the AC motor M on the load (not shown) by its angular velocity, plus losses, is the power that should be sent to the DC coupling unit 30 via the converter 10. To establish this relationship, the capacitor C, including the converter 10, is... c The DC component of the DC voltage section needs to be kept constant. Therefore, the control system of the converter 10 performs feedback control in such a way that the DC voltage of the aforementioned DC voltage section becomes a predetermined value, and its operating quantity becomes the power generation P of the AC power source G. Furthermore, this power generation P is controlled by causing a current to flow synchronously with the voltage of the AC power source G.

[0168] On the other hand, the control system of inverter 20 can operate the current of AC motor M while controlling the torque of AC motor M, and can operate the torque by performing speed feedback control while controlling the speed of AC motor M.

[0169] exist Figure 1B In this circuit, DC voltage control with a so-called current-controlled local loop is performed on the converter 10 side, and (rotational) speed control with a so-called current-controlled local loop is performed on the inverter 20 side. The methods for implementing these are well-known and can include direct control of AC quantities, DC control based on rotating coordinate transformation, vector control, sensorless vector control, etc. Further explanation is omitted here.

[0170] As explained through various embodiments, in order to reduce current ripple in the DC coupling section 30, it is effective to set the fundamental frequencies of the AC power supply G and the AC motor M to be the same. Therefore, in Figure 1B In the control circuit, the frequency of the current of the AC power supply G is set to an amount equivalent to the frequency command value of the AC motor M and given to the inverter 20.

[0171] In other words, since current control of the AC power supply G is performed on the converter 10 side, information about the frequency of the AC power supply G's current is available. On the other hand, speed control of the AC motor M is performed on the inverter 20 side, and the command value for this speed control is directly related to the frequency of the AC motor M's current. This relationship is determined by the type of AC motor M (synchronous motor, asynchronous motor, etc.), the number of poles, etc. Therefore, the frequency information of the current on the converter 10 side is sent to the inverter 20 side in a way that the frequency of the AC motor M's current matches the frequency of the AC power supply G's current—that is, in a synchronized manner—and the inverter 20 side generates, for example, a speed command value based on this information.

[0172] Figure 1B The specific control method is as follows. Here, both the AC power supply G and the AC motor M are set to be synchronous motors.

[0173] That is, in the case of a synchronous motor, the frequency of the current and the rotational frequency of the motor are consistent with the number of pole pairs (an integer) of the motor. Therefore, in both the converter 10 and the inverter 20, the current frequency and the rotational frequency of the motor are consistent with the number of pole pairs (an integer) of the motor. g p e Will be respectively via position sensor SENS g SENS m The phase angle θ of the synchronous motor being tested g θ m Converted to electrical angle θ ge θ me Using these θ ge θ me The rotational coordinate transformation unit VR and the inverse transformation unit VRI in the control system are respectively assigned. It should be noted that, in the so-called sensorless control scenario, θ is... ge θ me The estimated value is calculated using the voltage and current information provided by the control system.

[0174] Next, the local current control loops on the converter 10 side and the inverter 20 side will be described. For both the converter 10 side and the inverter 20 side, the AC power supply G and the AC current I of the AC motor M are controlled through the rotating coordinate transformation unit VR. g I m Converted into DC current I gd I gq and I md I mq The difference between the current command value and the current input is input to the current regulating unit ACR to generate a DC voltage command value. This DC voltage command value is then converted back to an AC voltage command V by the inverting unit VRI. c V iThrough comparator COMP c COMP i A PWM pulse (command value) is generated by comparing it with the carrier wave and is given to the power conversion units 11 and 21 respectively. Here, although the carrier wave frequencies are the same in the converter 10 and the inverter 20, the carrier waves of the two can be made to have a specified phase difference as needed by the phase shifting unit F.

[0175] First, the DC voltage control on the 10 side of the converter will be explained.

[0176] The voltage E of the DC voltage section of the converter 10 c The detection is performed to generate the voltage E passing through the low-pass filter LPF. c-lpf The structure is used to make the voltage E c-lpf With target value E cref A consistent feedback control system is used, where the difference between the two is input to the voltage regulation unit (AVR) to obtain a power command value. The power command value conversion unit "P→I" generates a current command value for outputting an equivalent amount of generated electricity to the AC power source G, and provides it to the current control local loop on the converter 10 side.

[0177] Next, the speed control on the inverter 20 side will be explained.

[0178] With the rotational frequency ω of the AC motor M m Multiply by the extreme logarithm p e The electrical angular frequency ω me The target value ω of the rotation frequency output by the phase synchronization unit 43 described later. meref Feedback control is performed in a consistent manner. Electrical angular frequency ω me With the target value ω meref The difference is input to the speed regulation unit ASR to obtain the torque command value. The torque command value conversion unit "T→I" generates a current command value to make the AC motor M produce a torque corresponding to the torque command value, and provides it to the current control local loop on the inverter 20 side.

[0179] Here, the method described above for making the frequencies of the AC power supply G and the current of the AC motor M the same will be explained.

[0180] exist Figure 1B In this process, not only is the frequency consistent, but the phase angle of the current can also be controlled. This control is achieved by the phase synchronization unit 43. In this phase synchronization unit 43, in order to synchronize the phase angle θ of the AC power supply G's current... ge The current phase angle θ of the AC motor M is used as a reference. me Synchronization, and feedback control. That is, θ ge With θ meThe difference plus the specified offset angle θ adf The obtained value is input into the phase adjustment unit PLL, and its output is used as the target value ω of the rotational frequency of the AC motor M. meref In other words, it is constituted by θ ge and θ me Synchronization is achieved by adjusting ω, which is proportional to the rotational speed of the AC motor M. meref Adjustments will be made.

[0181] The operation of the aforementioned control circuit enables the fundamental frequency of the AC power supply G and the AC motor M to be synchronized.

[0182] It should be noted that, conversely, the frequency of the current of the AC motor M can be determined according to the load condition of the AC motor M, and the frequency of the current of the AC power supply G can be made consistent with that frequency. Specifically, the rotational speed of the rotating power that serves as the driving source of the AC power supply G can be adjusted in a manner consistent with the frequency of the current of the AC motor M.

[0183] exist Figure 11 The block diagram of the control circuit for this situation is shown below. Figure 11 In the middle, the phase synchronization unit 43A is used to synchronize the electrical phase angle θ of the AC motor M. me The electric phase angle θ of the AC power supply G is used as a reference. ge Synchronize, θ me With θ ge The difference plus the specified offset angle θ adf The obtained value is input into the phase adjustment unit PLL, and its output is used as the target value ω of the electrical angular frequency of the AC power supply G. geref That is, it is constituted with θ me and θ ge The synchronization method is proportional to the rotational speed ω of the AC power supply G. geref Adjustments are made. Since the AC power supply G is driven by an external force (not shown), such as an engine, ω... geref The control system is subjected to an external force, which is not illustrated. Using this control circuit, it is also possible to synchronize the fundamental frequency of the AC power supply G and the AC motor M. It should be noted that in this control system, the target value ω of the electrical angular frequency of the AC motor M is... meref It depends on the load.

[0184] Next, a method for stabilizing the power supply from the AC power source G to the AC motor M and maintaining a high degree of synchronization between the fundamental frequency of the AC power source G and the AC motor M will be explained. To achieve this goal, it is effective for the AC power source G to supply the power required by the AC motor M in a timely and undelayed manner. This function is implemented by the motor power calculation unit 41 and the power feedforward unit (power FF unit) 42.

[0185] That is, due to the power P of the AC motor M m It is the product of the output torque and the mechanical angular frequency. The command values ​​of both are known in the control circuit. Therefore, using this information, the motor power calculation unit 41 can control the power P of the AC motor M. m The calculation is performed. The power feedforward unit 42 obtains the power P... m The power command value is added to the converter 10, thereby enabling the AC motor M to generate the required power from the AC power source G without delay. Strictly speaking, the AC power source G needs to include the losses of the converter 10, inverter 20, and AC motor M to generate power, but since these are generally small relative to the amount of power generated, the impact on the control circuit is also small, and the feedback control system compensates for the differences in these losses, so there is no problem.

[0186] Without the aforementioned AC motor M's power P m In the case of feedforward, if, for example, the power of the AC motor M increases rapidly, the inverter 20 wants to obtain more power from the converter 10 than the current power generation, resulting in a decrease in the DC voltage section. As a result, the power generation increases through the operation of the DC voltage control system of the converter 10. In other words, it becomes an operation that takes the change in the DC voltage section as a prerequisite.

[0187] Therefore, by conducting power P m The feedforward control minimizes voltage fluctuations in the DC voltage section, thus ensuring a stable power supply and stable synchronization of the fundamental frequency of the AC power supply G and the AC motor M.

[0188] (12) Embodiment of the Sixteenth Method

[0189] As described in the various embodiments, the current pulsation of the DC coupling section 30, such as the cable, varies according to the voltage fundamental, current fundamental, and carrier state changes of the AC side of the converter 10 and inverter 20. Basically, it is effective to keep the frequencies of these AC side voltage fundamental, current fundamental, and carrier consistent on both the converter 10 and inverter 20 sides. Furthermore, since the current pulsation of the DC coupling section 30 can be reduced by adjusting the phase of the carrier, a control system that automatically adjusts these can be configured. In particular, the time constant for the automatic adjustment is set to be larger than the response time constant when driving the AC motor M, for example, approximately five times or more, so as not to interfere with the system's original purpose of driving the AC motor M. This allows for stable system operation and reduces the computational load on the control circuit.

[0190] (13) Embodiment of the Seventeenth Method

[0191] The power conversion system of the present invention has a wide range of applications; as one example, it will be used in various fields. Figure 12 The invention is illustrated as an example of its application in a known aircraft propulsion system.

[0192] exist Figure 12 In this configuration, EN1 and EN2 are the aircraft's jet engines, and generators G1 and G2 are connected to them. Generators G1 and G2 are connected to converters CON1 and CON2, and further connected via converters CON1a and CON2a for charging and discharging batteries BAT1 and BAT2, and inverters INV1 and INV2 for driving motors M1 and M2. The aforementioned motors M1 and M2 constitute the electric fans that propel the aircraft. It should be noted that when batteries BAT1 and BAT2 are not used, converters CON1a and CON2a are not required, and converters CON1 and CON2 are directly connected to inverters INV1 and INV2 via cables serving as DC coupling sections.

[0193] The basic components of this aircraft propulsion system can be considered to include two... Figure 1A as well as Figure 1B The power conversion system of the present invention is shown.

[0194] Furthermore, since the propulsion motors for aircraft require large outputs ranging from hundreds of kW to several MW, improving system efficiency and reducing heat generation are important. Additionally, due to the extreme importance of lightweight design, it is necessary to minimize the electrostatic capacitance of the capacitors used for voltage smoothing in the converter and inverter, thereby preventing large current ripples in the cables flowing between the converter and inverter.

[0195] Therefore, by applying the present invention, by using small-capacity capacitors in the DC voltage section of the converter and inverter while suppressing current pulsation in the cable, it is possible to reduce heat generation and improve efficiency.

[0196] It should be noted that superconducting cables can also be considered as DC coupling components. In this case, the superconductor suffers losses due to the flow of high-frequency current, which hinders the maintenance of the superconducting state. Therefore, the application of the present invention is particularly effective.

[0197] The above mainly describes a three-phase power conversion system, but the present invention is not limited thereto and can also be applied to power conversion systems with other phases.

[0198] This international application claims priority to Japanese Patent Application No. 2021-023112, filed on February 17, 2021, and the entire contents of Japanese Patent Application No. 2021-023112 are incorporated herein by reference.

[0199] Explanation of reference numerals in the attached figures

[0200] 10: Converter

[0201] 11, 12: Power Conversion Department

[0202] 20: Inverter

[0203] 21, 22: Power Conversion Department

[0204] 30: DC coupling section

[0205] 41: Electric motor power calculation unit

[0206] 42: Power feedforward unit

[0207] 43, 43A: Phase synchronization unit

[0208] S1~S12, S21~S32: Semiconductor switching elements

[0209] Cc, Ci: Capacitors

[0210] SENSg, SENSm: Position sensor

Claims

1. A power conversion system, comprising: AC power supply; The converter converts the AC power from the aforementioned AC power source into DC power through PWM control; The inverter converts the DC power output from the converter into AC power and supplies it to the AC motor through PWM control. The first capacitor is connected to the DC voltage section of the aforementioned converter; The second capacitor is connected to the DC voltage section of the inverter described above; as well as A DC coupling section connects the first capacitor and the second capacitor, and has an inductive component. The switching frequencies of the converter and the inverter are set to be equal, and the switching frequency is set to be higher than the resonant frequency of the resonant circuit including the first capacitor, the second capacitor and the DC coupling section. By adjusting at least one of the phase relationship between the carrier waves used for PWM control of the converter and the inverter, and the phase relationship between the fundamental AC voltage waves of the converter and the inverter, or the phase relationship between the fundamental AC current waves of the converter and the inverter, the phase of a predetermined component of the voltage ripple of the first capacitor and the second capacitor generated by the switching operation of the converter and the inverter becomes approximately in phase, the switching operation of at least one of the converter and the inverter is controlled.

2. The power conversion system according to claim 1, wherein, PWM pulses are generated by comparing the voltage command value and the carrier wave, respectively, and the carrier wave on the converter side and the carrier wave on the inverter side are set to the same frequency, and the two carrier waves have a specified phase relationship.

3. The power conversion system according to claim 2, wherein, The number of phases of the aforementioned AC power supply and the aforementioned AC motor are set to be equal. At least one of the converter and the inverter is controlled such that the fundamental frequencies of the voltages of one phase of the AC power supply and the AC motor are the same and approximately in phase. Furthermore, the phases of the carrier waves on the converter side and the inverter side are reversed.

4. The power conversion system according to claim 2, wherein, The number of phases of the aforementioned AC power supply and the aforementioned AC motor are set to be equal. At least one of the converter and the inverter is controlled such that the fundamental frequencies of the voltages of one phase of the AC power supply and the AC motor are the same and approximately out of phase. Furthermore, the phases of the carrier waves on both the converter side and the inverter side are set to be in phase.

5. The power conversion system according to claim 1, wherein, At least one of the aforementioned converter and inverter has multiple power conversion sections connected in parallel to the DC voltage section.

6. The power conversion system according to claim 5, wherein, The switching frequencies of the aforementioned multiple power conversion units are set to be the same, and the timing of the pulse generation of the DC bus current of the aforementioned multiple power conversion units constituting the aforementioned converter or the aforementioned inverter is staggered.

7. The power conversion system according to claim 6, wherein, This ensures that the timing of the pulse generation of the DC bus current in the aforementioned multiple power conversion units is approximately evenly distributed.

8. The power conversion system according to claim 6, wherein, By setting the switching frequency component in the voltage pulsation of the first capacitor or the second capacitor to be approximately in phase in the converter and the inverter, the timing of the pulse generation of the DC bus current of the plurality of power conversion units is staggered for the higher harmonic components of the switching frequency, thereby canceling them out.

9. The power conversion system according to claim 6, wherein, By adjusting the switching frequency component in the voltage pulsation of the first capacitor or the second capacitor, the timing of the pulse generation of the DC bus current of the plurality of power conversion units is staggered, and the higher harmonic components of the switching frequency are set to be approximately in phase in the converter and the inverter, thereby canceling them out.

10. The power conversion system according to claim 6, wherein, PWM pulses are generated by comparing voltage command values ​​and carrier waves, and these carrier waves are set to the same frequency and have a specified phase relationship. Furthermore, the carriers used in the aforementioned converter and inverter will be set to the same frequency and have a specified phase relationship between them.

11. The power conversion system according to claim 10, wherein, The number of phases of the AC power supply and the AC motor are set to be equal, and the fundamental frequency of the AC side voltage is set to be the same. In the aforementioned converter and inverter, the fundamental amplitudes of the AC side voltage and AC side current of the aforementioned plurality of power conversion units are set to be approximately equal.

12. The power conversion system according to claim 5, wherein, When a portion of the aforementioned multiple power conversion units is stopped, control is performed such that the phase of a predetermined component of the voltage pulsation caused by the switch in the first or second capacitor is substantially in phase in the converter and the inverter.

13. The power conversion system according to claim 1, wherein, The aforementioned AC power source is an AC generator driven by external force. The current of the AC generator is controlled by the converter in such a way that the average DC voltage of the converter or inverter is a predetermined value, and the frequency of the AC generator current is given to the inverter as a quantity equivalent to the frequency command of the AC motor.

14. The power conversion system according to claim 1, wherein, The aforementioned AC power source is an AC generator driven by external force. A control device that controls the current of the AC generator by means of the average DC voltage of the converter or inverter being a predetermined value, and applies the frequency of the AC motor current as a quantity equivalent to the frequency command of the AC generator to the external force.

15. The power conversion system according to claim 13, wherein, The amount equivalent to the output power of the inverter is added to the amount equivalent to the input power command value of the converter.

16. The power conversion system according to claim 14, wherein, The amount equivalent to the output power of the inverter is added to the amount equivalent to the input power command value of the converter.

17. The power conversion system according to claim 1, wherein, By adjusting at least one of the phase relationship between the carrier waves used for PWM control of the converter and the inverter, and the phase relationship between the fundamental AC voltage waves of the converter and the inverter, or the phase relationship between the fundamental AC current waves of the converter and the inverter, the current ripple flowing through the DC coupling section is reduced.

18. A power conversion system, wherein, The power conversion system of claim 1 is a system for driving the aforementioned AC motor of the electric fan of a propulsion aircraft.

Images