Topology optimization and frequency matching method of contactless power supply system

Abstract

The present invention proposes a method for topology optimization and frequency matching of a contactless power supply system, comprising the following steps: constructing a topology-optimized contactless power supply circuit: adding an auxiliary bridge arm to a single-phase full-bridge inverter to form a three-phase inverter, with a primary coil coupled to a secondary coil; calculating the impedance of the composite resonant network based on Kirchhoff's law and transformer principles; using the composite resonant network impedance to determine the composite resonant network frequency; treating the primary circuit as a bandpass filter to determine the natural resonant frequency of the primary circuit; treating the secondary circuit as a low-pass filter to determine the natural resonant frequency of the secondary circuit; and limiting each frequency to place the switches of the three-phase inverter in a soft-switching state. The present invention utilizes a frequency-boosting method to improve light-load efficiency by approximately 3-5% during the light-load phase, and simplifies the circuit structure to improve light-load efficiency by approximately 3%. The present invention exhibits robust natural soft-switching capabilities within the switching frequency range and during the frequency conversion process.

Description

Technical Field

[0001] The present invention relates to the technical field of contactless power supply and automatic control, and in particular to a method for topological structure optimization and frequency matching of a contactless power supply system. Background Art

[0002] Contactless power supply systems are used for inductive wireless charging of robots and electric vehicles in underground coal mines, as well as in mining applications where plug-in wiring is unsuitable. These systems offer significant advantages in terms of explosion-proof performance and intelligent control. However, the disadvantages of plug-in wiring include unsuitability for explosion-proof environments and reliance on manual wiring.

[0003] Single-phase square-wave inverters are commonly used as the primary inverter in contactless power supply circuits. These inverters typically undergo both longitudinal and transverse commutation, making it easy for the switches to enter a hard-switching chopping state. Because the resonant network of a contactless power supply circuit exhibits frequency-selective characteristics, selecting a frequency with high efficiency under heavy loads results in high circulating current under light loads, significantly lowering efficiency under heavy loads. For circuits with power levels of several kilowatts or even higher, hard-switching the inverter's switches at light loads can cause rapid heating and potentially damage the device.

[0004] Domestic and international researchers have conducted extensive research on single-phase inductive power transmission systems using LCC compensation circuits. However, their efficiency is still lower under light loads than under heavy loads, as the inverter switches typically operate in a hard-switching state. While frequency conversion can improve the light-load efficiency of inductive power transmission systems using LCC compensation circuits, achieving soft switching during transients and light-load conditions, while maintaining ideal efficiency, requires a more scientific approach. Summary of the Invention

[0005] In response to the technical problems that the switching tubes of existing single-phase square wave inverters are prone to hard switching states, and that three-phase inverters are prone to hard switching states when lightly loaded, which may lead to device damage, the present invention proposes a method for optimizing the topology of a contactless power supply system and matching the frequency. The method optimizes the topology of the contactless power supply system and provides a frequency matching method, thereby adjusting the transmission power when the load changes to improve light load efficiency and transmission efficiency.

[0006] In order to achieve the above object, the technical solution of the present invention is implemented as follows: a method for topology optimization and frequency matching of a contactless power supply system, the steps of which are as follows:

[0007] Step 1: Construct a contactless power supply circuit with optimized topology: Add an auxiliary bridge arm to a single-phase full-bridge inverter to form a three-phase inverter. The three-phase inverter is connected to the three-phase primary coils through a three-phase compensation circuit. One primary coil is coupled to the secondary coil, and the secondary coil is connected to the load through a single-phase rectifier circuit.

[0008] Step 2: Equivalently treat the single-phase rectifier circuit as a current source, and the three-phase inverter and three-phase compensation circuit as a half-bridge circuit. Calculate the composite resonant network impedance Z of the primary and secondary coils and the three-phase compensation circuit based on Kirchhoff's law and transformer principle. in ;

[0009] Step 3: The impedance Z of the composite resonant network in Find the imaginary part and set it equal to 0 to solve the frequency f of the composite resonant network T Without considering electromagnetic coupling, the primary resonant circuit composed of the three-phase primary coil and the three-phase compensation circuit is regarded as a bandpass filter, and the inherent impedance Z of the primary circuit is solved. p , according to the primary side inherent impedance Z p , solve the natural resonant frequency f of the primary circuit p ; Consider the secondary resonant circuit composed of the secondary coil and the compensation capacitor as a low-pass filter, and solve the natural resonant frequency f of the secondary circuit s ;

[0010] Step 4: To keep the switch tube of the three-phase inverter in the soft switching state:

[0011] (1) The chopping frequency f of the switching tube v and the natural resonant frequency f of the secondary circuit s The relationship should satisfy the following conditions: vmax ≤f s ≤2·f vmin ;f vmin and f vmax Represent the minimum switching frequency and the maximum switching frequency respectively;

[0012] (2) Natural resonant frequency f p The two positive real roots f p1 and f p2 , let f p1 <f p2 , calculate the parameters of the inductor and capacitor components, and select the components to make the switching frequency f v Satisfy the conditions: f p1 ≤f vmax ≤f s ≤2·f vimn ;

[0013] (3) Select components to make the composite resonant network frequency f T The main resonant frequency f T1 and chopping frequency f v Satisfies the relationship: f T1 ≤f vmin .

[0014] Preferably, the three-phase inverter is a three-phase square wave inverter, which converts direct current into high-frequency alternating current, which is transmitted to the single-phase rectifier circuit by the primary coil, the three-phase compensation circuit, and the secondary coil; the auxiliary bridge arm is the bridge arm where the switch tube S1 and the switch tube S2 are located, and the single-phase full-bridge inverter includes switch tubes S3-S6, the switch tubes S3-S4 form a bridge arm, the switch tubes S5-S6 form a bridge arm, the three bridge arms are connected in parallel, and diodes are connected in anti-parallel on the switch tubes S3-S6; during each working mode conversion, a corresponding power supply circuit and freewheeling circuit are generated, the power supply circuit supplies power to the freewheeling circuit, and the freewheeling circuit makes the anti-parallel diode of the switch tube to be switched always conductive during the working mode conversion process; a compensation capacitor is connected in parallel on the secondary coil.

[0015] Preferably, the composite resonant network impedance Z in The calculation method is: the current i of the single-phase rectifier circuit s32 The average value of the current I s32 Indicates that the current i s32 The positive half cycle flows through diode D2 and output load R O , capacitor C5, current i s32 The negative half cycle flows through capacitor C4 and output load R O , diode D5, so the output current I DC =I s32 / 2; Ignoring the energy loss of the single-phase rectifier circuit, according to the law of conservation of power: I s32 2 ·R eq2 =I DC 2 ·R O , then the equivalent resistance of the secondary circuit is R eq2 for: Among them, R O is the output load;

[0016] Use Kirchhoff's current and voltage laws to get the impedance Z of the secondary circuit r32 (jω T ):

[0017]

[0018] Among them, r s32 is the inductance L s32 The internal resistance, ω T is the switching frequency of the switches S1 to S6;

[0019] Using Kirchhoff's current and voltage laws, we can get the equivalent impedance Z from the primary coil to the load of the three-phase inverter. 32 (jω T )for:

[0020]

[0021] Among them, r p32 is the inductance L of the primary coil p32 The internal resistance; M is the inductance L p32 and secondary coil L s32 The mutual inductance is proportional to the coupling coefficient k, and Represents capacitance C s32 The voltage across the terminals, Indicates the current flowing through the inductor L p32 The inductance and capacitance C at both ends s32 The secondary coil L s32 Parallel filter capacitors.

[0022] Equivalent impedance Z 32 (jω T ) and capacitor C s32 Parallel triangle impedance Z Δ32 (jω T ) indicates that the triangle impedance Z Δ32 (jω T )for:

[0023]

[0024] The three-phase primary coil and the three-phase compensation circuit are regarded as the triangle load of the three-phase inverter. The triangle load is converted into a star load. The center point o of the star load is regarded as the output zero line. The three-phase inverter is equivalent to the topology of three independent half-bridges. According to the circuit theory, the star load Z 12 (jω T )for:

[0025]

[0026] According to the impedance calculation method in circuit theory, the impedance Z of the composite resonant network including the primary and secondary coils and the three-phase compensation circuit is obtained. in for:

[0027]

[0028] Among them, the capacitor C p32 is the capacitance and inductance L in the three-phase compensation circuit p32 For the coupled primary coil, capacitor C s32 The secondary coil L s32 The parallel capacitor, r s32 is the inductance L s32 internal resistance.

[0029] Preferably, the method for solving the composite resonant network frequency f T The method is: the composite resonant network impedance Zin Calculate the imaginary part Im(Z in ) can be obtained:

[0030]

[0031] Let the imaginary part Im(Z in )=0Ω, use the calculation method to find the angular frequency ω T , according to ω T =2πf T , derive the composite resonant network frequency f T The expression of , discarding the solution containing imaginary numbers, obtains 2 real solutions:

[0032]

[0033] Among them, f T1 is the composite resonant network frequency f T The main resonant frequency of T2 is the composite resonant network frequency f T High resonant frequency.

[0034] Preferably, the natural resonant frequency f of the primary circuit is obtained p The method is: regard the primary resonant circuit as a bandpass filter and obtain the primary inherent impedance Z p for:

[0035]

[0036] Among them, ω p Represents the natural resonant angular frequency of the primary coil and its compensation circuit, ω p =2πf p ;

[0037] Calculate the primary side inherent impedance Z p The imaginary part of :

[0038]

[0039] Let the imaginary part Im(Z p ) is zero, and the natural resonant frequency f is derived p Solution:

[0040]

[0041] Among them, f p1 and f p2 is the natural resonant frequency f p two frequencies.

[0042] Preferably, the natural resonant frequency f of the secondary circuit is obtained sThe method is: consider the secondary resonant circuit as a low-pass filter, the inherent impedance Z of the secondary coil and the single-phase rectifier circuit s for:

[0043]

[0044] Let the inherent impedance Z s The imaginary part of is 0, and the natural resonant frequency of the secondary circuit is obtained:

[0045]

[0046] Among them, ω s Represents the natural resonant angular frequency of the secondary coil and its compensation circuit, ω s =2πf s .

[0047] Preferably, the main resonant frequency f T1 It should be slightly lower than the chopping frequency f v , and must be within the passband of the filter; the natural resonant frequency f p The amplification factor is bandpass; the natural resonant frequency f s The amplification factor is low-pass; the main resonant frequency f T1 and the natural resonant frequency f p1 Closer.

[0048] Preferably, a circuit diagram is drawn and simulated using computer software to verify the composite resonant network frequency f T and chopping frequency f v The effective regulation area, load power P(R O ) corresponds to the relationship between the chopping frequency f v ≈85kHz, when the chopping frequency f v Reduce, load power P(R O )rise.

[0049] Preferably, when the duty cycle of all switches of the three-phase inverter is 40-42%, the chopping frequency f of the switch is v It is adjusted between 80 and 120kHz, and the load resistance R O Under the condition of 50Ω~600Ω, the switch tube works in the soft switching state and has the strongest robustness.

[0050] Preferably, the load power W(R O )With the chopping frequency f v When the chopping frequency f v Fixed, with the load resistor R O The resistance value increases from the heavy load stage to the light load stage, the load power W(R O ) first rise and then fall;

[0051] In the heavy load stage, the switching frequency is between 80 and 85kHz, which is more efficient. Constant current regulation is achieved by reducing the frequency and increasing the transmission power. In the light load stage, the switching frequency is between 90 and 120kHz, which is more efficient. Constant voltage regulation is achieved by increasing the frequency.

[0052] The present invention 1) adds an auxiliary bridge arm to a single-phase full-bridge inverter to form a three-phase inverter, using a three-phase primary coil and a three-phase compensation circuit on the primary side, and a single-phase rectifier circuit on the secondary side. The function of adding the auxiliary bridge arm is to form a charging loop with the help of the diode reverse freewheeling circuit during the working mode switching process, thereby providing a relatively long soft switching condition for the switch tube.

[0053] 2) The three-phase inverter is used for inductive wireless charging of electric vehicles. It has the characteristics of high power and high heavy-load efficiency. However, this system is not suitable for adjusting the transmission power using a variable duty cycle method, and the resonant network has a frequency-selective characteristic. When a frequency with high heavy-load efficiency is selected, the light-load circulating current is large and the light-load efficiency is significantly lower than the heavy-load efficiency. In order to further improve the transmission efficiency and transmission power, the present invention adopts a variable frequency fixed duty cycle control strategy for a three-phase inductive power transmission system based on LCC compensation, and summarizes the soft switching frequency band optimization and impedance matching method. First, the calculation formulas for the impedance and resonant frequency of the composite resonant network, the inherent impedance and resonant frequency of the primary coil and its compensation circuit, and the inherent impedance and resonant frequency of the secondary coil and its compensation circuit are derived; then, the coordination principles between these three resonant frequencies and the switching frequency are summarized, and the primary resonant circuit is regarded as a bandpass filter; the secondary resonant circuit is regarded as a low-pass filter, and the main resonant frequency of the composite resonant network is set to be slightly lower than the switching frequency and in a reasonable position in the filter passband to obtain higher efficiency and greater transmission power. In this way, the soft switching frequency range of the switching tube is optimized, and the component parameters are determined based on this relationship and combined with simulation.

[0054] 3) Using a frequency conversion method to match impedance, the transmission power is adjusted and efficiency is improved when the load changes. Circuit analysis and simulation verify that the circuit has strong robust natural soft switching capability within the switching frequency range and during frequency conversion.

[0055] The present invention adopts a variable frequency constant duty cycle control strategy for a three-phase inductive power transmission system based on LCC compensation, and summarizes a soft switching frequency band optimization and impedance matching method, which has the following beneficial effects:

[0056] 1) The three-phase LCC compensation circuit, combined with a three-phase inverter, generates both a power supply circuit and a freewheeling circuit during commutation. The power supply circuit provides freewheeling energy for a long period of time. This freewheeling method is the key factor in enabling the inverter's switches to automatically achieve zero-current shutdown and zero-voltage conduction within the switching frequency and load range.

[0057] 2) The frequency matching method is used to ensure that the inverter switch tube always operates in a natural soft switching state and has strong robustness.

[0058] 3) Utilizing the frequency selection characteristics of the contactless power supply system, the appropriate transmission power is selected while optimizing the transmission efficiency. The frequency reduction method is used in the heavy-load phase to increase the transmission power; the frequency increase method is used in the light-load phase to reduce the transmission power and reduce the circulating current of the primary resonant circuit, without the need for an additional resonant circuit or complex control method. Compared with the parameter design method in the literature [Zhou Chenghu, Huang Mingming, Gao Zhendong, et al. Design and modeling analysis of a three-phase LCL compensation wireless charging system [J]. China Testing. 2022, 48(8): 35-43.], the frequency increase method used in the light-load phase improves the light-load efficiency of the present invention by approximately 3-5%, and further simplifies the circuit structure. Circuit analysis and simulation verify that the optimized circuit of the present invention has a strong and robust natural soft switching capability within the switching frequency range and during the frequency conversion process. BRIEF DESCRIPTION OF THE DRAWINGS

[0059] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the following briefly introduces the drawings required for use in the embodiments or the description of the prior art. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying any creative work.

[0060] Figure 1 FIG. 4 is a schematic diagram of an example circuit of the contactless power supply system of the present invention.

[0061] Figure 2 for Figure 1 The equivalent circuit of the primary and secondary coils and compensation circuit is shown.

[0062] Figure 3 It is the B-phase half-bridge circuit equivalent to the three-phase inverter.

[0063] Figure 4 Schematic diagram of matching of four frequencies of the present invention.

[0064] Figure 5 The composite resonant network frequency and the switching frequency f v Simulation diagram of effective regulation area and load power.

[0065] Figure 6 1 and 2 are the operating modes of the three-phase inverter of the present invention, wherein (a) represents the six basic operating modes and (b) represents the switching between the basic operating mode and the freewheeling mode.

[0066] Figure 7This is a schematic diagram of the process of switching from working mode ST1 to working mode ST2 of the present invention, wherein (a) is working mode ST1, (b) is the first transition process of switching from working mode ST1 to working mode ST2, and (c) is the second transition process of switching from working mode ST1 to working mode ST2.

[0067] Figure 8 This is a simulation result diagram of the present invention.

[0068] Figure 9 This is the experimental waveform diagram of the present invention.

[0069] Figure 10 The load power and efficiency curves of the variable frequency control of the present invention are shown in FIG. 1 , where (a) is the load power curve and (b) is the efficiency curve. DETAILED DESCRIPTION

[0070] The technical solutions in the embodiments of the present invention will be described clearly and completely below with reference to the accompanying drawings in the embodiments of the present invention.

[0071] A method for optimizing the topology structure and frequency matching of a contactless power supply system. On the basis of optimizing the topology structure, the calculation formulas of the impedance of the composite resonant network and its resonant frequency, the inherent impedance of the primary coil and its compensation circuit and its resonant frequency, and the inherent impedance of the secondary coil and its compensation circuit and its resonant frequency are first derived; then the coordination principles between these three resonant frequencies and the switching frequency are summarized to optimize the soft switching frequency range of the switching tube, and the component parameters are determined based on this relationship and in combination with simulation; the frequency conversion method is used to match the impedance, adjust the transmission power when the load changes, and improve the light load efficiency and transmission efficiency. The present invention regards the primary resonant circuit as a bandpass filter; the secondary resonant circuit as a low-pass filter, and sets the main resonant frequency of the composite resonant network to a reasonable position slightly lower than the switching frequency and in the filter passband to obtain higher efficiency and greater transmission power. Circuit analysis and simulation verify that the optimized contactless power supply circuit has a strong and robust natural soft switching capability within the switching frequency range and during the frequency conversion process. The specific steps of the present invention are:

[0072] Step 1: Construct a contactless power supply circuit with optimized topology: Add an auxiliary bridge arm to the single-phase full-bridge inverter to form a three-phase inverter. The three-phase inverter is connected to the three-phase primary coils through a three-phase compensation circuit. A primary coil L p32 With the secondary coil L s32 Phase coupling, the secondary coil is connected to the load through a single-phase rectifier circuit.

[0073] The present invention is applicable to a contactless power transmission topology. An embodiment of the topology is as follows: Figure 1As shown, it consists of a three-phase inverter, a three-phase primary coil, a secondary coil, a three-phase compensation circuit and a rectifier circuit. The three-phase inverter converts DC power into high-frequency AC power, which is transmitted to the secondary circuit by the three-phase primary coil, the secondary coil and the compensation circuit. The present invention adds an auxiliary bridge arm to the single-phase full-bridge inverter to form a three-phase inverter. The three-phase inverter is a three-phase square wave inverter. The DC power supply U L It is connected to the three-phase square wave inverter through the inductor L1. The function of the inductor L1 is to reduce the switching process of the switch tube to the DC power supply U L The auxiliary bridge arm is the bridge arm where switch tubes S1 and S2 are located. The single-phase full-bridge inverter includes switch tubes S3-S6. Switch tubes S3-S4 form one bridge arm, and switch tubes S5-S6 form another bridge arm.

[0074] The primary circuit uses a three-phase primary coil and a three-phase compensation circuit. The three-phase primary coil includes the primary coil L p31 , primary coil L p32 and the primary coil L p33 , the three-phase compensation circuit consists of capacitor C p31 ~C p33 and capacitor C 31 ~C 33 Composition. Primary coil L p32 With the secondary coil L s32 Phase coupling, the other two phase primary coils L p31 、L p33 Not with the secondary coil L s32 Coupling can be replaced by inductance.

[0075] Secondary coil L s32 With capacitor C s32 The secondary circuit uses a single-phase rectifier circuit, which includes diodes D2, D5 and capacitors C4, C5. The function of the auxiliary bridge arm is to make the freewheeling diode connected in parallel with the switch tube reversely flow to form a charging circuit during the working mode switching process, providing a longer soft switching condition for the switch tube. p31 、L p33 Ordinary core or coreless inductor can be used, the primary coil L p32 To the secondary coil L s32 Induction transfers electromagnetic field energy.

[0076] Three-phase inverters, used for inductive wireless charging of electric vehicles, offer high power and high heavy-load efficiency. However, these systems are not suitable for adjusting transmission power using a variable duty cycle. Furthermore, the resonant network exhibits frequency-selective characteristics. When selecting a frequency with high heavy-load efficiency, the light-load circulating current is large, significantly lower than the heavy-load efficiency. To further improve transmission efficiency and power, this paper proposes a variable-frequency, constant-duty-cycle control strategy for a three-phase inductive power transmission system based on LCC compensation, resulting in a soft-switching frequency band optimization and impedance matching method.

[0077] Step 2: Equivalently treat the single-phase rectifier circuit as a current source, and the three-phase inverter and three-phase compensation circuit of the primary circuit as a half-bridge circuit. According to Kirchhoff's law and transformer principle, the composite resonant network impedance Z of the primary and secondary coils and the three-phase compensation circuit is solved. in .

[0078] Figure 1 The voltage doubler rectifier circuit of the secondary circuit shown has the characteristics of a current source and can be equivalent to a current source. The current i of the single-phase rectifier circuit is s32 The average value of the current I s32 Indicates that the current i s32 The positive half cycle flows through diode D2 and output load R O , capacitor C5, current i s32 The negative half cycle flows through capacitor C4 and output load R O , diode D5, so the output current I DC =I s32 / 2. Ignoring the energy loss of diode D2, diode D5, capacitor C4, and capacitor C5, according to the law of conservation of power: I s32 2 ·R eq2 =I DC 2 ·R O , then the equivalent resistance of the secondary circuit is R eq2 It can be expressed as:

[0079]

[0080] Among them, R O is the output load.

[0081] Because the current i of the single-phase rectifier circuit s32 Always only at voltage u s32 Turn on near the peak, voltage u s32 The phase and current i s32 The fundamental wave phase is close to the same phase, so the rectifier circuit, filter capacitor and load resistor can be equivalent to a resistor. Under moderate load conditions, the voltage U load It can be expressed as:

[0082] U load ≈1.2U2(2)

[0083] Among them, U2 represents the effective value of voltage u2. Load changes will cause voltage U load The ratio of the voltage U2 has a slight change, but it does not affect the correctness of the theoretical derivation of this article. U can be measured in real time in the application circuit. load The instantaneous value of the voltage u2 is calculated to get the accurate ratio. p32 , compensation capacitor C p32 The equivalent circuit of the secondary circuit is as follows Figure 2 shown.

[0084] according to Figure 1 、 Figure 2 Using Equations (1) and (2), we can get the secondary circuit (the secondary circuit includes the secondary coil L) by using Kirchhoff’s current and voltage laws. s32 , compensation capacitor C s32 and the equivalent resistance R eq2 ) impedance Z r32 (jω T ):

[0085]

[0086] Among them, r s32 is the inductance L s32 The internal resistance, ω T is the switching frequency of the switch tubes S1 to S6. Using Kirchhoff's current and voltage laws, according to Figure 1 、 Figure 2 The equivalent impedance Z from the primary coil to the load of the three-phase inverter is obtained by using equations (1) and (3): 32 (jω T )for:

[0087]

[0088] Among them, r p32 is the inductance L of the primary coil p32 The internal resistance; M is the inductance L p32 and secondary coil L s32 The mutual inductance is proportional to the coupling coefficient k, and According to the simulation optimization results, k=0.21 is set. Represents capacitance C s32 The voltage across the terminals, Indicates the current flowing through the inductor L p32 The inductance and capacitance C at both ends s32 The secondary coil L s32 Parallel filter capacitors.

[0089] Equivalent impedance Z 32(jω T ) and capacitor C s32 Parallel triangle impedance Z Δ32 (jω T ) represents the triangle impedance Z Δ32 (jω T ) is:

[0090]

[0091] Will Figure 1 In the circuit shown, the three-phase primary coils and the three-phase compensation circuit after points a, b, and c are regarded as the triangle load of the three-phase inverter. After the triangle load is converted into a star load using the triangle and star load conversion method, the center point o of the star load can be regarded as the output neutral line. Therefore, the three-phase inverter can be equivalent to a topology of three independent half-bridges, where the equivalent B-phase half-bridge circuit is as follows: Figure 3 As shown. According to the conversion method of circuit theory, they are converted into star loads, where star load Z 12 (jω T ) can be expressed as:

[0092]

[0093] According to expressions (1) to (6) and the impedance calculation method in circuit theory, the impedance Z of the composite resonant network including the primary and secondary coils and the three-phase compensation circuit can be obtained: in The expression is:

[0094]

[0095] Among them, the capacitor C p32 is the capacitance and inductance L in the three-phase compensation circuit p32 For the coupled primary coil, capacitor C s32 The secondary coil L s32 The parallel capacitor, r s32 is the inductance L s32 internal resistance.

[0096] Step 3: The impedance Z of the composite resonant network in Find the imaginary part and set it equal to 0 to derive the frequency f of the composite resonant network T Without considering electromagnetic coupling, the three-phase primary coil and the three-phase compensation circuit form a primary resonant circuit. The primary resonant circuit is regarded as a bandpass filter, and the inherent impedance Z of the primary resonant circuit is solved. p , according to the primary side inherent impedance Z p , solve the natural resonant frequency f of the primary coil and the three-phase compensation circuit p Secondary coil L s32 and compensation capacitor Cs32 Construct a secondary resonant circuit, treat the secondary resonant circuit as a low-pass filter, and solve the natural resonant frequency f of the secondary circuit s .

[0097] Since the three-phase inverter, primary resonant circuit and secondary resonant circuit are cascaded, the system has three resonant frequencies: composite resonant network frequency f T , the natural resonant frequency f of the primary coil and its compensation circuit p and the natural resonant frequency f of the secondary coil and its compensation circuit s The following are three methods for calculating the resonant frequency.

[0098] Solve the impedance Z of the composite resonant network in The corresponding composite resonant network frequency f T First, the composite resonant network impedance Z in Calculate the imaginary part Im(Z in ) can be obtained:

[0099]

[0100] Then let the imaginary part Im(Z in )=0Ω, use the calculation method to find the angular frequency ω T , according to ω T =2πf T , the composite resonant network frequency f can be derived T The expression of , discarding the solution containing imaginary numbers, obtains two real number solutions as follows:

[0101]

[0102] Define f T1 is the composite resonant network frequency f T The main resonant frequency of T2 is the composite resonant network frequency f T According to circuit theory, let the imaginary part be equal to 0, discard the solution containing imaginary numbers, and the real number solution is the composite resonant network frequency f T .

[0103] Calculate the natural resonant frequency f of the primary coil and three-phase compensation circuit p : Without considering electromagnetic coupling, the primary resonant circuit can be regarded as a bandpass filter, and the primary inherent impedance Z is obtained p for:

[0104]

[0105] Among them, ω p Represents the natural resonant angular frequency of the primary coil and its compensation circuit, ω p=2πf p .

[0106] The primary side inherent impedance Z is calculated from formula ((10) p The imaginary part Im(Z p ):

[0107]

[0108] Let the imaginary part Im(Z p ) is zero, the primary side natural resonant frequency f can be derived p The solution f p1 and f p2 :

[0109]

[0110] Calculate the natural resonant frequency f of the secondary coil and its compensation circuit s Without considering electromagnetic coupling, the secondary resonant circuit can be regarded as a low-pass filter, and the inherent impedance Z of the secondary coil and its compensation circuit is s The expression is:

[0111]

[0112] Let the inherent impedance Z s The imaginary part is 0, and the natural resonant frequency f of the secondary coil and its compensation circuit is obtained. s Solution:

[0113]

[0114] Among them, ω s Represents the natural resonant angular frequency of the secondary coil and its compensation circuit, ω s =2πf s .

[0115] Step 4: To keep the switch in soft switching mode:

[0116] (1) Switching frequency and the natural resonant frequency f of the secondary coil and its compensation circuit s The relationship should satisfy the following conditions: vmax ≤f s ≤2·f vmin ; Define the chopping frequency f of the switching tube v , use f vmin and f vmax Represent the minimum switching frequency and maximum switching frequency respectively. Chopping frequency f v It is the frequency of driving the switching tube set by the controller through software.

[0117] (2) Natural resonant frequency f phas two positive real roots f p1 and f p2 , let f p1 <f p2 , calculate the parameters of components such as inductance and capacitance according to equations (9), (12), and (14), and select components to make the switching frequency f v Satisfy the conditions: f p1 ≤f vmax ≤f s ≤2·f vimn ;

[0118] The method of selecting components to meet the above conditions is: use the method of calculating the parameters of components such as inductance and capacitance to select components so that the switching frequency f v Satisfy the conditions: f p1 ≤f vmax ; Then select the component to make f vmax ≤f s ≤2·f vmin Calculate the parameters of components such as inductance and capacitance according to expressions (9), (12), and (14). Reasonable selection of components can broaden the soft switching frequency range of the switch tube and improve transmission efficiency and transmission power.

[0119] (3) According to equations (9), (12), and (14), the method of calculating the parameters of components such as inductance and capacitance is used to select components so that the frequency f of the composite resonant network is T The main resonant frequency f T1 With the switching frequency f v Satisfies the relationship: f T1 ≤f vmin ;

[0120] Main resonant frequency f T1 It should be slightly lower than the chopping frequency f v , and must be within the filter's passband to achieve greater transmission power. The characteristic diagrams of the four frequencies are as follows Figure 4 As shown. Since the primary resonant circuit is regarded as a bandpass filter, Figure 4 The natural resonant frequency f p The amplification factor is bandpass; since the secondary resonant circuit is regarded as a low-pass filter, Figure 4 The natural resonant frequency f s The amplification factor is low-pass. In addition, Figure 4 It can be seen intuitively that the main resonant frequency f T1 and the primary side natural resonant frequency f p1 If the switching frequency f v The upper and lower limits cannot fall within the natural resonant frequency f p Or the natural resonant frequency f s Within the passband, it will directly affect the transmission power and filtering effect.

[0121] Step 5: Determine component parameters through simulation, use frequency conversion method to match impedance, and adjust transmission power when load changes.

[0122] Use computer software to draw the circuit diagram and simulate to verify the frequency f of the composite resonant network T and the switching (chopping) frequency f v The effective regulation area, load power P(R O ) is shown in the corresponding diagram as Figure 5 As shown, where the load power P(R O ) uses logarithmic coordinates. Figure 5 It can be seen that at the chopping frequency f v ≈85kHz, when the chopping frequency f v Reduce, load power P(R O ) rises, which is the basic principle and theoretical basis of control system design. When the composite resonant network frequency f T and chopping frequency f v If the coordination deviation is too large, it will result in too low transmission power or out of the soft switching state. v Cross Figure 5 The monotonically decreasing range (80-120kHz) shown in the figure may enter the monotonically increasing range. The frequency drop in the monotonically increasing range will cause the load power P(R O )decline.

[0123] Based on the analysis of the working mode, the principle that the switching tube of the three-phase inverter works in a natural soft switching state with strong robustness is explained.

[0124] Figure 1 In the contactless power supply circuit shown, a single-phase full-bridge inverter is composed of switches S3-S6, with switches S1 and S2 forming an auxiliary bridge arm. The addition of the auxiliary bridge arm allows switches S1-S6 to form a three-phase inverter. During each operating mode transition, a corresponding power supply circuit and freewheeling circuit are generated. These two circuits and their generation are described in detail below, as the two switching processes from operating mode ST1 to operating mode ST2 progress. The power supply circuit supplies power to the freewheeling circuit. Because the power supply circuit supplies power to the freewheeling circuit, the freewheeling circuit keeps the antiparallel diodes of the switching switches to be switched on throughout the operating mode transition, resulting in robust soft switching.

[0125] The working mode of the three-phase inverter set in this embodiment is as follows Figure 6 shown. Figure 6 (a) shows six basic working modes ST1 to ST6. In each basic working mode, two upper bridge arms or two lower bridge arms are turned on at the same time, and the phase angle θ changes in the clockwise direction as shown in FIG. Figure 6(b) shown.

[0126] Taking the switching from working mode ST1 to working mode ST2 as an example, the working principle of soft switching is analyzed. In working mode ST1, the switch tube S1 of the upper bridge arm is turned on, and the switch tubes S4 and S6 of the lower bridge arm are turned on at the same time. Figure 7 (a) In this state, the power supply current i1 flows through the switch tube S1 and the capacitor C p31 , then split into two paths, one path through impedance Z Δ31 , capacitor C p32 , the switch tube S4 returns to the negative pole of the power supply; the other path passes through the impedance Z Δ33 , capacitor C p33 , the switch tube S6 returns to the negative pole of the power supply. The two parallel voltages are equal, and the voltage drops of the switch tubes S4 and S6 are close to 0V. In this process, the capacitor C p31 The voltage is positive on the left and negative on the right, and the capacitor C p32 、C p33 The voltage is positive on the right and negative on the left.

[0127] Switching from working mode ST1 to working mode ST2 requires two processes:

[0128] 1) The first transition process is as follows Figure 7 (b) As shown. During this process, the switch tube S4 is turned off, and the voltage across the drain and source of the switch tube S4 rises slightly, causing the two voltages to no longer be equal. Therefore, the current of the switch tube S4 quickly returns to zero and is turned off in the zero current state. The power supply current i1 flows through the switch tube S1 and the capacitor C p31 , impedance Z Δ33 , capacitor C p33 Together with the switch tube S6, it forms a power supply circuit.

[0129] 2) The second transition process is shown in Figure (c). During this process, the capacitor C p31 Voltage is positive on the left and negative on the right, capacitance C p32 The voltage on the right is positive and on the left is negative. p31 , impedance Z Δ31 , capacitor C p32 and diode D s3 In the freewheeling circuit formed by the diode D s3 The switch S3 is turned on and the current is continued, so that the switch S3 to be turned on is in the zero voltage soft switch on state. In addition, this process also includes the power supply current i1 passing through the switch S1 and the capacitor C p31 , impedance Z Δ33 , capacitor C p33 and the switch tube S6 form a power supply circuit. Because the capacitor C p31 It is in the discharge state in the freewheeling circuit and in the charging state in the power supply circuit, so the diode D s3 The conduction time is long.s3 When the freewheeling circuit is turned on, the turn-on voltage of the switch tube S3 is always kept at 0V, so the switch tube S3 is closed in the soft switching state with strong robustness. The mode of forming the freewheeling circuit and the power supply circuit before the switch tube S3 is turned on in the second transition process is called the freewheeling mode.

[0130] From the above analysis, it can be seen that due to the function of the auxiliary bridge arm, a freewheeling mode will be generated every time during the second transition process of switching between two working modes.

[0131] In order to further analyze the working principle of achieving natural soft switching state, Figure 1 The circuit shown is used for full system simulation. Set the input voltage U L =220V, load R O =100Ω, the simulation results are as follows Figure 8 As shown. Figure 8 It can be seen that:

[0132] During the period t0 to t1, the switches S1, S4, and S5 are turned on, and the operating mode is ST6.

[0133] During the period t1 to t′1, the switches S1 and S4 are turned on, and the operating mode is the first transition phase from ST6 to ST1. The driving voltage u of the switch S5 is g5 is low level, the output current i c It drops to 0A instantly, and the switch tube S5 is turned off with zero current. From the simulation waveform of the power W(S5) of the switch tube S5, it can be determined that the switch tube S5 is turned off in the soft switching state.

[0134] During the period t′1 to t2, the switches S1 and S4 are turned on, the switch S5 is turned off, and the diode D s6 Freewheeling: This stage is the second transition stage from working mode ST6 to ST1 and the working mode ST1 stage; when the duty cycle of the switches S5 and S6 is between 36% and 50%, the driving voltage u of the switch S6 during this period is g6 The voltage W(S6) of the switch tube S6 changes to a high level. From the waveform of the power W(S6) of the switch tube S6, it can be determined that the switch tube S6 is turned on in the soft switching state.

[0135] During the shutdown process of each switch tube S1~S6, the output current i a 、i b 、i c The waveform quickly drops to 0A; in each process of the switch tube S1~S6 being turned on, the diode connected in anti-parallel with the switch tube conducts and continues the current, making the switch tube in the zero voltage soft switching state. p1 ≤f vmax ≤f s ≤2·f vimnWhen , the switch tube has natural soft switching conditions.

[0136] When the load box resistor R O =100Ω, switching frequency f v =84kHz, the experimental waveform is as follows Figure 9 As shown. During each basic working mode switching process of the three-phase inverter of the contactless power supply circuit, the current i a 、i b 、i c In the first transition stage, the current drops to 0A instantly, which is a typical characteristic of zero-current shutdown of the switching tube.

[0137] from Figure 7 (b) It can be seen that in the first transition stage when the working mode switches from ST6 to ST1, the current i c Drops to 0A, when the current i c The switch S5 is turned off during the process of 0A. s5 =0A, the instantaneous power p5 when the switch tube S5 is turned off =u cn ·i s5 =0W, so the switch tube S5 is soft-off.

[0138] from Figure 7 (c) It can be seen that in the second transition stage when the working mode switches from ST6 to ST1, the diode D s6 Freewheeling, its current i ds6 The time is long enough to ensure that the switch tube S6 is turned on in the zero-voltage soft switching state.

[0139] Because the three-phase inverter's horizontal commutation offers the robustness of soft switching, and its vertical commutation has sufficient dead time, ensuring that the power supply current i1 always has a conductive path at the moment of commutation, it offers significant advantages over the commonly used single-phase inverter commutation. Dead time refers to the inability of the upper and lower switches in the same arm to conduct or switch state at the same time.

[0140] The experimental results show that (the chopping frequency is controlled by the controller and is programmed; the load resistance can be obtained by measuring the resistance value or the ratio of voltage and current online). When the duty cycle of all switches S1 to S6 is 40-42%, the chopping frequency f of the switch is v It is adjusted between 80 and 120kHz, and the load resistance R OUnder conditions of 50Ω to 600Ω, the switches operate in a soft switching state, achieving maximum robustness. To simplify analysis, the present invention sets the duty cycle of all switches S1 to S6 to 41%. The variable frequency control of the present invention does not require complex control methods or additional resonant or control circuits. The switches can be any switch capable of chopping, such as MOS transistors, IGBTs, or other switching transistors.

[0141] The charging process of electric vehicle batteries includes constant current and constant voltage charging stages. The load is heavier during the constant current charging stage and lighter during the constant voltage charging stage. The load power curve of variable frequency control is as follows: Figure 10 (a) shows the load power W(R O )With the switching frequency f v When the switching frequency f v Fixed, with load R O As the resistance value increases (from heavy load to light load), the load power first increases and then decreases.

[0142] The efficiency curve of variable frequency control is as follows: Figure 10 (b) shows the peak value of the efficiency curve changes with the change of switching frequency and load resistance. By comparing the switching frequency f v The efficiency curves of 80kHz, 85kHz, 90kHz, 100kHz and 120kHz respectively show that when the DC power supply U L Under constant conditions, the transmission efficiency peak of each frequency corresponds to different load resistance. By properly selecting components, the switch tube of the contactless power supply system can select the appropriate switching frequency range. When the load resistance R O Below 170Ω, the switching frequency f v When the load resistance R is equal to 85kHz, the transmission efficiency of the system is relatively high; when the load resistance R O At around 200Ω, the switching frequency f v When the load resistance R is equal to 80kHz, the transmission efficiency of the system is relatively high; when the load resistance R O At around 300Ω, the switching frequency f v When the load resistance R is equal to 90kHz, the transmission efficiency of the system is relatively high; when the load resistance R O At around 400Ω, the switching frequency f v When the load resistance R is equal to 100kHz, the transmission efficiency of the system is relatively high; when the load resistance R O Above 500Ω, the switching frequency f v When the DC power supply U is equal to 120kHz, the transmission efficiency of the system is relatively high. L If the frequency is changed, the load resistance value corresponding to the peak transmission efficiency of each frequency will also change accordingly.

[0143] Based on the above analysis, a switching frequency between 80 and 85 kHz achieves higher efficiency during heavy loads, using frequency reduction to increase transmission power to achieve constant current regulation. A switching frequency between 90 and 120 kHz achieves higher efficiency during light loads, using frequency increase to achieve constant voltage regulation. Selecting the appropriate switching frequency based on load power requirements maintains high load efficiency.

[0144] Set the load resistance to 50Ω≤R O ≤200Ω is the heavy load stage, the load current I DC =4.77A, with R O As the resistance value increases, the load power increases and the switching frequency f v When adjusting from 85kHz to 80kHz, the transmission power increases from 1.14kW to 3.87kW. O =100Ω, f v =84kHz, from DC power supply U L To load R O The transmission efficiency is 94.4%.

[0145] Set the load resistance to 200Ω <R O ≤600Ω is the light load stage, the load voltage U DC ≈600V, with load R O As the resistance value increases, the load power decreases and the switching frequency f v Gradually rising from 84kHz to 120kHz, the transmission efficiency power gradually decreased to 0.57kW. L To load R O The maximum light-load transmission efficiency is 96.3%. Compared with the parameter design method in the literature [Zhou Chenghu, Huang Mingming, Gao Zhendong, et al. Design and modeling analysis of three-phase LCL compensation wireless charging system [J]. China Testing. 2022, 48(8): 35-43.], the light-load efficiency is improved by about 3-5%.

[0146] To improve the light-load efficiency of a three-phase LCC wireless charging system, the impedance of the composite resonant network and its resonant frequency, the inherent impedance of the primary coil and compensation circuit, and the inherent impedance of the secondary coil and compensation circuit, as well as their resonant frequencies, were derived for the three-phase LCC wireless charging system. The coordination principles of these three resonant frequencies and switching frequencies were summarized to select the soft-switching frequency range of the switch tube. A variable-frequency constant-duty-cycle control strategy was then used to summarize the soft-switching frequency band optimization and impedance matching methods. The present invention has a highly robust natural soft-switching capability within the frequency and load ranges, and a frequency-boosting method can be used to reduce the circulating current in the resonant network during the light-load phase. The feasibility of the present invention was verified using a prototype with a transmission power of 3.87kW. When the load was adjusted between 100 and 600Ω, the transmission efficiency varied between 94.4% and 96.3%, which is approximately 3% higher than that of similar three-phase systems.

[0147] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. All other embodiments obtained by other technicians without creative work are within the scope of protection of the present invention; any modifications, equivalent substitutions, improvements, etc. made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A method for topology optimization and frequency matching of a contactless power supply system, characterized in that: The steps are as follows: Step 1: Construct a contactless power supply circuit with optimized topology: Add an auxiliary bridge arm to a single-phase full-bridge inverter to form a three-phase inverter. The three-phase inverter is connected to the three-phase primary coils through a three-phase compensation circuit. One primary coil is coupled to the secondary coil, and the secondary coil is connected to the load through a single-phase rectifier circuit. Step 2: Equivalently treat the single-phase rectifier circuit as a current source, and the three-phase inverter and three-phase compensation circuit as a half-bridge circuit. Calculate the composite resonant network impedance Z of the primary coil, secondary coil, and three-phase compensation circuit based on Kirchhoff's law and transformer principle. in ; Step 3: The impedance Z of the composite resonant network in Find the imaginary part and set it equal to 0 to solve the frequency f of the composite resonant network T Without considering electromagnetic coupling, the primary resonant circuit composed of the three-phase primary coil and the three-phase compensation circuit is regarded as a bandpass filter, and the inherent impedance Z of the primary circuit is solved. p , according to the primary side inherent impedance Z p , solve the natural resonant frequency f of the primary circuit p ; Consider the secondary resonant circuit composed of the secondary coil and the compensation capacitor as a low-pass filter, and solve the natural resonant frequency f of the secondary circuit s ; The solution of the composite resonant network frequency f T The method is: the composite resonant network impedance Z in Calculate the imaginary part Im(Z in ) can be obtained: Let the imaginary part Im(Z in )=0Ω, use the calculation method to find the angular frequency ω T , according to ω T =2πf T , derive the composite resonant network frequency f T The expression of , discarding the solution containing imaginary numbers, obtains 2 real solutions: Among them, f T1 is the composite resonant network frequency f T The main resonant frequency of T2 is the composite resonant network frequency f T High resonant frequency; r p32 is the inductance L of the primary coil p32 Internal resistance; capacitance C p32 is the capacitance in the three-phase compensation circuit, capacitance C s32 is the inductance L of the secondary coil s32 Capacitors in parallel; r s32 is the inductance L s32 The internal resistance, M is the inductance L p32 and secondary coil L s32 Mutual inductance; R O is the output load; Step 4: To keep the switch tube of the three-phase inverter in the soft switching state: (1) The chopping frequency f of the switching tube v and the natural resonant frequency f of the secondary circuit s The relationship should satisfy the following conditions: vmax ≤f s ≤2·f vmin ;f vmin and f vmax Represent the minimum switching frequency and the maximum switching frequency respectively; (2) Natural resonant frequency f p The two positive real roots f p1 and f p2 , let f p1 <f p2 , calculate the parameters of the inductor and capacitor components, and select the components to make the switching frequency f v Satisfy the conditions: f p1 ≤f vmax ≤f s ≤2·f vimn ; (3) Select components to make the composite resonant network frequency f T The main resonant frequency f T1 and chopping frequency f v Satisfies the relationship: f T1 ≤f vmin .

2. The method for topology optimization and frequency matching of a contactless power supply system according to claim 1, characterized in that: The three-phase inverter is a three-phase square wave inverter, which converts direct current into high-frequency alternating current, which is transmitted to the single-phase rectifier circuit through the primary coil, the three-phase compensation circuit, and the secondary coil. The auxiliary bridge arm is the bridge arm where the switch tube S1 and the switch tube S2 are located. The single-phase full-bridge inverter includes switch tubes S3-S6. The switch tubes S3-S4 form a bridge arm, and the switch tubes S5-S6 form a bridge arm. The three bridge arms are connected in parallel, and diodes are connected in anti-parallel to the switch tubes S3-S6. During each working mode conversion process, a corresponding power supply circuit and freewheeling circuit are generated. The power supply circuit supplies power to the freewheeling circuit. The freewheeling circuit ensures that the anti-parallel diodes of the switch tube to be switched are always turned on during the working mode conversion process. A compensation capacitor is connected in parallel to the secondary coil.

3. The method for topology optimization and frequency matching of a contactless power supply system according to claim 1 or 2, characterized in that: The composite resonant network impedance Z in The calculation method is: the current i of the single-phase rectifier circuit s32 The average value of the current I s32 Indicates that the current i s32 The positive half cycle flows through diode D2 and capacitor C5, and the current i s32 The negative half cycle flows through capacitor C4 and output load R O , diode D5, so the output current I DC =I s32 / 2; Ignoring the energy loss of the single-phase rectifier circuit, according to the law of conservation of power: I s32 2 ·R eq2 =I DC 2 ·R O , then the equivalent resistance of the secondary circuit is R eq2 for: Among them, R O is the output load; Use Kirchhoff's current and voltage laws to get the impedance Z of the secondary circuit r32 (jω T ): Among them, ω T is the switching frequency of the switches S1 to S6; Using Kirchhoff's current and voltage laws, we can get the equivalent impedance Z from the primary coil to the load of the three-phase inverter. 32 (jω T )for: Among them, the mutual inductance M is proportional to the coupling coefficient k, and Represents capacitance C s32 The voltage across the terminals, Indicates the current flowing through the inductor L p32 The inductance at both ends; Equivalent impedance Z 32 (jω T ) and capacitor C s32 Parallel triangle impedance Z Δ32 (jω T ) indicates that the triangle impedance Z Δ32 (jω T )for: The three-phase primary coil and the three-phase compensation circuit are regarded as the triangle load of the three-phase inverter. The triangle load is converted into a star load. The center point o of the star load is regarded as the output zero line. The three-phase inverter is equivalent to the topology of three independent half-bridges. According to the circuit theory, the star load Z 12 (jω T )for: According to the impedance calculation method in circuit theory, the impedance Z of the composite resonant network including the primary and secondary coils and the three-phase compensation circuit is obtained. in for:

4. The method for topology optimization and frequency matching of a contactless power supply system according to claim 3, characterized in that: The solution to the natural resonant frequency f of the primary circuit is p The method is: regard the primary resonant circuit as a bandpass filter and obtain the primary inherent impedance Z p for: Among them, ω p Represents the natural resonant angular frequency of the primary coil and its compensation circuit, ω p =2πf p ; Calculate the primary side inherent impedance Z p The imaginary part of : Let the imaginary part Im(Z p ) is zero, and the natural resonant frequency f is derived p Solution: Among them, f p1 and f p2 is the natural resonant frequency f p two frequencies.

5. The method for topology optimization and frequency matching of a contactless power supply system according to claim 3, characterized in that: The solution to the natural resonant frequency f of the secondary circuit is s The method is: consider the secondary resonant circuit as a low-pass filter, the inherent impedance Z of the secondary coil and the single-phase rectifier circuit s for: Let the inherent impedance Z s The imaginary part of is 0, and the natural resonant frequency of the secondary circuit is obtained: Among them, ω s Represents the natural resonant angular frequency of the secondary coil and its compensation circuit, ω s =2πf s .

6. The method for topology optimization and frequency matching of a contactless power supply system according to any one of claims 1-2, 4-5, characterized in that: The main resonant frequency f T1 It should be slightly lower than the chopping frequency f v , and must be within the passband of the filter; the natural resonant frequency f p The amplification factor is bandpass; the natural resonant frequency f s The amplification factor is low-pass; the main resonant frequency f T1 and the natural resonant frequency f p1 Closer.

7. The method for topology optimization and frequency matching of a contactless power supply system according to claim 6, characterized in that: Use computer software to draw the circuit diagram and simulate to verify the frequency f of the composite resonant network T and chopping frequency f v The effective regulation area, load power P(R O ) corresponds to the relationship between the chopping frequency f v ≈85kHz, when the chopping frequency f v Reduce, load power P(R O )rise.

8. The method for topology optimization and frequency matching of a contactless power supply system according to claim 7, characterized in that: When the duty cycle of all switches in the three-phase inverter is 40-42%, the chopping frequency f v It is adjusted between 80 and 120kHz, and the load resistance R O Under the condition of 50Ω~600Ω, the switch tube works in the soft switching state and has the strongest robustness.

9. The method for topology optimization and frequency matching of a contactless power supply system according to claim 8, characterized in that: Load power W(R O )With the chopping frequency f v When the chopping frequency f v Fixed, with the load resistor R O The resistance value increases from the heavy load stage to the light load stage, the load power W(R O ) first rise and then fall; In the heavy-load stage, the switching frequency is between 80 and 85 kHz, which is more efficient. Constant current regulation is achieved by reducing the frequency and increasing the transmission power. In the light-load stage, the switching frequency is between 90 and 120 kHz, which is more efficient. Constant voltage regulation is achieved by increasing the frequency.

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