A slow-wave patch resonator, band-pass filter and design method

Abstract

This invention provides a slow-wave patch resonator, a bandpass filter, and a design method. The method includes: sequentially stacking a first metal layer, a first dielectric substrate, a patch layer composed of a metal patch array, a second dielectric substrate, and a second metal layer, totaling five layers; each layer is rectangular in shape; a metallized via array is formed in the second dielectric substrate, and the metallized via array is used to connect the patch layer and the second metal layer, thereby forming a slow-wave patch resonator between the five layers. Based on this, by etching different grooves in the first metal layer and setting different feed line excitations, different types of bandpass filters can be obtained.

Description

Technical Field

[0001] This invention relates to the field of electromagnetic fields and microwave technology, specifically to a high-frequency filter, and more particularly to a slow-wave patch resonant cavity, a bandpass filter, and a design method thereof. Background Technology

[0002] From "seeing words as if seeing faces" to "the Internet of Everything," the past few decades have witnessed the rapid development of wireless communication systems. High-frequency filters, as an indispensable and fundamental component of wireless communication systems, not only need to fulfill functions such as allocating spectrum resources, filtering unwanted signals, and suppressing interference signals, but also face increasingly stringent requirements in terms of physical topology, integrated design methods, design and manufacturing processes and efficiency, device size, and cost in response to the rapid development demands of miniaturization, high power capacity, high selectivity, and low cost in wireless communication systems. For high-frequency band-pass filters (BPFs), researching their lightweight, miniaturization, and high selectivity to match the rapidly evolving wireless communication systems is an inevitable development trend.

[0003] In existing technologies, microstrip or substrate integrated waveguide (SIW) technology is generally used to implement BPF design.

[0004] Although BPFs implemented with single-mode microstrip resonators have a small size and structure and are easy to implement with multiple coupling paths or hybrid electromagnetic coupling, and can introduce multiple finite frequency transmission zeros to improve their selectivity, their low quality factor and low power capacity have limited their development and are not conducive to matching the rapidly developing wireless communication systems.

[0005] While BPFs implemented using SIW structures offer high Q values, their designs typically employ single-mode cavities, leading to excessive size and hindering miniaturization. Although designs utilize incomplete-mode SIW cavities (e.g., half-mode, quarter-mode, or even eighth-mode) or multi-mode SIW structures for miniaturization, SIW's structural limitations restrict miniaturization and make it difficult to introduce multiple finite-frequency transmission zeros through multiple coupling paths to improve selectivity. Furthermore, SIW-implemented BPFs generally suffer from poor passband selectivity and controllability.

[0006] Filters based on dual-mode patch resonators (BPFs) possess the advantages of microstrip structures, such as wide operating frequency coverage, light weight, low cost, and easier integration with other devices, as well as good power capacity, high selectivity, and relatively high Q value. However, existing dual-mode BPFs generally suffer from poor bandwidth tunability and weak zero-point controllability. Even some dual-mode or multi-mode BPFs have been shown to possess a certain degree of selectivity, but the adjustable range is generally narrow. Facing increasingly demanding high-performance wireless communication systems, dual-mode patch filters need further improvements in miniaturization and selectivity to meet current development needs.

[0007] Meanwhile, in existing technology research, most studies only focus on single-cavity dual-mode second-order filters. However, in practical engineering applications, single-cavity dual-mode second-order filters struggle to meet the stringent technical specifications of wireless communication systems. Even when single-cavity multi-mode filters of third order or higher are designed, their implementation is difficult and requires high precision in manufacturing. Research on implementing higher-order filters based on single-cavity multi-mode filters mostly avoids or only proposes simple cascading methods. In fact, significant challenges exist, especially in the cascading of dual-mode or multi-mode BPFs. Simple cascading is generally achieved through gap coupling, which suffers from weak coupling between filters and high energy loss, hindering the full utilization of the superior performance of single-cavity filters. Only a few studies use coupled-line connections, but these also suffer from excessively large overall dimensions. In the design of higher-order BPFs based on single-cavity multi-mode (dual-mode and above) filters, coupled lines achieve effective coupling between the two cavities, but impedance matching presents a trade-off between length and coupling strength. Therefore, existing technologies generally employ a combination of gap coupling and straight-line coupling, but the essence of such methods remains gap coupling. Studies on cascading using only coupling lines are limited, and even when successful, they come at the cost of size, hindering miniaturization. Most importantly, simple cascading does not necessarily achieve highly selective stacking of BPFs, thus negating the purpose of cascading to some extent. Summary of the Invention

[0008] To further improve the miniaturization and high selectivity of patch filters, this invention provides a slow-wave patch resonant cavity, a bandpass filter, and a design method.

[0009] In a first aspect, the present invention provides a design method for a slow-wave patch resonant cavity, comprising:

[0010] The structure consists of five layers stacked sequentially: a first metal layer, a first dielectric substrate, a patch layer composed of a metal patch array, a second dielectric substrate, and a second metal layer; each layer is rectangular in shape.

[0011] A metallized via array is formed in the second dielectric substrate, and the via array is used to connect the patch layer and the second metal layer, thereby forming a slow-wave patch resonant cavity between the 5-layer structure.

[0012] In one possible implementation, two grooves are etched on the first metal layer, and the two grooves are arranged in an intersecting manner.

[0013] In combination with the first possible implementation, in the second possible implementation, the two groove lines are parallel to two adjacent sides of the first metal layer.

[0014] In combination with the first possible implementation, in the third possible implementation, the two groove lines are parallel to the two diagonals of the first metal layer.

[0015] Secondly, the present invention provides a slow-wave patch resonant cavity, which is obtained by any of the design methods described above.

[0016] Thirdly, the present invention provides a bandpass filter based on a slow-wave patch resonant cavity, comprising: a resonant cavity unit and two port feed lines; the resonant cavity unit is a slow-wave patch resonant cavity formed by the second possible implementation of the first aspect; the two port feed lines are respectively disposed at a non-central position on a set of opposite sides of a first metal layer to connect to the resonant cavity unit.

[0017] Fourthly, the present invention provides a bandpass filter based on a slow-wave patch resonator, comprising: a resonator unit and two port feed lines; the resonator unit is a slow-wave patch resonator formed by the third possible implementation of the first aspect; the two port feed lines are respectively disposed on a set of adjacent sides of a first metal layer and at a distance D from the center point of the side to connect to the resonator; wherein, |D|≥0.

[0018] Fifthly, the present invention provides a bandpass filter based on a slow-wave patch resonator, comprising: two identical resonator units, a folded stepped impedance coupling line, and two port feed lines; wherein the resonator units are slow-wave patch resonators formed using the second possible implementation of the first aspect;

[0019] A port feed line is provided at the off-center position of the first side of the first metal layer of the two resonant cavity units; the folded stepped impedance coupling line is connected at the off-center position of the second side of the first metal layer of the two resonant cavity units respectively; wherein the first side and the second side are opposite sides of each other;

[0020] The bandpass filter as a whole is centrally symmetrical about the center of the folded stepped impedance coupling line.

[0021] In a sixth aspect, the present invention provides a bandpass filter based on a slow-wave patch resonator, comprising: two identical resonator units, a folded stepped impedance coupling line, and two port feed lines; wherein the resonator units are slow-wave patch resonators formed using the third possible implementation of the first aspect.

[0022] A port feeder is provided on the first side of the first metal layer of the two resonant cavity units at a distance D from the center point of the side; the folded stepped impedance coupling line is connected on the second side of the first metal layer of the two resonant cavity units at a distance D from the center point of the side; wherein the first side and the second side are adjacent to each other; wherein |D|≥0.

[0023] The bandpass filter as a whole is centrally symmetrical about the center of the folded stepped impedance coupling line.

[0024] In a seventh aspect, the present invention provides a bandpass filter based on a slow-wave patch resonator, comprising: a first resonator unit, a second resonator unit, a folded stepped impedance coupling line, and two port feed lines; the first resonator unit is a slow-wave patch resonator formed by a second possible implementation of the first aspect, and the second resonator unit is a slow-wave patch resonator formed by a third possible implementation of the first aspect.

[0025] A port feeder is provided at a non-central position on the first side of the first metal layer of the first resonant cavity unit, and a port feeder is provided at a position D on the third side of the first metal layer of the second resonant cavity unit, at a distance D from the center point of the side. One end of the folded stepped impedance coupling line is connected at a non-central position on the second side of the first metal layer of the first resonant cavity unit, and the other end of the folded stepped impedance coupling line is connected at a position D on the fourth side of the first metal layer of the second resonant cavity unit, at a distance D from the center point of the side. Wherein, the first side and the second side are opposite sides, and the third side and the fourth side are adjacent sides. Wherein, |D|≥0.

[0026] The beneficial effects of this invention are:

[0027] (1) The design method provided by the present invention can construct a slow-wave patch resonant cavity; by adjusting the distribution or size of the metallized vias and patches of the slow-wave layer, the cutoff frequency of the resonant cavity can be controlled and reduced according to actual needs without changing the cavity size, thus effectively achieving miniaturization. Compared with the prior art, the present invention can only control the resonant frequency of the cavity by adjusting the slow-wave layer without affecting the electromagnetic field distribution of each mode.

[0028] (2) The design method provided by the present invention involves etching two grooves of unequal size in the first metal layer, and the two grooves of different sizes respectively divide TM 100and TM 010 The resonant frequency of the mode. Different intersection types of the two slot lines can also excite different electromagnetic field distributions of non-degenerate modes, further reducing the resonant frequency of the non-degenerate modes and further realizing miniaturization.

[0029] (3) Combining slot lines and diffused layers can achieve a significant reduction in cutoff frequency, and then combining single-cavity dual-mode can greatly reduce the size. Compared with existing single-cavity single-mode BPFs, the size can be reduced by three-quarters.

[0030] (4) The two types of BPF implemented through slotted lines each have their own characteristics. One type can have strong coupling and achieve a wider bandwidth; the other type can achieve a high degree of freedom in feeder design, allowing for larger processing tolerances, and can also control the bandwidth and the frequency position of the finite frequency transmission zero over a wider range. Compared with the prior art, the present invention has better engineering applicability and higher design freedom in engineering applications.

[0031] (5) The two types of second-order filters proposed in this invention employ different feed positions and feed methods, effectively achieving high selectivity with controllable bandwidth and adjustable zero point. Among them, the "cross-shaped" dual-mode patch filter has a large adjustable range of bandwidth and transmission zero point position, which is rare in the prior art. Moreover, the bandwidth and transmission zero point position adjustment exhibit regular and continuous changes, making it more suitable for engineering design. This also strongly demonstrates its superior design effect of high selectivity.

[0032] (6) A cascaded structure using a folded stepped impedance coupling line to connect the two cavities was proposed, which solved the problem of the contradiction between the coupling strength and impedance matching of the two cavities in the existing dual-mode cascade technology. In addition, the use of a folded stepped impedance coupling line can also achieve miniaturization to a certain extent. At the same time, the size of the folded stepped impedance coupling line can be finely adjusted, as can the bandwidth and the position of the transmission zero of the filter, which also means an improvement in its selectivity.

[0033] (7) Based on the proposed folded stepped impedance coupling line, the high-order filter proposed in this invention adopts a centrally symmetrical cavity layout. Compared with the periodic ordinary cascading method used in the prior art, this invention can achieve the superposition of the superior performance of a single-cavity dual-mode second-order filter, especially in terms of selectivity, which is intuitively demonstrated, and can show the transmission zeros of finite frequencies in S. 21 The zero-point characteristics are obvious on the function curve. Furthermore, by employing a folded stepped impedance coupling line direct connection method, a combination of excellent performance and high selectivity is achieved.

[0034] (8) The present invention adopts a design in which the power supply structure is set in the first metal layer (also known as the top metal layer). The power supply structure is set in the top metal layer, and the power supply structure adopts a microstrip line design, which can make up for the problem of not being easy to integrate with other devices, and makes impedance matching easier, making the overall structure design more flexible and the device applicable to a wider range.

[0035] (9) The design of slow wave structure can confine electromagnetic waves in the resonant cavity to a certain extent, thereby reducing the energy radiation leakage of electromagnetic waves.

[0036] (10) The resonant cavity and filter in this invention do not require special dielectric substrates and present an overall planar structure design, which is easy to integrate and process, and is also conducive to lightweighting. It is applicable to the 5G-Sub 6 working frequency band. Therefore, it can meet the needs of miniaturized, low-cost and lightweight wireless communication systems, and further enhance the prospects and value of market applications. Attached Figure Description

[0037] Figure 1 This is a three-dimensional structural schematic diagram of the slow-wave patch resonator proposed in this invention;

[0038] Figure 2 This is a side view of the slow-wave patch resonator proposed in this invention;

[0039] Figure 3 This is a schematic diagram of the slow-wave layer in the slow-wave patch resonator proposed in this invention;

[0040] Figure 4 The first metal layer of the dual-mode slow-wave patch resonator proposed in this invention is shown in plan view: (a) is a "cross-shaped" groove line, and (b) is a "cross-shaped" groove line;

[0041] Figure 5 This is a planar schematic diagram of the "cross-shaped" SW patch second-order BPF proposed in this invention;

[0042] Figure 6 This is a planar schematic diagram of the "cross-type" SW patch second-order BPF proposed in this invention;

[0043] Figure 7 This is a planar schematic diagram of a "cross-shaped" SW patch fourth-order BPF with a centrally symmetrical layout, based on a direct connection of folded stepped impedance coupling lines proposed in this invention.

[0044] Figure 8 This is a planar schematic diagram of a "cross-type" SW patch fourth-order BPF with a centrally symmetrical layout, based on a direct connection of folded stepped impedance coupling lines proposed in this invention.

[0045] Figure 9This is a planar schematic diagram of the hybrid cavity fourth-order BPF based on the direct connection of the folded stepped impedance coupling line proposed in this invention.

[0046] Figure 10 The above is a simulation diagram of the electric field distribution of the "cross-shaped" SW patch resonator proposed in this invention.

[0047] Figure 11 The above is a simulation diagram of the electric field distribution of the "cross-type" SW patch resonator proposed in this invention.

[0048] Figure 12 The simulated frequency response curve of Embodiment 3 proposed in this invention;

[0049] Figure 13 This is an example of adjustable bandwidth and zero point in Embodiment 3 of the present invention;

[0050] Figure 14 The simulated frequency response curve for Embodiment 4 of this invention;

[0051] Figure 15 This is an example of adjustable bandwidth and zero point in Embodiment 4 of the present invention (adjustment parameter D);

[0052] Figure 16 This is an example of adjustable bandwidth and zero point in Embodiment 4 of the present invention (adjustment parameter g1).

[0053] Figure 17 The simulated frequency response curve of Embodiment 5 proposed in this invention;

[0054] Figure 18 The simulated frequency response curve of Embodiment 6 proposed in this invention;

[0055] Figure 19 This is the simulated frequency response curve of Embodiment 7 proposed in this invention. Detailed Implementation

[0056] To make the objectives, technical solutions, and advantages of this invention clearer, the technical solutions of the embodiments of this invention will be clearly described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of this invention. All other embodiments obtained by those skilled in the art based on the embodiments of this invention without creative effort are within the scope of protection of this invention.

[0057] Example 1

[0058] This invention provides a slow-wave (SW) patch resonator, combined with Figure 1 , Figure 2 and Figure 3As shown, its design process is as follows: Figure 1 As shown, a five-layer structure is constructed by sequentially stacking a first metal layer, a first dielectric substrate, a patch layer composed of a metal patch array, a second dielectric substrate, and a second metal layer; each layer is rectangular in shape. Figure 2 As shown, a metallized via array is formed in the second dielectric substrate, and the metallized via array is used to connect the patch layer and the second metal layer, thereby forming a slow-wave patch resonant cavity between the 5 layers. The patch layer and the second dielectric substrate layer are collectively referred to as the slow-wave layer. It should be noted that the first dielectric substrate and the second dielectric substrate can be different dielectric substrates. Preferably, the first dielectric substrate and the second dielectric substrate are dielectric substrates with the same dielectric constant.

[0059] like Figure 2 As shown, the thickness of the second dielectric substrate is h 1. The thickness of the patch is h 2. The thickness of the first dielectric substrate is h 3.

[0060] like Figure 3 As shown, the patches on the patch layer are circular patches, and the diameter of each circular patch is [missing information]. d 2. The distance (or patch spacing) between two adjacent circular patches is d 3; The diameter of the metallized via (or metal pillar) disposed in the second dielectric substrate is d 1. d 1. d 2 and d 3 are collectively referred to as slow wave layer structure parameters.

[0061] Specifically, by adjusting the size or thickness of the patches on the patch layer, the diameter or height of the metal vias in the second dielectric substrate, and the spacing of the patches (metallized vias), the cutoff frequency of the entire resonant cavity can be reduced, enabling smaller sizes to be used at lower frequencies. In other words, by adjusting the distribution or size of the patches and metallized vias in the slow-wave layer (the patches and metal vias are connected as a whole and cannot be separated), the tunability of the SW patch resonant cavity can be achieved without changing the overall resonant cavity size, effectively realizing miniaturization.

[0062] The function of the second dielectric substrate is to isolate the first and second metal layers, forming an effective slow-wave structure. The height of this layer can also be adjusted to regulate the cutoff frequency of the resonant cavity. Adjusting the height of this layer, while keeping other conditions constant, is essentially equivalent to changing the height of the slow-wave layer. The second metal layer is a metal backplate, which can be fully copper-clad; its function is to confine electromagnetic waves to propagate within the resonator.

[0063] The SW patch resonant cavity provided in this embodiment of the invention can effectively reduce its cutoff frequency without affecting the electromagnetic wave resonance mode of the traditional patch resonator, thus achieving the goal of miniaturization.

[0064] Example 2

[0065] Based on the above embodiment 1, in order to further miniaturize, improve high selectivity, and set the slow-wave patch resonator as a dual-mode slow-wave patch resonator, two groove lines are etched on the first metal layer, and the two groove lines are arranged in an intersecting manner.

[0066] like Figure 4 As shown, the two groove lines are distributed in two ways relative to the first metal layer: as Figure 4 As shown in (a), the two groove lines are parallel to two adjacent sides of the first metal layer. For ease of description, this distribution of groove lines is referred to as "cross-shaped" groove lines; Figure 4 As shown in (b), the two groove lines are parallel to the two diagonals of the first metal layer. For ease of description, the groove lines with this distribution pattern are referred to as "cross-type" groove lines.

[0067] Preferably, both types of grooves extend symmetrically outward from the center of the patch, with the two grooves having unequal lengths.

[0068] In this embodiment, "cross-shaped" or "intersecting" etched grooves are added to the first metal layer. These grooves, on the one hand, disrupt the degenerate mode, creating a non-degenerate dual-mode, making a single cavity equivalent to two single-mode resonators. This effectively halves the size, further achieving miniaturization. On the other hand, the grooves can cut the patch resonator TM. 100 and TM 010 The current in the resonant mode effectively reduces the resonant frequency of both modes, further enabling miniaturization.

[0069] Example 3

[0070] Based on the dual-mode slow-wave patch resonator provided in the above embodiments, this invention provides a bandpass filter based on the dual-mode slow-wave patch resonator, which is a second-order filter with two finite-frequency transmission zeros.

[0071] like Figure 5 As shown, this miniaturized, highly selective bandpass filter is constructed using a dual-mode slow-wave patch resonator with a cross-shaped slot. It consists of a dual-mode SW patch resonator, an input port feed line, and an output port feed line.

[0072] Specifically, both port feeds are located on the first metal layer and are connected to the dual-mode SW patch resonator. The two port feeds are positioned on opposite sides of the first metal layer, at non-central locations on the connected sides. The bandpass filter as a whole is axially symmetric about one of the center lines of the first metal layer. In this embodiment, each layer structure is square with a side length of L1. D represents the distance of the port feed from the center, and L represents the distance of the port feed from the center. p1 and L p2 The length of the two slot lines is represented by W0, and the width of the port feeder is represented by W0.

[0073] By adjusting the distribution and size of the slow-wave layer metal vias and patches, the desired resonant frequency (lower than the original cavity cutoff frequency) can be obtained. The "cross-shaped" groove can segment the degenerate mode, transforming it into a non-degenerate mode™ suitable for BPF design. 100 and TM 010 (The electric field distribution diagram of the non-degenerate mode after cutting is shown in the figure below) Figure 10 (As shown), the groove line can also further reduce TM. 100 and TM 010 The resonant frequency of the mode can be further miniaturized. By setting a reasonable slot line size, the resonant frequencies of the non-degenerate dual modes can be made similar, which can form an effective passband when used in BPF design.

[0074] Based on the distribution of non-degenerate modes, a reasonable feed structure can simultaneously excite both modes to resonate at similar frequencies, thus obtaining the desired single-cavity dual-mode filter. To facilitate easier cascading or integration with other devices, a microstrip line structure is adopted for the feed structure.

[0075] Preferably, W0=2.25, L1=21, L p1 =17.3, L p2 =19.3、 d 1 = 0.8 d 2 = 2.3 d 3=0.3, D=4.75, all units: mm.

[0076] The parameters of the dielectric substrates used are as follows: the relative permittivity of the first dielectric substrate is 2.2 and the thickness is 0.254 mm; the relative permittivity of the second dielectric substrate is 2.2 and the thickness is 0.508 mm; and the thickness of all metal layers is 18 μm.

[0077] Based on this preferred size, the microstrip feed lines are positioned on opposite sides parallel to the shorter slot line. The two feed lines are positioned symmetrically about the shorter slot line. Feed line positions at any other location on this side of the patch (except the exact center) can effectively excite the patch dual-mode resonance. With the longer slot line as the center line, the feeding effect on both sides is symmetrical and equal. By setting different feed line positions, the passband bandwidth and the position of the transmission zero can be adjusted. Controllable bandwidth and adjustable zero demonstrate its high selectivity.

[0078] Ultimately, without changing the patch cavity size, the TM originally resonating at 4.81GHz was... 100 and TM 010 The mode was reduced to resonant around 2.40 GHz, and a highly selective second-order filter with two finite-frequency transmission zeros was successfully designed.

[0079] Based on the above-mentioned preferred dimensions, simulation was performed using electromagnetic simulation software. Figure 12 These are the simulation results of the bandpass filter frequency scattering parameters (S-parameters) in this embodiment. The measured center frequency... f The in-band insertion loss (IL), return loss (RL), and 3-dB bandwidth (BW) are 2.4 GHz, -0.52 dB, -22 dB, and 370 MHz, respectively. The frequency locations of the finite frequency transmission zeros are 2.11 GHz and 3.09 GHz, respectively.

[0080] To demonstrate the high selectivity of this filter, the frequency response was simulated using software to measure variations in size parameters, such as... Figure 13 As shown, when D changes, its bandwidth and transmission zeros change accordingly, exhibiting a regular pattern. Starting from the center of one side of the patch, the greater the offset distance of D, the bandwidth gradually decreases, and the two transmission zeros gradually shrink towards the center frequency, resulting in better passband rectangularity. Controllable bandwidth and adjustable zeros imply high selective filtering characteristics, and the adjustable range is relatively large, which is rare in dual-mode filters. This also indicates that the method of this invention has greater practical engineering applicability.

[0081] Example 4

[0082] Based on the dual-mode slow-wave patch resonator provided in the above embodiments, this invention provides a bandpass filter based on the dual-mode slow-wave patch resonator, which is a second-order filter with two finite-frequency transmission zeros.

[0083] like Figure 6 As shown, this miniaturized, highly selective single-cavity dual-mode filter is constructed using the aforementioned "cross-shaped" slotted dual-mode patch resonator. It consists of a dual-mode SW patch resonator, an input port feed line, and an output port feed line.

[0084] Specifically, both port feeds are located on the first metal layer and connected to the SW patch resonant cavity. The two port feeds are positioned on a set of adjacent edges of the first metal layer, at a distance D from the center point of each edge; the bandpass filter as a whole is axially symmetric about one diagonal of the first metal layer. The position of the transmission zero is adjusted by changing the value of D. Preferably, when D=0, i.e., when the two feeds are located at the center of a set of adjacent edges, the TM is most effectively excited simultaneously. 100 and TM 010 Mode resonance yields the best results.

[0085] In this embodiment, each layer structure is square in shape with a side length of L1, and uses L... p1 and L p2 The length of the two slot lines is represented by W0, and the width of the port feeder is represented by W0.

[0086] Furthermore, small slots extending into the patch are etched on both sides of each port feeder. The length of the slot is represented by g1, allowing for adjustment of bandwidth and the position of the finite frequency transmission zero within a certain range by controlling the size of the slots while keeping the feeder fixed.

[0087] By adjusting the distribution and size of the slow-wave layer metal vias and patches, the required resonant frequency (lower than the original cavity cutoff frequency) can be obtained. "Cross-shaped" slot lines can segment degenerate modes, transforming them into non-degenerate modes™ suitable for BPF design. 100 and TM 010 (The electric field distribution diagram of the non-degenerate mode after cutting is shown in the figure below) Figure 11 As shown, the non-degenerate two-mode electric field distribution of this type of filter is symmetrically distributed along the slot line (the two modes are diagonally symmetrical), and the slot line can further reduce the TM. 100 and TM 010 The resonant frequency of the mode can be further miniaturized.

[0088] Based on the distribution of non-degenerate modes, a reasonable feed structure can simultaneously excite both modes to resonate at similar frequencies, thus obtaining the desired single-cavity dual-mode filter. To facilitate easier cascading or integration with other devices, a microstrip line structure is adopted for the feed structure.

[0089] Preferably, W0=2.25, L1=19, Lp1=24, Lp2=19.8, d1=0.8, d2=1.7, d3=0.9, g1=1, all units: mm.

[0090] The parameters of the dielectric substrates used are as follows: the relative permittivity of the first dielectric substrate is 2.2 and the thickness is 0.254 mm; the relative permittivity of the second dielectric substrate is 2.2 and the thickness is 0.508 mm; and the thickness of all metal layers is 18 μm.

[0091] Ultimately, without changing the patch cavity size, the TM originally resonating at 5.32GHz was... 100 and TM 010 The mode was improved to reduce resonance near 2.40 GHz, and a highly selective second-order filter with two finite frequency transmission zeros was successfully designed.

[0092] Based on the preferred dimensions described above, simulation tests were conducted using electromagnetic simulation software. Figure 14 These are the simulation results of the bandpass filter frequency scattering parameters (S-parameters) in this embodiment. The measured center frequency... f The in-band insertion loss, return loss (RL), and 3-dB bandwidth (BW) are 2.4 GHz, -0.41 dB, -23.8 dB, and 500 MHz, respectively. The frequency locations of the finite frequency transmission zeros are 1.82 GHz and 2.89 GHz, respectively.

[0093] Compared to second-order filters with "cross-shaped" slot lines, this filter has lower insertion loss and a wider bandwidth. This is because the non-degenerate mode electromagnetic field distribution caused by the "cross-shaped" slot lines is more easily excited by the feed line at the center, and the excitation resonance intensity is stronger. At the same time, the cross-shaped slot lines are set along the diagonal of the patch, which allows for a lower resonant frequency within the same cavity size, which is more conducive to miniaturization and a smaller overall size.

[0094] To demonstrate the high selectivity of this filter, the frequency response was simulated using software to measure variations in size parameters, such as... Figure 15 and Figure 16 As shown. By Figure 15 It can be seen that when D changes, its bandwidth and transmission zero point also change. From... Figure 16 It can be seen that when g1 changes, its bandwidth and transmission zero point change accordingly. The above data also directly demonstrates that the BPF designed in this embodiment has highly selective filtering characteristics.

[0095] Example 5

[0096] Based on the dual-mode slow-wave patch resonator provided in the above embodiments, this invention provides a bandpass filter based on a dual-mode slow-wave patch resonator, which is a fourth-order BPF with four finite-frequency transmission zeros.

[0097] like Figure 7As shown, this miniaturized, highly selective fourth-order BPF is constructed using a folded stepped impedance coupling line directly connecting two "cross-shaped" slot lines in a dual-mode patch resonator. It consists of two identical "cross-shaped" slot lines in a dual-mode SW patch resonator, a folded stepped impedance coupling line, and two port feed lines.

[0098] Specifically, the input port feed line, the output port feed line, and the folded stepped impedance coupling line are all located in the first metal layer, such as... Figure 7 As shown, a port feed line is provided at a non-central position on the first side of the first metal layer of each of the two dual-mode patch resonators; the folded stepped impedance coupling line is connected at a non-central position on the second side of the first metal layer of each of the two dual-mode patch resonators; wherein the first side and the second side are opposite sides of each other. The entire fourth-order BPF is centrally symmetrical about the center of the folded stepped impedance coupling line.

[0099] Furthermore, the folded stepped impedance coupling line mainly comprises three components: a first stepped segment, a straight segment, and a second stepped segment. One end of the straight segment connects to the first stepped segment, and the other end connects to the second stepped segment. The folded stepped impedance coupling line is centrally symmetrical about the center of the straight segment. The first and second stepped segments have identical structures, both being two-stage stepped structures. One end of the first stepped segment connects to one of the patch resonators, and the other end connects to the straight segment; one end of the second stepped segment connects to another patch resonator, and the other end connects to the straight segment. In this embodiment, DL represents the distance between two adjacent sides of the two patch resonators, DW1 represents the feed line width at the end of the first or second stepped segment connected to the patch resonator, and DW2 represents the width of the straight segment. The meanings of other parameters can be found in Embodiment 3, and will not be repeated here. Using this folded stepped impedance coupling line for cascading can effectively maintain the high performance of a single-cavity dual-mode second-order filter and achieve high-selectivity superposition without sacrificing too much size. This folded stepped impedance coupling line effectively increases the coupling strength between the two cavities and resolves the contradiction between the coupling line length and the coupling strength between the two cavities, achieving both impedance matching and enhanced coupling. The BPF obtained based on this cascaded method possesses advantages such as high selectivity, high performance, and high power capacity, while also exhibiting unique advantages, particularly in miniaturization.

[0100] It should be noted that the position of the port feeder and the position of the folded stepped impedance coupling line are basically consistent with the pattern of the second-order BPF in the above embodiment 3.

[0101] The bandwidth and the frequency position of the finite-frequency transmission zeros can be adjusted by modifying the position of the port feeder and the folded stepped impedance coupling line. Using a centrally symmetrical cascaded dual-cavity layout, compared to ordinary translational cascading, four distinct finite-frequency transmission zeros can be obtained.

[0102] Preferred values ​​are: W0=2.25, L1=21, Lp1=19.38, Lp2=17.45, d1=0.8, d2=2.26, d3=0.49, D=5.25, DW1=2.25, DW2=1.24, DL=5.5. All units are in mm.

[0103] The parameters of the dielectric substrates used are as follows: the relative permittivity of the first dielectric substrate is 2.2 and the thickness is 0.254 mm; the relative permittivity of the second dielectric substrate is 2.2 and the thickness is 0.508 mm; and the thickness of all metal layers is 18 μm.

[0104] Based on the preferred dimensions described above, simulation was performed using simulation software. Figure 17 These are the simulation results of the bandpass filter frequency scattering parameters (S-parameters) in this embodiment. The measured center frequency... f The in-band insertion loss, return loss (RL), and 3-dB bandwidth (BW) are 2.4 GHz, -0.77 dB, -17.2 dB, and 340 MHz, respectively. The frequency locations of the finite frequency transmission zeros are 2.08 GHz, 2.17 GHz, 2.97 GHz, and 3.18 GHz, respectively.

[0105] Compared to cascaded dual-cavity single-mode patch filters, the overall size is reduced by nearly three-quarters, while also offering a wider range of bandwidth adjustability. Furthermore, fine-tuning the dimensions of the folded stepped impedance coupling line allows for further fine-tuning of the bandwidth and the frequency position of the finite-frequency transmission zero. Controllable bandwidth and adjustable finite-frequency transmission zero position demonstrate its high selectivity.

[0106] Example 6

[0107] Based on the dual-mode slow-wave patch resonator provided in the above embodiments, this invention provides a bandpass filter based on a dual-mode slow-wave patch resonator, which is a fourth-order BPF with four finite-frequency transmission zeros.

[0108] like Figure 8 As shown, this miniaturized, highly selective fourth-order BPF is constructed using a folded stepped impedance coupling line directly connecting two "cross-shaped" slot lines in a dual-mode patch resonator. It consists of two identical "cross-shaped" slot lines in a dual-mode SW patch resonator, a folded stepped impedance coupling line, and two port feed lines.

[0109] Specifically, the input port feed line, the output feed line port, and the folded stepped impedance coupling line are all located in the first metal layer, such as... Figure 8As shown, a port feed line is set at a distance D from the center point of the first metal layer of each of the two dual-mode patch resonators; the folded stepped impedance coupling line is connected at a distance D from the center point of the second metal layer of each of the two dual-mode patch resonators; wherein the first and second sides are adjacent to each other. The entire fourth-order BPF is centrally symmetrical about the center of the folded stepped impedance coupling line. In this embodiment, D=0; it can be understood that, similar to embodiment 4 above, the zero-point position and bandwidth can be adjusted by adjusting the value of D.

[0110] Furthermore, small slots extending into the patch are etched on both sides of each port feed line. The length of the slot is represented by g1.

[0111] Furthermore, the structure of the folded stepped impedance coupling line can be referred to in Embodiment 5 above, and will not be repeated here.

[0112] By fine-tuning the dimensions of the slotted wire and the folded stepped impedance coupling wire, the bandwidth and the frequency position of the finite-frequency transmission zeros can be adjusted. Similarly, by employing a centrally symmetrical layout and cascading dual cavities, compared to the ordinary translational cascading method, the method of this invention can obtain four distinct finite-frequency transmission zeros.

[0113] Preferred values ​​are: W0=2.25, L1=19, Lp1=20.46, Lp2=24, d1=0.8, d2=1.59, d3=1, g1=1, DW1=2.25, DW2=1.2, DL=6.5. All units are in mm.

[0114] The parameters of the dielectric substrates used are as follows: the relative permittivity of the first dielectric substrate is 2.2 and the thickness is 0.254 mm; the relative permittivity of the second dielectric substrate is 2.2 and the thickness is 0.508 mm; and the thickness of all metal layers is 18 μm.

[0115] The preferred dimensions described above were then simulated using simulation software. Figure 18 These are the simulation results of the bandpass filter frequency scattering parameters (S-parameters) in this embodiment. The measured center frequency... f The in-band insertion loss, return loss (RL), and 3-dB bandwidth (BW) are 2.4 GHz, -0.59 dB, -20.3 dB, and 430 MHz, respectively. The frequency locations of the finite frequency transmission zeros are 1.76 GHz, 1.94 GHz, 2.77 GHz, and 2.91 GHz.

[0116] Compared to dual-cavity cascaded single-mode patch filters, the overall size is reduced by nearly three-quarters. Compared to the same type of "cross-shaped" BPF, its overall size and layout size are smaller, making it more suitable for miniaturization. Similarly, the dimensions of the fine-tuning folded stepped impedance coupling line and the slot line can be adjusted within a certain range to control the bandwidth and the frequency position of the finite frequency transmission zero.

[0117] Example 7

[0118] Based on the dual-mode slow-wave patch resonator provided in the above embodiments, this invention provides a bandpass filter based on a dual-mode slow-wave patch resonator, which is a fourth-order BPF with four finite-frequency transmission zeros.

[0119] like Figure 9 As shown, this miniaturized, highly selective fourth-order BPF is constructed using a folded stepped impedance coupling line directly connected to a dual-mode patch resonator with a "cross-shaped" slot and a dual-mode patch resonator with a "cross-shaped" slot. It consists of a dual-mode SW patch resonator with a "cross-shaped" slot, a dual-mode patch resonator with a "cross-shaped" slot, a folded stepped impedance coupling line, and two port feed lines.

[0120] Specifically, the input port feed line, the output port feed line, and the folded stepped impedance coupling line are all located in the first metal layer, such as... Figure 9 As shown. A port feed line is set at a non-central position on the first side of the first metal layer of the dual-mode SW patch resonator with a "cross-shaped" slot. One end of the folded stepped impedance coupling line is connected at a non-central position on the second side of the first metal layer of the dual-mode SW patch resonator with a "cross-shaped" slot. The first and second sides are opposite to each other. A port feed line is set at a position D away from the center point of the third side of the first metal layer of the dual-mode patch resonator with a "cross-shaped" slot. The other end of the folded stepped impedance coupling line is connected at a position D away from the center point of the fourth side of the first metal layer of the dual-mode patch resonator with a "cross-shaped" slot. The third and fourth sides are adjacent to each other. Without considering the slot type in the first metal layer, the overall layout of the two dual-mode patch resonators is centrally symmetrical about the center of the folded stepped impedance coupling line. In this embodiment, D=0; it can be understood that, similar to Embodiment 4 above, the zero-point position and bandwidth can be adjusted by adjusting the value of D.

[0121] The positions of the feed lines and the folded stepped coupling lines are basically consistent with the patterns of the "cross-shaped" SW patch BPF in Embodiment 5 and the "cross-shaped" SW patch BPF in Embodiment 6.

[0122] The bandwidth and frequency position of the finite-frequency transmission zeros are adjusted by modifying relevant dimensions. A centrally symmetrical cascaded dual-cavity layout, compared to ordinary translational cascading, yields four distinct finite-frequency transmission zeros. Furthermore, the combination of two types of patch resonators effectively combines the high-performance characteristics of two dual-cavity systems, resulting in a high-order BPF with high selectivity, strong coupling, and miniaturization.

[0123] Preferably, W0=2.25, L=19, L1=21, Lp1=20.45, Lp2=24, Lp3=17.4, Lp4=19.35, d1=0.8, d2=1.65, d3=1, d ， 1=0.8, d ， 2 = 2.28, d ， =0.3, DW1=2.25, DW2=1.3, DL=7.1 (All units: mm). Among them, "d1=0.8, d2=1.65, d3=1" are the slow-wave layer structure parameters of the resonant cavity with "cross-type" slot lines; "d ， 1=0.8, d ， 2 = 2.28, d ， "3=0.3" represents the slow-wave layer structure parameter of the resonant cavity with a "cross-shaped" groove.

[0124] The parameters of the dielectric substrates used are as follows: the relative permittivity of the first dielectric substrate is 2.2 and the thickness is 0.254 mm; the relative permittivity of the second dielectric substrate is 2.2 and the thickness is 0.508 mm; and the thickness of all metal layers is 18 μm.

[0125] The preferred dimensions described above were then simulated using simulation software. Figure 19 These are simulation results of the bandpass filter frequency scattering parameters (S-parameters) in the above embodiments. The measured center frequency... f The in-band insertion loss, return loss (RL), and 3-dB bandwidth (BW) are 2.4 GHz, -0.74 dB, -15 dB, and 390 MHz, respectively. The frequency locations of the finite frequency transmission zeros are 1.89 GHz, 2.07 GHz, 2.84 GHz, and 3.03 GHz, respectively.

[0126] By utilizing the two types of single-cavity dual-mode BPFs proposed in this invention, and directly connecting them via folded stepped impedance coupling lines, a highly selective fourth-order BPF can be obtained. Specifically, the bandwidth and finite-frequency transmission zero can be adjusted by changing the position of the feed line connected to the "cross-shaped" resonant cavity unit, achieving controllable and adjustable bandwidth and finite-frequency transmission zero over a wide range. Simultaneously, the strong coupling of the "cross-shaped" single-cavity dual-mode second-order filter effectively broadens the passband bandwidth, providing more design options for engineering applications.

[0127] In this embodiment, experimental data demonstrating adjustable zero-point frequency position and controllable bandwidth for finite frequency transmission also proves its high selectivity. Furthermore, by employing this type of hybrid-mode cascaded high-order filter, it is possible to obtain the excellent performance and high selectivity of two types of single-cavity dual-mode second-order filters.

[0128] It should be noted that by employing a patch resonator based on a slow-wave structure and folded stepped impedance coupling lines, and utilizing non-degenerate modes, an Nth-order highly selective dual-mode bandpass filter with N finite-frequency transmission zeros can be constructed, where N = 2, 4, 6, 8, 10, etc. Furthermore, the resonant mode and / or the cascaded layout of the folded stepped impedance coupling lines can be changed according to actual engineering needs.

[0129] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, and not to limit them; although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features; and these modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A design method for a slow-wave patch resonant cavity, characterized in that, include: The structure consists of five layers stacked sequentially: a first metal layer, a first dielectric substrate, a patch layer composed of a metal patch array, a second dielectric substrate, and a second metal layer. Each layer is rectangular in shape. Two grooves are etched on the first metal layer and are arranged in an intersecting manner. A metallized via array is formed in the second dielectric substrate, and the metallized via array is used to connect the patch layer and the second metal layer, thereby forming a slow-wave patch resonant cavity between the 5 layers; the patch layer and the second dielectric substrate layer are collectively referred to as the slow-wave layer; by adjusting the distribution or size of the patches and metallized vias in the slow-wave layer, the tunability of the slow-wave patch resonant cavity is achieved without changing the overall size of the resonant cavity, effectively realizing miniaturization; the cutoff frequency of the resonant cavity is adjusted by adjusting the height of the second dielectric substrate layer.

2. The design method of a slow-wave patch resonator according to claim 1, characterized in that, The two groove lines are parallel to the two adjacent sides of the first metal layer, respectively.

3. The design method of a slow-wave patch resonator according to claim 1, characterized in that, The two groove lines are parallel to the two diagonals of the first metal layer, respectively.

4. A slow-wave patch resonant cavity, characterized in that, It is obtained by using the design method described in any one of claims 1 to 3.

5. A bandpass filter based on a slow-wave patch resonant cavity, characterized in that, include: A resonant cavity unit and two port feed lines; the resonant cavity unit is a slow-wave patch resonant cavity formed using the design method described in claim 2. The two port feed lines are respectively located at non-central positions on a pair of opposite sides of the first metal layer to connect to the resonant cavity.

6. A bandpass filter based on a slow-wave patch resonant cavity, characterized in that, include: A resonant cavity unit and two port feed lines; the resonant cavity unit is a slow-wave patch resonant cavity formed by the design method described in claim 3; the two port feed lines are respectively located on a set of adjacent sides of the first metal layer and at a distance D from the center point of the side to connect to the resonant cavity; wherein, |D|≥0.

7. A bandpass filter based on a slow-wave patch resonant cavity, characterized in that, include: Two identical resonant cavity units, folded stepped impedance coupling lines, and two port feed lines; the resonant cavity units are slow-wave patch resonant cavities formed using the design method described in claim 2. A port feed line is provided at the off-center position of the first side of the first metal layer of the two resonant cavity units; the folded stepped impedance coupling line is connected at the off-center position of the second side of the first metal layer of the two resonant cavity units respectively; wherein the first side and the second side are opposite sides of each other; The bandpass filter as a whole is centrally symmetrical about the center of the folded stepped impedance coupling line.

8. A bandpass filter based on a slow-wave patch resonant cavity, characterized in that, include: Two identical resonant cavity units, folded stepped impedance coupling lines, and two port feed lines; the resonant cavity units are slow-wave patch resonant cavities formed using the design method described in claim 3. A port feeder is provided on the first side of the first metal layer of the two resonant cavity units at a distance D from the center point of the side; the folded stepped impedance coupling line is connected on the second side of the first metal layer of the two resonant cavity units at a distance D from the center point of the side; wherein the first side and the second side are adjacent to each other; wherein |D|≥0. The bandpass filter as a whole is centrally symmetrical about the center of the folded stepped impedance coupling line.

9. A bandpass filter based on a slow-wave patch resonant cavity, characterized in that, include: The system comprises a first resonant cavity unit, a second resonant cavity unit, a folded stepped impedance coupling line, and two port feed lines; the first resonant cavity unit is a slow-wave patch resonant cavity formed by the design method described in claim 2, and the second resonant cavity unit is a slow-wave patch resonant cavity formed by the design method described in claim 3. A port feeder is provided at a non-central position on the first side of the first metal layer of the first resonant cavity unit, and a port feeder is provided at a position D on the third side of the first metal layer of the second resonant cavity unit, at a distance D from the center point of the side. One end of the folded stepped impedance coupling line is connected at a non-central position on the second side of the first metal layer of the first resonant cavity unit, and the other end of the folded stepped impedance coupling line is connected at a position D on the fourth side of the first metal layer of the second resonant cavity unit, at a distance D from the center point of the side. Wherein, the first side and the second side are opposite sides, and the third side and the fourth side are adjacent sides. Wherein, |D|≥0.

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