A beam-level compensation method for IQ imbalance in zero-IF receiving array

Abstract

The present invention discloses a zero-IF receiving array IQ imbalance beam-level compensation method, comprising: obtaining a radio frequency signal received by each array element; obtaining M intermediate frequency analog signals; obtaining M baseband analog in-phase signals and M baseband analog quadrature signals; performing A / D sampling on the M baseband analog in-phase signals and the M baseband analog quadrature signals respectively to obtain M baseband digital complex signals; obtaining M receiving channel amplitude and phase error compensation values; performing channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and summing the values to obtain a beamformed output complex signal; and utilizing the circular characteristics of the received baseband digital complex signals to estimate and compensate for the IQ imbalance error after beamforming to obtain a final baseband digital complex signal of the digital array. The present invention has the advantages of providing a multi-channel IQ imbalance compensation method that does not require channel-by-channel estimation and compensation for IQ imbalance errors and protects the circular characteristics of communication signals from being destroyed.

Description

Technical Field

[0001] The present invention relates to the field of IQ imbalance compensation, and in particular to a zero intermediate frequency receiving array IQ imbalance beam-level compensation method. Background Art

[0002] With the continuous advancement of digital signal processing technology and the corresponding continuous improvement of processing capabilities, digital arrays, with their advantages such as multiple scanning beams and high design flexibility, have gradually replaced analog array antennas and become a major research direction in electronic information technology fields such as communications, countermeasures, and radar. Digital arrays typically use superheterodyne IF sampling receivers (including multi-stage analog downconversion and filtering, A / D IF sampling, and digital downconversion) to convert the received RF analog signals of each array antenna into baseband digital signals.

[0003] Compared to superheterodyne IF sampling receivers, zero-IF receivers offer advantages such as simple circuit structure, low power consumption, ease of integration, compact size, and low cost. Digital arrays constructed using multi-channel zero-IF receivers offer low cost, low power consumption, and high integration, making them a key development direction for digital arrays. In practical applications, due to current device process limitations, the orthogonal local oscillator frequency sources used in the in-phase (I branch) and quadrature (Q branch) branches of zero-IF receiver chips cannot guarantee absolute orthogonality. Furthermore, the amplitude-frequency responses of analog components such as mixers and low-pass filters in each branch cannot be completely consistent. This results in amplitude and phase errors (commonly referred to as IQ imbalance) between the in-phase and quadrature branches. Excessive IQ imbalance can severely distort the signals received by each channel of the digital array, significantly impacting the overall performance of the receiver array. Therefore, research on IQ imbalance compensation methods for zero-IF receiver arrays has significant practical value.

[0004] Currently, there are numerous research papers on IQ imbalance compensation methods for single-channel zero-IF receivers, such as Chinese Patent Publication No. CN115833957A, which discloses a method for correcting IQ imbalance in a zero-IF receiver. However, there is little discussion of IQ imbalance compensation methods for multi-channel zero-IF receivers in digital arrays. If single-channel IQ imbalance image suppression methods are directly applied to digital arrays, the IQ imbalance errors of all channels must be estimated and compensated separately before beamforming and other processing is performed. This results in very high system calibration complexity and computational effort. Furthermore, in the field of cooperative communications, there are a class of communication signals with modulation types such as QPSK and 8PSK. These signals exhibit circular properties (the real and imaginary parts of the complex envelope have equal and statistically independent energies) and are widely used in various communication scenarios, such as satellite data transmission. If these circular communication signals are received by a digital array with IQ imbalance errors, their circular properties will be destroyed.

[0005] In order to reduce the complexity of digital array channel calibration and avoid distortion of communication signals with circular characteristics, it is necessary to find an IQ imbalance beam-level compensation method that can fully utilize the circular characteristics of the signal. This method can protect the circular characteristics of the communication signal as much as possible without the need to estimate and compensate the IQ imbalance error channel by channel, thus solving the IQ imbalance compensation problem in zero-IF receiving arrays. Summary of the Invention

[0006] The technical problem to be solved by the present invention is to provide a multi-channel IQ imbalance compensation method that does not require channel-by-channel estimation and compensation of IQ imbalance errors, and fully utilizes the circular characteristics of signals to protect the circular characteristics of communication signals from being destroyed.

[0007] The present invention solves the above technical problems through the following technical means: a zero intermediate frequency receiving array IQ imbalance beam-level compensation method, comprising the following steps:

[0008] Step 1: The digital array has M array elements, and the RF signal received by each array element is obtained;

[0009] Step 2: Filter and amplify the RF signal to obtain M intermediate frequency analog signals;

[0010] Step 3: Mix the M intermediate frequency analog signals with the M in-phase local oscillator signals and the M quadrature local oscillator signals and perform low-pass filtering to obtain M baseband analog in-phase signals and M baseband analog quadrature signals.

[0011] Step 4: A / D sampling is performed on the M baseband analog in-phase signals and the M baseband analog quadrature signals to obtain M baseband digital complex signals;

[0012] Step 5: Before beamforming, the digital array calibrates the amplitude and phase consistency of the M receiving channels to obtain amplitude and phase error compensation values for the M receiving channels.

[0013] Step 6: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the beamforming output complex signal;

[0014] Step 7: Utilize the circular characteristics of the received baseband digital complex signal to estimate and compensate for the IQ imbalance error after beamforming, and obtain the final baseband digital complex signal of the digital array.

[0015] Furthermore, the step 1 includes:

[0016] The RF signal received by the mth array element is

[0017] x m (t)=A(t)cos[Ω c (t-τ m)+φ(t)]m=1,…,M

[0018] Among them, A(t) is the amplitude modulation information, φ(t) is the phase modulation information, Ω c is the carrier analog angular frequency of the received signal, τ m It is the time delay difference determined by the array element coordinates and the signal incident direction, t is the current time, and M is the total number of array elements.

[0019] Furthermore, the step 2 includes:

[0020] The intermediate frequency analog signal obtained after filtering and amplification of the mth array element is:

[0021]

[0022] Among them, A cm and The amplitude and phase errors are caused by the preselection filter and low noise amplifier of each channel respectively.

[0023] Furthermore, the step 3 includes:

[0024] The baseband analog in-phase signal I of the mth channel m (t) and the baseband analog quadrature signal Q of the mth channel m (t) are

[0025]

[0026]

[0027] in, and They are respectively composed of the low-pass filter of the in-phase branch in the mth channel Amplitude and phase errors caused by and They are respectively the local oscillator frequency sources of the in-phase branch in the mth channel Amplitude and phase errors caused by and They are respectively composed of the low-pass filters of the orthogonal branches in the mth channel Amplitude and phase errors caused by and They are the local oscillator frequency sources of the orthogonal branches in the mth channel Amplitude and phase errors are caused.

[0028] Furthermore, the step 4 includes:

[0029] The baseband digital complex signal z of the mth channel m (n) is

[0030]

[0031] in, T s is the A / D sampling period, n is nT s The index of the sample value at the moment.

[0032] Furthermore, the step 5 includes:

[0033] Step 5-1: Place the single-frequency correction source in the normal direction of the digital array, calculate the single-frequency signal radiated by the correction source, and based on the influence of the single-frequency signal radiated by the correction source, collect the baseband digital complex signals of the M channels through each channel of the zero-IF receiving array to obtain the baseband digital complex signals of the M receiving channels. N samples of;

[0034] Step 5-2: For the mth channel, first calculate the fast Fourier transform of N samples, and then extract the d The spectrum value Sp at m And the corresponding amplitude value Ap m ;

[0035] Step 5-3: For the M channel amplitude values Sort from small to large, select the channel number r∈[1,M] corresponding to the median amplitude after sorting, select the rth channel as the reference channel, and select the rth channel amplitude value Ap r As the average value of normal channel judgment;

[0036] Step 5-4: Given the threshold A for normal channel judgment thr , calculate the upper limit of the amplitude of the normal channel judgment Ap U and the lower limit Ap L , generate M channel normal identification CFg m ;

[0037] Step 5-5, take the spectrum value Sp of the rth channel r For reference, the formula Calculate the amplitude and phase error correction of each receiving channel relative to the rth channel The superscript * is a complex number conjugation operation;

[0038] Step 5-6: For the mth channel, the channel amplitude and phase error correction is rewritten as

[0039] Step 5-7: Calculate the amplitude and phase error correction values for M channels The maximum amplitude And with this maximum value through the formula Normalize the amplitude and phase error correction values of all channels to obtain the final channel amplitude and phase error compensation values

[0040] Furthermore, the step 5-1 includes:

[0041] The single frequency signal radiated by the correction source is s c (t)=A0cos[(Ω c +Ω d )t+φ0]

[0042] Among them, φ0 is the initial phase, A0 is the amplitude of the single frequency signal, Ω d is the frequency offset of the single-frequency signal, Ω d =2πk0 / (NT s )>0, k0 is a positive integer, and 0 <k0<N / 2,N=2 b is the number of samples collected, b is a positive integer;

[0043] Baseband digital complex signals of M receiving channels The N samples of

[0044]

[0045] Furthermore, the step 5-4 includes:

[0046] Given the threshold A for normal channel judgment thr , through the formula Calculate the upper limit of the amplitude of the normal channel judgment Ap U and the lower limit Ap L , and then by the formula Generate M channel normal identifications.

[0047] Furthermore, the step 6 includes:

[0048] The beamforming output complex signal z(n) is expressed as

[0049]

[0050] in, is the beam weighting coefficient of the mth array element, α m is the amplitude weighted value of the mth array element in the array, and its maximum value satisfies s(n) is the target baseband digital complex signal and λ1 and λ2 are the first coefficient and the second coefficient, respectively, expressed as

[0051]

[0052] Furthermore, the step 7 includes:

[0053] Step 7-1, by formula and formula Calculate the autocorrelation function value γ of the beamforming output complex signal z(n) s and complementary autocorrelation function c s , where L is the number of samples required to estimate the IQ imbalance error of the beamforming output;

[0054] Step 7-2: Use the autocorrelation function value γ s and complementary autocorrelation function c s Compensation coefficient w for IQ imbalance error opt Make an estimate, that is

[0055] Step 7-3: Use the compensation coefficient w for IQ imbalance error opt Compensate the beamforming output data to obtain the final baseband digital complex signal Right now

[0056] The advantages of the present invention are:

[0057] (1) The present invention provides a beam-level IQ imbalance estimation and compensation method. This method does not require channel-by-channel IQ imbalance error estimation and compensation. Instead, it only requires estimation and compensation of inter-channel amplitude and phase inconsistencies and IQ imbalance errors after beamforming. Furthermore, the method fully utilizes the circular characteristics of the received baseband digital complex signal to protect the circular characteristics of the communication signal from being destroyed. Therefore, the method is particularly suitable for situations where digital arrays process communication signals with circular characteristics.

[0058] (2) The zero-IF receiving channel calibration algorithm and the IQ imbalance estimation and compensation algorithm provided by the present invention have simple principles, small computational complexity, and are easy to implement in engineering.

[0059] (3) The method provided by the present invention is not limited by the array structure and is applicable to both planar digital arrays and conformal digital arrays. In addition, the method provided by the present invention is applicable not only to digital receiving arrays but also to digital transmitting arrays. BRIEF DESCRIPTION OF THE DRAWINGS

[0060] Figure 1 This is a flow chart of a method for compensating IQ imbalance beam level in a zero-IF receiving array disclosed in an embodiment of the present invention;

[0061] Figure 2 This is a schematic diagram of the estimation accuracy of the channel amplitude error when the signal-to-noise ratio is 15dB in a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention;

[0062] Figure 3This is a schematic diagram of the estimation accuracy of the channel phase error when the signal-to-noise ratio is 15dB in a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention;

[0063] Figure 4 The target and mirror beam patterns are shown when the IQ imbalance error is not compensated and the beam pointing angle is 20 degrees in a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention.

[0064] Figure 5 In a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention, when the beam pointing is 60 degrees, the IQ imbalance error is not compensated, and the target and mirror beam patterns are shown;

[0065] Figure 6 In a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention, when the beam pointing is 20 degrees, the IQ imbalance error is compensated, and the target and mirror beam patterns are obtained;

[0066] Figure 7 In a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention, when the beam pointing is 60 degrees, the IQ imbalance error is compensated, and the target and mirror beam patterns are obtained;

[0067] Figure 8 This is a curve showing how the image rejection ratio changes with the number of sampling points when the signal-to-noise ratio is 15dB in a beam-level compensation method for IQ imbalance in a zero-IF receiving array disclosed in an embodiment of the present invention;

[0068] Figure 9 A curve showing how the estimation accuracy of the IQ imbalance amplitude error varies with the number of sampling points in a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention;

[0069] Figure 10 This is a curve showing how the estimation accuracy of the IQ imbalance phase error varies with the number of sampling points in a zero-IF receiving array IQ imbalance beam-level compensation method disclosed in an embodiment of the present invention. DETAILED DESCRIPTION

[0070] To make the objectives, technical solutions, and advantages of the embodiments of the present invention more clear, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. All other embodiments obtained by ordinary technicians in this field based on the embodiments of the present invention without making any creative efforts shall fall within the scope of protection of the present invention.

[0071] like Figure 1As shown, the present invention provides a zero-IF receiving array IQ imbalance beam-level compensation method, comprising the following steps:

[0072] Step 1: The digital array has M array elements. Get the RF signal received by each array element. The specific process is as follows:

[0073] The RF signal received by the mth array element is

[0074] x m (t)=A(t)cos[Ω c (t-τ m )+φ(t)]m=1,…,M

[0075] Among them, A(t) is the amplitude modulation information, φ(t) is the phase modulation information, Ω c is the carrier analog angular frequency of the received signal, τ m It is the time delay difference determined by the array element coordinates and the signal incident direction.

[0076] Step 2: Filter and amplify the RF signal to obtain M intermediate frequency analog signals. The specific process is as follows:

[0077] M RF signals received by the digital array After the preselection filter and low noise amplifier, M filtered and amplified RF signals are obtained. Right now

[0078]

[0079] Among them, A cm and The amplitude and phase errors are caused by the preselection filter and low noise amplifier of each channel respectively.

[0080] Step 3: Mix the M intermediate frequency analog signals with the M in-phase local oscillator signals and the M quadrature local oscillator signals and perform low-pass filtering to obtain M baseband analog in-phase signals and M baseband analog quadrature signals. The specific process is as follows:

[0081] M intermediate frequency analog signals With M in-phase local oscillator signals and M orthogonal local oscillator signals Perform mixing and low-pass filtering respectively to obtain M baseband analog in-phase signals and M baseband analog quadrature signals Right now

[0082]

[0083]

[0084] in, and They are respectively composed of the low-pass filter of the in-phase branch in the mth channel Amplitude and phase errors are caused. and They are respectively the local oscillator frequency sources of the in-phase branch in the mth channel Amplitude and phase errors are caused. and They are respectively composed of the low-pass filters of the orthogonal branches in the mth channel Amplitude and phase errors are caused. and They are the local oscillator frequency sources of the orthogonal branches in the mth channel Amplitude and phase errors are caused.

[0085] Step 4: A / D sampling is performed on the M baseband analog in-phase signals and the M baseband analog quadrature signals to obtain M baseband digital complex signals. The specific process is as follows:

[0086] M baseband analog in-phase signals and M baseband analog quadrature signals Perform A / D sampling separately to obtain M baseband digital complex signals Right now

[0087]

[0088] in, T s is the A / D sampling period.

[0089] Step 5: Before beamforming, the digital array calibrates the amplitude and phase consistency of the M receiving channels to obtain the amplitude and phase error compensation values of the M receiving channels. The specific process is as follows:

[0090] Step 5-1: The single-frequency correction source is placed in the normal direction of the digital array to meet the array far-field condition. The single-frequency signal radiated by the correction source is

[0091] s c (t)=A0cos[(Ω c +Ω d )t+φ0]

[0092] Among them, φ0 is the initial phase, A0 is the amplitude of the single frequency signal, Ω d is the frequency offset of the single frequency signal, and Ω is required d =2πk0 / (NT s )>0, k0 is a positive integer, and 0 <k0<N / 2,N=2 bis the number of samples collected, and b is a positive integer greater than 0. After being collected by each channel of the zero-IF receiving array, the baseband digital complex signal of M receiving channels is obtained. There are N samples of

[0093]

[0094] Step 5-2: Correction samples in M channels The processing method is the same. For the mth channel, first calculate the fast Fourier transform (FFT) of N correction samples, that is,

[0095]

[0096] Then extract the frequency Ω d The spectrum value Sp at m And the corresponding amplitude value Ap m ,Right now

[0097]

[0098] Ap m =20log 10 (|Z cm (k0)|)

[0099] Step 5-3: Amplitude values of M channels Sort from small to large, select the channel number r∈[1,M] corresponding to the median amplitude after sorting, select the rth channel as the reference channel, and select the rth channel amplitude value Ap r As the average value of normal channel judgment.

[0100] Step 5-4: Given the threshold A for normal channel judgment thr , first calculate the upper limit of the amplitude of the normal channel judgment Ap U and the lower limit Ap L ,Right now

[0101] Ap U =Ap r +A thr

[0102] Ap L =Ap r -A thr

[0103] Then generate M channel normal identifications That is, the normal identification of the mth channel is

[0104]

[0105] Step 5-5: Take the spectrum value Sp of the rth channel rAs a reference, calculate the amplitude and phase error correction of each receiving channel relative to the rth channel Right now

[0106]

[0107] The superscript * indicates the complex number conjugation operation.

[0108] Step 5-6: Normal identification according to the channel Channel amplitude and phase error correction Processing is performed to eliminate the influence of abnormal channels, that is, for the mth channel, the channel amplitude and phase error correction amount is rewritten as

[0109]

[0110] Step 5-7: Calculate the amplitude and phase error corrections for M channels The maximum amplitude Right now

[0111]

[0112] The maximum value is used to normalize the amplitude and phase error correction values of all channels to obtain the final channel amplitude and phase error compensation value. Right now

[0113]

[0114] Step 6: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the beamforming output complex signal. The specific process is as follows:

[0115] For the M baseband digital complex signals of the digital array Perform channel error compensation and beam weight compensation, and sum them up to obtain the beamforming output complex signal z(n), that is,

[0116]

[0117] in, is the beam weight coefficient of the mth array element. m is the amplitude weighted value of the mth array element in the array, and its maximum value satisfies The target baseband digital complex signal s(n) can be expressed as

[0118]

[0119] The coefficients λ1 and λ2 are expressed as

[0120]

[0121] Step 7: Using the circular characteristics of the received baseband digital complex signal, estimate and compensate for the IQ imbalance error after beamforming to obtain the final baseband digital complex signal of the digital array. The specific process is as follows:

[0122] Step 7-1: Calculate the autocorrelation function value γ of the beamforming output signal z(n) s and complementary autocorrelation function c s ,Right now

[0123]

[0124]

[0125] Where L is the number of samples required to estimate the IQ imbalance error of the beamforming output.

[0126] Step 7-2: Using the autocorrelation function value γ s and complementary autocorrelation function c s Compensation coefficient w for IQ imbalance error opt Make an estimate, that is

[0127]

[0128] Step 7-3: Use the compensation coefficient w for IQ imbalance error opt Compensate the beamforming output data to obtain the final baseband digital complex signal Right now

[0129]

[0130] Based on the detailed technical solution of this invention, IQ imbalance error estimation and compensation can be performed on the zero-IF digital array beamforming output. The correctness of the beam-level compensation method for zero-IF digital array IQ imbalance was verified through three simulation scenarios.

[0131] In the simulation experiment, the system operates at a radio frequency of 8 GHz, a system sampling rate of 30 MHz, a uniform linear array with an element spacing of 11 mm and 64 elements, and the uniform linear array uses uniform weighting.

[0132] Scenario 1: Channel Amplitude and Phase Error Estimation Accuracy

[0133] For a zero-IF receiving array, the channel amplitude error A m The channel phase error is uniformly distributed within 0dB to 1dB. The sample is uniformly distributed within ±45 degrees. During channel calibration, the baseband digital complex signal samples of each receiving channel of the array are collected, and the signal-to-noise ratio of the sample is taken as 15dB. The number of collected samples is 10 0 to 106 After 1000 simulation experiments, the estimation accuracy of channel amplitude and phase error under different numbers of samples is as follows: Figure 2 and 3 shown.

[0134] As shown in the figure, when the number of sampling points is 100, the mean + 3 times the standard deviation of the channel amplitude error estimate is equal to 0.7284dB, and the mean + 3 times the standard deviation of the channel phase error estimate is equal to 3.18 degrees. When the number of sampling points is 1024, the mean + 3 times the standard deviation of the channel amplitude error estimate is better than 0.1dB, and the mean + 3 times the standard deviation of the channel phase error estimate is better than 1 degree. When the signal-to-noise ratio of the channel samples is 15dB, the number of samples collected for channel amplitude and phase error correction, N = 1024, is selected. The proposed zero-IF receive channel calibration algorithm has high amplitude and phase error estimation accuracy.

[0135] Scenario 2: Array beamforming patterns before and after IQ imbalance compensation

[0136] For a zero-IF receiving array, the channel amplitude error A m The channel phase error is uniformly distributed within 0dB to 1dB. The IQ imbalance amplitude error g is uniformly distributed within ±45 degrees. m The phase error θ is uniformly distributed between 0.5dB and 1.5dB. m The distribution is uniform between 0 and 10 degrees, so the image rejection ratio brought by the average value of the IQ imbalance amplitude and phase error is 22.8 dB. To obtain the beam pattern of the digital array, the target signal uses a single-frequency signal with a radio frequency of 8.001 GHz. The number of sampling points after beam synthesis is 3×10 6 , ensuring an integer number of single-frequency cycles within the sampling duration, satisfying the circular characteristic condition. The beamforming output signal-to-noise ratio is 30dB. The target signal is incident on the digital array from a 20-degree or 60-degree angle.

[0137] When the beam synthesis output does not perform IQ imbalance error estimation and compensation, the beam pointing to the target and mirror array at 20 degrees and 60 degrees respectively is as follows Figure 4 and 5 As shown in the figure, compared with the IQ imbalance image suppression level of a single channel, the beamforming of the digital array has a certain ability to suppress the image components caused by IQ imbalance, and the image suppression ratio after beamforming is about 32dB. When the array scale is small, the image suppression level brought by array beamforming is limited, and the IQ imbalance error needs to be further estimated and compensated after beamforming. After the IQ imbalance beam level compensation, the beam pointing to the target and mirror array with a beam angle of 20 degrees and 60 degrees respectively is as shown in the figure. Figure 6 and 7As shown in the figure, beam-level IQ imbalance compensation improves the image rejection ratio from approximately 32 dB before compensation to over 64 dB. The proposed beam-level IQ imbalance compensation method can further improve the image rejection ratio of the beamforming output.

[0138] Scenario 3: Beam-level IQ imbalance error compensation accuracy

[0139] For a zero-IF receiving array, the channel amplitude error A m The channel phase error is uniformly distributed within 0dB to 1dB. The amplitude error of the IQ imbalance of the beam synthesis output is g. m is 1dB, the phase error θ m The image rejection ratio of the IQ imbalance amplitude and phase error is 22.8dB. The array beamforming output signal-to-noise ratio is 15dB, the sampling frequency is 30MHz, the target signal is a single-frequency signal, and its RF frequency is 8.001GHz. The number of sampling points after beamforming is 3×10 2 to 3×10 6 The image rejection ratio curves corresponding to different sampling points are shown in Figure 1. Figure 8 The image rejection ratio (IRR) of the beamforming output (before compensation) is 22.8 dB. When the number of sampling points is 3×10 6 When the number of sampling points is small (for example, 3000), the mean - 3 times the standard deviation of the image rejection ratio is approximately 31 dB.

[0140] Figure 9 and Figure 10 The curves of the estimation accuracy of IQ imbalance amplitude error and phase error as the number of sampling points are given respectively. 6 When the number of sampling points is 3000, the amplitude error estimation accuracy is 0.0012dB, and the phase error estimation accuracy is 0.0080 degrees. When the number of sampling points is small (for example, 3000), the amplitude error estimation accuracy is 0.08dB, and the phase error estimation accuracy is 0.5 degrees. Therefore, to obtain an accurate estimate of the IQ imbalance error, a large amount of sample data must be collected. This makes IQ imbalance error estimation more computationally intensive than inter-channel amplitude and phase error estimation. If IQ imbalance error estimation is performed in every receive channel of the digital array, the computational complexity is enormous, which is not conducive to practical applications.

[0141] In summary, the beam-level IQ imbalance compensation method for zero-IF receive arrays is simple in principle, requires minimal computation, and is easily implemented. Its performance has been verified through simulation experiments. This method eliminates the need for channel-by-channel IQ imbalance estimation and compensation; instead, it only requires IQ imbalance estimation and compensation at the beamforming output. This reduces the computational complexity of error estimation and compensation for digital arrays and is particularly suitable for digital arrays processing communication signals with circular characteristics.

[0142] The above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit the same. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or make equivalent replacements for some of the technical features therein. However, these modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the various embodiments of the present invention.

Claims

1. A zero-IF receiving array IQ imbalance beam-level compensation method, characterized in that: The following steps are involved: Step 1: The digital array has M array elements, and the RF signal received by each array element is obtained; Step 2: Filter and amplify the RF signal to obtain M intermediate frequency analog signals; Step 3: Mix the M intermediate frequency analog signals with the M in-phase local oscillator signals and the M quadrature local oscillator signals and perform low-pass filtering to obtain M baseband analog in-phase signals and M baseband analog quadrature signals. Step 4: A / D sampling is performed on the M baseband analog in-phase signals and the M baseband analog quadrature signals to obtain M baseband digital complex signals; Step 5: Before beamforming, the digital array calibrates the amplitude and phase consistency of the M receiving channels to obtain amplitude and phase error compensation values for the M receiving channels. Step 6: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the beamforming output complex signal; Step 7: Utilize the circular characteristics of the received baseband digital complex signal to estimate and compensate for the IQ imbalance error after beamforming, and obtain the final baseband digital complex signal of the digital array.

2. The method for compensating IQ imbalance beam level in a zero-IF receiving array according to claim 1, wherein: The step 1 comprises: The RF signal received by the mth array element is x m (t)=A(t)cos[Ω c (t-t m )+φ(t)]m=1,…,M Among them, A(t) is the amplitude modulation information, φ(t) is the phase modulation information, Ω c is the carrier analog angular frequency of the received signal, τ m It is the time delay difference determined by the array element coordinates and the signal incident direction, t is the current time, and M is the total number of array elements.

3. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 2, wherein: The step 2 includes: The intermediate frequency analog signal obtained after filtering and amplification of the mth array element is: Among them, A cm and The amplitude and phase errors are caused by the preselection filter and low noise amplifier of each channel respectively.

4. The method for compensating IQ imbalance beam level in a zero-IF receiving array according to claim 3, wherein: The step 3 comprises: The baseband analog in-phase signal I of the mth channel m (t) and the baseband analog quadrature signal Q of the mth channel m (t) are in, and They are respectively composed of the low-pass filter of the in-phase branch in the mth channel Amplitude and phase errors caused by and They are respectively the local oscillator frequency sources of the in-phase branch in the mth channel Amplitude and phase errors caused by and They are respectively composed of the low-pass filters of the orthogonal branches in the mth channel Amplitude and phase errors caused by and They are the local oscillator frequency sources of the orthogonal branches in the mth channel Amplitude and phase errors are caused.

5. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 4, characterized in that: The step 4 comprises: The baseband digital complex signal z of the mth channel m (n) is in, T s is the A / D sampling period, n is nT s The index of the sample value at the moment.

6. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 5, characterized in that: The step 5 comprises: Step 5-1: Place the single-frequency correction source in the normal direction of the digital array, calculate the single-frequency signal radiated by the correction source, and based on the influence of the single-frequency signal radiated by the correction source, collect the baseband digital complex signals of the M channels through each channel of the zero-IF receiving array to obtain the baseband digital complex signals of the M receiving channels. N samples of; Step 5-2: For the mth channel, first calculate the fast Fourier transform of N samples, and then extract the d The spectrum value Sp at m And the corresponding amplitude value Ap m ; Step 5-3: For the M channel amplitude values Sort from small to large, select the channel number r∈[1,M] corresponding to the median amplitude after sorting, select the rth channel as the reference channel, and select the rth channel amplitude value Ap r As the average value of normal channel judgment; Step 5-4: Given the threshold A for normal channel judgment thr , calculate the upper limit of the amplitude of the normal channel judgment Ap U and the lower limit Ap L , generate M channel normal identification CFg m ; Step 5-5, take the spectrum value Sp of the rth channel r For reference, the formula Calculate the amplitude and phase error correction of each receiving channel relative to the rth channel The superscript * is a complex number conjugation operation; Step 5-6: For the mth channel, the channel amplitude and phase error correction is rewritten as Step 5-7: Calculate the amplitude and phase error correction values for M channels The maximum amplitude And with this maximum value through the formula m=1,…,M normalizes all channel amplitude and phase error correction values to obtain the final channel amplitude and phase error compensation value 7. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 6, wherein: The step 5-1 includes: The single frequency signal radiated by the correction source is s c (t)=A0cos[(Ω c +Ω d )t+φ0] Among them, φ0 is the initial phase, A0 is the amplitude of the single frequency signal, Ω d is the frequency offset of the single-frequency signal, Ω d =2πk0 / (NT s )>0, k0 is a positive integer, and 0 <k0<N / 2,N=2 b is the number of samples collected, b is a positive integer; Baseband digital complex signals of M receiving channels The N samples of 8. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 6, wherein: The step 5-4 includes: Given the threshold A for normal channel judgment thr , through the formula Calculate the upper limit of the amplitude of the normal channel judgment Ap U and the lower limit Ap L , and then by the formula Generate M channel normal identifications.

9. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 6, wherein: The step 6 comprises: The beamforming output complex signal z(n) is expressed as in, is the beam weighting coefficient of the mth array element, α m is the amplitude weighted value of the mth array element in the array, and its maximum value satisfies s(n) is the target baseband digital complex signal and λ1 and λ2 are the first coefficient and the second coefficient, respectively, expressed as 10. The method for compensating IQ imbalance beam level of a zero intermediate frequency receiving array according to claim 9, characterized in that: The step 7 comprises: Step 7-1, by formula and formula Calculate the autocorrelation function value γ of the beamforming output complex signal z(n) s and complementary autocorrelation function c s , where L is the number of samples required to estimate the IQ imbalance error of the beamforming output; Step 7-2: Use the autocorrelation function value γ s and complementary autocorrelation function c s Compensation coefficient w for IQ imbalance error opt Make an estimate, that is Step 7-3: Use the compensation coefficient w for IQ imbalance error opt Compensate the beamforming output data to obtain the final baseband digital complex signal Right now

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