All-digital phase noise measurement method with millihertz-level frequency resolution

Abstract

The application discloses a kind of full digital phase noise measurement methods of millihertz level frequency resolution, belong to phase noise measurement field, the method includes: the signal to be measured is converted into digital signal, and according to the frequency control word of the digital signal generation two-way phase quadrature and no phase truncation spur digital signal, after being mixed with digital signal respectively, carry out multi-section extraction filtering, phase unwrapping, frequency offset elimination, power spectrum estimation and segmented splicing processing.The method can measure the phase noise of signal using only one ADC, the principle is simple, easy to realize the miniaturization of system;By multi-section extraction filtering, while filtering high-frequency components, the data rate after mixing can be reduced, the number of operation points required for power spectrum estimation is reduced, and the phase noise measurement of signal extremely close millihertz level frequency resolution is realized.

Description

Technical Field

[0001] This invention belongs to the field of phase noise measurement, and more specifically, relates to a fully digital phase noise measurement method with millihertz-level frequency resolution. Background Technology

[0002] In certain special applications, there is a need for measurements of signals in the 10 kHz–500 kHz frequency band with millihertz-level frequency resolution. For example, in the Tianqin Project, several key technologies needed to be overcome to detect cosmic gravitational waves in the 0.001–0.1 Hz frequency range. Capacitive displacement sensors are a crucial component, and their sensing performance plays a significant role in the sensitivity of gravitational wave detection. In capacitive displacement sensors, a carrier wave is injected onto multiple capacitor plates surrounding a test mass (TM). If the frequency stability of the carrier wave used for modulation is not high, the low-frequency differential capacitance signal at the millihertz level may be masked by the phase noise of the carrier wave. Therefore, measuring the phase noise of the carrier wave at its very near end (millihertz frequency offset) is of significant research importance.

[0003] However, none of the phase noise instruments currently on the market can measure phase noise at the very near-end millihertz level for signals ranging from 10 kHz to 500 kHz. Summary of the Invention

[0004] In view of the above-mentioned defects or improvement needs of the existing technology, the present invention provides a fully digital phase noise measurement method with millihertz-level frequency resolution, thereby solving the technical problem that existing phase noise instruments cannot achieve phase noise measurement of 10k to 500kHz signals at the very near-end millihertz level.

[0005] To achieve the above objectives, according to a first aspect of the present invention, a fully digital phase noise measurement method with a frequency resolution at the millihertz level is provided, comprising:

[0006] S1, convert the signal under test into a digital signal, and obtain the frequency control word of the digital signal;

[0007] S2, based on the frequency control word, generate two digital signals V with orthogonal phases and no phase cutoff spurious signals. h1 V h2 V h1 V h2 V is obtained by mixing the digital signal with the digital signal respectively. h1 `、V h2 `;

[0008] S3, respectively for V h1 `、V h2 Perform multi-stage decimation filtering to obtain V at each decimation factor. h1 `、Vh2 The filtered signal; and at each decimation factor, V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence.

[0009] S4. Perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; stitch together the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve.

[0010] According to a second aspect of the present invention, a fully digital phase noise measurement device with a frequency resolution at the millihertz level is provided, comprising:

[0011] The first processing module is used to convert the signal under test into a digital signal and obtain the frequency control word of the digital signal;

[0012] The second processing module is used to generate two digital signals V with orthogonal phases and no phase truncation spurious signals, based on the frequency control word. h1 V h2 V h1 V h2 V is obtained by mixing the digital signal with the digital signal respectively. h1 `、V h2 `;

[0013] The third processing module is used to process V respectively. h1 `、V h2 Perform multi-stage decimation filtering to obtain V at each decimation factor. h1 `、V h2 The filtered signal; and at each decimation factor, V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence.

[0014] The fourth processing module is used to perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; and to splice the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve.

[0015] According to a third aspect of the present invention, a fully digital phase noise measurement system with a frequency resolution of millihertz is provided, comprising: a computer-readable storage medium and a processor;

[0016] The computer-readable storage medium is used to store executable instructions;

[0017] The processor is configured to read executable instructions stored in the computer-readable storage medium and execute the method as described in the first aspect.

[0018] According to a fourth aspect of the invention, a computer-readable storage medium is provided, the computer-readable storage medium storing computer instructions for causing a processor to perform the method as described in the first aspect.

[0019] In summary, compared with the prior art, the above-described technical solutions conceived by this invention can achieve the following beneficial effects:

[0020] 1. The method provided by this invention can measure the phase noise of a signal using only one ADC. The principle is simple, and the measurement system is easy to miniaturize, making it convenient for field testing. Through multi-segment decimation filtering, high-frequency components can be filtered out while reducing the data rate after mixing, reducing the number of computation points required for power spectrum estimation, thereby achieving phase noise measurement at the millihertz level frequency resolution of the signal at the very near end. For example, it can achieve the very near end phase noise measurement of signals in the 10k to 500kHz frequency band, making up for the inability of commercial instruments to achieve millihertz level frequency resolution for signals in this frequency band.

[0021] 2. The method provided by the present invention can generate two mutually orthogonal, phase-truncation-free spurious signals through low spurious orthogonal digital signal synthesis technology, which can reduce measurement interference caused by spurious signals.

[0022] 3. The method provided by this invention can reduce the influence of small frequency differences between the measured signal and the synthesized signal by using phase unwinding and frequency offset elimination algorithms. Attached Figure Description

[0023] Figure 1 A flowchart of a fully digital phase noise measurement process provided for an embodiment of the present invention.

[0024] Figure 2 This is a schematic diagram of the fully digital phase noise measurement principle provided in an embodiment of the present invention.

[0025] Figure 3 This is a schematic diagram of the principle of equal-precision frequency measurement.

[0026] Figure 4 This is a schematic diagram of a low-spurious-frequency orthogonal digital signal synthesis method.

[0027] Figure 5 This is a schematic diagram of the comparison of single-sideband phase noise spectral density of a 500kHz signal provided in an embodiment of the present invention.

[0028] Figure 6 This is a schematic diagram of single-sideband phase noise spectral density splicing of a 500kHz signal provided in an embodiment of the present invention. Detailed Implementation

[0029] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention. Furthermore, the technical features involved in the various embodiments of this invention described below can be combined with each other as long as they do not conflict with each other.

[0030] Among various phase noise measurement techniques, the direct spectrum analyzer method is affected by the inherent noise of the spectrum analyzer itself and is not suitable for measuring signals with high frequency stability; the beat method is difficult to achieve millihertz-level frequency resolution measurement at the very near end of the signal due to counter limitations; the frequency discrimination method has poor sensitivity when measuring near-end phase noise; the phase discrimination method has many physical components, a large phase discrimination system, and requires compensation for low-frequency loss phenomena; while the fully digital method has no of the above limitations and has the advantages of simple implementation principle, easy miniaturization of the system, and convenient field testing.

[0031] Based on this, embodiments of the present invention provide a fully digital phase noise measurement method with a frequency resolution at the millihertz level, such as... Figure 1 As shown, it includes:

[0032] S1, convert the signal under test into a digital signal, and obtain the frequency control word of the digital signal.

[0033] Specifically, the signal under test (e.g., a sine wave) is sampled by an ADC and converted into a digital signal. The frequency control word of the digital signal is obtained by an equal-precision frequency measurement method, a direct measurement method, or a periodic measurement method.

[0034] When using equal-precision frequency measurement, the frequency control word of the signal under test can be expressed by the count value of the standard signal and the signal under test.

[0035] S2, based on the frequency control word, generate two digital signals V with orthogonal phases and no phase cutoff spurious signals. h1 V h2 V h1 V h2 V is obtained by mixing the digital signal with the digital signal respectively. h1 `、V h2 `.

[0036] Preferably, in step S2, V is generated using a phase-truncation-free frequency synthesis method, a ROM table compression frequency synthesis method, or a CORDIC frequency synthesis method. h1 Vh2 .

[0037] Specifically, based on the frequency control word generated by equal-precision frequency measurement, the low-spurious orthogonal digital signal synthesis technology can generate two low-spurious digital signals with orthogonal phases and frequencies similar to the signal under test, which can reduce the interference of spurious signals on the phase noise measurement of the signal under test.

[0038] S3, respectively for V h1 `、V h2 Perform multi-stage decimation filtering to obtain V at each decimation factor. h1 `、V h2 The filtered signal; and at each decimation factor, V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence.

[0039] Specifically, the two signals synthesized from low-spurious quadrature digital signals are mixed with the signal under test in the digital domain. After multi-segment decimation filtering, high-frequency components after mixing can be filtered out, and the low-frequency signal after mixing can be obtained. In addition, multi-segment decimation can realize multi-resolution measurement, which can reduce the number of data points required to achieve millihertz-level frequency resolution.

[0040] S4. Perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; stitch together the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve.

[0041] Specifically, dividing the two signals after multiple decimation and filtering and performing arctangent in the four quadrants yields a preliminary phase difference sequence. By unwinding this phase difference sequence, the phase difference sequence's transitions at ±π can be recovered.

[0042] The digital signal generated by low spurious orthogonal digital signal synthesis technology has a small frequency difference with the measured signal. In order to eliminate the measurement error caused by this frequency difference, the frequency offset elimination of the unwound sequence needs to be performed.

[0043] Preferably, in step S4, the off-center carrier frequency range is segmented, and different sampling frequencies are selected for different off-center carrier frequency bands to achieve power spectrum estimation.

[0044] The following is combined Figure 2 The method provided by the present invention will be described in further detail.

[0045] like Figure 2As shown, the fully digital phase noise measurement method with a measurement frequency resolution at the millihertz level provided by the present invention mainly includes the following steps: ADC acquisition, equal-precision frequency measurement, low-spurious orthogonal digital signal synthesis, multi-segment decimation filtering, four-quadrant arctangent, phase unwinding, frequency offset elimination, power spectrum estimation, and segment splicing.

[0046] Step 1: Use an ADC to convert the signal under test into a digital signal;

[0047] Step 2: Use the equal-precision frequency measurement method to obtain the frequency control word of the digital signal, providing a variable frequency reference for the next stage of low-spurious quadrature digital signal synthesis;

[0048] Step 3: Based on the frequency control word provided in Step 2, use a low-spurious quadrature digital signal synthesis method to generate two digital signals with a phase difference of π / 2 and no phase truncation spurious signals. These two signals will be mixed with the signal under test in the digital domain.

[0049] Among them, the low spurious orthogonal digital signal synthesis technology in step three is a digital signal synthesis technology that can generate two mutually orthogonal, phase-truncation-free spurious signals.

[0050] Step 4: The mixed signal undergoes multi-stage decimation filtering to remove high-frequency components. Decimation also reduces the data rate after mixing, thus reducing the number of computation points required for power spectrum estimation in Step 8, thereby enabling phase noise measurement at the millihertz frequency resolution level.

[0051] Step 5: After dividing the data output from the upper and lower channels (i.e., I-channel and Q-channel), perform arctangent calculation in the four quadrants to obtain a preliminary phase difference sequence;

[0052] Step 6: Perform phase unwrapping on the preliminary phase difference sequence to recover the phase difference sequence jumps at ±π.

[0053] Step 7: Apply a frequency offset elimination algorithm to the phase difference sequence after phase unwrapping to eliminate the influence of small frequency differences, thereby obtaining a low-error phase difference sequence;

[0054] Among them, the phase unwinding and frequency offset elimination algorithms for the preliminary phase difference sequence described in steps six and seven work together to reduce the measurement error caused by small frequency differences.

[0055] Step 8: Perform power spectrum estimation on the phase difference sequence after frequency offset elimination to obtain the phase fluctuation spectral density, and then obtain the single-sideband phase noise spectral density of the measured signal. By splicing the single-sideband phase noise spectral density of each segment in the multi-segment decimation filter, a complete phase noise measurement curve can be obtained.

[0056] To balance the advantages and disadvantages of FFT point count and resolution (higher resolution requires more FFT points), a segmented approach is adopted, selecting different sampling frequencies (higher sampling multiples result in lower sampling rates) for different frequency bands to achieve power spectrum estimation. For example, for the very near-end (i.e., near the origin) frequency band requiring millihertz-level resolution (such as...),... Figure 5 The first frequency band shown has a high decimation factor, a low sampling rate, and fewer FFT points, making it easy to implement.

[0057] Understandably, during the multi-segment decimation filtering in step four, since there are multiple decimation factors, steps five through seven involve processing at each decimation factor. That is, after steps five through seven, the resulting sequence is the phase difference sequence after frequency offset elimination at each decimation factor. For example, when the decimation factors are 16, 64, 256, and 2048, step seven yields the corresponding phase difference sequences after frequency offset elimination for each decimation factor. Accordingly, step eight involves concatenating the single-sideband phase noise spectral density of the phase difference sequences after frequency offset elimination at each decimation factor.

[0058] Taking a sine wave as the measured signal as an example, the measured signal after ADC sampling is as follows:

[0059]

[0060] Where w0 = 2πf0. A0, w0, and f0 represent the ideal amplitude, angular frequency, and frequency of the signal, respectively, and ε(n) and These represent the random amplitude fluctuations and phase fluctuations of the signal, respectively.

[0061] The principle of equal precision frequency measurement is as follows: Figure 3 Two counters are used to count the measured signal and the standard signal respectively. When the first rising edge of the measured signal is detected after the preset gate is high, the actual gate is opened, and the two counters begin counting. After a period of time, the preset gate is pulled low. When the rising edge of the measured signal is detected, the actual gate is set to low, and the counters stop counting. The count values ​​of the standard signal and the measured signal are denoted as N0 and N, respectively. x .

[0062] The principle of low spurious quadrature digital signal synthesis is as follows: Figure 4 Where N is the bit width of the phase accumulator, and the frequency control word K is as shown in the formula.

[0063] K=2 N N x / N0 (2)

[0064] The reference frequency source uses an internal clock with a frequency of f.clk The generated signal frequency f x with f clk The relationship is:

[0065]

[0066] The two signals generated by the low spurious orthogonal digital signal synthesis technique are shown in equations 1 and 2, respectively.

[0067]

[0068]

[0069] Ideally, the amplitude, angular frequency, initial phase, and random phase fluctuations of the two signals satisfy A h1 =A h2 w h1 =w h2 ,

[0070] After mixing and decimation filtering, the high-frequency components in the mixing can be filtered out, resulting in I and Q signals as shown in equations 1 and 2.

[0071]

[0072]

[0073] By quoting Q(n) and I(n) and taking the arctangent of the fourth quadrant, we can obtain the preliminary phase difference sequence θ(n) between the measured signal and the two orthogonal signals, as shown in the equation.

[0074]

[0075] If the NCO generates two digital signals V h1 (n) and V h2 (n) has a small frequency difference with the measured signal, i.e., w0 ≠ w h2 Furthermore, since the four-quadrant arctangent restricts θ(n) to the range [-π, π], it will make θ(n) appear jagged. Directly estimating the power spectrum of θ(n) will distort the phase noise.

[0076] Before frequency offset elimination of θ(n), phase unwrapping of θ(n) is required. The specific steps are as follows:

[0077] 1) Starting from the second phase difference sequence, calculate the phase difference between the current data and the previous data;

[0078] 2) If the phase difference is greater than π or less than -π, then add or subtract 2π to that point and all subsequent data.

[0079] 3) Repeat step 2) until all data has been traversed.

[0080] The linear trend error after phase unwinding was fitted using a simple linear regression model.

[0081] Subtracting the fitted linear error from the unwrapped phase difference sequence yields the phase difference sequence after frequency shift elimination. Power spectrum estimation is then performed on this phase difference sequence, and the single-sideband phase noise spectral density is calculated.

[0082] The method provided in this invention is used to measure the phase noise of a 500kHz signal. In the multi-segment decimation filter, four decimation filters with decimation factors of 16, 64, 256, and 2048 are used. The number of computation points, frequency resolution, and splicing range for each segment are shown in Table 1. For each frequency band, data at different sampling frequencies (the sampling factor is inversely proportional to the sampling frequency, original sampling frequency / decimation factor = current sampling frequency) are selected to achieve power spectrum estimation. For example, for a frequency band with an off-carrier frequency range of 10kHz to 100kHz, data with a decimation factor of 16 is selected, and 2... 12 The power spectrum estimation is performed using the number of computational points; similarly, for the off-center carrier frequency range of 1 kHz to 10 kHz, data with a decimation factor of 64 are selected, and 2 11 The power spectrum estimation is performed using the number of computation points.

[0083] Table 1500kHz Signal Segment Extraction and Splicing Range

[0084]

[0085] Figure 5 This is the phase noise test result spliced ​​for this example. The curve marked "A" represents the measurement result after data processing using the algorithm of this invention, and the curve marked "B" represents the measurement result from a commercial instrument. Figure 5 It is known that, compared with commercial instruments, the present invention can measure phase noise in the range of [0.0006, 1] Hz, and the frequency resolution at the very near end is 0.0006 Hz. Figure 6 for Figure 5 The example is a stitching diagram, which includes the target phase difference sequence a. I a II a III a IVThe frequency bands are I, II, III, and IV; the decimation factors for frequency bands I, II, III, and IV are 2048, 256, 64, and 16, respectively. For the very near-end frequency band, i.e., frequency band I, to save computational resources, it is further divided into 5 segments according to the different number of FFT points. Then, the phase noise power spectral density is stitched together to obtain the phase noise power spectral density with an off-carrier frequency range of [0.0006, 100] Hz, and finally, the stitched image of the phase noise power spectral density with an off-carrier frequency range of [0.0006, 100 kHz] Hz is obtained.

[0086] This invention provides a fully digital phase noise measurement device with a frequency resolution at the millihertz level, comprising:

[0087] The first processing module is used to convert the signal under test into a digital signal and obtain the frequency control word of the digital signal;

[0088] The second processing module is used to generate two digital signals V with orthogonal phases and no phase truncation spurious signals, based on the frequency control word. h1 V h2 V h1 V h2 V is obtained by mixing the digital signal with the digital signal respectively. h1 `、V h2 `;

[0089] The third processing module is used to process V respectively. h1 `、V h2 Perform multi-stage decimation filtering to obtain V at each decimation factor. h1 `、V h2 The filtered signal; and at each decimation factor, V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence.

[0090] The fourth processing module is used to perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; and to splice the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve.

[0091] This invention provides a fully digital phase noise measurement system with a frequency resolution of millihertz, comprising: a computer-readable storage medium and a processor;

[0092] The computer-readable storage medium is used to store executable instructions;

[0093] The processor is configured to read executable instructions stored in the computer-readable storage medium and execute the method as described in any of the above embodiments.

[0094] This invention provides a computer-readable storage medium storing computer instructions that cause a processor to perform the method described in any of the above embodiments.

[0095] Those skilled in the art will readily understand that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A fully digital phase noise measurement method with a frequency resolution at the millihertz level, characterized in that, include: S1, convert the signal under test into a digital signal, and obtain the frequency control word of the digital signal; S2, Based on the frequency control word, generate two digital signals with orthogonal phases and no phase cutoff spurious signals. V h1 , V h2 ,Will V h1 , V h2 Each was mixed with the digital signal to obtain V h1 `、 V h2 `; S3, respectively for V h1 `、 V h2 Perform multi-stage decimation filtering to obtain the results at each decimation factor. V h1 `、 V h2 The filtered signal; and at each decimation factor, the... V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence. S4, perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; splice the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve; This involves segmenting the off-center carrier frequency range and selecting different sampling frequencies for different frequency bands to achieve power spectrum estimation, including: For the off-center carrier frequency range of 10k~100k Hz, data with a decimation factor of 16 are selected for power spectrum estimation; For the off-center carrier frequency range of 1k~10k Hz, data with a decimation factor of 64 are selected for power spectrum estimation; For the off-center carrier frequency range of 100~1kHz, data with a decimation factor of 256 are selected for power spectrum estimation. For the very near-end frequency band with a bias carrier frequency range of 0.0006 to 100 Hz, data with a decimation factor of 2048 are selected for power spectrum estimation to achieve a frequency resolution at the millihertz level.

2. The method as described in claim 1, characterized in that, In step S1, the frequency control word of the digital signal is obtained by using the equal precision frequency measurement method, the direct measurement method, or the periodic measurement method.

3. The method as described in claim 1, characterized in that, In step S2, a frequency synthesis method without phase cutoff, a ROM table compression frequency synthesis method, or a CORDIC frequency synthesis method is used to generate the frequency. V h1 , V h2 .

4. A fully digital phase noise measurement device with a frequency resolution at the millihertz level, characterized in that, include: The first processing module is used to convert the signal under test into a digital signal and obtain the frequency control word of the digital signal; The second processing module is used to generate two digital signals with orthogonal phases and no phase cutoff spurious signals according to the frequency control word. V h1 , V h2 ,Will V h1 , V h2 The digital signal is mixed with the digital signal respectively to obtain V h1 `、 V h2 `; The third processing module is used to process separately V h1 `、 V h2 Perform multi-stage decimation filtering to obtain the results at each decimation factor. V h1 `、 V h2 The filtered signal; and at each decimation factor, the... V h1 `'' filter signal and V h2 The filtered signal is divided into quotient and four-quadrant arctangent to obtain the corresponding initial phase difference sequence. Then, phase unwrapping and frequency offset elimination are performed on it to obtain the corresponding target phase difference sequence. The fourth processing module is used to perform power spectrum estimation on the target phase difference sequence at each decimation factor to obtain its single-sideband phase noise spectral density; and to splice the single-sideband phase noise spectral densities of the target phase difference sequence at each decimation factor to obtain a complete phase noise measurement curve. This involves segmenting the off-center carrier frequency range and selecting different sampling frequencies for different frequency bands to achieve power spectrum estimation, including: For the off-center carrier frequency range of 10k~100k Hz, data with a decimation factor of 16 are selected for power spectrum estimation; For the off-center carrier frequency range of 1k~10k Hz, data with a decimation factor of 64 are selected for power spectrum estimation; For the off-center carrier frequency range of 100~1kHz, data with a decimation factor of 256 are selected for power spectrum estimation. For the very near-end frequency band with a bias carrier frequency range of 0.0006 to 100 Hz, data with a decimation factor of 2048 are selected for power spectrum estimation to achieve a frequency resolution at the millihertz level.

5. A fully digital phase noise measurement system with a frequency resolution at the millihertz level, characterized in that, include: Computer-readable storage media and processors; The computer-readable storage medium is used to store executable instructions; The processor is configured to read executable instructions stored in the computer-readable storage medium and execute the method as described in any one of claims 1-3.

6. A computer-readable storage medium, characterized in that, The computer-readable storage medium stores computer instructions for causing a processor to perform the method as described in any one of claims 1-3.

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