Double-sideband FSI ranging system and method based on MZM-PM cascade modulation

Abstract

The invention discloses a double-sideband FSI ranging system and method based on MZM-PM cascade modulation, and mainly solves the problems that in the prior art, absolute distance and relative displacement can only be measured singly, and the ranging precision is not ideal. Comprising the following steps: 1) constructing an optical frequency scanning interference ranging system based on cascading of a Mach-Zehnder modulator and a phase modulator, and outputting an interference signal; 2) obtaining a group of orthogonal components by using the interference signal, the carrier wave and the frequency doubling carrier wave; 3) obtaining a sum value of upper and lower sideband phases of the interference signal by using the orthogonal component, calculating a phase variation, and obtaining relative displacement of an object; (4) constructing a complex signal by using the orthogonal component, and obtaining an interference signal phase containing absolute distance information through all-phase FFT (Fast Fourier Transform); and 5) the absolute distance is measured by solving the phase difference of the interference signals. According to the invention, integration of high-precision absolute distance measurement and relative displacement measurement is realized, and meanwhile, the real-time performance of displacement measurement is improved.

Description

Technical Field

[0001] The present invention belongs to the field of optical precision measurement, and further relates to an optical frequency scanning interferometry ranging technology, specifically a double-sideband optical frequency scanning interferometry FSI (Frequency Scanning Interferometry) ranging system and method based on cascade modulation of a Mach Zwedel modulator and a phase modulator MZM-PM (Mach Zwedel modulator & Phase modulator), which realizes the integration of high-precision absolute distance measurement and relative displacement measurement, and improves the real-time performance of displacement measurement. Background Art

[0002] In recent years, the range and accuracy of distance measurement have become increasingly demanding in many fields such as autonomous driving, high-end equipment manufacturing, and aerospace measurement. The development of advanced optical, electronic, and computing technologies has provided new opportunities for distance measurement technology. FSI laser distance measurement technology, which uses advanced sensors and signal processing methods, can achieve accurate absolute distance measurement in large, railless, and high-precision environments. However, in many of the fields mentioned above, relative displacement measurement is equally important. For example, in vehicle-mounted laser radar, it is necessary to identify micro-movement targets; in lithography machine manufacturing, it is necessary to continuously monitor the tiny displacement of components to ensure processing accuracy; in gravitational wave detection, the system needs to monitor tiny deformations in space with extremely high precision to capture gravitational wave signals. These application scenarios show that a single measurement method is often difficult to meet actual needs because they either focus on absolute distance or relative displacement. Therefore, an integrated distance measurement system that can simultaneously complete absolute distance and relative displacement measurements is needed to meet the multiple needs of different application scenarios. At the same time, under the premise of integrating distance and displacement measurements, new measurement methods are needed to achieve high precision.

[0003] In order to achieve the integration of high-precision absolute distance measurement and relative displacement measurement, the following common technologies are available: 1) Dual-wavelength laser interferometry, which uses two lasers of different wavelengths for interference measurement, can achieve a wide range of absolute distance measurement and has the ability to measure relative displacement with high precision. However, this method requires a high-stability dual-wavelength light source, and the system structure is complex and the cost is high. 2) Optical coherence tomography (OCT), which uses a low-coherence light source to reflect in the sample to form an interference signal, uses the depth information of the interference fringes to measure the absolute distance, and can achieve high-precision micro-displacement measurement. However, the measurement range of this method is limited and is usually used for measurements in the millimeter to centimeter range. 3) Self-mixing interferometry (SMI), which uses a wavelength-modulated laser source and the principle of self-mixing interference, is a compact and low-cost distance measurement method suitable for various high-precision absolute distance and relative displacement measurement applications. However, self-mixing interference requires that the object to be measured has a certain reflectivity, and the measurement effect of objects with low reflectivity is poor. Due to the limitation of the wavelength modulation frequency, the measurement range of the system may not be suitable for very long distances. In addition, the displacement measurement accuracy of this method is lower than the absolute distance measurement accuracy. Summary of the invention

[0004] The purpose of the present invention is to address the defects of the above-mentioned prior art and to propose a double-sideband FSI ranging system and method based on MZM-PM cascade modulation, so as to realize the integration of high-precision absolute distance measurement and relative displacement measurement.

[0005] The basic idea of ​​implementing the present invention is as follows: first, an FSI ranging system based on a cascade of a Mach-Zehnder modulator and a phase modulator (MZM-PM) is constructed to generate an interference signal; then, a set of orthogonal components are obtained by using the interference signal, a carrier and a double frequency carrier, which are used to construct a complex signal in distance measurement and the interference signal phase containing absolute distance information is obtained through an all-phase fast Fourier transform APFFT, and the distance measurement result is obtained by solving the phase difference of the interference signal that changes continuously at two different times under a fixed optical frequency variation range; in displacement measurement, the interference signal phase containing relative displacement information is obtained by dividing it and taking the inverse tangent, and the relative displacement is calculated using the phase change.

[0006] To achieve the above-mentioned purpose, the present invention proposes a double-sideband FSI ranging system based on MZM-PM cascade modulation, comprising: a frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber beam combiner FC, a photodetector PD and a data acquisition device; wherein the frequency-stabilized laser is used to generate an optical signal, and the optical signal is divided into two light beams for output after passing through the MZM and FS in sequence, the first light beam passes through the PM and enters the FC, and the second light beam passes through the EDFA and enters the circulator;

[0007] The second light beam entering the circulator is sent to the COL and emitted by the COL, and is reflected by the corner cube prism and received by the COL to obtain a laser echo; the laser echo is returned to the circulator and output to the FC through the circulator;

[0008] The FC is used to re-converge the two incoming light beams and interfere with each other and output them to the PD, and obtain the interference signal through the PD detection and data acquisition device;

[0009] Furthermore, the above-mentioned Mach-Zehnder modulator MZM is used to modulate the intensity of the optical signal generated by the frequency-stabilized laser and output it to the fiber splitter FS; the fiber splitter FS is used to split the optical signal to obtain the first light beam and the second light beam after decomposition; the phase modulator PM is used to phase modulate the first light beam and output it to the fiber combiner FC; the erbium-doped fiber amplifier EDFA is used to power amplify the second light beam and output it to the circulator.

[0010] At the same time, the present invention also proposes a double-sideband FSI ranging method based on MZM-PM cascade modulation, and the implementation steps include:

[0011] (1) The frequency-stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM;

[0012] (2) MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t);

[0013] (3) Optical signal S DSB (t) The optical signal L1 and L2 are divided into two optical signals L1 and L2 through the optical fiber splitter FS, where L1 points to the phase modulator PM and L2 points to the circulator through the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA amplifies the power of L2;

[0014] (4) The power-amplified L2 is used as the measurement light signal, which is emitted through the collimator COL, reflected after reaching the corner cube prism and received by the collimator to obtain the laser echo signal S MS (t);

[0015] (5) The reference optical signal S RS (t) and the laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photoelectric detector PD and the data acquisition device.M (t):

[0016]

[0017] Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interference; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of a circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS The time delay corresponding to the optical path difference of (t), v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration;

[0018] (6) Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω c t) Calculate the quadrature component S1(t) and the in-phase component S2(t);

[0019] (7) Using S1(t) and S2(t) to obtain the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re :

[0020] (8) Using S1(t) and S2(t), and using full-phase FFT to achieve the absolute distance L ab The measurement steps are as follows:

[0021] (8.1) Using S1(t) and S2(t), construct the complex signal E1(t);

[0022] (8.2) Using the complex signal E1(t) through full phase FFT, we can get the phase change of the upper and lower sidebands. and And calculate the absolute distance L ab .

[0023] Compared with the prior art, the present invention has the following advantages:

[0024] First, in the optical path part of the ranging system that generates the interference signal, the present invention uses a single detector to collect one interference signal, thereby avoiding the problem of unequal or non-ideal orthogonality of the two detectors in the double-detector orthogonal detection in the traditional ranging system.

[0025] Second, compared with the existing integrated system of absolute distance measurement and relative displacement measurement, the present invention proposes a ranging system structure of cascaded intensity modulator MZM and phase modulator PM to generate an electro-optical frequency comb, and then uses a solution that retains positive and negative first-order sidebands to achieve absolute distance measurement and relative displacement measurement, effectively simplifying the structure of the ranging system and reducing construction costs. At the same time, the relative displacement measurement accuracy of the present invention is higher than the absolute distance measurement accuracy, and is more practical in the application scenario of absolute relative ranging integration.

[0026] Third, the present invention generates two pairs of orthogonal components by mixing the interference signal with the carrier signal and the fundamental wave and second harmonic of the sinusoidal carrier signal. The phase of the interference signal containing relative displacement information is extracted by the PGC-Atan demodulation method. The calculation of the instantaneous phase in the phase demodulation algorithm does not need to rely on subsequent sampling points, thereby improving the real-time performance of displacement measurement.

[0027] Fourth, the present invention uses two pairs of orthogonal components to construct a complex signal, and then uses the full-phase FFT to perform phase discrimination on the complex signal to extract the phase of the interference signal containing absolute distance information. Compared with the situation where direct spectrum analysis of real signals may result in a double-sided spectrum, the present invention uses complex signals for spectrum analysis, which can effectively obtain a single-sided spectrum, reduce the risk of spectrum aliasing, and improve the phase discrimination accuracy, thereby significantly improving the absolute distance measurement accuracy of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0028] Figure 1 It is a structural schematic diagram of a double-sideband FSI system based on MZM-PM cascade modulation in the present invention;

[0029] Figure 2 It is a flowchart of the algorithm for distance measurement and relative displacement measurement in the present invention;

[0030] Figure 3 is a simulation result diagram of relative displacement measurement using the method of the present invention;

[0031] Figure 4This is a simulation result diagram of absolute distance measurement using the method of the present invention. DETAILED DESCRIPTION

[0032] The present invention will be further described below in conjunction with the accompanying drawings.

[0033] Example 1: Reference Figure 1 In this embodiment, the specific structure of the double-sideband FSI ranging system based on MZM-PM cascade modulation proposed in the present invention is further described as follows:

[0034] The FSI ranging system in the present invention is as follows: Figure 1 As shown; specifically includes a frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber beam combiner FC, a photodetector PD and a data acquisition device;

[0035] The frequency-stabilized laser is used to generate an optical signal and output it to a Mach-Zehnder modulator MZM. The MZM modulates the intensity of the light beam passing through and outputs it to a fiber beam splitter, which splits the light to obtain a first light beam and a second light beam after decomposition; and the two decomposed light beams are respectively output to a phase modulator PM and an erbium-doped fiber amplifier EDFA; wherein PM performs phase modulation on the light beam passing through and outputs it to a fiber beam combiner FC; the EDFA performs power amplification on the light signal passing through and outputs it to a circulator, which is output from the circulator to a collimator COL, which is emitted to a corner cube prism, reflected, received by the COL, and transmitted back to the circulator, and the circulator outputs a laser echo to the fiber beam combiner FC; in the FC, the two light beams converge again to interfere and are output to a photoelectric detector, which is detected by the photoelectric detector and converted by a data acquisition device to obtain an interference signal.

[0036] Example 2: Reference Figure 1-2 The present invention proposes a double-sideband FSI ranging method based on MZM-PM cascade modulation, which is implemented by a double-sideband FSI ranging system based on cascade modulation of an intensity modulator and a phase modulator, such as Figure 1 The system includes a frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber beam combiner FC, a photodetector PD and a data acquisition device; a method for distance measurement using the system specifically includes the following steps:

[0037] Step 1. The frequency-stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM; in this embodiment, the optical signal output by the laser is expressed as:

[0038] SL (t) = Acos(2πf L t),

[0039] Where A is the optical signal S L (t) AC amplitude, f L The center frequency of the laser emitted by a narrow linewidth laser.

[0040] Step 2. MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t):

[0041] S DSB (t) = a dsb [cos(2π(f L +f0)t+πKt 2 )+cos(2π(f L -f0)t-πKt 2 )],

[0042] Among them, a dsb is the power of the modulated optical signal.

[0043] Step 3. Optical signal S DSB (t) The optical signal L1 and L2 are divided into two optical signals L1 and L2 through the optical fiber splitter FS, where L1 points to the phase modulator PM and L2 points to the circulator through the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA amplifies the power of L2.

[0044] The reference optical signal S RS (t), is obtained according to the following steps:

[0045] (3.1) Apply a high-frequency carrier signal V to PM PM (t):

[0046]

[0047] Among them, α represents the amplification factor of the high-voltage amplifier, V0 and ω C is the modulation signal The amplitude and angular frequency of

[0048] (3.2) PM modulates the phase of L1, causing the beam to shift by The phase shift is obtained to obtain the reference optical signal S after phase modulation RS (t):

[0049]

[0050] Among them, V π is the half-wave voltage of the phase modulator, m = παV0 / V π is the phase modulation depth.

[0051] Step 4. The power-amplified L2 is used as the measurement light signal, which is emitted through the collimator COL. After reaching the corner cube prism, it is reflected and received by the collimator to obtain the laser echo signal S. MS (t):

[0052]

[0053] Step 5. Transform the reference optical signal S RS (t) and the laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photoelectric detector PD and the data acquisition device. M (t):

[0054]

[0055] Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interference; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of a circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS The time delay corresponding to the optical path difference of (t), v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration;

[0056] Step 6. Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω ct) Calculate the quadrature component S1(t) and the in-phase component S2(t):

[0057]

[0058] Wherein, J1(m) and J2(m) are first-order and second-order Bessel functions of the first kind, respectively.

[0059] Step 7. Use S1(t) and S2(t) to get the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re :

[0060] (7.1) Using S1(t) and S2(t), and setting the modulation depth, we can get the phase demodulation result. The implementation is as follows:

[0061] (7.1.1) The output signal S of the demodulation scheme is obtained according to the following formula out :

[0062]

[0063] (7.1.2) Set the modulation depth m so that the value of J1(m) / J2(m) changes to the minimum, that is, set J1(m) / J2(m) = 1, and get the demodulation result

[0064]

[0065] (7.2) Utilization Calculate the phase change at different instantaneous values And calculate the relative displacement L of the object re , implemented as follows:

[0066] (7.2.1) Yes The instantaneous phase values ​​corresponding to time t1 and t2 and Do the difference and get the phase change

[0067]

[0068] in, and They respectively represent the phase change of the upper and lower sidebands from time t1 to t2.

[0069] (7.2.2) Using the phase change Calculate the relative displacement L of the objectre :

[0070]

[0071] Step 8. Use S1(t) and S2(t) and obtain the absolute distance L through full phase FFT ab The measurement steps are as follows:

[0072] (8.1) Using S1(t) and S2(t), construct the complex signal E1(t):

[0073]

[0074] Where j represents the imaginary unit in the complex number, J1(m) and J2(m) are the first-order and second-order Bessel functions of the first kind, respectively.

[0075] (8.2) Using the complex signal E1(t) through full phase FFT, we can get the phase change of the upper and lower sidebands. and And calculate the absolute distance L ab :

[0076] (8.2.1) Using the complex signal E1(t) through full phase FFT, we can get the phase change of the upper and lower sidebands. and The implementation is as follows:

[0077] (8.2.1.1) The convolution window W with a length of 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point and the first N-1 sample points and the last N-1 sample points of the complex signal E1(t), which is a total of 2N-1 sample points. 1w (i) is expressed as follows:

[0078] E 1w (i) = W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)],

[0079] Wherein, N represents the number of sampling points of the full-phase FFT algorithm, i = 1, 2, ..., T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter;

[0080] (8.2.1.2) After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i):

[0081] E1(i)=E 1w (i)+E 1w(i+N);

[0082] (8.2.1.3) Perform a fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum; obtain the instantaneous phases of the upper and lower sidebands corresponding to the two spectrum peaks from the full phase spectrum of E1(i) and

[0083] (8.2.1.4) Yes and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within the time T and Do the difference and get the phase change and

[0084] (8.2.2) Calculate the absolute distance L according to the following formula ab (i)

[0085]

[0086] Embodiment 3: The overall implementation steps of the ranging method proposed in this embodiment are the same as those of Embodiment 2. Figure 2 The acquisition of the interference signal in the present invention is further described as follows:

[0087] The phase modulator modulates the phase of the optical signal passing through, causing the optical signal to shift by The phase shift of the reference optical signal S after phase modulation is obtained. RS (t):

[0088]

[0089]

[0090] Among them, V π is the half-wave voltage of the phase modulator, m = παV0 / V π is the phase modulation depth.

[0091] Expand the above formula to get:

[0092]

[0093] The part pointing to the circulator is used for distance measurement, thereby obtaining a laser echo signal S containing distance measurement information. MS (t):

[0094]

[0095] Where, τ(t)=(L OPD +2νt)n / c is the reference optical signal SRS (t) and the measured optical signal S MS The time delay corresponding to the optical path difference of (t), v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L represents the reference optical signal S RS (t) and the measured light signal (laser echo signal) S MS (t), L represents the optical path difference between the two, and the distance measurement value.

[0096] The reference optical signal S output by the phase modulator RS (t) and the circulator output S MS (t) The optical fibers are re-converged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photoelectric detector and the analog-to-digital converter. M (t):

[0097]

[0098] Where D is the interference signal I M (t) DC offset; is the reference optical signal S RS (t) and the upper sideband of the measured optical signal S MS The interference signal I obtained by interfering the upper sideband of M The upper sideband phase of (t), is the reference optical signal S RS (t) and the lower sideband of the measured optical signal S MS The interference signal I obtained by interfering the lower sideband of M (t) is the lower sideband phase. Expanding the above equation yields:

[0099]

[0100] Embodiment 4: The overall implementation steps of the ranging method proposed in this embodiment are the same as those of Embodiment 2. Figure 2 The present invention utilizes S1(t) and S2(t) and realizes the absolute distance L through full phase FFT (APFFT) ab The measurement process is described in further detail as follows:

[0101] The modulation depth m is set to minimize the change in the value of J1(m) / J2(m), that is, J1(m) / J2(m)=1; construct the complex signal E1(t):

[0102]

[0103] Here, j represents the imaginary unit in the complex number.

[0104] Using the complex signal E1(t) through APFFT, we can get the instantaneous phase of the upper and lower sidebands. and Here are the steps:

[0105] The convolution window W with a length of 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point and the first N-1 sample points and the last N-1 sample points of the complex signal E1(t), which is a total of 2N-1 sample points. 1w (i) is expressed as follows:

[0106] E 1w (i) = W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)],

[0107] Wherein, N represents the number of sampling points of the full-phase FFT algorithm, i = 1, 2, ..., T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter;

[0108] After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i):

[0109] E1(i)=E 1w (i)+E 1w (i+N);

[0110] Perform a fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum. From the full phase spectrum of E1(i), the instantaneous phases of the upper and lower sidebands corresponding to the two spectrum peaks are obtained. and The implementation is as follows:

[0111] From the full phase spectrum of E1(i), the discrete phase information of the upper and lower sidebands corresponding to the two spectrum peaks is obtained. and

[0112] The discrete phase information is phase unwrapped to obtain the instantaneous phases corresponding to the two peaks in the full phase spectrum of E1(i). and

[0113] Discrete phase information Taking the phase unwrapping as an example, the absolute phase sequence P1 of each sampling point starting from the phase value of the initial sampling point is obtained. + (i), P1 - (i) Similarly:

[0114]

[0115] Instantaneous phase and It is expressed as follows:

[0116]

[0117] right and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within the time T and Do the difference and get the phase change and

[0118]

[0119] The absolute distance L is calculated according to the following formula ab :

[0120]

[0121] The effect of the present invention can be further illustrated by the following simulation results.

[0122] 1. Simulation conditions

[0123] Using MATLAB simulation software, the wavelength of the laser is set to 1550nm, the sweep range of the sweep signal source is 5GHZ to 10GHZ, the scanning period of the sweep signal source is 1ms, the sampling rate of the analog-to-digital converter AD is 100MHZ, the signal-to-noise ratio is 40dB, the sampling point number of the full-phase FFT algorithm is set to 512, the absolute distance of the object to be measured is 10m, and it vibrates in a sinusoidal form with an amplitude of 10 nanometers and a frequency of 2000HZ. The sampling resolution is 10bit, and the simulation time is 0.5s.

[0124] 2. Simulation content

[0125] Simulation 1: Under the above simulation conditions, the relative displacement measurement accuracy diagram is realized by using the method of the present invention. The results are as follows: Figure 3 , where the horizontal axis represents the simulation time and the vertical axis represents the ranging accuracy. Figure 3 It can be seen that the relative displacement measurement accuracy achieved by the method of the present invention is 0.53 nanometers.

[0126] Simulation 2: Under the above simulation conditions, the absolute distance measurement accuracy diagram is realized by using the method of the present invention. The results are as follows: Figure 4 , where the horizontal axis represents the simulation time and the vertical axis represents the ranging accuracy. Figure 4 It can be seen that the method of the present invention achieves an absolute distance measurement standard deviation of 7.60 microns.

[0127] The above simulation analysis proves the correctness and effectiveness of the method proposed in the present invention.

[0128] Parts of the present invention that are not described in detail belong to common knowledge among those skilled in the art.

[0129] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Obviously, for professionals in this field, after understanding the content and principles of the present invention, they may make various modifications and changes in form and details without departing from the principles and structures of the present invention. However, these modifications and changes based on the ideas of the present invention are still within the scope of protection of the claims of the present invention.

Claims

1. A double-sideband FSI ranging system based on MZM-PM cascade modulation, characterized in that: include: A frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber beam combiner FC, a photodetector PD, and a data acquisition device; wherein the frequency-stabilized laser is used to generate an optical signal, and the optical signal is divided into two light beams for output after passing through the MZM and FS in sequence, the first light beam passes through the PM and enters the FC, and the second light beam passes through the EDFA and enters the circulator; The second light beam entering the circulator is sent to the COL and emitted by the COL, and is reflected by the corner cube prism and received by the COL to obtain a laser echo; the laser echo is returned to the circulator and output to the FC through the circulator; The FC is used to re-converge the interference of the two incoming light beams and output them to the PD, and the interference signal is converted by the PD detection and data acquisition device.

2. The system according to claim 1, characterized in that: The Mach-Zehnder modulator MZM is used to modulate the intensity of the optical signal generated by the frequency-stabilized laser and output it to the optical fiber splitter FS; the optical fiber splitter FS is used to split the optical signal to obtain the first and second optical beams after decomposition; the phase modulator PM is used to phase modulate the first optical beam and output it to the optical fiber combiner FC; the erbium-doped fiber amplifier EDFA is used to amplify the power of the second optical beam and output it to the circulator.

3. A double-sideband FSI ranging method based on MZM-PM cascade modulation, which is implemented by a double-sideband FSI ranging system based on cascade modulation of an intensity modulator and a phase modulator, the system comprising a frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber beam combiner FC, a photodetector PD and a data acquisition device; characterized in that, The steps include: (1) The frequency-stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM; (2) MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t); (3) Optical signal S DSB (t) The optical signal L1 and L2 are divided into two optical signals L1 and L2 through the optical fiber splitter FS, where L1 points to the phase modulator PM and L2 points to the circulator through the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA amplifies the power of L2; (4) The power-amplified L2 is used as the measurement light signal, which is emitted through the collimator COL, reflected after reaching the corner cube prism and received by the collimator to obtain the laser echo signal S MS (t); (5) The reference optical signal S RS (t) and the laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photoelectric detector PD and the data acquisition device. M (t): Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interference; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of a circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS The time delay corresponding to the optical path difference of (t), v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration; (6) Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω c t) Calculate the quadrature component S1(t) and the in-phase component S2(t); (7) Using S1(t) and S2(t) to obtain the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re : (8) Using S1(t) and S2(t), the absolute distance Lab is measured by full-phase FFT. The steps are as follows: (8.1) Using S1(t) and S2(t), construct the complex signal E1(t); (8.2) Using the complex signal E1(t) through full phase FFT, we can get the phase change of the upper and lower sidebands. and And calculate the absolute distance L ab .

4. The method according to claim 3, characterized in that: The optical signal S in step (1) L (t) and the optical signal S containing upper and lower sidebands in step (2) DSB (t), are expressed as follows: S L (t)=Acos(2πf L t), S DSB (t)=a dsb [cos(2π(f L +f0)t+πKt 2 )+cos(2π(f L -f0)t-πKt 2 )], Among them, a dsb is the power of the modulated optical signal.

5. The method according to claim 3, characterized in that: Step (3) The reference optical signal S RS (t), is obtained according to the following steps: (3.1) Apply a high-frequency carrier signal V to PM PM (t): Among them, α represents the amplification factor of the high-voltage amplifier, V0 and ω C is the modulated signal The amplitude and angular frequency of (3.2) PM modulates the phase of L1, causing the beam to shift by The phase shift is obtained to obtain the reference optical signal S after phase modulation RS (t): Among them, V π is the half-wave voltage of the phase modulator, m = παV0 / V π is the phase modulation depth.

6. The method according to claim 3, characterized in that: The laser echo signal S in step (4) MS (t), expressed as follows:

7. The method according to claim 3, characterized in that: The orthogonal component S1(t) and the in-phase component S2(t) in step (6) are expressed as follows: Wherein, J1(m) and J2(m) are first-order and second-order Bessel functions of the first kind, respectively.

8. The method according to claim 7, characterized in that: Step (7) calculates the phase change and the relative displacement L of the object re , as follows: (7.1) Using S1(t) and S2(t), and setting the modulation depth, we can get the phase demodulation result. The implementation is as follows: (7.1.1) The output signal S of the demodulation scheme is obtained according to the following formula out : (7.1.2) Set the modulation depth m so that the value of J1(m) / J2(m) changes to the minimum, that is, set J1(m) / J2(m) = 1, and get the demodulation result (7.2) Utilization Calculate the phase change at different instantaneous values And calculate the relative displacement L of the object re , implemented as follows: (7.2.1) Yes The instantaneous phase values ​​corresponding to time t1 and t2 and Do the difference and get the phase change in, and They respectively represent the phase change of the upper and lower sidebands from time t1 to t2. (7.2.2) Using the phase change Calculate the relative displacement L of the object re :

9. The method according to claim 3, characterized in that: Using S1(t) and S2(t) as described in step (8.1), construct the complex signal E1(t), which is implemented as follows: Wherein, j represents the imaginary unit in the complex number, J1(m) and J2(m) are the first-order and second-order Bessel functions of the first kind, respectively.

10. The method according to claim 3, characterized in that: The absolute distance L mentioned in step (8.2) ab , calculated according to the following steps: (8.2.1) Using the complex signal E1(t) through full phase FFT, we can get the phase change of the upper and lower sidebands. and The implementation is as follows: (8.2.1.1) The convolution window W with a length of 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point of the complex signal E1(t) and the first N-1 sample points and the last N-1 sample points, a total of 2N-1 sample points. 1w (i) is expressed as follows: HAVE BEEN 1w (i)=W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)], Wherein, N represents the number of sampling points of the full-phase FFT algorithm, i = 1, 2, ..., T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter; (8.2.1.2) After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i): E1(i)=E 1w (i)+E 1w (i+N); (8.2.1.3) Perform a fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum; obtain the instantaneous phases of the upper and lower sidebands corresponding to the two spectrum peaks from the full phase spectrum of E1(i) and (8.2.1.4) Yes and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within the time T and Do the difference and get the phase change and (8.2.2) Calculate the absolute distance L according to the following formula ab (i)

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