Air-ground communication method adaptive to double-selection fading channel

Abstract

The invention belongs to the field of wireless communication, particularly relates to wireless communication of aerospace crafts, and provides an air-ground communication method adaptive to a double-selection fading channel. The method comprises the following steps: step 1, designing an anti-doubly selective channel parameter; 2, framing is carried out according to the sequence of power control, a synchronization head, a pilot frequency X and a data segment, and a communication frame is generated; step 3, carrying out secondary correlation peak detection based on differential operation on the received data; step 4, performing coarse carrier synchronization based on peak detection; 5, performing fine carrier synchronization based on peak detection; step 6, carrying out joint transform domain and time-frequency interpolation channel estimation; and step 7, carrying out channel equalization.

Description

Technical Field

[0001] The invention belongs to the field of wireless communication, and in particular relates to wireless communication of aerospace vehicles, and is an air-to-ground communication method adapting to dual-selection fading channels. Background Art

[0002] With the rapid development of spacecraft, the demand for wireless communication is getting higher and higher, and it faces multipath fading and high-speed movement. Multipath fading brings frequency selective fading, and high-speed movement brings time selective fading, thus synthesizing severe time-frequency dual-selective channels, that is, the channel changes dramatically in both frequency and time dimensions. For quasi-static channels such as multipath channels, we can regard the wireless channel as a linear time-invariant system. However, for time-varying channels such as dual-selective channels, the linear time-invariant system model is no longer applicable, and the linear time-varying model is more in line with the actual situation. In a high-speed mobile environment, large Doppler frequency shift will cause the channel to change rapidly. The impact of rapidly changing channels on multi-carrier communication systems can be divided into three aspects: first, inter-subcarrier interference is generated within the symbol, second, selective fading between symbols, and third, deterioration of channel estimation performance. The channels of each subcarrier within a symbol and between symbols change dramatically, resulting in greater interference between each subcarrier within a symbol. At the same time, large Doppler frequency shift causes faster channel changes and stronger selective fading between symbols, which seriously deteriorates the receiver performance. Therefore, it is necessary to find an air-to-ground communication design method that adapts to dual-selective fading channels and effectively combats the time and frequency dual-selective channel fading caused by high-speed movement. Summary of the invention

[0003] The purpose of the present invention is to propose an air-to-ground communication method that is adaptable to dual-selective fading channels. The method adopts a distributed block pilot structure, determines the subcarrier spacing according to the multipath delay, and reduces the influence of frequency selective fading; determines the receiving equalization block length according to the coherence time to combat time selective fading; at the same time, the synchronization head designed by the present invention is less affected by the Doppler frequency shift, can complete the carrier synchronization more accurately, and has a larger frequency deviation correction range; secondly, the equalization block at the receiving end adopts the front and rear pilots Y for precise carrier synchronization, which can improve the frequency deviation estimation accuracy and the equalization effect.

[0004] The technical solution of the present invention:

[0005] An air-to-ground communication method adapted to a dual-selection fading channel, the steps are as follows:

[0006] Step 1: Design anti-double selection channel parameters;

[0007] Step 2: Frame according to the order of power control, synchronization header, pilot X, and data segment to generate a communication frame;

[0008] Step 3: Perform secondary correlation peak detection based on differential operation on the received data;

[0009] Step 4: Perform coarse carrier synchronization based on peak detection;

[0010] Step 5: Perform precise carrier synchronization based on peak detection;

[0011] Step 6: Perform joint transform domain and time-frequency interpolation channel estimation;

[0012] Step 7: Perform channel equalization.

[0013] Further, the pilot X includes two CPs and a pilot symbol;

[0014] The data segment includes N data blocks of equal length;

[0015] Each data block includes two data symbols of equal length and a pilot Y;

[0016] Each data symbol includes: CP and OFDM symbol;

[0017] Pilot Y includes CP and pilot symbols.

[0018] Furthermore, the anti-dual selection channel parameters include:

[0019] The anti-frequency selective fading parameter requirements are: the subcarrier spacing Δf of multi-carrier modulation is less than the coherence bandwidth ΔB; the coherence bandwidth ΔB is equal to the maximum multipath delay τ max Inversely proportional,

[0020] The subcarrier spacing Δf is:

[0021] The time selective fading parameter requirement is: the length of an OFDM symbol of multi-carrier modulation is T s Less than the coherence time T c ; Coherence time T c and the maximum Doppler shift ν max Inversely proportional,

[0022] An OFDM symbol length is T s The values ​​are:

[0023] Further, the synchronization header generation process is as follows:

[0024] S2.1, the synchronization header is generated by the PN sequence and the ZC sequence;

[0025] S2.2, the length of the ZC sequence in the synchronization header is S2; the length of the PN sequence in the synchronization header is S1;

[0026] S2.3, the PN sequence is generated by bit-by-bit differential coding of the original PN code; the differential coding rule is: set the PN sequence reference bit d0 to 0, and perform differential coding on it bit by bit, the formula is as follows:

[0027] d k =c k ⊕d k-1

[0028] In the formula, ⊕ represents XOR, c k Represents the original PN code, d k represents the PN sequence after differential encoding;

[0029] S2.4, synchronization header generation rule: use a ZC sequence with a length of S2 as the spreading code, perform S2 times phase modulation and spreading on the generated PN sequence, and generate a synchronization header with a length of S1*S2.

[0030] Further, the data segment generation process is as follows:

[0031] The data segment is decomposed into N / 4 interleaved groups according to the number of data blocks;

[0032] Telemetry data and mission data adopt variable interleaving groups and different modulation modes, and the interleaving groups and modulation modes are dynamically allocated according to the telemetry and mission data transmission rate requirements;

[0033] M interleaved groups transmit telemetry data using BPSK modulation;

[0034] N / 4-M interleaved groups transmit task data, using BPSK or QPSK modulation;

[0035] Every 8 data blocks in N / 4-M interleaving groups form an interleaving block, and a two-dimensional random interleaving method is used between the interleaving blocks and the interleaving groups to reduce the correlation between the information bits.

[0036] The equalization block at the receiving end includes: pilot Y, data symbol 1, data symbol 2 and pilot Y.

[0037] Furthermore, the secondary correlation peak detection based on differential operation includes signal quantization based on variable phase, primary peak detection and secondary differential peak detection, and the process is as follows:

[0038] S5.1, variable phase signal quantization performs phase quantization on the received signal, and maps the signals in different angle ranges to Q phases respectively, and the phase quantization Q takes an even number and is greater than or equal to 4;

[0039] S5.2, the first level peak detection is to correlate the phase quantization signal with the local ZC to obtain a correlation value P1;

[0040] S5.3, secondary differential peak detection is to de-differentiate P1, shift it by one spread spectrum code length for conjugate multiplication, and then correlate it with the local PN sequence. A larger peak will appear in the secondary differential, and the maximum peak detection is performed to determine the timing synchronization position timing_loc1.

[0041] Further, the coarse carrier synchronization process is as follows:

[0042] Use the correlation value P1 to perform order difference, and then perform M-point FFT, take the modulus to find the maximum value, calculate the frequency deviation based on the maximum value and its corresponding subscript position, and estimate the frequency deviation fre_set1. The formula is as follows:

[0043]

[0044] Where b is the position where the FFT output modulus value is the largest, F symb The synchronization header symbol rate, L is the spreading code length, and M is the number of FFT points.

[0045] Further, the precise carrier synchronization process is as follows:

[0046] The received signal is reorganized according to the timing synchronization position, and a copy of the pilot Y in each data block is placed in each data block to form an equalization block. Each equalization block consists of the front and rear two blocks of pilot Y and the middle two blocks of OFDM data; the front and rear pilot Y of each equalization block are correlated, and the frequency deviation result is used for correction. The relevant formula is as follows:

[0047] Related formula:

[0048]

[0049] Where LL is the pilot length, F d is the data symbol rate, M is the number of FFT points in S6, pilot_zc1 and pilot_zc2 are the pilots Y of the equalization block;

[0050] The estimated total frequency deviation is:

[0051] total_offset_esti=exact_offset_esti+fre_set1;

[0052] in

[0053] exact_esti_angle=angle(exact_esti_corr).

[0054] Furthermore, the joint transform domain and time-frequency interpolation channel estimation process is as follows:

[0055] S8.1, perform FFT transformation and IFFT transformation on the received signal:

[0056] (1) Perform FFT transformation on the received signal rm(i), where L is the length of the pilot Y, and the result frm is:

[0057] FFT(rm)=[frm0, frm1,..., frm L-1 ]

[0058] The pilot Y sequence is:

[0059] fm(i)=[fm(0), fm(1), ..., fm(L-1)](2) Taking the conjugate of fm is:

[0060] fm*(i)=[conj(fm(0)), conj(fm(1)),..., conj(fm(L-1))]

[0061] (3) Perform IFFT transformation on frm*fm* to obtain the signal impulse response:

[0062] h=IFFT(frm*fm*)=[h(0), h(1),,...,h(L-1)];

[0063] S8.2, performing channel impulse response processing according to signal power and noise threshold;

[0064] (1) Calculate the power of the strongest signal path:

[0065] Pow max =max[|h(0)| 2 ,|h(1)| 2 , …, |h(P-1)| 2 ], P = L

[0066] (2) Set the noise threshold to

[0067] a=Pow max / 16

[0068] (3) Perform noise reduction on the signal impulse response, set the data less than the noise threshold to zero, and obtain the channel impulse time domain response h_esti, and then perform L-point FFT transformation to obtain H_esti;

[0069] S8.3, interpolation processing is performed on H_esti. Since there are two pilot signals Y in the equalization block, the channel impulse responses of the two pilot signals Y before and after the equalization block are set to H1 and H4, and then linear interpolation is performed to obtain the channel responses H2 and H3 of the two middle data blocks.

[0070]

[0071] Thus, the channel response of the equalization block is H = [H1 H2 H3 H4].

[0072] Furthermore, channel equalization is to perform frequency domain equalization processing according to the channel response of the equalization block.

[0073] Beneficial effects of the present invention:

[0074] The present invention discloses an air-to-ground communication method adapting to a dual-selective fading channel, which adopts a distributed block pilot structure, determines the subcarrier spacing according to the multipath delay, and reduces the influence of frequency selective fading; determines the receiving equalization block length according to the coherence time to combat time selective fading; at the same time, the synchronization head designed by the present invention is less affected by the Doppler frequency shift, can complete the carrier synchronization more accurately, and has a larger frequency deviation correction range; secondly, the equalization block at the receiving end adopts the front and rear pilots Y to perform precise carrier synchronization, which can improve the frequency deviation estimation accuracy and the equalization effect. BRIEF DESCRIPTION OF THE DRAWINGS

[0075] Figure 1 is a schematic diagram of a signal transmission process in an embodiment of the present invention;

[0076] Figure 2 is a schematic diagram of a frame structure design in an embodiment of the present invention;

[0077] Figure 3 A schematic diagram of a signal receiving process in an embodiment of the present invention;

[0078] Figure 4 is a schematic diagram of primary correlation peak detection in an embodiment of the present invention;

[0079] Figure 5 is a schematic diagram of secondary differential peak detection in an embodiment of the present invention;

[0080] Figure 6 is a schematic diagram of the structure of the equalization block in an embodiment of the present invention;

[0081] Figure 7 is a constellation diagram of two data blocks after equalization without interpolation in an embodiment of the present invention;

[0082] Figure 8 4 is a constellation diagram of two data blocks after frequency domain interpolation equalization in an embodiment of the present invention. DETAILED DESCRIPTION

[0083] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0084] The present invention discloses an air-to-ground communication method adapted to a dual-selection fading channel, comprising the following steps:

[0085] The signal transmission process includes the design of anti-dual selection channel waveform parameters, synchronization header, data segment, etc. Figure 1 shown.

[0086] The communication frame consists of a power control, synchronization header, pilot X and data segment. Figure 2 shown.

[0087] The pilot X consists of two CPs and a pilot symbol;

[0088] The data segment consists of N data blocks of the same length;

[0089] The data block consists of two data symbols (data symbol 1, data symbol 2) of the same length and a pilot Y;

[0090] The data symbol is composed of CP and OFDM symbol;

[0091] The pilot Y consists of a CP and a pilot symbol.

[0092] S1 Anti-Dual Selection Channel Parameter Design

[0093] The anti-frequency selective fading parameter requirement in S1.1 is that the subcarrier spacing Δf of multi-carrier modulation must be smaller than the coherence bandwidth ΔB; the coherence bandwidth ΔB is related to the maximum multipath delay τ max Inversely proportional, that is: In order to ensure that a single subcarrier in a multi-carrier modulation system is not affected by frequency selective fading, the subcarrier spacing Δf is taken as:

[0094] S1.2 The time selective fading parameter requirement is that the length of an OFDM symbol of multi-carrier modulation is T s Need to be less than the coherence time T c ; Coherence time T c and the maximum Doppler shift ν max Inversely proportional, that is In order to ensure that a single OFDM symbol in a multi-carrier modulation system is not affected by time selective fading, an OFDM symbol length T s The values ​​are:

[0095] S2 generates synchronization header

[0096] S2.1 The synchronization header is generated by a PN sequence and a ZC sequence;

[0097] S2.2 The length of the ZC sequence in the synchronization header is S2; the length of the PN sequence in the synchronization header is S1;

[0098] S2.3 The PN sequence is generated by bit-by-bit differential coding of the original PN code;

[0099] The differential encoding rule described in S2.3.1 is: set the PN sequence reference bit d0 to 0, and perform differential encoding on it bit by bit. The formula is as follows:

[0100] d k =c k ⊕d k-1

[0101] In the formula, ⊕ represents XOR, c k Represents the original PN code, d k Represents the PN sequence after differential encoding.

[0102] The generation rule of the synchronization header described in S2.4 is: using a ZC sequence with a length of S2 as a spreading code, performing S2 times phase modulation spreading on the PN sequence generated in S2.3.1 to generate a synchronization header with a length of S1*S2.

[0103] S3 generates data segments

[0104] The data segment consists of N data blocks of the same length;

[0105] The data block consists of two data symbols of the same length and a pilot Y;

[0106] The data symbol is composed of CP and OFDM symbol;

[0107] The pilot Y consists of a CP and a pilot symbol.

[0108] The data segment can be decomposed into N / 4 interleaving groups according to the number of data blocks;

[0109] The telemetry data and mission data adopt variable interleaving groups and different modulation modes, so that under the same synchronization head detection probability condition, the interleaving groups and modulation modes can be dynamically allocated according to the telemetry and mission data transmission rate requirements, ensuring that the telemetry data receiving sensitivity is improved by more than 3dB compared with the mission data receiving sensitivity;

[0110] The M interleaved groups transmit telemetry data using BPSK modulation;

[0111] The N / 4-M interleaving groups transmit task data using BPSK or QPSK modulation;

[0112] Every 8 data blocks in the N / 4-M interleaving groups form an interleaving block, and a two-dimensional random interleaving method is adopted between the interleaving blocks and the interleaving groups to reduce the correlation between the information bits.

[0113] The equalization block at the receiving end may be composed of pilot Y, data symbol 1, data symbol 2 and pilot Y.

[0114] S4 output communication frame

[0115] The communication frame is generated by framing according to the order of power control, synchronization header, header and data segment.

[0116] The signal receiving process includes two-level correlation peak detection based on differential operation, coarse carrier synchronization based on peak detection, fine carrier synchronization based on peak detection, channel estimation and MMSE equalization based on joint transform domain and frequency interpolation, etc. Figure 3 .

[0117] The secondary correlation peak detection based on differential operation described in S5 includes signal quantization based on variable phase, primary peak detection correlation and secondary differential peak detection, etc.

[0118] The variable phase signal quantization in S5.1 performs phase quantization on the received signal, that is, mapping the signals in different angle ranges to Q phases respectively, and the phase quantization Q value is generally an even number and is greater than or equal to 4;

[0119] The first level peak detection described in S5.2 is that the phase quantization signal is correlated with the local ZC to obtain the correlation value P1, such as Figure 4 As shown;

[0120] The secondary differential peak detection described in S5.3 is to perform differential analysis on P1, that is, to perform conjugate multiplication by shifting one spread spectrum code length, and then to perform correlation with the local PN sequence. A larger peak will appear in the secondary differential. Figure 5 As shown, the maximum peak value is detected to determine the timing synchronization position timing_loc1;

[0121] S6 describes the coarse carrier synchronization based on peak detection: using the correlation result P1 obtained in 5.2 to perform order differentiation, and then performing M-point FFT, taking the modulus to find the maximum value, and calculating the frequency deviation based on the maximum value and its corresponding subscript position to estimate the frequency deviation fre_set1. The estimation formula is shown in the following formula:

[0122]

[0123] Where b is the position where the FFT output modulus value is the largest, F symb The synchronization header symbol rate, L is the spreading code length, and M is the number of FFT points.

[0124] The precise carrier synchronization based on peak detection described in S7: the received signal is reorganized into data blocks according to the timing synchronization position in S5.3, that is, a copy of the pilot Y in each data block is placed in each data block to form a balanced block, such as Figure 6 As shown, each equalization block consists of two blocks of pilot signals Y before and after and two blocks of OFDM data in the middle. The pilot signals Y before and after each equalization block are correlated, and the frequency deviation result obtained by S6 is used for correction. The relevant formula is as follows:

[0125] Related formula:

[0126]

[0127] Where LL is the pilot length, F d is the data symbol rate, M is the number of FFT points in S6, pilot_zc1 and pilot_zc2 are the pilot Y of the equalization block.

[0128] The estimated total frequency offset is: total_offset_esti=exact_offset_esti+fre_set1;

[0129]

[0130] exact_esti_angle=angle(exact_esti_corr)

[0131] The channel estimation of the joint transform domain and time-frequency interpolation in S8 mainly includes signal noise reduction processing, frequency domain interpolation, etc.

[0132] S8.1 The signal noise reduction processing includes FFT and IFFT transformation of the received signal and signal noise reduction processing.

[0133] S8.1.1 The received signal FFT transformation and IFFT transformation are as follows:

[0134] (1) Perform FFT transformation on the received signal rm(i), where L is the length of the pilot Y, and the result frm is:

[0135] FFT(rm)=[frm0, frm1,..., frm L-1 ]

[0136] The pilot Y sequence is:

[0137] fm(t)=[fm(0),fm(1),…,fm(L-1)](2) Taking the conjugate of fm is:

[0138]

[0139] (3) Perform IFFT transformation on frm·*fm* to obtain the signal impulse response:

[0140] h=IFFT(frm.*fm*)=[h(0), h(1),,...,h(L-1)]

[0141] The signal noise reduction processing described in S8.1.2 is to perform channel impulse response processing based on signal power and noise threshold.

[0142] (1) Calculate the power of the strongest signal path:

[0143] Pow max =max[|h(0)| 2 ,|h(1)| 2 , …, |h(P-1)| 2 ], P = L

[0144] (2) Set the noise threshold to

[0145] a=Pow max / 16

[0146] (3) The signal impulse response is subjected to noise reduction processing, that is, the following processing is performed to obtain the channel impulse time domain response he sti, and then an L-point FFT transform is performed to obtain He esti.

[0147] for n-1;P

[0148] if|h(n)| 2 <α

[0149] h(n)=0

[0150] end

[0151] The frequency domain interpolation processing described in S8.2 is performed on H_estl obtained in S8.1.2. Since there are two pilot signals Y in the equalization block, the channel impulse responses of the two pilot signals Y before and after the equalization block are set to H1 and H4, and then the channel responses H2 and H3 of the two middle data blocks are obtained by linear interpolation, that is,

[0152]

[0153] Thus, the channel response of the equalization block is H = [H1 H2 H3 H4].

[0154] The MMSE equalization described in S9 is to perform frequency domain equalization processing according to the channel impulse response obtained in S8.2, and output a signal value.

[0155] Example

[0156] Assume that in the present invention max =20us, then ΔB = 50kHz, so the subcarrier spacing Δf must satisfy

[0157] Δf≤12.5kHz

[0158] Assume that the subcarrier spacing is Δf = 10kHz; then the length of an OFDM symbol is 1 / Δf = 100us; then the CP length is 1 / 4 of the OFDM symbol, which is T CP =25us.

[0159] Assume that the maximum Doppler frequency shift ν of the present invention is max =500Hz, then T c =2ms, then the length of an OFDM symbol is T c The duration of one symbol is flat fading. The data symbol length is 125us, the pilot Y length is 125us, and the equalization block length is 500us.

[0160] Figure 7 is a constellation diagram of two data blocks after equalization without interpolation in an embodiment of the present invention; Figure 8 It is the constellation diagram of two data blocks after frequency domain interpolation equalization in the embodiment of the present invention. It can be seen from the two figures that after the introduction of frequency domain interpolation and the use of linear interpolation, the constellation diagram is basically straightened, and the influence on demodulation can be ignored.

[0161] The above is only a specific embodiment of the present invention, and the present invention is described in detail. The unexplained part is a conventional technology. However, the protection scope of the present invention is not limited to this. Any changes or substitutions that can be easily thought of by any technician familiar with the technical field within the technical scope disclosed by the present invention should be included in the protection scope of the present invention. The protection scope of the present invention shall be based on the protection scope of the claims.

Claims

1. An air-to-ground communication method adapted to a dual-selection fading channel, characterized in that: The steps are as follows: Step 1: Design anti-double selection channel parameters; Step 2: Frame according to the order of power control, synchronization header, pilot X, and data segment to generate a communication frame; Step 3: Perform secondary correlation peak detection based on differential operation on the received data; Step 4: Perform coarse carrier synchronization based on peak detection; Step 5: Perform precise carrier synchronization based on peak detection; Step 6: Perform joint transform domain and time-frequency interpolation channel estimation; Step 7: Perform channel equalization.

2. The method according to claim 1, characterized in that: The pilot X includes two CPs and a pilot symbol; The data segment includes N data blocks of equal length; Each data block includes two data symbols of equal length and a pilot Y; Each data symbol includes: CP and OFDM symbol; Pilot Y includes CP and pilot symbols.

3. The method according to claim 2, characterized in that: Anti-dual channel selection parameters include: The anti-frequency selective fading parameter requirements are: the subcarrier spacing Δf of multi-carrier modulation is less than the coherence bandwidth ΔB; the coherence bandwidth ΔB is equal to the maximum multipath delay τ max Inversely proportional, The subcarrier spacing Δf is: The time selective fading parameter requirement is: the length of an OFDM symbol of multi-carrier modulation is T s Less than the coherence time T c ; Coherence time T c and the maximum Doppler shift ν max Inversely proportional, An OFDM symbol length is T s The values ​​are:

4. The method according to claim 3, characterized in that: The synchronization header generation process is as follows: S2.1, the synchronization header is generated by the PN sequence and the ZC sequence; S2.2, the length of the ZC sequence in the synchronization header is S2; the length of the PN sequence in the synchronization header is S1; S2.3, the PN sequence is generated by bit-by-bit differential coding of the original PN code; the differential coding rule is: set the PN sequence reference bit d0 to 0, and perform differential coding on it bit by bit, the formula is as follows: d k =c k ⊕d k-1 In the formula, ⊕ represents XOR, c k Represents the original PN code, d k represents the PN sequence after differential encoding; S2.4, synchronization header generation rule: use a ZC sequence with a length of S2 as the spreading code, perform S2 times phase modulation and spreading on the generated PN sequence, and generate a synchronization header with a length of S1*S2.

5. The method according to claim 4, characterized in that: The data segment generation process is as follows: The data segment is decomposed into N / 4 interleaved groups according to the number of data blocks; Telemetry data and mission data adopt variable interleaving groups and different modulation modes, and the interleaving groups and modulation modes are dynamically allocated according to the telemetry and mission data transmission rate requirements; M interleaved groups transmit telemetry data using BPSK modulation; N / 4-M interleaved groups transmit task data, using BPSK or QPSK modulation; Every 8 data blocks in N / 4-M interleaving groups form an interleaving block, and a two-dimensional random interleaving method is used between the interleaving blocks and the interleaving groups to reduce the correlation between the information bits. The equalization block at the receiving end includes: pilot Y, data symbol 1, data symbol 2 and pilot Y.

6. The method according to claim 5, characterized in that: The secondary correlation peak detection based on differential operation includes signal quantization based on variable phase, primary peak detection and secondary differential peak detection. The process is as follows: S5.1, variable phase signal quantization performs phase quantization on the received signal, and maps the signals in different angle ranges to Q phases respectively, and the phase quantization Q takes an even number and is greater than or equal to 4; S5.2, the first level peak detection is to correlate the phase quantization signal with the local ZC to obtain a correlation value P1; S5.3, secondary differential peak detection is to de-differentiate P1, shift it by one spread spectrum code length for conjugate multiplication, and then correlate it with the local PN sequence. A larger peak will appear in the secondary differential, and the maximum peak detection is performed to determine the timing synchronization position timing_loc1.

7. The method according to claim 6, characterized in that: The coarse carrier synchronization process is as follows: Use the correlation value P1 to perform differential analysis, and then perform M-point FFT, take the modulus to find the maximum value, calculate the frequency offset based on the maximum value and its corresponding subscript position, and estimate the frequency offset fre_set1. The formula is as follows: Where b is the frequency point where the maximum modulus value of FFT output is located, which is the sequence value of M / 2. symb The synchronization header symbol rate, L is the spreading code length, and M is the number of FFT points.

8. The method according to claim 7, characterized in that: The precise carrier synchronization process is as follows: The received signal is reorganized according to the timing synchronization position, and a copy of the pilot Y in each data block is placed in each data block to form an equalization block. Each equalization block consists of the front and rear two blocks of pilot Y and the middle two blocks of OFDM data; the front and rear pilot Y of each equalization block are correlated, and the frequency deviation result is used for correction. The relevant formula is as follows: Related formula: Where LL is the pilot length, F d is the data symbol rate, M is the number of FFT points in S6, pilot_zc1 and pilot_zc2 are the pilots Y of the equalization block; The estimated total frequency deviation is: total_offset_esti=exact_offset_esti+fre_set1; in exact_esti_angle=angle(exact_esti_corr).

9. The method according to claim 8, characterized in that: The process of joint transform domain and time-frequency interpolation channel estimation is as follows: S8.1, perform FFT transformation and IFFT transformation on the received signal: (1) Perform FFT transformation on the received signal rm(t), where L is the length of the pilot Y. The result frm is: FFT(rm)=[frm0,frm1,...,frm L-1 ] The pilot Y sequence is: fm(i)-[fm(0), fm(1),..., fm(L-1)] (2) Taking the conjugate of fm is: (3) Perform IFFT transformation on frm*fm* to obtain the signal impulse response: h=IFFT(frm*fm * )=[h(0),h(1),...,h(L-1)]; S8.2, performing channel impulse response processing according to signal power and noise threshold; (1) Calculate the power of the strongest signal path: pow max =max[|h(0)| 2 ,|h(1)| 2 ,...,|h(p-1)| 2 ],p=L (2) Set the noise threshold to a=Pow max / 16 (3) Perform noise reduction on the signal impulse response, set the data less than the noise threshold to zero, and obtain the channel impulse time domain response h_esti, and then perform L-point FFT transformation to obtain H_esti; S8.3, interpolation processing is performed on H_esti. Since there are two pilot signals Y in the equalization block, the channel impulse responses of the two pilot signals Y before and after the equalization block are set to H1 and H4, and then linear interpolation is performed to obtain the channel responses H2 and H3 of the two middle data blocks. Thus, the channel response of the equalization block is H = [H1 H2 H3 H4].

10. The method according to claim 9, characterized in that: Channel equalization is a frequency domain equalization process based on the channel response of the equalization block.

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