Integrated lithium niobate photon millimeter wave radar

Abstract

A centimeter resolution integrated photonic millimeter wave (mmWave) radar chip operating in the millimeter wave V band based on 4 inch wafer level thin film lithium niobate (TFLN) technology is provided. The manufactured TFLN photonic millimeter wave integrated circuit has a first electro-optical modulator (EOM) capable of generating a broadband linear frequency modulated millimeter wave radar waveform by optical frequency multiplication of a low frequency input signal and a second EOM for frequency de-chirp of the reflected echo, therefore, the bandwidth requirements on the digital-to-analog converter in the transmitter and the analog-to-digital converter in the receiver are greatly reduced. In the absence of optical and electrical filters, the radar chip exhibits the following characteristics: continuous on-demand tunability of the center frequency and bandwidth, while now being limited by the bandwidth of the electrical amplifier. The radar chip provides a promising way for realizing high-resolution sensing and detection of vehicle-mounted radars, airborne radars and smart homes in the 6G era.

Description

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] This application claims priority to and the benefit of U.S. Provisional Patent Application Serial No. 63 / 592,962, filed on October 25, 2023, the disclosure of which is incorporated herein by reference in its entirety.

[0003] abbreviation

[0004] 6G Sixth Generation Wireless

[0005] AC

[0006] ADC Analog-to-Digital Converter

[0007] ANT Antenna

[0008] AWG Arbitrary Waveform Generator

[0009] CMOS Complementary Metal Oxide Semiconductor

[0010] CPI Coherent Processing Interval

[0011] CS-DSB Carrier Suppressed Double Side Band

[0012] DAC Digital to Analog Converter

[0013] DC Direct Current

[0014] DPMZM Dual Parallel Mach-Zehnder Modulator

[0015] EA Electric Amplifier

[0016] EOM Electro-Optic Modulator

[0017] FFT Fast Fourier Transform

[0018] FPC Fiber Polarization Controller

[0019] FPGA Field Programmable Gate Array

[0020] ICP-RIE Inductively Coupled Plasma-Reactive Ion Etching

[0021] IF Intermediate Frequency

[0022] ISAC Integrated Sensing and Communication

[0023] ISAR Inverse Synthetic Aperture Radar

[0024] LD Laser Diode

[0025] LFMW Linear Frequency Modulation Waveform

[0026] LNA Low Noise Amplifier

[0027] LNOI Lithium Niobate on Insulator

[0028] LNPIC Lithium Niobate Photonic Integrated Circuit

[0029] MMI Multimode Interferometer

[0030] mmWave

[0031] MZM Mach-Zehnder Modulator

[0032] OSA Optical Spectrum Analyzer

[0033] OSC Oscilloscope

[0034] PHODIR Photon-Assisted Fully Digital Radar

[0035] PD Photodetector

[0036] RF Radio Frequency

[0037] Si

[0038] SWaPSC size, weight, power consumption, stability, and cost

[0039] TFLN Thin Film Lithium Niobate

[0040] TIA Transimpedance Amplifier

[0041] UV

[0042] VNA Vector Network Analyzer Technical Field

[0043] The present disclosure generally relates to photonic millimeter wave radars. In particular, the present disclosure relates to an integrated lithium niobate photonic millimeter wave radar implemented by utilizing LNPIC. Background Art

[0044] For decades, microwave frequency radio detection and ranging (radar) has been a foundational technology for a variety of applications such as aerial object detection, weather forecasting, resource exploration, and vital signs monitoring [1], [2]. In the upcoming 6G era, mmWave radars operating at frequencies above 30 GHz or even higher and with wider bandwidths are expected to play a key role in ISAC systems that require high-resolution detection and real-time situational awareness, enabling new application scenarios such as indoor sensing, autonomous driving, and vital signs monitoring [3]-[5]. However, the operating frequency and bandwidth of conventional radar systems are usually limited and often trade off against each other, making it challenging to accurately locate, identify, and image objects with both large detection range and high resolution at the same time.

[0045] The emergence of photonics-based radar technology offers a promising solution that overcomes these limitations by processing microwave signals in the optical domain. It fully exploits the advantages of photonic systems, including high frequency, large bandwidth, low transmission loss, reconfigurability, and immunity to electromagnetic interference [6]. [7]. In 2014, the first PHODIR based on mode-locked lasers and optical domain frequency conversion technology was demonstrated [8]. Phase-coded microwave radar waveforms were generated for the detection of non-cooperative aircraft, with a ranging resolution of 23 m, which was significantly limited by the narrow signal bandwidth of 200 MHz. To further improve the resolution, different microwave photonic radar architectures have been proposed, such as frequency doubling [9], optical injection of semiconductor lasers

[10] , photon stretching

[11] , cyclic frequency shifting

[12] , and frequency up-conversion

[13] -

[15] . However, to date, most microwave photonic radars are still built using discrete optoelectronic devices, which have significant disadvantages in terms of SWaPSC.

[0046] Recently, integrated photonics has opened up new opportunities for improving the SWaPSC performance of microwave photonic systems by miniaturizing and integrating multiple photonic devices in a chip-level system

[16] –

[22] . Thanks to compatibility with mature CMOS manufacturing technology, several radars based on Si photonic chips have been realized in the microwave S (2 GHz to 4 GHz), C (4 GHz to 8 GHz), X (8 GHz to 12 GHz), and Ku (12 GHz to 18 GHz) bands

[23] –

[25] , with the largest demonstrated bandwidth being 6 GHz (12 GHz to 18 GHz)

[23] . However, the modulation mechanism used in Si EOM, namely free-carrier depletion, imposes inherent limitations on the performance that can be achieved in Si photonic radars. These limitations include limited bandwidth, nonlinear electro-optical response, and limited extinction ratio, which significantly affect the radar’s operating frequency range and the signal quality of the radar waveform. In addition, active Si photonic devices are prone to two-photon and free-carrier absorption, especially at the excessive optical power levels typically required in microwave photonic systems

[26] . As a result, the integrated photonic radars demonstrated to date cannot reach mmWave frequency bands (>30 GHz), which are highly desirable in future indoor sensing, autonomous driving, and 6G-based imaging and sensing networks.

[0047] In short, conventional photonic radars are mostly implemented as tabletop systems consisting of bulky discrete components, while more compact integrated photonic radars have difficulty reaching the mmWave band due to unsatisfactory bandwidth and signal integrity of the underlying EOM. There is a need in the art to develop an integrated photonic radar to reach the mmWave band.

[0048] The TFLN platform is an excellent choice to address these challenges and bring the operating frequency of integrated photonic radar into the mmWave band. On one hand, TFLNs exhibit a fast and linear Pockels effect, making them well suited for implementing high-speed linear EOMs. On the other hand, the high refractive index contrast of TFLNs allows for tight confinement of optical modes, enabling the implementation of multiple photonic functions in a single TFLN photonic integrated circuit

[27] –

[35] . In recent years, a number of TFLN-based EOMs have been developed, achieving unprecedented performance metrics including modulation bandwidth deep into the mmWave band

[28] ,

[36] –

[38] , CMOS-compatible drive voltage

[27] ,

[39] ,

[40] , small size

[41] , and ultra-high modulation linearity

[42] . Furthermore, the TFLN platform is compatible with cost-effective wafer-level manufacturing techniques

[43] ,

[44] and is equipped with a full range of functional equipment toolboxes including low-loss waveguides

[44] , high-quality factor microresonators

[45] , waveguide crossbars

[46] , delay lines

[47] , etc. These combined achievements pave the way for the TFLN platform to be applied to a variety of future integrated microwave and photonic mmWave applications that require high frequency, large bandwidth, and compact form factor

[48] . Summary of the invention

[0049] This paper provides a photonic millimeter-wave radar system.

[0050] The system includes a light source, a signal generator, a frequency multiplication module, a separation device, a TIA, an LNA and a frequency de-chirping module.

[0051] The light source is used to generate an optical carrier.

[0052] The signal generator is used to generate a first electrical signal with LFMW.

[0053] The frequency multiplication module is used to multiply the frequency of the first electrical signal. The frequency multiplication module includes an amplitude modulation device and a first PD. The amplitude modulation device is a device used to amplitude modulate the optical carrier with the first electrical signal to generate the first optical signal. The first PD is used to convert the first attenuated copy of the first optical signal into a second electrical signal, so that the second electrical signal is generated according to the square of the envelope of the first attenuated copy, so that the instantaneous frequency of the second electrical signal is twice the instantaneous frequency of the first electrical signal. In this system, the amplification of the second electrical signal forms a radar signal for remote sensing, so that the frequency multiplication module reduces the bandwidth requirement of the signal generator in generating the radar signal.

[0054] The amplitude modulation device is selected from the following items and their equivalents: a first EOM configured to be biased at a zero transmission point of the first EOM; a cascade of a third EOM and a first microring filter, the output of the third EOM being connected to the input of the first microring filter, the third EOM being configured to be biased at a full transmission point of the third EOM; and a cascade of a DPMZM and a second microring filter, the output of the DPMZM being connected to the input of the second microring filter, wherein the DPMZM comprises a first MZM and a second MZM, the first MZM and the second MZM being connected in parallel, the first MZM being arranged to receive a first electrical signal, the second MZM being arranged to receive a 90° phase shifted version of the first electrical signal, and wherein the main DCs of the first MZM, the second MZM and the DPMZM are biased at respective maximum transmission points. Preferably, the signal generator is used to further generate a 90° phase shifted version of the first electrical signal received by the second MZM. Optionally, each of the first EOM, the third EOM and the DPMZM is a capacitively loaded traveling wave electrode EOM.

[0055] The separation device is a device for separating the first optical signal into the second optical signal and the third optical signal so that the second optical signal is a first attenuated copy of the first optical signal forwarded to the first PD. The separation device can be an MMI, a directional coupler, or an adiabatic coupler. In some embodiments, the MMI is a 50:50 MMI.

[0056] The TIA is used to amplify the second electrical signal to generate a radar signal, wherein the radar signal is a chirp signal for remote sensing.

[0057] The LNA is used to amplify the reflected echo signal to generate a third electric signal, wherein the reflected echo signal is a signal formed by one or more echoes respectively reflected from one or more objects illuminated by the radar signal in the remote sensing.

[0058] The frequency de-chirp module is used to de-chirp the reflected echo signal and resolve one or more echoes. The frequency de-chirp module includes a second EOM and a second PD. The second EOM is used to amplitude modulate the third optical signal with a third electrical signal to generate a fourth optical signal, wherein the second EOM is configured to be biased at an orthogonal transmission point. The second PD is used to convert the fourth optical signal into a fourth electrical signal, so that the fourth electrical signal is generated according to the square of the envelope of the fourth optical signal. Both biasing the second EOM at the orthogonal transmission point and generating the fourth electrical signal according to the square of the envelope of the fourth optical signal de-chirp the reflected echo signal, and enable the delay of a single echo in the reflected echo signal to be determined from the peak in the spectrum of the fourth electrical signal.

[0059] In addition, in the system, the amplitude modulation device, the second EOM and the separation device are TFLN photonic circuits integrated on the LNPIC.

[0060] In certain embodiments, the radar signal is in the frequency range of 1 GHz to 300 GHz.

[0061] In certain embodiments, the signal generator is a DAC arranged to be input with the digital signal of the LFMW.

[0062] In certain embodiments, the system further includes an ADC for digitizing a fourth electrical signal received from the second PD.

[0063] In certain embodiments, the system further includes a digital signal processor configured to receive the digitized fourth electrical signal and determine a delay of the single echo from the received digitized fourth electrical signal.

[0064] In some embodiments, the system further comprises a transmission antenna and a receiving antenna. The transmission antenna is used to transmit the radar signal to the outside of the system for remotely sensing one or more objects. The receiving antenna is used to receive the reflected echo signal.

[0065] In one embodiment, each of the transmission antenna and the receiving antenna is a horn antenna. In another embodiment, each of the transmission antenna and the receiving antenna is a microstrip antenna, a patch antenna, a slot antenna or a Goubau wire antenna.

[0066] In some embodiments, at least the light source, the signal generator, the LNPIC, the TIA, the LNA, the first PD, the second PD, and the ADC are integrated on an integrated photonic mmWave radar chip.

[0067] In some embodiments, at least the light source, the signal generator, the LNPIC, the TIA, the LNA, the first PD, the second PD, the ADC, and the DSP are integrated on an integrated photonic mmWave radar chip.

[0068] In some embodiments, at least the light source, DAC, LNPIC, TIA, LNA, first PD, second PD, ADC, DSP, transmit antenna and receive antenna are integrated on an integrated photonic mmWave radar chip.

[0069] As shown in the following examples, other aspects of the disclosure are disclosed. BRIEF DESCRIPTION OF THE DRAWINGS

[0070] Figures 1A to 1J 1 and 2 collectively show a conceptual diagram and working principle of an exemplary photonic mmWave radar system disclosed herein, wherein:

[0071] Figure 1A depicts a schematic diagram of an exemplary photonic mmWave radar system;

[0072] Figure 1Bshows a picture of a 4-inch TFLN wafer fabricated for a proof-of-concept demonstration of the disclosed system;

[0073] Figure 1C A comparison picture showing a photonic mmWave radar chip cut from a TFLN wafer and a Hong Kong ten-dollar coin;

[0074] Figure 1D The measurement results for determining the half-wave voltage of the fabricated TFLN EOM are plotted;

[0075] Figure 1E The measurement results used to determine the 3dB electro-optical bandwidth of the fabricated TFLN EOM at 50GHz are plotted;

[0076] Figure 1F The spectrum measured at the output of the first EOM used as an amplitude modulation device is shown (see Figure 1A ), showing a sideband carrier suppression ratio greater than 25dB;

[0077] Figure 1G A first spectrum of the generated radar signal is shown, showing an arbitrarily configurable bandwidth (2 GHz to 10 GHz, centered at 45 GHz);

[0078] Figure 1H shows a second spectrum of the generated radar signal, showing the arbitrarily configurable center frequency (41 GHz to 45 GHz, with a fixed bandwidth of 10 GHz);

[0079] Fig. 1I The measured time domain waveform of the radar signal is plotted; and

[0080] Figure 1J The frequency-time graph of the radar signal is plotted.

[0081] FIG. 2A to FIG. 2C Involving high-resolution photonic mmWave ranging radar, where:

[0082] Figure 2A Depicts the experimental set-up for radar range measurement, where Figure 2A The inset in the figure shows a microscope image of a photonic mmWave radar chip under test;

[0083] Figure 2B The measured distances are plotted against the actual distances for a single target placed in various positions, with inset i) showing the spectrum measured at the output of the second EOM (see Figure 1A), insets ii to viii) show the measured de-chirped electrical spectra for different target distances (the horizontal axis has been converted to distance values ​​for better visualization), inset ix) shows the measured ranging errors for different target distances, and the diagonal lines in the figures correspond to the ideal relationship; and

[0084] Figure 2C The measured distances versus the actual distances for two targets and three targets placed at various positions are plotted, where insets i to v) show the measured de-chirped electrical spectra for various test scenarios, inset vi) shows a top view of the test setup illustrating the relative positions of the antennas and targets, inset vii) shows the measured ranging errors for various test scenarios, and the diagonal lines in the figures correspond to the ideal relationship.

[0085] Figure 3 The experimental results obtained by the high-resolution photonic mmWave speed detection radar are plotted in sub-curves (a) to (j), where: sub-curve (a) plots the target speed measured at different set speeds, showing the marginal deviation from the ideal relationship (dashed line); sub-curve (b) plots the measured speed error at different set speeds; sub-curves (c) to (j) plot two-dimensional speed range diagrams measured at various distances and speeds, and the solid lines on sub-curves (c) to (j) show the set speed values.

[0086] 4A to 4F It involves high-resolution photonic mmWave imaging radar, which:

[0087] Figure 4A A schematic diagram depicting an imaging radar test scenario;

[0088] Figure 4B Radar imaging results of various numbers and arrangements of small metal corner reflectors with a size of 3 cm × 3 cm are provided;

[0089] Figure 4C The radar imaging results of a large aircraft model imaged at various azimuth rotation angles are plotted;

[0090] Figure 4D The radar imaging results of a medium-sized aircraft model imaged at various azimuth rotation angles are plotted;

[0091] Figure 4E Plots radar imaging results of a small aircraft model imaged at various azimuth rotation angles; and

[0092] Figure 4F The radar imaging results of the doll imaged at various azimuth rotation angles are plotted.

[0093] FIG. 5A to FIG. 5CDepicted are three implementations of amplitude modulation devices used in photonic mmWave radar systems:

[0094] Figure 5A depicts a first EOM biased at a zero transmission point;

[0095] Figure 5B depicts a third EOM cascaded with the first microring filter, wherein the third EOM is biased at a full transmission point; and

[0096] Figure 5C A cascade of a DPMZM and a second microring filter is depicted, wherein the DPMZM includes a first MZM and a second MZM, and the first MZM and the second MZM and a main DC of the DPMZM are biased at respective maximum transmission points.

[0097] Figure 6 Depicted is a schematic diagram of a traveling wave electrode EOM that may be used to implement capacitive loading of a first EOM, a second EOM, and a third EOM.

[0098] Those skilled in the art will appreciate that elements in the figures are illustrated for simplicity and clarity and have not necessarily been drawn to scale. DETAILED DESCRIPTION

[0099] The present disclosure will be described more fully below with reference to the accompanying drawings. However, the present disclosure may be implemented in many different forms and should not be construed as limited to the embodiments described herein.

[0100] The present disclosure provides a photon mmWave radar system for remote sensing. The system uses mmWave signals (frequency range of 30 GHz to 300 GHz) for remote sensing. The photon mmWave radar system is implemented based on a TFLN photon circuit, and the TFLN photon circuit is integrated on an LNPIC to have the above-mentioned various advantages. By further integrating the LNPIC with other circuit components made of other semiconductor materials, an integrated photon mmWave radar system implemented as a TFLN photon radar chip is formed. In order to show and illustrate the disclosed system, an exemplary photon mmWave radar system operating in the mmWave V band (frequency range of 40 GHz to 75 GHz) is described in detail below. The exemplary system is preferably and advantageously implemented as a TFLN photon radar chip. However, the present disclosure is not limited to operating the disclosed system in the mmWave V band. The disclosed system can be implemented to operate in any segment of the mmWave band (30 GHz to 300 GHz).

[0101] For proof of concept demonstration and experimental verification, the example system is implemented as a TFLN photonic radar chip. The TFLN photonic radar chip is manufactured using a 4-inch wafer-level process and has a frequency multiplication module and a frequency de-chirp module. The frequency multiplication module is used for mmWave radar waveform generation, and the frequency de-chirp module is used for echo signal reception. Benefiting from the wide bandwidth of all photonic components in the disclosed chip with a systematic filter-free design, the center frequency and bandwidth of the generated radar waveform can be arbitrarily configured in a wide range. Therefore, the disclosed integrated photonic mmWave radar has the following characteristics: continuous on-demand tunability of center frequency and bandwidth, which is previously limited only by the bandwidth of the electrical amplifier. In the case of the current discussion, the radar waveform is located within 40 GHz to 50 GHz, which is only limited by the electrical amplifier used in the radar chip. The high carrier frequency and large bandwidth enable one to achieve multi-target ranging with a range resolution of 1.50 cm, velocity measurement with a resolution of 0.067 m / s, and ISAR imaging with a two-dimensional resolution of 1.50 cm×1.06 cm.

[0102] With the help of Figures 1A to 1J The disclosed system is illustrated. Figures 1A to 1J Collectively, a conceptual illustration and operating principles of an exemplary photonic mmWave radar system 100 are shown. Radar waveform generation is also shown.

[0103] A. Development of the Disclosed System

[0104] Figure 1A A schematic diagram of an exemplary photonic mmWave radar system 100 is depicted. It is preferred and advantageous to implement the system 100 on an integrated photonic mmWave radar chip, such as by Figure 1A The system 100 is highlighted by the dashed line defining the system 100 .

[0105] In the system 100, LFMW, which is a linear chirp signal, is selected as the radar waveform in the present disclosure, because LFMW is characterized by high ranging resolution, constant modulus and Doppler tolerance, and direct frequency de-linear chirp processing of the echo waveform, thereby greatly reducing the sampling rate requirement of the radar receiver. In order to generate the mmWave LFMW signal at the transmitter side, the low-frequency microwave LFMW signal as an electrical signal is frequency-up-converted to an optical frequency. Specifically, by using a first high-speed TFLN amplitude modulator, the microwave LFMW signal is modulated to a frequency of f cFor illustration purposes, the high-speed TFLN amplitude modulator is implemented as an EOM fabricated on an LNPIC. This basic microwave LFMW signal can be generated relatively easily by, for example, a CMOSDAC. The DAC exhibits the following characteristics: a linear variation from f1 to an instantaneous frequency f1+kt of f1+kT with a bandwidth of B1=kT, where T is the waveform period. Biasing the EOM at the zero transmission point results in a CS-DSB modulation process that projects the input microwave signal to frequencies f1+kT, respectively. c +f1+kt and f c -f1-kt in two LFMW optical sidebands. The modulated optical signal is then separated into two paths by a separation device. The separation device can be implemented as an MMI, a directional coupler, or an adiabatic coupler. If the separation device is an MMI, typically the MMI is a 50:50 MMI. The optical signal in the upper path is detected by the first high-speed PD to generate a mmWave radar waveform, whose initial frequency (2f1) and bandwidth (B2=2kT) are both twice the DAC input electrical signal. The generated radar waveform is then amplified by the TIA and transmitted into free space through the transmission antenna. When the transmitted radar waveform encounters a target, the waveform is reflected with a time delay of τ. The reflected echo waveform is collected by the receiving antenna, amplified by the LNA, and sent to a frequency de-chirp module, which includes a second high-speed TFLN amplitude modulator, which is implemented as an EOM manufactured on the same LNPIC. The input port of the EOM is connected to the lower output path of the separation device, so its characteristics are exhibited as having two carrier frequencies f c +f1+kt and f c +f1+kt, which are then modulated by the amplified reflected echo signal with an instantaneous frequency of 2f1+2kt-2kτ. By setting the EOM at the orthogonal transmission point, four new optical sidebands are generated, two of which are located near the two carriers with a frequency of f c +f1+kt-2kτ and f c The presence of the last two frequencies allows one to achieve frequency de-chirping and obtain the low frequency target information (2kτ) by beating the two related sidebands with two nearby carriers at the low speed PD and further processing the output of the low speed PD using a low speed ADC.

[0106] Based on the above explanation of the operating principle of the system 100 , an embodiment of the disclosed photonic mmWave radar system is described as follows.

[0107] The photonic mmWave radar system 100 is used to remotely sense one or more objects 198, such as nearby vehicles. Specifically, the photonic mmWave radar system 100 is configured to generate a radar signal 187 for remote sensing of one or more objects 198, and analyze the reflected echo signal 191 reflected from one or more objects 198 illuminated by the radar signal 187 in the remote sensing. Exemplarily, the photonic mmWave radar system 100 includes a light source 161, a signal generator 162, a frequency multiplication module 106, a separation device 115, a TIA 125, an LNA 135, and a frequency de-chirp module 107. The frequency multiplication module 106 includes an amplitude modulation device 110 and a first PD 120. The frequency de-chirp module includes a second EOM 140 and a second PD 145. Utilizing the advantages provided by using lithium niobate, the amplitude modulation device 110, the second EOM 140, and the separation device 115 are TFLN photonic circuits integrated on the LNPIC 105.

[0108] The light source 161 is used to generate the optical carrier 181. Generally, the light source 161 is a laser source, so that the optical carrier 181 is a monochromatic light beam. In some embodiments, the laser source is implemented by LD.

[0109] The signal generator 162 is used to generate a first electrical signal 182 having a LFMW. The first electrical signal 182 is a microwave LFMW signal as described above. In some embodiments, the signal generator 162 is a DAC arranged to input a digital signal of the LFMW.

[0110] The frequency multiplication module 106 is used to multiply the first electrical signal 182 (e.g., double, quadruple, etc.). The frequency-converted signal can be used to generate a radar signal 187. Advantageously, in the process of generating the radar signal 187, the frequency up-conversion of the first electrical signal 182 reduces the bandwidth requirement of the signal generator 162. This advantage is particularly important when the radar signal 187 is expected to operate in the mmWave frequency range (30 GHz to 300 GHz). Even if the bandwidth requirement is halved, it is generally a substantial reduction in demand. In addition, this advantage is particularly useful when the signal generator 162 is implemented by a DAC, where the aforementioned bandwidth requirement is a sampling rate requirement of the DAC.

[0111] The frequency multiplication module 106 includes an amplitude modulation device 110 and a first PD 120. The amplitude modulation device 110 is a first high-speed TFLN amplitude modulator as described above. The amplitude modulation device 110 may be implemented as an EOM (referred to as a first EOM). Figure 5AA first EOM 510 is depicted as an implementation of the amplitude modulation device 110. In the operation of the frequency multiplication module 106 as shown below, it is assumed that the amplitude modulation device 110 is implemented by the first EOM 510. Other implementations of the amplitude modulation device 110 will be discussed later.

[0112] The first EOM 510 is used to amplitude modulate the optical carrier 181 with the first electrical signal 182 to generate a first optical signal 183. The first optical signal 183, E1(t), is represented by

[0113]

[0114] Where: E0 and f c are the amplitude and frequency of the optical carrier 181; V1, f1+kt, k, T and B1 (B1=kT) are the amplitude, instantaneous frequency, chirp rate, waveform period and bandwidth of the first electrical signal 182; V DC1 is the DC bias voltage applied to the first EOM 510, and V π is the half-wave voltage of the first EOM 510. In particular, the first EOM 510 is configured to be biased at the zero transmission point so that the first optical signal 183 is a CS-DSB signal. (For background information on various settings of EOMs and transmission points, see

[50] ). Therefore, the first optical signal 183 can be written as

[0115]

[0116] Among them: J n is a first-order Bessel function of the first kind; and β1=πV1 / V π is the corresponding RF modulation index of the first EOM 510.

[0117] The first PD 120 is used to convert the first attenuated replica 184 of the first optical signal 183 into a second electrical signal 186. The first attenuated replica 184 of the first optical signal 183 is represented as the second optical signal 184. The second optical signal 184 is a replica of the first optical signal 183 with the signal power scaled down. Since the PD is a transducer that senses the intensity of an input optical signal to generate an output electrical signal, the second electrical signal 186 is generated according to the square of the envelope of the second optical signal 184. The nonlinear operation of the first PD 120 causes the instantaneous frequency of the second electrical signal 184 to be twice the instantaneous frequency of the first electrical signal 182. The second electrical signal 184, I1(t), represented by the current output by the first PD 120 is given by the following equation:

[0118]

[0119] The initial frequency (f2=2f1)) and bandwidth (B2=2kT=2V1)) of the generated radar waveform are twice that of the first electrical signal 182. Those skilled in the art will appreciate that the second electrical signal 184 given by EQN.(3) has the DC offset removed (e.g., by AC coupling at the output of the first PD 120) and is suitable for forming the radar signal 187.

[0120] The separation device 115 is a device for separating the first optical signal 183 received from the first EOM 510 into a second optical signal 184 and a third optical signal 185, so that the second optical signal 184 is a first attenuated replica 184 of the first optical signal 183. The second optical signal 184 generated by the separation device 115 is forwarded to the first PD 120. The third optical signal 185 is used as a reference signal for estimating the echo delay of the echo in remote sensing. The separation device 115 is an MMI, a directional coupler, an adiabatic coupler, or any equivalent thereof. In some embodiments, the separation device 115 is selected as an MMI, and the MMI is a 50:50 MMI, so that the second optical signal 184 and the third optical signal 185 have the same light intensity level.

[0121] The second electrical signal 186 provided in the form of an electric current is amplified by the TIA 125 to generate a radar signal 187 .

[0122] The radar signal 187 is a chirp signal for remote sensing of one or more objects 198. Therefore, the system 100 receives a reflection echo signal 191 for echo delay estimation. The reflection echo signal 191 is a signal formed by one or more echoes reflected from one or more objects 198 illuminated by the radar signal 187 in remote sensing.

[0123] The LNA 135 is used to amplify the reflected echo signal 191 to generate a third electrical signal 192. Advantageously, interfering outliers (such as out-of-band signals) may also be filtered out by the LNA 135 to enhance the detectability of one or more echoes.

[0124] The frequency de-chirp module 107 is used to de-chirp the reflected echo signal 191 and resolve one or more echoes. The frequency de-chirp module includes a second EOM 140 and a second PD 145. The second EOM 140 is a second high speed TFLN amplitude modulator as described above.

[0125] The second EOM 140 is used to amplitude modulate the third optical signal 185 received from the separation device 115 with the third electrical signal 192 to generate a fourth optical signal 193. The second EOM 140 is configured to be biased at an orthogonal transmission point. The second PD 145 is used to convert the fourth optical signal 193 into a fourth electrical signal 194. Similar to the first PD 120 generating the second electrical signal 186, the fourth electrical signal 194 is generated according to the square of the envelope of the fourth optical signal 193. In addition, the second EOM 140 is biased at the orthogonal transmission point, and the fourth electrical signal 194 is generated according to the square of the envelope of the fourth optical signal 193, both of which de-chirp the reflected echo signal 191 and enable the delay of a single echo in the reflected echo signal 191 to be determined from the peak in the spectrum of the fourth electrical signal 194.

[0126] The determination of the echo delay by the frequency dechirp module 107 is further explained as follows by mathematical derivation for the special case where the reflected echo signal 191 contains one echo. Extending the mathematical derivation to the case of multiple echoes is straightforward.

[0127] For the special case under consideration, let τ be the time delay of a single echo contained in the reflected echo signal 191. The reflected echo signal 191 is received and amplified by the LNA 135 to form a third electrical signal 192. By applying the third electrical signal 192 to the second EOM 140 and biasing the second EOM 140 at the orthogonal transmission point, the fourth optical signal 193 obtained at the output end of the second EOM 140 is represented as

[0128]

[0129] in

[0130]

[0131] as well as

[0132]

[0133] Where β2=πV2 / V π V1 and V2 are the corresponding RF modulation index of the second EOM 140 and the amplitude of the third electrical signal 192 (i.e., the amplified reflected echo signal 191). Subsequently, the second PD 145 generates a fourth electrical signal 194 according to the square of the envelope of the fourth optical signal 193. Using a nonlinear operation equivalent to obtaining the square of the signal envelope, the second PD 145 generates a fourth electrical signal 194 at f c +f1+kt and f c +f1+kt-2kτ(or f c -f1-kt+2kτ and f c= -f1-kt) under the beat frequency of the two optical side bands to restore the fourth electrical signal 194. It can be seen that it is expressed as I de The fourth electrical signal 194 of (t) satisfies the following relationship:

[0134] I de (t)∝4J0(β2)J1(β2)·cos(2π(2kτ)t+2π(2f1-kτ)τ). (5)

[0135] The frequency (f de ) is 2kτ. Therefore, the frequency f de Proportional to the time delay of the echo.

[0136] In addition to the first EOM 510, other implementations of the amplitude modulation device 110 are described in detail as follows.

[0137] Figure 5B A second implementation of the amplitude modulation device 110 is depicted. In this implementation, the amplitude modulation device 110 is a cascade 520 of a third EOM 521 and a first micro-ring filter 522. The output of the third EOM 521 is connected to the input of the first micro-ring filter 522. Unlike the first EOM 510, the third EOM 521 is configured to be biased at the full transmission point of the third EOM 521.

[0138] The operating principle and advantages of the cascade 520 are explained as follows. By biasing the third EOM 521 at the full transmission point (so as to suppress the first-order optical sideband in the signal modulation) and combining the first microring resonator 522 to filter out the optical carrier 181, a carrier-suppressed second-order sideband optical signal can be obtained as the first optical signal 183. After photodetection by the first PD 120, frequency quadrupling can be achieved by beating the two second-order sidebands. Subsequently, the generated frequency quadrupled LFMW is used to generate a radar signal 187, which is transmitted into free space for remote sensing.

[0139] Figure 5CA third implementation of the amplitude modulation device 110 is depicted. In this implementation, the amplitude modulation device 110 is a cascade 530 of a DPMZM 531 and a second microring filter 532. The output of the DPMZM 531 is connected to the input of the second microring filter 532. The DPMZM 531 includes a first MZM 535 and a second MZM 536. The first MZM 535 and the second MZM 536 are connected in parallel. The first MZM 535 is arranged to receive the first electrical signal 182. The second MZM 536 is arranged to receive a 90° phase shifted version 182s of the first electrical signal 182. In particular, the main DCs of the first MZM 535, the second MZM 536 and the DPMZM 531 are biased at respective maximum transmission points. Preferably, the signal generator 162 is configured to further generate a 90° phase-shifted version 182s of the first electrical signal 182 , such that the second MZM 536 receives the 90° phase-shifted version 182s from the signal generator 162 .

[0140] The operating principle and advantages of the cascade 530 are provided as follows. Replacing the first EOM 510 with a DPMZM 531 provides a factor of multiplying the signal frequency and bandwidth by eight times. It can greatly increase the frequency and bandwidth of the resulting radar signal 187. In detail, the first electrical signal 182 is processed with a 90° electrical mixer 537. The original first electrical signal 182 is sent to the RF input of the first MZM 535, and the 90° phase-shifted version of the first electrical signal 182 is sent to the corresponding RF input of the second MZM 536. By biasing the main DC bias of the two MZMs 535, 536 and the DPMZM 531 at the maximum transmission point, a fourth-order sideband signal added to the optical carrier 182 is obtained. The cascaded microring resonator acting as the second microring filter 532 is used to suppress the optical carrier 182. Therefore, an LFMW with a frequency eight times is generated as the second electrical signal 186 after photodetection, and can then be used to generate the radar signal 187.

[0141] It should be noted that the amplitude modulation device 110 can be composed of FIG. 5A to FIG. 5C The invention can be implemented by the circuits / devices / arrangements shown in the specification and the equivalents of these circuits / devices / arrangements.

[0142] In addition, it should be noted that FIG. 5A to FIG. 5CThe circuit / device / arrangement shown in provides amplitude modulation by summing different optical signals so that the first optical signal 183 is amplitude modulated by constructive interference or destructive interference. The signal summing process not only modulates the amplitude, but also modulates the phase of the optical carrier 181. It should be understood that the amplitude modulation device 110 as used herein provides amplitude modulation of the optical carrier 181 as a primary function and can modulate the phase of the optical carrier 181 in forming the first optical signal 183. Nevertheless, the undesirable phase modulation may not affect the intended operation of the system 100. The first optical signal 183 is processed in part by the first PD 120 and in part by the second EOM 140 (for amplitude modulation) after the second PD 145. The undesirable modulation in the phase of the optical carrier 181 is removed during the photoelectric detection process.

[0143] When implementing the amplitude modulation device 110, the first EOM 510 or the third EOM 521 may be involved. In some practical situations, it is necessary to achieve millimeter-level resolution in remote sensing. In some embodiments, each of the first EOM 510 and the third EOM 521 is a capacitively loaded traveling wave electrode EOM

[49] . This means that if the first EOM 510 (third EOM 521) is involved in implementing the amplitude modulation device 110, the first EOM 510 (third EOM 521) is implemented as a capacitively loaded traveling wave electrode EOM 610. Figure 6 A schematic diagram of a capacitively loaded traveling wave electrode EOM 610 is depicted in . By using a capacitively loaded traveling wave electrode EOM 610 with a greater bandwidth, the system 100 can be used for higher mmWave frequency bands with wider bandwidth, thereby achieving millimeter resolution in long-range sensing and imaging.

[0144] It should be noted that since the reflected echo signal 191 has the same bandwidth as the radar signal 187 , if the first EOM 510 (third EOM 521 ) is implemented as a capacitively loaded traveling wave electrode EOM 610 , it is preferred that the second EOM 140 is also implemented as a capacitively loaded traveling wave electrode EOM 610 .

[0145] Other implementation details of the system 100 are described as follows.

[0146] To determine the echo delay of one or more echoes, preferably, the system 100 further includes an ADC 171 for digitizing the fourth electrical signal 194 received from the second PD 145. The digitized fourth electrical signal 195 can be captured by a computer or computing processor for determining the echo delay.

[0147] Since the dechirping of the reflected echo signal 191 is handled by the frequency dechirp module 107 in the optical domain, the ADC 171 does not need to process the chirped signal having a significantly wider bandwidth than the fourth electrical signal 194. Advantageously, it reduces the bandwidth requirement or conversion rate requirement of the ADC 171.

[0148] In some embodiments, the system 100 further includes a DSP 172 for echo delay estimation. The DSP 172 is configured to receive the digitized fourth electrical signal 195 and determine the echo delay of one or more echoes from the received digitized fourth electrical signal 195.

[0149] The system 100 may further include a transmission antenna 127 and a receiving antenna 131. The transmission antenna 127 is used to transmit a radar signal 187 to the outside of the system 100 for remotely sensing one or more objects 198. The receiving antenna 131 is used to receive a reflected echo signal 191. The transmission antenna 127 and the receiving antenna 131 may be implemented as horn antennas for high directivity. The transmission antenna 127 and the receiving antenna 131 may also be implemented as microstrip antennas, patch antennas, slot antennas, or Goubau line antennas for integrating the system 100 on a radar chip co-packaged with the LNPIC 105, or even directly integrated on the LNPIC 105. Other suitable types of antennas may also be used.

[0150] The design of the transmission antenna 127 and the receiving antenna 131 is affected by the frequency selection of the radar signal 187. The radar signal 187 can be designed to have a signal frequency in the frequency range from 40 GHz to 75 GHz (i.e., V-band). However, the present disclosure is not limited to this frequency range in generating the radar signal 187. In general, the signal frequency of the radar signal 187 can be selected from the microwave and mmWave frequency ranges of 1 GHz and 300 GHz.

[0151] As described above, it is preferred and advantageous to implement system 100 on an integrated photonic mmWave radar chip. In one embodiment, at least light source 161, signal generator 162, LNPIC 105, TIA 125, LNA 135, first PD 120, second PD 145, and ADC 171 are integrated on the radar chip. In another embodiment, the radar chip is further integrated with DSP 172. In yet another embodiment, the radar chip is not only integrated with DSP 172, but also further integrated with transmit antenna 127 and receive antenna 131.

[0152] It should be noted that the PDs may be directly integrated on the LNPIC 105 using heterogeneous integration techniques like micro-transfer printing or die-to-wafer bonding. In some embodiments, the first PD 120 and the second PD 145 are implemented on the LNPIC 105 .

[0153] B. Applications of the Disclosed System

[0154] For illustration, consider again the special case where there is only a single echo in the reflected echo signal 191. After the echo delay is determined by the system 100, the range, speed, and imaging information of the detected target can be obtained through subsequent data processing. It is foreseeable that the application scenario of the disclosed integrated photonic mmWave radar system is to simultaneously provide high-resolution range / speed detection and imaging capabilities in future autonomous vehicles, which is the key to enhancing safety, perception, and decision-making processes in autonomous driving.

[0155] As mentioned above, the f of the de-chirped electrical signal de = 2kτ, which is proportional to the echo delay. Therefore, the range (R1) between the transmission antenna 127 (or the reception antenna 131) and the detected target can be equal to cτ / 2, which is written as

[0156]

[0157] Where c is the speed of light. Therefore, ranging detection can be achieved. If f de Equal to the minimum resolvable frequency difference (f 最小 =1 / T), the detected range is called the theoretical ranging resolution (ΔR), which can be expressed as

[0158]

[0159] In addition, the radar implemented by the system 100 has a speed detection capability. When a target moves a distance Δr toward or away from the radar line of sight, the received signal will experience an additional delay (Δτ) introduced by the Doppler shift compared to the echo signal received when the object is stationary. This additional phase change in the de-chirped electrical signal (i.e., the fourth electrical signal 194) can be expressed as

[0160]

[0161] where f rc is the center frequency of radar signal 187, λ is the wavelength of radar signal 187, and T is the period of the chirped transmission signal. The phase exhibits a linear response to small changes in the distance of the target. The phase difference measured across two consecutive chirps can be used to estimate the velocity of the target. Therefore, the estimated velocity obtained from the measured phase difference is expressed as

[0162]

[0163] Based on the characteristics of Fourier transform, the peak phase of the Fourier spectrum represents the initial phase of the signal. However, when using the "frame" (T f=1 / MT) to measure the velocities of multiple objects, the values ​​containing phasor components from different targets at the peaks can be distinguished by performing an FFT on the sequence corresponding to the range-FFT peaks, which is usually called a Doppler-FFT. Equal to the minimum resolvable phase difference The detected velocity is then called the theoretical velocity resolution (Δv), which can be calculated as

[0164]

[0165] Next, an ISAR system is demonstrated using an integrated photonic mmWave radar chip. A turntable model is used to analyze and simplify the motion of targets in ISAR imaging. The imaging process involves a relative rotation between the radar and the target, which can be observed in the joint range-Doppler domain. The detected target is detected with ω t To process the de-chirped electrical signal, we use a low-speed OSC to perform multi-cycle sampling equivalent to a CPI duration T i . The collected data is then rearranged into a two-dimensional matrix (M×N) of delay time (M) and number of pulses (N). By exploiting the relationship between delay time and distance, range compression can be achieved through FFT, which provides frequency domain information. Therefore, the distance can be calculated based on the distance formula (EQN.(6)). The position of the amplitude peak in the range envelope represents the range position of the main scattering point. The translation component of the target relative to the radar is useless for ISAR imaging, so the range compression data must be motion compensated (including range alignment and phase compensation) to eliminate the phase term. At each range unit, we perform Fourier transform on the reflected signal of N pulses, generate N-point Doppler frequency domain and achieve azimuth compression. Therefore, by applying Fourier transform to both range and azimuth dimensions, an image of the target under test can be obtained. The resolution in the range dimension remains the same as that in ranging detection, while the resolution in the azimuth dimension (ΔA) is expressed as

[0166]

[0167] where f a Equal to the minimum resolvable frequency difference (f 最小 =1 / T i ), f rc is the center frequency of the radar waveform, and Δθ is the accumulated angle during radar detection. The azimuth resolution is determined by the total number of pulses (N) within the CPI.

[0168] C. Proof of Concept Demonstration of the Disclosed System

[0169] Figure 1BShown is a picture of a 4 inch LNOI wafer we fabricated, containing 1.50 cm x 1.50 cm pieces (21 pieces in total) of various passive and active components. The wafer was patterned by a UV stepper lithography system and etched by an ICP-RIE system, followed by a standard metallization process. Figure 1C An example TFLN photonic radar chip cut from a 4-inch LNOI wafer is shown, showing the following features: it has a footprint much smaller than a Hong Kong $10 coin. Figure 1D and Figure 1E As shown, the fabricated TFLN EOM has a measured half-wave voltage of approximately 2.8 V and a 3dB electro-optical bandwidth of over 50 GHz, both of which are important indicators for achieving broadband and high-fidelity mmWave radar signal generation and echo processing.

[0170] We first demonstrated that our photonic mmWave radar chip is capable of generating high-quality LFMW signals in the mmWave V-band with arbitrarily configurable center frequency and bandwidth. Figure 1F FIG. 1 shows the CS-DSB spectrum measured at the output port of the first EOM 510 when the bias is zero, showing a sideband-carrier suppression ratio of more than 25 dB due to the excellent extinction ratio of the TFLN EOM. It ensures a low residual fundamental component in the radar signal 187. After the beat CS-DSB optical signal at the first PD 120, a LFMW signal doubled from the driving electrical signal is generated, as shown in FIG. Figure 1G and Figure 1H As shown in the electrical spectrum measured in . Since no optical or electrical filters are involved in our photonic mmWave radar, the bandwidth and center frequency of the generated radar signal 187 can be arbitrarily selected and continuously tuned over a wide range. Figure 1G 4 shows a generated radar signal having a fixed center frequency of 45 GHz and having a bandwidth increasing from 2 GHz to 10 GHz in steps of 2 GHz, while Figure 1H The corresponding signals with different center frequencies (41 GHz to 45 GHz with a step size of 1 GHz) and a fixed bandwidth of 10 GHz are depicted. In the remainder of this disclosure, we select the entire 41 GHz to 50 GHz range as the LFMW radar waveform to achieve the best detection resolution, which is inversely proportional to the bandwidth and center frequency of the radar waveform. Fig. 1I The time domain waveform of the radar signal 187 recorded directly using a real-time oscilloscope is shown. Figure 1J The corresponding frequency-time plot extracted from the time-domain radar waveform using a short-time Fourier transform is shown in . The frequency of the radar waveform increases linearly from 40 GHz to 50 GHz over a time period of 4 μs, corresponding to a linear frequency chirp of 2.5 GHz / μs, which matches the measured electrical spectrum very well.

[0171] D. Example of radar implemented with a public radar chip

[0172] D.1. Photonic mmWave ranging radar

[0173] We demonstrated high-resolution ranging using our photonic mmWave radar chip. FIG. 2A to FIG. 2C A high-resolution photonic mmWave ranging radar is collectively described, along with experimental results obtained with the radar.

[0174] Figure 2A The experimental assembly is shown, with the middle inset showing a light microscope image of the fabricated chip. Figure 2B and Figure 2C The measured range results for single and multiple (up to three) targets placed at various distances from the antenna are respectively illustrated. Figure 2B The inset i of FIG. 1 shows a representative spectrum measured at the output of the second EOM 140, wherein two spectral peaks separated by approximately 45 GHz each contain an optical carrier (f c -f1-kt and f c +f1+kt) and the sidebands generated by the echo modulation (f c -f1-kt+2kτ and f c +f1+kt-2kτ). Due to the limited resolution of the OSA, closely located carriers and sidebands cannot be distinguished in the spectrum, but can be easily de-chirped into low-frequency and narrow-band signals in the MHz level and detected using the low-frequency PD of the second PD 145. It greatly reduces the sampling rate requirement of the ADC in the oscilloscope that records the final time domain waveform of the frequency de-chirped IF signal. After FFT processing, the electrical spectrum of the de-chirped IF signal is obtained, which is directly converted into the target range (R1) ( Figure 2B Illustrations ii to viii and Figure 2C The average range ambiguity (3dB bandwidth) of the ranging measurement is 1.71cm (285kHz), which is very consistent with the theoretical range (frequency) resolution of 1.50cm (250kHz). In addition, the undesired sidelobe suppression ratios of the range spectra are all above 6.5dB, indicating low microwave signal crosstalk in the chip and good detection capability for small and weak targets. Figure 2B and Figure 2C In the main screen of , we summarize and compare the single target ( Figure 2B ) and multi-target ( Figure 2C) shows accurate and linear ranging performance. For a large dynamic range of 30 cm to 420 cm, the measured distance error for a single target (inset ix of Figure 2b) is within ±0.15 cm. This ranging error can be further reduced by reducing the linear chirp rate of the radar waveform. Our photonic mmWave radar also provides excellent ranging performance when detecting multiple targets ( Figure 2C Both the two-target and three-target can be clearly distinguished, and the measured distance error is within ±0.15cm ( Figure 2C Importantly, we show that the photonic mmWave radar can distinguish between two targets that are only 1.90 cm apart ( Figure 2C The yellow squares in the figure and inset i) are in good agreement with the theoretical resolution of 1.50 cm.

[0175] D.2. Photonic mmWave speed detection radar

[0176] In addition to distance detection, our photonic mmWave radar is also capable of high-resolution velocity detection, which relies on measuring the Doppler shift in the return signal caused by target motion. To verify this capability, Figure 3 The experimental results obtained by high-resolution velocity detection are plotted. Figure 3 As depicted in sub-curve (a), a small balancing car with adjustable speed is used as the detection target. Figure 3 The points in sub-graph (a) of the graph represent the relationship between the measured speed and different set speed values, while Figure 3 Sub-figure (b) shows the corresponding velocity measurement errors (all within ±0.017 m / s), revealing the high velocity detection fidelity over a wide velocity range of 0–1.2 m / s. Most notably, our photonic mmWave radar chip can successfully detect small velocities as low as 0.056 m / s (corresponding to a small Doppler shift). Figure 3 As shown in the sub-curves (c) to (j) of Fig. 1, further applying a two-dimensional Fourier transform to the de-chirped electrical waveform can simultaneously extract the target's speed and range information, where the vertical (speed) span of the signal is very consistent with the theoretical speed resolution of 0.067m / s. Such multi-dimensional environmental sensing capabilities can find applications in a variety of fields including automotive systems, traffic management, and object tracking.

[0177] D.3. Photonic mmWave ISAR for Imaging

[0178] Finally, we show that photonic mmWave radar can support high-resolution imaging tasks. 4A to 4F Together they demonstrated the purpose of this radar.

[0179] Figure 4AFigure 1 depicts the configuration of the ISAR used in the experiment, where the transceiver antenna is fixed and the target undergoes a rotation speed of 1 revolution per second during detection. To characterize the basic imaging capabilities of photonic mmWave ISAR, we first placed several small metal corner reflectors ( Figure 4B Illustration i) in different arrangements ( Figure 4B ) is placed on the turntable. The resulting image ( Figure 4B The insets ii to iv) of Figure 1 clearly reveal the number, size and relative positions of the corresponding metal plates, which are in good agreement with the actual setup shown in the upper left inset in each case. Figure 4B The radar image in the subsequent screen shows a top-down view of the target being imaged, where the vertical axis represents the radial distance from the radar antenna (range) and the horizontal axis corresponds to the azimuth position (cross-range). The results show that our photonic mmWave ISAR is able to simultaneously detect multiple closely spaced targets ( Figure 4B To further demonstrate the ability to image real-world objects of varying shapes, sizes, and poses, we used a number of more complex objects, including large aircraft (45 cm × 49 cm, Figure 4C ), medium-sized aircraft (33cm×34cm, Figure 4D ), small aircraft (21cm×24cm, Figure 4E ) and dolls (30cm×20cm, Figure 4F ), to replace the metal plate. FIG. 4C to FIG. 4F The processed images in show that our system can successfully resolve the structural outlines of these targets at various rotation angles. It should be noted that the imaging signals at the bottom of these images are weak and sometimes even non-existent, since they correspond to features far from the radar antenna and the return signals from these areas may be blocked by other thicker parts of the target. However, our results show that small features (such as the 5 cm tail of a small aircraft ( Figure 4E Illustration ii) and the doll's 0.7 cm wide arms and legs ( Figure 4F ), demonstrating the successful implementation of centimeter-resolution photonic mmWave ISAR using our integrated TFLN chip.

[0180] E. Experiment

[0181] E.1. Overview of the design and fabrication of the equipment used for testing

[0182] The devices were simulated using Ansys Lumerical mode and high frequency simulation software (Ansys HFSS). 500nm TFLN 4-inch from NANOLN was used to fabricate the designed devices. First, a SiO2 etch hard mask was deposited on the LNOI surface by plasma enhanced chemical vapor deposition. The waveguides, EOM, and MMI (1.5cm×1.5cm) were patterned die by die on a 4-inch LNOI wafer with a resolution of 400nm using an ASML UV stepper lithography system (NFF, HKUST). The exposed patterns were then transferred to the SiO2 hard mask using a standard fluorine-based dry etching process and the etched wafer was patterned using an optimized Ar + The auxiliary ICP-RIE process was transferred to the TFLN layer with a waveguide height of 250nm and a plate height of 250nm. Subsequently, the second stepper photolithography patterned the electrode layer after removing the residual SiO2 mask and re-depositing the SiO2 coating. After the metal evaporation and lift-off process, the ground-signal-ground electrodes with a gap of 5.5μm were obtained, which can ensure good electro-optical modulation efficiency and optical losses caused by lossy metal. Finally, the chip was carefully cleaved and end-polished, with a coupling loss of 4dB / facet.

[0183] E.2. Characteristics of devices on the TFLN platform

[0184] In order to test the optical performance of the TFLN chip manufactured, the optical input signal from a tunable telecommunication laser source (Santec TSL-510) is coupled to the chip using a lensed fiber. The optical signal of output is collected by another lensed optical fiber and sent to a low-speed PD (125MHz New Focus 1811). By measuring the optical transmission spectrum of the racetrack-shaped resonator and fitting with the Lorentz function to estimate the optical loss of the TFLN waveguide. The resonator load quality factor measured exceeds one million, indicating that the waveguide propagation loss is about 0.3dB / cm.

[0185] In order to measure the half-wave voltage (V π ), a kilohertz electrical triangle waveform generated by an arbitrary waveform generator (AWG, RIGOL DG4102) was applied to the ground-signal-ground electrodes of the EOM through a probe (GGBindustries, 50 GHz). The output optical signal of the EOM was detected using the same low-speed PD and monitored using a low-speed oscilloscope (RIGOL DS6104). For the electro-optical S 21For response measurement, a swept frequency electrical signal generated from a 53GHz VNA (Keysight E5080B) is sent to the EOM with a 50Ω load. A high-speed PD (XPDV2120R) is used to detect the modulated optical signal. The recovered electrical signal is sent back to the input port of the VNA. After calibrating the frequency response of the probe, cable, and PD, the S of the EOM can be obtained. 21 The frequency response, in this case shows a 3dB bandwidth greater than 50GHz.

[0186] E.3. Ranging, velocity and imaging measurements of photonic mmWave radar

[0187] In our proof-of-concept radar detection experiment, the continuous wave optical carrier emitted from the tunable laser is first amplified by an erbium-doped fiber amplifier (HaoMinOE EDFA-C-4) and coupled to the TFLN radar chip through a lensed fiber. Benefiting from the high power handling capability of the TFLN, the high input optical power can improve the signal-to-noise ratio of the modulated output optical signal from the chip. The FPC is used to tune the input optical signal to the TE mode to obtain the maximum EO modulation efficiency. The basic LFMW signal (the instantaneous frequency increases linearly from 20 GHz to 25 GHz in 4 μs) is generated by a high-speed AWG (Keysight M8196A, 33 GHz), amplified by an electrical power amplifier (Fairview microwave PE15A4021), combined with a DC voltage by a bias device (Marki microwave BT-0050), and used to drive the EOM 1 via a high-speed probe. Here for the CS-DSB modulation scheme, the DC voltage is used to bias the EOM 1 at the zero transmission point. The spectrum of the CS-DSB modulated signal was monitored using OSA (YOKOGAWA AQ6370D).

[0188] On the output side of the photonic mmWave radar chip, a lens fiber array is used to collect the output optical signals from the first EOM 510 and the second EOM 140, respectively. The second optical signal 184 obtained at the MMI (as a separation device 115) is amplified and detected by the first PD 120 as a high-speed PD to generate a radar signal 187, which can be monitored using an electrical spectrum analyzer (Agilent N9030A) and analyzed using a high-speed oscilloscope (Keysight UXR0404AP). In actual radar testing, the generated radar signal 187 is amplified by a two-stage electrical amplifier (Centellax OA4MVM3) before being transmitted to free space through a horn antenna (SAGEMillimeter InC WR-22SAZ-2410-22-S1) as a transmission antenna 127. The reflected echo signal 191 is collected by another horn antenna of the same type (i.e., receiving antenna 131), amplified first by an LNA (SHFS807) and then by a power amplifier (Centellax OA4MVM3) and then used to drive a second EOM 140 through another high-speed probe. The optical output of the second EOM 140 is amplified and detected by a second PD 145 as a low-speed PD. An oscilloscope (DSO-X91604A) is used to capture the recovered electrical signal at a sampling rate of 1GSa / s.

[0189] For the ranging experiment, a metal plate of size 7cm×10cm placed at different positions relative to the antenna was used as the target. For the velocity measurement, a toy balance car of size 9cm×6cm was used as the target, the speed of which could be programmed by FPGA. By sampling the recovered electrical signal for 50ms using an oscilloscope, the velocity information corresponding to a velocity resolution of 6.7cm / s was obtained after Doppler-FFT. For the ISAR demonstration, the target was placed on a homemade turntable, where the rotation speed was set to 1 revolution per second. After two-dimensional FFT processing of the collected 50ms recovered electrical signal, a detected target image with a two-dimensional resolution of 1.50cm×1.06cm was obtained.

[0190] E.4. Comparison with previous photon radar performance demonstrations

[0191] Table 1 shows the performance comparison between our work and previous photonic radar demonstrations, including integration, center frequency, bandwidth, range resolution, velocity resolution, and imaging resolution.

[0192] Table 1. Performance comparison of photonic microwave and millimeter wave radar.

[0193]

[0194] N / A = Information not available or applicable.

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Claims

1. A photonic millimeter wave (mmWave) radar system, comprising: A light source for generating an optical carrier wave; a signal generator for generating a first electrical signal having a linear frequency modulated waveform (LFMW); A frequency multiplication module, which is used to multiply the frequency of the first electrical signal, wherein the frequency multiplication module comprises: means for amplitude modulating the optical carrier with the first electrical signal to generate a first optical signal; and a first photodetector (PD) for converting a first attenuated replica of the first optical signal into a second electrical signal such that the second electrical signal is generated according to the square of an envelope of the first attenuated replica such that an instantaneous frequency of the second electrical signal is twice an instantaneous frequency of the first electrical signal; means for separating the first optical signal into a second optical signal and a third optical signal, wherein the second optical signal is a first attenuated copy of the first optical signal forwarded to the first photodetector; a transimpedance amplifier (TIA) for amplifying the second electrical signal to generate a radar signal, the radar signal being a chirp signal for remote sensing; a low noise amplifier (LNA) for amplifying a reflection echo signal to generate a third electrical signal, the reflection echo signal being a signal formed of one or more echoes respectively reflected from one or more objects illuminated by the radar signal in the remote sensing; and A frequency de-chirp module, which is used to de-chirp the reflected echo signal and analyze the one or more echoes, wherein the frequency de-chirp module comprises: a second electro-optical modulator (EOM) for amplitude modulating the third optical signal with the third electrical signal to generate a fourth optical signal, wherein the second electro-optical modulator is configured to be biased at a quadrature transmission point; and a second photodetector for converting the fourth optical signal into a fourth electrical signal, so that the fourth electrical signal is generated according to the square of the envelope of the fourth optical signal, whereby biasing the second electro-optical modulator at the orthogonal transmission point and generating the fourth electrical signal according to the square of the envelope of the fourth optical signal both de-chirp the reflected echo signal and enable the delay of a single echo in the reflected echo signal to be determined from a peak in the spectrum of the fourth electrical signal; Wherein, the amplitude modulation device, the second electro-optic modulator and the separation device are thin film lithium niobate (TFLN) photonic circuits integrated on a lithium niobate photonic integrated circuit (LNPIC). 2 . The photonic millimeter wave radar system according to claim 1 , wherein the frequency range of the radar signal is 1 GHz to 300 GHz.

3. The photonic millimeter wave radar system according to claim 1, wherein the signal generator is a digital-to-analog converter (DAC) arranged to input a digital signal of the linear frequency modulation waveform.

4. The photonic millimeter wave radar system according to claim 1, further comprising: An analog-to-digital converter (ADC) is used to digitize the fourth electrical signal received from the second photodetector.

5. The photonic millimeter wave radar system according to claim 4, further comprising: A digital signal processor (DSP) is configured to receive the digitized fourth electrical signal and determine a delay of the single echo from the received digitized fourth electrical signal.

6. The photonic millimeter wave radar system according to claim 5, further comprising: a transmission antenna for transmitting the radar signal to an exterior of the system for remotely sensing the one or more objects; as well as A receiving antenna is used to receive the reflected echo signal. 7 . The photonic millimeter wave radar system according to claim 6 , wherein each of the transmission antenna and the reception antenna is a horn antenna.

8. The photonic millimeter wave radar system according to claim 6, wherein each of the transmission antenna and the reception antenna is a microstrip antenna, a patch antenna, a slot antenna, or a Goubau line antenna.

9. The photonic millimeter-wave radar system according to claim 4, wherein at least the light source, the signal generator, the lithium niobate photonic integrated circuit, the transimpedance amplifier, the low-noise amplifier, the first photodetector, the second photodetector and the analog-to-digital converter are integrated on an integrated photonic millimeter-wave radar chip.

10. The photonic millimeter-wave radar system according to claim 5, wherein at least the light source, the signal generator, the lithium niobate photonic integrated circuit, the transimpedance amplifier, the low-noise amplifier, the first photodetector, the second photodetector, the analog-to-digital converter and the digital signal processor are integrated on an integrated photonic millimeter-wave radar chip. 11 . The photonic millimeter wave radar system according to claim 10 , wherein the first photodetector and the second photodetector are integrated on the lithium niobate photonic integrated circuit.

12. The photonic millimeter-wave radar system according to claim 6, wherein at least the light source, the analog-to-digital converter, the lithium niobate photonic integrated circuit, the transimpedance amplifier, the low-noise amplifier, the first photodetector, the second photodetector, the analog-to-digital converter, the digital signal processor, the transmission antenna and the receiving antenna are integrated on an integrated photonic millimeter-wave radar chip. 13 . The photonic millimeter wave radar system according to claim 12 , wherein the transmission antenna and the receiving antenna are integrated on the lithium niobate photonic integrated circuit.

14. The photonic millimeter wave radar system according to claim 1, wherein the amplitude modulation device is selected from: a first electro-optic modulator configured to be biased at a zero transmission point of the first electro-optic modulator; a cascade of a third electro-optic modulator and a first micro-ring filter, the output of the third electro-optic modulator being connected to the input of the first micro-ring filter, wherein the third electro-optic modulator is configured to be biased at a full transmission point of the third electro-optic modulator; as well as A cascade of a dual parallel Mach-Zehnder modulator (DPMZM) and a second microring filter, the output of the dual parallel Mach-Zehnder modulator being connected to the input of the second microring filter, wherein the dual parallel Mach-Zehnder modulator comprises a first Mach-Zehnder modulator and a second Mach-Zehnder modulator, the first Mach-Zehnder modulator being connected in parallel with the second Mach-Zehnder modulator, the first Mach-Zehnder modulator being arranged to receive the first electrical signal, the second Mach-Zehnder modulator being arranged to receive a 90° phase-shifted version of the first electrical signal, and wherein the main direct currents of the first Mach-Zehnder modulator, the second Mach-Zehnder modulator and the dual parallel Mach-Zehnder are biased at respective maximum transmission points. 15 . The photonic millimeter wave radar system of claim 14 , wherein each of the first electro-optic modulator and the third electro-optic modulator is a capacitively loaded traveling wave electrode electro-optic modulator.

16. The photonic millimeter wave radar system according to claim 1, wherein the separation device is selected from a multi-mode interferometer (MMI), a directional coupler, and an adiabatic coupler.

17. The photonic millimeter wave radar system according to claim 16, wherein the multi-mode interferometer is a 50:50 multi-mode interferometer.

18. A photon millimeter wave radar system for generating a radar signal and analyzing a reflected echo signal, the reflected echo signal being a signal formed by one or more echoes respectively reflected from one or more objects illuminated by the radar signal, the system comprising: A light source for generating an optical carrier wave; a signal generator for generating a first electrical signal having a linear frequency modulated waveform (LFMW); A frequency multiplication module, which is used to multiply the frequency of the first electrical signal, wherein the frequency multiplication module comprises: means for amplitude modulating the optical carrier with the first electrical signal to generate a first optical signal; and a first photodetector (PD) for converting a first attenuated replica of the first optical signal into a second electrical signal, such that the second electrical signal is generated according to the square of an envelope of the first attenuated replica, such that an instantaneous frequency of the second electrical signal is twice an instantaneous frequency of the first electrical signal, thereby forming the radar signal by amplifying the second electrical signal, such that the frequency multiplication module reduces a bandwidth requirement of the signal generator in generating the radar signal; and A transimpedance amplifier (TIA) is used to amplify the second electrical signal to generate the radar signal, wherein the radar signal is a chirp signal for remote sensing.

19. The photonic millimeter wave radar system according to claim 18, further comprising: means for separating the first optical signal into a second optical signal and a third optical signal, wherein the second optical signal is a first attenuated copy of the first optical signal forwarded to the first photodetector; a low noise amplifier (LNA), configured to amplify the reflected echo signal to generate a third electrical signal; as well as A frequency de-chirp module, which is used to de-chirp the reflected echo signal and analyze the one or more echoes, wherein the frequency de-chirp module comprises: a second electro-optical modulator for amplitude modulating the third optical signal with the third electrical signal to generate a fourth optical signal, wherein the second electro-optical modulator is configured to be biased at a quadrature transmission point; as well as a second photodetector for converting the fourth optical signal into a fourth electrical signal, so that the fourth electrical signal is generated according to the square of the envelope of the fourth optical signal, whereby biasing the second electro-optical modulator at the orthogonal transmission point and generating the fourth electrical signal according to the square of the envelope of the fourth optical signal both de-chirp the reflected echo signal and enable the delay of a single echo in the reflected echo signal to be determined from a peak in the spectrum of the fourth electrical signal; The amplitude modulation device, the second electro-optic modulator and the separation device are thin film lithium niobate (TFLN) photonic circuits integrated on a lithium niobate photonic integrated circuit (LNPIC).

20. The photonic millimeter wave radar system of claim 18, wherein the signal generator is a digital-to-analog converter (DAC) arranged to input a digital signal of the linear frequency modulation waveform.

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