Transmitter and receiver for transmitting and receiving symbols on time-varying channels with Doppler spread and methods thereof

Abstract

A communication frame for an orthogonal time-frequency space (OTFS) transmission system includes at least one block of a first class and at least one block of a second class. At least the first class of blocks includes data signals arranged two-dimensionally along a delay domain and a Doppler domain, wherein at least one data signal has a superimposed pilot signal. The blocks of the second class include data signals arranged two-dimensionally along the delay domain and the Doppler domain, which may or may not have superimposed pilot signals. At least one second class block is surrounded by the first class blocks in the delay domain, and one or more first class blocks surrounding the second class blocks in the front-back direction have at least one same data symbol and associated same superposed pilot symbols at the same position of the two-dimensional arrangement. The OTFS transmitter generates and transmits a communication frame, and the receiver performs oscillator frequency offset compensation and channel estimation by using the characteristics of the communication frame.

Description

Technical Field

[0001] The present invention relates to a method for transmitting and receiving symbols on an orthogonal time-frequency-space (OTFS) communication channel affected by Doppler spread, and a transmitter and a receiver for implementing the method.

[0002] Definitions

[0003] Throughout this specification, bold symbols represent vectors or matrices. Superscripts T, H, and represent the transpose, complex conjugate transpose, and pseudo-inverse of a vector or matrix, respectively. diag{a} is a diagonal matrix with vector a on its diagonal, and diag{A} is a vector whose elements are from the diagonal of matrix A. is the Kronecker product. Background Art

[0004] The sixth-generation (6G) wireless communication and subsequent technologies are expected to serve a large number of high-speed mobile users, such as vehicles, subways, highways, trains, drones, low Earth orbit (LEO) satellites, etc.

[0005] The previous fourth-generation and fifth-generation (5G) wireless communications adopt orthogonal frequency-division multiplexing (OFDM) technology, which has high spectral efficiency and high robustness to frequency-selective fading channels, and also allows the use of low-complexity equalizers. However, due to speed-related Doppler frequency shift or spread and fast-changing multipath reception, high-speed mobile communications suffer from severe time and frequency dispersion. Both time and frequency dispersion can cause signal fading at the receiver, so this kind of fading is also called double-selective channel fading. Double-selective channel fading will seriously damage the performance of OFDM communication.

[0006] As an alternative to OFDM, OTFS modulation is proposed as a solution to cope with double-selective fading channels.

[0007] OTFS modulation is a 2D modulation scheme that multiplexes information QAM symbols onto carrier waveforms corresponding to local pulses in the signal representation, which is called the delay-Doppler representation. OTFS waveforms are distributed in both time and frequency, but under general delay-Doppler channel impairments, they roughly remain orthogonal to each other. Theoretically, OTFS combines the reliability and robustness of spread spectrum and the high spectral efficiency and low complexity of narrowband transmission.

[0008] Figure 1A block diagram of a general OTFS transmission system is shown. The transmitter 200 includes a first transmitter-side transformation unit 202 and a second transmitter-side transformation unit 204. Serial binary data is input to a signal mapper (not shown in the figure), which outputs a two-dimensional sequence x[k, l] of information symbols, where QAM symbols are arranged along the delay periods and Doppler periods in the delay-Doppler domain.

[0009] Using the delay-Doppler channel representation has advantages due to its compactness and sparsity. Since there are usually only a small number of physical reflectors and their associated reflected signals, the number of parameters required for channel modeling and estimation in the delay-Doppler domain is much less than that required in the time-frequency domain.

[0010] The two-dimensional sequence x[k, l] of information symbols is input to the first transmitter-side transformation unit 202 and undergoes an inverse finite symplectic Fourier transform (iSFFT) to produce a matrix X[n, m], which represents the two-dimensional sequence of information symbols x[k, l] in the time-frequency domain. When the transmitter transmits in the time domain, a further transformation is required in the second transmitter-side transformation unit 204 to generate a signal s[t] in the time domain, such as a Heisenberg transform. Then, the signal s[t] is transmitted via the antenna 206 over the communication channel.

[0011] In a real environment, when the transmitted signal reaches the receiver from the transmitter through the communication channel, it is subject to doubly selective fading with Doppler spread. The received signal is a superposition of the direct copy of the transmitted signal and multiple reflected copies, where each copy is delayed due to the path delay that depends on the length of the signal path and frequency-shifted due to the Doppler shift that depends on the differential velocity between the transmitter, reflector, and receiver. Each signal copy is weighted according to its specific path delay and differential velocity. Typical Doppler shifts are on the order of 10 Hz - 1 kHz, but larger values may occur in cases of extremely high mobility (such as high-speed trains) and / or high carrier frequencies. Since it is very likely that there are multiple reflectors and / or moving reflectors in a real environment, the received superimposed signal is spread over a frequency range, not just frequency-shifted, so the signal distortion is also called Doppler shift. In the following description, the real communication channel is also referred to as the actual communication channel.

[0012] In Figure 1 it, the actual communication channel is represented by the unperturbed radio waves transmitted by the transmitter antenna 206 and various disordered radio waves from different directions and at different distances from each other at the receiver antenna 302. The radio waves can reach the receiver's antenna directly or after being reflected one or more times at one or more stationary and / or moving objects, which may introduce Doppler shifts and different delays to the reflected radio waves.

[0013] The receiver 300 picks up the received signal r[t] in the time domain and supplies it to the first receiver-side transformation unit 304, where a Wigner transform is performed on it to transform the received signal r[t] into a matrix Y[n,m] representing the received signal r[t] in the time-frequency domain. To be able to detect signals in the delay-Doppler domain, the matrix Y[n,m] is then supplied to the second receiver-side transformation unit 306, where a finite symplectic Fourier transform (SFFT) is performed on it, and the SFFT outputs a two-dimensional sequence of information symbols y[k,l] in the delay-Doppler domain. The two-dimensional sequence of information symbols y[k,l] is input to the channel estimation and equalization block 310, which performs channel estimation CE and signal detection SD and reconstructs the originally transmitted symbol, which is finally input to a demapper to output the originally transmitted binary data (the demapper is not shown in the figure).

[0014] To perform channel estimation in the receiver, the transmitter may add pilots. These pilots are known at the receiver and are located at known positions in the two-dimensional sequence of information symbols that are finally transmitted. However, the pilots replace the positions of the data symbols and do not carry any data, which results in a reduction in the spectral efficiency of the system.

[0015] Some known OTFS receivers utilize the characteristics of the delay-Doppler channel representation and adopt the basis expansion model (BEM) to parameterize the time-varying channel in OTFS as a weighted combination of multiple basis functions. The advantage of this method is that BEM can help reduce the number of unknown channel coefficients that need to be estimated.

[0016] There are various types of BEM, including complex exponential BEM (CE-BEM), generalized CE-BEM (GCE-BEM), non-critical sampling CE-BEM (NCS-CE-BEM), polynomial BEM, discrete prolate spheroidal (DPS) BEM, Karhunen-Loeve BEM (KL-BEM), spatio-temporal BEM, etc.

[0017] Among them, CE-BEM is the simplest model, but it has a large modeling error. On the other hand, CE-BEM and its variants GCE-BEM and NCS-CE-BEM do not depend on the channel statistical characteristics. GCE-BEM has simplicity and analyzability. To approach the optimal performance, its BEM order should be at least greater than 1, that is, T≥2, where T is the modeling resolution parameter. Specifically, when T = 1, GCE-BEM has a large modeling error, while when T>1, although the modeling error is small, a higher BEM order and complexity are required.

[0018] In known OTFS receivers that use complex exponential basis expansion for channel estimation, to achieve acceptable performance, the pilot overhead must increase with the increase in the maximum channel delay and Doppler spread, thus reducing the spectral efficiency. Although many OTFS channels may have a known maximum channel delay and may even have a known maximum Doppler spread, the design of practical systems takes into account higher maximum delays and Doppler spreads to provide a certain safety margin. This will further reduce the spectral efficiency of such practical systems.

[0019] The spectral efficiency can be improved by using superimposed pilots (SP) and using the released space for data symbols. Superimposed pilots use low-power pilots that are superimposed on data symbols in the delay-Doppler domain. After further transformation of the data symbols and the pilots superimposed on them, they are converted into the final transmitted OTFS signal vector x.

[0020] Figure 2 A general schematic of the superimposed pilots in the transmission frame is shown. As Figure 2 shown in the left part, the pilots can be arranged on the entire plane of the two-dimensional sequence of information symbols, which are arranged along the delay periods and Doppler periods in the delay-Doppler domain. The pilots with power much lower than that of the information symbols are represented by an ordered checkerboard pattern, indicating that the pilots are known in advance at the receiver. The data is represented by a random pattern, indicating the variability of the transmitted data. The power allocation is represented by the distance in the delay-Doppler plane. Figure 2 The right part of

[0021] shows an exemplary power allocation of pilots and data symbols. It can be seen that the power of the pilots is much lower than that of the data.

[0022] In practical application scenarios, there are constraints on the transmit power, and the needs of both data and pilot transmissions need to be covered simultaneously, that is, the data symbols and pilots need to share the total available power of the transmitter.

[0022] In the subsequent sections, M and N represent the dimensions of the transmission frame in the delay grid and Doppler grid where the symbols are arranged, respectively. The transmitted complex-valued OTFS vector x can be represented in the delay-Doppler domain as the sum of the superimposed pilot vector x sp and the data vector x d , which is defined as:

[0023] x sp = [x sp [0,0], x sp [0,1], …, x sp [0,M - 1], …, x sp [N - 1,0], x sp [N - 1,1], …, x sp [N - 1,M - 1]] T , and

[0024] x d = [x d [0,0], x d [0,1], …, x d [0,M - 1], …, x d [N - 1,0], x d [N - 1,1], …, x d [N - 1,M - 1]] T 。

[0025] Define P T as the total transmit power, α(α ∈ (0,1)) as the pilot power allocation ratio, αP T and (1 - α)P T are used to transmit pilots and data symbols respectively. Therefore, the transmitted OTFS signal vector x can be expressed as:

[0026]

[0027] Generally, if more power is used for pilot transmission, i.e., α is larger, better channel estimation performance can be expected. However, the power available for data transmission will be reduced, resulting in a lower signal - to - noise ratio (SNR) of the data signal and thus lower reliability. On the contrary, if less power is allocated to pilots, i.e., α is smaller, it will lead to poor channel estimation and signal estimation. Therefore, appropriate power allocation between data and pilots is crucial for achieving high reliability.

[0028] In the delay - Doppler domain, the received signal vector y can be expressed as:

[0029]

[0030] where b BEM,q and c q are the q - th BEM basis function and coefficient respectively, Q represents the BEM order related to f D , F MN is the discrete Fourier transform (DFT) matrix, w is the additive white Gaussian noise vector, and z is the received OTFS signal error due to BEM modeling. It should be noted that the above equation is equivalent to:

[0031]

[0032] Express the received signal vector y as the vector y d representing data spThe combination, along with noise and errors, allows data symbols to be treated as interference in channel estimation. Unlike pilot signals, data signals may have higher variability and may exhibit rather random characteristics, which can be further enhanced by appropriately arranging data symbols in a two-dimensional OTFS transmission frame, especially in the case where there are repeating structures in the data signal.

[0033] Now, while the superiority of OFTS modulation is indisputable under ideal conditions, its practical limitations in cost-effective applications may pose obstacles to its widespread adaptation, especially carrier frequency offset (CFO).

[0034] CFO is the carrier frequency mismatch between the transmitter and the receiver and is caused by the Doppler effect and radio frequency (RF) device components. The CFO caused by the Doppler effect is usually referred to as Doppler frequency shift or Doppler spread, while the CFO caused by the local crystal oscillator (XO) is called oscillator frequency offset (OFO).

[0035] There are mainly three types of crystal oscillators, namely free-running XO, temperature-controlled XO (TCXO), and oven-controlled XO (OCXO). Free-running XO is the cheapest but has the largest frequency error, for example, between ±10 to ±20 ppm, where ppm represents one part per million. OCXO can reduce the frequency error to 0.0015 ppm, but the device cost is very high, 2000 times that of free-running XO, and it also increases the power cost. TCXO is a good balance between free-running XO and OCXO. It is much cheaper than OCXO, about one-fifth of the cost, and the frequency error can be reduced to ±1.5 ppm.

[0036] OFO is usually much larger than the Doppler frequency shift or spread. For example, when the carrier frequency is 4 GHz, a TCXO with a frequency error of ±1.5 ppm may result in an OFO of ±6 kHz, which is more significant than the Doppler frequency shift of 0.5 to 2 kHz at speeds between 125 and 500 km / h. It should be noted that the Doppler frequency shift is added to the OFO, further increasing the maximum Doppler frequency shift at the receiver.

[0037] Therefore, in the presence of OFO Doppler frequency shift spread, it is necessary to optimize the performance of the OTFS communication system in terms of spectral efficiency, performance, and reliability. Summary of the Invention

[0038] The present invention aims to solve this problem and, as defined in the appended claims, proposes a communication frame with dual-rate data that is capable of estimating and compensating for large OFO (e.g., greater than 0.5 ppm) in an OTFS receiver, particularly in OTFS transmissions with low pilot overhead and high spectral efficiency. The expression "dual-rate" refers to blocks in the communication frame with low-rate data and high-rate data, where the low-rate data is particularly suitable for in-band signaling transmission in both the control plane and the user plane. In-band signaling can be used to transmit UE-specific control information and common control information. Other claims of the present invention relate to a time-frequency distortion resilient OTFS (TFDR-OTFS) transmitter and a receiver of a corresponding OTFS transmission system, respectively, for transmitting and receiving binary data sequences in a communication frame according to the present invention for high-speed mobile communication over an OTFS communication channel with long delay spread and large Doppler spread, particularly in the presence of OFO. There are other claims relating to methods for transmitting and receiving data symbols, respectively, where the receiving method includes using the proposed dual-rate data communication frame in an OTFS receiver to estimate and compensate for OFO. Further claims relate to a wireless device of an OTFS transmission system comprising the proposed receiver and / or transmitter, and a computer program product. Advantageous embodiments and extensions are given in the corresponding dependent claims.

[0039] In the following sections, an OTFS analysis system model considering OFO will be proposed, and then a communication frame, a transmitter, and a receiver of an OTFS transmission system for use in the method of the present invention will be proposed. Finally, the proposed TFDR-OTFS receiver with dual-rate data and SP (superimposed pilots) will be discussed in detail.

[0040] Among the various types of BEMs introduced above, KL-BEM is considered the most accurate BEM model when there is a good understanding of the channel statistical characteristics. However, its performance is not optimal when the assumed channel characteristics do not match the actual channel, nor is it optimal in the case of residual OFO after OFO estimation and compensation. Therefore, the present invention relies on GCE-BEM rather than KL-BEM. The present invention further uses superimposed pilots for initial channel estimation and detected symbols as additional pseudo-pilots in repeated iterative channel estimation. The superimposed pilots can be used for only a part of the communication frame or for the entire communication frame. The specific construction of the communication frame using a first type of block and a second type of block allows for the estimation of OFO in the receiver.

[0041] Similar to the OTFS system model discussed above, in the OTFS system model incorporating OFO, N and M respectively represent the dimensions of the delay grid and Doppler grid in which the symbols are arranged. Since the observations of OFO apply equally to dedicated pilots and superimposed pilots, the following discussion adopts the general representation of the OTFS signal vector. The transmitted complex OTFS matrix x is defined as

[0042] x = [x[0,0], x[0,1],..., x[0,M-1],..., x[N-1,0], x[N-1,1],..., x[N-1,M-1]] T

[0043] The received OTFS matrix y in the delay-Doppler domain is defined as

[0044] y = [y[0,0], y[0,1],.., y[0,M-1],..., y[N-1,0], y[N-1,1], …, y[N-1,M-1]] T

[0045] H t is an MN×MN time-varying channel matrix in the time domain, where the Jakes model is considered and the maximum Doppler frequency is denoted as f D . Let φ (φ ∈ [-e ofo , e ofo )) be defined as OFO, where e ofo is the maximum frequency error of XO in ppm. In the presence of OFO, the received OTFS signal vector y can be written as

[0046]

[0047] where is the OFO matrix, F N is the DFT (Discrete Fourier Transform) matrix, I M is the M×M identity matrix, and w is the additive white Gaussian noise vector. By applying GCE-BEM (Generalized Complex Exponential Basis Expansion Modeling) to the channel model H t , for example, as described in German Patent Publication No. DE 10 2022 106 409 A1, the entire content of which is incorporated herein by reference, y is further expressed as

[0048]

[0049] where b q and c q are the q-th GCE-BEM basis function and coefficient respectively, Q represents the GCE-BEM order related to f D , and F MNis the DFT matrix, z mod is the GCE - BEM modeling error.

[0050] Next, by considering Complex - Exponential - Basis - Expansion Modeling (CE - BEM) - whose BEM resolution is 1, the element - wise input - output relationship between x[k, l] and y[k, l] is derived. At the transmitter side, after applying the inverse symplectic finite Fourier transform (iSFFT) and the Heisenberg transform, the time - domain symbol s[n, m] is written as

[0051]

[0052] Define h[t, l′] as the channel gain of the l′ - th path (l′ = 0, 1,..., L) at the t - th (t = 0, 1,…, MN - 1) time instant, where L represents the channel length. After propagation through the doubly - selective fading channel, the received time - domain symbol r[n, l] is expressed as

[0053]

[0054] where it is assumed that e[n, l] is the time - domain modeling error due to GCE - BEM modeling. At the receiver side, after performing the SFFT and the Wigner transform, the received symbol y[k, l] in the delay - Doppler domain is given by

[0055]

[0056] where z[k, l] is the modeling error in the delay - Doppler domain due to GCE - BEM modeling. Compared with the ideal system without OFO, there are two problems:

[0057] -i) More symbols interfere with each other in the Doppler domain, so the inter - Doppler interference is stronger, and

[0058] ii) The phase of the received signal is changed by introducing an additional exponential term to change the phase of the received signal.

[0059] Therefore, if the OFO problem is not properly solved, OTFS modulation will lose its excellent performance in high - mobility communications. In addition, due to the existence of OFO, OTFS channel estimation becomes more challenging.

[0060] Referring to Figure 3 and assuming such a scenario where the carrier frequency f c is set to 4 GHz, the speed is v = 125 km / h, and the frequency error of XO is ±1.5 ppm. The maximum Doppler frequency is calculated as f D = 500 Hz, and the OFO frequency is f OFO = ±6 KHz. Figure 3depicts an example of a possible Doppler spectrum, where f D = 500 Hz, f OFO = -6 KHz, 0, 6 KHz. It can be easily seen that OFO will shift the Doppler spectrum to the left or right, so the maximum Doppler frequency increases to f D + f OFO . In known OTFS channel estimators, such as those discussed in German Patent Publication DE 10 2022 106 409 A1 or German Patent Publication DE 10 2021 126 321 A1, it is necessary to regenerate the subspace or basis expansion modeling (BEM) basis functions based on the new maximum Doppler frequency (i.e., f D + f OFO ). Since the frequency offset caused by OFO is much larger than the frequency offset caused by the Doppler effect, the number of required subspace or BEM basis functions will increase significantly, resulting in high pilot overhead, higher pilot power, and high computational complexity in an OTFS frame with dedicated pilots. Therefore, it is very important to estimate and compensate for OFO before implementing an OTFS channel estimator.

[0061] The present invention achieves OFO estimation and ultimate compensation by using a specific arrangement of first-class blocks and second-class blocks in a transmission frame, and by a specific arrangement of symbols in the first-class blocks, especially by having at least some identical symbols in the first-class blocks immediately before and after the second-class blocks.

[0062] Figure 4 a) shows a first exemplary representation of the data symbols of the first-class blocks and the second-class blocks in a communication frame. The repetition of at least one data symbol in the same position in the leading and trailing first-class (or low-rate) blocks is represented as a rectangular cross-hatched pattern, while the completely random nature of the data symbols in the second-class (or high-rate) blocks is represented as a pseudo-random pattern. The length of the communication frame in the delay domain is M, while the "height" in the Doppler domain is N. It should be noted that the pilot symbols in the first-class blocks are also the same. The distance between the same columns in subsequent first-class blocks is D. At least two first-class blocks and one or more second-class blocks are provided within one communication frame. Therefore, the length of the second-class blocks may vary. In Figure 4 a), the first-class blocks are arranged around the center of the transmission frame, while the second-class blocks are placed at the front, middle, and rear of the transmission frame.

[0063] Figure 4 b) shows a second exemplary arrangement of the first-class blocks and the second-class blocks in a communication frame. Here, the first-class blocks and the second-class blocks are placed alternately, i.e., one first-class block is followed by one second-class block, then one first-class block, and finally one second-class block.

[0064] Figure 4c) shows a third exemplary arrangement of the first type of block and the second type of block in the communication frame. This arrangement corresponds to Figure 4 the arrangement in b), but the positions of the first type of block and the second type of block are swapped.

[0065] Figure 4 d) shows a fourth exemplary arrangement of the first type of block and the second type of block in the communication frame. Here, the first type of blocks are placed at the front and the rear of the transmission frame, separated by one second type of block in the middle.

[0066] The length L + 1 of the first type of block depends on the longest path that the signal propagates in the channel. The channel matrix H t represents the channel length, as described below.

[0067] As described above, the received OTFS vector y in the delay - Doppler domain is defined as

[0068] y = [y[0,0], y[0,1], …, y[0,M - 1], …, y[N - 1,0], y[N - 1,1], …, y[N - 1,M - 1]] T .

[0069] Considering that superimposed pilots (SP) are used in the transmission frame, after propagation through a doubly - selective fading channel with Doppler spread, the received signal vector y can be regarded as the sum of a vector representing the received data and the pilots superimposed thereon in the delay - Doppler domain, and can be expressed as:

[0070]

[0071] where F N is the discrete Fourier transform (DFT) matrix, I M is the M×M identity matrix, w is the additive white Gaussian noise (AWGN) vector, H t is the MN×MN time - varying channel matrix in the time domain, defined as:

[0072]

[0073] where h[t,l] represents the channel gain of the l - th path at the t - th time instant, t = 0,1, …, MN - 1, l = 0,1, …, L, and L represents the channel length. The maximum Doppler frequency where f c is the carrier frequency, v is the vehicle speed, and c is the speed of light. According to the Jakes model and the U - shaped Doppler spectrum, the correlation function of the l - th path is defined as J 0 (2πnf max T s ), where J 0 (·) represents the zero - order Bessel function, Ts is the sampling period.

[0074] The ratio relationship between the spacing D between the respective identical columns of the first type of block in the delay dimension of the high-rate block that separates the low-rate blocks and the length M of the communication frame determines the OFO range that can be estimated. Since the distance D can be adjusted according to the requirements of the transmission system, and thanks to the use of SP, it is possible to support a wide and flexible OFO estimation range on the premise of only a moderately reduced spectral efficiency. This enables the use of a lower-cost free-running crystal oscillator (XO) in the transmitter and receiver, thereby reducing the equipment cost.

[0075] The oscillator frequency offset (OFO) can be estimated by utilizing the autocorrelation of the first type of block (this data block contains the superimposed pilot SP) of the two received low-rate data in the OTFS frame.

[0076] Define g k =[y[k,L],y[k,M-1]] T . The autocorrelation matrix R is calculated as: Next, perform eigenvalue decomposition on R to obtain the signal eigenvector u of length 2. Define u 1 and u 2 as the first and last elements of u respectively. Due to the constant phase shift characteristic, u 1 and u 2 satisfy the following relationship: Therefore, OFO can be easily estimated as:

[0077]

[0078] Therefore, the OFO estimation adopts a closed-form solution without additional pilots. It should be noted that the complexity of the OFO estimation process is 4N.

[0079] By incorporating the OFO estimation result into the GCE-BEM basis function b q , a new GCE-BEM basis function is obtained:

[0080]

[0081] Considering (Q + 1) BEM basis functions, the OFO compensation method has a linear complexity of MN(Q + 1). Therefore, the above system model can be re-modeled as:

[0082]

[0083] And in the delay-Doppler domain, the received signal vector y of the data and the superimposed pilot can be expressed as:

[0084]

[0085] After the initial OFO estimation and compensation, due to the BEM modeling error and noise, a small amount of OFO remains, which will be addressed in the joint estimation of the residual OFO and the communication channel.

[0086] Therefore, in a first aspect of the present invention, a dual-rate communication frame for an OTFS transmission system is proposed. The communication frame includes at least one first type of block and at least one second type of block. One or more first type of blocks include data signals arranged two-dimensionally along the delay domain and the Doppler domain. At least one of the data signals in one or more first type of blocks has a superimposed pilot signal. It should be noted that generally the entire block has a superimposed pilot signal. The size of one or more first type of blocks is N×(L + 1). One or more second type of blocks include data signals arranged two-dimensionally along the delay domain and the Doppler domain, or include data signals arranged two-dimensionally along the delay domain and the Doppler domain, where at least one of the data signals has a superimposed pilot signal. In other words, the second type of block may or may not have a superimposed pilot. In the delay domain, at least one second type of block of the communication frame is surrounded by first type of blocks in front and behind. The first type of blocks surrounding the second type of block in front and behind have at least one same data symbol and associated same superimposed pilot symbol at the same position in the two-dimensional arrangement. It should be noted that generally the first type of blocks surrounding the second type of block in front and behind are the same. These same elements of the first type of blocks surrounding the second type of block in front and behind allow or facilitate the initial oscillator frequency offset (OFO) estimation with high reliability. Compared with the second type of block, using the same symbols in the first type of blocks in front and behind reduces the effective data rate of the first type of blocks. Therefore, the first type of block can be referred to as a low-rate block, and the second type of block can be referred to as a high-rate block.

[0087] Figure 4 e) shows the first example of the superimposed pilot in the proposed dual-rate OTFS frame with superimposed pilots (SP), where both the data symbols of the first type of block and the second type of block have superimposed pilot symbols. The arrangement / configuration of the first type of block and the second type of block corresponds to Figure 4 the arrangement shown in d). The low-rate data symbols are shown in a cross-hatched pattern, emphasizing their same repetition in the leading and trailing first type of blocks. The high-rate data symbols are shown in a pseudo-random pattern, emphasizing the randomness of the data. The pilot symbols are shown in a regular checkerboard pattern, emphasizing the receiver's prior knowledge of the pilot symbols. The pilot symbols cover the entire communication frame. According to the arrangement of the first type of block and the second type of block shown in d) here, the frame contains two first type of blocks (or low-rate blocks), which are placed at the front end and the back end of the OTFS frame along the delay dimension, surrounding the second type of block. As mentioned before, the first type of block is used for the initial OFO estimation. Figure 4 ​

[0088] Figure 4 f) shows a second example of the proposed dual-rate OTFS frame with SP, where only the data symbols of the first type of blocks have superimposed pilot symbols. Similar to Figure 4 d), the low-rate data symbols are shown in a cross-hatched pattern, emphasizing their same repetition in the leading and trailing first type of blocks. The high-rate data symbols are shown in a pseudo-random pattern, emphasizing the randomness of the data. The pilot symbols are shown in a regular checkerboard pattern, emphasizing the receiver's prior knowledge of the pilot symbols. Clearly, the pilot symbols only cover the first type of blocks or the low-rate blocks. The data symbols of the second type of blocks do not have superimposed pilots, so all the power can be allocated to these data symbols.

[0089] It should be noted that, compared with the first example, the pilot symbols may require higher power in the second example, and their convergence performances may be different. In addition, the corresponding superimposed pilot arrangement can be used for Figure 4 any one of the other exemplary arrangements of the first type of blocks and the second type of blocks shown in a) to 4c).

[0090] Therefore, in one or more embodiments, at least the length of the second type of blocks arranged between the leading and trailing first type of blocks is variable in the delay domain.

[0091] According to a second aspect of the present invention, a transmitter of an OTFS transmission system includes a signal mapper arranged upstream of a first transmitter-side transformation unit and a second transmitter-side transformation unit. The signal mapper is adapted to receive a binary data sequence and output a two-dimensional arrangement / array of data symbols of a dual-rate communication frame in the delay-Doppler domain according to the first aspect of the present invention described above and / or data symbols with superimposed pilot symbols. The first transmitter-side transformation unit is adapted to receive, at an input, the two-dimensional dual-rate communication frame in the delay-Doppler domain output from the signal mapper and output a two-dimensional arrangement of information symbols in the time-frequency domain. The two-dimensional arrangement of information symbols in the time-frequency domain includes and represents both data symbols and data symbols with pilot symbols (SP). The output of the first transmitter-side transformation unit is provided to the input of the second transmitter-side transformation unit, and the second transmitter-side transformation unit is adapted to output a continuous time-domain signal representing the communication frame for transmission over a communication channel.

[0092] In one or more embodiments, the first transmitter-side transformation unit is adapted to perform precoding and / or inverse symplectic finite Fourier transform.

[0093] In one or more embodiments, the second transmitter-side transformation unit is adapted to perform a Heisenberg transform or an inverse finite Fourier transform (IFFT).

[0094] In one or more embodiments, the transmitter is arranged to allocate 50% to 99% (preferably 90% to 99%) of the total transmit power of a block with superimposed pilots to data symbols and the remaining transmit power to pilot symbols.

[0095] In one or more embodiments, the transmitter is arranged to adapt the power allocated to data symbols and pilot symbols, respectively, in a first type (or low rate) block according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver. For example, if the communication channel, the carrier frequency, and / or the speed difference between the transmitter and the receiver change, the adaptation can be dynamic for individual or grouped subsequent communication frames.

[0096] In one or more embodiments, the signal mapper is arranged to adapt the size of the first type of block and / or the second type of block, in particular the length in the delay domain, according to the communication channel used, the carrier frequency used, the maximum delay, and / or the speed difference between the transmitter and the receiver. For example, if the communication channel, the carrier frequency, and / or the speed difference between the transmitter and the receiver change, the adaptation can be dynamic for individual or grouped subsequent communication frames.

[0097] The static determination of the size of the first type of block and / or the second type of block, and / or the power allocation ratio in the transmitter can be based on the assumption that the communication channel used, the carrier frequency, and / or the speed difference between the transmitter and the receiver are static or vary negligibly within tolerable limits. The static determination can also consider worst-case scenarios, for example, when the receiver and the transmitter are in communication connection (e.g., within the radio range), the maximum expected or allowed speed difference between the transmitter and the receiver, the maximum expected delay, etc. The expected maximum speed difference can be provided by external input data - for example, the speed limit of a moving entity (such as a car or a train) within the coverage area of a fixed transmitter.

[0098] The dynamic adaptation of the size of the first type of block and / or the second type of block, and / or the power allocation ratio in the transmitter can be based on the actual speed difference between the transmitter and the receiver. Such information can be provided from the receiver, for example, the speed vector of the receiver, or based on information available at the transmitter (such as the number of receivers within the range of the transmitter). In a fixed transmitter (such as a base station, etc.), this number can correspond to the current or average number of receivers attached to or in communication connection with the transmitter. The dynamic adaptation can also be based on information received from the receiver, such as channel state information, bit error rate, or the number of iterations of the channel estimation required to decode a previously received signal.

[0099] However, the power allocation in the transmitter and / or the sizes of the first type of blocks and / or the second type of blocks and / or the power allocation ratio can also be adjusted for special requirements. For example, to achieve fast convergence performance of the receiver, the power allocation ratio α can be selected to be less than the power allocation ratio when the data SNR and the pilot SNR are equal, and / or the BEM resolution of the first channel estimation unit can be set to be greater than 1. Simulation experiments show that a power allocation ratio of about 95% can achieve an optimal balance between the bit error rate (BER) and the convergence performance.

[0100] By performing training before the actual transmission starts, a near-optimal power allocation for dynamically obtaining the ratio of the pilot signal and the data signal can be achieved. As further mentioned above, when the average data SNR and the average pilot SNR are equal, a favorable power allocation ratio can be obtained.

[0101] According to a third aspect of the present invention, there is provided a receiver for an OTFS transmission system, including a first receiver-side transformation unit and a second receiver-side transformation unit. The receiver is configured to receive, at an input part of the first receiver-side transformation unit, a time-domain signal representing a communication frame according to the first aspect of the present invention, the time-domain signal being transmitted over an actual communication channel (i.e., a communication channel with Doppler spread), and the first receiver-side transformation unit outputs a two-dimensional representation of the received communication frame in the time-frequency domain. The output of the first receiver-side transformation unit is provided to the input of the second receiver-side transformation unit, and the second receiver-side transformation unit outputs a two-dimensional representation of the received communication frame including the first type of blocks and the second type of blocks in the delay-Doppler domain. The receiver includes an OFO estimator configured to perform an initial OFO estimation and compensation using symbols carried in the first type of blocks of the received communication frame, and further includes an iterative two-stage channel estimation and equalization block configured to perform a joint estimation of the residual OFO and the communication channel based on symbols carried in all blocks of the received communication frame.

[0102] Therefore, according to one or more embodiments, the initial OFO estimation includes: performing an autocorrelation process on the received OTFS symbols (corresponding to low-rate data and SP) carried in the first type of blocks of the received communication frame. The initial OFO compensation includes providing the initial OFO estimation to the iterative two-stage channel estimation and equalization block to be combined with the BEM basis functions for channel estimation. Combining the oscillator frequency offset (OFO) estimation result with the BEM basis functions generates new BEM basis functions As discussed in detail above.

[0103] According to one or more embodiments, the joint estimation of the residual OFO and the channel uses an iterative two-stage channel estimation architecture: performing an initial channel estimation and subsequent initial equalization and symbol estimation; and performing iterative channel estimation and subsequent corresponding equalization and symbol estimation.

[0104] According to one or more embodiments, at least the pilot signal output from the second receiver-side transformation unit is provided to the first channel estimation unit, and the first channel estimation unit outputs a first estimation result of the time-domain channel matrix The first estimation result of the time-domain channel matrix And at least the data signal output from the second receiver-side transformation unit or the pilot signal and data signal output from the second receiver-side transformation unit are provided to the equalizer unit, and the equalizer unit outputs an estimated set of at least the data signal. The estimated set of at least the data signal and at least the first type of block or the first and second types of blocks output from the second receiver-side transformation unit are provided to the second channel estimation unit, and the second channel estimation unit outputs a second estimation result of the time-domain channel matrix The output of the second channel estimation unit And at least the data signal output from the second receiver-side transformation unit or the pilot signal and data signal output from the second receiver-side transformation unit are provided to the equalizer unit, which outputs a further estimated set of at least the data signal. The channel estimation in the second channel estimation unit and the estimation of the estimated set of at least the data signal in the equalizer unit are iteratively repeated until a termination criterion is satisfied. In other words, the estimation of the time-domain channel matrix in the second channel estimation unit is repeated And the estimation of the set of at least data symbols in the equalizer unit, and the process of feeding back the corresponding latest output of the equalizer unit and at least the pilot signal output from the second receiver-side transformation unit or the pilot signal and data signal output from the second receiver-side transformation unit to the second receiver-side channel estimation unit until the termination criterion is satisfied.

[0105] The termination criterion may include the convergence of the output of the equalizer unit. For example, when the bit error rate of the decoded output of the equalizer unit for two successive iterations is lower than a predetermined threshold, such convergence may be assumed. The threshold may be, for example, a difference in bit error rate less than 10 -6 Another conceivable termination criterion may be a predetermined number of iterations. A maximum number of iterations may also be set, and the iteration is terminated after this maximum number of iterations, but the iteration is terminated prematurely when the bit error rate in two successive iterations is lower than the predetermined threshold before reaching the maximum number of iterations.

[0106] In one or more embodiments of the receiver, the first receiver-side transformation unit is adapted to perform a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform.

[0107] In one or more embodiments of the receiver, the second receiver-side transformation unit is adapted to perform a symplectic finite Fourier transform.

[0108] In one or more embodiments of the receiver, the first channel estimation unit is adapted to perform channel estimation based on basis expansion modeling of a first BEM order of a time-varying communication channel. The first BEM order refers to the order of the basis expansion used to model the communication channel. The first channel estimation is preferably pilot-assisted channel estimation, i.e., estimation is performed using the known positions and / or other attributes of pilot signals in a communication frame.

[0109] In one or more embodiments of the receiver, the equalizer performs message passing, zero-forcing, and / or minimum mean square error equalization.

[0110] In one or more embodiments of the receiver, the second channel estimation unit is adapted to perform channel estimation based on basis expansion modeling of a second BEM order of a time-varying communication channel. The second BEM order refers to the order of the basis expansion used to model the communication channel. The second channel estimation is preferably data-assisted channel estimation, i.e., in addition to pilot signals in a communication frame, signals estimated in the equalizer unit are also used for estimation.

[0111] The first and second BEM orders of the first and second channel estimation units may be the same or different. It should be noted that when lower pilot power is required, a smaller BEM order Q and a lower BEM resolution T can be used. However, a smaller BEM order generally results in slower convergence. A higher BEM order with a higher resolution can bring excellent performance and faster convergence, but may require higher pilot power. For example, when the BEM order increases from Q = 2 to Q = 4, the resolution T can advantageously increase from 1 to 2.

[0112] Since OFO is estimated in the receiver, the residual OFO is relatively small. This allows a smaller BEM order Q and a smaller resolution T to be used in the initial channel estimation while still achieving near-optimal performance and fast convergence.

[0113] One or more embodiments of the receiver further include a control unit, which is adapted to receive information about the direction and absolute speed of the receiver on the ground, the direction and absolute speed of the transmitter on the ground, and / or the relative speed between the receiver and the transmitter, and is also adapted to determine the BEM order Q S , and / or is adapted to receive the BEM order Q used at the transmitter for composing a communication frame S . The received or determined BEM order Q Sand / or the received information is passed to the first channel estimation unit and / or the second channel estimation unit for determining the respective order of the BEM to be applied or used. For example, when the mobile terminal attaches to the base station, or more generally, when a communication connection is established between the transmitter and the receiver, information about the BEM order Q S used at the transmitter can be sent. Although such a transmission only requires a few bytes, this can improve performance and / or spectral efficiency.

[0114] If the dynamic adaptation of the BEM order Q S is not used, the receiver can adopt a predefined default value.

[0115] The various elements of the above-mentioned transmitter and receiver can be implemented in a hardware manner (i.e., hardware controlled and / or parameterized by software), software modules, or a combination thereof. Specifically, the first channel estimation unit and the second channel estimation unit of the receiver can rely on the same hardware or software module, and can be parameterized for the respective pilot-assisted or data-assisted channel estimation by using the corresponding input data and GCE-BEM parameters.

[0116] A wireless device according to a fourth aspect of the present invention includes a transmitter and / or a receiver for an OTFS transmission system as described above.

[0117] According to a fifth aspect of the present invention, a method for transmitting a binary data sequence through an OTFS communication channel includes mapping a previously received binary data sequence in a signal mapper into a two-dimensional arrangement of data symbols and data symbols with superimposed pilot symbols of a dual-rate communication frame in the delay-Doppler domain as described in the first aspect of the present invention above. The two-dimensional arrangement forms a communication frame of the OTFS transmission system. The mapping may include receiving the binary data sequence at an input part of the signal mapper and providing the two-dimensional arrangement of data symbols and data symbols with SP at an output part of the signal mapper.

[0118] The method further includes mapping the two-dimensional communication frame in the delay-Doppler domain into a two-dimensional arrangement of information symbols in the time-frequency domain in a first transmitter-side transformation unit. The first transformation may include receiving the two-dimensional communication frame in the delay-Doppler domain at an input part of the first transmitter-side transformation unit and providing the two-dimensional arrangement of information symbols at an output part of the first transmitter-side transformation unit.

[0119] The method further includes transforming the two-dimensional arrangement of information symbols into a continuous time-domain signal representing the communication frame in a second transmitter-side transformation unit. The second transformation may include receiving the two-dimensional arrangement of information symbols in the time-frequency domain at an input part of the second transmitter-side transformation unit and providing a continuous time-domain signal representing the communication frame at an output part of the second transmitter-side transformation unit.

[0120] The method further includes transmitting a continuous time-domain signal representing a communication frame via a communication channel. The transmission may include steps known to conventional transmitters, such as amplification, beamforming, and pointing, etc.

[0121] In one or more embodiments of the method, the first transformation step includes performing an inverse symplectic finite Fourier transform on a two-dimensional communication frame in the delay-Doppler domain.

[0122] In one or more embodiments of the method, the second transformation step includes performing a Heisenberg transform or an inverse finite Fourier transform (IFFT) on a two-dimensional arrangement of information symbols.

[0123] In one or more embodiments, the method further includes setting the power allocation ratio between data symbols and pilot symbols to be in the range of 0.5 to 0.99, preferably in the range of 0.9 to 0.99.

[0124] In one or more embodiments, the method further includes adapting the power allocation ratio between data symbols and pilot symbols according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

[0125] In one or more embodiments, the method further includes adapting the number of data symbols with superimposed pilots according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

[0126] According to a sixth aspect of the present invention, a method for receiving a binary data sequence via an actual OTFS communication channel includes receiving, via the communication channel, a continuous time-domain signal representing a communication frame according to the first aspect of the present invention. The method further includes transforming, in a first receiver-side transformation unit, the continuous time-domain signal representing the communication frame into a two-dimensional arrangement of information symbols in the time-frequency domain obtainable at an output portion of the first receiver-side transformation unit. In the next step of the method, the two-dimensional arrangement of information symbols including pilot signals and data signals in the time-frequency domain is transformed in a second receiver-side transformation unit into a two-dimensional communication frame including pilot signals and data signals in the delay-Doppler domain, the two-dimensional communication frame being obtainable at an output portion of the second receiver-side transformation unit. Next, the OFO is estimated from a first type of block of the communication frame, and the OFO is provided to a channel estimation unit to incorporate the OFO estimation into the applied channel estimation (CE) function.

[0127] In order to obtain a first estimation result of the time-domain channel matrix at an output portion of the first channel estimation unit the first type of block or the first and second types of blocks in the two-dimensional communication frame in the delay-Doppler domain are provided to the first channel estimation unit. Then the first estimation result of the time-domain channel matrix The communication frame output from the second receiver-side transformation unit (with the superimposed pilot symbols removed) is provided to the equalizer unit to obtain at least a set of estimates of the data signal at the output section of the equalizer unit.

[0128] Next, the first and second type blocks output from the second receiver-side transformation unit and at least the set of estimates of the data signal output from the equalizer unit are provided to the second channel estimation unit to estimate the time-domain channel matrix for further estimation. Then, the further estimation of the time-domain channel matrix obtainable at the output section of the second channel estimation unit and the communication frame output from the second receiver-side transformation unit (with the superimposed pilot symbols removed) are provided to the equalizer unit to obtain a further set of estimates of at least the data signal. The estimation of the time-domain channel matrix in the second channel estimation unit and the estimation of at least the set of estimates of the data signal in the equalizer unit are iteratively repeated until a termination criterion is met. During the iterative process, the respective latest further set of estimates of the data signal and the superimposed pilot signal are used to estimate the time-domain channel matrix and the respective latest estimated time-domain channel matrix and the communication frame output from the second receiver-side transformation unit are used for the next estimation of at least the data signal.

[0129] In one or more embodiments of the method, transforming the continuous time-domain signal representing the communication frame into a two-dimensional arrangement of information symbols in the time-frequency domain includes performing a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform on the continuous time-domain signal representing the communication frame.

[0130] In one or more embodiments of the method, transforming the two-dimensional arrangement of information symbols including the data signal and the data signal with SP in the time-frequency domain into a two-dimensional communication frame including the data signal and the data signal with SP in the delay-Doppler domain includes performing a symplectic finite Fourier transform on the two-dimensional arrangement of information symbols including the data signal and the data signal with SP in the time-frequency domain.

[0131] In one or more embodiments of the method, estimating the OFO from the first type block includes separating the first type block from the received communication frame, performing autocorrelation calculation on at least the SP symbols contained therein, and extracting the OFO information from the autocorrelation calculation result.

[0132] In one or more embodiments of the method, obtaining a first estimation result of the time-domain channel matrix includes performing channel estimation based on the basis expansion modeling of the time-varying communication channel with the first BEM order.

[0133] In one or more embodiments of the method, the estimation of the time-domain channel matrix in the second channel estimation unit comprises performing channel estimation based on the basis expansion modeling of the time-varying communication channel of the second BEM order.

[0134] As further mentioned above, the first and second BEM orders Q, and / or the resolutions T of the first and second channel estimation units may be the same or different, respectively.

[0135] In one or more embodiments of the method, obtaining an estimated set of at least data signals in the equalizer unit comprises performing message passing, zero-forcing, and / or minimum mean square error equalization processing on at least the data signals obtained after the second transformation in the second receiver-side transformation unit. This may include pre-removing any superimposed pilot signals from the transmission frame.

[0136] In one or more embodiments, the method further comprises receiving in the control unit information about the direction and absolute velocity of the receiver on the ground, the direction and absolute velocity of the transmitter on the ground, and / or the relative velocity between the receiver and the transmitter, and determining the BEM order Q S , and / or receiving the BEM order Q used at the transmitter for composing the communication frame S . The received information can be used to determine the corresponding BEM orders to be used in the first channel estimation unit and / or the second channel estimation unit. The correspondingly received or determined BEM orders are provided to the first channel estimation unit and / or the second channel estimation unit.

[0137] The method of transmitting and / or receiving can be represented by computer program instructions that, when executed by a microprocessor, respectively cause the computer and / or control hardware components of the transmitter or receiver of the OFTS transmission system as described above to perform the transmitting or receiving method as described above.

[0138] The computer program instructions can be stored or transmitted in a retrievable manner on a computer-readable medium or data carrier. The medium or data carrier can be physically embodied in the form of, for example, a hard disk, a solid-state disk, a flash device, etc. However, the medium or data carrier can also include a modulated electromagnetic, electrical, or optical signal that is received by the computer by means of a corresponding receiver and is transmitted to and stored in the memory of the computer.

[0139] The OTFS receiver described above can effectively resist time-frequency distortion caused by oscillator frequency offset (OFO) of a free-running crystal oscillator (XO) by adopting a dual-rate data and superimposed pilot (SP) technique.

[0140] By performing an autocorrelation operation on two identical first - type blocks carrying data symbols and superimposed pilots, whose design is optimized to support a wide range of OFO estimation, an initial OFO estimation result with only a small residual OFO can be obtained. This OFO estimation result is fused into the BEM basis functions, and by using the superimposed pilots and pseudo - pilots in the iterative process, a joint estimation of the residual OFO and the OTFS channel due to BEM modeling errors and noise is performed. Using the superimposed pilots helps to improve the spectral efficiency, while OFO estimation allows reducing the signal power allocated to the SP, thus leaving more signal power for the data signal, which improves the reliability of signal estimation.

[0141] It should be noted that OFO has an impact on the bit error rate (BER) and mean square error (MSE) of symbol detection. As the residual OFO increases, the BER deteriorates slightly. This can be compensated for by increasing the signal - to - noise ratio (SNR) of the data symbols in the first - type blocks.

[0142] XOs with very small OFO are usually more expensive and require more power to operate. In the transmitters and receivers implementing the corresponding method according to the present invention, the proposed communication frame allows processing a larger OFO in cheaper XOs with larger OFO with lower power requirements, thus also allowing reducing power consumption while reducing device costs. Due to the flexible design of the first - type and second - type blocks, a wide estimation range of OFO can be adjusted according to the needs of the system.

[0143] Using the first - type and second - type blocks and adopting superimposed pilots (SP), OFO can be estimated and OTFS channel estimation can be performed while reducing the cost of the frame data rate. By carefully designing the positions of the first - type blocks, OFO estimation does not require an additional dedicated pilot signal.

[0144] Benefiting from the careful design of the communication frame with the first - type and second - type blocks, where the first - type blocks provide a low - rate data rate and SP, the proposed OTFS receiver performs excellently in terms of bit error rate (BER), OFO estimation mean square error (MSE), and MSE of channel estimation, while not requiring a dedicated pilot signal. Its BER performance is close to the BER lower bound assuming perfect estimation and compensation of OFO and the channel.

[0145] The communication frame, transmitter, receiver, and corresponding method according to the present invention can be advantageously used for high - speed mobile communication. Description of the Drawings

[0146] In the following sections, exemplary embodiments of the present invention will be described in more detail with reference to the drawings. In the drawings,

[0147] Figure 1 a block diagram of a general OTFS transmission system is shown,

[0148] Figure 2 shows the superimposed pilots and their power allocation,

[0149] Figure 3 shows an exemplary visualization of the Doppler spectral shift in the OTFS communication channel in the presence of OFO,

[0150] Figure 4 shows an exemplary OTFS frame pattern at the transmitter according to the present invention.

[0151] Figure 5 shows a block diagram of OFO estimation, channel estimation, and equalization of an exemplary receiver according to the present invention,

[0152] Figure 6 shows a flowchart of a method for transmitting a binary data sequence through an OTFS communication channel,

[0153] Figure 7 shows a flowchart of a method for receiving a binary data sequence through an OTFS communication channel vulnerable to double-selective fading,

[0154] Figure 8 shows an exemplary block diagram of an apparatus for performing a transmission method according to the present invention, and

[0155] Figure 9 shows an exemplary block diagram of an apparatus for performing a reception method according to the present invention..

[0156] In all the figures, the same reference numerals may be used to refer to the same or similar elements. Detailed Description

[0157] Figures 1 to 4 has been further described above and will not be discussed further.

[0158] Figure 5 shows a schematic block diagram of initial OFO estimation and compensation and joint residual OFO and channel estimation in an exemplary receiver 300 according to the present invention. After performing the SFFT and the Wigner transform, the received symbols in the delay-Doppler domain y[k,l] can be used for further processing.

[0159] The OFO estimation 312, the channel estimation 320 with two-stage iterative CE blocks 321, 322, and the equalization 326 replace Figure 1 the general channel estimation and equalization block 310 shown in Figure 1 All other elements of the receiver 300 shown in

[0160] First, the first type of blocks carrying data symbols and superimposed pilots and the second type of blocks carrying data symbols and superimposed pilots or only data symbols in the two-dimensional arrangement output from the second receiver-side transformation unit 306 (as the signal y[k, l] in the delay-Doppler domain) are provided to the OFO estimation unit 312. The OFO estimation unit 312 includes a block separation unit 314 that separates the first type of blocks of the communication frame from the second type of blocks and provides the first type of blocks to the autocorrelation unit 316. The autocorrelation processing may include eigenvalue decomposition (not shown in the figure), or eigenvalue decomposition is performed after the autocorrelation processing. The result of the autocorrelation processing is provided to the OFO extraction unit 318, which determines the OFO and provides it to the BEM basis generation unit 319. Based on the OFO estimate output from the OFO extraction unit 318 The BEM basis generation unit 319 determines the BEM basis functions to be used in the channel estimation units 321 and 322, and forwards the corresponding information thereto to the first channel estimation unit 321 and the second channel estimation unit 322 accordingly.

[0161] The first type of blocks carrying data symbols and superimposed pilots output from the low-rate block extraction unit 314, or the entire frame represented by the signal y[k, l] including the first type of blocks and the second type of blocks in the delay-Doppler domain, is provided to the first channel estimation unit 321, which performs a superimposed pilot-assisted first channel estimation including OFO using a GCE-BEM channel model with a first BEM order Q S The first BEM order Q S can be very small, using a low resolution T, although at the cost of slower convergence. However, the first BEM order Q S can also be quite large, using a higher resolution T, resulting in faster convergence. The first channel estimation unit 321 outputs a first or initial channel estimation result This estimation result is provided to the input part of the pilot removal unit 324.

[0162] The communication frame represented by the signal y[k, l] includes the first type of blocks and the second type of blocks in the delay-Doppler domain, and this communication frame is also provided to the input part of the pilot removal unit 324 and the input part of the second channel estimation unit 322.

[0163] The pilot removal unit 324 uses the latest channel estimation output from the first channel estimation unit 321 or the second channel estimation unit 322 to remove the superimposed pilot signals from the signal y[k, l] according to the known structure of the transmitted communication frame (i.e., its arrangement and power level), and only retains the data signals. The output signal of the pilot removal unit 324 is provided to the equalizer unit 326. The equalizer unit 326 determines and outputs an estimation result of the transmitted data symbol

[0164] Initial channel estimation is performed in the first channel estimation unit 321 and an initial estimation of the transmitted data symbol is obtained Thereafter, the group of the latest estimated data symbols is provided to the input section of the second channel estimation unit 322 in an iterative manner. The second channel estimation unit 322 uses the GCE - BEM channel model (which has the second BEM order Q L and the second resolution T), and performs data - aided channel estimation including OFO on the superimposed pilots in the signal y[k, l] and the transmitted symbol estimation (as pseudo - pilots) fed back from the equalizer unit 326 to the second channel estimation unit 322. The second channel estimation unit 322 may use a higher BEM order QL and a higher resolution T than the first channel estimation unit, although it is also possible to use the same BEM order Q L and resolution T.

[0165] In each iteration, the latest output of the second channel estimation unit 322 (representing the channel estimation ) is input to the pilot removal unit 324. Based on this, the pilot removal unit 324 removes the superimposed pilots from the received signal vector y[k, l] in the delay - Doppler domain and provides the estimation result of the signal representing only the received data signal to the input section of the equalizer unit 326. The equalizer unit 326 outputs a more accurate estimation result of the transmitted data symbol The process is repeated until a termination criterion is met.

[0166] Figure 6 FIG. shows a flowchart of a method 400 for transmitting a binary data sequence through an OTFS communication channel. In step 402, the binary data sequence is mapped to a two - dimensional communication frame in the delay - Doppler domain, and the two - dimensional communication frame includes a first type of block and a second type of block according to the first aspect of the present invention. In step 404, the two - dimensional communication frame in the delay - Doppler domain is transformed into a two - dimensional arrangement of information symbols in the time - frequency domain. In step 406, the two - dimensional arrangement of information symbols in the time - frequency domain is transformed into a continuous time - domain signal representing the communication frame, and the continuous time - domain signal is transmitted through the channel in step 408. Before transforming the two - dimensional arrangement of information symbols in the delay - Doppler domain into the two - dimensional arrangement of information symbols in the time - frequency domain, the power allocation ratio between the pilot signal and the data signal and / or the number of data symbols with superimposed pilots may be determined or adapted in an optional step 410, and these parameters are set in an optional step 412.

[0167] Figure 7 FIG. 500 is a flowchart of a method for receiving a binary data sequence carried in a communication frame according to a first aspect of the present invention over an OTFS communication channel vulnerable to double-selective fading. In step 502, a continuous time-domain signal representing a communication frame is received over the communication channel. In step 504, the continuous time-domain signal representing the communication frame is transformed into a two-dimensional arrangement of information symbols in the time-frequency domain. In step 506, the two-dimensional arrangement of information symbols including pilot signals and data signals in the time-frequency domain is transformed into a two-dimensional communication frame in the delay-Doppler domain, the two-dimensional communication frame including a first type of block and a second type of block according to the first aspect of the present invention. In step 508, the OFO from the first type of block of the communication frame is estimated, and in step 510, the OFO is provided from the OFO estimator 312 to the channel estimation units 321, 322 so that the OFO estimation result φ^ is incorporated into the applied CE function. In step 512, an initial estimation result of the time-domain channel matrix is obtained in the first channel estimation unit 320, and the first channel estimation unit 321 performs channel estimation on the first type of block or the first and second types of blocks based on the basis expansion modeling of the time-varying communication channel with a first BEM order and a first resolution. In step 514, pilot symbols are removed from the received communication frame represented by the first type and the second type of blocks, and the resulting signal is provided to the equalizer unit 326. In step 516, an estimated set of at least the data signals is determined in the equalizer unit 326 based on the channel estimation and the communication frame in the delay-Doppler domain. Step 518 checks whether a termination criterion is satisfied. If satisfied (the "yes" branch of step 518), then in step 522, it is signaled that the estimated received symbols can be output to the demapper and ultimately output as the received binary sequence. If the termination criterion is not satisfied (the "no" branch of step 518), then in step 520, a further estimation of the time-domain channel matrix is obtained in the second channel estimation unit 322, and the second channel estimation unit 322 performs channel estimation using, in addition to the pilot signals, the previously estimated data signals based on the basis expansion modeling of the time-varying communication channel with a second BEM order and at a second resolution. The result of the further channel estimation is provided to the pilot symbol removal step 514, and the equalization step 516 and the check step 518 for compliance with the termination criterion are repeated.

[0168] Optionally, in step 524, the BEM order Q used at the transmitter may be received S , or information allowing determination of the BEM order to be used in channel estimation. In step 526, the BEM order Q to be used is determined S , and in step 528, the order Q S is provided to the channel estimation unit.

[0169] Figure 8Fig. 0 shows an exemplary block diagram of a transmitter 200 according to a second aspect of the present invention, which is configured to perform a transmission method 400 according to a fourth aspect of the present invention. The transmitter 200 includes a microprocessor 220, a volatile memory 222, a non-volatile memory 224, and a communication interface 226 for transmitting signals to a receiver 300 via an antenna 206. The above elements are communicatively connected via at least one data connection or bus 228. The non-volatile memory 224 stores computer program instructions, which, when executed by the microprocessor 220, cause the transmitter 200 to perform the transmission method 400 according to the fourth aspect of the present invention as described above. It should be noted that, with reference to Figure 1 The various functional modules of the transmitter described (e.g., the first transmitter-side transformation unit 202 and the second transmitter-side transformation unit 204) can be implemented in whole or in part by software executed by the microprocessor 220.

[0170] Figure 9 Fig. 6 shows an exemplary block diagram of a receiver 300 according to a third aspect of the present invention, which is configured to perform a reception method 500 according to a fifth aspect of the present invention. The receiver 300 includes a microprocessor 330, a volatile memory 332, a non-volatile memory 334, and a communication interface 336 for receiving signals from a transmitter via an antenna 302. The above elements are communicatively connected via at least one data connection or bus 338. The non-volatile memory 334 stores computer program instructions, which, when executed by the microprocessor 330, cause the receiver 300 to perform the reception method 500 according to the fifth aspect of the present invention as described above. It should be noted that, with reference to Figure 1 The various functional modules of the receiver described (e.g., the symbol estimation and detection functions in the first receiver-side transformation unit 304, the second receiver-side transformation unit 306, the OFO estimation unit 312, the first channel estimation unit 321, the second channel estimation unit 322, the pilot removal unit 324, and the equalizer unit 326) can be implemented in whole or in part by software executed by the microprocessor 330.

[0171] Definition and List of Reference Numerals (Part of the Specification)

[0172] f c Carrier frequency

[0173] Δf Sub-carrier spacing

[0174] L Channel length

[0175] M Number of delay bins of the communication frame

[0176] N Number of Doppler bins of the communication frame

[0177] P T Total transmit power

[0178] α pilot power allocation ratio

[0179] Q S BEM order in initial channel estimation

[0180] Q L BEM order in subsequent iterative channel estimation

[0181] AWGN Additive White Gaussian Noise

[0182] BEM Basis Expansion Model

[0183] CE-BEM Complex Exponential Basis Expansion Model

[0184] GCE-BEM Generalized Complex Exponential Basis Expansion Model

[0185] DFT Discrete Fourier Transform

[0186] KL-BEM Karhunen-Loeve BEM

[0187] MSE Mean Square Error

[0188] OTFS Orthogonal Time Frequency Space

[0189] SNR Signal-to-Noise Ratio

[0190] BER Bit Error Rate

[0191] OFDM Orthogonal Frequency Division Multiplexing

[0192] MP Message Passing

[0193] SFFT Symplectic Finite Fourier Transform

[0194] 200 Transmitter

[0195] 202 First Transmitter-Side Transformation Unit

[0196] 204 Second Transmitter-Side Transformation Unit

[0197] 206 Antenna

[0198] 220 Microprocessor

[0199] 222 Volatile Memory

[0200] 224 Non-Volatile Memory

[0201] 226 Communication Interface

[0202] 228 Data Link / Bus

[0203] 300 Receiver

[0204] 302 Antenna

[0205] 304 First Receiver - side Transformation Unit

[0206] 306 Second Receiver - side Transformation Unit

[0207] 310 Channel Estimation and Equalization Block

[0208] 312 OFO Estimator

[0209] 314 Low - rate Block Extraction

[0210] 316 Autocorrelation

[0211] 318 OFO Extraction

[0212] 319 BEM Basis Generation Containing OFO

[0213] 320 Two - stage CE and EQ

[0214] 321 First Channel Estimation Unit

[0215] 322 Second Channel Estimation Unit

[0216] 324 Pilot Removal Unit

[0217] 326 Equalizer Unit

[0218] 326 Pilot Extraction Unit

[0219] 330 Microprocessor

[0220] 332 Volatile Memory

[0221] 334 Non - volatile Memory

[0222] 336 Communication Interface

[0223] 338 Data Link / Bus

[0224] 500 Receiving Method

[0225] 502 Receive Continuous - time Domain Signal

[0226] 504 Transform the Continuous - time Domain Signal into a Two - dimensional Arrangement of Information Symbols in the Time - frequency Domain

[0227] 506 Transform the Two - dimensional Arrangement of Information Symbols in the Time - frequency Domain into a Two - dimensional Communication Frame in the Delay - Doppler Domain

[0228] 508 Estimate OFO

[0229] 510 Provide the OFO Estimation Result to Channel Estimation

[0230] 512 Initial channel estimation

[0231] 514 Pilot removal

[0232] 516 Symbol estimation in the equalizer unit

[0233] 518 Does the termination criterion hold?

[0234] 522 Output the latest estimation result to the demapper

[0235] 520 Channel estimation in the second channel estimation unit

[0236] 524 Receive the BEM order used in the transmitter

[0237] 526 Determine the BEM order to be used in the receiver

[0238] 528 Provide the BEM order to the channel estimation unit

Claims

1. A communication frame for an Orthogonal Time-Frequency Space (OTFS) transmission system, wherein, the communication frame includes at least one first type of block and at least one second type of block. At least one or more first type of blocks include multiple data signals arranged two-dimensionally along the delay domain and the Doppler domain, where at least one of them has a superimposed pilot signal. The one or more second type of blocks include data signals arranged two-dimensionally along the delay domain and the Doppler domain, or include multiple data signals arranged two-dimensionally along the delay domain and the Doppler domain, where at least one of them has a superimposed pilot signal. In the delay domain, at least one second type of block is surrounded by first type of blocks before and after, and the first type of blocks surrounding the second type of block before and after have at least one same data symbol and associated same superimposed pilot symbol at the same position in the two-dimensional arrangement.

2. The communication frame for an orthogonal time-frequency space transmission system according to claim 1, wherein, the data symbols in the first type of blocks surrounding the second type of block before and after in the communication frame carry only data for controlling the communication connection between the transmitter and the receiver.

3. The communication frame for an orthogonal time-frequency space transmission system according to claim 1 or 2, wherein, in the delay domain, the spacing between at least two first type of blocks is variable.

4. A transmitter (200) for an orthogonal time-frequency space transmission system, the transmitter includes a first transmitter-side transformation unit (202), a second transmitter-side transformation unit (204), and a signal mapper arranged upstream of the first transmitter-side transformation unit and the second transmitter-side transformation unit, wherein, the signal mapper is adapted to receive a binary data sequence and output the two-dimensional double-rate communication frame (x[k,l]) in the delay-Doppler domain according to claim 1, the communication frame includes data symbols and superimposed pilot symbols, wherein the first transmitter-side transformation unit (202) is adapted to receive, at an input part, the two-dimensional double-rate communication frame in the delay-Doppler domain output from the signal mapper, and output a two-dimensional arrangement of information symbols in the time-frequency domain, and wherein the output of the first transmitter-side transformation unit (202) is provided to the input part of the second transmitter-side transformation unit (204), and the second transmitter-side transformation unit is adapted to output a continuous time-domain signal representing the communication frame for transmission over a communication channel.

5. The transmitter (200) for an orthogonal time-frequency space transmission system according to claim 4, wherein, the first transmitter-side transformation unit (202) is adapted to perform precoding and / or inverse symplectic finite Fourier transform.

6. The transmitter (200) for an orthogonal time-frequency space transmission system according to claim 4 or 5, wherein, the second transmitter-side transformation unit (204) is adapted to perform a Heisenberg transform or an inverse finite Fourier transform (IFFT).

7. The transmitter (200) for an orthogonal time-frequency space transmission system according to any one of claims 4 to 6, wherein, The transmitter (200) is arranged to allocate 50% to 99%, preferably 90% to 99%, of the total transmit power of the blocks with superimposed pilots to data symbols, and the remaining transmit power to pilot symbols.

8. The transmitter (200) for an orthogonal time-frequency-space transmission system according to any one of claims 4 to 7, wherein, the transmitter (200) is arranged to adapt the power allocated to data symbols and superimposed pilot symbols respectively according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

9. The transmitter (200) for an orthogonal time-frequency-space transmission system according to any one of claims 4 to 8, wherein, the signal mapper is arranged to adapt the size of the first type of blocks and / or the second type of blocks according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

10. A receiver (300) for an orthogonal time-frequency-space transmission system, the receiver comprising a first receiver-side transformation unit (304) and a second receiver-side transformation unit (306), wherein, the receiver (300) is adapted to receive, at an input part of the first receiver-side transformation unit (304), a time-domain signal representing a communication frame according to any one of claims 1 to 3 transmitted through a communication channel, the first receiver-side transformation unit outputs a two-dimensional representation of the received communication frame in the time-frequency domain, wherein the output of the first receiver-side transformation unit (304) is provided to an input part of the second receiver-side transformation unit (306), the second receiver-side transformation unit outputs a two-dimensional representation of the received communication frame including pilot signals and data signals in the delay-Doppler domain, wherein the receiver (300) includes an oscillator frequency offset (OFO) estimator (312), the oscillator frequency offset estimator being configured to perform an initial oscillator frequency offset estimation and compensation using the symbols carried in the first type of blocks of the received communication frame, the receiver further includes an iterative two-stage channel estimation and equalization block (320), the iterative two-stage channel estimation and equalization block being configured to determine a residual oscillator frequency offset and perform channel estimation using all the symbols of the received communication frame.

11. The receiver (300) according to claim 10, wherein, the oscillator frequency offset estimator (312) is configured to perform an autocorrelation process (316) on the received orthogonal time-frequency-space symbols carried in the first type of blocks of the received communication frame, wherein the initial oscillator frequency offset estimation is provided to the iterative two-stage channel estimation and equalization block (320).

12. The receiver (300) according to claim 10 or 11, including an iterative two-stage channel estimator for jointly estimating the residual oscillator frequency offset and the channel by performing an initial channel estimation (321) and subsequent initial equalization and symbol estimation (326); and performing iterative channel estimation (322) and subsequent corresponding equalization and symbol estimation (326).

13. The receiver (300) according to claim 12, wherein, At least a pilot signal output from the second receiver-side conversion unit (306) is provided to a first channel estimation unit (321), and the first channel estimation unit outputs an initial estimation result of a time-domain channel matrix. wherein the initial estimation result of the time-domain channel matrix and at least a data signal output from the second receiver-side conversion unit (306) or a pilot signal and a data signal output from the second receiver-side conversion unit (306) are provided to an equalizer unit (326), and the equalizer unit outputs an estimation set of at least the data signal, wherein the estimation set of at least the data signal and at least a pilot signal output from the second receiver-side conversion unit (306) or a pilot signal and a data signal output from the second receiver-side conversion unit (306) are provided to a second channel estimation unit (322), and the second channel estimation unit outputs a second estimation result of the time-domain channel matrix. wherein the output of the second channel estimation unit (322) and at least the data signal output from the second receiver-side conversion unit (306) or a pilot signal and a data signal output from the second receiver-side conversion unit (306) are provided to the equalizer unit (326), and the equalizer unit outputs a further estimation set of at least the data signal, wherein the receiver (300) is adapted to iteratively repeat the channel estimation in the second channel estimation unit (322) and the estimation of the estimation set of at least the data signal in the equalizer unit (326) until a termination criterion is satisfied.

14. The receiver (300) according to claim 13, wherein, the first channel estimation unit (321) is adapted to perform channel estimation based on basis expansion modeling with a first basis expansion modeling (BEM) order of a time-varying communication channel.

15. The receiver (300) according to claim 13 or 14, wherein, the second channel estimation unit (322) is adapted to perform channel estimation based on basis expansion modeling with a second basis expansion modeling order of a time-varying communication channel.

16. The receiver (300) according to any one of claims 10 to 15, wherein, the first receiver-side transformation unit (304) is adapted to perform a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform.

17. The receiver (300) according to any one of claims 10 to 14, wherein, the second receiver-side transformation unit (306) is adapted to perform decoding and / or a symplectic finite Fourier transform.

18. The receiver (300) according to any one of claims 10 to 16, wherein, the equalizer unit (326) performs message passing, zero-forcing, and / or minimum mean square error equalization processing.

19. The receiver (300) according to any one of claims 10 to 18, the receiver further comprising a control unit adapted to receive information about the direction and absolute velocity of the receiver (300) on the ground, the direction and absolute velocity of the transmitter (200) on the ground, and / or the relative velocity between the receiver (300) and the transmitter (200), and the control unit is further adapted to determine the basis expansion modeling order (Q S ) ， and / or adapted to receive the basis expansion modeling order (Q S ) used at the transmitter (200) for composing communication frames, and adapted to transfer the received information and / or the basis expansion modeling order (Q S ) to the first channel estimation unit and / or the second channel estimation unit (320, 322).

20. A wireless device for an orthogonal time-frequency-space transmission system, the wireless device comprising a transmitter (200) according to any one of claims 4 to 9 and / or a receiver (300) according to any one of claims 10 to 19.

21. A method (400) for transmitting a binary data sequence over an orthogonal time-frequency-space communication channel, the method comprising: - mapping (402) the binary data sequence in a signal mapper into a two-dimensional communication frame in a delay-Doppler domain according to any one of claims 1 to 3, - transforming (404) the two-dimensional communication frame in the delay-Doppler domain in a first transmitter-side transformation unit (202) into a two-dimensional arrangement of information symbols in a time-frequency domain, - transforming (406) the two-dimensional arrangement of information symbols in the time-frequency domain in a second transmitter-side transformation unit (204) into a continuous time-domain signal representing the communication frame, and - transmitting (408) the continuous time-domain signal representing the communication frame over the communication channel.

22. The method (400) according to claim 21, wherein, the first transformation step (404) comprises performing an inverse symplectic finite Fourier transform on the two-dimensional communication frame in the delay-Doppler domain.

23. The method (400) according to claim 21 or 22, wherein, the second transformation step (406) comprises performing a Heisenberg transform or an inverse finite Fourier transform (IFFT) on the two-dimensional arrangement of information symbols.

24. The method (400) according to any one of claims 21 to 23, further comprising setting (410) a power allocation ratio between data symbols and pilot symbols to be in a range of 0.5 to 0.99, preferably in a range of 0.9 to 0.

99.

25. The method (400) according to any one of claims 21 to 24 further comprises adapting (412) the power distribution ratio between data symbols and pilot symbols according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

26. The method (400) according to any one of claims 21 to 25 further comprises adapting (412) the number of data symbols with superimposed pilots according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

27. A method (500) of receiving a binary data sequence through an orthogonal time-frequency-space communication channel vulnerable to double-selective fading, the method comprises: - receiving (502) a continuous time-domain signal representing a communication frame according to any one of claims 1 to 3 through the communication channel, - transforming (504) the continuous time-domain signal representing the communication frame in a first receiver-side transformation unit (304) into a two-dimensional arrangement of information symbols in the time-frequency domain obtainable at the output of the first receiver-side transformation unit (304), - transforming (506) the two-dimensional arrangement of information symbols including pilot signals and data signals in the time-frequency domain in a second receiver-side transformation unit (306) into a two-dimensional communication frame including pilot signals and data signals in the delay-Doppler domain obtainable at the output of the second receiver-side transformation unit (306), - estimating (508) an oscillator frequency offset according to a first type of block of the communication frame, - The estimated oscillator frequency offset is provided (510) to the channel estimation units (321, 322) for incorporating the oscillator frequency offset estimate into the applied channel estimation function - Provide the first type of blocks or the first type of blocks and the second type of blocks output from the second receiver-side transformation unit (306) to the first channel estimation unit (321) so as to obtain (512) a first estimation result of the time-domain channel matrix at the output part of the first channel estimation unit (321). -Using a first estimation result of a time-domain channel matrix Removing (514) pilot symbols from the received communication frame - The communication frame with the superimposed pilot symbols removed and the first estimation result of the time-domain channel matrix are provided to the equalizer unit (326) to obtain (516) at the output section of the equalizer unit (326) an estimation set of at least the data signal - In the second channel estimation unit (322), an estimation result of a time-domain channel matrix is estimated (520) based on the first type of blocks and the second type of blocks and based on an estimation set of at least data signals most recently output from the equalizer unit (326). - The estimation result of the time-domain channel matrix obtained at the output section of the second channel estimation unit (322) and the first type of blocks and the second type of blocks output from the second receiver-side transformation unit (306) are provided to the pilot removal unit (324), - Using the estimation result of the time-domain channel matrix obtained at the output section of the second channel estimation unit (322) Remove (514) pilot symbols from the received communication frame - The communication frame from which the superimposed pilot symbols have been removed and the estimation result of the time-domain channel matrix obtained at the output section of the second channel estimation unit (322) are provided to the equalizer unit (326) to obtain (516) at the output section of the equalizer unit (326) a further set of estimates of at least the data signal, and - Iteratively repeat the following operations until the termination criterion is met: Estimate (520) the time-domain channel matrix in the second channel estimation unit (322). Remove (514) pilot symbols and estimate (516) at least a set of data signals in the equalizer unit (326).

28. The method (500) according to claim 27, wherein, the first transformation step (504) comprises performing a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform on the continuous time-domain signal representing the communication frame.

29. The method (500) according to claim 27 or 28, wherein, the second transformation step (506) comprises performing a symplectic finite Fourier transform on the two-dimensional arrangement of information symbols including pilot signals and data signals in the time-frequency domain.

30. The method (500) according to any one of claims 27 to 29, wherein, estimating (508) an oscillator frequency offset according to a first type of block of the communication frame comprises: - separating the first type of block from the received communication frame, - performing autocorrelation processing on at least the superimposed pilot symbols included in the first type of block, and - extracting oscillator frequency offset information from the autocorrelation processing.

31. The method (500) according to any one of claims 27 to 30, wherein, The first estimation result of the time-domain channel matrix is obtained (512) in the first channel estimation unit (321). It includes performing channel estimation based on basis expansion modeling of a time-varying communication channel having a first basis expansion modeling order.

32. The method (500) according to any one of claims 27 to 31, wherein, Obtain (520) an estimation result of the time-domain channel matrix in the second channel estimation unit (322). Including performing channel estimation based on basis expansion modeling of a time-varying communication channel having a second basis expansion modeling order.

33. The method (500) according to any one of claims 27 to 32, wherein, obtaining (516) an estimation set of at least data signals in the equalizer unit (326) comprises: performing message passing, zero-forcing, and / or minimum mean square error equalization processing on at least the data signals.

34. The method (500) according to any one of claims 27 to 33 further comprises: - Receive (524) in the control unit information on the direction and absolute velocity of the receiver (300) on the ground, the direction and absolute velocity of the transmitter (200) on the ground, and / or the relative velocity between the receiver (300) and the transmitter (200), and / or receive the basis expansion modeling order (Q used at the transmitter (200) for composing the communication frame S ) - Determine (526) the respective basis expansion modeling order to be used in the first channel estimation unit and / or the second channel estimation unit (321, 322), and - Provide (528) the respective determined basis expansion modeling order to the first channel estimation unit and / or the second channel estimation unit (321, 322).

35. A computer program product comprising computer program instructions which, when executed by a microprocessor, cause a computer and / or a control hardware component of a transmitter (200) for an orthogonal time-frequency-space transmission system according to any one of claims 4 to 9 or of a receiver (300) for an orthogonal time-frequency-space transmission system according to any one of claims 10 to 19 to perform a method (400, 500) according to one or more of claims 21 to 26 or according to one or more of claims 27 to 34, respectively.

36. A computer-readable medium or data carrier for retrievably transmitting or storing the computer program product according to claim 35.

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