Radar unit, circuit for radar transceiver and method thereof

Abstract

The invention relates to a radar unit comprising a radar transceiver comprising: a reference local oscillator (LO); and at least two transmitter paths arranged to transmit the reference LO, one of the at least two transmitter paths having a frequency shift of at least an ADC sampling frequency. A receiver is coupled to two receiver paths including a down-conversion circuit configured to receive a reflected radar signal and a reference signal and to provide a down-converted baseband signal to a bandpass filter, an ADC, and a DSP configured to process the down-converted, filtered baseband signal in digital form. A frequency shifter circuit applies a frequency shift to the reference signal that shifts the transmit signal by an amount wherein a first down-converted baseband signal is passed by a first bandpass filter in a first receiver path and a second down-converted baseband signal is passed by a second bandpass filter in a second receiver path.

Description

Technical Field

[0001] The present invention is applicable to (but not limited to) utilizing orthogonal 'I' and 'Q' branches of a radar receiver (Rx) to support simultaneous transmission from different transmitter (Tx) antennas (or groups of Tx antennas). Background Art

[0002] The simplest radar unit consists of a single, static transmit antenna and a single, static receive antenna. This type of radar unit provides information about the distance between the radar unit and other objects, and can calculate the absolute velocity of those objects within the radar range. However, this simple radar unit is unable to distinguish the angular direction of detected objects. To enable radar units to additionally obtain angular information, the following solutions are known. In radar applications, such as on ships, directional transmit and / or directional receive antennas can be mechanically rotated to obtain additional angular information. Alternatively, in other radar units, the transmit and / or receive antennas may consist of arrays of antenna elements, allowing the transmit and / or receive 'beams' to be steered in different directions by varying the phase offsets between the antenna elements.

[0003] Some radars are known to use linear frequency modulated chirps as sensing signals and mix the received chirp signal with a transmitted replica to produce a beat signal containing information about the target range of its frequency (called the 'beat frequency'). Some conventional radar transceivers use an orthogonal (IQ) receiver (Rx) with two receiver branches per receiver antenna. The IQ Rx should theoretically expand the acquisition frequency interval of the measured beat frequency from f b ∈[0,f s / 2] doubled up to f b ∈[0,f s ], where f s is the analog-to-digital converter (ADC) sampling frequency in each ('I' and 'Q') branch. In practice, due to hardware imperfections (e.g., IQ imbalance / non-orthogonality), full utilization of the complex ADC bandwidth is difficult to achieve, resulting in an image rejection ratio of the order of 30-40 dB. This image rejection ratio range is unacceptable for distance measurement in most practical millimeter wave (mmw) radar units, where a high dynamic range (of the order of 60-100 dB due to path loss at mm wavelengths) is required. This is typically driven by limitations of current technology and coupling at radio frequency (RF) for the compact size of the device.

[0004] Reference Figure 1, shows a known radar unit 100 operating at mmW frequencies, where radar signal processing utilizes a straightforward implementation of a quadrature (IQ) receiver and is based on a linear frequency modulated continuous wave (FMCW) transmit and receive system. The known radar unit 100 includes one or more receive antennas 102 for receiving radar signals and two or more antennas 103 for transmitting FMCW radar signals. The FMCW radar signals are generated by processing a high-frequency reference local oscillator (RefLO) signal 128 using a phase rotator 122 programmed by dedicated circuitry to apply any modulation, particularly frequency shift, to one of the two transmitters. Phase rotator 122 applies a predefined additional phase shift per transmit (Tx) channel / antenna. The high-frequency signal is then passed to power amplifier circuitry 124, which amplifies the radar transmit signals and routes them to the two transmitter antennas 103, thereby generating FMCW transmissions with the additional modulation implemented by phase rotator 122. The number of antennas used may depend on the number of radar receiver and transmitter channels supported / implemented in a given radar unit.

[0005] The receiver includes receiver front-end circuitry that effectively provides frequency conversion in low-noise amplification circuitry 104, down-mixer circuitry 106 (operably coupled to RefLO 128), and filtering (in this example, bandpass filtering circuitry 109 having a high-pass filter (HPF) and a low-pass filter (LPF). In this IQ receiver implementation, the received signal is shown as being divided into two receiver paths and then down-converted by down-mixer circuitry 106, which applies a 90-degree phase shift 127 to one path of the reference signal provided by RefLO 128. A reference clock from reference clock circuitry 126 controls the phase rotation of the Tx phase rotator. Finally, the two down-converted IQ receive signals are input to analog-to-digital converter circuitry 110 to convert the received analog radar signal into a digitized version, which is provided to a digital signal processor (DSP) 108. The DSP processes the received and digitized symbols with the known transmitted symbols to derive radar information, such as the range and velocity of the detected object. In this known standard IQ receiver radar unit 100, the DSP 108 includes a quadrature summing circuit 112 that sums the I+jQ receiver signals. In theory, this can provide twice the maximum range of a real (e.g., 'I' branch only) receiver. However, in practice, due to IQ imbalance and other imperfections in the received signal, visibility of distant targets with a low signal-to-noise ratio (SNR) of the received radar signal is not feasible / achievable in the presence of spurious signals from nearby strong targets. Therefore, doubling the maximum range using real hardware is not possible. The quadrature summing circuit 112 then passes the summed received radar signal to a circuit or function that performs a range fast Fourier transform (FFT) 114 on the IQ signal, followed by a Doppler FFT 116 and DOA processing 118.

[0006] An alternative known technique is for DSP 108 to employ a Hilbert filter (not shown) between quadrature summation circuit 112 and range FFT 114. Using a Hilbert filter allows the positive and negative spectra to be coherently combined, and then the negative portion is removed (where the real portion of the signal is obtained without sacrificing SNR). Thus, the noise figure is improved (+3 dB) by coherently combining the two IQ branches in the positive portion of the spectrum (where real radar targets are expected).

[0007] Radar systems that operate with multiple transmit and receive channels are known as multiple-input, multiple-output (MIMO) radars. It is known that angular scan time in radar applications can be minimized by applying a MIMO scheme, rather than sequentially scanning in each direction. A dedicated decoder can be employed to retrieve angular information through post-processing of the received signal. Implementing MIMO radars presents the challenge of allocating sufficient time-frequency resources to the transmit waveform to accommodate each transmitter, allowing them to be distinguished during reception.

[0008] Known MIMO radar multiplexing schemes that are considered to operate 'K' Tx channels include:

[0009] 1) Time Division Multiple Access (TDMA), which separates Tx channels in the time domain. TDMA limits the instantaneous transmit power to that of a single Tx (which is not the case for other schemes) and reduces the explicit speed range by 'K' times.

[0010] 2) Doppler Division Multiplexing (DDMA), which separates Tx signals via Doppler shift and thus results in the same unambiguous speed interval degradation as TDMA.

[0011] 3) Range Division Multiplexing (RDMA), which separates Tx channels using large range (or beat frequency) offsets, reducing the maximum measurable distance by a factor of K. RDMA also introduces additional challenges to direct Tx-Rx coupling suppression (where a simple high-pass filter can be replaced with a multi-notch filter) and may result in range ambiguity.

[0012] 4) (Range) Circular Linear Frequency Modulation (LFM), another approach to implementing MIMO radar. Here, the Txs are typically placed in adjacent range bins, which directly reduces the range resolution by a factor of 'K' and requires a high-accuracy phase rotator for hardware implementation.

[0013] 5) Code division multiplexing (CDMA) can also be implemented in slow time, fast time, or both. In all three cases, given that the signal processing is a linear transform, the observation time determined by the system update requirements limits the total code duration and imposes a limit on the achievable dynamic range of the system. Direct leakage cancellation of fast-time coding presents another problem with this approach. Although some improvement in dynamic range can be achieved using nonlinear signal processing, this operation is generally computationally significantly more expensive than conventional two-dimensional (2D) fast Fourier transform (FFT) range-Doppler processing.

[0014] The present inventors have recognized and appreciated that the implementation of a practical radar unit comes at the expense of some performance degradation in the range-Doppler image compared to single Tx operation (phased array radar).

[0015] US20210173069A1 describes a transmission scheme with a small frequency shift that can be captured by an ADC without the need for an additional mixer or a change in reference frequency. A dedicated multi-notch filter is required to remove bumper response and leakage. US20080112519A1 describes a circuit that uses multiple reference LOs to sample different parts of the spectrum. Mixing is performed with a certain frequency tone, and both the 'I' and 'Q' branches are used to capture data in each sub-band.

[0016] Therefore, there is a need for an improved radar unit, a circuit, and a method to utilize two (eg, 'I' and 'Q') branches of a radar receiver to support different Tx antennas (or groups of Tx antennas) transmitting simultaneously. Summary of the Invention

[0017] The examples described herein provide a radar unit, a circuit therefor, and a method therefor, as described in the accompanying claims. Specific embodiments are set forth in the dependent claims. These and other aspects will be apparent from and elucidated with reference to the embodiments described hereinafter.

[0018] In a first aspect, a radar unit is described, comprising a transceiver having a reference local oscillator (RefLO) configured to generate a reference signal; a reference clock circuit configured to output a clock signal; and at least two transmitter paths, each of which includes a phase rotator configured to receive the reference signal and a clock signal, wherein the clock signal rotates the phase of the reference signal in at least one transmitter path, wherein a first frequency-modulated continuous wave (FMCW) radar signal in a first transmitter path is shifted by a sampling frequency Fs relative to a second FMCW radar signal in a second transmitter path. Each of the at least two transmitter paths includes a power amplifier configured to amplify the corresponding first or second FMCW radar signal and apply the amplified FMCW radar signal to a corresponding first and second transmit antennas. At least one receiver is coupled to a receive antenna that receives reflections of the FMCW radar signal, the receive antenna being coupled to the two receiver paths. Downconversion circuitry, coupled to the RefLO, is configured to downconvert the reflected FMCW radar signal in each receiver path to a baseband signal. The downconversion circuitry includes a frequency shifter circuit located in a first receiver path and configured to apply a frequency shift to a received reflection of an FMCW radar signal or a reference signal in one path. The reference signal downconverts the reflection of the FMCW radar signal in the first receiver path to a baseband signal in the frequency band represented by the FMCW transmit signal after the frequency shift applied by the phase rotator. A bandpass filter is configured to filter the baseband signal in each receiver path, wherein a first bandpass filter in the first receiver path is configured to pass the baseband signal, and a second bandpass filter in the second receiver path is configured to pass the baseband signal in the second transmitter path after the applied frequency shift has been removed. An analog-to-digital converter (ADC) circuit is configured to convert the filtered baseband signal in each receiver path to digital form; and a digital signal processor (DSP) operatively coupled to at least one receiver is configured to process the filtered baseband signal in digital form and, in response thereto, determine the angular dimension of the received reflected radar signal.

[0019] Illustratively, the frequency shift applied by the phase rotator in one of the at least two transmitter paths and the frequency shift applied by the frequency shifter circuit in the first receiver path are each configured to be greater than or equal to an ADC sampling frequency δ f ≥f s , the ADC sampling frequency is applied to one of the at least two transmitter paths, and the received beat signal associated with the one of the at least two transmitter paths falls outside a passband frequency of one receiver path.

[0020] Illustratively, the phase rotator is configured to apply a frequency shift to the at least one transmitter signal as a phase ramp at a clock rate of the reference clock circuit.

[0021] Illustratively, the downconversion circuit includes a first downconversion mixer in the first receiver path and a second downconversion mixer in the second receiver path, wherein the clock signal from the reference clock circuit produces a frequency shift input to a baseband mixer coupled to an output of one of the two receiver paths in the downconversion circuit.

[0022] For example, δ f The frequency modulation is applied to the baseband mixer before the bandpass filter and the ADC, where the δ f The frequency tone is configured to frequency compensate for the frequency shift applied in the at least one of the transmitter paths.

[0023] Illustratively, the frequency shifter circuit includes a fixed π / 2 phase shift, and the baseband mixer is selectively configured to receive the down-converted signal in response to a control signal applied to a bypass switch, wherein the control signal is configured to switch the radar unit to operate as a multiple-input multiple-output (MIMO) mode of operation or as a conventional radar receiver.

[0024] Illustratively, the frequency shifter circuit includes a phase rotator operably coupled to a down-conversion mixer located in one of the two receiver paths, and the frequency shifter circuit is configured to apply a frequency shift to the reference signal, which frequency compensates for the frequency shift after processing by the phase rotator and the bandpass filter and the ADC circuit.

[0025] Illustratively, the at least one receiver is a quadrature IQ receiver.

[0026] For example, the DSP is configured to:

[0027] summing the digital versions of the filtered baseband signals provided by the ADC circuit as real and complex parts and performing a range Fast Fourier Transform (FFT) across both positive and negative frequency spectra; and

[0028] The real and complex parts are separated using the FFT, which applies:

[0029]

[0030] where S(f)=F(s(t)) is the frequency and time representation of the received digital signal, () *represents the complex conjugate, and S(f + )、S(f - ) represent the positive and negative spectrum parts respectively.

[0031] Illustratively, the ADC circuit is configured to operate at a sampling frequency f_s configured to capture a signal component corresponding to each of the at least two transmitter paths, wherein each of the at least two transmitter paths occupies a frequency band of fs / 2.

[0032] Illustratively, the radar transceiver is configured to support one of: multiple-input single-output (MISO) communication or multiple-input multiple-output (MIMO) communication.

[0033] For example, an integrated circuit for a radar unit includes a radar transceiver, the radar transceiver including:

[0034] a reference local oscillator (Ref LO) configured to generate a reference signal;

[0035] a reference clock circuit configured to output a clock signal;

[0036] at least two transmitter paths, wherein each transmitter path includes a phase rotator configured to receive the reference signal and the clock signal, wherein the clock signal rotates a phase of the reference signal in at least one transmitter path, wherein a first frequency modulated continuous wave (FMCW) radar signal in a first transmitter path is shifted by a sampling frequency Fs relative to a second FMCW radar signal in a second transmitter path, wherein each of the at least two transmitter paths includes a power amplifier configured to amplify the respective first FMCW radar signal or the second FMCW signal and apply the respective amplified FMCW radar signal to the respective first transmit antenna or the second transmit antenna;

[0037] at least one receiver coupled to a receive antenna that receives reflections of the FMCW radar signal, the receive antenna coupled to two receiver paths;

[0038] a downconversion circuit coupled to the Ref LO and configured to downconvert the reflected FMCW radar signal in each receiver path to a baseband signal, wherein the downconversion circuit includes a frequency shifter circuit located in a first receiver path and configured to apply a frequency shift to a received reflection of the FMCW radar signal or one path of the reference signal, the reference signal downconverting the reflection of the FMCW radar signal in the first receiver path to a baseband signal of a frequency band represented by the FMCW transmit signal after the frequency shift applied by the phase rotator;

[0039] a bandpass filter configured to filter the baseband signal in each receiver path, wherein a first bandpass filter in a first receiver path is configured to pass the baseband signal, and a second bandpass filter in a second receiver path is configured to pass the baseband signal of the second transmitter path after an applied frequency shift has been removed; and

[0040] analog-to-digital converter (ADC) circuitry configured to convert the filtered baseband signal in each receiver path into digital form; and

[0041] a digital signal processor (DSP) operatively coupled to the at least one receiver and configured to process the digital form of the filtered baseband signal and to determine an angular dimension of a received reflected radar signal in response to the digital form.

[0042] In a second aspect, a method for receiving two transmitted radar signals in a radar unit is described. The method includes generating a reference signal using a reference local oscillator (RefLO); rotating the phase of the reference signal in one of at least two transmitter paths providing a frequency modulated continuous wave (FMCW) radar signal using a clock signal applied to a phase rotator, wherein the signal of the first transmitter is shifted by a sampling frequency Fs relative to the second transmitter. The method further includes amplifying and transmitting a first FMCW radar signal from a first transmit antenna and a second FMCW radar signal from a second transmit antenna; receiving reflected radar signals, and amplifying the received reflected radar signals in two receiver paths. The method further includes down-converting the received amplified reflected radar signals to baseband signals in the two receiver paths using the reference signal, including applying a frequency shift to any received reflection of the FMCW radar signal in the first receiver path; or applying a frequency shift to the reference signal used to down-convert a reflection of the FMCW radar signal in the first receiver path. The method further comprises: bandpass filtering the baseband signals, wherein a first down-converted baseband signal of the received reflected radar signal is passed by a first bandpass filter in a first receiver path, and a second down-converted baseband signal of the received reflected radar signal is passed by a second bandpass filter in a second receiver path; analog-to-digital converting the down-converted, filtered baseband signals; and digitally processing the down-converted, filtered baseband signals in digital form.

[0043] Exemplarily, the method further comprises configuring the frequency shift applied by the phase rotator in one of the at least two transmitter paths and the frequency shift applied by the frequency shifter circuit in the first receiver path to be each greater than or equal to an ADC sampling frequency δ applied to one of the at least two transmitter paths. f ≥f s , and the received beat signal associated with the one of the at least two transmitter paths falls outside a passband frequency of one receiver path.

[0044] Illustratively, the method further comprises applying a frequency shift to the at least one transmitter signal as a phase ramp at a clock rate of the reference clock circuit.

[0045] Illustratively, the down-conversion circuit includes a first down-conversion mixer in the first receiver path and a second down-conversion mixer in the second receiver path, and the method further includes generating a frequency shift input to a baseband mixer by the clock signal from the reference clock circuit, the baseband mixer being coupled to an output of one of the two receiver paths in the down-conversion circuit.

[0046] For example, the method further comprises: f The frequency modulation is applied to the baseband mixer before the bandpass filter and the ADC, where the δ f The frequency tone is configured to frequency compensate for the frequency shift applied in the at least one of the transmitter paths.

[0047] Illustratively, the frequency shifter circuit includes a fixed π / 2 phase shift, wherein the method further includes selectively configuring the baseband mixer to receive the down-converted signal in response to a control signal applied to a bypass switch, and configuring the control signal to switch the radar unit to operate as a multiple-input multiple-output (MIMO) operating mode or as a conventional radar receiver.

[0048] Illustratively, the method further comprises:

[0049] operatively coupling a phase rotator of the frequency shifter circuit to a downconversion mixer in one of the two receiver paths; and

[0050] A frequency shift is applied to the reference signal by the frequency shifter circuit, which is frequency compensated for the frequency shift after processing by the phase rotator and the bandpass filter and ADC circuit.

[0051] Illustratively, the at least one receiver is a quadrature IQ receiver. BRIEF DESCRIPTION OF THE DRAWINGS

[0052] Additional details, aspects, and embodiments will be described by way of example only with reference to the drawings. In the drawings, like reference numerals are used to identify similar or functionally similar elements. The elements in the drawings are shown for simplicity and clarity and are not necessarily drawn to scale.

[0053] Figure 1 A simplified block diagram of a known quadrature (IQ) transceiver is shown.

[0054] Figure 2 A simplified example block diagram of a radar transceiver adapted according to the examples described herein is shown.

[0055] Figure 3Various example implementations of frequency shifter circuits according to the examples described herein are shown.

[0056] Figure 4 Further various example implementations of frequency shifter circuits according to the examples described herein are shown.

[0057] Figure 5 An example flow diagram of a quadrature (IQ) multiplexing transceiver adapted according to the examples described herein is shown.

[0058] Figure 6 A series of example spectra of a beat signal for IQ MIMO adapted according to examples described herein are shown (top spectrum is a full IQ receiver without baseband mixers; middle spectrum is the 'I' branch; and bottom spectrum is the 'Q' branch after the baseband mixers).

[0059] Figure 7 An example graph illustrating beamformed DOA estimation for a point target with two Tx by one Rx MIMO radar using an IQ MIMO scheme according to examples described herein is shown. DETAILED DESCRIPTION

[0060] The motivation for alleviating the aforementioned issues with IQ receivers is that, for an ideal IQ receiver with infinite spacing, the negative spectrum of the received radar signal can be exploited to double the maximum radar range (e.g., using the folded portion of the digital spectrum to represent the frequency [0, f_s]), thereby achieving twice the sampling frequency compared to a real receiver using [0, f_s / 2]. However, in practice, and due to the high path loss at millimeter-wave frequencies (on the order of 70-100 dB at maximum radar range), this means that radar targets at distances with high beat frequencies [f_s / 2, f_s] will be much weaker than radar targets at closer ranges. The inventors have recognized and appreciated that this dictates high IQ isolation requirements for detecting these targets below the IQ imbalance image of the signal at close ranges [0, f_s / 2].

[0061] To address these requirements, it is proposed to avoid these large power differences between the two frequency intervals (i.e., [0, f_s / 2] and [f_s / 2, f_s]) by acquiring data from (at least) two transmitters (or transmitter groups), between which the power difference should be minimal. Specifically, for example, typical dynamic range requirements in the 'range' dimension are on the order of 70 dB, while the inventors have recognized and appreciated that for the 'angle' dimension, typical dynamic range requirements are approximately 30 dB, determined by array calibration accuracy and RF imperfections. It is important to note that this suppression ratio is typically unacceptable in the range domain due to the significant received power variation relative to target range according to the radar equation at mmWave. The same 30-40 dB suppression ratio is typically sufficient in the angular domain, where RF impairments limit the dynamic range by a similar value. Therefore, the inventors have recognized and appreciated that by adding IQ imbalance constraints to this domain, system performance is not degraded.

[0062] Thus, the examples described herein propose a radar unit, a circuit and a method to utilize substantially the full frequency band [0, f s ]. In the example, the 'I' and 'Q' branches of the radar receiver are also used to support different Tx antennas (or groups of Tx antennas) with simultaneous transmission. In this example, an IQ receiver with an ADC with a sampling frequency f_s captures the signal components corresponding to both Tx-1 and Tx-2 so that each of them occupies a frequency band of fs / 2 without compromising the range and Doppler dynamic range (i.e., without any loss in pre-detection signal-to-noise ratio (SNR)). The proposed method consists in using a frequency shift in a set of transmitter signals so that they are rejected by the receiver bandpass filter in one branch (I), but can be observed in the other branch (Q) by an additional mixer located in this (Q) branch after removing the applied frequency shift. Thus, a portion of the Tx channel is acquired via the I branch, while another portion of the Tx channel is acquired in the Q branch of each receiver. In this way, any hardware imperfections (IQ imbalance / non-orthogonality) are therefore moved to the corner processing domain, where the requirements on dynamic range are weaker and therefore IQ hardware imperfections are not critical (or are of the same order of magnitude or lower than other imperfections, such as array coupling).

[0063] To ensure Tx separation at the receiver, the frequency shift δ greater than or equal to the ADC sampling frequency can be f ≥f sA set of Tx channels (e.g., 40 MHz) is applied to the MIMO radar so that the received beat signal associated with this Tx channel falls outside the frequency band acquired in one (e.g., 'I') branch. This frequency shift can be achieved using an additional analog baseband mixer with δ f The frequency shift is compensated for by the frequency tone at ' ' (after mixing with the RF chirp, but before applying the filters in the receiver path to the ADC). Alternatively, this frequency shift can be compensated by shifting the reference chirp frequency (Rx-LO) with a phase rotator in the other (Q) branch so that the useful signal passes through a high-pass filter (HPF) and a low-pass filter (LPF) in the same way as in the 'I' branch. In particular, direct coupling and bumper reflections are rejected by both the HPF (for the wanted band) and the LPF (for the unwanted band). RF leakage combines the effects due to many phenomena occurring at RF frequencies. RF leakage also includes multipath reflections from the radar housing (radome and / or bumper in the case of automotive applications). Therefore, bumper reflections can be considered a special case of RF leakage.

[0064] It should be noted that some key features that distinguish the examples described herein from conventional LFM radars with IQ receivers include, among others: a frequency shift greater than or equal to the ADC sampling frequency δ f ≥f s The MIMO multiplexing scheme of FIG1 is described, and the presence of an additional analog baseband mixer in the Q branch before the bandpass (HPF / LPF) filter, or a shift in the frequency reference chirp in one of the (I or Q) branches. In other examples, the proposed modification may also include a baseband switch (with frequency modulation or DC mixing) so that the additional MIMO capability of the proposed scheme is disabled (for example, for Tx beamforming mode). It is envisioned that yet another embodiment may also incorporate similar baseband mixers in both the 'I' and 'Q' branches for balancing the outputs of the two branches.

[0065] The examples described herein create a virtual array in a MIMO radar by correlating the received signal with the transmitter that transmits the radar signal. In this context, a virtual array can be viewed as an abstraction describing a phased array that will provide a phase response of a target equivalent to that obtained by combining coherent MIMO channels. Thus, the combination of a MIMO waveform and specific signal processing for linear frequency modulated (LFM) signals is designed to exploit separation in the virtual antennas of the co-located Tx and Rx arrays. For the co-located arrays, the spatial convolution of the Tx and Rx arrays creates a virtual array with more virtual elements than physical channels. Consequently, the virtual array has higher angular resolution and / or fewer grating lobes than the Tx and Rx arrays, respectively.

[0066] In some examples, an additional baseband mixer or additional phase rotator can be used to bring the signal from another Tx (or Tx group) into the passband of the ADC in a receiver with a DSP configured to perform stretch processing. A DSP configured to perform stretch processing (which may be referred to as a de-skew receiver or de-chirp receiver) can be implemented using an IQ receiver, meaning there are two branches per receive antenna that are coherently combined. In this way, the radar receiver is configured to transmit a frequency modulated signal (typically a linear frequency modulated signal, such as a chirp signal) and mix the received signal with the transmitted chirp in the receiver. In this scenario, information about the target range is then embedded in the beat frequency difference between the Tx chirp and the Rx chirp (where the Rx signal is a linear combination (e.g., a sum) of multiple responses from point targets, and each individual point target response is a time-delayed and scaled copy of the Tx RF chirp). In this way, information about the range of a target can be captured with an ADC operating at a sampling frequency significantly smaller than the bandwidth of the chirp (typically 2-50 times, depending on the application). Although examples are described herein for radar units operating in the millimeter-wave frequency range, it is envisioned that the concepts described herein will operate over a very wide frequency range (e.g., 1 MHz to 1 THz).

[0067] Now refer to Figure 2 , shows a simplified example block diagram of a radar unit having an example radar transceiver 200 adapted according to the examples described herein and configured to operate at mmw frequencies. The radar unit includes one receive antenna 202 for receiving radar signals and two or more transmitter antennas 203 for transmitting FMCW radar signals. The FMCW radar signals are generated by a reference local oscillator (Ref. LO) 228 and have additional modulation implemented by a phase rotator 222. The signals are then passed to a power amplifier circuit 224, which amplifies the radar transmit signals and routes them to multiple (e.g., two) transmitter antennas 203. In one example, the phase of the reference LO signal 228 in at least one of the two transmitter paths is rotated in response to a reference clock signal from a reference clock circuit 226 applied as a phase ramp to the phase rotator 222 at a clock rate. Thus, both phase rotators 222 are controlled by a control sequence synchronously aligned with the reference clock signal. However, the first phase rotator is configured to apply phase to achieve frequency shift, and the second phase rotator is configured to always apply a constant phase (and therefore no frequency shift). In this way, a frequency modulated continuous wave (FMCW) radar signal is generated, in which the signal of the first transmitter is shifted by the sampling frequency Fs relative to the second transmitter. The vertical arrows show the timing of the phase rotators by the reference clock signal, while the arrows with the clock frequency offset provide the control input.

[0068] The receiver includes receiver front-end circuitry operatively coupled to a receiver antenna 202, effectively providing a low-noise amplifier circuit 204. In this IQ receiver implementation, the received signal is shown as being divided into two receiver paths and then downconverted by downconversion circuitry 206, which performs a frequency / phase shift on one path of a reference signal provided by Ref LO 228 by a frequency shifter 229. Downconversion circuitry 206 outputs the downconverted radar signal to baseband filter 209 (which, in this example, is a bandpass filtering circuit having a high-pass filter (HPF) and a low-pass filter (LPF)). Finally, the two downconverted IQ received radar signals are input to analog-to-digital converter (ADC) circuitry 210, which converts the received IQ-converted analog radar signals into digitized versions and provides the digitized versions to a digital signal processor (DSP) 208. The reference clock signal of the reference clock circuit 226 is coupled to the ADC circuit 210 and the frequency shifter 229 and is configured to synchronize the ADC sampling, the frequency shift applied by the phase rotator on Tx, and the frequency shift applied by the frequency shifter in Rx. As will be appreciated by those skilled in the art, the Ref LO 228 should operate at a radio frequency (e.g., in this example, at 77 GHz), while the reference clock should be configured to operate at the sampling rate of the ADC circuit 210 (e.g., at 40 MHz or 80 MHz).

[0069] The DSP processes the received and digitized symbols with the known transmitted symbols to derive radar information about, for example, the range and velocity of detected objects. In this known standard IQ receiver radar unit, the DSP 208 includes a quadrature summing circuit 212 that sums the I+jQ receiver signals. To correct for IQ imbalance and other imperfections, the DSP 208 has been modified as follows.

[0070] The quadrature summation circuit 212 then passes the summed received radar signal to a circuit or function that performs a range fast Fourier transform (FFT) 240 on the IQ signal and splits the output FFT signal into two spectral components, a positive spectrum S+ and a negative spectrum S-, which are then combined by range FFT algebraic operators 252 and 254, respectively, each of which corresponds to a particular Tx channel being utilized. In some examples, the signal processing performed by the range FFT 240 for both the 'I' and 'Q' branches can be performed using a single FFT by combining the signal into real and complex components and separating the real and complex components using the properties of the Fourier transform after the FFT, namely:

[0071]

[0072] Where S(f)=F(s(t)) is the frequency and time representation of the signal, () * represents the complex conjugate, and S(f + )、S(f - ) represent the positive and negative parts of the spectrum, respectively.

[0073] Based on this processed information, the frequency shifter circuit 229 is then adapted. In this way, the processed information serves the dual purpose of downconverting the combined LO and demodulating the frequency shift applied in Tx1. The positive S+ algebraic operator 252 and the negative S- algebraic operator 254 of the spectrum of the range FFT 240 are followed by the corresponding Doppler FFTs 262 and 264, and the phase correction circuit 217, which digitally corrects for the phase differences between the channels due to the application of the Tx frequency and any phase offsets. The received IQ range and Doppler FFT signals are then passed to the DOA processing circuit 218.

[0074] In some examples, the transmitted signal proposed herein may be similar to a frequency division (or distance division) multiplexing scheme (FDMA), where the transmitted signal is configured to utilize a frequency domain with respect to the ADC sampling frequency f ADC Here, the proposed IQ receiver uses δ f ≥f ADC , while distance division multiplexing uses less than δ f ≤f ADC / K or for real receivers even where K is the number of Tx channels utilized in this (MIMO) mode.

[0075] In some examples, the phase term -2πδ may be performed after a range FFT 240 with an estimated value τ per range bin and ignoring target displacement within one chirp. f A range-dependent correction of τ(t) is performed. In some examples, those skilled in the art will appreciate that this can be performed in a manner similar to the methods used in FDMA processing for range DOA decoupling. Here, in some examples, the phase correction can also compensate for any additional phase shift introduced into the signal at the Tx side (e.g., by PR) or in the receiver baseband mixer.

[0076] The modulation applied on the Tx side and the baseband mixer input on the Rx side can differ from the pure COS frequency tone previously considered. Due to limited hardware accuracy or the update rate of the PR or the analog implementation of the baseband mixer (for example, by switching the sign, that is, equivalent to mixing with a square wave), both signals may have some harmonics (for example, square waves). It is conceivable that in some cases, there may be slight differences in performance when mixing with a square wave instead of a COS wave.

[0077] In some examples, it is contemplated that the proposed IQ receiver can be a multiple-input single-output (MISO) system corresponding to a single Rx channel. In some examples, it is contemplated that the proposed IQ receiver can be a multiple-input multiple-output (MIMO) system corresponding to multiple Rx channels. In some examples, the proposed IQ receiver concept employs a frequency shifter 229 in the 'Q' branch, which is used to shift the transmitter's spectrum, which has additional frequency shifts implemented by the HFP and LPF in the baseband bandpass filter 209 and is acquired by the ADC circuit 210.

[0078] It is also contemplated that, in other examples, two linear chirp generators (e.g., phase locked loops (PLLs) as respective RefLOs 228) may be utilized that, together with the frequency shifter 229, apply the desired frequency shift δ in both the I and Q branches of the receiver. f These waveform generators can also be placed in different integrated circuits (ICs) that are remotely synchronized for coherent operation. In this example, the phase rotator 222 that generates the frequency shift on the two Tx paths can be omitted.

[0079] The proposed transmission scheme, receiver structure and digital signal processing Figure 2 The principle of can be illustrated without loss of generality using the example of a single chirp and two Tx channels operating in IQ MIMO mode. More Tx channels can be multiplexed by combining IQ MIMO with other multiplexing schemes such as TDMA, DDMA, FDMA, CDMA, etc. Assume a MIMO radar with K = 2 simultaneously active channels (from each Tx) at carrier frequency f c Chirp with emission duration T and bandwidth B:

[0080]

[0081] Where β = B / T is the chirp slope, is the phase of the reference chirp at the carrier frequency, and k defines the transmitter index.

[0082] To illustrate the operating principle, a single Rx channel (MISO) is considered, but those skilled in the art will recognize that the generalization to multiple Rx channel operation is simple. The response of the target at a time delay located at a time delay τ(t) = 2r(t) / c (where r(t) is the distance to the target as a function of time and c is the speed of light) then becomes:

[0083]

[0084] Where: η k (t) is the baseband phase of the Rx signal from Txk, λ is the wavelength at the carrier frequency, and α is a constant proportional to the complex backscatter coefficient of the target, propagation loss, and other constant terms. The received RF signal is mixed with the reference RF chirp in the IQ mixer:

[0085]

[0086] The second term in both the I and Q chains corresponds to twice the carrier frequency and is filtered out by the receiver chain. For simplicity, the cutoff frequency of the LPFs in both branches is assumed to be And the signal in the suppression band is fully rejected. Then, the filtered signal in the I branch becomes:

[0087]

[0088] This corresponds to the beat signal corresponding to Tx-0.

[0089] The signal in the 'Q' branch goes to the second mixer which affects only the low frequency components (twice the carrier frequency is removed by the receiver chain). The second mixer mixes the signal with the frequency modulated cos(2πδ f t) mixing, and its output is:

[0090]

[0091]

[0092] The first three components correspond to frequencies shifted by δ f and 2δ f These components should be suppressed by the low-pass filter and anti-aliasing filter of the ADC, which captures the Therefore, only the last component is captured by the ADC after the filter in the Q branch, which is:

[0093]

[0094] The signal in the Q branch appears at the same beat frequency as in the I branch and therefore passes through the LPF and HPF in a similar manner (with the same amplitude and phase response). The phase of the signal in the Q branch contains information about the target angle as observed from Tx-1 and can therefore be used for angle processing.

[0095] It is conceivable that the frequency shifter 229 can be Figure 3 and Figure 4 The invention can be implemented in a variety of ways as shown in and described below. Figure 3 , shows various example implementations of frequency shifter circuits according to the examples described herein. A first example frequency shifter circuit 327A shows an input radar receive signal 310 provided to the down-conversion circuit 206. For example, Figure 2 A suitably configured downconverted signal 322 configured in accordance with the DSP 208 in the embodiment of the present invention is generated by the phase rotator 222, as influenced by the reference clock 326A and the Ref LO 328A. In this example, the phase rotator 222 shifts the signal from the Ref LO 328A by a frequency offset determined by the reference clock and provides this reference mixer signal to the downconversion circuit 206. This example implementation also supports dual-mode operation (if the phase rotator 222 is set to a fixed phase shift of π / 2, this example embodiment can be considered equivalent to a conventional IQ receiver).

[0096] The second example frequency shifter circuit 327B shows an input radar receive signal 360 provided to the down conversion circuit 206. Figure 2 The appropriately configured down-converted signal 322 configured by the DSP 208 in the embodiment is directly generated by the Ref LO 328B, so that the down-conversion circuit 206 outputs a baseband signal 362. The baseband signal 362 is then phase-adjusted by the reference clock 326B, such as by Figure 2 In the DSP 208 configuration in FIG, the reference clock 326B is applied to the (second) baseband mixer 366 in the Q path, for example, operating at baseband. Figure 2 Compared to the example architecture of , with the presence of an additional baseband mixer 366 in one of the quadrature branches, the IQ receiver utilizes each branch of the IQ receiver to operate a different Tx channel and, therefore, fully utilizes the signal acquired by the ADC, even in the presence of moderate impairments such as IQ imbalance in the receive chain. It is contemplated that, in some examples, additional baseband mixers may be employed in both quadrature branches, as it is easier to calibrate the system when it is fully symmetrical in terms of hardware (HW), even when the full power of this HW is not utilized in the other branch. In this case, having a symmetrical implementation for the 'I' and 'Q' branches can benefit HW calibration. Notably, the Rx branch of the IQ receiver is configured with a frequency equal to or greater than the ADC sampling frequency, along with modulation of the transmitted chirp in some Tx channels from the RF chirp generator.

[0097] Figure 3One benefit of the second example frequency shifter circuit 327B is that it can capture both sets of Tx channels simultaneously and utilize the full ADC sampling rate of the IQ pair in the presence of typical hardware impairments. Thus, the practically achievable image rejection ratio (IRR) of approximately 30-40 dB for a zero-IF (IF) or homodyne radio architecture is moved from the range measurement dimension to the array or angular dimension (requiring MISO or MIMO separation only in the DOA processing 218 in the DSP).

[0098] Now refer to Figure 4 , shows various other example implementations of frequency shifter circuits according to the examples described herein. The third example frequency shifter circuit 327C shows an input radar receive signal 410 provided to the down-conversion circuit 206. For example, Figure 2 The appropriately configured down-converted signal 322 configured by the DSP 208 in FIG. 1 is generated by Ref LO 328C and routed through fixed π / 2 phase shift 427. The subsequent down-converted signal 462 is then selectively phase-adjusted by reference clock 326C, such as by Figure 2 In the DSP 208 configuration in FIG. 4 , the reference clock 326C is applied to a second mixer 466 in the Q path, for example, operating at baseband. In this example, selective phase adjustment of a subsequent down-converted (e.g., baseband) signal is controlled via a bypass switch 422, which in some examples is controlled by a Figure 2 4. The example mode-selection embodiment is implemented by control of a switch 422, which can be used to switch the IQ MIMO mode of operation 'on' / 'off' (where, in this example, a fixed π / 2 phase shift 427 facilitates backward compatibility with conventional IQ receivers) as output 420. Thus, in this example, when the IQ MIMO mode is switched 'off' (thus operating in Mode 1), the receiver behaves as a conventional IQ receiver, where the 'I' and 'Q' branches are shifted by π / 2 relative to each other, allowing the two branches to be combined to double the range or improve the noise figure. Conventional IQ operation (Mode 1) may be beneficial in some radar modes where the field of view is limited and beamforming on the Tx is used instead of the MIMO mode (e.g., Lane Change Assist (LCA) mode of corner radar). In Mode 2, several Txs operate simultaneously in IQ MIMO mode. In other examples, it is contemplated that switching between modes may be accomplished differently, such as via controlling a signal that enters the baseband mixer along with the Rx signal (eg, between a reference clock signal and DC).

[0099] Figure 4 Also shown is a fourth example frequency shifter circuit 327D including an input radar receive signal 460 provided to the down-conversion circuit 206. Figure 2 The appropriately configured down-converted signal 322 configured by the DSP 208 in the embodiment is directly generated by the second Ref LO 328D. In this example, it is assumed that the signal from the second reference Ref LO 328D is shifted by δ relative to the reference LO from the second phase locked loop (PLL). f frequency shifted to provide baseband output 470.

[0100] While the examples described herein relate to an orthogonal (IQ) radar receiver, in which a single Rx antenna receives reflected radar signals from at least two transmitter paths, which are then split into two receiver paths, it is contemplated that the concepts described herein are equally applicable to any radar unit or radar system employing two signal receiver paths per antenna. For example, in some examples, the two receiver paths may be conventional IQ (phase 0 degrees, 90 degrees of Ref LO 328), or in other examples, the two receiver paths may simply be (phase 0 degrees, 0 degrees), as long as frequency shifting is still applied.

[0101] Now refer to Figure 5 , shows an example flow chart 500 of a radar transceiver according to examples described herein, which in one example includes a quadrature (IQ) receiver. At 510, the flow chart includes a reference local oscillator Ref LO (e.g., Figure 2 At 515, the flow chart includes the steps of applying a phase rotator (e.g., Figure 2A reference clock signal (e.g., a phase rotator 222 in the transmitter) is used to rotate the phase of a reference signal in one of at least two transmitter paths that provide a frequency-modulated continuous wave (FMCW) radar signal, wherein the signal of the first transmitter is shifted by a sampling frequency Fs relative to the second transmitter. At 520, the flowchart includes amplifying and transmitting a first FMCW radar signal from a first transmit antenna and a second FMCW radar signal from a second transmit antenna. At 530, the flowchart includes receiving a reflected radar signal and amplifying the received reflected radar signal in two receiver paths. At 540, the flowchart includes downconverting the received, amplified, reflected radar signal to baseband signals in both receiver paths using a reference signal, including: applying a frequency shift to any received reflection of the FMCW radar signal in the first receiver path; or applying a frequency shift to the reference signal used to downconvert a reflection of the FMCW radar signal in the first receiver path; wherein the frequency shift shifts the reflection of the FMCW radar signal to a baseband signal of the frequency band represented by the FMCW transmit signal after compensating for a frequency shift applied by a phase rotator in one of the transmitted FMCW radar signals. At 550, the flowchart includes bandpass filtering the baseband signals, wherein a first downconverted baseband signal of the received reflected radar signal is passed by a first bandpass filter in the first receiver path. Additionally, a second downconverted baseband signal of the received reflected radar signal is passed by a second bandpass filter in the second receiver path (i.e., configured to pass the baseband signal of the second transmitter path after the applied frequency shift has been removed). For example, after removing the frequency offset by mixing with the reference clock signal (40 MHz), the useful signals (of different TXs) in the two paths are in the same frequency band. At 560, the flowchart includes performing analog-to-digital conversion on the down-converted, filtered baseband signal; and at 570, the flowchart includes performing digital processing on the digital form of the down-converted, filtered baseband signal.

[0102] simulation

[0103] Figure 6 Shown is a series of example spectra 600 (dB 602 versus frequency shift 604) of a beat signal for IQ MIMO adapted according to examples described herein.

[0104] In the first (top) example spectrum 610, the spectrum obtained from a typical IQ receiver with some imbalance after the RF down-conversion mixer is depicted. The positive spectrum has a first response 611 of the first Tx at a beat frequency of about 4 MHz, and leakage 614 to the negative spectrum at -4 MHz due to hardware imperfections. The IQ imbalance causes about -35 dB leakage of the positive spectrum into the negative range, which ideally should be null. The second Tx has a second response 613 at 44 MHz (due to the applied Tx-2 frequency shift δ f=40MHz==f ADC ADC sampling rate) and the corresponding IQ leakage 612 at -44Mhz in the negative spectrum. Here, the dotted line 615 shows the passband of the anti-aliasing filter and the ADC, so that the signal of the second Tx appears in this band Besides.

[0105] In the second (middle) example spectrum 640, the spectrum obtained in the real branch shows a symmetrical response 641 at 4 MHz and -4 MHz. In addition, the second example spectrum 640 shows the response 643 of the second transmit Tx-2 signal at +44 MHz and the response at -44 MHz 642. Here, the dotted line 644 shows the cutoff frequency of the LP anti-aliasing filter. Therefore, any signal with an absolute frequency above these limits will be rejected by the ADC and not captured by the ADC. Therefore, the signal of Tx-2 is outside the ADC passband and will be removed by the LP anti-aliasing filter. Therefore, in this way, the I branch only captures the signal of Tx-1.

[0106] In the third (bottom) example spectrum 670, the obtained spectrum is shown after mixing with the applied frequency shift (i.e., at Figure 3 The second baseband mixer 366 in Figure 4 466 in FIG. Here, the lines labeled 673 at 36 MHz and 44 MHz and 672 at -36 MHz and -44 MHz show the signals of Tx-1 shifted there by the real baseband mixer. These signals, along with the signals of Tx-2 at twice the baseband mixer frequency 674 at -76 and 76 MHz shifts, are filtered out by the LPF. Therefore, only Tx-2 appears in the ADC's passband, and the 'Q' branch captures Tx-2's signals only at 4 and -4 MHz 675 (since the branches are real channels, the signals are symmetrical). Thus, in this way, the signals of both Txs can be isolated and processed for DOA estimation.

[0107] Now refer to Figure 7 , dB 710 versus spatial frequency 720, an example graph 700 shows a beamformed direction of arrival (DOA) angle FFT estimate graph 724 for a point target with two Tx by one Rx MIMO radars using an IQ MIMO scheme according to the examples described herein. The example graph 700 demonstrates the ability to estimate target DOA using the proposed scheme, with a peak location 722 of -2 dB identified at an identified true location 726 with a spatial frequency 720 of +0.2.

[0108] It is contemplated that the examples described herein can be used within MISO or in combination with other MIMO multiplexing techniques, such as CDMA, FDMA, and Doppler Division Multiple Access (DDMA). In DDMA, each transmitter is assigned a subband in the Doppler spectrum estimated via a slow-time FFT. Consequently, transmitters are shifted by different frequencies relative to each other in slow time (where the phase varies linearly with slow time, varying differently for each transmitter). Furthermore, in some examples, it is desirable to implement imaging radars with a large number of simultaneously operating transmitter channels and equipped with IQ receivers. Therefore, in some examples, all supported transmitters are divided into two groups, where different multiplexing schemes can be applied to distinguish between transmitters within each group. For example, a radar with eight simultaneously operating transmitters can be split into two groups of four transmitters using IQ MIMO. Within each group of four transmitters, for example, DDMA coding can be applied to distinguish between the four transmitters within the group. Effectively, the number of transmitters to be multiplexed is reduced by a factor of two compared to current (DDMA-only) solutions.

[0109] Although examples have been described with reference to MISO or MIMO radar units, it is contemplated that the concepts described herein, utilizing similarly generated waveforms and receiver techniques, are equally applicable to acoustic navigation and ranging (sonar) or light detection and ranging (lidar) communication units that are affected by quadrature imbalance or leakage.

[0110] In the foregoing specification, examples have been described with reference to specific exemplary embodiments. However, it will be apparent that various modifications and changes may be made herein without departing from the scope of the invention as set forth in the appended claims and the claims are not limited to the specific examples described above.

[0111] As discussed herein, the connection can be any type of connection suitable for, for example, transmitting a signal from a corresponding node, unit or device via an intermediate device or transmitting a signal to a corresponding node, unit or device. Accordingly, unless otherwise implied or stated, the connection can be, for example, a direct connection or an indirect connection. The connection can be shown or described as a single connection, multiple connections, a unidirectional connection or a bidirectional connection. However, different embodiments can change the implementation scheme of the connection. For example, a separate unidirectional connection can be used instead of a bidirectional connection, and vice versa. In addition, multiple connections can be replaced by a single connection that transmits multiple signals in a continuous manner or in a time-division multiplexing manner. Similarly, a single connection carrying multiple signals can be divided into various different connections that carry a subset of these signals. Therefore, there are many options for transmitting signals. Those skilled in the art will recognize that the architecture described herein is only exemplary, and in fact, many other architectures that realize the same functionality can be implemented.

[0112] Any arrangement of components that achieve the same functionality is effectively 'associated' so as to achieve the desired functionality. Thus, any two components herein combined to achieve a particular functionality can be considered to be 'associated' with each other so as to achieve the desired functionality, regardless of architectures or intermediary components. Likewise, any two components so associated can also be considered to be 'operably connected' or 'operably coupled' with each other so as to achieve the desired functionality.

[0113] Furthermore, those skilled in the art will recognize that the boundaries between the operations described above are merely illustrative. Multiple operations may be combined into a single operation, a single operation may be dispersed within additional operations, and the execution of operations may at least partially overlap in time. Furthermore, alternative embodiments may include multiple instances of a particular operation, and the order of the operations may be altered in various other embodiments. Moreover, for example, in one embodiment, the illustrated examples may be implemented as circuitry located on a single integrated circuit or within the same device.

[0114] In some examples, various components within the receiver may be implemented as discrete or integrated components, so the final structure is a matter of application or design choice. Since the examples shown can largely be implemented using electronic components and circuits known to those skilled in the art, in order to understand and appreciate the basic concepts of the described examples and to avoid obscuring or detracting from the teachings described herein, no further details will be explained than deemed necessary as described below. It will be appreciated by those skilled in the art that the level of integration of the receiver circuits or components may, in some cases, depend on the implementation.

[0115] In the claims, any reference numerals placed between parentheses should not be construed as limiting the claim. The word 'comprising' does not exclude the presence of elements or steps other than those listed in the claim. In addition, as used herein, the terms 'a' or 'an' are defined as one or more than one. Moreover, the use of introductory phrases such as 'at least one' and 'one or more' in the claims should not be interpreted as implying that another claim element introduced by the indefinite article 'a' or 'an' will include such introduced claim element, and any particular claim is limited to an invention containing only one such element, even when the same claim includes the introductory phrases 'one or more' or 'at least one' and an indefinite article such as 'a' or 'an'. The same applies to the use of definite articles. Unless otherwise stated, terms such as 'first' and 'second' are used to arbitrarily distinguish between the elements described by such terms. Therefore, these terms are not necessarily intended to indicate a temporal or other priority of such elements. The mere fact that certain measures are recited in different claims does not indicate that a combination of these measures cannot be used to advantage.

Claims

1. A radar unit, characterized in that: A transceiver is included, the transceiver comprising: a reference local oscillator (Ref LO) configured to generate a reference signal; a reference clock circuit configured to output a clock signal; at least two transmitter paths, wherein each transmitter path includes a phase rotator configured to receive the reference signal and the clock signal, wherein the clock signal rotates a phase of the reference signal in at least one transmitter path, wherein a first frequency modulated continuous wave (FMCW) radar signal in a first transmitter path is shifted by a sampling frequency Fs relative to a second FMCW radar signal in a second transmitter path, wherein each of the at least two transmitter paths includes a power amplifier configured to amplify the respective first FMCW radar signal or the second FMCW signal and apply the respective amplified FMCW radar signal to the respective first transmit antenna or the second transmit antenna; at least one receiver coupled to a receive antenna that receives reflections of the FMCW radar signal, the receive antenna coupled to two receiver paths; a downconversion circuit coupled to the Ref LO and configured to downconvert the reflected FMCW radar signal in each receiver path to a baseband signal, wherein the downconversion circuit includes a frequency shifter circuit located in a first receiver path and configured to apply a frequency shift to a received reflection of the FMCW radar signal or one path of the reference signal, the reference signal downconverting the reflection of the FMCW radar signal in the first receiver path to a baseband signal of a frequency band represented by the FMCW transmit signal after the frequency shift applied by the phase rotator; a bandpass filter configured to filter the baseband signal in each receiver path, wherein a first bandpass filter in a first receiver path is configured to pass the baseband signal, and a second bandpass filter in a second receiver path is configured to pass the baseband signal of the second transmitter path after an applied frequency shift has been removed; and analog-to-digital converter (ADC) circuitry configured to convert the filtered baseband signal in each receiver path into digital form; and a digital signal processor (DSP) operatively coupled to the at least one receiver and configured to process the digital form of the filtered baseband signal and to determine an angular dimension of a received reflected radar signal in response to the digital form.

2. The radar unit according to claim 1, characterized in that The frequency shift applied by the phase rotator in one of the at least two transmitter paths and the frequency shift applied by the frequency shifter circuit in the first receiver path are each configured to be greater than or equal to an ADC sampling frequency δ f ≥f s , the ADC sampling frequency is applied to one of the at least two transmitter paths, and the received beat signal associated with the one of the at least two transmitter paths falls outside a passband frequency of one receiver path.

3. The radar unit according to claim 1, wherein: The downconversion circuit includes a first downconversion mixer in the first receiver path and a second downconversion mixer in the second receiver path, wherein the clock signal from the reference clock circuit produces a frequency shift input to a baseband mixer coupled to an output of one of the two receiver paths in the downconversion circuit.

4. The radar unit according to claim 1, wherein: The frequency shifter circuit includes a phase rotator operably coupled to a down-conversion mixer located in one of the two receiver paths, and the frequency shifter circuit is configured to apply a frequency shift to the reference signal, which frequency compensates for the frequency shift after processing by the phase rotator and the bandpass filter and the ADC circuit.

5. The radar unit according to claim 1, wherein: The DSP is configured to: summing the digital versions of the filtered baseband signals provided by the ADC circuit as real and complex parts and performing a range Fast Fourier Transform (FFT) across both positive and negative frequency spectra; and The real and complex parts are separated using the FFT, which applies: where S(f)=F(s(t)) is the frequency and time representation of the received digital signal, () * represents the complex conjugate, and S(f + )、S(f - ) represent the positive and negative spectrum parts respectively.

6. An integrated circuit for a radar unit, characterized in that The radar unit includes a radar transceiver, and the radar transceiver includes: a reference local oscillator (Ref LO) configured to generate a reference signal; a reference clock circuit configured to output a clock signal; at least two transmitter paths, wherein each transmitter path includes a phase rotator configured to receive the reference signal and the clock signal, wherein the clock signal rotates a phase of the reference signal in at least one transmitter path, wherein a first frequency modulated continuous wave (FMCW) radar signal in a first transmitter path is shifted by a sampling frequency Fs relative to a second FMCW radar signal in a second transmitter path, wherein each of the at least two transmitter paths includes a power amplifier configured to amplify the respective first FMCW radar signal or the second FMCW signal and apply the respective amplified FMCW radar signal to the respective first transmit antenna or the second transmit antenna; at least one receiver coupled to a receive antenna that receives reflections of the FMCW radar signal, the receive antenna coupled to two receiver paths; a downconversion circuit coupled to the Ref LO and configured to downconvert the reflected FMCW radar signal in each receiver path to a baseband signal, wherein the downconversion circuit includes a frequency shifter circuit located in a first receiver path and configured to apply a frequency shift to a received reflection of the FMCW radar signal or one path of the reference signal, the reference signal downconverting the reflection of the FMCW radar signal in the first receiver path to a baseband signal of a frequency band represented by the FMCW transmit signal after the frequency shift applied by the phase rotator; a bandpass filter configured to filter the baseband signal in each receiver path, wherein a first bandpass filter in a first receiver path is configured to pass the baseband signal, and a second bandpass filter in a second receiver path is configured to pass the baseband signal of the second transmitter path after an applied frequency shift has been removed; and analog-to-digital converter (ADC) circuitry configured to convert the filtered baseband signal in each receiver path into digital form; and a digital signal processor (DSP) operatively coupled to the at least one receiver and configured to process the digital form of the filtered baseband signal and to determine an angular dimension of a received reflected radar signal in response to the digital form.

7. A method for receiving two transmitted radar signals in a radar unit, characterized in that The method comprises: A reference signal is generated by a reference local oscillator Ref LO; Rotating the phase of the reference signal in one of at least two transmitter paths providing a frequency modulated continuous wave (FMCW) radar signal by a clock signal applied to a phase rotator, wherein the signal of the first transmitter is shifted by a sampling frequency Fs relative to the second transmitter; amplifying and transmitting a first FMCW radar signal from the first transmitting antenna and a second FMCW radar signal from the second transmitting antenna; receiving reflected radar signals and amplifying the received reflected radar signals in two receiver paths; Down-converting the received amplified reflected radar signal to a baseband signal in the two receiver paths using the reference signal, comprising: applying a frequency shift to any received reflection of the FMCW radar signal in a first receiver path; or applying a frequency shift to the reference signal that downconverts the reflection of the FMCW radar signal in the first receiver path; wherein the frequency shift shifts the reflection of the FMCW radar signal to a baseband signal of a frequency band represented by the FMCW transmit signal after compensating for a frequency shift applied by the phase rotator caused by the sampling frequency Fs applied in one of the transmit FMCW radar signals; performing bandpass filtering on the baseband signals, wherein a first downconverted baseband signal of the received reflected radar signal is passed by a first bandpass filter in the first receiver path and a second downconverted baseband signal of the received reflected radar signal is passed by a second bandpass filter in the second receiver path; Performing analog-to-digital conversion on the down-converted, filtered baseband signal; and The down-converted, filtered baseband signal in digital form is digitally processed.

8. The method according to claim 7, characterized in that Further comprising configuring the frequency shift applied by the phase rotator in one of the at least two transmitter paths and the frequency shift applied by the frequency shifter circuit in the first receiver path to be each greater than or equal to an ADC sampling frequency δ applied to one of the at least two transmitter paths. f ≥f s , and the received beat signal associated with the one of the at least two transmitter paths falls outside a passband frequency of one receiver path.

9. The method according to claim 7, characterized in that The downconversion circuit includes a first downconversion mixer in the first receiver path and a second downconversion mixer in the second receiver path, and the method further includes generating a frequency shift input to a baseband mixer by the clock signal from the reference clock circuit, the baseband mixer being coupled to an output of one of the two receiver paths in the downconversion circuit.

10. The method according to claim 7, characterized in that The method further comprises: operatively coupling a phase rotator of the frequency shifter circuit to a downconversion mixer in one of the two receiver paths; and A frequency shift is applied to the reference signal by the frequency shifter circuit, which is frequency compensated for the frequency shift after processing by the phase rotator and the bandpass filter and ADC circuit.

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