Three-phase single-stage isolated matrix converter circuit and driving method

Abstract

This application discloses a three-phase single-stage isolated matrix converter circuit and its driving method. The three-phase single-stage isolated matrix converter circuit generates primary-side control quantities and secondary-side control quantities based on grid-side electrical parameters and DC-side electrical parameters. It then generates primary-side drive signals and secondary-side drive signals based on the primary-side control quantities and secondary-side control quantities, respectively. The primary-side drive signal drives the three-phase matrix converter, and the secondary-side drive signal controls the transformer current through the full-bridge inverter to be at its minimum value within the power frequency cycle. This minimizes the transformer current throughout the entire power frequency cycle, thereby reducing device conduction losses and transformer losses.

Description

Technical Field

[0001] This application relates to the field of power electronics technology, specifically to a three-phase single-stage isolated matrix converter circuit and its driving method. Background Technology

[0002] Three-phase single-stage isolated AC / DC matrix converters are easy to achieve high power levels and high power density. However, due to the large number of control variables and complex operating modes, simplified control schemes are usually adopted, such as fixing the primary-side modulation ratio and controlling the secondary-side phase shift angle, or fixing the secondary-side phase shift angle and controlling the primary-side modulation ratio. These control schemes result in larger transformer currents, which in turn lead to higher device conduction losses and transformer losses.

[0003] However, the transformer current optimization control method, which simultaneously controls the primary-side modulation ratio and the primary-secondary-side phase shift angle, is only applicable to a specific grid voltage phase angle. Since the grid voltage phase angle changes periodically over time, it cannot achieve transformer current optimization throughout the entire power frequency cycle. Summary of the Invention

[0004] This application provides a three-phase single-stage isolated matrix converter circuit and driving method to alleviate the above-mentioned technical problems.

[0005] In a first aspect, this application provides a three-phase single-stage isolated matrix converter circuit, which includes a sampling module, a digital controller, and a three-phase matrix converter, a first inductor, a transformer, and a full-bridge inverter connected in sequence. The three-phase matrix converter is used to connect to a three-phase AC power grid. The sampling module is connected to the three-phase AC power grid and the full-bridge inverter to obtain the grid-side electrical parameters of the three-phase AC power grid and the DC-side electrical parameters of the full-bridge inverter. The digital controller is connected to the sampling module, the three-phase matrix converter, and the full-bridge inverter to generate primary-side control quantities and secondary-side control quantities based on the grid-side electrical parameters and the DC-side electrical parameters, and to generate primary-side drive signals and secondary-side drive signals based on the primary-side control quantities and secondary-side control quantities, respectively. The primary-side drive signals are used to drive the three-phase matrix converter, and the secondary-side drive signals are used to control the transformer current to be at its minimum value within the power frequency cycle through the full-bridge inverter.

[0006] Secondly, this application also provides a driving method applied to a three-phase single-stage isolated matrix converter circuit. The three-phase single-stage isolated matrix converter circuit includes a three-phase matrix converter, a first inductor, a transformer, and a full-bridge inverter connected in sequence. The three-phase matrix converter is used to connect to a three-phase AC power grid. The driving method includes: acquiring the grid-side electrical parameters of the three-phase AC power grid and the DC-side electrical parameters of the full-bridge inverter; generating primary-side control quantities and secondary-side control quantities based on the grid-side electrical parameters and the DC-side electrical parameters; and generating primary-side drive signals and secondary-side drive signals based on the primary-side control quantities and secondary-side control quantities, respectively. The primary-side drive signal is used to drive the three-phase matrix converter, and the secondary-side drive signal is used to control the transformer current to be at its minimum value within the power frequency cycle through the full-bridge inverter.

[0007] The three-phase single-stage isolated matrix converter circuit and driving method provided in this application generate primary-side control quantities and secondary-side control quantities based on grid-side electrical parameters and DC-side electrical parameters. Primary-side drive signals and secondary-side drive signals are generated based on the primary-side control quantities and secondary-side control quantities, respectively. The primary-side drive signal drives the three-phase matrix converter, and the secondary-side drive signal controls the transformer current through the full-bridge inverter to be at its minimum value within the power frequency cycle. This can minimize the transformer current within the entire power frequency cycle, thereby reducing the conduction losses of the devices and the losses of the transformer. Attached Figure Description

[0008] The technical solution and other beneficial effects of this application will become apparent from the following detailed description of specific embodiments in conjunction with the accompanying drawings.

[0009] Figure 1 A circuit diagram of a three-phase single-stage isolated matrix converter circuit provided in an embodiment of this application.

[0010] Figure 2 The schematic diagram of the sampling module and digital controller provided in the embodiments of this application is shown.

[0011] Figure 3 The circuit schematic of the second computing unit provided in the embodiments of this application.

[0012] Figure 4 This is a schematic diagram of the first modulation sequence of the above three-phase single-stage isolated matrix converter circuit.

[0013] Figure 5 This is a schematic diagram of the second modulation sequence for the three-phase single-stage isolated matrix converter circuit described above.

[0014] Figure 6 This is a flowchart illustrating the driving method provided in an embodiment of this application. Detailed Implementation

[0015] The technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of this application, and not all of them. All other embodiments obtained by those skilled in the art based on the embodiments of this application without creative effort are within the scope of protection of this application.

[0016] Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of technical features indicated. Features thus defined as "first" or "second" may explicitly or implicitly include one or more of the stated features. In the description of this invention, "a plurality of" means two or more unless otherwise explicitly specified.

[0017] With the development of the new energy industry, more and more DC loads such as energy storage and electric vehicles are being connected to the AC power grid, and bidirectional isolated AC / DC converters are widely used in the AC / DC conversion field. The mainstream topology of bidirectional isolated AC / DC converters usually adopts a two-stage topology scheme of non-isolated AC / DC plus isolated DC / DC, which is beneficial for widening the voltage gain range and achieving input-output electrical isolation. However, the two-stage topology has lower efficiency, more components, and higher cost.

[0018] The mainstream topology of single-stage isolated AC / DC converters is the single-phase dual active bridge converter (DAB). It is flexible in control and has fewer components. However, single-phase converters are suitable for lower power levels, and the DC side usually requires a large filter capacitor, which limits the improvement of the converter's power density.

[0019] Three-phase single-stage isolated AC / DC matrix converters are easy to achieve high power levels and high power density. However, due to the large number of control variables and complex operating modes, simplified control schemes are usually adopted, such as fixing the primary-side modulation ratio and controlling the secondary-side phase shift angle, or fixing the secondary-side phase shift angle and controlling the primary-side modulation ratio. These control schemes result in larger transformer currents, which in turn lead to higher device conduction losses and transformer losses.

[0020] The transformer current optimization control method, which simultaneously controls the primary-side modulation ratio and the primary-secondary-side phase shift angle, is only applicable to specific grid voltage phase angles. Since the grid voltage phase angle changes periodically over time, it cannot achieve transformer current optimization throughout the entire power frequency cycle.

[0021] This embodiment provides a three-phase single-stage isolated matrix converter circuit, such as... Figure 1 and Figure 2As shown, the three-phase single-stage isolated matrix converter circuit includes a sampling module 30, a digital controller 40, and a three-phase matrix converter 10, a first inductor Lk, a transformer T, and a full-bridge inverter 20 connected in sequence. The three-phase matrix converter 10 is used to connect to a three-phase AC power grid 110. The sampling module 30 is connected to the three-phase AC power grid 110 and the full-bridge inverter 20. The sampling module 30 is used to obtain the grid-side electrical parameters of the three-phase AC power grid 110 and the DC-side electrical parameters of the full-bridge inverter 20. The digital controller 40 is connected to the sampling module 30, the three-phase matrix converter 10, and the full-bridge inverter 20. The digital controller 40 is used to generate primary-side control quantities and secondary-side control quantities based on grid-side electrical parameters and DC-side electrical parameters, and to generate primary-side drive signal PWM1 and secondary-side drive signal PWM2 based on the primary-side control quantities and secondary-side control quantities, respectively. The primary-side drive signal PWM1 is used to drive the three-phase matrix converter 10, and the secondary-side drive signal PWM2 is used to control the current of the transformer T to be at its minimum value within the power frequency cycle through the full-bridge inverter 20.

[0022] It is understood that the three-phase single-stage isolated matrix converter circuit provided in this embodiment generates primary-side control and secondary-side control quantities based on grid-side electrical parameters and DC-side electrical parameters, and generates primary-side drive signal PWM1 and secondary-side drive signal PWM2 based on the primary-side control and secondary-side control quantities, respectively. The primary-side drive signal PWM1 drives the three-phase matrix converter 10, and the secondary-side drive signal PWM2 controls the current of transformer T to be at its minimum value within the power frequency cycle through the full-bridge inverter 20. This can minimize the current of transformer T within the entire power frequency cycle, thereby reducing the conduction loss of the devices and the loss of transformer T.

[0023] It should be noted that the grid-side electrical parameters include grid-side voltage and grid-side current, while the DC-side electrical parameters include DC-side voltage Vdc and DC-side current Idc. Specifically, the grid-side voltage is the phase voltage (ua, ub, uc / ua~uc) of the three-phase AC power grid 110, and the grid-side current is the phase current (ia, ib, ic / ia~ic) of the three-phase AC power grid 110, where a, b, and c represent the three phases of the three-phase AC power grid 110.

[0024] In some embodiments, such as Figure 1 As shown, the three-phase matrix converter 10 includes three parallel bridge arms, each of which includes two bidirectional switches, such as Sa1 and Sa2, Sb1 and Sb2, and Sc1 and Sc2. The primary-side drive signal PWM1 drives the three-phase matrix converter 10 by driving these bidirectional switches.

[0025] In some embodiments, such as Figure 1As shown, the full-bridge inverter 20 includes two parallel bridge arms, each of which includes two switching transistors, such as S1 and S2, and S3 and S4. The secondary-side drive signal PWM2 drives the full-bridge inverter 20 by driving these switching transistors.

[0026] It should be noted that since both the three-phase matrix converter 10 and the full-bridge inverter 20 use fully controlled switching transistors, on the one hand, they can provide the degree of freedom required for current optimization control, and on the other hand, they can realize bidirectional energy transmission, allowing the DC side to be connected to a DC source or load.

[0027] In some embodiments, such as Figure 1 As shown, the three-phase single-stage isolated matrix converter circuit also includes a filter 120 and a first capacitor Cdc. The filter 120 is connected between the three-phase AC power grid 110 and the three-phase matrix converter 10. The first capacitor Cdc is connected to the DC side of the full-bridge inverter 20.

[0028] In some embodiments, such as Figure 2 As shown, the sampling module 30 includes a grid-side sampling unit 31 and a DC-side sampling unit 32. The grid-side sampling unit 31 is connected to the three-phase AC power grid 110 and the digital controller 40, and is used to obtain the grid-side voltage and grid-side current. The DC-side sampling unit 32 is connected to the full-bridge inverter 20 and the digital controller 40, and is used to obtain the DC-side voltage Vdc and DC-side current Idc.

[0029] It should be noted that this embodiment only uses sampled values ​​such as grid-side voltage, grid-side current, DC-side voltage Vdc, and DC-side current Idc to minimize the current of transformer T, which can reduce the number of sampled values ​​and simplify control.

[0030] In some embodiments, such as Figure 2 As shown, the digital controller 40 includes a computing module 41, a primary-side control module 42, and a secondary-side control module 43. The computing module 41 is connected to the grid-side sampling unit 31 and the DC-side sampling unit 32. The primary-side control module 42 is connected to the computing module 41 and the three-phase matrix converter 10. The secondary-side control module 43 is connected to the computing module 41 and the full-bridge inverter 20.

[0031] It should be noted that the calculation module 41 is used to obtain the phase voltage amplitude Vm, grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ based on the grid-side voltage, grid-side current, DC-side voltage Vdc, and DC-side current Idc. The primary-side control module 42 is used to generate the primary-side drive signal PWM1 based on the grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ. The secondary-side control module 43 is used to generate the secondary-side drive signal PWM2 based on the phase voltage amplitude Vm, grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ.

[0032] In some embodiments, such as Figure 2 As shown, the secondary control module 43 includes a current optimization unit 431 and a secondary pulse width modulation unit 432. The current optimization unit 431 is connected to the calculation module 41, and the secondary pulse width modulation unit 432 is connected to the current optimization unit 431 and the full-bridge inverter 20.

[0033] It should be noted that the current optimization unit 431 is used to generate the secondary-side control quantity based on the phase voltage amplitude Vm, the grid voltage phase angle θg, the modulation ratio m, and the modulation phase angle θ. The secondary-side control quantity is the phase shift angle of the secondary-side voltage us of transformer T relative to the primary-side voltage up of transformer T. The secondary-side pulse width modulation unit 432 is used to generate the secondary-side drive signal PWM2 based on the secondary-side control quantity and the sawtooth carrier 44.

[0034] In some embodiments, such as Figure 2 As shown, the primary-side control module 42 includes a time calculation unit 421 and a primary-side pulse width modulation unit 422. The time calculation unit 421 is connected to the calculation module 41, and the primary-side pulse width modulation unit 422 is connected to the time calculation unit 421 and the three-phase matrix converter 10.

[0035] It should be noted that the time calculation unit 421 is used to generate multiple voltage action times for the primary-side voltage up based on the grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ. These multiple voltage action times include the first action time for the maximum line voltage, the second action time for the second largest line voltage, and the third action time for zero voltage. The primary-side pulse width modulation unit 422 is used to generate the primary-side drive signal PWM1 based on the first action time, the second action time, the third action time, and the sawtooth carrier wave 44. The primary-side control quantities include the first action time, the second action time, and the third action time.

[0036] In some embodiments, such as Figure 2 As shown, the calculation module 41 includes a first calculation unit 411 and a second calculation unit 412. The first calculation unit 411 is connected to the grid-side sampling unit 31, and the second calculation unit 412 is connected to the grid-side sampling unit 31 and the DC-side sampling unit 32.

[0037] It should be noted that the first calculation unit 411 is used to generate the phase voltage amplitude Vm and the grid voltage phase angle θg based on the grid-side voltage. The second calculation unit 412 is used to obtain the modulation ratio m and the modulation phase angle θ based on the grid-side voltage, grid-side current, DC-side voltage Vdc, and DC-side current Idc.

[0038] In some embodiments, such as Figure 3As shown, the second computing unit 412 includes a first computing subunit 4121, a second computing subunit 4122, and a third computing subunit 4123. The first computing subunit 4121 is connected to the DC-side sampling unit 32, the second computing subunit 4122 is connected to the grid-side sampling unit 31, and the third computing subunit 4123 is connected to the first computing subunit 4121 and the second computing subunit 4122.

[0039] It should be noted that the first calculation subunit 4121 is used to generate the reference current, i.e., id_ref, based on the DC-side voltage Vdc and the DC-side current Idc. The second calculation subunit 4122 is used to generate the grid phase, i.e., ωt, the d-axis current component, i.e., id, and the q-axis current component, i.e., iq, based on the grid-side voltage, i.e., ua~uc, and the grid-side current, i.e., ia~ic. The third calculation subunit 4123 is used to generate the modulation ratio m and the modulation phase angle θ based on the reference current, i.e., id_ref, the grid phase, i.e., ωt, the d-axis current component, i.e., id, and the q-axis current component, iq.

[0040] In some embodiments, such as Figure 3 As shown, the first calculation subunit 4121 includes a multiplier CF, a first subtractor JF1, a first PI controller PI_P, a second subtractor JF2, a second PI controller PI_V, and a switching unit K1. The multiplier CF is connected to the DC-side sampling unit 32, the first subtractor JF1 is connected to the multiplier CF, the first PI controller PI_P is connected to the first subtractor JF1, the second subtractor JF2 is connected to the DC-side sampling unit 32, the second PI controller PI_V is connected to the second subtractor JF2, and the switching unit K1 is connected to the first PI controller PI_P and the second PI controller PI_V.

[0041] It should be noted that the multiplier CF is used to obtain the real-time power based on the product of the DC-side voltage Vdc and the DC-side current Idc. The first subtractor JF1 is used to obtain the power error based on the difference between the reference power (Pref) and the real-time power. The first PI controller PI_P is used to obtain the reference current based on the power error. The second subtractor JF2 is used to obtain the voltage error based on the difference between the reference voltage (Vref) and the DC-side voltage Vdc. The second PI controller PI_V is used to obtain the reference current based on the voltage error. The switching unit K1 is used to control the output of the first PI controller PI_P or the output of the second PI controller PI_V to connect to the second calculation subunit 4122 according to the DC-side load conditions of the full-bridge inverter 20.

[0042] In some embodiments, such as Figure 3As shown, the second calculation subunit 4122 includes a phase-locked loop (PLL) and a conversion unit abc / dq. The PLL is connected to the grid-side sampling unit 31, and the conversion unit abc / dq is connected to both the grid-side sampling unit 31 and the PLL.

[0043] It should be noted that the phase-locked loop (PLL) is used to obtain the grid phase based on the grid-side voltage. The conversion unit abc / dq is used to obtain the d-axis current component and the q-axis current component based on the grid-side current and the grid-side current.

[0044] In some embodiments, such as Figure 3 As shown, the third calculation subunit 4123 includes a third subtractor JF3, a third PI controller PI_I3, a fourth subtractor JF4, a fourth PI controller PI_I4, and an amplitude angle calculator FJ1. The third subtractor JF3 is connected to the switching unit K1, the third PI controller PI_I3 is connected to the third subtractor JF3, the fourth PI controller PI_I4 is connected to the fourth subtractor JF4, and the amplitude angle calculator FJ1 is connected to the third PI controller PI_I3 and the fourth PI controller PI_I4.

[0045] It should be noted that the third subtractor JF3 is used to obtain the d-axis current error based on the difference between the reference current and the d-axis current component. The third PI controller PI_I3 is used to obtain the d-axis adjustment amount, xd, based on the d-axis current error. The fourth subtractor JF4 is used to obtain the q-axis current error based on the difference between the zero reference current and the q-axis current component. The fourth PI controller PI_I4 is used to obtain the q-axis adjustment amount, xq, based on the q-axis current error. The amplitude-angle calculator FJ1 is used to obtain the modulation ratio m and the modulation phase angle θ based on the d-axis and q-axis adjustment amounts.

[0046] The multiplier CF, the first subtractor JF1, and the first PI controller PI_P are used to implement power control, while the second subtractor JF2 and the second PI controller PI_V are used to implement voltage control. The second calculation subunit 4122 and the third calculation subunit 4123 are used to implement power factor correction control.

[0047] Figure 3 The implementation steps of the second computing unit 412 shown are as follows:

[0048] Step 1: Based on the load conditions on the DC side, select the control mode as power control or voltage control through the switching unit K1. The error between the sampled and calculated DC side power / voltage and the corresponding reference power (Pref) or reference voltage (Vref) is used to calculate the reference current (id_ref) through the corresponding first PI controller PI_P or second PI controller PI_V.

[0049] Step 2: The grid phase ωt is calculated by ua~uc through a phase-locked loop (PLL), and id and iq are obtained by rotating coordinate transformation of ia~ic.

[0050] Step 3: Calculate the errors of id and iq with id_ref and 0 respectively, and obtain the adjustment values ​​xd and xq in the dq coordinate system through their respective third PI controller PI_I3 and fourth PI controller PI_I4.

[0051] Step 4: Calculate the corresponding modulation ratio m and modulation phase angle θ from xd and xq.

[0052] In some embodiments, such as Figure 4 and Figure 5 As shown, in each switching cycle Ts, the maximum line voltage, the second largest line voltage, and the zero voltage are symmetrically distributed about the midpoint of the switching cycle Ts. The phase of the secondary voltage us is delayed by φ×Ts÷(2π) relative to the phase of the primary voltage up.

[0053] It should be noted that at any given time, one switch in each of the upper (Sa1 / Sb1 / Sc1) and lower (Sa2 / Sb2 / Sc2) arms of the primary side is turned on, making the primary voltage up the corresponding line voltage or zero voltage. On the secondary side, the upper and lower switches of each arm conduct complementaryly, with an inward phase shift angle φ of 180° between the arms, resulting in a secondary voltage us that is a square wave voltage with a duty cycle of 50% and an amplitude of Vdc. Under the influence of the primary voltage up and the secondary voltage us, the current flowing through the first inductor Lk, i.e., the current of the transformer T, periodically charges and discharges, realizing power transfer between the grid side (AC side) and the DC side.

[0054] Figure 4 shows the modulation sequence of the above-mentioned three-phase single-stage isolated AC / DC matrix converter within one switching period Ts. Figure 4 In this equation, ux and uy represent the maximum and second-largest line voltages of the power grid at the current moment, respectively. For example, when the phase voltages satisfy ua>0>uc>ub, ux=ua-ub and uy=ua-uc. Based on the relationship between the phase voltages, one power frequency cycle can be divided into 12 modes. The operating conditions in different modes are the same, so only one mode needs to be analyzed, and the other modes are analyzed similarly. Tx, Ty, and T0 represent the duration of line voltage ux, uy, and zero voltage within one switching cycle Ts, respectively. Within the power frequency cycle, ux, uy, Tx, Ty, and T0 vary with the phase angle θg of the power grid voltage, as shown in Equation 1 below:

[0055]

[0056] Where ux is the maximum line voltage, uy is the second largest line voltage, Vm is the phase voltage amplitude, θg is the grid voltage phase angle, θ is the modulation phase angle, m is the modulation ratio, Ts is the switching period, Tx is the first action time, Ty is the second action time, and T0 is the third action time.

[0057] Within the power frequency cycle, θg changes continuously with time. 0≤m≤1, -π / 6≤θg+θ≤0.

[0058] In the driving method proposed in this application, the modulation ratio m and modulation phase angle θ are generated by closed-loop regulation to achieve DC-side power or voltage control, while simultaneously achieving grid-side power factor correction control. Based on this, the secondary-side control quantity (phase shift angle φ) serves as the optimized control quantity, used to regulate the current of transformer T during operation, and is calculated by the current optimization algorithm in the current optimization unit 431. The current optimization algorithm is derived with the goal of minimizing the current of transformer T. Within the power frequency cycle, the control quantities, namely the modulation ratio m, modulation phase angle θ, and phase shift angle φ, are all adjusted in real time according to the grid voltage phase angle θg, ensuring that the current flowing through transformer T is at its minimum at any given moment within the entire power frequency cycle.

[0059] The following is a detailed process for deriving the current optimization algorithm:

[0060] (1) Fourier decomposition:

[0061] The primary voltage up and the secondary voltage us are represented by Fourier series, as shown in Equation 2. Here, ωs is the angular frequency corresponding to the switching period Ts, and an is the Fourier coefficient of the primary voltage up, given by Equation 3.

[0062]

[0063]

[0064] Furthermore, the current ip flowing through the first inductor Lk can also be expressed in Fourier series form, as shown in Equation 4, where Lk is the inductance value of the first inductor Lk and N is the turns ratio of the transformer T.

[0065]

[0066] The square of the amplitude of the nth harmonic of the current is Ipn 2 As shown in Equation 5:

[0067]

[0068] The average power P during the switching cycle Ts is shown in Equation 6:

[0069]

[0070] Where n represents the harmonic order of the decomposed current, such as Ip1 when n=1 representing the fundamental component of the current, Ip2 when n=2 representing the second harmonic component of the current, Ip3 when n=3 representing the third harmonic component of the current, and so on. ws is the angular frequency, ws=2πfs, and fs is the switching frequency.

[0071] (2) Fundamental wave approximation

[0072] In ip and P, the fundamental component (n=1) is the main component, therefore the square of the current amplitude is Ip. 2 Both P and P are approximated as their fundamental components, as shown in Equation 7:

[0073]

[0074] When n=1, a1 is obtained from equation (3), as shown in equation (8):

[0075]

[0076] (3) Current optimization algorithm

[0077] The goal of this current optimization algorithm is to minimize the required current amplitude Ip under the given transmission power (Po), which is equivalent to Ip 2 Minimum. Based on the Lagrange multiplier method, the Lagrange function L(a1,φ,λ) is constructed as shown in Equation 9:

[0078]

[0079] Here, λ is a variable introduced by the constructor and has no practical meaning.

[0080] Setting the partial derivatives of the Lagrange function L(a1,φ,λ) with respect to a1, φ, and λ to 0, the resulting system of equations can be solved to obtain the inner phase shift angle φ, as shown in Equation 10:

[0081]

[0082] Equation 10 is the phase shift angle calculation formula used in the current optimization algorithm. When the transmission power, i.e., Po, is constant, for any grid voltage phase angle θg within the power frequency cycle, the inner phase shift angle φ is adjusted according to the relationship shown in Equation 10 with the control quantity, i.e., the modulation ratio m and the modulation phase angle θ, so that the current flowing through the transformer T is minimized, thereby significantly improving the efficiency of the three-phase single-stage isolated AC / DC matrix converter.

[0083] Compared to Figure 4 , Figure 5 The modulation sequence was modified, where ux, uy, Tx, Ty, and T0 are the same as in the previous scheme, only the segmented sequence of the primary voltage up within the switching period Ts was changed. Figure 5The corresponding current optimization algorithm provides the inner phase shift angle φ, as shown in Equation 11:

[0084]

[0085] Where e1 is an intermediate quantity, as shown in Equation 12:

[0086]

[0087] Based on the above analysis and derivation, the flowchart of the current optimization control method is as follows: Figure 6 As shown, the specific steps are as follows:

[0088] Step 1: The grid-side sampling unit 31 and the DC-side sampling unit 32 sample the grid phase voltage (ua~uc), phase current (ia~ic), DC-side voltage (Vdc), and DC-side current (Idc), and after processing, transmit them to the digital controller 40.

[0089] Step 2: In the digital controller 40, the phase voltage amplitude Vm and the current grid voltage phase angle θg are calculated from the grid phase voltage ua~uc.

[0090] Step 3: In the digital controller 40, the modulation ratio m and modulation phase angle θ are calculated by the power / voltage control loop and the PFC control loop.

[0091] Step 4: In the digital controller 40, the corresponding voltage application times, namely Tx, Ty, and T0, are calculated using the formula 1 above.

[0092] Step 5: In the digital controller 40, the inner phase shift angle φ between the primary voltage up and the secondary voltage us is calculated using the above formula 10.

[0093] Step 6: In the digital controller 40, the primary-side control quantity, the secondary-side control quantity, and the sawtooth carrier 44 are modulated by PWM to generate the primary-side drive signal PWM1 and the secondary-side drive signal PWM2.

[0094] In summary, this application utilizes the fundamental frequency analysis method to establish a mathematical model of a three-phase single-stage isolated AC / DC matrix converter, obtaining the relationship between the converter's transmission power, current, and control quantities. By analyzing the conditions under which the current of transformer T reaches its minimum value under given operating conditions, the relationship between control quantities that must be satisfied at any moment within the power frequency cycle when the current of transformer T reaches its minimum value is derived. A closed-loop control scheme is designed so that the current optimization algorithm maintains the converter's current at its minimum value based on real-time calculations of the current operating state, thereby achieving the current optimization objective and improving converter efficiency.

[0095] Therefore, this application can achieve the following effects:

[0096] The current optimization goal of minimizing the current of transformer T can be achieved throughout the entire power frequency cycle.

[0097] The primary and secondary sides are coordinated to achieve basic control objectives such as power / voltage / power factor correction (PFC), while the secondary side control is used to achieve current optimization objectives.

[0098] The proposed current optimization algorithm provides the relationship between the secondary-side control quantity and the primary-side control quantity, without limiting the control strategy used to generate the primary-side control quantity, thus exhibiting high adaptability.

[0099] In some embodiments, this embodiment also provides a driving method applied to the above-described three-phase single-stage isolated matrix converter circuit. The driving method includes: acquiring the grid-side electrical parameters of the three-phase AC power grid 110 and the DC-side electrical parameters of the full-bridge inverter 20; generating primary-side control signals and secondary-side control signals based on the grid-side and DC-side electrical parameters; and generating primary-side drive signal PWM1 and secondary-side drive signal PWM2 based on the primary-side and secondary-side control signals, respectively. Primary-side drive signal PWM1 is used to drive the three-phase matrix converter 10, and secondary-side drive signal PWM2 is used to control the current of transformer T through the full-bridge inverter 20 to be at its minimum value within the power frequency cycle.

[0100] It is understood that the driving method provided in this embodiment generates primary-side control and secondary-side control quantities based on grid-side electrical parameters and DC-side electrical parameters, and generates primary-side drive signal PWM1 and secondary-side drive signal PWM2 based on the primary-side control and secondary-side control quantities, respectively. The primary-side drive signal PWM1 drives the three-phase matrix converter 10, and the secondary-side drive signal PWM2 controls the current of transformer T to be at its minimum value throughout the power frequency cycle through the full-bridge inverter 20. This can minimize the current of transformer T throughout the entire power frequency cycle, thereby reducing the conduction loss of the device and the loss of transformer T.

[0101] In some embodiments, the grid-side electrical parameters include grid-side voltage and grid-side current, and the DC-side electrical parameters include DC-side voltage Vdc and DC-side current Idc. The driving method further includes: obtaining the phase voltage amplitude Vm, grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ based on the grid-side voltage, grid-side current, DC-side voltage Vdc, and DC-side current Idc. Generating a primary-side driving signal PWM1 based on the grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ. Generating a secondary-side driving signal PWM2 based on the phase voltage amplitude Vm, grid voltage phase angle θg, modulation ratio m, and modulation phase angle θ.

[0102] In some embodiments, the driving method further includes: generating a secondary-side control quantity based on the phase voltage amplitude Vm, the grid voltage phase angle θg, the modulation ratio m, and the modulation phase angle θ, wherein the secondary-side control quantity is the phase shift angle of the secondary-side voltage us of transformer T relative to the primary-side voltage up of transformer T. A secondary-side driving signal PWM2 is generated based on the secondary-side control quantity and the sawtooth carrier wave 44.

[0103] In some embodiments, the driving method further includes: generating multiple voltage action times for the primary-side voltage up based on the grid voltage phase angle θg, the modulation ratio m, and the modulation phase angle θ, wherein the multiple voltage action times include a first action time for the maximum line voltage, a second action time for the second largest line voltage, and a third action time for zero voltage. A primary-side driving signal PWM1 is generated based on the first action time, the second action time, the third action time, and the sawtooth carrier wave 44.

[0104] In some embodiments, the driving method further includes: configuring the maximum line voltage, the second largest line voltage, and the zero voltage to be symmetrically distributed about the midpoint of the switching period Ts in each switching cycle Ts; configuring the phase of the secondary voltage us to be delayed by φ×Ts÷(2π) relative to the phase of the primary voltage up.

[0105] In some embodiments:

[0106]

[0107] Where ux is the maximum line voltage, uy is the second largest line voltage, Vm is the phase voltage amplitude, θg is the grid voltage phase angle, θ is the modulation phase angle, m is the modulation ratio, Ts is the switching period, Tx is the first action time, Ty is the second action time, and T0 is the third action time.

[0108] In some embodiments, the driving method further includes: generating a phase voltage amplitude Vm and a grid voltage phase angle θg based on the grid-side voltage. The modulation ratio m and modulation phase angle θ are obtained based on the grid-side voltage, grid-side current, DC-side voltage Vdc, and DC-side current Idc.

[0109] In some embodiments, the driving method further includes: generating a reference current based on the DC-side voltage Vdc and the DC-side current Idc; generating a grid phase, a d-axis current component, and a q-axis current component based on the grid-side voltage and the grid-side current; and generating a modulation ratio m and a modulation phase angle θ based on the reference current, the grid phase, the d-axis current component, and the q-axis current component.

[0110] In some embodiments, the driving method further includes: obtaining real-time power based on the product of DC-side voltage Vdc and DC-side current Idc; obtaining power error based on the difference between reference power and real-time power; obtaining reference current based on power error; obtaining voltage error based on the difference between reference voltage and DC-side voltage Vdc; obtaining reference current based on voltage error; and connecting the output of the first PI controller PI_P or the output of the second PI controller PI_V to the second calculation subunit 4122 according to the DC-side load conditions of the full-bridge inverter 20.

[0111] In some embodiments, the driving method further includes: obtaining the grid phase based on the grid-side voltage; and obtaining the d-axis current component and the q-axis current component based on the grid-side current and the grid-side current.

[0112] In some embodiments, the driving method further includes: obtaining a d-axis current error based on the difference between a reference current and a d-axis current component; obtaining a d-axis adjustment amount based on the d-axis current error; obtaining a q-axis current error based on the difference between a zero reference current and a q-axis current component; obtaining a q-axis adjustment amount based on the q-axis current error; and obtaining a modulation ratio m and a modulation phase angle θ based on the d-axis adjustment amount and the q-axis adjustment amount.

[0113] In the above embodiments, the descriptions of each embodiment have different focuses. For parts not described in detail in a certain embodiment, please refer to the relevant descriptions in other embodiments.

[0114] This document uses specific examples to illustrate the principles and implementation methods of this application. The descriptions of the above embodiments are only for the purpose of helping to understand the technical solutions and core ideas of this application. Those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. These modifications or substitutions will not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of this application.

Claims

1. A three-phase single-stage isolated matrix converter circuit, characterized by, Comprise: a three-phase matrix converter, a first inductor, a transformer and a full-bridge inverter connected in sequence, the three-phase matrix converter being used for connecting a three-phase alternating current power grid; a sampling module connected with the three-phase alternating current power grid and the full-bridge inverter, and used for obtaining grid-side electrical parameters of the three-phase alternating current power grid and direct-current side electrical parameters of the full-bridge inverter; a digital controller connected with the sampling module, the three-phase matrix converter and the full-bridge inverter, and used for generating primary side control quantity and secondary side control quantity based on the grid-side electrical parameters and the direct-current side electrical parameters, and generating primary side driving signal and secondary side driving signal based on the primary side control quantity and the secondary side control quantity respectively, the primary side driving signal being used for driving the three-phase matrix converter; the digital controller comprises a secondary side control module, the secondary side control module comprises a current optimization unit, the current optimization unit is used for generating the secondary side control quantity according to phase voltage amplitude, grid voltage phase angle, modulation ratio and modulation phase angle, the secondary side control quantity is a phase shift angle of the secondary side voltage of the transformer relative to the primary side voltage of the transformer, the secondary side driving signal is generated based on the phase shift angle, and is used for controlling the current of the transformer to be minimum in a power frequency period through the full-bridge inverter.

2. The three-phase single-stage isolated matrix converter circuit according to claim 1, characterized in that, The grid-side electrical parameters comprise grid-side voltage and grid-side current, the direct-current side electrical parameters comprise direct-current side voltage and direct-current side current, and the sampling module comprises: a grid-side sampling unit connected with the three-phase alternating current power grid and the digital controller, and used for obtaining the grid-side voltage and the grid-side current; a direct-current side sampling unit connected with the full-bridge inverter and the digital controller, and used for obtaining the direct-current side voltage and the direct-current side current.

3. The three-phase single-stage isolated matrix converter circuit according to claim 2, characterized in that, The digital controller further comprises: a calculation module connected with the grid-side sampling unit and the direct-current side sampling unit, and used for obtaining the phase voltage amplitude, the grid voltage phase angle, the modulation ratio and the modulation phase angle based on the grid-side voltage, the grid-side current, the direct-current side voltage and the direct-current side current; a primary side control module connected with the calculation module and the three-phase matrix converter, and used for generating the primary side driving signal based on the grid voltage phase angle, the modulation ratio and the modulation phase angle; the secondary side control module is connected with the calculation module and the full-bridge inverter, and used for generating the secondary side driving signal based on the phase voltage amplitude, the grid voltage phase angle, the modulation ratio and the modulation phase angle.

4. The three-phase single-stage isolated matrix converter circuit according to claim 3, characterized in that, The secondary side control module further comprises: a secondary side pulse width modulation unit connected with the current optimization unit and the full-bridge inverter, and used for generating the secondary side driving signal according to the secondary side control quantity and sawtooth carrier.

5. The three-phase single-stage isolated matrix converter circuit according to claim 4, characterized in that, The primary side control module comprises: a time calculation unit connected with the calculation module, and used for generating multiple voltage action times of the primary side voltage according to the grid voltage phase angle, the modulation ratio and the modulation phase angle, the multiple voltage action times comprising a first action time of maximum line voltage, a second action time of sub-maximum line voltage and a third action time of zero voltage; The primary side pulse width modulation unit is connected with the time calculation unit and the three-phase matrix converter, and is configured to generate the primary side driving signal according to the first action time, the second action time, the third action time and the sawtooth carrier.

6. The three-phase single-stage isolated matrix converter circuit of claim 5, wherein, In each switching cycle, the maximum line voltage, the second maximum line voltage and the zero voltage are symmetrically distributed about the midpoint of the switching cycle. The phase of the secondary side voltage is delayed by φ×Ts÷(2π) with respect to the phase of the primary side voltage.

7. The three-phase single-stage isolated matrix converter circuit according to any one of claims 3-6, characterized in that, The calculation module comprises: The first calculation unit is connected with the grid side sampling unit, and is configured to generate the phase voltage amplitude and the grid voltage phase angle according to the grid voltage. The second calculation unit is connected with the grid side sampling unit and the DC side sampling unit, and is configured to obtain the modulation ratio and the modulation phase angle based on the grid voltage, the grid current, the DC voltage and the DC current.

8. The three-phase single-stage isolated matrix converter circuit of claim 7, wherein, The second calculation unit comprises: The first calculation subunit is connected with the DC side sampling unit, and is configured to generate a reference current according to the DC voltage and the DC current. The second calculation subunit is connected with the grid side sampling unit, and is configured to generate a grid phase, a d-axis current component and a q-axis current component according to the grid voltage and the grid current. The third calculation subunit is connected with the first calculation subunit and the second calculation subunit, and is configured to generate the modulation ratio and the modulation phase angle according to the reference current, the grid phase, the d-axis current component and the q-axis current component.

9. The three-phase single-stage isolated matrix converter circuit of claim 8, wherein, The first calculation subunit comprises: The multiplier is connected with the DC side sampling unit, and is configured to obtain real-time power according to the product of the DC voltage and the DC current. The first subtractor is connected with the multiplier, and is configured to obtain a power error according to the difference between a reference power and the real-time power. The first PI controller is connected with the first subtractor, and is configured to obtain the reference current according to the power error. The second subtractor is connected with the DC side sampling unit, and is configured to obtain a voltage error according to the difference between a reference voltage and the DC voltage. The second PI controller is connected with the second subtractor, and is configured to obtain the reference current according to the voltage error. The switching unit is connected with the first PI controller and the second PI controller, and is configured to control the output end of the first PI controller or the output end of the second PI controller to be connected to the second calculation subunit according to the DC side load working condition of the full-bridge inverter.

10. The three-phase single-stage isolated matrix converter circuit of claim 9, wherein, The second calculation subunit comprises: The phase-locked loop is connected with the grid side sampling unit, and is configured to obtain the grid phase according to the grid voltage. The conversion unit is connected with the grid side sampling unit and the phase-locked loop, and is configured to obtain the d-axis current component and the q-axis current component according to the grid current and the grid phase.

11. The three-phase single-stage isolated matrix converter circuit of claim 10, wherein, The third calculation subunit comprises: The third subtractor is connected with the switching unit, and is configured to obtain a d-axis current error according to the difference between the reference current and the d-axis current component. a third PI controller, connected with the third subtractor, for obtaining a d-axis regulation quantity according to the d-axis current error; a fourth subtractor, for obtaining a q-axis current error according to a difference between a zero reference current and the q-axis current component; a fourth PI controller, connected with the fourth subtractor, for obtaining a q-axis regulation quantity according to the q-axis current error; an amplitude and angle calculator, connected with the third PI controller and the fourth PI controller, for obtaining the modulation ratio and the modulation phase angle according to the d-axis regulation quantity and the q-axis regulation quantity.

12. The three-phase single-stage isolated matrix converter circuit according to any one of claims 1-6, characterized in that, The three-phase single-stage isolation type matrix conversion circuit further comprises: a filter, connected between the three-phase alternating current power grid and the three-phase matrix converter; a first capacitor, connected to a direct current side of the full-bridge inverter.

13. A driving method, comprising: The driving method is applied to a three-phase single-stage isolation type matrix conversion circuit, the three-phase single-stage isolation type matrix conversion circuit comprising a three-phase matrix converter, a first inductor, a transformer and a full-bridge inverter connected in sequence, the three-phase matrix converter being used for connecting a three-phase alternating current power grid, the driving method comprising: obtaining grid-side electrical parameters of the three-phase alternating current power grid and direct current side electrical parameters of the full-bridge inverter; generating a primary side control quantity and a secondary side control quantity based on the grid-side electrical parameters and the direct current side electrical parameters, and generating a primary side driving signal and a secondary side driving signal based on the primary side control quantity and the secondary side control quantity respectively, the primary side driving signal being used for driving the three-phase matrix converter; wherein the step of generating the secondary side control quantity comprises: generating the secondary side control quantity according to a phase voltage amplitude, a power grid voltage phase angle, a modulation ratio and a modulation phase angle, the secondary side control quantity being a phase shift angle of a secondary side voltage of the transformer relative to a primary side voltage of the transformer, the secondary side driving signal being generated based on the phase shift angle and being used for controlling a current of the transformer to be a minimum value in a power frequency period through the full-bridge inverter.

14. The driving method according to claim 13, wherein The grid-side electrical parameters comprise a grid-side voltage and a grid-side current, and the direct current side electrical parameters comprise a direct current side voltage and a direct current side current; the driving method further comprises: obtaining the phase voltage amplitude, the power grid voltage phase angle, the modulation ratio and the modulation phase angle based on the grid-side voltage, the grid-side current, the direct current side voltage and the direct current side current; generating the primary side driving signal based on the power grid voltage phase angle, the modulation ratio and the modulation phase angle; generating the secondary side driving signal based on the phase voltage amplitude, the power grid voltage phase angle, the modulation ratio and the modulation phase angle.

15. The driving method according to claim 14, wherein The driving method further comprises: generating the secondary side driving signal according to the secondary side control quantity and a sawtooth carrier.

16. The driving method according to claim 15, wherein The driving method further comprises: generating a plurality of voltage action times of the primary side voltage according to the power grid voltage phase angle, the modulation ratio and the modulation phase angle, the plurality of voltage action times comprising a first action time of a maximum line voltage, a second action time of a sub-maximum line voltage and a third action time of a zero voltage; generating the primary side driving signal according to the first action time, the second action time, the third action time and the sawtooth carrier.

17. The driving method according to claim 16, wherein The driving method further comprises: The maximum line voltage, the second maximum line voltage and the zero voltage are symmetrically distributed about a midpoint of the switching period in each switching period; A phase of the secondary side voltage is delayed by φ×Ts÷(2π) with respect to a phase of the primary side voltage.

18. The driving method according to any one of claims 14 to 17, wherein The driving method further comprises: generating the phase voltage amplitude and the grid voltage phase angle according to the grid side voltage; obtaining the modulation ratio and the modulation phase angle based on the grid side voltage, the grid side current, the DC side voltage and the DC side current.

19. The driving method according to claim 18, wherein The driving method further comprises: generating a reference current according to the DC side voltage and the DC side current; generating a grid phase, a d-axis current component and a q-axis current component according to the grid side voltage and the grid side current; generating the modulation ratio and the modulation phase angle according to the reference current, the grid phase, the d-axis current component and the q-axis current component.

20. The driving method according to claim 19, wherein The driving method further comprises: obtaining real-time power according to a product of the DC side voltage and the DC side current; obtaining a power error according to a difference between a reference power and the real-time power; obtaining the reference current according to the power error; obtaining a voltage error according to a difference between a reference voltage and the DC side voltage; obtaining the reference current according to the voltage error; controlling an output terminal of the first PI controller or an output terminal of the second PI controller to be connected to the second calculation subunit according to a DC side load condition of the full-bridge inverter.

21. The driving method according to claim 20, wherein The driving method further comprises: obtaining the grid phase according to the grid side voltage; obtaining the d-axis current component and the q-axis current component according to the grid side current and the grid side current.

22. The driving method according to claim 21, wherein The driving method further comprises: obtaining a d-axis current error according to a difference between the reference current and the d-axis current component; obtaining a d-axis adjustment amount according to the d-axis current error; obtaining a q-axis current error according to a difference between a zero reference current and the q-axis current component; obtaining a q-axis adjustment amount according to the q-axis current error; obtaining the modulation ratio and the modulation phase angle according to the d-axis adjustment amount and the q-axis adjustment amount.

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