Broadband electromagnetic dipole antenna based on X-shaped polarization conversion metasurface

Abstract

The invention discloses a broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface, and belongs to the technical field of electromagnetic dipole antennas. In order to solve the problems that an existing metamaterial wave absorber RCS is narrow in reduction bandwidth, an RCS antenna of a PCM chessboard structure is concentrated at the frequency band below 20 GHz and the like, the electromagnetic dipole antenna combines a 1 * 8 electromagnetic dipole array with the PCM chessboard structure, and the RCS is minimized while high gain and wide frequency band are kept; by using the X-shaped PCM unit and the mirror image unit array thereof, cross polarization conversion of reflected waves is realized in a working frequency band, a 180-degree phase difference is generated, the PCM disperses back scattering energy to a surrounding area, and normal reflection is reduced, so that the RCS of the antenna is effectively reduced, and the RCS is reduced by more than 10 dB in 74.1% of bandwidth (19.07-41.52 GHz). The electromagnetic dipole antenna disclosed by the invention has the characteristics of high gain, wide bandwidth and low RCS (radar cross section), has a wide application prospect in the fields of high-frequency wireless communication and radar stealth, and shows huge potential of future development of a stealth technology.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of electromagnetic dipole antennas, and particularly relates to a broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface. BACKGROUND

[0002] With the rapid development of stealth technology, the stealth performance of antennas as key components of stealth platforms becomes crucial. In particular, for planar antenna arrays, due to their large size and significant physical aperture, effectively reducing the radar cross section (RCS) has become a major challenge. In recent years, metasurfaces (MS) have shown great potential in the field of antenna stealth due to their design flexibility. Metasurfaces can flexibly manipulate multiple degrees of freedom of electromagnetic fields, and by reasonably designing metasurface structures, not only can the RCS of the antenna be effectively reduced, but also its good electromagnetic performance can be maintained.

[0003] Currently, metamaterial wave absorbers and reflective phase cancellation are two widely used RCS reduction techniques. However, metamaterial wave absorbers have the problem of narrow RCS reduction bandwidth, and may affect the radiation efficiency of the antenna. In addition, metamaterial wave absorbers usually need to integrate lumped resistors or varactor diode elements, resulting in difficult processing and high manufacturing cost. In contrast, reflective phase cancellation metasurfaces achieve wideband RCS reduction through chessboard structures, such as artificial magnetic conductors (AMC), electromagnetic band gaps (EBG), and polarization conversion metasurfaces (PCM), with significant advantages. Among them, PCM, with its unique polarization conversion characteristics, can achieve wideband RCS reduction, is flexible in design and has significant effects, and has become an important choice for antenna stealth technology.

[0004] Although there have been numerous studies proposing PCM-based chessboard structures to design low-RCS antennas, most designs focus on frequencies below 20 GHz, with relatively limited research on frequencies above 20 GHz. In addition, designing an antenna that simultaneously achieves high-frequency operating capability, wideband performance, and wide-angle low-RCS characteristics remains a key issue. SUMMARY

[0005] In view of the above problems, the present application aims to provide a broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface, which achieves cross-polarization conversion of reflected waves within the operating frequency band by utilizing an array of X-shaped PCM units and their mirror units, producing a 180° phase difference. The PCM disperses the backscattered energy to the surrounding area while reducing the normal reflection, thereby effectively reducing the RCS of the antenna.

[0006] To achieve the above purpose, the technical solution adopted by the present application is as follows: The wideband electromagnetic dipole antenna based on X-shaped polarization conversion metasurface comprises a first dielectric substrate and a second dielectric substrate, and an intermediate layer is coated between the first dielectric substrate and the second dielectric substrate; a radiation dipole antenna array and a scattering PCM checkerboard array are arranged on the top of the first dielectric substrate, the radiation dipole antenna array is located at the center of the scattering PCM checkerboard array, and the scattering PCM checkerboard array comprises a plurality of scattering PCM checkerboard array units arranged in a mirror-symmetrical checkerboard array, and the scattering PCM checkerboard array unit is in an X shape. The second dielectric substrate is staggered left and right with the first dielectric substrate, and a T-shaped power divider feed network is arranged on the bottom of the second dielectric substrate, and a coupling slot array corresponding to the radiation dipole antenna array is arranged on the intermediate layer.

[0007] Further, the radiation dipole antenna array comprises 1x8 radiation dipole antenna array units, the scattering PCM checkerboard array comprises 10x16 scattering PCM checkerboard array units, and 2x12 scattering PCM checkerboard array units are removed from the center region of the scattering PCM checkerboard array for arranging the radiation dipole antenna array.

[0008] Further, above and below the center region of the scattering PCM checkerboard array, each 4x4 scattering PCM checkerboard sub-array with the same handedness is formed every 4x4 scattering PCM checkerboard array units, the handedness of the adjacent two 4x4 scattering PCM checkerboard sub-arrays is opposite, and the handedness of the 4x4 scattering PCM checkerboard sub-arrays corresponding to each other above and below the center region is also opposite. There are four scattering PCM checkerboard array units on the left and right sides of the center region respectively: on the left and right sides of the center region, the two scattering PCM checkerboard array units close to the upper side have the same handedness as the 4x4 scattering PCM checkerboard sub-arrays corresponding to the left and right edges of the upper side respectively; the two scattering PCM checkerboard array units close to the lower side have the same handedness as the 4x4 scattering PCM checkerboard sub-arrays corresponding to the left and right edges of the lower side respectively.

[0009] Further, eight metallized vias corresponding to the radiation dipole antenna array are arranged on the first dielectric substrate.

[0010] Further, the coupling slot array comprises 1x8 coupling slots, and the coupling slot array corresponds to the eight metallized vias.

[0011] Further, the T-shaped power divider feed network is a 1:8 T-shaped power divider feed network.

[0012] Further, the 1:8 T-type power divider feeding network comprises a first-stage T-type power divider, a second-stage T-type power divider, a third-stage T-type power divider and eight feed lines, the input end of the first-stage T-type power divider is used as a feeding port, two outputs of the first-stage power divider are divided into four outputs through two second-stage T-type power dividers, and the four outputs of the second-stage T-type power dividers are connected with the eight feed lines through four third-stage T-type power dividers respectively.

[0013] Further, the 1:8 T-type power divider feeding network adopts 50Ω coaxial feeding.

[0014] Further, the electromagnetic dipole antenna has a size of 86.25 mm × 56.18 mm × 1.892 mm.

[0015] The present application has the following advantages: 1. The scattering PCM checkerboard array unit in the present application adopts an X-shaped structure, and high-efficiency polarization conversion can be realized by adjusting only two parameters (the radius and width of the circular arc), and the polarization conversion rate (PCR) is more than 90% within a bandwidth of 77.4% (19-43 GHz). Compared with the existing PCM unit structure which mainly adopts a multilayer dielectric substrate, even a single-layer basic structure often contains an air layer to enhance the polarization conversion rate, the structure is simpler and has low manufacturing complexity; compared with an anisotropic artificial magnetic conductor (AMC) which needs two different structure designs, the X-shaped scattering PCM checkerboard array unit proposed in the present application is configured in a mirror-symmetrical checkerboard array mode, which significantly simplifies the design process.

[0016] 2. The electromagnetic dipole antenna in the present application adopts a simple double-layer dielectric substrate configuration, two radiation patches are used as electric dipoles, and eight metalized through holes and a central coupling slot are used as magnetic dipoles. The 1×8 antenna array realizes an impedance matching bandwidth of about 34.56% (-10 dB) (22.60-32.04 GHz), and the maximum realized gain is 14.84 dBi. In addition, the relative bandwidth of the RCS reduction value exceeding 10 dB is about 74.1% (19.07-41.52 GHz).

[0017] 3. The electromagnetic dipole antenna in the present application has the advantages of high gain, wide bandwidth and low RCS, has high application value in the fields of high-frequency wireless communication and radar stealth, and has broad development prospects. BRIEF DESCRIPTION OF DRAWINGS

[0018] Figure 1 It is a schematic diagram of the overall structure of the electromagnetic dipole antenna in the present application.

[0019] Figure 2 It is a schematic diagram of the positional relationship of the first dielectric substrate, the second dielectric substrate and the intermediate layer in the present application.

[0020] Figure 3 Top view of the first dielectric substrate in the present application.

[0021] Figure 4 Top view of the middle layer in the present application.

[0022] Figure 5 Bottom view of the second dielectric substrate in the present application.

[0023] Figure 6 Structure diagram of the T-shaped power divider feeding network in the present application.

[0024] Figure 7 Top view of the two different-rotation scattering PCM chessboard array units in the present application.

[0025] Figure 8 Performance detection results of the same polarization and cross polarization of the X-shaped scattering PCM chessboard array unit in the present application.

[0026] Figure 9 Structure diagram of the 1x2 antenna array in the simulation experiment of the present application.

[0027] Figure 10 Current distribution of the 1x2 antenna array radiation patch and feed line structure under the working frequency of 27 GHz in the simulation experiment of the present application.

[0028] Figure 11 S parameter and normalized radiation pattern of the 1x2 antenna array in the simulation experiment of the present application.

[0029] Figure 12 Physical diagram of the electromagnetic dipole antenna in the simulation experiment of the present application.

[0030] Figure 13 Performance comparison results of the antenna integrated with PCM, the antenna without PCM and the electromagnetic dipole antenna prototype in the present application in the simulation experiment of the present application.

[0031] Figure 14 Simulation results of the far field radiation and scattering characteristics of the antenna integrated with PCM in the simulation experiment of the present application.

[0032] Figure 15 Comparison results of the single station RCS and RCS reduction of the electromagnetic dipole antenna and the equal-area metal plate in the simulation experiment of the present application.

[0033] Wherein, 1, the first dielectric substrate; 101, the radiating dipole antenna array; 102, the scattering PCM checkerboard array; 103, the metallized via; 2, the second dielectric substrate; 201, the T-shaped power divider feed network; 2010, the feed port; 2011, the first-stage T-shaped power divider; 2012, the second-stage T-shaped power divider; 2013, the third-stage T-shaped power divider; 2014, the feeder; 3, the intermediate layer; 301, the coupling slot array; 4, the connecting hole; 5, the aluminum block. DETAILED DESCRIPTION

[0034] In order for those skilled in the art to better understand the technical solutions of the present application, the technical solutions of the present application will be further described below in combination with the drawings and examples.

[0035] Referring to the drawings Figure 1 And 2 As shown in the drawings, the broadband electromagnetic dipole antenna based on X-shaped polarization conversion metasurface includes a first dielectric substrate 1, a second dielectric substrate 2 and an intermediate layer 3. The first dielectric substrate 1 and the second dielectric substrate 2 both use Rogers 5880 material with loss, and the first dielectric substrate 1 and the second dielectric substrate 2 are staggered left and right. The intermediate layer 3 is a copper layer coated on the top of the second dielectric substrate 2 and the bottom of the staggered part of the first dielectric substrate 1 and the second dielectric substrate 2. The first dielectric substrate 1, the second dielectric substrate 2 and the intermediate layer 3 are all provided with corresponding connecting holes 4, which are used to connect the first dielectric substrate 1, the second dielectric substrate 2 and the intermediate layer 3 by bolts to form a whole. In addition, the top of the second dielectric substrate 2 corresponding to the staggered part of the first dielectric substrate 1 is also provided with an aluminum block 5 for fixing the coaxial connector on the corresponding intermediate layer 3. The aluminum block 5, the intermediate layer 3 and the second dielectric substrate 2 are also provided with corresponding connecting holes 4 for connecting the aluminum block 5 and the second dielectric substrate 2 by bolts.

[0036] Specifically, the top of the first dielectric substrate 1 is provided with a radiating dipole antenna array 101 and a scattering PCM checkerboard array 102, and the radiating dipole antenna array 101 is located at the center of the scattering PCM checkerboard array 102. The radiating dipole antenna array 101 includes 1x8 radiating dipole antenna array units, and the scattering PCM checkerboard array 102 includes 10x16 scattering PCM checkerboard array units. In order to reduce the mutual coupling between the radiation and scattering parts, 2x12 scattering PCM checkerboard array units are removed at the center of the scattering PCM checkerboard array 102 for arranging the radiating dipole antenna array 101, as shown in the drawings. Figure 3 Eight metallized vias 103 corresponding to the radiating dipole antenna array 101 are also provided on the first dielectric substrate 1.

[0037] The first dielectric substrate 1 and the second dielectric substrate 2 are offset left and right, that is, the left edge of the second dielectric substrate 2 is shifted to the left relative to the left edge of the first dielectric substrate 1, and the right edge of the first dielectric substrate 1 is shifted to the right relative to the right edge of the second dielectric substrate 2, as shown in the attached figure. Figure 2 As shown. Correspondingly, the intermediate layer 3 includes two parts: the left part corresponds to the second dielectric substrate 2 and is directly coated on the top of the second dielectric substrate 2; the right part corresponds to the right offset portion of the first dielectric substrate 1 and the second dielectric substrate 2 and is directly coated on the bottom of the right offset portion of the first dielectric substrate 1 and the second dielectric substrate 2.

[0038] A coupling slot array 301 is provided on the left side of the intermediate layer. The coupling slot array 301 includes 1×8 coupling slots, which correspond to eight metallized vias 103 on the first dielectric substrate 1, forming a magnetic dipole. This is used to couple and feed the signal from the bottom of the second dielectric substrate 2 into the electromagnetic dipole antenna, as shown in the attached diagram. Figure 4 As shown. The radiating dipole antenna array 101 on the first dielectric substrate 1 serves as an electrode, and the electrode and magnetic pole cooperate to form a magnetoelectric dipole.

[0039] As attached Figure 5 As shown, a T-type power divider feed network 201 is provided at the bottom of the second dielectric substrate 2. The T-type power divider feed network 201 is a 1:8 T-type power divider feed network, and the output end of the 1:8 T-type power divider feed network corresponds to the coupling slot array 301. Specifically, as shown in the attached diagram... Figure 6 As shown, the 1:8 T-type power divider feed network includes a first-stage T-type power divider 2011, a second-stage T-type power divider 2012, a third-stage T-type power divider 2013, and eight feed lines 2014. The input terminal of the first-stage T-type power divider 2011 serves as the feed port 2010. The two output terminals of the first-stage power divider 2011 are divided into four output terminals through two second-stage T-type power dividers 2012. The four output terminals of the second-stage T-type power dividers 2012 are then connected to the eight feed lines 2014 through four third-stage T-type power dividers 2013. The output terminals of the eight feed lines 2014 correspond to the coupling slot array 301. This network uses a 50Ω coaxial feed and, combined with a stepped impedance matching structure and chamfered edge design, effectively improves the impedance matching performance of the antenna.

[0040] In the present application, the scattering PCM checkerboard array unit is X-shaped, and 10*16 scattering PCM checkerboard array units are arranged in a mirror-symmetrical checkerboard array. Above and below the central region, every 4*4 scattering PCM checkerboard array units form a 4*4 scattering PCM checkerboard sub-array with the same chirality, the adjacent two 4*4 scattering PCM checkerboard sub-arrays have opposite chirality, and the 4*4 scattering PCM checkerboard sub-arrays corresponding to each other above and below the central region also have opposite chirality; on the left and right sides of the central region, there are four scattering PCM checkerboard array units respectively: on the left side of the central region, the two scattering PCM checkerboard array units close to the top have the same chirality as the 4*4 scattering PCM checkerboard sub-array at the top left edge, and the two scattering PCM checkerboard array units close to the bottom have the same chirality as the 4*4 scattering PCM checkerboard sub-array at the bottom left edge; on the right side of the central region, the two scattering PCM checkerboard array units close to the top have the same chirality as the 4*4 scattering PCM checkerboard sub-array at the top right edge, and the two scattering PCM checkerboard array units close to the bottom have the same chirality as the 4*4 scattering PCM checkerboard sub-array at the bottom right edge.

[0041] The specific principle of the electromagnetic dipole antenna in the present application is as follows: the radiating dipole antenna array 101 acts as an electric dipole, when high-frequency current reaches the radiating dipole antenna array 101, an equivalent oscillating current source is formed at the edge of the radiating dipole antenna array, generating a vertical electric field radiation component, and the radiation wave propagates outward, a plurality of radiating dipole antenna array units form an array, realizing high-gain directional radiation. The metallized via hole 103 and the coupling slot array 301 jointly constitute a magnetic dipole, wherein the coupling slot array 301 is located on the middle layer 3, according to the Babinet equivalent principle, the coupling slot can be equivalent to a magnetic current source, generating a horizontal equivalent magnetic current radiation; the metallized via hole 103 penetrates through the first dielectric substrate 1, forming a vertical current channel, and cooperates with the slot array 301 to form an equivalent vertical current loop, jointly generating a magnetic dipole radiation. In the process of feeding, the radio frequency signal enters the T-type power divider feeding network 201 from the feeding port 2010, the first-stage T-type power divider 2011, the second-stage T-type power divider 2012 and the third-stage T-type power divider 2013 equally and in phase distribute energy to each output port, each channel of signal excites the coupling slot array 301 to generate a magnetic dipole radiation through the second dielectric substrate 2, at the same time, the energy is transmitted upward to the radiating dipole antenna array 101 through electromagnetic coupling through the metallized via hole 103, exciting the electric dipole to radiate, finally the electric dipole and the magnetic dipole work together, the radiation field in the front of the antenna is in phase superposition, enhancing the radiation, and the back radiation is in opposite phase and is cancelled, thereby realizing the characteristics of one-way radiation with wideband, high gain and low back lobe.

[0042] The energy transmission path of the electromagnetic dipole antenna in the application is: feed port 2010→T-shaped power divider feed network 201→second dielectric substrate 2→coupling slot array 301→metalized via hole 103→radiating dipole antenna array 101→free space.

[0043] As shown in the accompanying drawings Figure 7 are top views of two different rotation direction scattering PCM checkerboard array units, where (a) is left-handed, and (b) is right-handed.

[0044] The linearly polarized wave of the incident wave along the x direction is: ; In the formula, is the linearly polarized wave of the incident wave along the x direction; is the electric field amplitude of the incident field, represents the electric field along the x direction or the x polarized wave, is a plane wave propagation factor, k is a wave vector, j represents an imaginary unit, z is a propagation direction, w is an angular frequency, and t is a time variable.

[0045] The linearly polarized wave can be decomposed into the superposition of left-handed LCP and right-handed RCP circularly polarized waves: ; According to the geometric phase theory, when the scattering PCM checkerboard array unit rotates an angle θ, the LCP is converted to RCP to obtain a phase 2θ, and the RCP is converted to LCP to obtain a phase -2θ.

[0046] The reflected electric field is: where r represents a complex reflection coefficient, and the reflected wave is still a linearly polarized wave, but the polarization direction is rotated by 2θ.

[0047] Therefore, when the scattering PCM checkerboard array unit is rotated by 90°, .

[0048] The linearly polarized wave is subjected to polarization rotation, and the x polarized incident wave is converted to a y polarized reflected wave, thereby realizing polarization rotation. The scattering PCM checkerboard array unit shown in (a) of Figure 7 is rotated by 90° to obtain the scattering PCM checkerboard array unit shown in (b) of Figure 7 , and at the same time, a reflection phase shift of 180° is introduced according to the geometric phase relationship Δφ=2θ. As Figure 7As shown by the red arrow, when an electromagnetic wave is incident perpendicularly onto a checkerboard structure formed by alternating arrangement of these two types of scattering PCM checkerboard array elements, adjacent scattering PCM checkerboard array elements generate two equal-amplitude, out-of-phase reflected waves. The two waves interfere destructively, thereby reducing the radar cross section (RCS).

[0049] After CST simulation optimization, the geometric parameters of the electromagnetic dipole antenna in this embodiment are shown in Table 1 below (unit: mm). The final dimensions determined by parameter scanning optimization are 86.25 mm × 56.18 mm × 1.892 mm (length × width × thickness).

[0050] The performance test results of the homopolarization and crosspolarization of the X-type scattering PCM checkerboard array elements are attached. Figure 8 The above refers to the reflection amplitude and polarization conversion rate (PCR) of the X-type scattering PCM checkerboard array elements with co-polarization and cross-polarization; and the phase and phase difference of the X-type scattering PCM checkerboard array elements with co-polarization and cross-polarization. (See attached...) Figure 8 As can be seen, the scattering PCM checkerboard array elements exhibit polarization conversion capability within 77.4% of the bandwidth (19-43 GHz), with a polarization conversion ratio (PCR) exceeding 90%, indicating its efficient and stable polarization conversion characteristics. The phase difference remains almost within ±90° within the 15-45 GHz range.

[0051] Table 1 Optimized geometric parameters of the electromagnetic dipole antenna in Example 1

[0052] Simulation verification: To compare the characteristics with the electromagnetic dipole antenna (1×8 antenna array) proposed in this invention, an electromagnetic simulation of a 1×2 antenna array was first performed. A structural diagram of the 1×2 antenna array is attached. Figure 9 As shown, (a) is a three-dimensional structural schematic diagram of a 1×2 radiating dipole antenna array element, (b) is a top view of a 1×2 radiating dipole antenna array element, (c) is a top view of 1×2 coupling slots, and (d) is the feed network of a 1×2 antenna array. (See attached diagram.) Figure 10 The current distribution of the 1×2 antenna array structure, radiating patch and feeder structure at the operating frequency of 27 GHz is shown, where (a) corresponds to the operating state with port 1 fed and port 2 connected to a 50Ω load; (b) corresponds to the configuration with both ports fed simultaneously.

[0053] Current vector analysis shows that the current density of the radiating element along the x-axis exhibits strong in-phase superposition, while the currents along the y-axis cancel each other out due to their opposite phase. This optimized current distribution gives the antenna excellent in-phase radiation characteristics.

[0054] Appendix Figure 11 The S-parameters and normalized radiation pattern of a 1×2 antenna array are shown, where (a) represents the S-parameters, (b) is the E-plane of the 27 GHz normalized radiation pattern, and (c) is the H-plane of the 27 GHz normalized radiation pattern. During the simulation, one port was excited, and the other port was connected to a 50Ω matched load. (See attached diagram.) Figure 11 As shown in (a), the reflection coefficient The -10 dB impedance bandwidth is 30.5% (25–34 GHz). From The curve reveals two resonant points within the operating frequency band: the first resonant point is located at 27.6 GHz, generated by the mixed-mode excitation of the magnetoelectric dipole; the second resonant point is located at 31.33 GHz, originating from the impedance matching effect of the stepped microstrip line in the feed network. The two resonant points work together to effectively broaden the bandwidth. To suppress mutual coupling between array elements, the element spacing is designed as follows: mm. From As can be seen from the curve, the isolation remains below -18.9 dB within the -10 dB impedance bandwidth.

[0055] Appendix Figure 11 Figures (b) and (c) show the normalized radiation pattern at 27 GHz. Due to the main radiated current along the x-axis and the mutual cancellation of the y-axis currents, the single-port feed configuration results in increased beamwidth in the low elevation region of the E-plane. Simultaneously, the normalized cross-polarization level of the E-plane remains below -20 dB.

[0056] To verify the performance of the electromagnetic dipole antenna proposed in this invention, a prototype was fabricated using PCB manufacturing technology and tested in this simulation experiment. Figure 12 As shown in the figure, (a) is a top view of the machined object; (b) is a bottom view of the machined object; (c) is the far-field radiation pattern in the microwave anechoic chamber; and (d) is the RCS measurement environment. The first dielectric substrate 1 and the second dielectric substrate 2 are fixed with nylon screws and excited by a 2.92-KFD3G coaxial connector (rated frequency: 40 GHz) connected via a microstrip feed network. The S-parameters of the prototype were measured using a network analyzer, and the radiation pattern, antenna gain, and RCS were obtained using a compact field antenna test system.

[0057] To evaluate the radiation performance and impedance matching characteristics of the electromagnetic dipole antenna proposed in this invention, simulation analysis was performed. For a direct comparison of performance differences, analyses were conducted on antennas with integrated PCM, antennas without PCM, and fabricated prototypes. The comparison results are attached. Figure 13 As shown, (a) is the S-parameter curve, and (b) is the measured gain characteristic.

[0058] As attachedFigure 13 As shown in (a) of FIG. 18, the influence of the PCM chessboard structure on the S parameter can be ignored, and both configurations achieve a -10 dB impedance bandwidth of 34.56% (22.60-32.04 GHz). Compared with the S parameter of the 1x2 antenna array shown in FIG. 17, the -10 dB impedance bandwidth of the 1x8 array is shifted downward by about 2 GHz. This phenomenon is mainly due to the increase in the effective electrical length of the current transmission path when using a 1:8 T-type power divider for feeding, which causes the operating frequency to systematically drift to low frequencies. Figure 11

[0059] The measured data show that the third resonance point shifts to high frequencies, and the fourth resonance point shifts to low frequencies, and the two eventually merge into a single resonance point, resulting in a decrease in the number of observed resonance points by one compared to the simulation. The measured impedance bandwidth is 33.55% (22.65-31.78 GHz), with a deviation of less than 1.5% from the simulation results. This high consistency strongly verifies the wideband characteristics of the electromagnetic dipole antenna in the present application.

[0060] As shown in (b) of FIG. 18, the introduction of polarization conversion metasurfaces produces additional resonance modes, effectively expanding the radiation aperture, and the gain of the antenna loaded with PCM is improved compared to the antenna without PCM. Specifically, the measured peak gain of the antenna loaded with PCM at 30 GHz reaches 14.84 dBi, which is 1.01 dBi higher than that without PCM (13.83 dBi). The measured maximum gain at 28 GHz is 14.57 dBi. The simulation and measurement results are very close, which confirms that the design achieves both wideband and high gain. Figure 13 FIG. 19 shows the simulation results of the far-field radiation and scattering characteristics of the antenna integrated with PCM in the present simulation experiment, where (a) is the three-dimensional gain pattern, i.e., the far-field radiation pattern, under coaxial feeding, and (b) is the bistatic RCS pattern. As shown in (a) of FIG. 19, under the specified frequency band or operating conditions, the radiation gain always maintains stable high gain characteristics (peak gain > 13.7 dBi). In contrast, (b) of FIG. 19 shows that the incident wave in the normal direction is scattered in all directions, which reduces the energy in the main radiation direction, thereby effectively reducing the RCS.

[0061] Figure 14 To verify the RCS reduction effect of the X-shaped PCM structure proposed in the present application, the electromagnetic dipole antenna proposed in the present application is simulated and experimentally measured, and compared with an equal-area metal plate, and the results are shown in FIG. 20. Figure 14 Figure 14 As shown in (a) of FIG. 20, under the specified frequency band or operating conditions, the radiation gain always maintains stable high gain characteristics (peak gain > 13.7 dBi). In contrast, (b) of FIG. 20 shows that the incident wave in the normal direction is scattered in all directions, which reduces the energy in the main radiation direction, thereby effectively reducing the RCS.

[0062] To verify the RCS reduction effect of the X-shaped PCM structure proposed in the present application, the electromagnetic dipole antenna proposed in the present application is simulated and experimentally measured, and compared with an equal-area metal plate, and the results are shown in FIG. 20. Figure 15 ​​​The measured results show that the RCS values are all below -10 dBsm in the range of 17.5-40 GHz.

[0063] From the above Figure 15 As can be seen from (b), compared with the metal reference body, the designed antenna achieves significant RCS reduction in the range of 16-45 GHz. Among them, the 10-dB RCS reduction bandwidth reaches 74.1% (19.07-41.52 GHz), which confirms the ultra-wideband RCS suppression capability.

[0064] To more accurately reflect the excellent performance of the antenna, Table 2 below gives the performance comparison of the electromagnetic dipole antenna in the present application and the existing array antennas. The RCS reduction relative bandwidths of documents [1], [2], [3], [6] are better than this work, but document [2] only considers the scattering characteristics and does not involve the radiation performance, and the RCS reduction values of documents [4], [5] and [7] are all lower than 10 dB. The PCM units of documents [3] and [4] all use single-layer structure design, and although document [4] achieves about 10 dB RCS reduction in multiple frequency bands, its gain is low. Although the antenna of document [7] has excellent gain performance, it has defects such as large radiation aperture and high antenna profile. The comparative analysis shows that the electromagnetic dipole antenna proposed in the present application not only has excellent radiation performance in the Ka band (26.5-40 GHz), but also has outstanding scattering characteristics, showing good comprehensive performance.

[0065] Table 2 Performance comparison results of electromagnetic dipole antenna in the present application and existing low RCS antennas

[0066] Among them, documents [1]-[7] are: Document [1]: M. Li, Z. Huang, X.-X. Yang, et al. Broadband In-Band and Out-of-Band RCS Reduction of Antenna Based on Multiple Mechanisms[J]. IEEE Antennas and Wireless Propagation Letters, 2024, 23(12): 4673-4677. Document [2]: Soliman S A, El-Desouki E M, El-Nady S M, et al. Broadband low RCS based on polarization-dependent artificial magnetic conductor metasurface [J]. IEEE Access, 2023, 11: 53176-53184. Document [3]: Y. Liu, K. Li, Y. Jia, et al. Wideband RCS Reduction of a Slot Array Antenna Using Polarization Conversion Metasurfaces [J]. IEEE Transactions on Antennas and Propagation, 2016, 64(1): 326-331. Document [4]: T. Hong, S. Wang, Z. Liu, et al. RCS Reduction and Gain Enhancement for the Circularly Polarized Array by Polarization Conversion Metasurface Coating [J]. IEEE Antennas and Wireless Propagation Letters, 2019, 18(1): 167-171. Document [5]: H. Chen et al. X-Band and Low-RCS Flexible Wideband Antenna Array Based on Metasurface [J]. IEEE Antennas and Wireless Propagation Letters, 2025, 24(3): 567-571. Document [6]: W. Yao, H. Gao, Y. Tian, et al. Wideband Low-RCS Linear Polarized Array Based on Miniaturized Polarization Conversion Metasurface[J]. IEEE Transactions on Antennas and Propagation, 2023, 71(7): 5663-5674. Document [7]: Y. Hei, M. Wang, Z. Shen, et al. Radial Line Slot Array of Low Radar Cross Section for Target Detection and Radar Stealth in Missile-Borne Sensing Systems[J]. IEEE Sensors Journal, 2023, 23(22): 28084-28094.

[0067] In summary, the present application proposes an X-shaped scattering PCM chessboard array unit, which is applied to antenna design to effectively reduce the RCS of the antenna. The overall size of the antenna is 7.85x5.12x0.172 λ0 3 , the -10 dB impedance matching bandwidth is about 34.56% (22.60-32.04 GHz), and the maximum realized gain is 14.84 dBi. In addition, the relative bandwidth of the RCS reduction value exceeding 10 dB is about 74.1% (19.07-41.52 GHz). These characteristics show that the electromagnetic dipole antenna designed by the present application has the advantages of high gain, wide bandwidth and low RCS, and has high application value in the fields of high-frequency wireless communication and radar stealth, and shows broad development prospects.

[0068] The above shows and describes the basic principles, main features and advantages of the present application. Those skilled in the art should understand that the present application is not limited to the above examples, and the above examples and descriptions in the specification are only to illustrate the principles of the present application. Without departing from the spirit and scope of the present application, various changes and improvements can be made to the present application, and these changes and improvements all fall within the scope of the claimed present application. The scope of protection of the present application is defined by the appended claims and their equivalents.

Claims

1. A broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface, comprising a first dielectric substrate (1) and a second dielectric substrate (2), wherein an intermediate layer (3) is coated between the first dielectric substrate (1) and the second dielectric substrate (2); characterized in that: The top of the first dielectric substrate (1) is provided with a radiating dipole antenna array (101) and a scattering PCM checkerboard array (102). The radiating dipole antenna array (101) is located at the center of the scattering PCM checkerboard array (102). The scattering PCM checkerboard array (102) includes a plurality of scattering PCM checkerboard array units arranged in a mirror-symmetric checkerboard array configuration. The scattering PCM checkerboard array units are X-shaped. The second dielectric substrate (2) is offset from the first dielectric substrate (1) to the left and right, and a T-type power divider feed network (201) is provided at the bottom of the second dielectric substrate (2). A coupling slot array (301) corresponding to the radiating dipole antenna array (101) is provided on the intermediate layer (3).

2. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 1, characterized in that: The radiating dipole antenna array (101) includes 1×8 radiating dipole antenna array elements, and the scattering PCM checkerboard array (102) includes 10×16 scattering PCM checkerboard array elements. Two×12 scattering PCM checkerboard array elements are removed from the central region of the scattering PCM checkerboard array (102) to accommodate the radiating dipole antenna array (101).

3. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 2, characterized in that: Above and below the central region of the scattering PCM checkerboard array (102), each 4×4 scattering PCM checkerboard array unit forms a 4×4 scattering PCM checkerboard subarray with the same rotation direction. The rotation directions of two adjacent 4×4 scattering PCM checkerboard subarrays are opposite, and the rotation directions of the 4×4 scattering PCM checkerboard subarrays at corresponding locations above and below the central region are also opposite. There are four scattering PCM checkerboard array elements on the left and right sides of the central region: the two upper scattering PCM checkerboard array elements on the left and right sides of the central region have the same rotation direction as the corresponding 4×4 scattering PCM checkerboard subarrays at the upper left and upper right edges, respectively; the two lower scattering PCM checkerboard array elements have the same rotation direction as the corresponding 4×4 scattering PCM checkerboard subarrays at the lower left and lower right edges, respectively.

4. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 2, characterized in that: The first dielectric substrate (1) is also provided with eight metallized vias (103) corresponding to the radiating dipole antenna array (101).

5. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 4, characterized in that: The coupling slot array (301) includes 1×8 coupling slots, and the coupling slot array (301) corresponds to eight metallized vias (103).

6. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 1, characterized in that: The T-type power divider feed network (201) is a 1:8 T-type power divider feed network.

7. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 6, characterized in that: The 1:8 T-type power divider feed network includes a first-stage T-type power divider (2011), a second-stage T-type power divider (2012), a third-stage T-type power divider (2013), and eight feeders (2014). The input terminal of the first-stage T-type power divider (2011) serves as the feed port (2010). The two output terminals of the first-stage power divider (2011) are divided into four output terminals through two second-stage T-type power dividers (2012). The four output terminals of the second-stage T-type power divider (2012) are then connected to the eight feeders (2014) through four third-stage T-type power dividers (2013).

8. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 7, characterized in that: The 1:8 T-type power divider feed network uses a 50Ω coaxial feed.

9. The broadband electromagnetic dipole antenna based on an X-shaped polarization conversion metasurface according to claim 1, characterized in that: The electromagnetic dipole antenna has dimensions of 86.25 mm × 56.18 mm × 1.892 mm.

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