Transimpedance amplifier

Abstract

This application relates to a transimpedance amplifier circuit, comprising a front-end amplifier that receives an input current and outputs a corresponding voltage, and a processing stage configured to process the voltage output from the front-end amplifier. The application also relates to an amplifier circuit where the processing stage has a filter having a frequency response with a local minimum. The transimpedance amplifier circuit can be used to process signals including data within a data band, and the filter can be configured such that the local minimum occurs within that data band. The filter response at least partially compensates for resonance peaks in the frequency response of the front-end amplifier caused by anticipated resonance at the input of the front-end amplifier.

Description

Technical Field

[0001] This disclosure relates to apparatus and methods for transimpedance amplifiers, as well as transimpedance amplifier circuits. Background Technology

[0002] Transimpedance amplifiers (TIAs) are used in a variety of different applications. For example, TIAs are commonly used in optical receivers for optical data communications, where the receiver receives data signals in the form of modulated optical radiation. An optical receiver typically includes a photodiode arranged to convert the received optical signal into a photocurrent and a TIA arranged to convert the photocurrent into a voltage, typically a differential voltage, which can be processed by downstream circuitry to recover the encoded data.

[0003] A key performance metric for TIA is its sensitivity, i.e., how small an input signal it can recover with an acceptable error rate or signal-to-noise ratio (SNR). The sensitivity of a TIA can generally be largely determined by its input reference noise (IRN) characteristics. Therefore, it is generally advantageous to minimize the IRN number of the TIA—specifically, within the frequency band of the signal of interest for a given application.

[0004] Embodiments of this disclosure relate to methods and apparatus for transimpedance amplifiers having improved sensitivity and / or noise performance. Summary of the Invention

[0005] According to some embodiments, a transimpedance amplifier circuit is provided, comprising: a front-end amplifier configured to receive an input current from an input terminal and output a corresponding voltage; and at least one processing stage configured to process the voltage output from the front-end amplifier. The at least one processing stage includes a filter configured to provide a filter response having a frequency response having local minima.

[0006] A transimpedance amplifier circuit can be configured to process signals including data within a data band, and a filter can be configured such that the local minimum occurs within that data band. The filter can be configured such that the filter response at least partially compensates for a resonant peak in the frequency response of the front-end amplifier due to an expected resonance at the input of the front-end amplifier. The filter can be configured such that the local minimum substantially corresponds to the inverse of the resonant peak in the frequency response of the front-end amplifier, which would be expected in use with a series inductance of a first-defined value connected to the input terminal. The filter can be configured such that the filter response includes two complex conjugate zeros in the s-domain that match two complex conjugate poles in the transfer characteristics of the front-end amplifier, which would occur in use with a series inductance of a first-defined value connected to the input terminal.

[0007] In some implementations, the transimpedance amplifier circuit may further include an inductor connected in series between the input terminal and the front-end amplifier. The inductor connected in series between the input terminal and the front-end amplifier includes at least a portion of the input resonant network. The filter may be configured such that the local minimum point substantially corresponds to the inverse of the resonant peak of the input resonant network.

[0008] In some examples, the filter may include a notch filter having a non-zero transfer function at its notch frequency. In some examples, the filter may include a passive LC filter. In some examples, the filter may include a common-source or common-emitter amplifier and may further include a frequency-dependent degradation network comprising a parallel combination of inductors and capacitors.

[0009] In some examples, the filter may include a first capacitor and an active inductor network, each connected to a first filter node. The first filter node may be coupled to the source of a common-source amplifier or the emitter of a common-emitter amplifier. The active inductor network may include a second capacitor. At least one of the first and second capacitors may be configured to be controllably variable to alter the filter response. At least one of the first and second capacitors may include a varactor diode.

[0010] In some implementations, at least one parameter of the filter is controllably variable to alter the filter response. The transimpedance amplifier circuit may also include a controller for controlling at least one parameter of the filter during use, such that local minima in the filter response at least partially compensate for peaks in the frequency response of the front-end amplifier caused by resonance at the input of the front-end amplifier.

[0011] The aspects also include a data receiver device comprising: a transimpedance amplifier circuit of any embodiment discussed herein; a data receiver configured to receive a modulated data signal and generate a corresponding current; and an electrical interconnect for connecting the output of the data receiver to an input terminal of the transimpedance amplifier circuit. The data receiver and the electrical interconnect may form at least a portion of an input resonant network, and the data receiver device may be configured such that local minima in the filter response correspond inversely to resonant peaks in the frequency response of the input resonant network. The data receiver device may be configured as an optical receiver, in which case the data receiver may be a photodiode. In some implementations, the electrical interconnect may have an inductance value such that the resonant peak lies within the frequency band of the data signal.

[0012] On the other hand, a transimpedance amplifier circuit is provided, comprising: a front-end amplifier configured to receive an input current from an input terminal and output a corresponding voltage; and at least one processing stage configured to process the voltage output from the front-end amplifier. The at least one processing stage includes a filter having a filter frequency response having a local minimum, the filter frequency response being configured to compensate for a resonant peak in the frequency response of the front-end amplifier, which would occur in use with a series inductance of a first defined value connected to the input terminal.

[0013] In a further aspect, a transimpedance amplifier circuit is provided, comprising: a front-end amplifier; and at least one processing stage in a signal path downstream of the front-end amplifier. The at least one processing stage includes a filter having a frequency response configured to at least partially compensate for a desired peaking generated in use from a resonant network connected to the input of the front-end amplifier. Attached Figure Description

[0014] To better explain the various embodiments and examples of this disclosure, their principles, exemplary implementations, and operations, reference will now be made to the following figures by way of example, wherein: Figure 1 This section shows an example of a conventional TIA as part of an optical receiver; Figure 2 The model of TIA input is shown; Figure 3 An example of a TIA with a compensation filter according to an embodiment is shown; Figures 4a to 4d Different examples of alternative universal compensation filters are shown; Figure 5 An example of a suitable compensated filter with an active inductor is shown; Figure 6 An example of a suitable compensated filter with a tunable filter response is shown; and Figure 7 An example of a differential compensation filter with a tunable filter response is shown. Detailed Implementation

[0015] Embodiments of this disclosure relate to transimpedance amplifiers (TIAs), and more specifically to TIAs having some form of filtering for mitigating the effects of peaking response associated with a given input inductor.

[0016] Figure 1 An example of at least a portion of a data receiver device 100 is shown, in which the at least a portion is an optical receiver including TIA 101. Figure 1 The device 100 is shown to have a data receiver, in this case a photodiode 102, configured to receive a modulated optical data signal Opt-In from an optical data path (not shown), which may be at least partially implemented by an optical fiber cable. The photodiode 102 converts the optical signal Opt-In into a corresponding photocurrent I. IN The photocurrent is supplied to TIA 101.

[0017] The TIA 101 features a front-end amplifier 103, in Figure 1 In the example, the front-end amplifier has a feedback resistor R FB This is implemented in a parallel feedback configuration, as those skilled in the art will understand. The TIA front-end amplifier 103 is configured to input current I... IN Convert to the corresponding voltage V TIA The TIA 101 also includes at least one back-end processing stage, in Figure 1 In the example, this back-end processing stage can provide single-ended to differential conversion to provide component S OUTP and S OUTN The differential output signal. The back-end stage 104 may additionally or alternatively provide functions such as additional amplification, filtering, equalization, automatic gain control, limiting, etc., as will be understood by those skilled in the art. The back-end stage 104 may include an output stage with an impedance-matched differential line driver for outputting the signal for downstream processing, such as demodulation or clocking and data recovery.

[0018] The key performance indicator of the TIA 101 is sensitivity, which is how small an input signal it can recover with an acceptable error rate or SNR. A relatively important parameter for TIA sensitivity is the input reference noise (IRN).

[0019] Figure 2 It shows Figure 1 The simplified small-signal model of the TIA input is shown below. The photodiode 102 is modeled as providing the input current I. IN Together with the parallel capacitor C representing the capacitance of the photodiode PD The current source 201. The model also includes the input capacitor C of the front-end amplifier 103. IN And the series inductance L in the coupling between the photodiode and the front-end amplifier 103 PEAK Inductor L PEAK This may be due to the electrical interconnection between the photodiode and the front-end amplifier 103, for example, from the bonding wires that couple the photodiode to the integrated circuit including the TIA, and the inductor L. PEAKThe value of IRN can therefore depend on the length of the associated interconnect. A major contributor to IRN is modeled as the voltage Vn applied to the input terminals of the front-end amplifier 103 (another major contributor is the feedback resistor R). FB Johnson noise).

[0020] The feedback loop surrounding the front-end amplifier 103 causes the noise voltage Vn to be higher than the voltage V at the input terminal of the front-end amplifier 103. IN This noise voltage causes voltage fluctuations. This noise voltage induces an equivalent noise current, but has a complex transfer function, determined by the input network at the input of the front-end amplifier—that is, including the photodiode capacitor C. PD Inductor L PEAK and the input capacitance C of the front-end amplifier itself IN The impedance of the network is determined.

[0021] The impedance of this input network has a certain relationship with respect to the inductor L. PEAK The dependence of the value, and the inductance L PEAK A higher value can increase the magnitude of the impedance at the input of the TIA front-end amplifier 103 for at least a certain frequency range. This can reduce the conversion ratio between the noise voltage Vn and the equivalent noise current, and thus reduce the amplifier's noise contribution in that frequency range.

[0022] However, due to inductance L PEAK and surrounding capacitance (i.e., C) PD and C IN The resulting input network is a resonant network. The resonance of this network can be damped to some extent by the resistive input impedance of the front-end amplifier 103; however, generally speaking, the resonance effect will cause a resonance peak in the frequency response of the TIA. While this resonance peak can potentially help to effectively extend the system bandwidth, resonance can cause ringing in the transient response. Ringing can last for several bit cycles, which may cause the contribution from data symbols in one bit cycle to contribute to subsequent bit cycles. If ringing occurs within the frequency band of interest of the data signal, it can thus degrade the quality of the data link by introducing ISI (inter-symbol interference) and / or DDJ (data-dependent jitter) into the data signal.

[0023] Therefore, conventionally, the TIA device 100 will be configured to cause the inductor L PEAKA value is specified such that the resonant frequency is outside and sufficiently outside the data band—that is, the band of the data signal—so that almost no energy in the data signal is available to induce resonance. Any resonant components present are outside the band of interest and can be blocked (i.e., removed) by appropriate filtering. This requires maintaining a sufficiently high resonant frequency to avoid data signal degradation from transient ringing; therefore, conventionally, this involves limiting the permissible inductance value between the photodiode and the TIA front-end amplifier.

[0024] Embodiments of this disclosure apply filtering within the back-end processing stage of the TIA, wherein the filter is configured to have characteristics that at least partially compensate for resonance effects caused by expected resonance at the input of the TIA. Specifically, the filtering can be applied together with filter transfer characteristics, which are typically the inverse of the expected resonance peak characteristics, in order to flatten the frequency response of the system. Thus, the filter can be configured to have a local minimum in its frequency response that substantially coincides with the expected frequency of the resonance peak.

[0025] Therefore, the applied filtering is not merely a filter used to block or remove any signal components in the frequency band including the expected resonant frequency of the input network and outside the data band. Rather, the filter has a transfer characteristic that is tuned with respect to the expected resonant response of the input network at the TIA's input to effectively provide a substantially inverse frequency response.

[0026] Figure 3 An example of a TIA device 300 according to an embodiment is shown, in which the TIA device 300 is configured as part of an optical receiver, wherein the same reference numerals as used above are used for similar elements.

[0027] In this example, the TIA device 300 is configured to be connected via an inductor L PEAK Receive input current I from photodiode 102 IN The TIA 300. The TIA 301 has the features described in the reference. Figure 1 The described front-end amplifier 103 and at least one back-end stage 302, which may be similar to the back-end stage 104 described above, differ in that the back-end stage 302 includes a filter 303, wherein the filter characteristics are tuned to at least partially compensate for the expected resonant characteristics at the input of the TIA front-end amplifier 103. Therefore, the filter 303 will be referred to herein as a compensation filter.

[0028] Figure 3 An example of the frequency response 304 of the front-end amplifier 103 to the input signal is shown, i.e., the transfer function. H(s)The amplitude. The frequency response 304 has a signal peak at frequency f1, which is caused by the input network (i.e., inductor L) at the input of the front-end amplifier 103. PEAK and capacitor C PD and C IN The resonance of the input network causes the signal to be inverted. To compensate for this resonance characteristic, the compensation filter 303 should ideally have a transfer function that is the inverse of the resonance characteristic. Therefore, the frequency response of the compensation filter 303 is configured to have a dip or trough at frequency f1, where the dip or trough is essentially the inverse of the resonance peak. In other words, the frequency response of the compensation filter 303 has a local minimum, where the local minimum is configured to substantially coincide with the expected resonant frequency of the input network. The amplitude of the local minimum, i.e., the depth of the dip or trough, can match the expected amplitude or peak height of the resonance peak. The frequency response of the compensation filter 303 can be substantially flat in other ways, at least in the data band of interest of the data signal. Within this data band, the local minimum can therefore also be a global minimum (however, those skilled in the art will understand that there may be some roll-off at frequencies outside the data band of interest, which is why the relevant minimum is described as a local minimum). Therefore, the overall frequency response 306 of the front-end amplifier 103 and the back-end stage 302 including the filter 303 is adjusted for the inductance of the input network due to the inductance of the input network. PEAK The resulting resonance is compensated for, and can be, for example, generally flat (up to the cutoff frequency). Note that the shape of the depression around the local minimum may be symmetrical or asymmetrical.

[0029] Because the compensation filter 303 at least partially compensates for the inductance L between the photodiode and the TIA front-end amplifier. PEAK The resulting resonance allows the inductor to have a larger value than usual. As noted above, conventionally, the inductor value would be limited to ensure the resonant frequency is sufficiently far from the data band of interest in the data signal to avoid ringing effects that degrade the signal. In embodiments of this disclosure, the use of compensation filter 303 allows this limitation to be relaxed, and the inductor L... PEAK It can be configured to have a value that results in a resonant frequency that is relatively close to or even within the data band of interest of the data signal, because the compensation filter mitigates the resonance effect.

[0030] Allowable inductance L PEAK A larger value of L is beneficial in reducing IRN because the inductance L PEAK A larger value of can lead to an increase in the effective impedance of the input network over at least a portion of the band of interest in the data signal, thereby reducing noise and improving the sensitivity of the TIA.

[0031] Therefore, the TIA device 300 can be configured with an inductor L during use. PEAKThe value of L is beneficial in reducing IRN. As mentioned above, the inductance L PEAK The value can be configured to have a value that causes the peak of the expected resonant frequency within the data band of the received data signal. The compensation filter 303 is configured to compensate for this expected resonant peak.

[0032] In some embodiments, inductor L PEAK This can be provided solely by the connection wiring between the photodiode and the input of the TIA front-end amplifier 103. In such an embodiment, the inductance L can be set by adjusting the length of this wiring. PEAK The value of the inductance is determined, for example, by selecting the length of the bonding wire or other interconnect that connects the photodiode to the TIA integrated circuit. Therefore, the length of such a bonding wire can be chosen to be longer than required by the physical dimensions of the component to provide the desired inductance value. However, in some embodiments, an inductor—that is, an element specifically configured to act as an inductor—can be added in series in the path. Such an inductor can be formed as an integrated element that forms part of the integrated circuit along with the TIA 301, or it can be formed as a discrete off-chip element. Therefore, some embodiments relate to a TIA device having a front-end amplifier stage connected to an input node for receiving input current via an inductor.

[0033] As mentioned above, the compensation filter 303 causes a local minimum in the frequency response, i.e., a dip or trough. Note that at the normal expected input signal level, the compensation filter 303 does not result in a complete zero at the local minimum; that is, the compensation filter will allow a signal component at the frequency of the local minimum to pass through, but with only reduced gain compared to the rest of the frequency band. Therefore, the reduced gain at the local minimum is finite and is configured to counteract peaking caused by input network resonance.

[0034] Note that while in some embodiments, the compensation filter 303 may operate to provide attenuation across a frequency range to provide a local minimum in its frequency response, in other embodiments, the compensation filter 303 may provide some amplification to at least some signal components, and the degree of amplification and / or attenuation may vary with frequency such that the frequency response exhibits a local minimum, i.e., a trough or dip, at a desired frequency. As used herein, references to reduced gain will therefore be used to encompass increased attenuation or reduced amplification.

[0035] In some embodiments, the peaking caused by the resonance of the input network may be relatively mild, with a moderate Q factor, and therefore the amplitude of the local minimum of the compensation filter 303—i.e., the depth of the trough in the frequency response—may be relatively low or minimal. However, even if the compensation filter 303 has a low or minimal amplitude at its local minimum and therefore has a relatively slight effect on the signal amplitude, the presence of the compensation filter 303 with a local minimum tuned to the desired resonant peak frequency can, for example, improve the phase characteristics of the output of the compensation filter by compensating for group delay, and thus provide improved flatness of the phase response, which reduces signal degradation due to ISI or DDJ.

[0036] In some embodiments, the compensation filter 303 may be connected to the input network (i.e., the inductor L). PEAK and capacitor C PD and C IN A resonant filter is selected to match the resonant response of the TIA front-end amplifier 103. The parameters of the compensation filter 303 are chosen such that the local minimum point in the frequency response of the filter within the band of interest matches the resonant peak in the frequency response of the TIA front-end amplifier 103, i.e., effectively reflecting the resonant peak. In terms of the s-domain response, the inductance of the input network can be viewed as introducing a pair of complex conjugate poles into the transfer function of the front-end amplifier 103. Therefore, the compensation filter 303 can be configured to introduce corresponding pairs of complex conjugate zeros. Ideally, the compensation filter will be tuned so that the complex zeros and complex poles are perfectly matched, so that they cancel each other out, thereby eliminating the resonance introduced by the inductor. However, practical considerations of component tolerances and variations with operating conditions such as temperature may mean that perfect cancellation across the expected range of operating conditions may not be possible. However, as long as the Q factor of the resonant peak is moderate, even an approximate match between the complex zeros and the expected complex poles is sufficient to substantially flatten the frequency response of the outputs of the compensation filter 303 and the TIA 301 and suppress ringing in the transient response.

[0037] The compensation filter 303 can be arranged as one of a plurality of back-end processing stages of the TIA 301, and can be located in the single-ended or differential signal domain, i.e., upstream or downstream of a single-ended to differential conversion stage if one exists. The compensation filter 303 can be configured as part of a back-end processing stage that provides more than one function, such as amplification in addition to filtering.

[0038] There are multiple ways to implement a compensation filter. In some examples, the compensation filter 303 can be based on the principle of a notch filter. Those skilled in the art will understand that a notch filter is a band-stop filter that provides a relatively narrow stopband in the filter response, thus exhibiting a dip or trough in the frequency response. However, conventional notch filters typically provide a zero response at the notch frequency, i.e., a total attenuation at the relevant frequency. As mentioned above, the frequency response of the compensation filter 303 does not provide a completely zero response at local minima. In some respects, the compensation filter 303 can be considered a notch filter that provides partial or limited notch characteristics, such that a relatively significant signal component still passes through at the notch frequency. However, in general, the general design principles of notch filters can be applied to the compensation filter 303.

[0039] Figures 4a to 4d Four examples of general filter designs are shown, which can serve as the basis for compensation filter 303, depending on the specific use case. Figure 4a and 4b Passive LC (inductor-capacitor) resonant filters in parallel and series configurations are shown respectively. Figure 4c A dual-T filter architecture with a non-inverting operational amplifier buffer and feedback via a voltage divider is shown. Figure 4d An active filter design is shown that uses a parallel LC network as the frequency-dependent degradation of a common-source FET amplifier, and the same applies to a common-emitter amplifier. The parallel LC network has impedance characteristics that include a local maximum at its resonant frequency. When such a network is configured for the degradation of a common-source or common-emitter amplifier, the amplifier's frequency response will include a local minimum corresponding to the local maximum of the LC network's impedance. It is assumed that the power supply and DC voltage component Vin are appropriately chosen to bias the transistor at a favorable operating point. Those skilled in the art will appreciate how such filters can be implemented to match the desired resonant characteristics, and how the width and depth of the local minimum—i.e., the Q-factor and relative attenuation of the dip in the frequency response—can be modified using additional series or parallel resistors.

[0040] Figure 5 A practical example of a suitable compensation filter is shown, which is compared with... Figure 4dThe degraded common-source (or common-emitter) amplifier shown is similar, but the inductor is replaced by an active inductor 501. As those skilled in the art will understand, an active inductor is an arrangement or network of elements that do not actually contain an inductor but provide small-signal impedance characteristics that include virtual inductance elements. For integrated circuit applications, this can be advantageous in some cases because providing an inductor with the required inductance may require a relatively significant circuit area and thus increase the circuit area and / or cost, while an active inductor network can be implemented in a smaller circuit area and / or at a lower cost. Figure 5 An example of an active inductor with a FET as the active device and a capacitance Cind is shown. Although Figure 5 The transistors shown in the related figures are depicted as FETs, but the techniques and topologies considered are equally applicable to circuits using bipolar transistors instead of FETs. The active inductor 501 can be approximated as a small-signal equivalent network 502 comprising an inductance Lfil with a value approximately proportional to the capacitance Cind and a series resistor providing damping elements. Furthermore, it is assumed that a DC bias of Vin is suitable for biasing the transistor at a suitable operating point. Those skilled in the art will recognize other examples of active inductors that can be used in other implementations.

[0041] For reference Figure 4a to d and Figure 5 The compensation filter discussed can have its response characteristics tuned by appropriately selecting the parameters of various components (e.g., inductors, capacitors, resistors, etc.), as those skilled in the art will understand. These parameters can be selected to match the filter characteristics to the desired resonant characteristics, i.e., to make the filter transfer function substantially correspond to the inverse of the resonant peaking at the front end.

[0042] In some cases, a TIA device can be designed for applications where all parameters of the input network for the TIA are known in advance—for example, where it is known that the TIA will be used with a specific type of photodiode with a known photodiode capacitance and also that the type and length of the interconnect between the photodiode and the TIA are known—preferably chosen to provide an inductance value L that results in low IRN and thus high sensitivity. PEAK In such cases, the resonant characteristics of the input network can be determined, for example, through appropriate modeling / simulation, and the parameters of the compensation filter can be selected accordingly.

[0043] However, in some cases, TIA circuits can be fabricated for applications where a range of possible input networks can be implemented—for example, where the photodiodes used can be selected by the device manufacturer from a range of different photodiodes and the type and / or length of the interconnects can be varied. In this case, a typical photodiode capacitance can be assumed, and an inductor L that is feasible and results in good noise performance can be selected. PEAK The values ​​of TIA can then be used to tune the compensation filter to the desired resonance based on these defined values ​​(i.e., assumed values), and the device manufacturer can be provided with recommendations for the value of the input inductance or the length of the interconnect that will result in optimal performance.

[0044] However, in some embodiments, the compensation filter can be configurable, allowing the filter response to be tuned in use, i.e., allowing the shape of the filter response (i.e., local minima) to be controllably changed. For example, the compensation filter can be configured such that at least one of the center frequency of the dip or trough, the width of the dip or trough, and the depth of the dip or trough is variable, or in other words, at least some of the poles and zeros in the s-domain can be changed. This allows for efficient configuration of the filter response of the compensation filter at the device manufacturer's level to better tune it to compensate for the actual resonant characteristics of the input network.

[0045] The filter parameters can be tuned by using components with variable values. For example, for Figure 5 In the example, where one terminal of each of the capacitors Cfil and Cind is AC ground, one or both of these capacitors Cfil and Cind can be implemented as variable capacitors. This would allow the value of Cfil to be changed separately, as well as the value of the active inductor or effective inductance Lfil. Various ways exist in which variable capacitors can be implemented, for example, by controlling the switching of capacitor arrays with different values.

[0046] Figure 6 An example of a tunable compensation filter is shown, which is related to Figure 5 Similar to the description in [the text], but in this case, the capacitances Cfil and Cind are each variable. In [the text] Figure 6 In the example, capacitors Cfil and Cind are provided by corresponding varactor diodes 601 and 602, which may include, for example, MOS varactor diodes whose capacitance can be changed by a DC control voltage. The DC control voltage of varactor diodes 601 and 602 can be controlled by controller 603 based on control signal Scon.

[0047] As described above, controller 603 can control the DC control voltage of varactor diodes 601 and 602 based on control signals, and thereby control the capacitance values ​​of Cfil and Cind to tune the compensation filter response to appropriately match the resonant characteristics of the resonant network at the input of the TIA. Therefore, the control signals may include settings that can be programmed by the device manufacturer and / or can be determined as part of a calibration process, which may be factory calibration during device manufacturing and / or part of an automated calibration process that may be repeated periodically to accommodate any changes. Alternatively, controller 603 can be configured to dynamically change the capacitance values ​​of Cfil and Cind during use, for example, to accommodate changes in the characteristics of the input resonance and / or filter response due to changes in operating conditions such as temperature or supply voltage or input signal level. Therefore, control signal Scon may indicate the relevant operating conditions, such as temperature, to allow control to appropriately adapt to the compensation filter. In some implementations, the control signals may include a feedback signal indicating how effectively the compensation filter compensates for the resonant peak.

[0048] As mentioned above, compensation filters can be implemented in the single-ended signal domain or the differential signal domain, and the differential implementation can be advantageous for noise suppression in some applications. Figure 7 An example of a differential filter suitable for use as compensation filter 303 is shown. Figure 7 The compensation filter essentially includes, as in the reference... Figure 6 The discussion focuses on filters for each of the positive and negative signal components of the differential input, and thus on filters with variable capacitance implemented by varactor diodes to allow for tuning of the filter characteristics. Figure 7 The differential compensation filter also includes a current source 701 shared between the two active inductors 501, which provides a convenient means of controlling the DC bias current of the active devices.

[0049] Therefore, generally speaking, embodiments involve including a compensation filter in a TIA device, specifically in the back-end stage of the TIA circuit (i.e., downstream of the front-end TIA amplifier), wherein the compensation filter has a frequency response that has a local minimum. This allows the input series inductor L PEAK The value is larger than in other appropriate cases, thus allowing for greater suppression of the input reference noise contributed by the TIA front-end amplifier. If the compensation filter is well optimized to counteract the noise caused by L... PEAKThe introduced peaking in the TIA frequency response allows the overall sensitivity of the TIA to be improved due to reduced input noise without compromising signal integrity. This optimization can be facilitated by providing power adjustment or tuning compensation filter parameters (e.g., via a variable capacitor). Increased TIA input series inductance can be achieved by increasing the length of the interconnect wiring between the photodiode and the TIA IC and / or by including a discrete inductor (e.g., an on-chip inductor) coupled in series with the photodiode.

[0050] A TIA circuit system can be integrated with a suitable photodiode to form part of a light receiver. However, these principles apply to any TIA device in which a resonant network may exist at the TIA input, and other embodiments relate to other applications.

[0051] It will be understood that the examples and embodiments described above are given by way of example only, and those skilled in the art will understand that modifications, variations, additions or alterations can be made to the specific embodiments described, or alternative embodiments can be implemented without departing from the scope of the appended claims.

[0052] It should be noted that, as used herein, unless otherwise expressly stated, the word "comprising" does not exclude the presence of other elements or steps besides those listed, references to a singular element or feature do not exclude the possibility of multiple such elements or features, and the recitation of different features or elements in the appended claims does not necessarily imply a single element; a single element or unit can perform the function of several elements recited in the claims. Any reference numerals in the appended claims should not be construed as limiting their scope.

Claims

1. A transimpedance amplifier circuit, comprising: A front-end amplifier configured to receive input current from an input terminal and output a corresponding voltage; as well as At least one processing stage, the at least one processing stage being configured to process the voltage output from the front-end amplifier; The at least one processing stage includes a filter configured to provide a filter response having a frequency response having a local minimum.

2. The transimpedance amplifier circuit of claim 1, wherein the transimpedance amplifier circuit is configured to process a signal comprising data within a data band, and wherein the filter is configured such that the local minimum point is within the data band.

3. The transimpedance amplifier circuit of claim 1 or claim 2, wherein the filter is configured such that the filter response at least partially compensates for the resonant peak in the frequency response of the front-end amplifier due to the expected resonance at the input of the front-end amplifier.

4. The transimpedance amplifier circuit of any one of claims 1 to 3, wherein the filter is configured such that the local minimum point substantially corresponds to the inverse of the resonant peak in the frequency response of the front-end amplifier, the resonant peak being anticipated in use with a series inductance of a first defined value connected to the input terminal.

5. The transimpedance amplifier circuit of any one of claims 1 to 4, wherein the filter is configured such that the filter response includes two complex conjugate zeros in the s-domain that match two complex conjugate poles in the transfer characteristics of the front-end amplifier, the complex conjugate zeros occurring in use with a series inductance of a first defined value connected to the input terminal.

6. The transimpedance amplifier circuit as claimed in any one of claims 1 to 5, further comprising an inductor connected in series between the input terminal and the front-end amplifier.

7. The transimpedance amplifier circuit of claim 6, wherein the inductor connected in series between the input terminal and the front-end amplifier includes at least a portion of the input resonant network, and wherein the filter is configured such that the local minimum point substantially corresponds to the inverse of the resonant peak of the input resonant network.

8. The transimpedance amplifier circuit according to any one of claims 1 to 7, wherein the filter includes a notch filter, the notch filter having a non-zero transfer function at the notch frequency of the notch filter.

9. The transimpedance amplifier circuit according to any one of claims 1 to 8, wherein the filter comprises a passive LC filter.

10. The transimpedance amplifier circuit of any one of claims 1 to 9, wherein the filter comprises a common-source or common-emitter amplifier, and further comprises a frequency-dependent degradation network, wherein the frequency-dependent degradation network comprises a parallel combination of an inductor and a capacitor.

11. The transimpedance amplifier circuit of any one of claims 1 to 10, wherein the filter comprises a first capacitor and an active inductor network, each connected to a first filter node.

12. The transimpedance amplifier circuit of claim 11, wherein the first filter node is coupled to the source of a common-source amplifier or the emitter of a common-emitter amplifier.

13. The transimpedance amplifier circuit of claim 11 or claim 12, wherein the active inductor network includes a second capacitor.

14. The transimpedance amplifier circuit of claim 13, wherein at least one of the first capacitor and the second capacitor is configured to be controllably variable in order to change the filter response of the filter.

15. The transimpedance amplifier circuit of claim 14, wherein at least one of the first capacitor and the second capacitor comprises a varactor diode.

16. The transimpedance amplifier circuit of any one of claims 1 to 15, wherein at least one parameter of the filter is controllably variable to alter the filter response.

17. The transimpedance amplifier circuit of claim 16, further comprising a controller configured to control the at least one parameter of the filter in use such that the local minima in the filter response at least partially compensate for peaks in the frequency response of the front-end amplifier caused by resonance at the input of the front-end amplifier.

18. A data receiver device, comprising: The transimpedance amplifier circuit as described in any one of claims 1 to 17; A data receiver configured to receive modulated data signals and generate corresponding current; as well as Electrical interconnection, the electrical interconnection being used to connect the output of the data receiver to the input terminal of the transimpedance amplifier circuit; The data receiver and the electrical interconnect form at least a portion of the input resonant network, and the device is configured such that the local minima in the filter response correspond inversely to the resonant peaks in the frequency response of the input resonant network.

19. The data receiver device of claim 18, wherein the data receiver device is configured as an optical receiver, wherein the data receiver is a photodiode.

20. The data receiver device of claim 18 or claim 1, wherein the electrical interconnect has an inductance value such that the resonant peak is within the frequency band of the data signal.

21. A transimpedance amplifier circuit, comprising: A front-end amplifier configured to receive input current from an input terminal and output a corresponding voltage; as well as At least one processing stage, the at least one processing stage being configured to process the voltage output from the front-end amplifier; The at least one processing stage includes a filter with a filter frequency response having a local minimum point, the filter frequency response being configured to compensate for a resonant peak in the frequency response of the front-end amplifier, the resonant peak being present in use with a series inductance connected to the input terminal of a first defined value.

22. A transimpedance amplifier circuit, comprising: Front-end amplifier; as well as At least one processing stage, wherein the at least one processing stage is in the signal path downstream of the front-end amplifier; The at least one processing stage includes a filter with a frequency response configured to at least partially compensate for a desired peaking that is generated in use from a resonant network connected to the input of the front-end amplifier.

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