A D-band FMCW radar transceiver front-end based on high-power frequency multiplier

Abstract

The application discloses a D-wave FMCW radar transceiver front end based on a high-power frequency multiplier, and relates to the technical field of radar communication.The technical problems of insufficient output power, high receiving noise, large signal transmission loss and difficult spectrum purity control of the existing terahertz FMCW radar transceiver front end under high-frequency conditions are solved.The application comprises a chirp signal generator used for inputting a same chirp local signal to a transmitting module and a receiving module respectively; the transmitting module is used for generating a D-wave terahertz transmitting signal based on the chirp local signal and transmitting the signal to a detection target; the receiving module is used for receiving and down-conversion processing a signal reflected by the detection target and the chirp local signal to obtain an IF beat signal and outputting the IF beat signal; an electric control module is used for communicating with an upper computer and controlling each device in other modules through a communication protocol; and the application realizes high output power, low phase noise and good spectrum performance.

Description

Technical Field

[0001] This invention relates to the field of radar communication technology, and more specifically to a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier. Background Technology

[0002] Frequency-modulated continuous wave (FMCW) radar transmits a continuous signal whose frequency varies linearly with time, and uses the frequency difference between the echo signal and the transmitted signal to obtain target range and velocity information. Due to its low probability of intercept and high measurement accuracy, it is widely used in radar detection systems. With the increasing demand for high-resolution detection, the terahertz band (0.1–10 THz), with its wider bandwidth and shorter wavelength, can significantly improve the range resolution and imaging capabilities of radar systems. Currently, terahertz FMCW radar transceiver front-ends mainly adopt monolithic integrated architecture or modular architecture. Monolithic integrated solutions have the advantages of small size and high integration, but under high-frequency conditions, the power amplifier output power is limited, the performance of low-noise amplifiers and mixers degrades, and isolation between the transmit and receive channels is difficult. While modular architectures are relatively larger, they can fully utilize the performance advantages of different components and have certain advantages in high-power output and system expansion. However, existing terahertz FMCW radar transceiver front-ends still suffer from insufficient output power, high receive noise, and significant high-frequency signal transmission loss. Summary of the Invention

[0003] To address the problems existing in the prior art, this invention provides a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, which solves the technical problems of insufficient output power, high receiving noise, large signal transmission loss, and difficulty in controlling spectral purity in existing terahertz FMCW radar transceiver front-ends under high-frequency conditions.

[0004] A D-band FMCW radar transceiver front-end based on a high-power frequency multiplier includes a transmitting module, a receiving module, a chirped signal generator, and an electronic control module;

[0005] The chirp signal generator is used to input the same chirp local oscillator signal to both the transmitting module and the receiving module.

[0006] The transmission module is used to generate a D-band terahertz transmission signal based on the chirp local oscillator signal and transmit it to the detection target.

[0007] The receiving module is used to receive and down-convert the signal reflected back from the detection target and the chirp local oscillator signal to obtain the IF beat frequency signal and output it.

[0008] The electronic control module is used to communicate with the host computer and control the various devices in other modules through the communication protocol.

[0009] Furthermore, the chirped signal generator includes a frequency modulation (FM) signal link, which includes: a temperature-controlled crystal oscillator generating a stable reference frequency signal; the reference frequency signal is first split into two paths by a first power divider, one path enters the phase-locked loop (PLL) control circuit through a second power divider, and the other path enters the comb spectrum generator through a third power divider; then, it enters the digital-to-analog converter (DAC) through a first bandpass filter; subsequently, the outputs of the PLL control circuit and the DAC are mixed in an up-converter mixer 1 to obtain an FM signal; the FM signal enters the second power divider after passing through a second bandpass filter and is split into two chirp local oscillator signals, which are respectively sent to the transmitting module and the receiving module.

[0010] Furthermore, the chirped signal generator also includes a multi-point frequency signal link, which includes: one output signal of the third power divider generating multiple sets of point frequency signals via the first voltage-controlled oscillator synthesizer and sending them to the transmitting module; and one output signal of the second power divider generating multiple sets of point frequency signals via the second voltage-controlled oscillator synthesizer and sending them to the receiving module.

[0011] Furthermore, the transmitting module includes a transmitting local oscillator link, and the receiving module includes a receiving local oscillator circuit. The transmitting local oscillator circuit includes: the chirp local oscillator signal sequentially passes through a first 6x frequency multiplier, a second waveguide filter, and a second drive amplifier, and then is mixed with a point frequency signal in a second up-conversion mixer to form an E-band drive frequency band, and finally outputs through a second waveguide filter; the receiving local oscillator circuit includes: the chirp local oscillator signal sequentially passes through a second 6x frequency multiplier, a third waveguide filter, and a third drive amplifier, and then is mixed with a point frequency signal in a third up-conversion mixer to form an E-band drive frequency band, and finally outputs the transmitting local oscillator signal through a fourth waveguide filter.

[0012] Furthermore, the transmitting module also includes a transmitting link, which includes a cascaded power amplifier, a frequency multiplier, and an antenna connected in sequence. The transmitting local oscillator signal is sent to the input terminal of the cascaded power amplifier and then transmitted to the detection target by the transmitting antenna.

[0013] Furthermore, the frequency multiplier adopts a frequency multiplication structure based on GaN Schottky diodes and combines it with a suspended microstrip transmission structure to achieve low-loss signal transmission. At the same time, it enhances local electric field coupling through resonant microstructures, thereby improving frequency conversion efficiency and output power.

[0014] Furthermore, the receiving module also includes a receiving link, which includes: receiving a signal reflected back from the target by a receiving feed, amplifying it by a low-noise amplifier, and then entering a second harmonic mixer to perform frequency down-conversion with the received local oscillator signal to obtain an IF beat frequency signal. The IF beat frequency signal passes sequentially through a third bandpass filter, a cascaded first variable gain amplifier, and a second variable gain amplifier before being output from the output port. A cascaded first signal switching switch and a second signal switching switch are provided between the first variable gain amplifier and the second variable gain amplifier. A fourth bandpass filter and a fifth bandpass filter are provided in parallel between the first signal switching switch and the second signal switching switch.

[0015] Furthermore, the operational process includes:

[0016] Step 1: The electronic control module provides a stable power supply to each module and completes the temperature-stabilized clock and reference source locking;

[0017] Step 2: The chirp signal generator outputs the corresponding point frequency signal to the transmitting module and the receiving module respectively, and the transmitting module and the receiving module select the target center frequency point working sub-band;

[0018] Step 3: The chirp signal generator sets the chirp signal sweep bandwidth and sweep duration and synchronously switches the IF filter bandwidth path, and then sends the chirp signal to the transmitting module and the receiving module.

[0019] Step 4: The transmitting module mixes the point frequency signal and the chirped signal to form an E-band drive signal. The E-band drive signal is then passed through a power amplifier and frequency multiplier in sequence to output a D-band terahertz signal, which is transmitted to the horn antenna through a waveguide and then reflected by the parabolic reflector antenna and directed toward the detection target.

[0020] Step 5: The terahertz signal reflected back from the target enters the RX link through the horn feeder, is first amplified by the LNA, and then mixed with the LO in the second harmonic mixer to obtain the IF beat frequency signal.

[0021] Step 6: The IF beat frequency signal is amplified by two stages of VGA and then band-limited by a dual-bandwidth bandpass filter before being output to the back end;

[0022] Step 7: The backend calculates and analyzes the IF beat frequency to obtain the corresponding information.

[0023] The beneficial effects of this invention include: addressing the problems of insufficient output power, high receiving noise, large signal transmission loss, and difficulty in controlling spectral purity in existing terahertz FMCW radar transceiver front-ends under high-frequency conditions, this invention proposes a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier. This front-end adopts a modular system architecture. In the transmit link, a high-power frequency multiplier based on a GaN Schottky diode is used to achieve high-power terahertz signal output. In the receive link, a low-noise amplifier and a subharmonic mixer are combined to achieve low-noise signal down-conversion. Furthermore, reasonable local oscillator spectrum planning and a low-loss transmission structure reduce system spurious emissions and transmission loss, thereby achieving a D-band FMCW radar transceiver front-end system with high output power, low noise, and good spectral performance. Attached Figure Description

[0024] Figure 1 This is a circuit architecture diagram of a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, according to an embodiment of this application.

[0025] Figure 2 This is a block diagram of the frequency and phase noise of the radio frequency link involved in the embodiments of this application.

[0026] Figure 3 These are measured data of the transmission power involved in the embodiments of this application.

[0027] Figure 4 This refers to the output intermediate frequency signal test data involved in the embodiments of this application. Figure 4 (a) is the spectrum of the intermediate frequency signal when it is saturated. Figure 4 (b) is the test data for intermediate frequency signal spurious signals.

[0028] Figure 5 This is the test data for the phase noise of the transmit excitation signal involved in the embodiments of this application.

[0029] Figure 6 This is a schematic diagram of the structure of a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, according to an embodiment of this application.

[0030] Figure label:

[0031] 11-Horn antenna, 12-Frequency multiplier, 13-First power amplifier, 14-First waveguide filter, 15-Second up-conversion mixer, 16-Second driver amplifier, 17-Second waveguide filter, 18-First 6th harmonic multiplier, 21-Third waveguide filter, 22-Third driver amplifier, 23-Third up-conversion mixer, 24-Fourth waveguide filter, 25-Second power amplifier, 26-Second harmonic mixer, 27-Horn feed, 28-Low noise amplifier, 29-Third bandpass filter, 30-First variable gain amplifier, 31-First signal switching switch, 32-Fourth bandpass filter, 33-Fifth bandpass filter, 34-Second signal switching switch, 35-Second variable gain amplifier. 36-Second 6th frequency multiplier, 40-Temperature-controlled crystal oscillator, 41-First power divider, 42-Second power divider, 43-Third power divider, 44-Comb spectrum generator, 45-First bandpass filter, 46-First voltage-controlled oscillator synthesizer, 47-Digital-to-analog converter, 48-Phase-locked loop control circuit, 49-First driver amplifier, 50-First up-conversion mixer, 51-Second bandpass filter, 52-Fourth power divider, 53-Second voltage-controlled oscillator, 61-Power supply module, 62-Control module, 71-Transmitter arm, 72-Receiver arm, 73-Top cover, 74-First transmitter module, 75-First receiver module, 76-Second transmitter module, 77-Second receiver module, 100-Chirp generator, 101-Electronic control module. Detailed Implementation

[0032] To make the objectives, technical solutions, and advantages of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of this application, and not all of the embodiments. Therefore, the following detailed description of the embodiments of this application provided in the accompanying drawings is not intended to limit the scope of the claimed application, but merely represents selected embodiments of this application. All other embodiments obtained by those skilled in the art based on the embodiments of this application without creative effort are within the scope of protection of this application.

[0033] Example 1

[0034] The following is in conjunction with the appendix Figure 1 Specific embodiments of the present invention will be described in detail;

[0035] This embodiment provides a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, with an operating frequency band of 139.5–148.5 GHz, and supports three switchable center frequency points (e.g., 143.5 GHz, 144 GHz, 144.5 GHz and ±4 GHz coverage).

[0036] This embodiment employs the FMCW (Frequency Modulated Continuous Wave) system: a frequency modulation signal source generates a linear frequency modulated continuous wave. Its instantaneous frequency is within the sweep bandwidth The frequency sweep duration varies linearly with time. .

[0037] The transmitted signal is output to the antenna via a terahertz transmission link and radiated into space; the target echo has a propagation delay. (R is distance, c is speed of light), echo signal ( (This is the attenuation coefficient).

[0038] In the receiver, the echo signal and the local oscillator signal are multiplied in the mixer to form the beat signal. Its main component lies within the intermediate frequency (IF) bandwidth, and the beat frequency is approximately proportional to the distance, which can be expressed as:

[0039]

[0040] Therefore, the target distance information can be obtained by bandpass filtering and spectrum / FFT processing of the IF beat frequency signal.

[0041] Based on the conventional relationship of the FMCW system, the maximum unambiguous detection distance and the IF filter bandwidth in this embodiment are... Sweep bandwidth Frequency sweep duration Related; distance resolution is determined by bandwidth, and is approximately:

[0042]

[0043] For example, when the sweep bandwidth is approximately 8 GHz, the distance resolution can reach the order of approximately 18.75 mm (based on...). (Estimation).

[0044] To adapt to different ranging requirements, this embodiment supports two frequency sweep duration modes: and The receiver's IF filter provides two switchable bandwidth modes (e.g., 50 MHz and 90 MHz) to achieve a trade-off between maximum detection range and ranging refresh rate.

[0045] A D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, such as Figures 1-2 As shown, it includes a transmitting module, a receiving module, a chirp signal generator, and an electronic control module;

[0046] The chirp signal generator is used to modulate and generate a chirp local oscillator signal shared by the transmitting and receiving modules, and sends it to the transmitting and receiving modules respectively. The frequency modulation signal link of the chirp signal generator is as follows: at the beginning of the link, a stable 100MHz reference frequency signal is generated by a temperature-controlled crystal oscillator 40 (OCXO), which first passes through a first power divider 41, and then through a second power divider 42 to enter a phase-locked loop control circuit 48 to stabilize the output frequency of the microwave oscillator.

[0047] In this embodiment, the phase-locked control circuit 48 uses a phase-locked dielectric resonant oscillator (PLLDRO) as a low-phase-noise reference source to generate an 8.2 GHz reference signal. Because the PLLDRO has a high resonance quality factor (Q value), its output signal has extremely low phase noise in the microwave band, thus providing a high-quality frequency reference for the subsequent frequency synthesis process.

[0048] Subsequently, a linear frequency modulation (chirp) signal is generated at the low-frequency end. Specifically, the 100MHz signal generated by the OCXO passes through the first power divider 41 and the third power divider 43 in sequence and enters the comb spectrum generator to obtain a signal group with an integer multiple of the fundamental frequency. Then, it passes through the first bandpass filter 45 to obtain a 6GHz signal, which enters the digital-to-analog converter 47 (DAC) as a clock synchronization reference.

[0049] As a frequency modulation (FM) signal generation unit, the DAC directly generates a linear frequency modulated (LFM) signal with a center frequency of approximately 2.375 GHz and a sweep range of approximately ±0.34 GHz. The frequency of this signal changes linearly with time according to a predetermined slope. Because the modulation process occurs within a lower frequency range, it is easier to achieve good linearity and low phase noise, thereby improving the ranging accuracy and range resolution of the FWCW radar system. The core DAC chip used in the specific DAC module is the AD9164.

[0050] After obtaining the low-frequency FM signal, the frequency is first boosted through mixing. The low-phase-noise 8.2GHz reference signal generated by PLLDRO is amplified by the first driver amplifier 49 and then mixed with the approximately 2.375 ± 0.34 GHz chirp signal output from the DAC in the first upconverter mixer 50. The frequencies are then added together to obtain an FM signal of approximately 10.575 ± 0.34 GHz.

[0051] Compared to directly obtaining high-frequency signals through high-frequency harmonics, using a mixer-upconversion method can significantly reduce the degradation of phase noise. This is because the frequency multiplier amplifies the phase noise by a multiplication factor, while the mixing process has a relatively small impact on phase noise. Theoretically, when a signal undergoes N-fold frequency harmonics, its phase noise typically increases by approximately 20log. 10(N)dB, therefore, by generating chirp at the low-frequency end and using mixing to boost the frequency, the phase noise performance of the system can be effectively maintained.

[0052] The 10.575 ± 0.34 GHz chirp signal output from the first upconverter mixer 50 is filtered out by the second bandpass filter 51 to remove noise generated during mixing, and then enters the power distribution network, namely the fourth power divider 52, to distribute and output two signals, which are sent to the transmit local oscillator link and the receive local oscillator link respectively, so that the transmit module and the receive module share the same chirp local oscillator signal.

[0053] This shared local oscillator structure ensures that the transmitted signal and the received demodulated signal have strictly consistent frequency modulation characteristics, which is an important condition for FMCW radar to achieve beat frequency detection and accurate distance calculation.

[0054] In addition, to enable flexible switching between multiple working subbands, multiple sets of fixed-frequency spot signals are generated as frequency offsets during the up-conversion process. Specifically, the 100MHz signal generated by the temperature-controlled crystal oscillator 40 passes through the first power divider 41 and the third power divider 43 to enter the first voltage-controlled oscillator synthesizer 46 in the transmit link, and passes through the second power divider 42 to enter the second voltage-controlled oscillator synthesizer 53 in the receive link. Corresponding to the transmit link, it can generate, for example, three sets of spot frequency signals of 8.3 GHz, 8.55 GHz, and 8.8 GHz; corresponding to the receive link, it generates three sets of spot frequency signals of 8.4875 GHz, 8.7375 GHz, and 8.9875 GHz.

[0055] By selecting different point-frequency signals and adding or subtracting them from a shared chirp local oscillator signal in a mixer, intermediate local oscillator signals with different center frequencies can be obtained. Subsequently, these signals undergo further frequency multiplication or mixing to be converted to the E-band or higher millimeter-wave bands, ultimately forming the transmit and receive signals required for the target D-band operating subband.

[0056] The local oscillator link of the transmitting module includes a first 6x frequency multiplier 12, a second waveguide filter 17, a second drive amplifier 16, a second up-conversion mixer 15, a first waveguide filter 14, and a first power amplifier 13.

[0057] The local oscillator link of the receiving module includes a second 6x frequency multiplier 12, a third waveguide filter 21, a third drive amplifier 22, a third up-conversion mixer 23, a fourth waveguide filter 24, and a second power amplifier 25.

[0058] Since frequency multipliers and mixers are nonlinear devices and are the main sources of spurious emissions, this embodiment performs spectrum planning on the LO frequency and intermediate frequency point to keep high-order spurious emissions as far away from the operating bandwidth as possible, and places waveguide filters at key nodes for suppression.

[0059] The signal processing procedures for the transmit and receive local oscillator links are the same. Taking the transmit local oscillator link as an example, the process includes: a chirp signal of 10.575 ± 0.34 GHz is passed through a first 6x frequency multiplier 12 to obtain a chirp LO of approximately 63.45 ± 2 GHz. This LO is then passed through a second waveguide filter 17 covering approximately 61–65 GHz to suppress harmonic-related spurious signals. After passing through a second drive amplifier 16, the signal is mixed with a point frequency signal via a second up-conversion mixer 15 to form an E-band drive frequency. Subsequently, the signal passes through a first waveguide filter 14 covering approximately 69.75–74.25 GHz to further suppress mixing and higher-order spurious signals, while also considering the LO power requirements of the TX and RX links. The first waveguide filter 14 and the second waveguide filter 17 employ multi-cavity coupled waveguide filters (Chebyshev response) to achieve a trade-off between insertion loss and out-of-band suppression.

[0060] The transmitting module includes a power amplification link and a frequency multiplier 12. The power amplification circuit adopts a two-stage cascaded power amplifier architecture (which can be one large and one small stage) to amplify the E-band signal to a power level capable of driving the frequency multiplier 12. To reduce the transmission loss of the E / D band PCB, it is preferable to use CNC-machined copper parts with gold plating to construct the waveguide / cavity structure; an E-plane probe is used for the waveguide-microstrip / suspended microstrip transition between the waveguide and the planar transmission line, and the chip is connected to the transition structure by gold wire bonding.

[0061] Specifically, the power amplification link is the first power amplifier 13 shown in the figure. The first power amplifier 13 is composed of two cascaded power amplifiers. The first power amplifier 13 provides a saturated output power of not less than 26.9 dBm in the range of approximately 69.75–74.25 GHz to compensate for approximately 1–2 dB of passive losses caused by package interconnects, filtering and waveguide bends, etc., and to ensure that the effective drive power of the frequency multiplier is close to 26 dBm.

[0062] The frequency multiplier 12 is a 72G to 144G double frequency multiplier. Its main function is to double the frequency of the input signal to twice and then output it, which is then transmitted through the horn antenna 11.

[0063] This 72GHz to 144GHz frequency multiplier is a key frequency boosting unit in the transmit link, preferably employing a balanced frequency multiplication structure with GaN Schottky diodes as its nonlinear components. The frequency multiplier consists of two sets of anti-tandem GaN diode pairs forming a balanced topology, and multiple small-sized diodes can be connected in series to form a multiplication unit, improving voltage withstand and power handling capabilities. The circuit preferably uses a suspended microstrip structure to reduce dielectric loss and widen the matching bandwidth. The input is coupled to the fundamental signal via a WR-12 waveguide, and the output uses a WR-7 waveguide to extract the second harmonic (2f) signal and suppress higher harmonics. This frequency multiplier operates in the 70–75 GHz input band and outputs a 140–150 GHz second harmonic signal, with a typical conversion efficiency on the order of 10%–20%. The input matching maintains an S11 better than −10 dB within the operating bandwidth. In addition, the circuit introduces DC bias through an external bias network and filter structure to ensure that the frequency multiplier operates stably under high input power conditions and achieves good frequency multiplication efficiency.

[0064] The receiving module is used to amplify, downconvert, and condition the millimeter-wave signal from the target echo to obtain a stable IF output suitable for back-end signal processing.

[0065] The receiver module's receiver link includes: a horn feed 27 (receiving antenna), a low-noise amplifier 28 (LNA), a second harmonic mixer, a variable gain amplifier (VGA), a bandpass filter, and an intermediate frequency output interface. The overall structure employs a step-by-step amplification and frequency down-conversion method to convert millimeter-wave signals in the approximately 144 GHz range into intermediate frequency signals in the hundreds of MHz range. Adjustable gain and selectable bandwidth enable flexible adaptation of the system's dynamic range and ranging modes.

[0066] The receiving link first receives the millimeter-wave echo signal from the space target via a horn feed 27. The signal frequency is located in the approximately 144 GHz operating band, and the signal power is typically low. A terahertz low-noise amplifier 28 (LNA) is placed near the antenna to minimize the front-end noise figure and reduce the impact of waveguide transmission loss on the system sensitivity.

[0067] In a preferred embodiment, the low-noise amplifier 28 is installed directly adjacent to the feed source or waveguide transition structure, thereby shortening the millimeter-wave transmission path and reducing insertion loss. After amplification by the low-noise amplifier 28, the received signal power is significantly improved, while the overall system noise figure is effectively controlled, providing sufficient signal level for subsequent mixing and downconversion.

[0068] After front-end amplification, the millimeter-wave signal enters a second harmonic mixer for frequency down-conversion. To reduce the local oscillator frequency requirement and simplify the millimeter-wave local oscillator link, this embodiment employs a second harmonic balanced subharmonic mixer structure. This structure typically includes a waveguide-to-suspended microstrip transition structure, a LO-IF duplex network, and a pair of Schottky diodes for mixing.

[0069] During operation: the RF signal (approximately 144 GHz) is input through the waveguide port; the LO signal is provided by the receiver local oscillator link and input to the mixer at approximately half the RF frequency; the diode generates the second harmonic under LO drive, thereby achieving... The mixing process. By using the second harmonic of the LO signal for mixing, the LO frequency requirement can be reduced while maintaining low conversion loss, thereby simplifying the millimeter-wave local oscillator design and improving system stability.

[0070] In a preferred embodiment, the second harmonic mixer is a 144GHz second harmonic mixer that achieves a single-sideband conversion loss of approximately 10 dB in the frequency range of 139.5–148.5 GHz, ensuring sensitivity while also reducing device complexity. The signal generated after mixing is an intermediate frequency (IF) beat frequency signal, the frequency of which is determined by the frequency difference between the transmitted chirp signal and the echo signal, and carries target distance information.

[0071] To adapt to different target distances and echo intensity conditions, this embodiment sets up a first variable gain amplifier 30 and a second variable gain amplifier 35 cascaded in the IF link. The variable gain amplifiers typically employ digital control, such as 6-bit gain control, with each gain step ranging from 0.5 dB to 1 dB, thereby achieving a wider dynamic range of gain adjustment.

[0072] Signal switching switches 1 and 2 (SPDT) are placed between the two stages of variable gain amplifiers to allow selection of different IF bandwidth paths. Specifically, the system selects different bandpass filters 4 or 5 via the switches to adapt to different frequency modulation modes.

[0073] Mode 1: Narrowband mode (approximately 50 MHz bandwidth), suitable for long-term frequency sweeping or long-distance detection scenarios, can achieve higher distance resolution and lower noise bandwidth.

[0074] Mode 2: Broadband mode (approximately 90 MHz bandwidth), suitable for fast scan or high refresh rate mode, to obtain greater IF bandwidth and faster target update rate.

[0075] Through the aforementioned reconfigurable IF link structure, the system can achieve a flexible trade-off between detection range, range resolution, and refresh rate, while maintaining a stable signal amplitude output.

[0076] After being amplified by the second variable gain amplifier 35, the IF signal is finally output from the system output port. In a preferred embodiment, the center frequency of the output intermediate frequency signal is approximately 375 MHz, and the output power can be stabilized at approximately 6–12 dBm, so that it can be directly fed into the subsequent analog-to-digital conversion and digital signal processing module for distance calculation and target detection.

[0077] The electronic control module includes a power supply module 61 and a control module 62. The control module 62 communicates with the host computer via a network port and controls various devices in the transceiver link internally via protocols such as SPI and I2C. For example, it controls the DAC module to change the sweep bandwidth and sweep period, controls the voltage-controlled oscillator synthesizer to switch frequency points, and controls signal switching switches 1 and 2 to switch the bandwidth of the output IF signal. The power supply module 61 is responsible for stepping down and filtering the externally input 24V DC power supply to generate the voltage required by each submodule of the system, thus meeting the power supply requirements of all modules in the whole machine.

[0078] The final test results of all indicators of the radar transmitting front end are shown in Table 1. It can be seen that the spurious components of the output intermediate frequency signal are suppressed to below −43.84 dBc, which verifies the effectiveness of the local oscillator spectrum planning and filter design and demonstrates the excellent spectrum performance of this embodiment.

[0079] Table 1 Performance Index Test Results

[0080]

[0081] The power of the terahertz signal ultimately transmitted by the transmission link is as follows: Figure 3 As shown, the output power is higher than 18.53 dBm throughout the frequency band, with a peak value of 19.36 dBm. The overall transmit power flatness is 0.83 dB, indicating that the radar front-end implemented in this embodiment achieves high output power and has good power consistency throughout the frequency band.

[0082] from Figure 4 (a) It can be seen that the measured saturated output power is 11.92 dBm, indicating that the IF amplifier can maintain linear operation even under maximum gain conditions; from Figure 4 (b) It can be seen that when the input signal power is -80.26dBm, the measured IF (intermediate frequency) output power is 5.1dBm, proving that the total gain of the entire receiving link is as high as 85.36dB.

[0083] Low phase noise refers to the phase noise of the final terahertz signal, but the final signal frequency is 144 GHz, and existing instruments cannot directly measure the phase noise of such a high-frequency signal. The measurement method used in this embodiment for acceptance testing is: first, measure the phase noise of the transmitted signal, such as... Figure 5 As shown, the phase noise is -115.45 dBc / Hz. Since the entire transmission link includes 12 times the frequency (6 times first and 2 times later), the phase noise will deteriorate by 20*log(12) = 21.58 dBc / Hz. Therefore, the final terahertz signal phase noise is -115.45 dBc / Hz + 20log(12) = -92.52 dBc / Hz@1kHz. This shows that the scheme in this embodiment achieves excellent low phase noise.

[0084] In another embodiment, the method for operating the radar front-end implemented in the foregoing embodiments includes the following steps:

[0085] S1) Power-on initialization: The electronic control module provides stable power to each stage of synthesizer, amplifier, frequency multiplier bias, LNA and VGA, and completes temperature-stable clock / reference source locking;

[0086] S2) Working sub-band selection: The chirped signal generator outputs the corresponding point frequency, with one set on the TX side and one set on the RX side, to select the working sub-band at the target center frequency point;

[0087] S3) Select sweep frequency mode: Set the chirp sweep frequency bandwidth and sweep frequency duration, optionally 600μs or 1200μs, and simultaneously switch the IF filter bandwidth path, optionally 90 MHz or 50 MHz;

[0088] S4) Transmit link output: after chirp is mixed with the point frequency to form an E-band drive signal, it is amplified by the E-band power amplifier chain to about 26.9 dBm and then drives the frequency multiplier to output a D-band terahertz signal, which is transmitted to the outside world through the waveguide to the horn antenna and then reflected by the parabolic reflector antenna and directed towards the detection target.

[0089] S5) Echo reception and downconversion: The terahertz signal reflected back by the target enters the RX link through the horn feeder, is first amplified by the LNA, and then mixed with the LO in the second harmonic mixer to obtain the IF beat frequency signal.

[0090] S6) IF Gain and Filtering: The IF signal is amplified by two stages of VGA and then limited by a dual-bandwidth bandpass filter before being output to the back-end acquisition / processing;

[0091] S7) Distance Calculation: The backend performs spectral analysis / FFT on the IF beat frequency, based on... and The corresponding relationship outputs target distance / velocity and other information. If velocity measurement is required, it is also necessary to combine multi-frame processing and Doppler analysis.

[0092] In another embodiment, such as Figure 6 As shown, a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier is constructed using a modular architecture. The modules from top to bottom are (the dimensions are length, width, and height):

[0093] 1) Launch arm 71 (85mm×10mm×117mm);

[0094] 2) Receiver arm 72 (120mm×10mm×117mm);

[0095] 3) Top cover plate 73 (182mm×142mm×5mm);

[0096] 4) Transmitter Module 100 (TX) (60mm×120mm×15mm)

[0097] 5) Receiver module 101 (RX) (60mm×120mm×15mm);

[0098] 6) Chirp generator 102 (160mm×120mm×23.5mm);

[0099] 7) Electrical control module 103 (160mm×120mm×16mm);

[0100] The horn antenna 11 and horn feed 27 are small components integrated on the transmitting arm 71 and the receiving arm 72, respectively, and are responsible for transitioning terahertz signals from the waveguide to the air / from the air to the waveguide.

[0101] All structures except for the transmitter arm 71 and receiver arm 72 are collectively referred to as the main structure, which is fixed to the upper cover plate from bottom to top by eight M4*7.5 through screws. The transmitter arm 71 and receiver arm 72 are individually fixed to the upper cover plate 73 by screws. The total thickness of the main structure can be obtained by adding the thicknesses of the four modules (15+15+23.5+16) to 69.5mm, and the overall dimensions are 160mm×120mm×69.5mm. The upper cover plate 73, transmitter arm 71, and receiver arm 72 are located on top of the main structure, with dimensions of 182mm×142mm×123mm. The overall transceiver dimensions are 182mm×142mm×192.5mm.

[0102] The modules are interconnected via coaxial cables, J30J connectors, and rectangular / bent waveguides. Waveguide interconnection is preferred in the terahertz high-frequency band to reduce losses.

[0103] In terms of mechanical structure, an aluminum alloy base plate / cavity structure is preferably used to support each module. The main body of the transceiver can be divided into four layers: the transmitting module 100 and the receiving module 101 occupy the first and second layers respectively, and the terahertz signal is transmitted between the two layers through a bent waveguide; the third layer houses the chirped signal generator 102; and the fourth layer houses the electronic control module 103. The empty space between the first and second layers is reserved for structures such as bent waveguides, coaxial cables, and power supply lines.

[0104] The embodiments described above merely illustrate specific implementation methods of this application, and while the descriptions are detailed and specific, they should not be construed as limiting the scope of protection of this application. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of the technical solution of this application, and these modifications and improvements all fall within the scope of protection of this application.

Claims

1. A D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, characterized in that, It includes a transmitting module, a receiving module, a chirp signal generator, and an electronic control module; The chirp signal generator is used to input the same chirp local oscillator signal to both the transmitting module and the receiving module. The transmission module is used to generate a D-band terahertz transmission signal based on the chirp local oscillator signal and transmit it to the detection target. The receiving module is used to receive and down-convert the signal reflected back from the detection target and the chirp local oscillator signal to obtain the IF beat frequency signal and output it. The electronic control module is used to communicate with the host computer and control the various devices in other modules through the communication protocol.

2. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 1, characterized in that, The chirped signal generator includes a frequency modulation (FM) signal link, which includes: a temperature-controlled crystal oscillator generating a stable reference frequency signal; the reference frequency signal is first split into two paths by a first power divider, one path enters the phase-locked loop (PLL) control circuit through a second power divider, and the other path enters the comb spectrum generator through a third power divider; then, it enters the digital-to-analog converter (DAC) through a first bandpass filter; the outputs of the PLL control circuit and the DAC are then mixed in an up-converter mixer 1 to obtain an FM signal; the FM signal passes through a second bandpass filter and enters the second power divider to be split into two chirp local oscillator signals, which are respectively sent to the transmitting module and the receiving module.

3. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 2, characterized in that, The chirped signal generator also includes a multi-point frequency signal link, which includes: one output signal of the third power divider generating multiple sets of point frequency signals via the first voltage-controlled oscillator synthesizer and sending them to the transmitting module; and one output signal of the second power divider generating multiple sets of point frequency signals via the second voltage-controlled oscillator synthesizer and sending them to the receiving module.

4. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 1, characterized in that, The transmitting module includes a transmitting local oscillator link, and the receiving module includes a receiving local oscillator circuit. The transmitting local oscillator circuit includes: the chirp local oscillator signal sequentially passes through a first 6x frequency multiplier, a second waveguide filter, and a second drive amplifier, and then is mixed with a point frequency signal in a second up-conversion mixer to form an E-band drive frequency band, and finally outputs through a second waveguide filter; the receiving local oscillator circuit includes: the chirp local oscillator signal sequentially passes through a second 6x frequency multiplier, a third waveguide filter, and a third drive amplifier, and then is mixed with a point frequency signal in a third up-conversion mixer to form an E-band drive frequency band, and finally outputs the transmitting local oscillator signal through a fourth waveguide filter.

5. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 4, characterized in that, The transmitting module also includes a transmitting link, which includes a cascaded power amplifier, a frequency multiplier, and an antenna connected in sequence. The transmitting local oscillator signal is sent to the input of the cascaded power amplifier and then transmitted to the detection target by the transmitting antenna.

6. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 5, characterized in that, The frequency multiplier adopts a frequency multiplication structure based on GaN Schottky diodes and combines it with a suspended microstrip transmission structure to achieve low-loss signal transmission. At the same time, it enhances local electric field coupling through resonant microstructures, thereby improving frequency conversion efficiency and output power.

7. The D-band FMCW radar transceiver front-end based on a high-power frequency multiplier according to claim 4, characterized in that, The receiving module further includes a receiving link, which includes: receiving a signal reflected from the target by a receiving feed, amplifying it by a low-noise amplifier, and then entering a second harmonic mixer to perform frequency down-conversion with the received local oscillator signal to obtain an IF beat frequency signal. The IF beat frequency signal passes sequentially through a third bandpass filter, a cascaded first variable gain amplifier, and a second variable gain amplifier before being output from the output port. A cascaded first signal switching switch and a second signal switching switch are provided between the first variable gain amplifier and the second variable gain amplifier. A fourth bandpass filter and a fifth bandpass filter are provided in parallel between the first signal switching switch and the second signal switching switch.

8. An operation method for a D-band FMCW radar transceiver front-end based on a high-power frequency multiplier, used in a D-band FMCW radar front-end based on a high-power frequency multiplier, characterized in that, include: Step 1: The electronic control module provides a stable power supply to each module and completes the temperature-stabilized clock and reference source locking; Step 2: The chirp signal generator outputs the corresponding point frequency signal to the transmitting module and the receiving module respectively, and the transmitting module and the receiving module select the target center frequency point working sub-band; Step 3: The chirp signal generator sets the chirp signal sweep bandwidth and sweep duration and synchronously switches the IF filter bandwidth path, and then sends the chirp signal to the transmitting module and the receiving module. Step 4: The transmitting module mixes the point frequency signal and the chirped signal to form an E-band drive signal. The E-band drive signal is then passed through a power amplifier and frequency multiplier in sequence to output a D-band terahertz signal, which is transmitted to the horn antenna through a waveguide and then reflected by the parabolic reflector antenna and directed toward the detection target. Step 5: The terahertz signal reflected back from the target enters the RX link through the horn feeder, is first amplified by the LNA, and then mixed with the LO in the second harmonic mixer to obtain the IF beat frequency signal. Step 6: The IF beat frequency signal is amplified by two stages of VGA and then band-limited by a dual-bandwidth bandpass filter before being output to the back end; Step 7: The backend calculates and analyzes the IF beat frequency to obtain the corresponding information.

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