A temperature compensation circuit and a bandpass filter

Abstract

This utility model relates to the field of microwave electronics technology, specifically a temperature compensation circuit and a bandpass filter, comprising a composite capacitor including a negative temperature coefficient (NTC) capacitor and a positive temperature coefficient (PTC) capacitor. The first terminals of the NTC capacitor and the PTC capacitor are connected together to a first port; the second terminals of the NTC capacitor and the PTC capacitor are connected together to a second port. By connecting the NTC capacitor and the PTC capacitor, which have opposite temperature characteristics, in parallel, a composite capacitor unit is formed. When the temperature changes, the capacitance of the NTC capacitor decreases, while the capacitance of the PTC capacitor increases or remains stable within a specific temperature range. The opposite trends of these two changes result in a total equivalent capacitance change that is much smaller than that of any single capacitor, directly compensating for the temperature drift of the lumped capacitor itself and reducing the impact of temperature on the filter's center frequency.

Description

Technical Field

[0001] This utility model relates to the field of microwave electronics technology, and in particular to a temperature compensation circuit and a bandpass filter. Background Technology

[0002] A bandpass filter is a key passive radio frequency device that allows signals within a specific frequency band to pass through while suppressing signals outside that band. It is widely used in wireless communication, radar, satellite navigation, and various electronic testing equipment, and its performance directly affects the signal quality, interference immunity, and reliability of the entire system.

[0003] One of the core metrics for evaluating the performance of a bandpass filter is the temperature stability of its passband frequency. Because the physical structures of the filter, such as the metal cavity and resonant pillars, as well as the internal capacitors and inductors, expand and contract with changes in ambient temperature, or experience parameter drift, the filter's center frequency and passband characteristics shift. This shift is unacceptable in high-precision applications.

[0004] To address the temperature drift problem, traditional methods primarily involve introducing compensation mechanisms externally or internally to the filter. Common approaches include using a single component with a specific temperature coefficient or adding a complex active compensation module with temperature sensors and control circuitry. However, these methods often face a dilemma: simple compensation schemes have limited effectiveness and cannot achieve accurate wide-temperature stability, while effective compensation schemes often make the circuit complex and expensive, and may introduce additional noise and fault points.

[0005] Therefore, a temperature compensation circuit and a bandpass filter are proposed. Utility Model Content

[0006] In this section, as well as in the abstract and title of this application, some simplifications or omissions may be made to avoid obscuring the purpose of this section, the abstract, and the title of this application. Such simplifications or omissions shall not be used to limit the scope of this utility model.

[0007] To address the shortcomings of existing technologies, one objective of this utility model is to provide a temperature compensation circuit.

[0008] To achieve the above objectives, this utility model adopts the following technical solution: a temperature compensation circuit, comprising a composite capacitor, which includes a negative temperature coefficient capacitor and a positive temperature coefficient capacitor; the first end of the negative temperature coefficient capacitor and the first end of the positive temperature coefficient capacitor are connected to a first port; the second end of the negative temperature coefficient capacitor and the second end of the positive temperature coefficient capacitor are connected to a second port; and a temperature compensation inductor is also included; the first end of the temperature compensation inductor is connected to the first port, and the second end of the temperature compensation inductor is connected to the second port, wherein the sum of the temperature coefficients of the composite capacitor and the temperature coefficients of the temperature compensation inductor is 0.

[0009] As a preferred embodiment of the temperature compensation circuit of this utility model, it further includes a varactor diode and an RF choke; the cathode of the varactor diode is connected to the first port, and the anode is connected to the second port; the first end of the RF choke is connected to the anode of the varactor diode, and the second end is led out as a bias port.

[0010] As a preferred embodiment of the temperature compensation circuit of this utility model, it further includes a DC blocking capacitor; the DC blocking capacitor is connected in series between the cathode of the varactor diode and the first port.

[0011] As a preferred embodiment of the temperature compensation circuit of this utility model, it further includes a microelectromechanical switch and a tuning capacitor; the microelectromechanical switch includes a common terminal, a first fixed terminal and a control terminal, the common terminal is connected to a first port, and the first fixed terminal is connected to a second port through the tuning capacitor.

[0012] In a preferred embodiment of the temperature compensation circuit described in this utility model, the microelectromechanical switch further includes a second fixed terminal, which is either suspended or connected to the second port via a damping resistor.

[0013] The temperature compensation circuit of this utility model has the following advantages: By connecting a negative temperature coefficient capacitor and a positive temperature coefficient capacitor with opposite temperature characteristics in parallel, a composite capacitor unit is formed. When the temperature changes, the capacitance of the negative temperature coefficient capacitor decreases, while the capacitance of the positive temperature coefficient capacitor increases or remains stable within a specific temperature range. The opposite trends of the two make the change in the total equivalent capacitance of the composite capacitor much smaller than the change in any single capacitor. This directly compensates for the temperature drift of the lumped capacitor itself, reducing the impact of temperature on the center frequency of the filter from the source. Only two capacitor elements and two ports are required, making the circuit structure extremely simple, easy to implement and integrate. The temperature compensation inductor and the composite capacitor form a parallel resonant circuit. The addition of the inductor has two main purposes: first, to utilize the resonant characteristics of the inductor and capacitor to create an operating point that is insensitive to parameter changes near the operating frequency, further improving frequency stability; second, to select an inductor with a specific temperature coefficient to achieve secondary temperature compensation of the circuit's resonant frequency, further improving the temperature stability of the circuit when operating at high frequencies, and more complex compensation curves can be achieved through the selection of the inductor.

[0014] To address the shortcomings of existing technologies, another objective of this invention is to provide a bandpass filter.

[0015] To achieve the above objectives, the present invention adopts the following technical solution: a bandpass filter, including the temperature compensation circuit, comprising a metal cavity, wherein a plurality of resonant pillars are disposed inside the metal cavity, the plurality of resonant pillars are arranged sequentially along the signal transmission direction, each resonant pillar has a loading end, each resonant pillar is electrically connected to a first port of a temperature compensation circuit through the loading end, and a second port of the temperature compensation circuit is electrically connected to the inner wall of the metal cavity.

[0016] In a preferred embodiment of the bandpass filter described in this utility model, the plurality of resonant pillars are arranged symmetrically about the central plane of the metal cavity.

[0017] In a preferred embodiment of the bandpass filter described in this utility model, the distance between the resonant pillar and the inner wall of the metal cavity decreases continuously from the center of the signal transmission direction towards both ends.

[0018] In a preferred embodiment of the bandpass filter of this utility model, the plurality of resonant pillars include a central equidistant section and two second decreasing sections. In the central equidistant section, the distance between adjacent resonant pillars and the inner sidewall is equal. In the second decreasing sections on both sides, the distance decreases gradually from the central equidistant section to both ends.

[0019] In a preferred embodiment of the bandpass filter described in this utility model, all components in the temperature compensation circuit are mounted on a circuit board, and the circuit board is fixed to the bottom wall of the metal cavity by conductive adhesive.

[0020] The beneficial effects of the bandpass filter of this utility model are the same as those of the temperature compensation circuit, and will not be described in detail here. Attached Figure Description

[0021] To more clearly illustrate the technical solutions of the embodiments of this utility model, the drawings used in the description of the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this utility model. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0022] Figure 1 This is a schematic diagram of the temperature compensation circuit of this utility model.

[0023] Figure 2 This is a schematic diagram of the overall temperature compensation circuit of this utility model.

[0024] Figure 3 This is a product diagram of the bandpass filter of this utility model.

[0025] Figure 4 The image shows the simulation test curves of the bandpass filter of this utility model.

[0026] Figure 5 This is a measured curve of the bandpass filter of this utility model during debugging.

[0027] Figure 6 This is a diagram showing the arrangement of the resonant pillars in Embodiment 4 of this utility model.

[0028] Figure 7 This is a diagram showing the arrangement of the resonant pillars in Embodiment 5 of this utility model.

[0029] In the diagram, A is the first decreasing segment; W is the equidistant middle segment; Y is the second decreasing segment; 100 is a composite capacitor; 101 is a negative temperature coefficient capacitor; 102 is a positive temperature coefficient capacitor; 103 is the first port; 104 is the second port; 200 is a temperature-compensated inductor; 300 is a varactor diode; 400 is an RF choke; 500 is a DC blocking capacitor; 600 is a microelectromechanical switch; 601 is a damping resistor; 700 is a tuning capacitor; 801 is a metal cavity; and 802 is a resonant pillar. Detailed Implementation

[0030] To make the objectives, features and advantages of this utility model more apparent and understandable, the specific embodiments of this utility model will be described in detail below with reference to the accompanying drawings.

[0031] Many specific details are set forth in the following description in order to provide a full understanding of the present invention. However, the present invention may also be implemented in other ways different from those described herein. Those skilled in the art can make similar extensions without departing from the spirit of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed below.

[0032] Secondly, the term "an embodiment" or "embodiment" as used herein refers to a specific feature, structure, or characteristic that may be included in at least one implementation of the present invention. The phrase "in one embodiment" appearing in different places in this specification does not necessarily refer to the same embodiment, nor is it a single or selective embodiment that excludes other embodiments.

[0033] Example 1

[0034] Reference Figure 1 This embodiment provides a temperature compensation circuit, including a composite capacitor 100, which includes a negative temperature coefficient capacitor 101 and a positive temperature coefficient capacitor 102. The first end of the negative temperature coefficient capacitor 101 and the first end of the positive temperature coefficient capacitor 102 are connected to a first port 103, and the second end of the negative temperature coefficient capacitor 101 and the second end of the positive temperature coefficient capacitor 102 are connected to a second port 104. A temperature compensation inductor 200 has its first end connected to the first port 103 and its second end connected to the second port 104. The winding inductance of the temperature compensation inductor 200 has a positive temperature coefficient, and its temperature coefficient value is greater than +50 ppm / ℃. The sum of the temperature coefficients of the composite capacitor 100 and the temperature compensation inductor 200 is 0.

[0035] By connecting a negative temperature coefficient capacitor 101 and a positive temperature coefficient capacitor 102 with opposite temperature characteristics in parallel, a composite capacitor 100 unit is formed. When the temperature changes, the capacitance of the negative temperature coefficient capacitor 101 decreases, while the capacitance of the positive temperature coefficient capacitor 102 increases or remains stable within a specific temperature range. The opposite trends of the two make the total equivalent capacitance of the composite capacitor 100 change much smaller than that of any single capacitor. This directly compensates for the temperature drift of the lumped capacitor itself, reducing the impact of temperature on the center frequency of the filter from the source. Only two capacitor elements and two ports are needed, making the circuit structure extremely simple, easy to implement and integrate. The temperature compensation inductor 200 and the composite capacitor 100 form a parallel resonant circuit. The inductor has two main purposes: first, to use the resonant characteristics of the inductor and capacitor to create an operating point that is insensitive to parameter changes near the operating frequency, further improving frequency stability; second, to select an inductor with a specific temperature coefficient to achieve secondary temperature compensation of the circuit resonant frequency, further improving the temperature stability of the circuit when operating at high frequencies, and to achieve more complex compensation curves through the selection of the inductor.

[0036] The operating point refers to the input impedance characteristic of the circuit. Without the temperature-compensated inductor 200, the circuit impedance is Zc = 1 / (jωC). When the temperature changes, the capacitance C changes, and Zc changes linearly and sensitively. After the temperature-compensated inductor 200 is connected in parallel, the circuit becomes an LC parallel resonant circuit. Its total impedance Z reaches its maximum value at the resonant frequency f0, and the phase is 0. This means that even if the temperature causes a small change in the inductance L and capacitance C, as long as the change does not push the resonant point too far, the circuit's performance (impedance magnitude and phase) at f0 remains almost unchanged.

[0037] The selection of a temperature coefficient greater than +50 ppm / ℃ is due to the following: Many precision inductors used at high frequencies (such as air-core coils and inductors with specific stable magnetic cores) may have very small temperature coefficients, within ±20 ppm / ℃. The temperature characteristics of such inductors are close to "stable" or "inert." If an inductor with a TC of only +10 ppm / ℃ is used in a circuit, the compensation effect it produces is negligible and can hardly offset the residual drift of the capacitor. From a technical point of view, it is not fundamentally different from using an inductor with a TC close to 0. Setting a value >50 ppm / ℃ is to exclude these conventional high-performance inductors with "insignificant compensation effects," thereby highlighting the special characteristics and technical contributions of the inductors selected in this solution. In engineering practice, an inductor with a temperature coefficient greater than +50 ppm / ℃ means that its magnetic core uses ferrite materials with strong positive temperature characteristics (such as certain nickel-zinc ferrite formulations). This numerical threshold precisely guides the protection range to this type of magnetic core material specifically selected for active temperature compensation, rather than all inductors.

[0038] Among them, the negative temperature coefficient capacitor 101 has a dielectric constant that decreases with increasing temperature, resulting in a decrease in capacitance. For example, the X7R material has a typical temperature coefficient of -(hundreds to thousands) ppm / ℃.

[0039] Positive temperature coefficient capacitor 102: Its dielectric constant increases with increasing temperature within a certain temperature range, or increases after remaining stable, resulting in an increase or stabilization of the capacitance value. For example, polypropylene film capacitors typically have a temperature coefficient of +100~200ppm / ℃.

[0040] In this context, the equivalent capacitance is represented by eq, the negative temperature coefficient capacitor 101 is represented by NTC, and the positive temperature coefficient capacitor 102 is represented by PTC.

[0041] When the negative temperature coefficient capacitor 101 and the positive temperature coefficient capacitor 102 are connected in parallel, their total equivalent capacitance C eq =C NTC +C PTC .

[0042] Assuming a temperature change of ΔT, the change in a single capacitor is: C NTC (T) = C NTC0 ×(1+α NTC ΔT), C PTC (T) = C PTC0 ×(1+α PTC ΔT), where α is the temperature coefficient (α NTC <0,α PTC >0).

[0043] The change in total capacitance is: ΔC eq =C NTC0α NTC ΔT+C PTC0 α PTC ΔT.

[0044] By carefully selecting C NTC0 and C PTC0 The initial capacitance ratio can make C NTC0 α NTC ≈−C PTC0 α PTC Under these conditions, ΔC eq Approaching zero means that although the capacitance of each capacitor changes drastically with temperature, the total equivalent capacitance of them connected in parallel remains highly stable over a wide temperature range.

[0045] In an LC resonant circuit, the temperature stability of the frequency requires the temperature coefficients of the inductor and capacitor to compensate for each other. Specifically, the overall resonant frequency of the circuit... The total equivalent capacitance C of inductor L and composite capacitor 100 eq Decide:

[0046] ,

[0047] Let the temperature coefficient of the composite capacitor 100 be... The temperature coefficient of the inductor is The temperature coefficient of frequency is then: .

[0048] in, The overall resonant frequency of the circuit. Let π be the mathematical constant, L be the inductance, and C be the inductance. eq Let d be the total equivalent capacitance of the composite capacitor 100, d be the differential operator, and T be the temperature. Indicates the composite capacitance value C eq The rate of change of temperature T, that is, the capacitance value with respect to a small change in temperature dT. The ratio of the small change in temperature dT multiplied by (The reciprocal of the initial equivalent value of the composite capacitor) is used to convert the absolute temperature change rate of the capacitor into the relative temperature change rate (the unit is 1 / ℃, which can also be converted into percentage % / ℃ or parts per million ppm / ℃). This represents the rate of change of inductance L with respect to temperature T. This represents the rate of change of the resonant frequency f with respect to temperature T. The temperature coefficient of the composite capacitor is 100. The temperature coefficient of inductance, The temperature coefficient of frequency.

[0049] In order to achieve temperature compensation (i.e. =0), must satisfy:

[0050] =0 or .

[0051] The relative rate of increase of the inductance value must exactly offset the relative rate of decrease of the capacitance value (or vice versa), and the two effects cancel each other out, thereby locking the resonant frequency unchanged.

[0052] When this composite capacitor 100 unit replaces the traditional single capacitor as the lumped loading element of the filter, its improved stability directly translates into improved stability of the filter's resonant frequency.

[0053] Furthermore, the dielectric material of the negative temperature coefficient capacitor 101 is X7R or X5R. Both X7R and X5R belong to Class II ceramic dielectrics, and their capacitance values ​​decrease significantly with increasing temperature above room temperature. For example, in the range of -55℃ to +125℃, the capacitance change rate of X7R can reach ±15%. This large-scale nonlinear change is precisely the "strong NTC signal source" needed. The dielectric constant (K value) of X7R is typically between 2000-4000, while X5R is slightly lower. This allows for the use of smaller packages (such as 0402, 0201) for the same capacitance requirement (e.g., 1nF), which is crucial for GHz band circuits (reducing parasitic inductance). Moreover, X7R / X5R are the most mainstream MLCC materials in consumer electronics and industrial fields, with huge production volumes, extremely low prices, and numerous suppliers. This ensures that the cost of the entire compensation circuit solution is controllable and easy to industrialize.

[0054] The dielectric material of the positive temperature coefficient capacitor 102 is polypropylene or polyphenylene sulfide. Polypropylene exhibits a stable, linear, and predictable positive temperature coefficient of capacitance change within its core operating range of -40°C to +85°C (or even higher), with typical values ​​of approximately +100 to +200 ppm / °C. This linearity and stability form the basis for calculations and proportioning compensation. Polyphenylene sulfide has similar temperature characteristics to polypropylene, also exhibiting a stable positive temperature coefficient, but the absolute value is typically slightly lower (approximately +50 to +150 ppm / °C), and it demonstrates excellent stability at higher temperatures.

[0055] Example 2

[0056] Reference Figure 2It also includes a varactor diode 300 and an RF choke 400. The cathode of the varactor diode 300 is connected to the first port 103, and the anode is connected to the second port 104. The first end of the RF choke 400 is connected to the anode of the varactor diode 300, and the second end is led out as a bias port. It adds an electrical tuning function on the basis of temperature compensation. The varactor diode 300 is connected in reverse parallel across the two ends of the circuit. Its junction capacitance changes with the reverse bias voltage. The RF choke 400 provides a DC bias path for the varactor diode 300 and at the same time prevents the RF signal from leaking into the bias circuit.

[0057] The varactor diode 300 acts as a voltage-controlled capacitor: it compensates for the center frequency dispersion of the filter caused by the manufacturing tolerance, so that each filter can be calibrated to the nominal value by electronic adjustment. In software radio, FM communication and other systems, it realizes real-time switching of filter channels. In extreme cases, it can be used as a manual or automatic fine-tuning tool to further correct residual temperature drift.

[0058] The RF choke 400 is a path for DC (low frequency), providing a stable DC bias current path for the varactor diode 300 to ensure accurate bias application. For RF (high frequency), it is a high impedance, and its high inductance prevents the main RF signal (GHz band) from leaking into the bias power supply and control circuit, avoiding two serious consequences: signal leakage leading to increased filter insertion loss and the possibility that the leaked RF signal may interfere with sensitive DC power supplies and control chips.

[0059] Furthermore, it also includes a DC blocking capacitor 500, which is connected in series between the cathode of the varactor diode 300 and the first port 103. The DC blocking capacitor 500 prevents the DC bias voltage from affecting the resonant pillar 802 or other sensitive parts of the preceding circuit, i.e. the filter, through the first port 103, thus ensuring the independence of the DC operating point.

[0060] Furthermore, it also includes a microelectromechanical switch 600 and a tuning capacitor 700. The microelectromechanical switch 600 includes a common terminal, a first fixed terminal, and a control terminal. The common terminal is connected to the first port 103, and the first fixed terminal is connected to the second port 104 through the tuning capacitor 700. The microelectromechanical switch 600 controls its on / off state through the signal of the control terminal, thereby connecting or disconnecting the tuning capacitor 700 from the circuit. Compared with analog tuning (varactor diode 300), digital tuning has the advantages of stable state, strong anti-interference, and low power consumption.

[0061] Furthermore, the microelectromechanical switch 600 also includes a second fixed terminal, which is either floating or connected to the second port 104 via a damping resistor 601. The addition of the second fixed terminal provides more switching state options, such as being designed to switch the circuit to different values ​​of the tuning capacitor 700, or to provide specific termination conditions such as grounding via the damping resistor 601 when disconnected, to improve circuit matching or suppress switching transients.

[0062] Example 3

[0063] Reference Figures 3-7 A bandpass filter is provided, including a temperature compensation circuit, a metal cavity 801, and a plurality of resonant pillars 802 disposed inside the metal cavity 801. The plurality of resonant pillars 802 are arranged sequentially along the signal transmission direction. Each resonant pillar 802 has a loading end. Each resonant pillar 802 is electrically connected to a first port 103 of a temperature compensation circuit through the loading end. The second port 104 of the temperature compensation circuit is electrically connected to the inner wall of the metal cavity 801.

[0064] By configuring a temperature compensation circuit independently for each resonant pillar 802, each resonant unit can be independently and accurately fine-tuned for frequency and temperature compensation. This distributed compensation architecture can ensure that the overall passband (center frequency, bandwidth, and in-band ripple) of the filter, which is determined by multiple resonant pillars 802, remains highly stable over a wide temperature range.

[0065] The second port 104 is connected to the inner wall of the metal cavity 801. In the filter, the metal cavity 801 is equivalent to the radio frequency ground / machine ground. Therefore, the second port 104 is grounded through the metal cavity 801.

[0066] Furthermore, the multiple resonant pillars 802 are symmetrically arranged about the central plane of the metal cavity 801. Due to the symmetrical arrangement of the resonant pillars 802, the electromagnetic field distribution and the coupling relationship between the resonators are mirror symmetrical. This directly and accurately realizes the symmetrical coupling coefficient sequence required by filter theory, thereby ensuring the electrical symmetry of the passband frequency response (such as in-band ripple and out-of-band suppression). At the same time, the symmetry of this physical structure allows any adjustment operation on one side (such as adjusting the tuning screw) to serve as a mirror reference for the corresponding position on the other side, greatly simplifying the adjustment process and improving the consistency and efficiency of design and production.

[0067] Furthermore, all components in the temperature compensation circuit are mounted on a circuit board, which is fixed to the bottom wall of the metal cavity 801 by conductive adhesive. The circuit board can be an FR4 epoxy board, a ceramic substrate, a flexible circuit board, or even a metal-based composite board.

[0068] To verify the performance of this invention, the 3.2–5.8 GHz filter was tested, and the results are as follows: Figure 5 (The test curves for the 3.2-5.8GHz filter are shown below:)

[0069] The X-axis represents frequency in GHz, the Y-axis represents the filter characteristic curve, and M1-M5 are the simulation settings for fixed frequency test values, namely 3.2GHz, 5.8GHz, 3.0GHz, 6.0GHz and 2.8GHz.

[0070] The red curve represents the S21 curve, which shows the filter insertion loss. The blue curve represents the S11 curve, which shows the filter input port reflected power. The red curves M1-M6 represent fixed test frequency points, with frequencies of 4.5GHz, 3.2GHz, 5.8GHz, 2.8GHz, 6.0GHz, and 3.0GHz respectively. Their test values ​​all meet the design specifications. The blue curves M1-M3 represent test values ​​at frequencies of 4.5GHz, 3.2GHz, and 5.8GHz, which also meet the design specifications.

[0071] Example 4

[0072] Reference Figure 6 The distance between the resonant pillar 802 and the inner wall of the metal cavity 801 decreases continuously from the center of the signal transmission direction to both ends. The multiple resonant pillars 802 form an "isosceles triangle", with the first decreasing segment A on both sides.

[0073] This arrangement method, through the deterministic physical relationship of "directly controlling the electromagnetic coupling strength by the monotonically decreasing mechanical distance," allows the non-uniform coupling coefficient sequence required by filter theory to be precisely pre-set as a manufacturable geometric gradient during the design phase. In this way, the core characteristics of the filter, such as bandwidth and in-band flatness, are "solidified" through the structure before manufacturing, ensuring a high degree of consistency between simulation and actual measurement, shortening the R&D cycle, and significantly reducing the reliance on human experience during debugging. As a result, a significant improvement in performance, efficiency, and consistency is achieved in mass production.

[0074] Example 5

[0075] Reference Figure 3 , 45. 7. The multiple resonant pillars 802 include a central equidistant segment W and two second decreasing segments Y. In the central equidistant segment W, the distance between adjacent resonant pillars 802 and the inner sidewall is equal. In the two second decreasing segments Y, the distance gradually decreases from the central equidistant segment W towards both ends. The central equidistant segment W corresponds to the weakest and most uniform coupling region, which is easy to process and ensures consistency. The two second decreasing segments Y enable a controllable increase in coupling strength. This "platform + slope" configuration... The combination of these two components represents the golden solution in engineering practice, achieving the best balance between performance, manufacturing difficulty, and cost. Its principle lies in the fact that it simultaneously satisfies the electrical requirements of the coupling coefficient distribution and the process constraints of the mechanical structure. Electrically, the equidistant middle section W provides uniform weak coupling, ensuring a flat response and stable group delay in the middle of the passband. The second decreasing section Y achieves a controllable increase in coupling strength towards the port to meet the requirements of input / output matching and out-of-band suppression. In terms of process, the platform section allows the use of repeated positioning references, significantly reducing manufacturing difficulty and the risk of tolerance accumulation. Although the second decreasing section Y needs to vary, it is also easy to control the manufacturing process because it is only located at both ends and the variation pattern is consistent.

[0076] The 802 resonant column has 17 orders. The parameters between the resonators are designed using coupling coefficients. The theoretical value of the coupling coefficient between the resonators is based on the formula Ki,i+1=W / W is the relative bandwidth of the bandpass filter, and gi and gi+1 are the normalized low-pass element values. That is, the coupling coefficients of the 17th stage of the filter are:

[0077] K1,2 = 0.529 = K16,17

[0078] K2,3 = 0.366 = K15,16

[0079] K3,4 = 0.336 = K14,15

[0080] K4,5 = 0.326 = K13,14

[0081] K5,6 = 0.321 = K12,13

[0082] K6,7 = 0.318 = K11,12

[0083] K7,8 = 0.317 = K10,11

[0084] K8,9 = 0.317 = K9,10.

[0085] The coupling coefficient is negatively correlated with the distance: the smaller the distance, the stronger the coupling of the edge field, and the larger the coupling coefficient; the larger the distance, the weaker the coupling, and the smaller the coupling coefficient.

[0086] The symmetrical gradient layout of the resonant pillar 802 precisely achieves the coefficient distribution of "strong coupling at both ends and weak coupling in the middle" required by filter theory. This directly solves the problems of passband performance degradation and port matching caused by edge effects and processing limitations. Through the built-in compensation in the structural design, while ensuring a flat frequency response and suppressing steepness, it significantly reduces the extreme dependence on processing accuracy and post-tuning, and improves the manufacturability and performance consistency of the design.

[0087] Compared with Example 4, this embodiment has the following advantages: because a central equidistant segment W is set, the distance between all adjacent resonant pillars 802 and the inner wall of the metal cavity 801 within this segment remains consistent. This means that the coupling coefficient remains uniform and weakest in this region, corresponding to the flat-topped passband region in filter theory. This is beneficial for achieving a flat in-band response and stable group delay. Only one repeated positioning reference is needed during processing, which greatly reduces the processing difficulty and tolerance accumulation risk, and improves the consistency between resonant units. During the debugging stage, the tuning rules of the central resonant pillars 802 are the same, which facilitates mass production and debugging, and reduces performance dispersion caused by structural asymmetry.

[0088] Meanwhile, a second decreasing segment Y is set, which gradually reduces the distance between the resonant pillar 802 and the cavity wall from the middle equidistant segment to both ends. This means that the coupling coefficient gradually increases from the center to the port, which conforms to the typical distribution of "strong coupling at both ends and weak coupling in the middle" in filter design. This is conducive to achieving good port matching and out-of-band suppression. At the same time, the decreasing change only occurs in the two end regions, rather than a continuous change throughout the entire process, which reduces the structural complexity and simulation optimization difficulty. In actual assembly, the decreasing segments at both ends can adopt a stepped or asymptotic layout, which makes it easy to achieve precise control of the coupling coefficient through a limited number of adjustment points (such as tuning screws) without having to fine-tune each resonant pillar 802 independently.

[0089] Because it adopts a segmented arrangement of "equidistant middle section + decreasing sections at both ends," this bandpass filter not only meets the theoretical requirements for coupling coefficient distribution in structure but also simplifies and standardizes the manufacturing process. This arrangement significantly improves the manufacturability, consistency, and reliability of the product while ensuring electrical performance (such as in-band flatness, out-of-band rejection, and port matching), making it more suitable for high-precision, mass-production microwave filter applications.

[0090] It should be noted that the above embodiments are only used to illustrate the technical solution of this utility model and are not intended to limit it. Although this utility model has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can be made to the technical solution of this utility model without departing from the spirit and scope of the technical solution of this utility model, and all such modifications or substitutions should be covered within the scope of the claims of this utility model.

Claims

1. A temperature compensation circuit, characterized by: include, A composite capacitor (100) includes a negative temperature coefficient capacitor (101) and a positive temperature coefficient capacitor (102). The first end of the negative temperature coefficient capacitor (101) and the first end of the positive temperature coefficient capacitor (102) are connected to the first port (103). The second end of the negative temperature coefficient capacitor (101) and the second end of the positive temperature coefficient capacitor (102) are connected together to the second port (104). A temperature-compensated inductor (200) has its first end connected to the first port (103) and its second end connected to the second port (104). The sum of the temperature coefficients of the composite capacitor (100) and the temperature-compensated inductor (200) is 0.

2. The temperature compensation circuit of claim 1, wherein: It also includes a varactor diode (300) and an RF choke (400); The cathode of the varactor diode (300) is connected to the first port (103), and the anode is connected to the second port (104). The first end of the radio frequency choke (400) is connected to the anode of the varactor diode (300), and the second end is led out as a bias port.

3. The temperature compensation circuit of claim 1 or 2, wherein: It also includes a DC blocking capacitor (500); The DC blocking capacitor (500) is connected in series between the cathode of the varactor diode (300) and the first port (103).

4. The temperature compensation circuit as described in claim 3, characterized in that: It also includes a microelectromechanical switch (600) and a tuning capacitor (700); The microelectromechanical switch (600) includes a common terminal, a first fixed terminal and a control terminal. The common terminal is connected to a first port (103), and the first fixed terminal is connected to a second port (104) through the tuning capacitor (700).

5. The temperature compensation circuit as described in claim 4, characterized in that: The microelectromechanical switch (600) also includes a second fixed terminal, which is either suspended or connected to the second port (104) via a damping resistor (601).

6. A bandpass filter comprising the temperature compensation circuit of claim 5, characterized in that: include, A metal cavity (801) is provided inside the metal cavity (801), and the multiple resonant pillars (802) are arranged sequentially along the signal transmission direction, and each resonant pillar (802) has a loading end; Each of the resonant pillars (802) is electrically connected to a first port (103) of a temperature compensation circuit via a loading end, and a second port (104) of the temperature compensation circuit is electrically connected to the inner wall of the metal cavity (801).

7. The bandpass filter as described in claim 6, characterized in that: The multiple resonant pillars (802) are arranged symmetrically about the central plane of the metal cavity (801).

8. The bandpass filter as described in claim 6 or 7, characterized in that: The distance between the resonant column (802) and the inner wall of the metal cavity (801) decreases continuously from the center of the signal transmission direction towards both ends.

9. The bandpass filter as described in claim 6 or 7, characterized in that: The plurality of resonant pillars (802) include a central equidistant segment (W) and two second decreasing segments (Y); In the central equidistant section (W), the distance between adjacent resonant pillars (802) and the inner sidewall is equal; In the second decreasing segment (Y) on both sides, the distance gradually decreases from the central equidistant segment (W) towards both ends.

10. The bandpass filter as described in claim 6 or 7, characterized in that: All components in the temperature compensation circuit are mounted on a circuit board, which is then bonded to the bottom wall of the metal cavity (801) with conductive adhesive.

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