A 10-gigabit ethernet port protection circuit

Abstract

The utility model relates to communication technical field, concretely is a kind of gigabit ethernet port protection circuit.The protection circuit includes: difference signal positive electrode circuit and difference signal negative electrode circuit, for forming signal transmission path;Difference impedance matching circuit is used to suppress signal reflection;Power filter network circuit is used to filter high-frequency noise and suppress common-mode interference.The circuit architecture design of the utility model systematically solves the core technical problem in the high-speed transmission of gigabit ethernet port.In the signal transmission level, two-stage resistance-capacitance isolation and dynamic impedance matching mechanism construct the parameter self-adaptive signal transmission path, through the impedance buffer of series resistance, the direct current isolation of coupling capacitor and the dynamic adjustment of difference pair network, effectively suppress signal reflection and direct current bias interference, ensure the integrity of 10Gbps high-frequency signal in long-distance transmission.

Description

Technical Field

[0001] This utility model relates to the field of communication technology, specifically to a 10 Gigabit Ethernet port protection circuit. Background Technology

[0002] With the rapid development of 5G, cloud computing, and data center technologies, 10 Gigabit Ethernet, with its high-speed characteristic of 10Gbps, has become the core interface of high-speed communication equipment. However, in practical applications, the complexity of high-frequency signal transmission poses a severe challenge to the reliability of interface circuits. Although traditional 10 Gigabit Ethernet port protection circuits can achieve basic signal transmission and overvoltage protection, they have technical bottlenecks in the coordinated design of high-frequency signal integrity maintenance, power supply noise coupling suppression, transient interference graded protection, and efficient PoE power conversion. Specifically, the impedance matching network of the differential signal path often adopts a fixed parameter design, which is difficult to dynamically adapt to impedance changes caused by factors such as cable attenuation and temperature drift, and is prone to high-frequency signal reflection; the power supply filtering and ground isolation schemes lack differentiated processing of noise characteristics on the chip side and the network port side, and the common-mode interference suppression effect is limited; the single configuration of electrostatic discharge and surge protection devices cannot meet the energy discharge requirements of instantaneous overvoltage at 10 Gigabit speeds, and the response timing of multi-level protection devices is prone to mismatch; the layout design and device selection of the PoE rectifier circuit do not fully consider high-frequency noise coupling, resulting in mutual constraints between power conversion efficiency and signal purity. Utility Model Content

[0003] This disclosure presents a 10 Gigabit Ethernet port protection circuit, which aims to overcome at least one of the defects existing in the prior art.

[0004] To achieve the above objectives, the technical solution disclosed in this utility model is as follows:

[0005] According to one aspect of this disclosure, a 10 Gigabit Ethernet port protection circuit is provided, the protection circuit comprising:

[0006] Differential signal positive and negative circuits are used to form a signal transmission path;

[0007] Differential impedance matching circuits are used to suppress signal reflections;

[0008] Power supply filter network circuit, used to filter out high-frequency noise and suppress common-mode interference;

[0009] Electrostatic discharge and surge protection circuits are used to absorb transient overvoltages;

[0010] The common-mode inductor is connected to the rectifier bridge circuit to provide the output voltage;

[0011] Signal ground and reference ground isolation circuit, used for isolation filtering.

[0012] Furthermore, the differential signal positive electrode circuit includes:

[0013] The PHY chip output node XGPHY_MDIAP_P0 is connected to one end of the series resistor 10GPR2, and the other end of the series resistor 10GPR2 is connected to the intermediate node XGE_AP_TSP0.

[0014] The intermediate node XGE_AP_TSP0 is connected to one end of the first coupling capacitor 10GPC22, and the other end of the first coupling capacitor 10GPC22 is connected to the intermediate node XGE_AP_TMP0.

[0015] The intermediate node XGE_AP_TMP0 is connected to the RJ45 interface node RJ45_AP_P0 through the second coupling capacitor 10GPC23 to form a signal transmission path;

[0016] The differential signal negative electrode circuit includes:

[0017] The PHY chip output node XGPHY_MDIAN_P0 is connected to one end of the series resistor 10GPR6, and the other end of the series resistor 10GPR6 is connected to the intermediate node XGE_AN_TSP0.

[0018] The intermediate node XGE_AN_TSP0 is connected to one end of the first coupling capacitor 10GPC26, and the other end of the first coupling capacitor 10GPC26 is connected to the intermediate node XGE_AN_TMP0.

[0019] The intermediate node XGE_AN_TMP0 is connected to the RJ45 interface node RJ45_AN_P0 via the second coupling capacitor 10GPC27.

[0020] Furthermore, the differential impedance matching circuit includes:

[0021] Impedance matching is achieved between the intermediate nodes XGE_AP_TSP0 and XGE_AN_TSP0 through differential pairs to suppress signal reflection;

[0022] The power supply filtering network circuit includes:

[0023] The chip-side power supply path is connected to signal ground (GND) through filter capacitors 10GPC14, 10GPC15 and 10GPC16 respectively, to filter out high-frequency noise;

[0024] The network port side signal ground (SGND) path is grounded through capacitors 10GPC17 and 10GPC35 to suppress common-mode interference.

[0025] Furthermore, the electrostatic discharge and surge protection circuit includes: the bidirectional TVS diode 10GPTVS1 connected between the intermediate node XGE_AP_TSP0 and the intermediate node XGE_AN_TSP0; and the bidirectional TVS diode 10GPTVS2 connected between the intermediate node XGE_AP_TMP0 and the intermediate node XGE_AN_TMP0, for absorbing transient overvoltages.

[0026] Furthermore, the common-mode inductor and rectifier bridge connection circuit includes:

[0027] The POE input node CT1 is coupled to the input terminal of the common-mode inductor PT1, and the output terminal of the common-mode inductor is connected to the rectifier bridge input node CMCT1;

[0028] The rectifier bridge includes Schottky diodes SD5, SD6, SD7, and SD8. CMCT1 is connected to the input terminal of the rectifier bridge through the Schottky diodes SD5 and SD6, and after rectification, it is output to the high voltage terminal VH and the low voltage terminal VL of the power supply.

[0029] The signal ground and reference ground isolation circuit includes:

[0030] The network port side signals RJ45_AP_P0 and RJ45_AN_P0 are connected to signal ground (SGND) through resistor 10GPR10 to achieve isolation and filtering between signal ground and system ground (GND). One end of resistor 10GPR10 is connected to the second coupling capacitor 10GPC27 of the RJ45_AN_P0 path, and the other end is connected to signal ground (SGND).

[0031] The beneficial effects of this utility model are:

[0032] The circuit architecture design of this invention systematically solves the core technical challenges in high-speed transmission of 10 Gigabit Ethernet ports. At the signal transmission level, a two-stage RC isolation and dynamic impedance matching mechanism constructs a parameter-adaptive signal transmission path. Through impedance buffering with series resistors, DC isolation with coupling capacitors, and dynamic adjustment of the differential pair network, signal reflection and DC bias interference are effectively suppressed, ensuring the integrity of 10Gbps high-frequency signals during long-distance transmission.

[0033] Furthermore, the composite filtering and ground isolation design addresses the noise characteristics of the chip side and the network port side by employing multilayer filter capacitors and common-mode rejection networks, combined with isolation filtering techniques for signal ground and system ground, significantly reducing the impact of power supply noise and ground loop interference on signal quality.

[0034] Furthermore, the PoE power module, through the optimized configuration of common-mode inductors and Schottky full-bridge rectifier circuits, improves power conversion efficiency while filtering out common-mode noise, meeting the high power requirements of the 802.3bt standard; the transient overvoltage protection scheme adopts a hierarchical combination of bidirectional TVS diodes and front-end protection devices, and achieves efficient absorption and clamping of transient interferences such as electrostatic discharge and surges by precisely controlling the response timing and energy distribution of the protection devices, raising the port protection level to an industry-leading level. Attached Figure Description

[0035] Figure 1 This is a flowchart of the 10 Gigabit Ethernet port signal processing method in this utility model;

[0036] Figure 2 This is a schematic diagram of the differential signal interface on the 10 Gigabit Ethernet port chip side of this utility model;

[0037] Figure 3 This is a schematic diagram of the filtering and common-mode rejection circuit of the POE power conversion module in this utility model;

[0038] Figure 4 This is a schematic diagram showing the pin definitions of the RJ45 interface differential signal and the T568B standard connection in this utility model;

[0039] Figure 5 This is a schematic diagram of the POE power rectifier bridge circuit in this utility model;

[0040] Figure 6 This is a schematic diagram of the dual RC isolation and protection circuit for PAIRA differential signals in this utility model;

[0041] Figure 7 This is a schematic diagram of the dual RC isolation and protection circuit for PAIR B differential signals in this utility model;

[0042] Figure 8 This is a schematic diagram of the dual RC isolation and protection circuit for PAIR C differential signals in this utility model;

[0043] Figure 9 This is a schematic diagram of the dual RC isolation and protection circuit for PAIR D differential signals in this utility model. Detailed Implementation

[0044] The technical solutions of the present utility model will be clearly and completely described below with reference to the accompanying drawings of the embodiments. Obviously, the described embodiments are only some embodiments of the present utility model, not all embodiments. Based on the embodiments of the present utility model, all other embodiments obtained by those skilled in the art without creative effort are within the protection scope of the present utility model.

[0045] The term "comprising," and any variations thereof, used in the specification and claims of this application, is intended to cover a non-exclusive inclusion. For example, a process, method, system, product, or apparatus that includes a series of steps or units is not necessarily limited to those explicitly listed, but may include other steps or units not explicitly listed or inherent to such process, method, product, or apparatus. Furthermore, the use of "and / or" in the specification and claims indicates at least one of the connected objects, such as A and / or B, indicating the inclusion of A alone, B alone, or both A and B.

[0046] In this embodiment of the invention, the terms "exemplary" or "for example" are used to indicate that something is an example, illustration, or description. Any embodiment or design described as "exemplary" or "for example" in this embodiment of the invention should not be construed as being more preferred or advantageous than other embodiments or designs. Specifically, the use of the terms "exemplary" or "for example" is intended to present the relevant concepts in a specific manner.

[0047] like Figures 1 to 9 As shown, the present invention provides the following preferred embodiments:

[0048] Example 1

[0049] To address the issues of signal integrity degradation, noise coupling interference, and insufficient power conversion efficiency faced by 10 Gigabit Ethernet ports in high-speed signal transmission, this embodiment provides a 10 Gigabit Ethernet port signal processing method. Through the coordinated design of signal transmission paths, impedance matching mechanisms, noise suppression schemes, power conversion circuits, and overvoltage protection links, reliable high-speed signal transmission and port protection are achieved. Figure 1 As shown, the process flow of the 10 Gigabit Ethernet port signal processing method is as follows:

[0050] S100: The positive terminal XGPHY_MDIAP_P0 and the negative terminal XGPHY_MDIAN_P0 of the differential signal output by the PHY chip are respectively buffered by series resistors to form a primary signal node. The primary signal node is DC isolated by the first coupling capacitor to generate an intermediate signal node, and then coupled to the RJ45 interface node by the second coupling capacitor.

[0051] S200: A differential impedance matching network is established between primary signal nodes. By acquiring the signal reflection intensity in real time, the resistance and capacitance parameters of the network are dynamically adjusted to stabilize the differential impedance at the standard value.

[0052] S300: Multi-layer filter capacitors are connected in parallel on the power supply path on the chip side to filter out high-frequency noise above 10GHz. A common-mode rejection capacitor network is set on the signal ground path on the network port side to attenuate common-mode interference signals. At the same time, the network port signal ground is electrically isolated from the system ground through isolation resistors to cut off the ground loop interference path and reduce the coupling noise between power supply and signal ground.

[0053] S400: The AC voltage signal input from POE is filtered to remove common-mode noise by a common-mode inductor and then input to a full-bridge rectifier circuit composed of Schottky diodes. By optimizing the diode layout and trace impedance of the rectifier bridge, the AC signal is converted into a standard DC voltage to reduce rectification losses.

[0054] S500: A bidirectional TVS diode is connected in parallel between the primary and intermediate nodes of the signal transmission path to clamp the instantaneous overvoltages caused by electrostatic discharge and surge impact to a safe threshold. At the same time, a pre-stage protection device is set between the RJ45 interface and the signal ground to form a multi-stage overvoltage protection link.

[0055] Specifically, in the signal transmission path construction stage, the positive terminal XGPHY_MDIAP_P0 and the negative terminal XGPHY_MDIAN_P0 of the differential signal output by the PHY chip are connected in series with a high-frequency, low-inductive resistor. The resistance value of this resistor is configured according to the characteristic impedance difference between the preceding and following circuits, aiming to perform preliminary impedance buffering of the 10 Gbps high-frequency signal and suppress reflection phenomena caused by impedance discontinuities during signal edge transitions. The signal after resistor buffering forms a primary signal node, which contains both DC bias and AC data components. To achieve DC component isolation, the primary signal node is connected to the input of a first coupling capacitor. This coupling capacitor is a thin-film capacitor with excellent high-frequency characteristics, and its capacitance value is selected based on the lowest operating frequency of the 10 Gbps signal to ensure low-loss transmission of AC signals at 10 Gbps, while simultaneously blocking the influence of DC bias on subsequent circuits, generating an intermediate signal node containing only the AC data component. The intermediate signal node is further coupled to the RJ45 interface node via a second coupling capacitor. The second coupling capacitor is positioned close to the RJ45 interface to shorten the transmission path of high-frequency signals and reduce the influence of parasitic parameters. The combination of the two-stage coupling capacitor and the series resistor forms a signal transmission path that combines impedance buffering and DC isolation, effectively matching the output impedance of the PHY chip with the characteristic impedance of the transmission cable, and suppressing signal baseline drift caused by DC bias.

[0056] Furthermore, a differential impedance matching network is constructed between the primary signal nodes. This network comprises a combination of adjustable resistors and fixed capacitors. The adjustable resistors are high-precision thin-film resistors, and their resistance values ​​can be dynamically adjusted by a microcontroller integrated on the motherboard. Signal reflection intensity is acquired through a directional coupler deployed on the RJ45 interface. This coupler extracts the power difference between the incident and reflected signals in real time, forming a feedback signal reflecting the impedance matching state. When the feedback signal indicates that the reflection intensity exceeds a preset threshold, the microcontroller triggers an impedance adjustment algorithm, adjusting the resistance parameters of the matching network using a successive approximation method to gradually bring the differential impedance closer to the 100Ω standard value. It is important to understand that the dynamic adjustment mechanism of the differential impedance matching network can compensate for impedance drift caused by factors such as temperature changes and cable aging in real time, ensuring that the high-frequency signal maintains a low reflection state during transmission, thereby improving signal integrity and reducing the bit error rate.

[0057] Furthermore, in terms of noise suppression and ground isolation design, a multilayer ceramic filter capacitor is connected in parallel along the chip-side power path. This capacitor array includes surface-mount capacitors of different capacitance values, which filter ultra-high frequency noise above 10GHz and intermediate frequency noise in the hundreds of megabits range, respectively. Its layout follows the principle of "dense arrangement near power pins" to filter out high-frequency noise coupling from the power port via the shortest path. The network port signal ground (SGND) path is configured with a common-mode rejection network composed of symmetrical capacitors. The capacitor values ​​are chosen to meet the high impedance characteristics for 10 Gigabit common-mode components, while presenting a low impedance path for power frequency and radio frequency common-mode interference, effectively attenuating common-mode interference signals introduced through cables. In addition, an isolation resistor electrically connects the network port signal ground to the system ground (GND). This resistor is a high-resistance, low-parasitic-inductance alloy resistor, designed to both cut off low-frequency interference paths formed by ground loops and provide a discharge path for high-frequency noise, thereby reducing coupling noise between power ground and signal ground and improving the system's electromagnetic compatibility.

[0058] Furthermore, to address the PoE power conversion requirements, the PoE input AC voltage signal first passes through a common-mode inductor PT1. This inductor uses a high-permeability core, which significantly suppresses common-mode noise while keeping the transmission loss of differential signals below 0.5dB. The signal, after common-mode filtering, is then input to a full-bridge rectifier circuit composed of four Schottky diodes. The selection of Schottky diodes emphasizes a balance between reverse recovery time and forward voltage drop, ensuring low-loss characteristics in high-frequency rectification scenarios. The rectifier bridge PCB layout employs a symmetrical trace design to reduce the difference in conduction impedance among the diodes and shorten the trace lengths at the input and output terminals, reducing the impact of parasitic inductance on rectification efficiency. Through this design, the PoE input AC signal is converted into DC voltages VH and VL conforming to the 802.3bt standard. The efficiency of this voltage conversion process can reach over 95%, meeting the high power requirements of the 10 Gigabit Ethernet port and subsequent circuits, while reducing heat accumulation caused by rectification losses.

[0059] Furthermore, in the transient overvoltage protection link construction, a bidirectional TVS diode is connected in parallel between the primary node of the signal transmission path and ground. This TVS diode has a response time of less than 1 ns and a clamping voltage threshold set at ±6kV, enabling it to quickly absorb the instantaneous energy generated by electrostatic discharge and surge impact, limiting the overvoltage amplitude within the safe operating range of the PHY chip. Further, a pre-stage protection device is connected in series between the RJ45 interface and signal ground. This device can employ a combination of a gas discharge tube and a varistor. The breakdown voltage of the gas discharge tube is set to 800V to discharge the main energy of the surge current, while the varistor provides secondary clamping for the residual voltage. The response timing of the two-stage protection devices is optimized through parameter matching to ensure that the gas discharge tube conducts first within nanoseconds, providing energy buffering for the TVS diode and preventing single device failure due to overload. This hierarchical protection architecture can effectively handle 8 / 20μs standard surge waveforms, achieving step-by-step discharge of differential-mode ±10kV and common-mode ±15kV transient interference, improving the long-term reliability of the port.

[0060] The advantage of this embodiment lies in its systematic design of key aspects such as signal transmission, impedance matching, noise suppression, power conversion, and overvoltage protection, which constructs a high-performance processing solution suitable for 10 Gigabit Ethernet ports. A two-stage RC isolation path effectively solves the problems of DC bias and impedance discontinuity; a dynamic impedance matching mechanism improves the stability of signal transmission; composite filtering and ground isolation technology reduces multi-source noise interference; a high-efficiency PoE rectifier circuit meets high power requirements; and multi-stage protection links enhance the port's anti-interference capability. These technical features, through reasonable layout design and parameter configuration, create a synergistic effect, providing a practical technical solution for the reliable operation of 10 Gigabit Ethernet ports in complex electromagnetic environments.

[0061] Example 2

[0062] To address the signal reflection problem caused by impedance drift in complex transmission environments, this embodiment provides a specific implementation method for the dynamic impedance matching adjustment stage. Through the coordinated design of real-time monitoring, adaptive adjustment, and stability control, the impedance consistency of differential signals during transmission is ensured.

[0063] In the signal reflection monitoring mechanism, a directional coupler is deployed at the RJ45 interface as a signal acquisition module. This coupler adopts a microstrip line coupling structure, capable of separating the incident and reflected signals with an insertion loss of less than 0.1dB. The reflected signals are then amplified by a low-noise amplifier and input to a high-speed analog-to-digital converter (ADC). The ADC has a sampling rate of at least 20GS / s, accurately capturing instantaneous power changes in a 10Gbps signal. The reflection intensity is obtained by calculating the ratio of the reflected signal power to the incident signal power. A preset threshold is set according to the return loss requirements of the IEEE 802.3ba protocol. When the reflection intensity exceeds the corresponding threshold, the subsequent adaptive adjustment process is triggered. It is important to understand that this real-time monitoring method based on the power ratio directly reflects the impedance matching status, providing accurate feedback signals for dynamic adjustment.

[0064] Furthermore, the differential matching network employs a parallel structure of adjustable resistors and adjustable capacitors. The adjustable resistor is a digital potentiometer (DCP) with a resistance adjustment resolution of 0.1Ω. The adjustable capacitor is a combination of a varactor diode and a fixed capacitor, with continuous capacitance adjustment achieved by applying a DC bias voltage. The adaptive adjustment mechanism is executed by a microprocessor integrated into the system control unit, using a successive approximation algorithm to optimize the matching network parameters: first, the capacitance value is fixed; then, the resistance value is adjusted using a binary method until the initial reflection intensity meets the target; finally, the capacitance value is fine-tuned to compensate for phase shifts in the high-frequency band, resulting in coordinated adjustment of the resistance and capacitance parameters. This step-by-step adjustment strategy can balance low-frequency impedance matching and high-frequency phase consistency, avoiding the limitations of single-parameter adjustment.

[0065] Furthermore, to suppress erroneous adjustments caused by environmental noise, this embodiment introduces a hysteresis comparison algorithm. A fluctuation range of 10% is set above and below the threshold. Parameter adjustment is initiated only when the reflection intensity continuously exceeds the upper fluctuation boundary by 500 μs. If, after adjustment, the reflection intensity falls below the lower fluctuation boundary and remains below it for 200 μs, the current parameter is considered stable. This dual criterion mechanism of time window and amplitude window effectively avoids frequent adjustments caused by instantaneous signal fluctuations, maintaining the stability of the impedance matching process. It should be noted that the parameter settings of the hysteresis comparison algorithm are matched to the symbol period of the 10 Gigabit signal to ensure that the adjustment action does not cause timing interference to data transmission.

[0066] Furthermore, at the hardware implementation level, the layout of the signal acquisition module and matching network follows high-speed circuit design specifications: the directional coupler is placed adjacent to the RJ45 interface to reduce parasitic parameters in the signal transmission path; the digital potentiometer and varactor diode are connected to the primary signal node via microstrip lines, with the trace length controlled within 500 mil and impedance compensation implemented. The control unit communicates with the adjustable device via the SPI bus, and the clock signal uses differential transmission mode to reduce electromagnetic interference. Through the above design, the return loss of the differential signal can be stably controlled below -20dB, and the eye diagram opening after transmission via 100 meters of CAT6a cable meets the protocol requirements, ensuring the integrity of high-speed signals during long-distance transmission.

[0067] The advantage of this embodiment is that by constructing a closed-loop system that includes real-time monitoring, adaptive adjustment, and stability control, it effectively solves the problem that traditional fixed matching networks cannot cope with dynamic impedance changes. The combination of a directional coupler and a high-speed ADC enables accurate acquisition of reflection intensity, and the combination of a successive approximation algorithm and a hysteresis comparison strategy improves adjustment accuracy while avoiding malfunctions. Optimized hardware layout further ensures the processing efficiency of high-frequency signals.

[0068] Example 3

[0069] To address the common-mode voltage drift problem caused by cable attenuation and electromagnetic interference during long-distance transmission of 10 Gigabit Ethernet signals, this embodiment provides a specific implementation method for adjustable bias voltage compensation. Through real-time monitoring of signal quality and dynamic voltage adjustment, it ensures that the common-mode component of the differential signal remains stable within the optimal operating range.

[0070] An adjustable bias circuit consisting of a voltage divider resistor network and a digital potentiometer (DCP) is installed between the intermediate signal node and system ground. The voltage divider resistor network uses high-precision thin-film resistors, whose fixed voltage division ratio converts the system power supply voltage into a basic bias voltage. The digital potentiometer is connected in series in the voltage divider branch, and its resistance value is changed to achieve fine adjustment of the output voltage. The signal quality monitoring module built into the PHY chip integrates a common-mode voltage detection unit and a noise power sensor. The former extracts the common-mode component of the received signal through a differential amplifier, while the latter uses a bandpass filter to separate noise signals in a specific frequency band and performs power integration, with monitoring accuracies of 0.01V and 0.5dBm, respectively.

[0071] Furthermore, when the monitoring module detects a common-mode voltage deviation of 1.2V ± 5% or a noise power exceeding -50dBm, it triggers the microcontroller to execute a compensation algorithm. The microcontroller first calculates the required voltage adjustment based on the degree of deviation, then uses a proportional-integral (PI) control algorithm to generate adjustment commands for the digital potentiometer, ensuring that each adjustment step does not exceed 0.1V to avoid the impact of voltage fluctuations on signal transmission. It's important to understand that this triggering mechanism, based on both common-mode voltage and noise criteria, can accurately identify signal baseline drift caused by cable attenuation and amplitude attenuation caused by external interference, enabling targeted compensation.

[0072] Furthermore, in terms of circuit structure design, a high-frequency ferrite bead is connected in series between the ground terminal of the adjustable bias circuit and the system ground to block radio frequency interference in the ground loop. Simultaneously, a 10nF ceramic capacitor is connected in parallel at the output terminal for high-frequency filtering to prevent voltage ripple during adjustment from affecting signal quality. The digital potentiometer uses non-volatile components and supports power-down memory function to ensure that optimal bias parameters are maintained after a system restart. The PHY chip communicates with the microcontroller via an I2C bus with a data transmission rate configured at 400kHz to meet the high-speed interaction requirements of real-time monitoring data.

[0073] Furthermore, the bias voltage adjustment range covers 0.8V to 1.6V to accommodate compensation requirements under varying cable lengths and environmental noise conditions. When the common-mode voltage is below 1.14V, the microcontroller controls the digital potentiometer to decrease its resistance, increasing the bias voltage to compensate for signal attenuation. When the common-mode voltage is above 1.26V or noise power exceeds the limit, the bias voltage is decreased by increasing its resistance, suppressing the impact of DC drift on the signal decision threshold. This bidirectional adjustment mechanism matches the AC coupling characteristics of 10 Gigabit signals, ensuring dynamic calibration of the common-mode component without affecting the AC component of the differential signal.

[0074] The advantage of this embodiment lies in constructing an adaptive long-distance transmission compensation system by combining the built-in monitoring function of the PHY chip with an external adjustable bias circuit. High-precision signal quality monitoring enables real-time quantization of common-mode voltage and noise, while voltage regulation based on a PI algorithm ensures the smoothness of the compensation process. The ferrite bead filtering and capacitor decoupling design enhance the anti-interference capability of the bias circuit itself. This solution effectively solves the problem that traditional fixed bias circuits cannot cope with environmental changes, ensuring the stability of the common-mode component of the differential signal over transmission distances exceeding 100 meters, and providing technical support for the reliable application of 10 Gigabit Ethernet ports in long-distance communication scenarios.

[0075] Example 4

[0076] To address the transient overvoltage threat faced by 10 Gigabit Ethernet ports in complex electromagnetic environments, this embodiment provides a specific implementation scheme for the graded protection of transient overvoltage. Through the coordinated design of high current discharge in the front stage and precise clamping in the back stage, a protection link with energy dissipation at each stage is constructed to improve the surge resistance of the port.

[0077] Between the RJ45 interface and signal ground, a gas discharge tube (GDT) and a varistor (MOV) are connected in series to form a pre-stage protection device. The gas discharge tube adopts a triode ceramic package structure, with a breakdown voltage set at 800V and a response time of less than 10ns, enabling it to conduct within nanoseconds and providing a low-impedance discharge path for surge current. The varistor is made of zinc oxide, with a nominal voltage matched to the residual voltage of the gas discharge tube. Its current carrying capacity is designed to withstand 8kA per pulse (8 / 20μs waveform), used to absorb the residual energy after the gas discharge tube conducts. The series connection of the two forms a composite protection unit of "voltage switching type + voltage limiting type", balancing high current discharge capability and voltage clamping accuracy.

[0078] When the instantaneous overvoltage peak exceeds the breakdown voltage of the gas discharge tube, the tube first ionizes and conducts, diverting more than 80% of the surge current to ground. Its dynamic resistance rapidly drops to the milliohm level after conduction, effectively suppressing the rate of overvoltage rise. The residual voltage is further clamped by a varistor, whose nonlinear characteristics stabilize the voltage across the terminals within the safe range that the PHY chip can withstand (typically ≤±6V). A bidirectional TVS diode is connected in parallel between the primary signal node and ground as a secondary protection device. Its response time is less than 1ns, enabling it to capture the instantaneous spikes in the response gap of the preceding protection devices, ensuring that the voltage transmitted to the PHY chip does not exceed the device's safe operating threshold.

[0079] Furthermore, to optimize the response timing of the protection devices, the discharge delay time of the gas discharge tube and the operating voltage of the varistor need to be matched: by adjusting the electrode spacing and charging pressure of the gas discharge tube, its breakdown voltage is made slightly lower than the varistor's start-up voltage, ensuring that the discharge tube conducts preferentially when an overvoltage occurs, preventing the varistor from aging and failing due to premature exposure to large currents. Simultaneously, in terms of PCB layout, the pre-stage protection devices are placed adjacent to the RJ45 interface, and the grounding pin uses wide copper foil directly connected to an independent grounding plane, shortening the surge current discharge path and reducing voltage bounce caused by grounding inductance.

[0080] Furthermore, the energy distribution strategy is optimized using simulation software. A surge impact model is established based on the Electromagnetic Transient Procedure (ATP-EMTP) to simulate the current distribution and voltage response of each stage of the device under an 8 / 20μs waveform. This ensures that the gas discharge tube undertakes the main discharge task, while the varistor and TVS diode handle residual energy and high-frequency components, respectively. This hierarchical protection architecture can withstand ±10kV differential-mode surges and ±15kV common-mode surges, meeting the highest protection level requirements of the IEC 61000-4-5 standard. At the same time, through reasonable configuration of device parameters, the insertion loss of the protection link is controlled within 0.3dB, avoiding adverse effects on 10 Gigabit signal transmission.

[0081] The advantage of this embodiment lies in achieving graded energy discharge and precise clamping of transient overvoltages by constructing a three-level protection system consisting of a gas discharge tube, a varistor, and a TVS diode. The high current discharge capability of the front-end devices effectively reduces the pressure on the subsequent protection stages, the timing matching design avoids response conflicts between devices, and simulation optimization ensures a balance between protection performance and signal integrity. This solution overcomes the limitations of traditional single protection devices in terms of energy handling capacity and response speed, providing highly reliable transient protection for 10 Gigabit Ethernet ports, enabling them to operate stably in harsh environments such as lightning strikes and industrial interference.

[0082] Example 5

[0083] To address the signal integrity and electromagnetic compatibility issues faced by 10 Gigabit Ethernet ports during high-speed signal transmission, this embodiment provides a 10 Gigabit Ethernet port protection circuit. Through the hierarchical design and coordinated layout of functional modules, an integrated protection system is constructed that includes signal transmission, impedance matching, noise suppression, overvoltage protection, and grounding isolation.

[0084] The protection circuit uses differential signal transmission as its core architecture, comprising symmetrically designed differential signal positive and negative circuits. These circuits form a complete signal transmission path through coupling capacitors and series resistors. A differential impedance matching circuit is integrated at the intermediate node of the signal path, employing a parallel network of adjustable resistors and fixed capacitors. Its impedance parameters match the characteristic impedance (100Ω±15%) of the CAT6a cable. By adjusting the equivalent impedance of the parallel network, the signal reflection coefficient is controlled below 0.1. It is important to understand that this symmetrical differential structure effectively suppresses common-mode noise and improves the anti-interference capability of signal transmission.

[0085] The power filtering network circuit is divided into two parts: the chip side and the network port side. On the chip side, the power path is connected in parallel with three layers of filter capacitors: 10nF, 100nF, and 1μF. These capacitors are used to filter out low-frequency noise below 10MHz, mid-frequency noise between 10-100MHz, and high-frequency noise above 100MHz, respectively. The capacitor layout follows the principle of "placing large-capacity capacitors near the power supply and small-capacity capacitors near the chip" to shorten the noise loop path. On the network port side, the signal ground (SGND) is connected to the system ground (GND) through a 1nF ceramic capacitor to form a common-mode noise discharge channel, suppressing common-mode interference introduced by cable coupling. The capacitor value is matched with the highest operating frequency of the 10 Gigabit signal (approximately 5GHz) to ensure noise suppression effect across the entire frequency band.

[0086] The electrostatic discharge (ESD) and surge protection circuit employs a multi-stage protection architecture: the front stage deploys a series combination of a gas discharge tube (GDT) and a varistor (MOV) to absorb kV-level transient overvoltage energy; the rear stage connects bidirectional TVS diodes in parallel between differential signal pairs, with a clamping voltage set at ±6V and a response time of less than 1ns, enabling rapid suppression of high-frequency spikes in residual voltage. The common-mode inductor and rectifier bridge connection circuit is used for PoE (Power over Ethernet) scenarios. The common-mode inductor adopts a dual-wire parallel winding structure with an inductance of 47μH, which can suppress common-mode current in the 10-100MHz frequency band. The rectifier bridge uses a Schottky diode array with a forward voltage drop of less than 0.5V, ensuring a power supply efficiency of no less than 90%.

[0087] The signal ground and reference ground isolation circuit connects the network port signal ground (SGND) and system ground (GND) through a 100Ω thin-film resistor, forming a grounding path with high-frequency isolation and low-frequency conduction. This avoids electromagnetic coupling interference caused by ground loops and provides a discharge path for electrostatic discharge current. During PCB layout, the differential signal pairs use microstrip lines with a 50Ω characteristic impedance. The line width and spacing are optimized using SI9000 software simulation to ensure differential impedance consistency. Protection devices are placed adjacent to the RJ45 interface, and the grounding pin is directly connected to the main ground plane through a 1mm diameter via, reducing the impact of grounding inductance on surge discharge.

[0088] The advantage of this embodiment lies in its integration of signal transmission, impedance matching, noise filtering, overvoltage protection, and grounding isolation functions into a unified circuit architecture, achieving multi-dimensional protection for the 10 Gigabit Ethernet port in high-speed transmission environments. The layered filtering network and symmetrical differential structure enhance signal integrity, the collaborative operation of multi-level protection devices strengthens transient overvoltage withstand capability, and the grounding isolation design effectively suppresses common-mode interference, providing systematic hardware support for the stable operation of the 10 Gigabit Ethernet port in complex electromagnetic environments.

[0089] Example 6

[0090] To address the issues of DC bias and impedance discontinuity during the transmission of 10 Gigabit Ethernet signals between the PHY chip and the RJ45 interface, this embodiment provides a refined implementation scheme for the specific connection relationship of the positive and negative circuits of the differential signal. By configuring the parameters of the series resistor and coupling capacitor, a reliable signal path that meets the characteristics of high-frequency signal transmission is constructed.

[0091] The signal transmission path of the differential signal positive circuit is as follows:

[0092] The PHY chip output node XGPHY_MDIAP_P0 is first connected to one end of the series resistor 10GPR2. This resistor is a 33Ω thin-film resistor in a 0402 package, used to match the PHY chip output impedance with the input impedance of the subsequent circuit, reducing signal reflection. The other end of the resistor is connected to the intermediate node XGE_AP_TSP0, which is connected to the intermediate node XGE_AP_TMP0 through the first coupling capacitor 10GPC22 (10nF capacitance, 50V withstand voltage). The coupling capacitor provides DC isolation, preventing the DC bias voltage of the PHY chip and the network port from interfering with each other, while allowing 10Gbps high-speed signals to pass through. XGE_AP_TMP0 is connected to the RJ45 interface node RJ45_AP_P0 through the second coupling capacitor 10GPC23 (same capacitance as the previous stage, made of NPO material with excellent high-frequency characteristics), completing the complete transmission link of the positive signal.

[0093] Furthermore, the differential signal negative terminal circuit adopts a symmetrical design. Specifically, the PHY chip output node XGPHY_MDIAN_P0 is connected to a series resistor 10GPR6 (with parameters identical to the positive resistor), which then passes through intermediate node XGE_AN_TSP0, the first coupling capacitor 10GPC26 (with electrical parameters identical to the positive capacitor), intermediate node XGE_AN_TMP0, and the second coupling capacitor 10GPC27, finally connecting to the RJ45 interface node RJ45_AN_P0. In other words, the symmetrical layout and consistent component parameter design of the positive and negative terminal circuits are crucial for ensuring differential signal phase synchronization and suppressing common-mode noise. The voltage rating and capacitance value of the coupling capacitors must simultaneously meet the requirements of DC bias isolation and high-frequency signal coupling efficiency.

[0094] Furthermore, in terms of signal integrity design, the series resistors are placed close to the PHY chip output pins, with a distance controlled within 2mm, to shorten the length of impedance discontinuities. The coupling capacitors adopt a surface-mount structure, with the pads directly connected to the signal traces, avoiding parasitic inductance introduced by vias. The traces from the RJ45 interface nodes to the PCB edge use differential pair routing, with a spacing of twice the trace width, ensuring the differential impedance remains stable at 100Ω±5%. In addition, the copper foil thickness of the intermediate nodes XGE_AP_TSP0 and XGE_AN_TSP0 is designed to be 35μm, meeting the skin effect requirements for 10Gbps signals and reducing high-frequency losses.

[0095] The advantage of this embodiment lies in resolving the DC bias conflict and impedance abrupt change issues in high-speed differential signal transmission by precisely setting the matching resistance value of the series resistor and the isolation characteristics of the coupling capacitor. The symmetrical circuit structure and strict layout rules ensure the phase consistency and amplitude balance of the differential signal, providing stable signal input conditions for subsequent impedance matching and noise suppression, and effectively improving the integrity guarantee capability of the 10 Gigabit Ethernet port at the beginning of the signal transmission link.

[0096] Example 7

[0097] To address the signal quality degradation issues caused by impedance mismatch and power supply noise in 10 Gigabit Ethernet ports, this embodiment provides a refined design scheme for the specific implementation of differential impedance matching circuits and power supply filtering networks. Through parameter optimization and structural configuration, it achieves synergistic optimization of signal reflection suppression and noise filtering.

[0098] Specifically, the differential impedance matching circuit is deployed between intermediate nodes XGE_AP_TSP0 and XGE_AN_TMP0, employing a "T-type" network structure. This means that an adjustable resistor (resistance range 20-180Ω, adjustment accuracy 0.5Ω) and a fixed capacitor (capacitance 10pF, ESR < 50mΩ) are connected in parallel between the two nodes. By adjusting the resistor value, the differential impedance is precisely matched to the cable characteristic impedance (100Ω), while the capacitor compensates for phase shift in the high-frequency band. In other words, the node position of this matching network is chosen after the coupling capacitor to avoid the influence of DC isolation devices on impedance adjustment. Its parameter adjustment is controlled by feedback from the signal quality monitoring module built into the PHY chip, forming a closed-loop adjustment system.

[0099] Furthermore, the power supply filtering network circuit is divided into two parts: the chip side and the network port side. Specifically, in the chip side power path, the filter capacitors 10GPC14 (1μF, electrolytic capacitor), 10GPC15 (100nF, ceramic capacitor), and 10GPC16 (10nF, high-frequency capacitor) are connected in a π-type configuration to filter out low-frequency ripple below 100kHz, intermediate-frequency noise from 100kHz to 10MHz, and high-frequency noise above 10MHz, respectively. The capacitors are installed according to the principle of "from large to small capacitance and from far to near" to ensure noise suppression across the entire frequency band. The network port side signal ground (SGND) path is grounded through 10GPC17 (1nF, feedthrough capacitor) and 10GPC35 (1nF, surface mount capacitor). The former is used to suppress common-mode noise above 1GHz, and the latter handles interference in the 100MHz-1GHz frequency band, forming a hierarchical filtering structure.

[0100] Furthermore, at the hardware implementation level, the adjustable resistors of the differential impedance matching network use digital potentiometers (DCPs) that support I2C interface control. The length of the traces connecting the pins to the intermediate nodes is controlled within 5mm to avoid introducing additional parasitic parameters due to excessively long traces. The grounding terminal of the power filter capacitor is connected to the main ground plane through an independent grounding copper foil with a width of not less than 3mm to reduce the grounding loop impedance. The signal traces and power paths adopt a layered design: the differential signal pairs are laid on the top layer of the PCB, and the power layer is located on the inner layer, isolated by a 20μm thick insulating medium to reduce the coupling interference of power supply noise on signal transmission.

[0101] The advantage of this embodiment lies in the combination of the dynamic adjustment capability of the differential impedance matching circuit and the hierarchical filtering characteristics of the power supply filter network, which effectively solves the problems of reflection loss and power supply noise in high-speed signal transmission. The optimization of the node positions and closed-loop control of the matching network improves the impedance adjustment accuracy, and the selection and layout design of the filter capacitors achieve full-band noise suppression, providing key hardware support for the stable operation of the 10 Gigabit Ethernet port and ensuring the integrity and purity of the signal in the transmission link.

[0102] Example 8

[0103] To address the risk of device damage to 10 Gigabit Ethernet ports under electrostatic discharge (ESD) and surge impact scenarios, this embodiment provides an optimized solution for the specific device configuration and connection relationship of the ESD and surge protection circuit. By differentiating the deployment of bidirectional TVS diodes, a precise clamping protection mechanism for high-frequency transient overvoltages is constructed.

[0104] Specifically, the protection circuit sets up two stages of bidirectional TVS diodes on the differential signal path. The first stage bidirectional TVS diode 10GPTVS1 is connected in parallel between the intermediate nodes XGE_AP_TSP0 and XGE_AN_TSP0. This node is located between the series resistor and the coupling capacitor at the output of the PHY chip. The breakdown voltage of the TVS diode is set to ±8V, and the response time is less than 500ps. It is used to clamp the common-mode and differential-mode transient voltages generated during signal transmission. The second stage bidirectional TVS diode 10GPTVS2 is connected between the intermediate nodes XGE_AP_TMP0 and XGE_AN_TMP0, close to the RJ45 interface. Its breakdown voltage is slightly lower than the previous stage (±6V), and its response time is less than 100ps. It can capture the instantaneous spikes of the action gap of the previous protection device and form a gradient clamping effect.

[0105] It's important to understand that the breakdown voltage gradient design of the two-stage TVS diodes (2V higher in the front stage than in the back stage) and their different placement (front stage closer to the chip side, back stage closer to the network port side) ensures the gradual absorption of overvoltage energy: when a transient overvoltage occurs, the back stage TVS diode closer to the network port activates before the front stage, clamping the voltage to a lower level and reducing energy transfer to subsequent circuits; if the overvoltage energy exceeds the processing capacity of the subsequent stage, the front stage TVS diode then conducts to share the remaining energy, preventing single device overload failure. The parasitic capacitance of the TVS diodes is controlled below 1pF (using silicon-based avalanche devices) to meet the high-frequency transmission requirements of 10Gbps signals and avoid signal distortion introduced by the capacitors.

[0106] In terms of PCB layout, the TVS diode leads feature a 45° chamfered design, with the pads directly connected to the signal traces to avoid impedance abrupt changes caused by right-angle bends. The ground pin is directly connected to a dedicated grounding island via a 0.5mm diameter via, which is connected to the system's main ground plane via a 1mm wide copper foil, forming a high-frequency, low-impedance discharge path. Furthermore, the signal trace length between the two TVS diode stages is controlled within 10mm to ensure that the time difference between the two stages of devices when the overvoltage wavefront arrives is less than the signal rise time (approximately 50ps), achieving a synchronous clamping effect.

[0107] The advantage of this embodiment lies in the construction of a dual protection barrier against high-frequency transient overvoltages through the gradient clamping design and precise layout of two-stage bidirectional TVS diodes. Differentiated breakdown voltage and response speed configurations achieve rational energy distribution and effective capture of instantaneous spikes. Strict control of parasitic parameters avoids adverse effects on high-speed signals, providing a reliable protection solution for 10 Gigabit Ethernet ports that balances protection effectiveness and signal integrity, enabling it to withstand standard ±15kV air discharge and ±8kV contact discharge tests without failure.

[0108] Example 9

[0109] To address the power stability and ground loop interference issues of 10 Gigabit Ethernet ports in PoE power supply scenarios, this embodiment provides specific implementation methods for the common-mode inductor and rectifier bridge connection circuit and the signal ground and reference ground isolation circuit. Through electromagnetic compatibility design and grounding strategy optimization, it ensures power supply efficiency and signal transmission independence.

[0110] Specifically, the common-mode inductor and rectifier bridge connection circuit constitute the core conversion unit for PoE input. That is, the PoE input node CT1 is connected to the input terminal of the common-mode inductor PT1. This inductor uses a ferrite core, has an inductance of 68μH, and a saturation current of no less than 1.5A, capable of suppressing common-mode noise in the 10-50MHz frequency band. The inductor output is connected to the rectifier bridge input node CMCT1. The rectifier bridge consists of a full-bridge structure composed of four Schottky diodes SD5-SD8, with a forward current of 5A and a reverse withstand voltage of 100V, ensuring stable operation in the Type 4 power supply mode of the 802.3bt standard. The rectifier bridge output generates a high-voltage terminal VH (48-54V) and a low-voltage terminal VL (GND). VH is output to the subsequent power conversion module through an LC filter circuit (10μH inductor + 100μF capacitor), with ripple voltage controlled within 100mV.

[0111] Furthermore, the signal ground and reference ground isolation circuit connects the network port signal ground (SGND) and system ground (GND) via a resistor 10GPR10 (1kΩ, 1% accuracy). This resistor is connected in series between the second coupling capacitor 10GPC27 and SGND in the RJ45_AN_P0 path, forming a high-frequency, high-impedance, low-frequency, low-impedance grounding path: for high-frequency components greater than 100MHz carried by the 10Gbps signal, the resistor exhibits high impedance, blocking high-frequency coupling interference in the ground loop; for low-frequency components such as electrostatic discharge and surge current, the resistor provides a low-impedance discharge path to ensure safe grounding. It is important to understand that the resistor value must balance isolation effect and discharge capability. Simulation verification shows that a 1kΩ resistor value can suppress ground loop interference in the 1GHz band by more than 30dB, while meeting the discharge requirements of 8 / 20μs surge current (voltage drop <50V).

[0112] Furthermore, in terms of electromagnetic compatibility design, the input and output terminals of the common-mode inductor are isolated by a metal shield, and the grounding pin of the shield is soldered to the system ground plane to reduce interference from leakage flux on the signal path. The diode layout of the rectifier bridge follows the "input-output separation" principle to avoid overlap between high-frequency current loops and signal traces. The isolation resistor between signal ground and reference ground uses a 0603 package, with a parasitic inductance of less than 5nH and a parasitic capacitance of less than 0.5pF, and its impact on the phase shift of 10 Gigabit signals is negligible. In the PCB stack-up design, a complete ground plane is set between the PoE power supply layer and the signal layer, connected by an array of more than 20 vias to reduce interlayer coupling noise.

[0113] The advantages of this embodiment are that the combination of common-mode inductor and rectifier bridge achieves efficient conversion and noise suppression of PoE input, while the signal ground isolation circuit effectively blocks ground loop interference, providing dual protection for the stable operation of the 10 Gigabit Ethernet port in PoE scenarios. Electromagnetic compatibility design and device parameter optimization ensure the independence of the power supply system and signal transmission system, meeting the high-power power supply requirements while avoiding power supply noise interference with high-speed signals, thus improving the applicability of the device in complex power supply environments.

[0114] Example 10

[0115] To address the signal quality degradation issue caused by channel variations during high-speed transmission of 10 Gigabit Ethernet ports, this embodiment provides a 10 Gigabit Ethernet port signal processing system. It offers an integrated implementation scheme for the functional architecture of the signal processing system, and constructs an intelligent processing system with adaptive adjustment capabilities through a closed-loop linkage mechanism of signal monitoring, logic control, and drive execution, ensuring signal integrity and transmission stability in complex environments.

[0116] The signal processing system is based on a three-layer architecture:

[0117] The signal quality monitoring unit employs the high-speed analog-to-digital converter (ADC) AD9625, whose 24-bit resolution and 25GS / s sampling rate meet the real-time acquisition requirements of 10 Gigabit signals (maximum frequency 5GHz). It can accurately capture signal amplitude (dynamic range ±5V), common-mode voltage (accuracy 0.1mV), and noise power spectrum (covering the 100MHz-10GHz frequency band) at the RJ45 interface. The analog front-end of the monitoring unit is equipped with a programmable gain amplifier (PGA) with a gain range of 0-60dB. The amplification factor is dynamically adjusted through an automatic gain control (AGC) algorithm to ensure that the ADC input signal is always within the optimal quantization range. It is important to understand that the front-end sample-and-hold circuit (S / H) of this unit adopts a differential input structure, matching the differential signal characteristics of the Ethernet port, which can suppress the impact of common-mode interference on monitoring accuracy.

[0118] The logic control unit implements a state machine algorithm based on a Field-Programmable Gate Array (FPGA). Preset signal quality thresholds include: signal eye diagram opening < 80%, common-mode voltage offset > 50mV, and noise power > -30dBm. When the monitored data triggers any of these thresholds, the state machine enters the adjustment process: first, it analyzes the noise spectrum using a Fast Fourier Transform (FFT) to determine the interference type (common-mode / differential-mode, narrowband / wideband), and then generates corresponding control commands—if the noise is due to impedance mismatch, it outputs an impedance adjustment command to the drive execution unit; if the signal offset is due to bias voltage drift, it generates a bias compensation command. The control logic adopts a pipelined architecture, with instruction processing latency controlled within 10ns, ensuring real-time synchronization between adjustment actions and signal changes.

[0119] The drive execution unit controls a digital potentiometer (DCP) and an adjustable capacitor array via an SPI interface to achieve fine adjustment of the matching network parameters. The digital potentiometer, an AD5290 (1024 taps, 0.1Ω resolution), is used to adjust the equivalent resistance of the differential impedance matching network, with an adjustment step of 0.2Ω to meet the impedance matching accuracy requirement of 100Ω ± 1%. The adjustable capacitor array consists of MEMS variable capacitors (capacitance range 1-10pF, accuracy 0.1pF) used to compensate for phase shifts in the high-frequency band. Its control voltage is provided by a dedicated DAC (16-bit, 50ns settling time). It is important to understand that the drive execution unit's adjustment actions follow a "resistor first, capacitor second" sequence to avoid adjustment oscillations caused by parameter coupling. Each adjustment cycle does not exceed 1μs to ensure that the continuity of signal transmission is not affected.

[0120] Furthermore, at the hardware design level, the system follows high-speed circuit layout specifications. Specifically, differential signal lines employ a 100Ω characteristic impedance design, with line length error controlled within 5mil (approximately 0.127mm), and spacing maintained at 3 times the line width (50μm line width corresponds to 150μm spacing) to reduce crosstalk and phase deviation. Matching networks and protection devices (such as TVS diodes and common-mode inductors) are placed adjacent to the RJ45 interface, with the shortest signal path length <20mm, reducing the impact of parasitic inductance and capacitance. The power module and signal module achieve physical isolation through a 4-layer PCB layout: the top layer is the signal layer, the middle two layers are the power plane and ground plane respectively, and the bottom layer is the shielding layer. The interlayer dielectric uses FR-4 material (dielectric constant 4.4, loss tangent 0.02). The power plane is connected to the ground plane through a 0.3mm diameter via array (2mm spacing), forming a low-impedance noise discharge channel.

[0121] The advantage of this embodiment lies in the fact that, through the deep integration of signal monitoring, logical decision-making, and hardware adjustment, a 10 Gigabit Ethernet signal processing system with autonomous optimization capabilities is constructed. The precise sampling of the high-speed ADC and the intelligent decision-making of the state machine algorithm enable dynamic evaluation of signal quality and accurate identification of interference types; the high-precision adjustment of digitally adjustable devices and strict hardware layout specifications ensure the real-time performance of parameter matching and the integrity of the signal path.

[0122] Although the present invention has been specifically described above with reference to preferred embodiments, it should be understood that the present invention is not limited to the embodiments described above. Rather, various modifications and variations can be made by those skilled in the art without departing from the essence of the present invention, and such modifications and variations should fall within the scope defined by the appended claims and their equivalents.

Claims

1. A 10 Gigabit Ethernet port protection circuit, characterized in that, The protection circuit includes: Differential signal positive and negative circuits are used to form a signal transmission path; Differential impedance matching circuits are used to suppress signal reflections; Power supply filter network circuit, used to filter out high-frequency noise and suppress common-mode interference; Electrostatic discharge and surge protection circuits are used to absorb transient overvoltages; The common-mode inductor is connected to the rectifier bridge circuit to provide the output voltage; Signal ground and reference ground isolation circuit, used for isolation filtering.

2. The Gigabit Ethernet port protection circuit of claim 1, wherein, The differential signal positive circuit includes: The PHY chip output node XGPHY_MDIAP_P0 is connected to one end of the series resistor 10GPR2, and the other end of the series resistor 10GPR2 is connected to the intermediate node XGE_AP_TSP0. The intermediate node XGE_AP_TSP0 is connected to one end of the first coupling capacitor 10GPC22, and the other end of the first coupling capacitor 10GPC22 is connected to the intermediate node XGE_AP_TMP0. The intermediate node XGE_AP_TMP0 is connected to the RJ45 interface node RJ45_AP_P0 through the second coupling capacitor 10GPC23 to form a signal transmission path; The differential signal negative electrode circuit includes: The PHY chip output node XGPHY_MDIAN_P0 is connected to one end of the series resistor 10GPR6, and the other end of the series resistor 10GPR6 is connected to the intermediate node XGE_AN_TSP0. The intermediate node XGE_AN_TSP0 is connected to one end of the first coupling capacitor 10GPC26, and the other end of the first coupling capacitor 10GPC26 is connected to the intermediate node XGE_AN_TMP0. The intermediate node XGE_AN_TMP0 is connected to the RJ45 interface node RJ45_AN_P0 via the second coupling capacitor 10GPC27.

3. The Gigabit Ethernet port protection circuit of claim 2, wherein, The differential impedance matching circuit includes: Impedance matching is achieved between the intermediate nodes XGE_AP_TSP0 and XGE_AN_TSP0 through differential pairs to suppress signal reflection; The power supply filtering network circuit includes: The chip-side power supply path is connected to signal ground (GND) through filter capacitors 10GPC14, 10GPC15 and 10GPC16 respectively, to filter out high-frequency noise; The network port side signal ground (SGND) path is grounded through capacitors 10GPC17 and 10GPC35 to suppress common-mode interference.

4. The Gigabit Ethernet port protection circuit of claim 2, wherein, The electrostatic discharge and surge protection circuit includes: a bidirectional TVS diode 10GPTVS1 connected between intermediate node XGE_AP_TSP0 and intermediate node XGE_AN_TSP0; and a bidirectional TVS diode 10GPTVS2 connected between intermediate node XGE_AP_TMP0 and intermediate node XGE_AN_TMP0, used to absorb transient overvoltages.

5. The Gigabit Ethernet port protection circuit of claim 2, wherein, The common-mode inductor and rectifier bridge connection circuit includes: The POE input node CT1 is coupled to the input terminal of the common-mode inductor PT1, and the output terminal of the common-mode inductor is connected to the rectifier bridge input node CMCT1; The rectifier bridge includes Schottky diodes SD5, SD6, SD7, and SD8. CMCT1 is connected to the input terminal of the rectifier bridge through the Schottky diodes SD5 and SD6, and after rectification, it is output to the high voltage terminal VH and the low voltage terminal VL of the power supply. The signal ground and reference ground isolation circuit includes: The network port side signals RJ45_AP_P0 and RJ45_AN_P0 are connected to signal ground (SGND) through resistor 10GPR10 to achieve isolation and filtering between signal ground and system ground (GND). One end of resistor 10GPR10 is connected to the second coupling capacitor 10GPC27 of the RJ45_AN_P0 path, and the other end is connected to signal ground (SGND).

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