Radar target with identification feature

Abstract

Radar target with identification feature, which is implemented as a passive retroreflector device for microwave-based monostatic radar systems, without a power supply required for its intended operation, - the signal frequencies emitted by the radar system vary cyclically and continuously in a frequency range between a low minimum frequency and a high maximum frequency, - whereby the receiving unit belonging to the radar system simultaneously evaluates both the frequency and the amplitude of the echo signal reflected from the radar target and feeds this information to a signal processing unit for further processing, which generates a digital identification signal from the respectively determined pattern of the frequency spectrum of the echo signal, - wherein the radar target has at least one retroreflector (1) for electromagnetic waves, in front of the aperture surface of which there is a coding filter (2) consisting of a partially periodic, alternating layer sequence of two dielectric materials each with a different permittivity, - wherein the respective layer thicknesses and the respective layer sequences are selected such that, during the manufacture of the coding filter (2), a specifically constructible frequency-selective transmission behavior is achieved, - depending on the arrangement and choice of layer thicknesses and layer sequences, only individual frequency ranges of the electromagnetic wave (7) emitted by the monostatic radar system and incident on the coding filter (2) can penetrate these layer sequences and thus reach the retroreflector (1), are retroreflected there and thereby penetrate the coding filter (2) again to be received and evaluated by the receiving unit of the radar system as an echo signal in the form of an electromagnetic wave (10) encoded by the frequency-selective transparency properties of the coding filter (2) fixed during the manufacture of the coding filter, - wherein the frequency-dependent pattern of the spectral profile impressed by the coding filter (2) serves as the identification feature of this radar target to be transmitted, characterized in that in addition to the retroreflector (1) and the coding filter (2) mounted in front of its aperture surface, outside the aperture surface of the retroreflector (1), a structure which corresponds to the coding filter (2) in terms of its reflection properties, as a compensator (3) in the same spatial orientation and in electrically effective spatial proximity but parallel to the direction of the normal (4) by an odd multiple of a quarter wavelength of the center frequency, which results as an arithmetic mean value, formed from the highest and the lowest design frequency of the respective coding filter (2), is mounted offset from the coding filter (2) and thus causes an additional reflection contribution of a wave (14) reflected from the surface of the compensator (3) upon incidence of an electromagnetic wave (7) in the direction of the normal (4),which, in terms of magnitude, almost coincides with the reflection contribution of a wave (13) reflected from the surface of the coding filter, originating directly from the coding filter (2), but is largely destructively phase-shifted relative to the latter, whereby the amplitudes of the wave (13) reflected from the surface of the coding filter (2) at normal incidence and the wave (14) reflected from the surface of the compensator (3) at normal incidence are significantly reduced by antiphase superposition.

Description

TECHNICAL FIELD OF THE INVENTIONThe invention relates to passive frequency-selective retroreflector devices for microwave-based monostatic radar systems, the reflection spectra of which have uniquely recognizable identification features.BACKGROUND OF THE INVENTION AND PRIOR ARTPassive retroreflective radar targets such as trihedral corner reflectors, Luneburg lenses, back metal coated spherical lenses, or Van Atta reflector arrays are known.By providing an object having a small radar cross section with such a passive retroreflective radar target, the radar cross section thereof can be significantly increased. This significantly improves its detection with the aid of a monostatic microwave-based radar system even over longer distances.Passive radar targets which do not require any auxiliary energy for their intended function are always preferred if they already sufficiently meet the respective requirements without additional electronic components, because of the freedom from maintenance, the reliability and the lower costs. However, the signal backscattered from the aforementioned conventional passive retroreflective radar targets is generally not provided with an individual identification feature which could be used for identifying different objects.Also known are so-called passive RFID tags which, without their own energy supply, respond to a high-frequency interrogation signal transmitted contactless by a reading device with a backscatter signal which contains an individual identification feature. However, such RFID tags are not retroreflective and can only be read out over relatively short distances of up to about 10 m for physical and regulatory reasons.In principle, a unique individual radar signature can be impressed on a passive retroreflective radar target, which can also be evaluated in stationary scenarios with the aid of a monostatic radar system, by virtue of its back-scattering behavior being designed specifically with regard to its frequency dependence, with regard to its time dependence or with regard to its polarization properties. Such solutions, which are based on a configuration of the polarization properties or those in which the configuration of the time dependence would not be equivalent to the configuration of the frequency dependence, such as, for example, so-called harmonic tags, are not used here for the problem solutions.In principle, technical solutions of this type in which a unique individual radar signature is impressed on a passive, retroreflective radar target by shaping the frequency dependence of its back-scattering behavior have already been known for a long time.In the publication DE000001279785A with the designation "System for the automatic wireless transmission of multi-digit information between mutually movable interrogators and responders, in particular the numbers of railway vehicles after stationary interrogators", an invention was already filed by Siemens AG in May 1966, in which "each interrogator transmits a periodically frequency-changing interrogation signal in a predetermined frequency band, from which the respectively moving responder selects the frequencies assigned to the information by filters and sends back a predetermined number of response frequencies to the interrogator for each location of the information". This invention is "characterized in that the interrogation signals of the interrogators have frequencies in the microwave range and the responders are provided with a high high-frequency line section to which the filters for the frequencies to be selected, which filters are primarily designed as stubs tuned to resonance, are coupled, and that the individual high-frequency line section is provided with at least one antenna, which is preferably designed as a directional antenna and is arranged on the respective responder such that the main maximum of its radiation pattern sweeps over its transmitting and receiving antenna, which is likewise designed as a directional antenna, when the responder moves past the interrogator."In an embodiment described in this publication, a interrogation signal continuously periodically changing its frequency is fed into a waveguide region at an input and is decoupled again at a distanced output. Within the waveguide connecting path between the input and the output are resonant circuits which act like frequency-selective filters and absorb the energy fed in at the input at the respective resonant frequencies, as a result of which those frequency ranges to which the resonant circuits are respectively tuned are coupled out at the output with reduced amplitude. As a result, a characteristic frequency pattern can be recognized for this arrangement, which can be interpreted, for example, as an identification feature for a digital password.In principle, such a method is suitable for enabling remote identification of an identification number with the aid of an interrogation device which transmits an interrogation signal periodically changing its frequency in a predetermined frequency band without additional electronics which must be supplied with the operating energy required for this purpose for proper operation.For short distances, both a targeted local feeding and a targeted local decoupling are possible and this method is also used exactly for this. For longer distances of a few meters and beyond, this is no longer possible, however, because the field density generated by the interrogation device can no longer be fed specifically only into the input of the waveguide arrangement, but at the same time illuminates both the input and the output, whereby clear signal recognition is no longer possible. It is also disadvantageous that the used waveguide components can be interconnected with one another with very low losses, but waveguides have relatively large geometric dimensions, are heavy in weight and are comparatively expensive. This applies both in terms of material costs and in terms of assembly costs.It is therefore obvious to dispense with connecting waveguides and to use only a single funnel-shaped retroreflector both as a receiving antenna and simultaneously as a transmitting antenna. Such implementations are also known and are implemented, for example, as trihedral retroreflectors. Likewise, solutions are known for modulating the radar beam emanating from a radar system by the retroreflector with a digital pattern before said beam is received by the radar system as an echo signal.The patent specification DE0000101046700B, with the designation "Modulated reflector in echo methods for radio locating or sound locating", with a filing date in October 1955, uses mechanical diaphragms to alternately either reflect or damp the radar beam between the radar system and the retroreflector and thereby generate an identifier. A "modulated reflector in echo methods for radio location or sound location" is claimed. This is "characterized in that a surface which reflects the wave energy well and predominantly absorbs it are arranged such that they are movable relative to one another in such a way that wave energy arriving from remote emitters is alternately reflected and absorbed when the surfaces move relative to one another."In the exemplary embodiments, arrangements of a plurality of trihedral retroreflectors are exclusively used which are overall geometrically larger than a single trihedral retroreflector.Added to this are the additional mechanical elements which modulate the radar beam as movable diaphragms. Overall, this invention is therefore not as well suited because it is bulky in comparison to the wavelength of the radar beam and cannot be constructed continuously without maintenance because of the movable mechanical diaphragms.A modulation of an electromagnetic beam reflected by a trihedral retroreflector without mechanical components is described in the publication U.S. Pat. No. 000906154299A with the designation "Modulating Reflector using Multiple Quantum Well Technology".In this case, a special coding filter with electronically controllable transmission behavior is mounted in front of the aperture surface of a trihedral retroreflector. This filter can transmit or reflect specific frequency ranges in a targeted manner by means of electronic pulses. As a result, serial data can be generated as amplitude-modulated light beams.This is therefore an electronically controllable filter disk mounted in front of the retroreflector.This is already very close to the method used with the invention presented here, but these special filter disks are currently available exclusively for electromagnetic beams in the range of infrared optical frequencies and therefore this method is currently not applicable to microwave-based systems.This is because the effects of quantum confinement occur in the filter material used only when the thickness of the quantum well of the heterostructure is in the nanometer range and therefore becomes comparable to the de-Broglie wavelength of the charge carriers in the semiconductor, because the charge carriers can only assume discrete energy values. Since the part of the heterostructure in the crystal that is active for the desired controllable quantum effect is only a few nanometers large, the structures are thus much too small for a suitable interaction with radar beams having wavelengths in the millimeter range.A further disadvantage is that an electronic circuit must always be used to effect modulation, as a result of which a permanent maintenance of operating energy is required.A number of other technologies are also known, such as activated micromirrors, electro-optical modulators, piezo-activated deflectors and liquid crystal modulators. All these theoretically conceivable solutions have in practice many advantages and disadvantages with respect to such features as power consumption, speed, modulation range, compactness, retroreflective divergence, cost and availability.In particular, in order to modulate the reflected beam with unique features, an additional electronic circuit is always required in these concepts, which requires an additional operating energy supply.Merely because of the need to provide an additional operating energy supply, these known solutions are not suitable for the object set here.As a further solution, the publication by Alejandro Jiménez-Sanez et al. entitled "Frequency Selective Surface Coded Reflectors for Chipless Indoor Localization Tag Landmarks", IEEE Antennas and Wireless Propagation Letters, vol. 19, no. 5, pp. 726-730, May 2020 proposes positioning a frequency-selective notch filter, consisting of a multiplicity of cross-shaped planar line resonators on a dielectric substrate as a coding filter directly in front of a retroreflector and continuously transmitting the reflected echoes of a suitable reading device which continuously frequency-variably transmits in the range from 65 GHz to 110 GHz, to be analyzed.By selecting different geometric dimensions of the resonant lines, these structures have respective self-resonances at different frequencies, at whose respective frequency detectable absorptions in the reflected frequency spectrum are detectable. In this publication, arrangements are thus described, consisting of a trihedral angle reflector, in front of the aperture surface of which a frequency-selective surface in the form of a periodically structured metallic layer is respectively arranged on a dielectric substrate, wherein crossed dipoles are used as unit cells.However, the frequency dependencies of the back scattering behavior shown in this publication do not show that a practical, reliable and unambiguously recognizable identification feature can be realized with this approach. Only by using a highly accurate measuring network analyzer, using horn radiators each with a gain of 25 dB and a previously calibrated, very short measurement distance of less than one meter, and aligning the surface normal of the structured substrate exactly in the direction of the horn radiator, it was possible to detect a frequency-selective absorption behavior by measurement technology.Unfortunately, when using a single horn radiator and a broadband measurement of the amount of the echo reflected back from the horn radiator, the respective notches expected at the resonant frequencies could no longer be unambiguously recognized in the large number of frequency-dependent power dips occurring during such free space measurements as a result of superimpositions of diffracted and reflected waves.Although the method described in this publication is not suitable for the object set forth here, it represents the current state of the art in this regard.SHORTCOMINGS OF THE PRIOR ART EMBODIMENTSIn the solutions known up to now, the devices are too bulky, too heavy or they use components that are unsuitable or not available for the respective frequency ranges, additionally use operating energy to be provided for electronic control circuits or do not generate significant signals that can also be clearly detected over longer distances of up to 100 m and beyond. In particular, those methods known hitherto which use coding filters and which manage without additional operating energy fail as soon as the angle of incidence is no longer perpendicular to the coding filter planes used in each case.DISCLOSURE OF THE INVENTIONIt is an object of the invention to improve the prior art under at least one of the following aspects:• Generation of a frequency-dependent, significant echo signal which can transmit information of at least one bit or more bits in predefined frequency ranges over an interval of angles of incidence and over a distance of up to 100 m and moreover to a reading device suitable for this purpose,• Generation of echo signals in the form of significant amplitude maxima, i.e. so-called peaks and expressly not as frequency-dependent amplitude minima, so-called notches, which are difficult to recognize and can be confused,• Realization in a compact, as small as possible and lightweight form,• Independence of any operating energy required for the intended function, and• Suppression of avoidable signal interference by means of measures for reducing reflections which could overlap the modulated echo signal without information content.The invention is defined in the claims. It allows in particular solutions as follows:Depending on requirements and application, a sufficient number of signal bits can be transmitted by frequency-selective echo signals over longer distances by suitable readers in the form of monostatic radar systems. The technology required for this is compatible, for example, with already available radar-based driver assistance systems in the microwave range. As a result, persons and objects from a relatively long distance can be detected by enlarging the radar cross section not only but can also be reliably and unambiguously identified or classified without an additional energy supply being necessary for this purpose for the marked persons or objects. This results in a permanent freedom from maintenance and also no charging or replacement of energy stores is necessary in order to maintain operation as intended. In this case, classification by the frequency range of the echo signals and identification by bit combination by further echo signals in the respective frequency ranges can be effected.Further details and advantages of the invention are evident from the following, purely exemplary and non-limiting description in conjunction with the respective figures.DESCRIPTION OF THE DRAWINGSFIG. 01 shows schematically in a side view a trihedral triangular corner reflector as a retroreflector (1), in front of the aperture surface of which there is a coding filter (2) and a compensator (3) laterally mounted thereto with an associated absorber (12). FIG. 02 shows schematically in plan view a trihedral corner reflector as a retroreflector (1) behind or below the coding filter (2), which has been assumed to be transparent here, and the compensator (3) positioned laterally thereto, which is assumed to be opaque here, as a result of which the absorber (12) located behind or below it cannot be seen. FIG. 03 shows schematically in a lateral view of the coding filter (2) with the input surface (5) and the output surface (6), an angle θ present with respect to the surface normal (4) and an incident wave (7), which is decisive both for the reflected wave (8) and for the transmitted wave (9). FIG. 04 shows, by way of example, schematically in side view the detailed structure of a coding filter, having regions of material with a high permittivity number, characterized by the letter "H" and having regions of material with a low permittivity number, characterized by the letter "L", wherein the layer thicknesses of the material with a high permittivity number are always of the same thickness and the layer thicknesses of the material with a low permittivity number are in places either twice as thick, marked by "L 2 ", or four times as thick, marked by "L 4 ", and therefore the coding filter has a periodic structure arranged in such a way that the coding filter has two pass regions in the transmission spectrum. FIG. 05 shows schematically, by way of example, in a side view the detailed structure of a coding filter which has three passbands in the transmission spectrum. FIG. 06 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 HLH. FIG. 07 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 2 HLH. FIG. 08 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 2 HLH. FIG. 09 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLH. FIG. 10 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 2 HLH. FIG. 11 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 2 HLH. FIG. 12 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 H(LH) 2 L 4 HLH. FIG. 13 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 H(LH) 2 L 2 HLH. FIG. 14 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 4 H(LH) 2 L 2 HLH. FIG. 15 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 2 H(LH) 2 L 2 HLH. FIG. 16 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 2 H(LH) 2 L 2 HLH. FIG. 17 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLHL 4 HLH. FIG. 18 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLHL 2 HLH. FIG. 19 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 4 HLHL 2 HLH. FIG. 20 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 2 HLHL 2 HLH. FIG. 21 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 2 HLHL 2 HLH. FIG. 22 schematically shows, by way of example, the change in the frequencies of the transmission maxima as a function of the magnitude of the angle of incidence θ and the polarization of the incident electromagnetic wave for a coding filter having three maxima. FIG. 23 shows schematically, by way of example for explanation, the profile of the propagation directions of the respective waves, namely the incident wave (7), the specular reflected wave (8), the transmitted wave (9) and the retro-reflected wave (10). FIG. 24 shows, by way of example, schematically in a lateral view, the device according to the invention with an absorber (12) for explaining the course of the propagation directions of the respective waves in the case of a perpendicular incidence of the incident wave (7), that is to say in the case where the angle of incidence θ is equal to zero, that is to say the retro-reflected wave (10), the wave (13) reflected by the coding filter (2) in the direction of the normal (4) in the case of a normal incidence of the wave (7), this wave being the mirror-reflected electromagnetic wave (8) for the special case where the angle of incidence θ is equal to zero, and the wave (14) reflected by the compensator (3) in the direction of the normal (4) in the case of a normal incidence of the wave. FIG. 25 shows schematically, by way of example, in plan view a possible arrangement of two retroreflector arrangements each having a coding filter (2) and (15) positioned in front thereof and having expediently different patterns in different frequency ranges having, for example, two or three maxima at different frequencies, as a result of which a digital signal having a plurality of bits can be transmitted remotely, and the respectively associated two compensators (3) and (16). FIG. 26 shows schematically, by way of example, in side view an arrangement according to FIG. 25, wherein, however, the normal (4) and the normal of a second arrangement (18) point in different directions in each case. FIG. 27 shows schematically by way of example a possible frequency pattern of an arrangement according to FIGS. 25 and 26, which can be generated with the aid of two coding filters, namely the coding filter ( 2) and the coding filter of a second arrangement ( 15).DESCRIPTION OF PREFERRED EMBODIMENTSFIG. 01 schematically shows a side view of a trihedral triangular corner reflector as retroreflector ( 1). In front of its aperture surface, which here is a triangular incident beam surface, which acts as the plane spanned by the three outer corner points of the trihedral triangular corner reflector, there is a coding filter ( 2) and a compensator ( 3) laterally attached thereto. The geometrical dimensions of the coding filter (2) and the compensator (3) are exactly the same. In particular, the heights or the material thicknesses of the two components are identical. Thus, the distance between the top and bottom of the encoding filter (2) is exactly the same as the distance between the top and bottom of the compensator (3).Below the compensator (3) is an absorber (12) made of a material which absorbs the electromagnetic waves that have passed through the compensator (3).Preferably, the compensator (3) is an exact copy of the laterally mounted coding filter (2). However, while the electromagnetic waves transmitted through the encoder filter (2) are retroreflected, the waves transmitted through the compensator (3) are absorbed and cannot cause interference.The coding filter ( 2) is located at a distance d 1 in front of the retroreflector ( 1). All sides of the coding filter project in geometric terms beyond the sides of the aperture surface formed by the edges of the boundary of the retroreflector ( 1) with a width of at least one quarter of the wavelength of the incident radar beam. This distance d1 is of little significance for the function of the device and can be selected such that it either completely disappears, i.e. d1=0.mu.m, which means that the coding filter rests directly on the outer metallic edge and thus on the aperture surface of the retroreflector (1) configured as a trihedral triangular corner reflector, or else that typically such a distance is selected which is less than a quarter of the wavelength of the incident radar beam in order to avoid disturbing diffraction or scattering effects on the metal edge of the aperture surface.Laterally of the coding filter (2) is a compensator (3) which has exactly the same geometrical dimensions as the coding filter and is positioned here in the exemplary embodiment laterally mirror-symmetrically at a distance d3. The respective normals (4) point in the same direction. The compensator (3) is made of a material whose surface has the same reflective properties as the surface of the coding filter (2), or the compensator (3) is preferably identical to the coding filter (2) in terms of material and structure in order to ensure a similar reflection behavior.The respective wave components used below for the discussion are shown in detail in FIGS. 23 and 24 and are also clearly denoted there.The following problem exists:For the special case of perpendicular irradiation, i.e. for an angle of incidence θ equal to zero, a part of the incident electromagnetic wave (7) of the radar system is already reflected directly from the surface of the coding filter (2), similar to in the optics for a planar mirror, and thus causes a wave (13) reflected from the surface of the coding filter (2) without having passed through the coding filter (2) beforehand. As a result, strong interference occurs during the evaluation of the returning coded electromagnetic wave ( 10), which has previously been spectrally marked by the coding filter ( 2), due to superposition or interference. This interference already occurs at small angles of incidence θ of approximately 10 degrees and becomes more severe the more precisely the radar target is aligned with the monostatic radar system and reaches a maximum at an angle of incidence θ of 0 degrees. In order to reduce these interference, the compensator (3) serves at the distance d2, as a result of which the difference (d2-d1) between the specular surface of the coding filter (2) on the one hand and the specular surface of the compensator (3) on the other hand arises with regard to the distance to the radar system. The distance d2 is the distance of the compensator (3) from the reference plane for the distance d1 according to FIG. 01.The distance d3 is also of little significance for the function of the device and can be selected such that it either completely disappears, i.e. d3=0 μm, which means that the compensator (3) is directly present at an outer edge of the coding filter (2), or else that typically a distance is selected which is less than half a wavelength of the incident radar beam.The distance d2 determines the difference in height (d2-d1) between the specular surface of the coding filter (2) and the specular surface of the compensator (3) with respect to the direction towards the incident radar beam.The distance d2 is of decisive importance for the device presented here! It is selected such that the two wave components reflected directly from the respective surfaces of the coding filter (2) on the one hand and of the absorber (12) on the other hand, which surfaces are at different heights, namely at a difference in distance (d2 - d1), cancel each other out. The respective dimensions of the distance d2 as a function of the distance d1 must therefore be observed as exactly as possible!The function of the compensator (3) in this arrangement is to mirror a radar beam directly reflected by the surface of the coding filter (2) likewise with its surface, wherein, however, owing to the different distance between the two surfaces, that is to say owing to the two distances d1 and d2, a difference in the propagation time results in the electromagnetic waves (13) and (14) directly reflected by the respective surfaces being canceled out, as a result of which these directly reflected waves (13) and (14) do not pass to the receiver input of the monostatic radar installation, or at least significantly reduced in amplitude.This is achieved by arranging the respective surfaces of the coding filter (2) and of the compensator (3) acting as mirror planes so offset at a distance that the respective directly mirrored electromagnetic wave components (13) and (14) cancel each other out in the direction of the radar system by the different phase positions.This is theoretically exactly possible with only a single frequency. However, if, as in this case, a larger frequency range is crossed over, the distances must be selected such that this applies exactly, for example, at the mean frequency of the selected frequency band and at least approximately applies at frequencies differing therefrom.In the normal incidence of an electromagnetic wave (7), there are thus three contributions to the overall reflected wave, i.e. firstly the contribution of the reflected wave (13) originating directly from the surface of the coding filter alone, then the contribution of the reflected wave (14) originating from the surface of the compensator, which contribution has an opposite phase position on account of the height offset (d2-d1) selected in a targeted manner, as a result of which both contributions ideally cancel out or at least largely compensate one another by superposition at exactly realized distances d1and d2. The remaining contribution to the overall reflected wave is then that contribution of the incident wave which, after passing through the coding filter, is reflected by the retroreflector (1) and, after passing once again through the coding filter (2), finally occurs as a spectrally coded retro-reflected wave (10).In FIG. 01, it is shown as an exemplary embodiment that the compensator ( 3) is positioned closer to the incident radar beam than the coding filter ( 2). The distance d 2 is therefore greater than the distance d 1. Of course, an arrangement would also be conceivable in which the distance d 2 is smaller than the distance d 1. However, it is important that the difference of the distance of the respective surfaces in the direction of the normal (4) is realized such that the respective non-spectrally encoded wave components (13) and (14) compensate each other. This is the case when the spatial maximum of the one wave component is superimposed with the in-phase spatial maximum of the other wave component, i.e. the spatial maximum shifted by 180°.The height offset (d2-d1) between the two mirror planes is thus set such that the respective mirror-reflected electromagnetic waves (13) and (14) of the two surfaces have as exactly as possible a phase difference of 180° and are thereby mutually cancelled out.This is achieved by the respective surfaces of the coding filter (2) and the compensator (3) acting as mirror planes differing by an odd multiple of a quarter wavelength of the radar beam.In FIG. 01, it is shown as an exemplary embodiment that the compensator ( 3) is positioned closer to the incident radar beam than the coding filter ( 2). The relationship would then apply to the device shown by way of example in this FIG. 01:In this case, the wavelength λ is preferably to be selected in accordance with the center frequency valid for the respective coding filter, referred to as "normalized frequency" in accordance with FIGS. 06 to 21, in order to achieve maximum effectiveness. The dimensions are therefore variably associated with the respective realization of a coding filter on the basis of the design and cannot be exactly named at this point.To get an approximate illustrative idea, reference is made to Fig. 27 which shows the frequency pattern of an arrangement of two coding filters with the associated compensator devices. There, the center frequencies of one arrangement are 73.5 GHz and those of the other arrangement 101.5 GHz. The associated wavelengths are thus 4082 μm and 2956 μm. At a distance d1=500 μm, the distance d2for the arrangement with the lower center frequency must then be 1520 μm, for example, and the distance d2for the arrangement with the higher center frequency must be 1239 μm, for example.As a result of the height difference selected in this way, the respective mirror-reflected electromagnetic waves (13) and (14) of the two surfaces have a phase difference of 180° and are thereby mutually canceled, whereas the retro-reflected coded electromagnetic wave (10) with unreduced amplitude is able to transmit the digital information back to the evaluation unit. A practical possible arrangement of a coding filter (2) and a compensator (3) is shown in FIG. 02 schematically in a plan view of a trihedral corner reflector behind the coding filter (2), which is shown transparently, and the compensator (3) positioned laterally thereto. The compact and expedient arrangement of the respective components of the device can be clearly seen. The exemplary embodiment presented hitherto shows an individual retroreflector ( 1) realized as a trihedral triangular angle reflector with a coding filter ( 2) positioned in front thereof and a compensator ( 3) positioned offset in height next thereto. For the purpose of illustration, the coding filter ( 2) is considered transparent in this figure, so that the contour of the retroreflector ( 1) situated underneath can be recognized. The compensator ( 2), on the other hand, is shown as non-transparent, so that the absorber ( 12) located underneath cannot be visualized. Of course, with these elements shown in FIG. 02, entire arrays can be designed by modules arranged in a suitable manner next to one another, as a result of which the quantity of information to be transmitted can be increased, because additional frequency marks can thereby be generated in the spectrum of the respectively reflected waves ( 10).For a detailed explanation of the mode of operation of the radar target according to the invention with identification feature, the coding filter ( 2), then the interaction of coding filter ( 2) and retroreflector ( 1), then the interaction of coding filter ( 2), retroreflector ( 1) and compensator ( 3) with absorber ( 12) are first considered below.The coding filter ( 2) used here is designed such that its transmission spectrum, i.e., the frequency dependence of its transmission factor with respect to an incident electromagnetic wave ( 7), represents a unique, individual identification feature within a selected operating frequency range. In particular, the transmission spectrum of a coding filter ( 2) has one or more passbands which, in the case of a plurality of passbands, are separated from one another by stopbands. When illuminating the radar target according to the invention with an identification feature with the radar beam of a monostatic radar system with a variable frequency range, individual echo signals, which are dependent on the respective coding filter ( 2) and are generated by the radar target according to the invention with an identification feature, can be detected only at discrete points of the frequency spectrum emitted by the radar system.For explanation, FIG. 03 is considered. This shows schematically in a lateral view of the coding filter (2) with the input surface (5) and the output surface (6) an angle θ present at an angle θ with respect to the surface normal (4) and an incident wave (7), which is decisive both for the reflected wave (8) and for the transmitted wave (9).Referring to FIG. 03, a pass-through region is characterized in that an approximately planar electromagnetic wave (7) incident on the input surface (5) of the coding filter (2) at an angle θ relative to the normal (4) predominantly exits again at this angle at the output surface (6) as a transmitted electromagnetic wave (9) and is only subject to a slight transmission attenuation in the process. In contrast, a frequency-dependent stopband is characterized in that said wave is subject to a large transmission attenuation at these frequency ranges and instead is predominantly converted into a mirrored wave (8), for example.A coding filter according to the invention according to the method presented here is formed from a multilayer dielectric layer structure which is composed of a plurality of planar layers of dielectric media, each of which has low dielectric losses and a linear behavior in electromagnetic terms, wherein the layer structure contains at least two dielectric media having different dielectric constants.Such or similarly periodic arrangements are known in very similar form as so-called photonic crystal structures which cause remarkable phenomena, particularly in connection with light. These structures are macroscopically simulated here for the microwave range.Such a coding filter is reciprocal in the sense that the same transmission spectrum is observed when the input surface (5) and the output surface (6) are interchanged. In addition, in the case of low dielectric losses of the dielectric media used, the transmission spectrum and the reflection spectrum are approximately complementary in the sense that the square of the magnitude of the transmission factor and the square of the magnitude of the reflection factor supplement one another approximately to one.The theoretical aids and design methods required for the configuration of the transmission spectrum of a multilayer planar layer structure are sufficiently known from applications in optics, so that the further description of the coding filters can be limited to the exemplary embodiments presented here by way of example. A large number of further embodiments are thus easily apparent to persons skilled in the art.In the preferred embodiment, the multilayer planar dielectric layer structure consists of a plurality of layers of a first dielectric medium having a lower dielectric constant, the thicknesses of which each correspond to a certain integer multiple of a quarter wavelength of an electromagnetic wave at a selected design frequency in just this first medium, and a plurality of identical layers of a second dielectric medium having a higher dielectric constant, the thickness of which corresponds to a quarter wavelength of an electromagnetic wave at the same design frequency in just this second medium. The individual transmission spectrum of a coding filter is then determined only by the individual sequence and number of said layers in the multilayer structure, which enables production by assembling similar basic components.For simplicity of illustration, a single layer of a first medium and a thickness corresponding to one quarter wave length at the design frequency in this lower dielectric medium will be referred to with the letter "L" hereinafter. Similarly, a single layer of a second medium having a thickness corresponding to one quarter of the wavelength in that higher dielectric medium at the same design frequency is referred to by the letter "H".Layer sequences of the form HLH, HLHLH or, more generally, of the form (HL) n H, i.e. an n-fold repetition of the layer sequence HL, followed by a layer H, realize an n-periodic so-called Bragg reflector. Layer sequences of the form LL, LLLL, or generally L 2m, each enclosed between two Bragg reflectors, realize a half-wave resonator, a full-wave resonator and an m-half-wave resonator, respectively. The enclosing Bragg reflectors may each have different numbers of periods.Overall, the particularly simple embodiments of coding filters shown by way of example below each consist of an alternating sequence of Bragg reflectors and resonators.The individual transmission spectrum of a coding filter is defined by the number "M" of the resonators, the numbers n 1, n 2 to n M+1, of the periods of the individual Bragg reflectors, the orders of the individual resonators, characterized by the parameters m 1, m 2, to m M and the quotient of the dielectric constants of the two media used, also referred to as dielectric contrast.The above-mentioned parameters primarily determine the number of passbands, the distances of the passbands on the frequency axis and the widths of the passbands.The absolute position of the transmission and blocking regions on the frequency axis relative to the design frequency also depends on the angle of incidence θ of the incident electromagnetic wave (7) and its polarization direction relative to the entry surface (5), as is shown by way of example in FIG. 22.The distinguishability of different encodings on the basis of the frequency dependence of the transmission factor is ensured by the selection of a suitable subset of all encodings that can be realized in this way.There are presented below, as a selection of possible examples, some variations of two different coding filters.FIG. 04 schematically shows, by way of example, in a side view, the detailed structure of a coding filter which has two passbands in the transmission spectrum. This exemplary embodiment shows a coding filter ( 2) consisting of a half-wave resonator, a full-wave resonator and three Bragg reflectors with n 1= 1, n 2= 2 and n 3= 1 in the layer sequence HLHL 2( HL) 2 HL 4 HLH. Its transmission spectrum has two passbands, combinations of which are shown in FIGS. 06 to 11.FIG. 05 schematically shows, by way of example, in a side view the detailed structure of a coding filter which has three passbands in the transmission spectrum. This embodiment shows a coding filter consisting of three half-wave resonators and three Bragg reflectors with n 1= 1, n 2= 2, n 3= 2 and n 4= 1. Its transmission spectrum has three passbands, combinations of which are shown in FIGS. 12 to 21.In order to clarify how a plurality of coding filters can be realized in the manner described above in such a way that they all have transmission spectra which can be unambiguously distinguished as individual identification means, the transmission spectra of a selection of 6 coding filters having in each case two passbands and a selection of 10 coding filters having in each case three passbands are reproduced by way of example in the following figures. The calculated results shown therein all relate to those exemplary embodiments in which the layers denoted by "L" have the dielectric constant ε r;L= 1 ( air) and the layers denoted by "H" have the complex dielectric constant ε r;H= 9,9 + j 0.004 (aluminum oxide ceramic). They also refer here to a design frequency in the range of 100 GHz and with a normal angle of incidence θ=0° of an incident electromagnetic wave ( 7). The dependence of the transmission spectra on the angle of incidence θ and the polarization is discussed further below. First, various possibilities for coding filters having two passbands in the transmission spectrum are presented.These coding filters consist of a half-wave resonator, a full-wave resonator and three Bragg reflectors.FIG. 06 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 HLH.FIG. 07 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 2 HLH.FIG. 08 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 2 HLH.FIG. 09 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLH.FIG. 10 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 2 HLH.FIG. 11 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 2 HLH.Various possibilities for coding filters which have three passbands in the transmission spectrum are now presented.These coding filters consist of three half-wave resonators and three Bragg reflectors with n 1= 1, n 2= 2, n 3= 2 and n 4= 1.FIG. 12 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 H(LH) 2 L 4 HLH.FIG. 13 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 4 H(LH) 2 L 2 HLH.FIG. 14 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 4 H(LH) 2 L 2 HLH.FIG. 15 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 H(LH) 2 L 2 H(LH) 2 L 2 HLH.FIG. 16 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 H(LH) 2 L 2 H(LH) 2 L 2 HLH.FIG. 17 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLHL 4 HLH.FIG. 18 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 4 HLHL 2 HLH,FIG. 19 shows, by way of example, the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 4 HLHL 2 HLH.FIG. 20 shows schematically by way of example the transmission spectrum of a coding filter with the layer sequence HLHL 4 HLHL 2 HLHL 2 HLH.FIG. 21 shows schematically by way of example the transmission spectrum of a coding filter with the layer sequence HLHL 2 HLHL 2 HLHL 2 HLH.With the ensemble of 16 different frequency-dependent transmission characteristics of coding filters presented by way of example from FIG. 06 to FIG. 21, a set of identification features which can likewise be unambiguously distinguished within a relative bandwidth of approximately 8% on the basis of their transmission spectra is available. These can be mapped, for example, to a 4-bit code.Persons skilled in the art will readily appreciate that using established methods of designing such filters, other, even larger, ensembles of coding filters with unambiguously distinguishable transmission spectra or those with other advantageous properties, such as broader passbands, can be realized in a similar manner.It is known that the transmission spectrum of a planar multilayer dielectric layer sequence has a dependence on the angle of incidence θ of the incident electromagnetic wave (7) with respect to the normal (4) and, to a lesser extent, on the polarization of the incident wave relative to the entry surface (5). These dependencies are theoretically well described and computationally predictable. They therefore do not present any impediment to the use of coding filters according to the invention.To illustrate these dependencies, FIG. 22 schematically illustrates, by way of example, the change in the frequencies of the transmission maxima as a function of the magnitude of the angle of incidence θ.This FIG. 22 shows, by way of example, the calculated typical frequency dependence of the transmission maxima of a coding filter on the angle of incidence and the polarization of a planar wave for a layer sequence HLHL 2 H(LH) 2 L 2 H(LH) 2 L 2 HLH with the dielectric constants ε r;L= 1 ( air), ε r;H= 9,9 + j 0.004 (aluminum ceramic) and for an average design frequency of 77 GHz. Three transmission maxima result at the frequencies 76.25 GHz, 77 GHz and 77.78 GHz for a perpendicular incidence of the radiated wave, that is to say for an amount of the angle of incidence θ equal to zero.In the case of non-perpendicular incidence, the transmission maxima of the arrangement used here by way of example for the calculation shift towards higher frequencies and their distances on the frequency axis change slightly depending on the polarization. They increase, in the case of the TM polarization, whose associated frequency-dependent profile is shown as a solid line, and decrease, in the case of the TE polarization, whose associated frequency-dependent profile is shown as a dashed line. Since each possible polarization of an incident wave is writable as a weighted superposition of a TM polarized wave and a TE polarized wave, the solid and dotted graphs in FIG. 22 represent the possible boundary cases, respectively.To explain further the interaction of the coding filter and the retroreflector, FIG. 23 serves.FIG. 23 shows schematically, by way of example for explanation, the profile of the propagation directions of the respective waves, namely the incident wave (7), the specular reflected wave (8), the transmitted wave (9) and the retro-reflected wave (10).A coding filter ( 2) with a retroreflector ( 1) mounted behind it is schematically shown, which here is shown by way of example as a trihedral triangular corner reflector. First, consider the case that the incident electromagnetic wave ( 7) is an approximately planar electromagnetic wave that is incident on the input surface ( 5) of the coding filter ( 2) at a finite angle of incidence θ with IθI≠0°. If the frequency of the incident electromagnetic wave (7) is within one of the passbands of the coding filter, the wave emerges again predominantly at the exit surface (6) of the coding filter (2) as a transmitted wave (9), is reflected back onto the coding filter (2) by the retroreflector (1) in the opposite propagation direction and, after renewed transmission through the coding filter (2), emerges again at the entry surface thereof as a retro-reflected wave (10) at the same angle as the angle of incidence θ.If, on the other hand, the frequency of the incident electromagnetic wave (7) is within one of the blocking ranges of the coding filter (2), then the latter is transferred directly from the coding filter (2) predominantly into a mirror-reflected wave (8) and only to a small extent into a transmitted wave (9).Overall, the monostatic backscatter cross section of the arrangement is proportional to the fourth power of the transmission factor amount of the coding filter ( 2) for angles of incidence θ with IθI≠0°.In the case of exactly normal incidence of the electromagnetic wave (7), that is to say for the case of angle of incidence θ with IθI=0°, on the other hand, the propagation direction of the retro-reflected wave (10) coincides with that of the wave (8) which is mirror-reflected directly by the coding filter (2).The desired frequency dependence of the backscatter cross section of the arrangement corresponding to the transmission spectrum of the coding filter (2) is covered in this case by the contribution of the mirror-reflected wave (8), whereby the reading out of the coding is strongly disturbed or even prevented.Depending on the distance between the radar target with identification feature and the associated monostatic radar system, this effect is also already disturbingly effective at small angles of incidence θ, i.e. close to zero.This limitation is remedied, as described below, by expanding the device by a compensator (3) together with an absorber (12), as will be explained below.FIG. 24 shows schematically, by way of example, in a side view, the entire device according to the invention with an absorber (12) necessary for perfect functioning in order to explain the course of the propagation directions of the respective waves, namely the incident wave (7), the retro-reflected wave (10), the wave (13) reflected by the coding filter (2) in the direction of the normal (4) in the case of normal incidence of the wave, that is to say at an angle of incidence θ of 0 degrees, and the wave (14) reflected by the compensator (3) in the direction of the normal (4) in the case of normal incidence of the wave.The above-mentioned problem that, in the event of an electromagnetic wave incident parallel to the common direction of the normal (4) of coding filter (2) and retroreflector (1), the frequency dependence of the amplitude of the retroreflected wave (10), which dependence is formed by the coding filter (2) together with the retroreflector (1), cannot always be determined unambiguously and reliably because significant contributions in the form of a mirror-reflected wave (8) are already generated by the surface of the coding filter (2), which contribute to interference due to superpositions, is solved in that the apparatus is extended by a compensator (3) together with an absorber (12).The compensator (3) according to the invention as claimed in claim 1 is a copy of the coding filter (2) which is as exact as possible, as a result of which its backscatter cross section corresponds in terms of magnitude, frequency dependence and polarization dependence to the respective corresponding variables of the coding filter (2) used.In this case, the specular surface of the compensator (3) is shifted with respect to the specular surface of the coding filter (2) with respect to the electromagnetic wave (7) incident at an angle of incidence θ=0 degrees in such a way that a phase difference results between the respective specular-reflected electromagnetic waves (8) emanating from the two surfaces, wherein, in the case of normal incidence, the phase position of the wave (14) reflected by the compensator (3) from the surface is shifted by 180° with respect to the wave (13) reflected directly from the surface of the coding filter (2).The contribution of the compensator (3) to the reflection of the overall arrangement thus compensates, in the case of normal incidence, the contribution which is already produced directly by the coding filter (2) by mirroring this contribution on its surface.In the preferred embodiment of the presented device, the compensator (3) shown in FIG. 24 consists of a copy of the coding filter (2) in the same spatial orientation with an absorber (12) for electromagnetic waves arranged behind it or below it. However, the compensator (3) is offset from the coding filter by an odd number multiple of a quarter wave length at the design frequency parallel to the common direction of the normal (4) of the coding filter (2) and the compensator (3).The absorber ( 12) is designed such that it has a high reflection attenuation in the intended operating frequency range. It can consist, for example, of commercially available foamed absorber material.For conceptual description of the relationships, the incident wave (7) and the various contributions to the reflected wave are visualized by arrows in Figure 24.In the event of an electromagnetic wave (7) incident parallel to the normal (4), three contributions to the reflected wave result, namely first the contribution (13) originating directly from the coding filter (2) solely by mirroring on the surface, then the contribution (14) originating from the compensator (3) likewise by mirroring on the surface, which contribution has an opposite phase position due to the offset in height and thereby compensates the former in the far field. The remaining contribution is that contribution of the incident wave (7) which, after passing through the coding filter (2), is reflected by the retroreflector (1) and, after passing once again through the coding filter (2), occurs as a retro-reflected wave (10). The frequency dependence of the retro-reflected contribution of the electromagnetic wave (10) can be influenced in the case of normal incidence by additional resonances which arise as a result of multiple reflections on the surface of the coding filter (2) facing the retroreflector (1). Due to the large dimension of the resonator forming in the spatial region between coding filter (2) and retroreflector (1) compared to the wavelength, the changes contributed by additional resonances are, however, very narrowband and do not interfere with the clear recognition of the identification feature given a suitable selection of the coding filters (2) used jointly in an application. In addition, the reflection spectra of the radar targets with identification features that are used together in an application can also be determined in advance by calculation or measurement even in the case of a wave that is incident parallel to the normal direction ( 4) and can therefore likewise be evaluated with respect to their correlation with the spectra of the radar targets with identification feature that are detected in use by means of a radar device.The coding filters presented in FIGS. 06 to 11 by way of example each have 2 significant signal maxima in the reflected frequency spectrum and the coding filters presented in FIGS. 12 to 21 even each have 3 significant signal maxima. As a result, for example, at least 3-bit coding could be realized in a specific frequency band.However, this is not the only possibility of coding. Depending on the frequency range in which the respective signal maxima can be detected, further information can be generated.For example, the frequency range from 65 GHz to 110 GHz can be divided into 9 frequency bands each having a width of 5 GHz.If, for example, a single significant signal maximum now appears exclusively at a frequency of 67.5 GHz, a bit sequence 100000000 could be derived therefrom. A single significant signal maximum at a frequency of 92.5 GHz would then result in the bit sequence 000001000, etc.A larger number of significant signal maxima is also conceivable. Thus, for example, two signal maxima could be generated simultaneously in the echo signal. If, for this purpose, the frequencies already mentioned above in the example are assumed to be 67.5 GHz and 92.5 GHz, then my bit sequence of 100001000 would result. Basically, such extensions of the information content are technically possible. However, this would require additional arrangements, depending on the intended number of additional significant bits, consisting of further coding filters (2), further retroreflectors (1), further compensators (3) and further absorbers (12), as are illustrated as a single arrangement in FIG. 01.The possibility of generating multiple signal maxima with a relatively small frequency interval with the aid of the coding filters increases both the possibility of a more complex interpretation of the frequency pattern and increases the reliability of the signal detection in the case of electromagnetically disturbed environmental conditions, in particular in the case of signal detection over relatively long distances and the signals received which are thus naturally weaker. For example, a twice significant echo signal as shown in Figs. 06 to 11 may be assigned to an automobile and a three times significant echo signal as shown in Figs. 12 to 21 may be assigned to a human person. Further classification possibilities result if, for example, the center frequencies of the triple significant echo signal are at different values.Thus, in the case where the center frequency of the triple significant echo signal is about 80 GHz, the information could be transmitted that it is a human person who is still a small child and, in the case where the center frequency of the triple significant echo signal is about 100 GHz, the information could be transmitted that it is a human person who has grown but is beta, etc.How these possibilities are used in the future is currently still unexplained. Individual arrangements according to FIG. 01 can already be produced so small for the frequency range considered here that they can be installed conveniently, for example, in a school broom. For future higher frequency bands, they would then have even significantly smaller dimensions.The smaller and the less expensive the arrangements according to FIG. 01 can be produced, the higher will be the acceptance for this possibility of increasing the safety in road traffic, in particular in autonomous road traffic.An additional simple possibility of transmitting further bit sequences with a higher number of bits is outlined below. However, as a result, the arrangements become more bulky and are then less suitable for persons than, for example, for use in vehicles or for the remote transmission of traffic signs or as beacons for orientation for autonomous vehicles.An exemplary extension with a second arrangement is discussed below. Naturally, extensions to an even higher number of arrangements are possible according to the same method, as is described here below by way of example for just a single further second arrangement.FIG. 25 shows schematically, by way of example, in plan view an arrangement of two retroreflectors each having a coding filter positioned in front thereof and having different patterns in the frequency spectrum at different frequencies, as a result of which a digital signal having additional bits can be transmitted remotely, and of two compensators having absorbers located beneath them.In contrast to the arrangement in FIG. 02, the coding filter ( 2) and the associated vertically offset compensator ( 3) of the first retroreflector devices are now arranged differently and supplemented by a second coding filter ( 15) and an associated second compensator ( 16) in the manner illustrated. This is one of numerous possibilities for supplementing the retroreflector devices, in this case by a further similar arrangement.The purpose of this modified arrangement is further illustrated by Figure 26.FIG. 26 shows schematically, by way of example, in a lateral partial view a part of the arrangement shown in FIG. 25 in the upper part of the figure with the coding filter (2) and the second compensator (16), wherein, however, the normal (4) and the normal of the further second arrangement (18) point in each case in slightly different directions. These normals with different directions for the respective two arrangements have the effect that possible problems due to directly mirrored electromagnetic waves are reduced even further.By using trihedral triangular angle reflectors as retroreflectors behind or under the coding filters, incident electromagnetic waves are also reflected back to the monostatic radar systems functioning as reading devices even if the respective angles of incidence θ vary in an angle range of approximately + / - 20 degrees.Therefore, the respective normals of the arrangements according to FIG. 25 discussed here by way of example, namely the normal (4) of a first arrangement, consisting inter alia of a coding filter (2) with an associated compensator (3) and the normal (18) of a second arrangement, consisting inter alia of a second coding filter (15) and a second compensator (16), can deviate from one another in their direction without a significant disadvantage with regard to the reflected echo signals thereby being caused. Rather, this deviation in the directions further reduces the probability of undesired interfering direct mirror reflection, because this would possibly prove to be relevant only for a single arrangement, depending on the orientation, but never simultaneously for two arrangements, whereby the information to be transmitted of the passive frequency-selective retroreflector devices with unambiguously recognizable identification feature is less strongly interfered with overall. The deviations of the directions of the respective normals (4) and (18) can be small in order to avoid undesired simultaneous direct mirror reflection of the respective two arrangements.In practice, an angular deviation of about 6 degrees at a reading distance of about 100 m between the passive frequency-selective retroreflector devices with a clearly recognizable identification feature and the microwave-based monostatic radar system initiating the echo signals will already prove to be very helpful. The distance d2 according to FIG. 01, which is decisive for the cancellation of the directly mirrored electromagnetic wave components, for example the wave component (13) caused by the coding filter (2) and the wave component (14) caused by the compensator (3), is expediently dependent on the center frequency of the respective frequency-dependent coding.In the case of a plurality of signal maxima at different center frequencies, the condition for optimum extinction cannot be fulfilled by a single distance d2which is the same for both different center frequencies.However, by using mutually diverging directions for the respective normals of the different arrangements, a distance d 2 exactly matching the respective correct center frequency range of the respective arrangement can be used. Due to the relatively wide opening angle of the trihedral corner reflector used here in this example, both arrangements nevertheless supply the desired coded echo signals back to the monostatic microwave-based radar system used here and the evaluation unit connected to it. Other arrangements can be easily supplemented by this approach if necessary or desirable.FIG. 27 schematically shows, by way of example, a possible frequency pattern of an arrangement according to FIGS. 25 and 26, which can be generated with the aid of two different coding filters, namely a coding filter ( 2) and a second coding filter ( 16).The respective center frequencies of the two frequency-dependent signal patterns are significantly different and therefore each require distances d2 which are matched thereto and can easily be realized in practice independently of one another.At present, such frequency patterns are not yet assigned any information. The respective interpretation of the respective frequency patterns and their decoding expediently requires standardization, before this method can be used in conjunction with an autonomous traffic situation in order to avoid undesired mis interpretations of the signals.ADVANTAGES ACHIEVED BY THE INVENTIONPortable retroreflective radar targets can be provided with identification features with a manageable technical outlay, which can be easily attached to persons or objects as compact, lightweight devices for marking and identification.The higher the operating frequencies of the radar systems used and the fewer digital bits have to be transmitted, the smaller the required dimensions of the respective radar targets with identification features can be. With an operating frequency of the radar system in the range of, for example, approximately 80 GHz and a digital information item to be transmitted of 3 bits, the entire device can be realized, for example, on a clip-on button having a diameter of approximately 5 cm. If these radar targets are illuminated with the aid of monostatic microwave-based radar systems, not only does the radar cross section appear to be significantly enlarged, but an identification feature can also be transmitted with suitable reading devices. The radar target does not require any operating energy for this purpose and is maintenance-free. In contrast to previous solutions, no frequency-dependent absorption patterns are generated which could easily be disturbed by superpositions and interference, but instead retro-reflections are carried out only at discrete frequencies.This means that, due to the presence of easily detectable, significant frequency patterns, characterized by maxima in the frequency profile, made of microwave signals in previously defined frequency ranges, digital information can be transmitted wirelessly over distances with substantially low interference.Additional measures according to the invention prevent disruptive superimpositions and thus significantly improve the interference safety. By selecting coding filters which simultaneously select a plurality of frequency ranges, the entire apparatus can be constructed in an extremely compact and cost-effective manner and, if appropriate, can be expanded with respect to the number of significant bits.LIST OF REFERENCE NUMERALS AND SIZE USED1 Retroreflector 2 Coding filter 3 Compensator 4 Normal 5 Entry surface of the coding filter 6 Exit surface of the coding filter 7 Incident electromagnetic wave 8 Mirror-reflected electromagnetic wave 9 Transmitted electromagnetic wave 10 Retro-reflected coded electromagnetic wave 11 Aperture surface of the retroreflector 12 Absorber 13 Wave 14 reflected in the case of normal incidence from the surface of the coding filter in the case of normal incidence from the wave 15 reflected in the case of normal incidence from the surface of the compensator Further coding filter 16 Further compensator 17 Further absorber 18 Normal of the further frequency-selective retroreflector device 19 Further retroreflector d 1 Distance between retroreflector (1) and coding filter (2) d 2 Distance of the compensator (3) from the reference plane for the distance d 1 d 3 Distance between coding filter (2) and compensator (3) H Identification of a layer with high permittivity L Identification a low-permittivity layer M number of the resonators in the periodic structure m M order of the respective resonators n natural number n M+1 number of periods of the individual Bragg reflectors ε r;L permittivity of the low-permittivity dielectric layer ε r;H permittivity of the high-permittivity dielectric layer θ angle of incidence with respect to the surface normal λ wavelength of the signal emitted from the radar apparatusNON-PATENT LITERATUREA. Jimenez-Saez, M. Schöeler, M. El-Absi, A. A. Abbas, K. Solbach, T. Kaiser, and R. Jacoby: "Frequency selective surface coded retroreflectors for chipless indoor localization tag landmarks," IEEE Antennas Wireless Propagate. Lett., vol. 19, no. 5, pp. 726-730, May 2020.

Claims

Radar target with an identification feature, which is realized as a passive retroreflector device for microwave-based monostatic radar systems, without an energy supply required for intended operation, - wherein the signal frequencies emitted by the radar system vary cyclically continuously in a frequency range between a low minimum frequency and a high maximum frequency, - wherein the receiving unit associated with the radar system simultaneously evaluates both the frequency and the amplitude of the echo signal reflected by the radar target and supplies this information for further processing to a signal processing unit which generates a digital identification signal from the respectively determined pattern of the frequency spectrum of the echo signal, - wherein the radar target has at least one retroreflector (1) for electromagnetic waves, in front of the aperture surface of which there is a coding filter (2) which consists of a partially periodic, alternating layer sequence of two dielectric materials each having a different permittivity, wherein the respective layer thicknesses and the respective layer sequences are selected such that during the production of the coding filter (2) a specifically configurable frequency-selective transmission behavior is effected, - wherein depending on the arrangement and selection of the layer thicknesses and the layer sequences only individual frequency ranges of the electromagnetic wave (7) emitted by the monostatic radar system and incident into the coding filter (2) can penetrate these layer sequences and thus enter the retroreflector (1), are retro-reflected there and in the process penetrate the coding filter (2) again in order to be received and evaluated by the receiving unit of the radar system as an echo signal in the form of an electromagnetic wave (10) encoded by the frequency-selective transparency properties of the coding filter (2) fixed during the production of the coding filter, - wherein the frequency-dependent pattern of the spectral profile impressed by the coding filter (2) is evaluated, as the identification feature of this radar target to be transmitted serves, characterized in that in addition to the retroreflector (1) and the coding filter (2) mounted in front of its aperture surface, a structure matching the coding filter (2) with respect to its reflection properties is mounted offset with respect to the coding filter (2) outside the aperture surface of the retroreflector (1), as a compensator (3) in the same spatial orientation and in electrically effective spatial proximity but parallel to the direction of the normal (4) by an odd multiple of a quarter wavelength of the center frequency, which results as an arithmetic mean value formed from the highest and the lowest design frequency of the respective coding filter (2), and thus brings about an additional reflection contribution of a wave (14) reflected from the surface of the compensator (3) upon the incidence of an electromagnetic wave (7) in the direction of the normal (4), which, according to its amount, almost corresponds to the reflection contribution of a wave (13) reflected by the surface of the coding filter (2) coming directly from the coding filter (2), but is largely destructively phase shifted with respect to the latter, whereby the amplitudes of the wave (13) reflected by the surface of the coding filter (2) in the case of normal incidence and of the wave (14) reflected by the surface of the compensator (3) in the case of normal incidence significantly decrease by means of counterphase superposition.Radar target with identification feature according to Claim 1, characterized in that, in addition to the retroreflector (1) and the coding filter (2) mounted in front of its aperture surface, a further retroreflector (19) is mounted, in front of its aperture surface a further coding filter (15) is located and, in addition to the compensator (3), a further compensator (16) is located, wherein the respective normals (4) of the planes of the coding filter (2) and of the associated compensator (3) on the one hand and the respective normals (18) of the coding filter (15) and of the associated compensator (16) on the other hand diverge in their direction from one another up to a value of 20 degrees.Radar target with identification feature according to Claim 1, characterized in that, in addition to the retroreflector (1) and the coding filter (2) mounted in front of its aperture surface, two or more further retroreflectors are mounted, in each case further coding filters are situated in front of their aperture surfaces and, in addition to the compensator (3), two or more further compensators are situated, wherein the respective normals (4) of the planes of the coding filter (2) and of the associated compensator (3), on the one hand, and the respective normals of the further coding filter and of the associated compensator, on the other hand, each diverge from one another in their direction up to a value of 20 degrees.

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