Demodulation device and program

Abstract

To suppress the degradation of BER characteristics in demodulation of an OFDM signal to which the ACE method is applied.SOLUTION: A demodulation device 300 according to the present disclosure includes an FFT unit 307 that performs an FFT on a time domain OFDM signal to convert it into a frequency domain carrier symbol, a transmission path equalization unit 311 that performs a transmission path equalization process on the frequency domain carrier symbol, a constellation clipping unit 317 that clips the carrier symbol after the transmission path equalization process to a certain range in the IQ signal space, a demapping / LLR calculation unit 318 that demaps the clipped carrier symbol and calculates a log-likelihood ratio for each bit, and an LDPC decoding unit 321 that performs error correction on the basis of the log-likelihood ratio for each bit.SELECTED DRAWING: Figure 26

Description

[Technical field]

[0001] The present invention relates to a demodulation device and program for terrestrial digital broadcasting that uses the Orthogonal Frequency Division Multiplexing (OFDM) system. [Background technology]

[0002] In recent years, terrestrial digital broadcasting has adopted the OFDM method, which is resistant to multipath interference, as a transmission method. OFDM signals for terrestrial digital broadcasting are composed of a very large number of mutually orthogonal subcarriers, ranging from several thousand to tens of thousands, and have a rectangular spectrum within the transmission band. On the other hand, the time waveform of an OFDM signal is almost the same as white Gaussian noise, and has a large ratio of average power to peak power, that is, a waveform with a high peak rate. When amplifying an OFDM signal composed of such a large number of subcarriers and with a high peak rate, if the amplifier has poor linearity (has nonlinear characteristics), intermodulation may occur between the subcarriers, causing interference. For this reason, it is necessary to use an amplifier with excellent linearity.

[0003] One method for ensuring the linearity of an amplifier is to increase the input back-off of the amplifier (the ratio of the input level at which the amplifier output saturates to the actual input signal level, the difference expressed in dB) to use the region of good linearity in the amplifier's input / output characteristics. However, this method results in low amplifier efficiency (the ratio of the power supplied to the amplifier to the power of the output signal extracted), high amplifier power consumption, and high heat generation.

[0004] As a method for solving this problem, power amplifiers with nonlinear distortion compensation functions using a predistortion method or a feedforward method have been developed and are used in transmitters for Japanese terrestrial digital broadcasting using the Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) system. On the other hand, as an approach to reduce the degradation of characteristics due to the nonlinear characteristics of such amplifiers, various methods (PAPR (Peak to Average Power Ratio) reduction methods) that suppress the peak rate of OFDM signals using signal processing have been devised. Representative methods include the TR (Tone Reservation) method and the ACE (Active Constellation Extension) method, which are adopted in DVB-T2, the second generation terrestrial digital broadcasting system in Europe, and ATSC-3.0, the terrestrial digital broadcasting system in the United States (see Non-Patent Documents 1-3). The introduction of such PAPR reduction methods is also desired for Japan's next-generation terrestrial digital broadcasting system (also called the advanced terrestrial system). [Prior art documents] [Non-patent literature]

[0005] [Non-Patent Document 1] DVB Document A133 Implementation guidelines for a second generation digital terrestrial television broadcasting system (DVB-T2) [Non-Patent Document 2] DVB Document A122 Frame structure channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2) [Non-Patent Document 3] ATSC Standard: Physical Layer Protocol Doc. A / 322:2017 Summary of the Invention [Problem to be solved by the invention]

[0006] There are two ACE methods: 1D-ACE and 2D-ACE. The 1D-ACE method is applied to OFDM signals (sometimes called "1D-ACE signals") with a constellation of UC (Uniform Constellation) in which signal points are uniformly arranged, and 1D-NUC (1 dimension Non Uniform Constellation) in which signal points are nonuniformly arranged in one dimension. The 2D-ACE method is applied to OFDM signals (sometimes called "2D-ACE signals") with a constellation of 2D-NUC (1 dimension Non Uniform Constellation) in which signal points are nonuniformly arranged in two dimensions. Both methods are applied to carrier symbols (sometimes called "data symbols") modulated by transmission data, excluding pilot symbols such as SP (Scattered Pilot) and CP (Continual Pilot), and control symbols such as TMCC (Transmission and Multiplexing Configuration and Control) and Lch.

[0007] Fig. 1 is a diagram showing a signal processing system 100 of the 1D-ACE method. Fig. 2 is a diagram showing a signal processing system 200 of the 2D-ACE method. First, the signal processing according to the 1D-ACE method will be described with reference to Fig. 1.

[0008] An OFDM frame consisting of carrier symbols (data symbols) modulated by transmission data, pilot symbols (SP, CP), and control symbols (TMCC, Lch for transmitting additional information) is input to the IFFT unit 101. The IFFT unit 101 converts the carrier symbols (hereinafter referred to as “original carrier symbols”) included in the OFDM frame into time domain waveforms by IFFT (Inverse Fast Fourier Transform), and outputs the waveforms to the upsampling unit 102.

[0009] The time domain waveform output from the IFFT unit 101 is converted into a time domain waveform oversampled at four times the sampling frequency by upsampling using zero interpolation and interpolation filtering by the upsampling unit 102 and the interpolation filter 103. The oversampled time domain waveform is input to the time domain clip processing unit 104.

[0010] The time domain clipping processor 104 clips the oversampled time domain waveform at a predetermined signal amplitude value (hereinafter referred to as the "ACE clip level") (hereinafter referred to as "ACE clip processing"). That is, the time domain clipping processor 104 suppresses the time domain waveform exceeding the ACE clip level to the ACE clip level. The time domain clipping processor 104 outputs the time domain waveform after the ACE clipping process to the pre-filter 105.

[0011] The pre-filter 105 removes unnecessary out-of-band components generated by the ACE clipping process from the time-domain waveform that has been subjected to the ACE clipping process, and outputs the result to the downsampling unit 106. The downsampling unit 106 converts the time-domain waveform from which the unnecessary components have been removed into a time-domain waveform in which the sampling frequency has been downsampled to ¼ by thinning out the waveform, and outputs the result to the FFT unit 107. The FFT unit 107 converts the downsampled time-domain waveform into a carrier symbol in the frequency domain by FFT (Fast Fourier Transform), and outputs the result to the comparator 108. Hereinafter, the carrier symbol after the ACE clipping process is referred to as the carrier symbol after the ACE clipping process.

[0012] The original carrier symbol and the ACE-clipped carrier symbol are input to the comparator 108. The comparator 108 outputs the difference between the ACE-clipped carrier symbol and the original carrier symbol to the extension unit 109.

[0013] The extension unit 109 extends the difference between the ACE clipped carrier symbol and the original carrier symbol for the extendable signal points (hereinafter referred to as "extendable signal points") among the multiple signal points of the constellation by a predetermined gain G (extension gain), and outputs the result to the adder 110.

[0014] The adder 110 receives the original carrier symbol and the error between the ACE clipped carrier symbol extended with the gain G and the original carrier symbol. The adder 110 adds the original carrier symbol and the error between the ACE clipped carrier symbol extended with the gain G and the original carrier symbol. This causes the signal point of the original carrier symbol to be extended towards the outside of the constellation.

[0015] FIG. 3 is a diagram for explaining a signal point extension rule according to the 1D-ACE method when the modulation method of the carrier symbols is 16QAM (Quadrature Amplitude Modulation).

[0016] As shown in Fig. 3, in the 1D-ACE method, among the multiple signal points that make up a square constellation, the signal points on the outer periphery of the constellation are considered to be extensible signal points, and the other signal points are considered to be non-extensible signal points (hereinafter referred to as "non-extensible signal points"). In the following drawings, the extensible signal points are illustrated with open circles, and the non-extensible signal points are illustrated with black circles.

[0017] In the 1D-ACE method, among the extendable signal points, the signal points whose ACE-clipped carrier symbols are shifted outward from the original carrier symbols are extended. That is, the signal points whose distortion vectors indicating the shift of the ACE-clipped carrier symbols from the original carrier symbols are directed outward from the constellation are extended. Therefore, even if they are extendable signal points, the signal points whose distortion vectors are directed inward from the constellation are not extended.

[0018] When an extendable signal point is extended, it is extended at a right angle to the straight line on the periphery of the constellation to which the signal point belongs (one of the four straight lines that make up the periphery of the constellation), as shown in Figure 3. However, for signal points at the four corners, diagonal extension is permitted, as shown in Figure 3. The difference between the extended signal point and the original signal point is called the extension vector.

[0019] Referring back to FIG. 1, the adder 111 adds the original carrier symbol and the error between the ACE-clipped carrier symbol extended with the gain G and the original carrier symbol, and outputs the result to the extension width limiting unit 111 .

[0020] The extension width limiting unit 111 limits the absolute value of the extension vector of each signal point to an amplitude value (extension limiting value L) that is determined independently for the I axis and the Q axis. The carrier symbol after the limiting by the extension amplitude value L is called an extension signal. The extension width limiting unit 111 outputs the extension signal to the signal selecting unit 112.

[0021] The signal selection unit 112 receives the original carrier symbol (original signal) and the extended signal. For each signal point, the signal selection unit 112 selects either the original signal or the extended signal, and outputs the selected signal to the IFFT unit 113. For an extendable signal point that has actually been extended, the signal selection unit 112 outputs the carrier symbol of the extended signal to the IFFT unit 113. Furthermore, for an extendable signal point that has not been extended and whose distortion vector faces inward in the constellation, and for an extendable signal point that cannot be extended, the signal selection unit 112 outputs the carrier symbol of the original signal to the IFFT unit 113 as is.

[0022] The IFFT unit 113 converts the carrier symbols output from the signal selection unit 112 into a time domain waveform by IFFT and outputs the time domain waveform. The time domain waveform after conversion by the IFFT unit 113 is subjected to predetermined processing such as addition of a GI (Guard Interval).

[0023] Next, a signal processing system 200 for the 2D-ACE method will be described with reference to Fig. 2. In Fig. 2, the same components as those in Fig. 1 are given the same reference numerals, and the description thereof will be omitted.

[0024] The expansion unit 201 expands the difference between the ACE-clipped carrier symbol and the original carrier symbol for an expandable signal point among a plurality of signal points of the constellation by a predetermined gain G, and outputs the expanded difference to the expansion region restriction processing unit 202 .

[0025] The extension region restriction processor 202 receives the original carrier symbol and the difference between the ACE clipped carrier symbol and the original carrier symbol extended by the gain G. The extension region restriction processor 202 restricts the region in which the signal points are extended based on the extension limit value L and the angle limit value θ.

[0026] FIG. 4 is a diagram for explaining the signal point extension rule according to the 2D-ACE method when the modulation method of the carrier symbols is NUC-64QAM.

[0027] As shown in Figure 4, in the 2D-ACE method, the signal points on the outermost circumference and the signal points close to that circumference are considered to be extensible signal points. In Figure 4, as in Figure 3, the extensible signal points are shown with open circles, and the non-extensible signal points are shown with black circles.

[0028] If the phase of the original carrier symbol relative to the origin of the IQ signal space is φ, and the phase of the difference vector (distortion vector) between the original carrier symbol and the ACE-clipped carrier symbol is φ+Δφ, then in the 2D-ACE method, the distortion vector is expanded with a predetermined gain only when -90 degrees < Δφ < 90 degrees. Here, in the 2D-ACE method, not only the magnitude of the expanded distortion vector but also an expandable region defined by an angle limit value θ is specified, and the expanded vector is the result of limiting the magnitude of the expanded distortion vector and the relative phase difference Δφ.

[0029] The extension area restriction processor 202 clips the relative phase difference Δφ to -θ / 2 when the relative phase difference Δφ of the extension vector is smaller than -θ / 2, and clips the relative phase difference Δφ to θ / 2 when the relative phase difference Δφ of the extension vector is larger than θ / 2. As a result, the extension vector is distributed in a sector-shaped extendable area with the signal point of the original carrier symbol as the apex, as shown in Fig. 4.

[0030] 2 again, the extension region restriction processing unit 202 outputs the clipped signal to the adder 203. The adder 203 receives the original signal (original carrier symbol) and the signal clipped by the extension region restriction processing unit 202. The adder 203 adds the original signal (original carrier symbol) and the clipped signal, and outputs the result to the carrier symbol selection unit 204. The signal obtained by adding the original signal (original carrier symbol) and the signal clipped by the extension region restriction processing unit 202 is referred to as an extension signal.

[0031] The carrier symbol selection unit 204 receives the original carrier symbol (original signal) and the extended signal as input. For each signal point, the carrier symbol selection unit 204 selects either the original signal or the extended signal, and outputs the selected signal to the IFFT unit 113. For actually extended signal points among the extendable signal points, the carrier symbol selection unit 204 outputs the carrier symbol of the extended signal to the IFFT unit 113. Furthermore, for signal points among the extendable signal points whose distortion vectors face inward in the constellation, and for non-extendable signals, the signal selection unit 112 outputs the carrier symbol of the original signal as is to the IFFT unit 113.

[0032] As described above, the ACE method is applied to carrier symbols (data symbols). Therefore, for pilot symbols and control symbols, the original symbols are output to the IFFT unit 113 as they are.

[0033] As described with reference to Fig. 1 to Fig. 4, in the ACE method, among the signal points of carrier symbols mapped to the IQ signal space by a predetermined modulation method, a predetermined signal point (among the extendable signal points, a signal point whose distortion vector faces the outside of the constellation) is extended toward the outside of the constellation of the predetermined modulation method. Here, as is clear from Figs. 3 and 4, even if both adjacent signal points are extended or even if one of the adjacent signal points is extended, the distance between adjacent signal points after extension will not be smaller than the distance between adjacent signal points before extension. In this way, in the ACE method, the PAPR is reduced by performing a process (extension process) of moving the signal point corresponding to the outermost circumference of the constellation toward the outside of the constellation so that the minimum signal point distance does not become small.

[0034] The above-mentioned extension process increases the signal power and moves the extended signal points away from their original positions, which affects the calculation of the log-likelihood ratio (LLR) calculated before the error correction process of the demodulator, resulting in degradation of the bit error rate (BER) characteristics.

[0035] Next, various characteristics of the ACE method, including the BER characteristics, will be explained with reference to the results of computer simulations.

[0036] Fig. 5A is a diagram showing a processing system of a computer simulation, and Fig. 5B and Fig. 5C show the specifications of the computer simulation shown in Fig. 5A.

[0037] Although a detailed explanation is omitted, the transmitting side generates a pseudo-random noise (PN) sequence, performs low-density parity-check (LDPC) coding on the PN sequence, bit interleaves the bit string after LDPC coding, generates carrier symbols by mapping the interleaved bit string to an IQ signal space, generates pilot symbols (SP, CP), constructs a frame including carrier symbols and pilot symbols, performs ACE processing on the carrier symbols, performs IFFT, and adds a GI.

[0038] Also, on the receiving side, the following are performed: setting of an FFT window, cutting out a signal in the FFT window, FFT processing, extraction of SP, channel estimation based on the SP, equalization processing based on the result of the channel estimation, calculation of LLR based on the signal after the equalization processing, bit deinterleaving opposite to the bit interleaving on the transmitting side, and LDPC decoding. Then, the BER of the signal after the LDPC decoding is measured. In this disclosure, between the equalization processing and the calculation of the LLR, a constellation clip (CC) processing is performed, the details of which will be described later. It is possible to switch whether or not to perform CC processing. Also, the complementary cumulative distribution function (CCDF) of the signal from the transmitting side to the receiving side is observed, and additive white Gaussian noise (AWGN) is added.

[0039] First, an example of characteristics when the 1D-ACE method is applied will be described with reference to Fig. 6 to Fig. 9. Fig. 6 is a diagram showing the extension gain-PAPR characteristics when the carrier modulation method is UC-1024QAM and the coding rate is 3 / 4. As shown in Fig. 6, the PAPR is smallest when the ACE clip level is 7 dB and the extension gain is about 70 (69). Fig. 7 is a diagram showing a constellation after ACE processing (1D-ACE processing) when the ACE clip level is 7 dB and the extension gain is 70.

[0040] Fig. 8 is a diagram showing the expansion gain-signal power increase characteristic when the carrier modulation method is UC-1024QAM. As shown in Fig. 8, the lower the ACE clip level and the larger the expansion gain, the larger the signal power increase due to ACE processing. When the ACE clip level is 7 dB and the expansion gain is 70, the signal power increase is about 0.3 dB.

[0041] FIG. 9 is a diagram showing the extension gain-required C / N characteristic when the carrier modulation method is UC-1024QAM. FIG. 9 shows the extension gain-required C / N characteristic when the constellation clip (CC) process described later is not performed. As shown in FIG. 9, the BER characteristic, i.e., the required C / N, does not degrade significantly when the extension gain is small, but degrades rapidly as the extension gain increases. When the ACE clip level is 7 dB, the required C / N increases at a faster rate from about 20 extension gain, and when the extension gain is 70, the required C / N increases by about 0.9 dB compared to when the extension gain is 0. The required C / N also degrades with an increase in signal power (because C(signal)) of C / N increases), but since the increase in signal power when the ACE clip level is 7 dB and the extension gain is 70 is about 0.3 dB (FIG. 8), the BER characteristic degrades by about 0.6 dB due to an error in the LLR calculation.

[0042] Next, an example of characteristics when the 2D-ACE method is applied will be described with reference to Fig. 10 to Fig. 13. Fig. 10 is a diagram showing the extended gain-PAPR characteristics when the carrier modulation method is NUC-256QAM and the coding rate is 3 / 4. As shown in Fig. 10, the PAPR is smallest when the ACE clip level is 7 dB and the extended gain is about 50 (49). Fig. 11 is a diagram showing a constellation after ACE processing (2D-ACE processing) when the ACE clip level is 7 dB and the extended gain is 49.

[0043] Fig. 12 is a diagram showing the characteristics of the expansion gain vs. the signal power increase amount when the carrier modulation method is NUC-256QAM. As shown in Fig. 12, the lower the ACE clip level and the higher the expansion gain, the larger the signal power increase amount due to ACE processing becomes. When the ACE clip level is 7 dB and the expansion gain is 49, the signal power increase amount is about 0.4 dB.

[0044] FIG. 13 is a diagram showing the expansion gain-required C / N characteristic when the carrier modulation method is NUC-256QAM. FIG. 13 shows the expansion gain-required C / N characteristic when CC processing is not performed. As shown in FIG. 13, the BER characteristic, that is, the increase in the required C / N with an increase in the expansion gain, is gentle when the expansion gain is small, but becomes rapid when the expansion gain is large. When the expansion gain is 49, the required C / N increases by about 0.75 dB compared to when the expansion gain is 0. Since the signal power increase amount when the ACE clip level is 7 dB and the expansion gain is 49 is about 0.4 dB (FIG. 12), the BER characteristic is degraded by about 0.35 dB due to an error in the LLR calculation.

[0045] The degradation of the required C / N as described above is a disadvantage in actually applying the ACE method. To simplify the following explanation, two types of C / N (observed C / N and set C / N) are specified here. The observed C / N is the C / N that includes the increase in signal power due to ACE processing in the carrier symbol (C), and is the C / N that is actually observed. The set C / N is the C / N that does not include the increase in signal power due to ACE processing (or "C" is the value when the expansion gain is zero).

[0046] Since ACE processing involves an increase in signal power, the set C / N is a virtual C / N that does not actually exist. However, it is specified here because it is useful for evaluating the degradation of the required C / N due to LLR errors.

[0047] Fig. 14 is a diagram showing the extended gain vs. required C / N characteristic when the carrier modulation method shown in Fig. 9 is UC-1024QAM, rewritten with the set C / N. That is, Fig. 14 is a diagram showing the extended gain vs. set C / N characteristic when the carrier modulation method is UC-1024QAM. Also, Fig. 15 is a diagram showing the extended gain vs. required C / N characteristic when the carrier modulation method is NUC-256QAM, shown in Fig. 13, rewritten with the set C / N. That is, Fig. 15 is a diagram showing the extended gain vs. set C / N characteristic when the carrier modulation method is NUC-256QAM.

[0048] As shown in FIGS. 14 and 15, as the expansion gain increases, the required C / N at the set C / N does not increase (deteriorate) at first, but once the expansion gain reaches a certain level, it begins to increase (deteriorate) rapidly.

[0049] As described above, the ACE method has the problem of degrading the BER characteristics. Therefore, there is a demand for technology that can reduce the PAPR using the ACE method while suppressing the degradation of the BER characteristics.

[0050] An object of the present invention is to provide a demodulation device and a program that can solve the above-mentioned problems and suppress deterioration of BER characteristics in demodulation of an OFDM signal to which the ACE method is applied. [Means for solving the problem]

[0051] (1) A demodulation device according to the present disclosure is a demodulation device that demodulates a time-domain OFDM signal that has been subjected to ACE processing in which a specific signal point among signal points of a carrier symbol mapped into an IQ signal space by a specific modulation scheme has been expanded toward the outside of a constellation of the specific modulation scheme, and the demodulation device includes an FFT unit that performs an FFT on the time-domain OFDM signal to convert it into a frequency-domain carrier symbol, a transmission path equalization unit that performs a transmission path equalization process on the frequency-domain carrier symbol, a constellation clipping unit that clips the carrier symbol after the transmission path equalization process to a certain range of the IQ signal space, a demapping / LLR calculation unit that demaps the clipped carrier symbol and calculates a log-likelihood ratio for each bit, and a decoding unit that performs error correction based on the log-likelihood ratio for each bit.

[0052] (2) In the demodulation device described in (1), the constellation is a Uniform Constellation (UC) in which signal points are uniformly arranged, or a Non-Uniform Constellation (1D-NUC) in which signal points are non-uniformly arranged in one dimension, and the constellation clipping unit clips the carrier symbol after the transmission path equalization processing so that, when the absolute value of the I-axis component of the carrier symbol after the transmission path equalization exceeds a first clip level defined for the I-axis, the absolute value of the I-axis component becomes the first clip level, and, when the absolute value of the Q-axis component of the carrier symbol after the transmission path equalization exceeds a second clip level defined for the Q-axis, the absolute value of the Q-axis component becomes the second clip level.

[0053] (3) In the demodulation device described in (1), the constellation is a 2D-NUC in which signal points are arranged unevenly in two dimensions, and the constellation clipping unit clips the carrier symbol after the transmission path equalization processing so that the absolute value of the carrier symbol after the transmission path equalization becomes the clip level without changing the phase angle of the carrier symbol after the transmission path equalization when the absolute value of the carrier symbol after the transmission path equalization exceeds a predetermined clip level.

[0054] (4) In the demodulation device described in (2), the first clip level is the maximum value of the I-axis component of the signal point on the constellation, and the second clip level is the maximum value of the Q-axis component of the signal point on the constellation.

[0055] (5) In the demodulation device described in (3), the clip level is the maximum value of the absolute values ​​of the signal points on the constellation.

[0056] (6) In the demodulation device according to (2), the first clip level is set to CL I The maximum value of the I-axis component of the signal point on the constellation is I MAX And CL I =I MAX ×α I (α I >1), and the second clip level is set to CL Q The maximum value of the Q-axis component of the signal point on the constellation is Q MAX And CL Q =Q MAX ×α Q (α Q >1), then the above α I and α Q is a constant determined so that the required C / N of the demodulated OFDM signal is equal to or less than a predetermined threshold.

[0057] (7) In the demodulation device according to (3), the clip level is set to CL R The maximum absolute value of the signal points on the constellation is RMAX And CL R =R MAX ×α R (α R >1), then the above α R is a constant determined so that the required C / N of the demodulated OFDM signal is equal to or less than a predetermined threshold.

[0058] (8) A program according to the present disclosure causes a computer to operate as the demodulation device according to any one of (1) to (7). Effect of the Invention

[0059] According to the demodulation device and program of the present invention, it is possible to suppress deterioration of the BER characteristics in demodulation of an OFDM signal to which the ACE method is applied. [Brief description of the drawings]

[0060] [Figure 1] FIG. 1 is a diagram showing a signal processing system of the 1D-ACE method. [Diagram 2] FIG. 1 is a diagram showing a signal processing system of the 2D-ACE method. [Diagram 3] FIG. 1 is a diagram illustrating an example of signal processing in the 1D-ACE method when the carrier modulation method is 16QAM. [Figure 4] FIG. 1 is a diagram illustrating an example of signal processing in the 2D-ACE method when the carrier modulation method is NUC-64QAM. [Figure 5A] FIG. 1 is a diagram showing a processing system of a computer simulation of the ACE method. [Figure 5B] FIG. 5B is a diagram (part 1) showing the specifications of the computer simulation shown in FIG. 5A. [Figure 5C] FIG. 5B is a diagram (part 2) showing the specifications of the computer simulation shown in FIG. 5A. [Figure 6] FIG. 13 is a diagram showing the extended gain-PAPR characteristics when the carrier modulation method is UC-1024QAM. [Figure 7]FIG. 13 is a diagram showing an example of a constellation after 1D-ACE processing when the carrier modulation method is UC-1024QAM. [Figure 8] FIG. 11 is a diagram showing the expansion gain vs. signal power increase characteristic when the carrier modulation method is UC-1024QAM. [Figure 9] FIG. 11 is a diagram showing the expansion gain vs. required C / N characteristic when the carrier modulation method is UC-1024QAM. [Figure 10] FIG. 11 is a diagram showing the extended gain-PAPR characteristics when the carrier modulation method is NUC-256QAM. [Figure 11] FIG. 13 is a diagram showing an example of a constellation after 2D-ACE processing when the carrier modulation method is NUC-256QAM. [Figure 12] FIG. 11 is a diagram showing the expansion gain-signal power increase characteristic when the carrier modulation method is NUC-256QAM. [Figure 13] FIG. 11 is a diagram showing the expansion gain-required C / N characteristic when the carrier modulation method is NUC-256QAM. [Figure 14] 13 is a diagram showing the extended gain-setting C / N characteristics when the carrier modulation method is UC-1024QAM. [Figure 15] FIG. 13 is a diagram showing the extended gain-setting C / N characteristics when the carrier modulation method is NUC-256QAM. [Figure 16] FIG. 11 is a diagram illustrating an example of CC processing for a 1D-ACE signal when the carrier modulation method is UC-16QAM. [Figure 17] 11 is a diagram showing an example of CC processing for a 2D-ACE signal when the carrier modulation method is NUC-64QAM. FIG. [Figure 18] 1 is a diagram showing the extended gain-PAPR characteristics when the carrier modulation method is UC-256QAM and the coding rate is 3 / 4. [Figure 19] FIG. 11 is a diagram showing the extended gain vs. required C / NPAPR characteristics when the carrier modulation method is UC-256QAM and the coding rate is 3 / 4. [Figure 20]FIG. 11 is a diagram showing another example of CC processing for a 1D-ACE signal when the carrier modulation method is UC-16QAM. [Figure 21] FIG. 13 is a diagram showing the expansion gain vs. required C / N characteristic when CC processing is applied to a 1D-ACE signal whose carrier modulation method is UC-1024QAM. [Figure 22] FIG. 13 is a diagram showing the extended gain vs. required C / N characteristic when CC processing is applied to a 2D-ACE signal whose carrier modulation method is NUC-256QAM. [Diagram 23] FIG. 13 is a diagram showing the expansion gain vs. required C / N characteristic when CC processing is applied to a 1D-ACE signal whose carrier modulation method is UC-256QAM. [Figure 24] 11 is a diagram showing the relationship between the CC coefficient and the required C / N ratio when CC processing is applied to a 1D-ACE signal whose carrier modulation method is UC-256QAM. FIG. [Diagram 25] FIG. 13 is a diagram showing the expansion gain vs. required C / N characteristic when CC processing is applied to a 1D-ACE signal whose carrier modulation method is UC-64QAM. [Figure 26] FIG. 1 is a diagram illustrating an example of the configuration of a demodulation device according to an embodiment of the present disclosure. [Figure 27] FIG. 27 illustrates an example of the configuration of a CC circuit illustrated in FIG. 26. [Figure 28] FIG. 27 is a diagram illustrating another example of the configuration of the CC circuit illustrated in FIG. 26. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0061] Hereinafter, an embodiment of the present invention will be described with reference to the drawings.

[0062] As mentioned above, there are two factors that cause the degradation of BER characteristics when the ACE method is applied. The first factor is due to the increase in power caused by moving the expandable signal points outward (in the direction in which the signal power increases) among the signal points on the periphery of the constellation based on the expansion gain. The second factor is due to an error in the LLR calculation result caused by the shifting of the expandable signal points from their original positions.

[0063] The first factor is an essential characteristic of the ACE method, so it is not easy to solve. On the other hand, there is a possibility of improving the second factor. In the ACE method, in order to prevent the deterioration of the BER characteristics as much as possible, the distance between signal points is considered not to be smaller than that of the original signal when the extendable signal points are extended outside the constellation. However, when trying to reduce the PAPR as much as possible, depending on the modulation method and the coding rate, the extension gain has to be considerably increased, resulting in deterioration of the BER characteristics. In this disclosure, the constellation of the received carrier symbol (data symbol) is clipped within a certain range to suppress errors occurring in the calculation result of the LLR and suppress deterioration of the BER characteristics. The process of clipping the constellation of the carrier symbol within a certain range is called constellation clip (CC) process.

[0064] When performing CC processing in a demodulation device that receives and demodulates an OFDM signal that has been subjected to ACE processing, the method differs depending on whether the constellation is 1D (UC or 1D-NUC) or 2D. First, a case where the constellation of carrier symbols is 1D (UC or 1D-NUC) will be described.

[0065] Fig. 16 is a diagram showing an example of CC processing when the carrier modulation method is UC-16QAM. As described with reference to Fig. 3, in 1D-ACE processing, extension processing is performed independently on the I axis and the Q axis. Therefore, CC processing is also performed independently on the I axis and the Q axis. Below, CC processing for a signal that has been subjected to 1D-ACE processing (1D-ACE signal) will be described using formulas.

[0066] The I-axis value and Q-axis value of the transmitted data symbol are respectively represented as I T ,Q T Then, the clipping levels of the I and Q axes in CC processing are I MAX ,Q MAX are expressed by the following equations (1) and (2), respectively. I MAX =max(|I T |) Formula (1) Q MAX =max(|Q T |) Formula (2)

[0067] The I-axis value and Q-axis value of the received data symbol, which are the sum of the expansion vector and the noise vector caused by amplification by the amplifier, are respectively defined as I R ,Q R Then, the I-axis value and Q-axis value of the carrier symbol after CC processing, I C ,Q C are expressed as follows. If equation (3) holds, then I C is expressed by equation (4). If equation (3) does not hold, I C is expressed by equation (5). Also, when equation (6) holds, Q C is expressed by equation (7). If equation (6) does not hold, Q C is expressed by equation (8). |I R |≧I MAX Formula (3) I C =sign(I R )·I MAX Formula (4) I C =I R Formula (5) |Q R |≧Q MAX Formula (6) Q C =sign(Q R )·Q MAX Formula (7) Q C =Q R Formula (8)

[0068] From equations (3) to (5), as shown in FIG. 16, the I-axis value I R The absolute value of is the clip level I on the I axis. MAX If it is greater than or equal to I, the I-axis value is I R If the value of is positive, the I-axis value of the carrier symbol after CC processing, I C IMAX The I-axis value of the received data symbol is clipped to I R The absolute value of is the clip level I on the I axis. MAX If it is greater than or equal to I, the I-axis value is I R If the value of is negative, the I-axis value of the carrier symbol after CC processing, I C -I MAX Also, the I-axis value of the received data symbol I R The absolute value of is the clip level I on the I axis. MAX If it is less than the I-axis value of the received symbol, I R is the I-axis value of the carrier symbol after CC processing, I C It becomes.

[0069] Furthermore, from equations (6) to (8), as shown in FIG. 16, the Q-axis value Q R The absolute value of is the clip level Q of the Q axis. MAX If it is greater than or equal to, the Q-axis value Q R If the value of is positive, the Q-axis value Q of the carrier symbol after CC processing C Q MAX The Q-axis value of the received data symbol, Q R The absolute value of is the clip level Q of the Q axis. MAX If it is greater than or equal to, the Q-axis value Q R If the value of is negative, the Q-axis value Q of the carrier symbol after CC processing C Ha-Q MAX In addition, the Q-axis value of the received data symbol I R The absolute value of is the clip level Q of the Q axis. MAX If it is less than, the Q-axis value I R is the Q-axis value Q of the carrier symbol after CC processing. C It becomes.

[0070] In this way, in 1D-ACE processing, the maximum value of the I axis is I MAX and the minimum I-axis value is -I MAX and the maximum value of the Q axis is Q MAX and the minimum value of the Q axis is -Q MAXWhen a received data symbol is located outside the clip line of a rectangle centered at the origin of the IQ signal space, the I-axis value and the Q-axis value of the received data symbol are clipped at the clip line.

[0071] Next, a case where the constellation of carrier symbols (data symbols) is 2D-NUC will be described. Fig. 17 is a diagram showing an example of CC processing when the carrier modulation method is NUC-64QAM. When 2D-ACE processing is applied to carrier symbols, as shown in Fig. 17, the maximum value R of the distance R between the origin and the signal point in the constellation of the transmission signal is MAX The signal point of the received data symbol outside the clip circle is clipped onto the circumference of a circle with a radius of . In other words, if the signal point of the received data symbol is outside the clip circle in the IQ signal space, the signal point of the received data symbol is moved to the point where the straight line connecting that signal point and the origin intersects with the clip circle. CC processing for 2D-ACE signals is explained below using mathematical expressions.

[0072] R MAX Let be the radius of the clip circle, R MAX is expressed by equation (9).

[0073]

number

[0074] The absolute value of the received data symbol (distance from the origin) is R R , the phase angle is θ R Then, R R ,θ R are expressed by equations (10) and (11), respectively.

[0075]

number

[0076] When equation (12) holds, the I-axis value and Q-axis value of the carrier symbol after CC processing, I C ,Q C are expressed by equations (13) and (14), respectively. R R ≧R MAX Formula (12) I C =R MAX cos(θ R ) Formula (13) Q C =R MAX sin(θ R ) Formula (14)

[0077] When equation (12) does not hold, the I-axis value and Q-axis value of the carrier symbol after CC processing, I C ,Q C are expressed by equations (15) and (16), respectively. I C =I R Formula (15) Q C =Q R Formula (16)

[0078] Next, a method for enhancing the effect of CC processing will be described.

[0079] 18 is a diagram showing the expansion gain-PAPR characteristic when the carrier modulation method is UC-256QAM and the coding rate is 3 / 4. As shown in FIG. 18, when the carrier modulation method is UC-256QAM, the PAPR is minimum when the ACE clip level is 7 dB and the expansion gain is 38.

[0080] Fig. 19 is a diagram showing the characteristics of the extension gain vs. required C / N (set C / N, observed C / N) when the carrier modulation method is UC-256QAM, the coding rate is 3 / 4, and the clip level is 7 dB. As shown in Fig. 19, when the extension gain at which the PAPR is the smallest is 38, the required C / N at the set C / N is almost the same as the required C / N when the extension gain is zero, and it is understood that further improvement of the required C / N by CC processing cannot be expected. However, the curve of the required C / N at the set C / N initially drops as the extension gain increases from zero, starts to rise when the extension gain is between 20 and 30, and becomes almost the same as when the extension gain is zero around the extension gain of 38 where the PAPR starts, and then continues to rise.

[0081] The movement of the required C / N curve at such a set C / N is thought to be due to the fact that as the extension gain increases, the distance between the extendable signal point on the outermost periphery of the constellation and the signal point inside the periphery of the constellation increases, making it less likely for bit errors to occur. However, as the extended signal point moves away from its original position, the calculated LLR value becomes smaller, making it more likely for bit errors to occur.

[0082] Therefore, it is possible to utilize the effect that the bit error rate is improved by increasing the distance between the extendable signal point and the signal point inside the constellation when the expansion gain is not so large. Specifically, in the 1D-ACE method, instead of clipping on the side of the outermost square of the constellation, it is possible to clip on the side of a square that is larger than the outermost square by δ (expand the clip line by δ) as shown in Fig. 20. In addition, in the 2D-ACE method, although not shown in the figure, the distance between the origin and the signal point that is the farthest from the origin is expressed as R MAX As R MAX It is possible to clip a circle with a radius of x(1+δ) as the clip circle (expand the clip circle by δ).

[0083] The expansion of the clip line and the clip circle will be described below using mathematical expressions.

[0084] Let δ>0, and α defined in equation (17) is called the CC coefficient. α=1+δ Equation (17)

[0085] In the 1D-ACE process, equation (18) is used instead of equation (1) to calculate I MAX Calculate Q using equation (19) instead of equation (2). MAX Calculate the calculated I MAX and Q MAX Then, the processing shown in the above equations (3) to (8) can be performed using the above equations. I MAX =α·max(|I T |) Formula (18) Q MAX =α·max(|Q T |) Formula (19)

[0086] In 2D-ACE processing, equation (20) is used instead of equation (9) to calculate R MAX Calculate the calculated R MAX Then, the processing shown in the above equations (10) to (16) can be performed using the above equations.

[0087]

number

[0088] As shown in Fig. 19, as the expansion gain increases from zero, the required C / N at the set C / N first drops, then reaches a minimum value, and then rises. This characteristic is not common to all cases. For example, when the carrier modulation method is UC-1024QAM or NUC-256QAM, the tendency shown in Fig. 19 was not clearly observed. In general, the smaller the modulation multi-level number of the carrier modulation and the lower the coding rate, the stronger the tendency shown in Fig. 19 appears. Note that the optimal value of the CC coefficient α differs depending on the carrier modulation method (multi-level number, UC or NUC, etc.), the coding rate, the ACE clip level, and the expansion gain, so it is sufficient to obtain it in advance by simulation, etc.

[0089] Next, the effect of the CC processing according to the present disclosure will be described.

[0090] Fig. 21 is a diagram showing the expansion gain vs. required C / N characteristics when the carrier modulation method is UC-1024QAM, the coding rate is 3 / 4, and the ACE clip level is 7 dB. Fig. 21 shows the expansion gain vs. required C / N characteristics with and without CC processing.

[0091] Without CC processing, as the expansion gain increases, both the required C / N at the set C / N and the required C / N at the observed C / N increase. On the other hand, with CC processing, the required C / N at the set CN barely changes even when the expansion gain increases, remaining a constant value (approximately 26.45 dB), while the required C / N at the observed C / N increases, reflecting only the increase in signal power. As a result, an improvement in the required C / N of approximately 0.6 dB was obtained.

[0092] Fig. 22 is a diagram showing the expansion gain vs. required C / N characteristics when the carrier modulation method is NUC-256QAM, the coding rate is 3 / 4, and the ACE clip level is 7 dB. Fig. 22 shows the expansion gain vs. required C / N characteristics with and without CC processing.

[0093] Without CC processing, as the expansion gain increases, both the required C / N at the set C / N and the required C / N at the observed C / N increase. On the other hand, with CC processing, the required C / N at the set C / N remains almost constant (about 20.5 dB) even when the expansion gain increases, while the required C / N at the observed C / N increases, reflecting only the increase in signal power. As a result, an improvement in the required C / N of about 0.35 dB was obtained.

[0094] Next, Fig. 23 shows the expansion gain vs. required C / N characteristics when the carrier modulation method is UC-256QAM, the coding rate is 3 / 4, the ACE clip level is 7 dB, and the CC coefficient α = 1. Fig. 23 shows the expansion gain vs. required C / N characteristics with and without CC processing.

[0095] When the carrier modulation method is UC-256QAM, the coding rate is 3 / 4, and the ACE clip level is 7dB, as shown in Fig. 23, the PAPR is minimum when the expansion gain is 38, and the improvement in the required C / N due to CC processing when the expansion gain is 38 is about 0.07dB. This is considerably smaller than when the carrier modulation method is UC-1024QAM as described with reference to Fig. 21 and when the carrier modulation method is NUC-256QAM as described with reference to Fig. 22.

[0096] FIG. 24 is a diagram showing the relationship between the CC coefficient α and the required C / N when the CC coefficient is changed at the expansion gain point of 38, when the carrier modulation method is UC-256QAM, the coding rate is 3 / 4, and the ACE clip level is 7 dB.

[0097] As shown in Fig. 24, when the CC coefficient α becomes larger than 1, the required C / N at both the set C / N and observed C / N suddenly drops by about 0.05 dB, and after maintaining a nearly constant value for a while, it rises by 0.07 dB when the CC coefficient α becomes about 1.8 (δ = 0.8), and then it remains at that value. This is because when the ACE clip level is 7 dB and the expansion gain is 38, the magnitude of the expanded signal point reaches the maximum value of the I axis and Q axis (I MAX ,Q MAX ) is rarely exceeded. It was confirmed that by performing CC processing with an adjusted CC coefficient α (α = 1) or CC processing with an adjusted CC coefficient α before demapping, it is possible to improve the PAPR while suppressing degradation of the required C / N (BER characteristics).

[0098] However, CC processing may increase the required C / N ratio. Fig. 25 shows the extension gain vs. required C / N ratio characteristics when the carrier modulation method is UC-64QAM and the ACE clip level is 6.5 dB. Fig. 25 shows the extension gain vs. required C / N ratio characteristics when CC processing is performed and when CC processing is not performed. When the carrier modulation method is UC-64QAM and the ACE clip level is 6.5 dB, the PAPR is minimized when the extension gain is 21. Comparing the required C / N ratio when CC processing is performed and when CC processing is not performed at the point where the extension gain is 21, the required C / N ratio when CC processing is performed is about 0.1 dB higher than when CC processing is not performed for both the set C / N ratio and the observed C / N ratio.

[0099] This is because, as described above, the CC processing cancels the advantage of extending signal points by ACE processing, that is, the outermost signal points of the constellation are moved away from the inner signal points of the constellation, making it less likely to cause errors. In the example shown in Fig. 25, at the extension gain of 21, the required C / N at the set C / N without CC processing is almost the minimum value, so there is no need to perform CC processing. If the extension gain at which the PAPR is minimum is larger (approximately 40 or more), it is considered that the effect of CC processing can be obtained.

[0100] In this way, it is clear that it is necessary to determine whether or not to perform CC processing and to adjust the CC coefficient α depending on the carrier modulation method, the coding rate, the ACE clip level, the expansion gain, and the like.

[0101] Next, the configuration of the demodulation device 300 according to the present disclosure will be described with reference to Fig. 26. The demodulation device 300 according to the present disclosure receives and demodulates a time domain OFDM signal that has been subjected to ACE processing. As described above, the ACE processing is a process of extending a predetermined signal point out of the signal points of the carrier symbol mapped to the IQ signal space by a predetermined modulation method toward the outside of the constellation of the predetermined modulation method.

[0102] As shown in FIG. 26 , a demodulation device 300 according to the present disclosure includes a band pass filter (BPF) 301, an analog-to-digital converter (ADC) 302, an orthogonal demodulation unit 303, a GI correlation unit 304, a clock recovery unit 305, a local frequency error correction unit 306, an FFT unit 307, a TMCC demodulation unit 308, a channel estimation unit 309, a weight calculation unit 310, a channel equalization unit 311, a noise variance calculation unit 312, The decoder includes a noise distribution frequency deinterleaving unit 313, a noise distribution time deinterleaving unit 314, a frequency deinterleaving unit 315, a time deinterleaving unit 316, a constellation clip (CC) unit 317, a demapping / LLR calculation unit 318, a bit deinterleaving unit 319, an FEC block conversion unit 320, and an LDPC decoding unit 321 and a BCH decoding unit 322 as decoding units.

[0103] An OFDM signal (analog format) in the time domain in the IF (Intermediate Frequency) band is input to the BPF 301. The BPF 301 removes unnecessary components from the input time domain OFDM signal and outputs the signal to the ADC 302.

[0104] The ADC 302 converts the time domain OFDM signal (analog format) output from the BPF 301 into an IF signal in digital format, and outputs it to the orthogonal demodulation unit 303 .

[0105] The orthogonal demodulation unit 303 converts the IF signal output from the ADC 302 into an equivalent baseband IQ signal, and outputs it to the GI correlation unit 304 and the local frequency error correction unit 306 .

[0106] The GI correlator 304 detects a frequency error by performing a GI correlation calculation on the IQ signal output from the orthogonal demodulator 303, and outputs the detection result to the local frequency error corrector 306. In addition, the GI correlator 304 outputs a clock reference signal obtained by the GI correlation calculation to the clock recovery unit 305.

[0107] The clock recovery unit 305 recovers the clock based on the clock reference signal output from the GI correlation unit 304 .

[0108] The local frequency error correction unit 306 corrects the frequency error of the IQ signal output from the orthogonal demodulation unit 303 based on the frequency error output from the GI correlation unit 304 , and outputs the corrected frequency error to the FFT unit 307 .

[0109] The FFT unit 307 performs FFT on the time domain OFDM signal to convert it into a carrier symbol in the frequency domain. Specifically, the FFT unit 307 converts the IQ signal output from the local frequency error correction unit 306 into a carrier symbol in the frequency domain by FFT. The FFT unit 307 outputs the TMCC signal, out of the carrier symbols in the frequency domain, to the TMCC demodulation unit 308, outputs reference signals such as SP and CP to the channel estimation unit 309, and outputs the data signal (data symbol) to the channel equalization unit 311.

[0110] The TMCC demodulation unit 308 demodulates the TMCC signal output from the FFT unit 307 to obtain the control information.

[0111] The channel estimation unit 309 performs channel estimation using the reference signal output from the FFT unit 307 , and outputs the result to the weight calculation unit 310 .

[0112] The weight calculation unit 310 calculates a weighting coefficient for each carrier based on the result of the propagation path estimation output from the propagation path estimation unit 309 , and outputs the calculated weighting coefficient to a transmission path equalization unit 311 and a noise variance calculation unit 312 .

[0113] The transmission channel equalization unit 311 performs a transmission channel equalization process on the carrier symbols in the frequency domain. Specifically, the transmission channel equalization unit 311 performs transmission channel equalization on the data signal (data symbol) output from the FFT unit 307 based on the weighting coefficient output from the weight calculation unit 310, and outputs the result to the noise variance calculation unit 312 and the frequency deinterleaving unit 315.

[0114] The noise variance calculation unit 312 calculates the noise variance σ for each carrier based on the weighting coefficient output from the weight calculation unit 310 and the data symbol after transmission path equalization processing output from the transmission path equalization unit 311, and outputs it to the noise variance frequency deinterleaving unit 313.

[0115] The noise variance frequency deinterleaving unit 313 performs frequency deinterleaving on the noise variance σ for each carrier output from the noise variance calculation unit 312 , and outputs the result to the noise variance time deinterleaving unit 314 .

[0116] The noise variance time deinterleaving unit 314 performs time deinterleaving on the noise variance σ for each carrier after frequency deinterleaving output from the noise variance frequency deinterleaving unit 313 , and outputs the result to the demapping / LLR calculation unit 318 .

[0117] The frequency deinterleaving unit 315 performs frequency deinterleaving on the data symbols after transmission path equalization output from the transmission path equalization unit 311 , and outputs the result to the time deinterleaving unit 316 .

[0118] The time deinterleaving unit 316 performs time deinterleaving on the data symbols after frequency deinterleaving output from the frequency deinterleaving unit 315 , and outputs the result to the CC unit 317 .

[0119] The CC unit 317 clips the data symbols after the transmission path equalization processing to a certain range in the IQ signal space, and outputs the data symbols after the CC processing to the demapping and LLR calculation unit 318. The CC unit 317 will be described in detail later.

[0120] The demapping / LLR calculation unit 318 demaps the data symbols clipped by the CC unit 317, and calculates a log-likelihood ratio (LLR) for each bit. Specifically, the demapping / LLR calculation unit 318 performs demapping processing using the data symbols after CC processing by the CC unit 317 and the noise variance after time deinterleaving by the noise variance time deinterleaving unit 314, and calculates an LLR for each bit. The demapping / LLR calculation unit 318 outputs the calculated LLR for each bit to the bit deinterleaving unit 319.

[0121] The bit deinterleaving unit 319 performs bit deinterleaving on the LLR for each bit output from the demapping and LLR calculation unit 318 , and outputs the result to the FEC block conversion unit 320 .

[0122] The FEC block conversion unit 320 divides the bit-deinterleaved LLR for each bit output from the bit deinterleaving unit 319 into blocks (FEC blocks) whose size is the LDPC code length, and outputs the blocks to the LDPC decoding unit 321.

[0123] The LDPC decoding unit 321 performs LDPC decoding on the FEC block output from the FEC block conversion unit 320 , and outputs a bit string to the BCH decoding unit 322 .

[0124] The BCH decoding unit 322 performs BCH decoding on the bit string output from the LDPC decoding unit 321, and outputs a bit stream.

[0125] Next, the processing of the CC unit 317 will be described in more detail.

[0126] When the received signal is a 1D-ACE signal, that is, when the constellation of the modulation method is UC in which signal points are uniformly arranged, or 1D-NUC in which signal points are non-uniformly arranged in one dimension, as described with reference to FIG. 16, when the absolute value of the I-axis component of the carrier symbol after the transmission path equalization exceeds a clip level (first clip level) defined for the I-axis, the CC unit 317 clips the carrier symbol after the transmission path equalization so that the absolute value of the I-axis component becomes the first clip level. Also, when the absolute value of the Q-axis component of the carrier symbol after the transmission path equalization exceeds a clip level (second clip level) defined for the Q-axis, the CC unit 317 clips the carrier symbol after the transmission path equalization so that the absolute value of the Q-axis component becomes the second clip level. Here, the first clip level is, for example, the maximum value of the I-axis component of the signal point on the constellation (I MAX ) The second clip level is, for example, the maximum value of the Q-axis component of the signal point on the constellation (Q MAX ).

[0127] Also, CC section 317 may expand the clip line (i.e., the first clip level and the second clip level) as described with reference to equations (17) to (19) and FIG. 20. In this case, CC section 317 expands the clip line by adjusting the CC coefficient. MAX The CC coefficient for the I axis is α I (α I >1), the first clip level CL I =I MAX ×α I In addition, the maximum value of the Q-axis component of the signal point on the constellation is Q MAX The CC coefficient for the Q axis is α Q (α Q >1), the second clip level CL Q =Q MAX ×α Q Here, the CC coefficient for the I axis is α I and CC coefficient α for the Q axis Qis a constant determined so that the required C / N of the demodulated OFDM signal is equal to or less than a predetermined threshold. In this way, an appropriate CC coefficient α (CC coefficient α I ,α Q ) can be set to further suppress the degradation of the BER characteristics.

[0128] When the received signal is a 2D-ACE signal, that is, when the constellation of the modulation method is 2D-NUC in which signal points are arranged non-uniformly in two dimensions, the CC unit 317 clips the carrier symbol after the transmission path equalization process so that the absolute value of the carrier symbol after the transmission path equalization becomes the clip level without changing the phase angle of the carrier symbol after the transmission path equalization, when the absolute value of the carrier symbol after the transmission path equalization exceeds a predetermined clip level, as described with reference to FIG. 17. Here, the clip level is, for example, the maximum value of the absolute values ​​of the signal points on the constellation (R MAX ).

[0129] Also, the CC unit 317 may expand the clip circle (i.e., the clip level) as described with reference to equations (17) and (20). In this case, the CC unit 317 expands the clip circle by adjusting the CC coefficient. MAX Let the CC coefficient be α R (α R >1), the clip level CL R =R MAX ×α R Here, the CC coefficient α R is a constant determined so that the required C / N of the demodulated OFDM signal is equal to or less than a predetermined threshold. In this way, an appropriate CC coefficient α (CC coefficient α R ) can be set to further suppress the degradation of the BER characteristics.

[0130] Next, we will explain the configuration of the CC section 317. The configuration of the CC section 317 differs depending on whether it is applied to a 1D-ACE signal or a 2D-ACE signal. First, we will explain the configuration of the CC section 317 when it is applied to a 1D-ACE signal.

[0131] FIG. 27 is a diagram showing an example of the configuration of the CC section 317 when applied to a 1D-ACE signal.

[0132] As shown in FIG. 27, the CC unit 317 includes an absolute value calculation unit 41i, a sign detection unit 42i, a maximum value generation unit 43i, a multiplier 44i, a comparator 45i, a switch 46i, an absolute value calculation unit 41q, a sign detection unit 42q, a maximum value generation unit 43q, a multiplier 44q, a comparator 45q, and a switch 46q. The absolute value calculation unit 41i, the sign detection unit 42i, the maximum value generation unit 43i, the multiplier 44i, the comparator 45i, and the switch 46i are provided corresponding to the I-axis component. The absolute value calculation unit 41q, the sign detection unit 42q, the maximum value generation unit 43q, the multiplier 44q, the comparator 45q, and the switch 46q are provided corresponding to the Q-axis component. Since the processing is similar for the I-axis component and the Q-axis component, the configuration provided for the I-axis component will be described below.

[0133] The I-axis component of the received data symbol is input to the absolute value calculation unit 41i. The absolute value calculation unit 41i calculates the absolute value of the I-axis component of the input received data symbol, and supplies the result to the B input of the comparator 45i.

[0134] The sign detection unit 42i receives the I-axis component of the received data symbol as input. The sign detection unit 42i detects whether the I-axis component of the input received data symbol is positive or negative, and outputs "1" to the multiplier 44i if the result is positive, and outputs "-1" to the multiplier 44i if the result is negative.

[0135] The maximum value generating unit 43i is a predetermined I-axis clip level I MAX The maximum value generating unit 43i records the clip level I MAX is supplied to the A input of comparator 45i.

[0136] The multiplier 44i multiplies the "1" or "-1" output from the sign detector 42i by the I-axis clip level I output from the maximum value generator 43i. MAX The result of the multiplication is supplied to the second input of the switch 46i.

[0137] The comparator 45i detects the clip level I of the I axis supplied to the A input. MAX and the absolute value of the I-axis component of the received data symbol supplied to the B input. The comparator 45i outputs a control signal to the switch 46i such that, when the A input is greater than the B input, the 1-system input of the switch 46i described below is selected, and when the A input is less than or equal to the B input, the switch 46i selects the 2-system input.

[0138] The switch 46i receives the I-axis component of the received data symbol as a 1-channel input, and receives the multiplication result by the multiplier 44i as a 2-channel input, as described above. The switch 46i outputs one of the 1-channel input and the 2-channel input as the I-axis component of the CC-processed data symbol according to the control signal output from the comparator 45i.

[0139] FIG. 28 is a diagram showing an example of the configuration of the CC unit 317 when applied to a 2D-ACE signal.

[0140] As shown in FIG. 28, the CC unit 317 includes coordinate converters 51 and 55, a maximum value generating unit 52, a comparator 53, and a switch .

[0141] The I-axis component and Q-axis component of the received data symbol are input to the coordinate converter 51. The coordinate converter 51 converts the received data symbol into an absolute value R and a phase angle θ by converting the coordinates from rectangular coordinates to polar coordinates. The coordinate converter 51 supplies the absolute value R of the received data symbol to the B input of the comparator 53 and to the 1-system input of the switch 54. The coordinate converter 51 also outputs the phase angle θ of the received data symbol to the coordinate converter 55.

[0142] The maximum value generating unit 52 generates a predetermined maximum value R of the absolute values ​​of the received data symbols.MAX is recorded and supplied to the A input of the comparator 53 and the 2-system input of the switch 54.

[0143] The comparator 53 detects the maximum absolute value R of the received data symbols supplied to the A input. MAX and the absolute value R of the received data symbol supplied to the B input. The comparator 53 outputs a control signal to the switch 54 such that, if the A input is greater than the B input, the 1-system input of the switch 46i is selected, and, if the A input is less than or equal to the B input, the 2-system input of the switch 46i is selected.

[0144] The switch 54 receives the absolute value R of the received data symbol as a first input and the maximum value R of the absolute value of the received data symbol as a second input. MAX The switch 54 outputs one of the first-system input and the second-system input to the coordinate converter 55 in accordance with the control signal output from the comparator 53.

[0145] Coordinate converter 55 receives the absolute value of the received data symbol output from switch 54 and the phase angle θ of the received data symbol output from coordinate converter 51. Coordinate converter 51 converts the received data symbol into I-axis and Q-axis components by coordinate conversion from polar coordinates to Cartesian coordinates, and outputs them as the I-axis and Q-axis components of the data symbol after CC processing.

[0146] As described with reference to FIGS. 26 and 27, the CC unit 317 can be realized with an extremely simple configuration, and problems such as an increase in the circuit scale of the demodulation device 300 do not occur.

[0147] As described above, the demodulation device 300 according to the present disclosure includes an FFT unit 307 that performs an FFT on a time domain OFDM signal to convert it into a frequency domain carrier symbol, a transmission path equalization unit 311 that performs a transmission path equalization process on the frequency domain carrier symbol, a constellation clipping unit 317 that clips the carrier symbol after the transmission path equalization process to a certain range in the IQ signal space, a demapping / LLR calculation unit 318 that demaps the clipped carrier symbol and calculates a log-likelihood ratio for each bit, and an LDPC decoding unit 321 that performs error correction based on the log-likelihood ratio for each bit.

[0148] By clipping the carrier symbol after the transmission path equalization process to a certain range in the IQ signal space, the deviation of the expandable signal point from its original position by the ACE method is reduced. As a result, the error caused by the LLR calculation result is also reduced, so the deterioration of the BER characteristics in the demodulation of the OFDM signal to which the ACE method is applied can be suppressed.

[0149] Although not specifically mentioned in the embodiment, a program for operating a computer as the demodulation device 300 may be provided. The program may be recorded in a computer-readable medium. The computer-readable medium can be used to install the program in a computer. Here, the computer-readable medium on which the program is recorded may be a non-transient recording medium. The non-transient recording medium is not particularly limited, and may be, for example, a recording medium such as a CD-ROM or a DVD-ROM.

[0150] Alternatively, a chip may be provided that is configured by a memory that stores a program for executing each process performed by the demodulation device 300 and a processor that executes the program stored in the memory and is mounted on the transmission device 10.

[0151] Although the above-mentioned embodiment has been described as a representative example, it is clear to those skilled in the art that many modifications and substitutions are possible within the spirit and scope of the present invention. Therefore, the present invention should not be interpreted as being limited by the above-mentioned embodiment, and various modifications and changes are possible without departing from the scope of the claims. For example, it is possible to combine multiple building blocks shown in the configuration diagram of the embodiment into one, or to divide one building block. [Explanation of symbols]

[0152] 100 Signal Processing System 101 IFFT section 102 Upsampling section 103 Interpolation Filter 104 Time domain clipping processing unit 105 Pre-filter 106 Downsampling section 107 FFT section 108 Comparator 109,201 Extension 110 Adder 111 Expansion width limit section 112 Signal selection section 113 IFFT section 202 Extended Area Restriction Section 203 Adder 204 Carrier Symbol Selection Unit 301 Bandpass Filter 302 Analog to Digital Converter 303 Quadrature demodulation section 304 GI Correlation Department 305 Clock recovery section 306 Local Frequency Error Correction Unit 307 FFT section 308 TMCC demodulation section 309 Propagation Path Estimation Unit 310 Weight calculation unit 311 Transmission line equalization section 312 Noise Variance Calculation Unit 313 Noise dispersion frequency deinterleave unit 314 Noise dispersion time deinterleave unit 315 Frequency Deinterleave Unit 316 Time Deinterleave Section 317 Constellation Clip 318 Demapping / LLR calculation section 319-bit deinterleave section 320 FEC block conversion unit 321 LDPC decoder 322 BCH Decoding Unit 41i, 41q Absolute value calculation section 42i, 42q Code detection section 43i, 43q Maximum value generating section 44i,44q multiplier 45i,45q comparator 46i,46q switch 51,55 Coordinate conversion section 52 Maximum value generating section 53 Comparator 54 Switch

Claims

1. A demodulation device that demodulates a time domain OFDM signal that has been subjected to ACE processing in which a predetermined signal point among signal points of carrier symbols mapped to an IQ signal space by a predetermined modulation method is expanded toward the outside of a constellation of the predetermined modulation method, an FFT unit that performs FFT on the time domain OFDM signal to convert it into a carrier symbol in the frequency domain; a transmission path equalization unit that performs a transmission path equalization process on the carrier symbol in the frequency domain; a constellation clipping unit that clips the carrier symbol after the transmission path equalization processing to a certain range of the IQ signal space; a demapping and LLR calculation unit that demaps the clipped carrier symbols and calculates a log-likelihood ratio for each bit; A demodulation device comprising: a decoding unit that performs error correction based on the log-likelihood ratio for each bit.

2. 2. The demodulation device according to claim 1, The constellation is a Uniform Constellation (UC) in which signal points are uniformly arranged, or a Non-Uniform Constellation (1D-NUC) in which signal points are non-uniformly arranged in one dimension, a demodulation device, wherein the constellation clipping unit clips the carrier symbol after the transmission path equalization processing so that, when an absolute value of an I-axis component of the carrier symbol after the transmission path equalization exceeds a first clip level defined for the I-axis, the absolute value of the I-axis component becomes the first clip level, and, when an absolute value of a Q-axis component of the carrier symbol after the transmission path equalization exceeds a second clip level defined for the Q-axis, the absolute value of the Q-axis component becomes the second clip level.

3. 2. The demodulation device according to claim 1, The constellation is a 2D-NUC in which signal points are non-uniformly arranged in two dimensions, a constellation clipping unit that clips the carrier symbol after the transmission path equalization processing when the absolute value of the carrier symbol after the transmission path equalization exceeds a predetermined clip level, without changing the phase angle of the carrier symbol after the transmission path equalization, so that the absolute value of the carrier symbol after the transmission path equalization becomes the clip level.

4. 3. The demodulation device according to claim 2, the first clip level is a maximum value of an I-axis component of a signal point on the constellation; A demodulation device, wherein the second clip level is the maximum value of the Q-axis component of the signal point on the constellation.

5. 4. The demodulation device according to claim 3, A demodulation device, wherein the clip level is the maximum absolute value of signal points on the constellation.

6. 3. The demodulation device according to claim 2, The first clip level is CL Ｉ The maximum value of the I-axis component of the signal point on the constellation is I ＭＡＸ And, CL Ｉ = I ＭＡＸ ×α Ｉ (α Ｉ > 1) and The second clip level is CL Ｑ The maximum value of the Q-axis component of the signal point on the constellation is Q ＭＡＸ And, CL Ｑ =Q ＭＡＸ ×α Ｑ (α Ｑ > 1) Then, The a Ｉ and α Ｑ is a constant determined so that a required C / N ratio of the OFDM signal after demodulation is equal to or less than a predetermined threshold.

7. 4. The demodulation device according to claim 3, The clip level is CL Ｒ The maximum absolute value of the signal points on the constellation is R ＭＡＸ And, CL Ｒ = R ＭＡＸ ×α Ｒ (α Ｒ > 1) Then, The a Ｒ is a constant determined so that a required C / N ratio of the OFDM signal after demodulation is equal to or less than a predetermined threshold.

8. A program for causing a computer to operate as the demodulation device according to claim 1.

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