Signal transmission, calibration, compensation and transmission / reception links, integrated circuits, electromagnetic wave sensors and devices

Abstract

This disclosure relates to the technical field of electromagnetic wave sensors, and more specifically to signal transmission, calibration, compensation, and transmission / reception links, integrated circuits, electromagnetic wave sensors, and devices. The transmission link includes an analog signal source and a digital phase shifter, the analog signal source being configured to supply an initial analog signal, and the digital phase shifter being configured to generate a phase shift signal in the digital domain to perform a preset phase shift operation on the initial analog signal, and to phase shift the initial analog signal based on the phase shift signal. This effectively improves phase modulation accuracy and precision, while also avoiding offline calibration operations on links and devices such as phase shifters in the transmission link, and further reducing the complexity and difficulty of process implementation. In addition, the transmission link area and loss of the phase shift architecture can be effectively reduced, improving system stability and reducing channel coupling.

Description

[Technical Field]

[0001] This application claims priority to a Chinese patent application filed with the China National Intellectual Property Office on June 14, 2023, with application number 202310702586.5 and title "Signal transmission, calibration, compensation and transceiver link, IQ mixer, integrated circuit, sensor and apparatus," and to a Chinese patent application filed with the China National Intellectual Property Office on May 14, 2024, with application number 202410599441.1 and title "Signal transmission link and method, transceiver link, integrated circuit and electromagnetic wave device," and it should be understood that the contents of these applications are incorporated into this disclosure by reference.

[0002] The embodiments of this disclosure relate to, but are not limited to, the technical field of electromagnetic wave sensors, and more specifically to signal transmission, calibration, compensation and transmission / reception links, integrated circuits, electromagnetic wave sensors and devices. [Background technology]

[0003] Signal transmission links employing analog phase shifter architectures suffer from problems with low phase modulation resolution and accuracy, and are unable to meet various demands for high precision and accuracy. [Overview of the project]

[0004] To solve the above technical problems, the embodiments of this disclosure provide technologies such as signal transmission, calibration, compensation, and transmission / reception links, IQ mixers, integrated circuits, electromagnetic wave sensors and devices, signal transmission / reception links based on a digital phase shifter architecture, and corresponding calibration and compensation links. This effectively improves phase modulation resolution and accuracy while avoiding offline calibration operations on links and devices such as phase shifters in the transmission link, and further reduces the complexity and difficulty of process implementation. In addition, it effectively reduces the transmission link area and loss of the phase shift architecture, improves system stability, and reduces channel coupling.

[0005] Embodiments of the present disclosure provide a signal transmission link applicable to an electromagnetic wave sensor, the transmission link comprising an analog signal source and a digital phase shifter, wherein the analog signal source is configured to supply an initial analog signal, and the digital phase shifter is configured to generate a phase shift signal in the digital domain to perform a preset phase shift operation on the initial analog signal, and to phase shift the initial analog signal based on the phase shift signal.

[0006] Exemplary, the signal transmission link further includes a transmitting antenna, which is configured to radiate the initial analog signal after phase shift into a predetermined spatial region.

[0007] Exemplary, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter, and a mixer, wherein the digital phase shifter is configured to generate a digital phase shift signal, the digital-to-analog converter is configured to convert the received digital phase shift signal into an analog phase shift signal, and the mixer is configured to perform a mixing operation on the received initial analog signal using the received analog phase shift signal in order to perform a preset phase shift operation on the initial analog signal.

[0008] Exemplarily, the digital phase shift signal source includes a direct digital frequency synthesizer, the digital-to-analog converter is an IQ digital-to-analog converter, and the mixer is an IQ mixer.

[0009] Exemplarily, the digital phase shift signal is a single tone signal, the initial analog signal is a frequency sweep signal, or the digital phase shift signal is a frequency sweep signal and the initial analog signal is a single tone signal.

[0010] Exemplarily, the signal transmission link transmits a frequency modulated continuous wave signal.

[0011] Embodiments of the present disclosure further provide a signal transmission link, which includes a main signal transmission path and a signal calibration link integrated in the same integrated circuit. The signal calibration link is configured to calibrate the main signal transmission path to obtain compensation information, and the main signal transmission path is configured to generate a radio frequency transmission signal after performing a compensation operation based on the compensation information to achieve target detection and / or communication.

[0012] Exemplarily, the compensation information includes at least one of harmonic distortion compensation parameters, local oscillator leakage compensation parameters, and quadrature imbalance compensation parameters.

[0013] Exemplarily, the main signal transmission path includes a first signal source and a phase shifter. The first signal source is configured to generate a first analog signal, and the phase shifter is configured to perform frequency shift and / or phase shift on the first analog signal to form a radio frequency transmission signal.

[0014] Exemplarily, when the phase shifter has a non-orthogonal architecture, the phase shifter includes a second signal source and a transmission mixer. The second signal source is configured to generate a second analog signal, and the transmission mixer is configured to perform mixing processing on the first analog signal and the second analog signal to form the radio frequency transmission signal. If the phase shifter has an orthogonal architecture, the phase shifter includes a second signal source, a digital-to-analog conversion module, and a transmit mixer, wherein the second signal source is configured to generate a first digital signal, the digital-to-analog conversion module is configured to convert the first digital signal to a second analog signal, and the transmit mixer is configured to frequency-shift and / or phase-shift the first analog signal based on the second analog signal in order to form the radio frequency transmit signal.

[0015] Exemplary, the transmitter path further includes a compensation circuit, the signal input terminal of the compensation circuit is connected to the second signal source, the signal input terminal is connected to the phase shifter, and the compensation circuit merges the compensation signal with the signal output from the second signal source and outputs it.

[0016] For example, if the signal output from the signal transmission path is a two-channel orthogonal signal, the compensation signal used in the compensation circuit is a two-channel orthogonal signal. If the signal output from the signal transmission path is not a two-channel orthogonal signal, the compensation signal used in the compensation circuit is a one-channel signal of the same type as the signal output from the second signal source, and the signal type is either a digital or analog signal.

[0017] Exemplary, the compensation circuit includes a compensation signal generator and an adder, wherein the compensation signal generator is configured to generate the compensation signal, and the adder is connected to the compensation signal generator and the second signal source, and performs a signal superposition operation between the signal output from the second signal source and the compensation signal output from the compensation signal generator.

[0018] For example, if the signal output from the signal transmission main path is a two-channel orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal unbalanced compensation signal unit, and a local oscillator leak compensation signal unit. If the signal output from the signal transmission main path is not a two-channel orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leak compensation signal unit. The compensation signal generated by the harmonic compensation signal unit is used to cancel the harmonic signals of the main frequency signal in the signal transmission main path. The compensation signal from the local oscillator leak compensation signal unit is used to compensate for the leak signals generated by the transmitting local oscillator signal in the signal transmission main path. The compensation signal generated by the orthogonal unbalanced compensation signal unit is used to compensate for the mirror signals of the main frequency signal in the signal transmission main path.

[0019] For example, the compensation signal generated by the harmonic compensation signal unit has the same frequency, the same amplitude, and the opposite phase to the harmonic signal in the signal transmission main path.

[0020] Exemplary, the harmonic compensation signal unit includes an n-th power module or an n-frequency signal generator, wherein the n-th power module's signal input terminal is connected to the signal output terminal of the second signal source, and it is used to obtain the compensation signal by generating a signal with a frequency n times the frequency of the signal output from the second signal source using the signal output from the second signal source. The n-frequency signal generator is used to obtain the compensation signal by generating a signal with a frequency n times the frequency of the signal output from the second signal source. The value of n is a positive integer. Exemplary, the value of n is an odd number. Exemplary, the value of n is 3.

[0021] For example, the compensation signal generated by the local oscillator leak compensation signal unit is generated based on a leak signal corresponding to the first analog signal used in the phase shifter.

[0022] For example, the compensation signal generated by the local oscillator leak compensation signal unit has the same frequency and amplitude as the leak signal, but in opposite phase.

[0023] For example, the compensation signal used in the orthogonal unbalanced compensation circuit has the same frequency, the same amplitude, and opposite phase between the mirror signal corresponding to the signal transmitter path and the mirror signal corresponding to the desired signal generated by the signal transmitter path.

[0024] For example, the compensation signal used in the orthogonal unbalanced compensation circuit is determined by the signal output from the second signal source and the complex conjugate signal obtained by inverting the frequency of the signal output from the second signal source.

[0025] For example, a method for obtaining a compensation signal generated by the orthogonal unbalanced compensation circuit includes obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal, and calculating the difference between the signal output from the second signal source and the adjustment signal to obtain a compensation signal generated by the orthogonal unbalanced compensation circuit.

[0026] For example, the pre-compensation coefficient is determined based on the ratio of the amplitude of the desired signal to the amplitude of the Miller signal corresponding to the desired signal.

[0027] For example, the signal calibration link includes a signal receiving link corresponding to the signal transmission main path, the signal receiving link is used to receive and process an echo signal corresponding to an FMCW radio frequency transmission signal, the signal receiving link is configured to acquire a sampled signal from the signal transmission main path and obtain configuration information for a compensation signal based on the sampled signal, and the transmitting local oscillator signal used in the signal transmission main path has a different frequency from the receiving local oscillator signal used in the signal receiving link.

[0028] Exemplary, the signal calibration link further includes a frequency adjustment circuit configured to adjust the frequency of at least one of the sampling signal and the received local oscillator signal.

[0029] Illustratively, the signal input terminal of the frequency adjustment circuit is connected between the signal transmission main path and the transmitting antenna, the signal output terminal of the frequency adjustment circuit is connected between the signal receiving link and the receiving antenna, and the frequency adjustment circuit is configured to adjust the frequency of the radio frequency transmission signal output from the signal transmission main path, and correspondingly, the signal receiving link is configured to acquire a sampled signal from the frequency adjustment circuit and obtain configuration information of a compensation signal based on the sampled signal. Alternatively, the frequency adjustment circuit is connected between the receiving mixer and the receiving local oscillator in the signal receiving link and is configured to adjust the frequency of the received local oscillator signal, the receiving local oscillator is used to generate the received local oscillator signal, the receiving mixer is used to demodulate the received signal using the received local oscillator signal, and the signal receiving link is connected between the signal transmission main path and the transmitting antenna and is configured to acquire a sampled signal from the signal transmission main path, process the sampled signal using the signal output from the frequency adjustment circuit, and obtain configuration information of a compensation signal.

[0030] For example, if the radio frequency transmission signal transmitted in the signal transmission main path is a two-channel orthogonal signal, the signal calibration link is a signal receiving link that supports the processing of echo signals containing the two-channel orthogonal signal.

[0031] For example, if the radio frequency transmission signal transmitted in the signal transmission main path is a two-channel orthogonal signal, the signal calibration link is equipped with an orthogonal processing circuit and two-channel signal receiving links, and neither of the signal receiving links supports processing of orthogonal echo signals. The orthogonal processing circuit is configured to perform radio frequency processing on the received orthogonal signal to obtain two-channel signals and transmit each of the two-channel signals to the two-channel signal receiving links.

[0032] Exemplary, the signal calibration link has a first input terminal connected between a voltage-current converter and a current switch in a transmitting mixer in the signal main path, a second input terminal connected between the signal main path and a transmitting antenna, and a signal output terminal connected to a compensation circuit in the signal main path, and acquires a signal in the signal main path from at least one of the first and second input terminals, and determines compensation information based on the acquired signal.

[0033] Exemplary, the signal calibration link includes a calibration demodulator, a multiplexer, and a calibration module. The calibration demodulator is configured to acquire a signal from the second input terminal in the main signal transmission path and perform demodulation. The multiplexer has two signal input terminals and one signal input terminal, one of which is connected between a voltage-current converter and a current switch in the transmit mixer, and the other signal input terminal is connected to the signal input terminal of the calibration demodulator and is configured to output a signal corresponding to one of the two signal input terminals. The calibration module is configured to determine the compensation information based on the signal output from the multiplexer.

[0034] For example, if the radio frequency transmission signal transmitted in the signal transmission main path is a two-channel orthogonal signal, then the acquired signal transmitted in the signal calibration link is also a two-channel orthogonal signal.

[0035] Exemplary, the acquired signal transmitted over the signal calibration link is a two-channel quadrature signal, and the signal calibration link further includes an analog-to-digital converter, which is connected between the signal acquisition circuit and the calibration module and is configured to perform analog-to-digital conversion on the acquired signal output from the signal acquisition circuit.

[0036] Embodiments of the present disclosure further provide a signal transmission link, comprising a signal transmission link and a signal reception link as described in any embodiment of the present disclosure, wherein the signal reception link comprises a receiving mixer, an analog-to-digital converter and a digital signal processing module, the receiving mixer configured to downconvert a received echo signal based on a received received local oscillator signal to obtain an analog intermediate frequency signal, the analog-to-digital converter configured to convert the received intermediate frequency signal from analog to digital to obtain a digital intermediate frequency signal, and the digital signal processing module configured to process the digital intermediate frequency signal to obtain a target parameter, wherein the echo signal is a signal formed by the reflection and / or scattering of a signal transmitted by the signal transmission link by an object.

[0037] For example, the receiving mixer is a real-number mixer and the analog-to-digital converter is a real-number analog-to-digital converter, or the receiving mixer is a quadrature mixer and the analog-to-digital converter is a quadrature analog-to-digital converter.

[0038] For example, the received local oscillator signal is a frequency sweep signal, or the received local oscillator signal is a monophonic signal.

[0039] Embodiments of the present disclosure further provide a signal calibration link, which includes a signal transmission / reception link as described in any embodiment of the present disclosure, wherein the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link, and the signal receiving link is configured to calibrate the signal transmitting link.

[0040] For example, there is a preset difference frequency between the local oscillator signal of the signal receiving link and the local oscillator signal of the signal transmitting link.

[0041] Exemplary, the system further includes a BIST module, which is installed between a local oscillator signal source and a receiving mixer, and is configured to mix the received local oscillator signal based on a preset frequency offset signal such that there is a preset difference frequency between the local oscillator signal received by the receiving mixer and the local oscillator signal of the signal transmission link.

[0042] Embodiments of the present disclosure further provide a signal calibration link comprising a signal transmission / reception link and a BIST module as described in any embodiment of the present disclosure, wherein the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link via the BIST module, and the signal receiving link is configured to calibrate the signal transmitting link.

[0043] Embodiments of the present disclosure further provide a signal calibration link comprising two signal receiving links, a BIST module, an auxiliary circuit unit, and a signal transmitting link as described in any embodiment of the present disclosure, wherein each of the signal receiving links comprises a real number mixer, a real number analog-to-digital converter, and a digital signal processing module, wherein the real number mixer is configured to downconvert a received echo signal based on a received local oscillator signal to obtain an analog intermediate frequency signal, the real number analog-to-digital converter is configured to convert the received intermediate frequency signal from analog to digital to obtain a digital intermediate frequency signal, the digital signal processing module is configured to process the digital intermediate frequency signal to obtain a target parameter, the echo signal being a signal formed by the reflection and / or scattering of a signal transmitted by the signal transmitting link by an object, the receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection port of the signal transmitting link via the auxiliary circuit unit and the BIST module, and the signal receiving links are configured to calibrate the intermediate frequency portion of the signal transmitting link.

[0044] Embodiments of the present disclosure further provide a signal calibration link for a signal transmitter path, the signal transmitter path generates a radio frequency transmission signal after performing a compensation operation on a signal generated based on a compensation coefficient in order to achieve target detection and / or communication, the signal calibration link is configured to acquire current observation information of the signal transmitter path at the current compensation coefficient, and if the current observation information satisfies an iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link, otherwise the current compensation coefficient is repeated until the obtained observation information satisfies the iteration condition.

[0045] Exemplary, the compensation coefficient includes at least one of the following: a harmonic distortion compensation parameter, a local oscillation leakage compensation parameter, and an orthogonal unbalance compensation parameter.

[0046] For example, the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

[0047] For example, the integrated circuit is a millimeter-wave radar chip, and / or the radio frequency transmission signal is an FMCW signal.

[0048] For example, the current compensation coefficients sequentially include the initial compensation coefficient h(0), the first compensation coefficient h(1), ..., the (k-1)th compensation coefficient h(k-1), and the kth compensation coefficient h(k), where the kth compensation coefficient h(k) is determined based on the (k-1)th compensation coefficient h(k-1) and the (k-1)th observation information O(k-1). k is an integer greater than or equal to 2.

[0049] For example, if the difference between the initial observation information O(0) and the first observation information O(1) is greater than a preset difference threshold, the k-th compensation coefficient h(k) is determined based on the k-1 compensation coefficient h(k-1) and the k-1 observation information O(k-1). If the difference between the initial observation information O(0) and the first observation information O(1) is less than a preset difference threshold, the initial observation information O(0) is adjusted based on a preset phase adjustment amount to obtain an adjusted first compensation coefficient h(1). A new first observation information O(1) is then obtained based on the adjusted first compensation coefficient h(1). The k-th compensation coefficient h(k) is determined based on the new first observation information O(1), and the difference between the initial observation information O(0) and the new first observation information O(1) is greater than the aforementioned difference threshold.

[0050] For example, determining the k-th compensation coefficient h(k) based on the k-1st compensation coefficient h(k-1) and the k-1st observation information O(k-1) includes: determining the k-th compensation coefficient h(k) by iteratively calculating the sum of the k-1st compensation coefficient h(k-1) and the k-1st observation information O(k-1) based on updating the first compensation coefficient h(1) to the initial observation O(0) if the difference between the absolute value of the initial observation information O(0) and the absolute value of the first observation information O(1) is greater than a preset difference threshold; and determining the k-th compensation coefficient h(k) by iteratively calculating the difference between the k-1st compensation coefficient h(k-1) and the k-1st observation information O(k-1) based on updating the first compensation coefficient h(1) to the inverse of the initial observation O(0) if the difference between the absolute value of the first observation information O(1) and the absolute value of the initial observation information O(0) is greater than a preset difference threshold.

[0051] For example, determining the k-th compensation coefficient h(k) based on the new first observation information O(1) means that if the difference between the absolute value of the initial observation information O(0) and the absolute value of the new first observation information O(1) is greater than the difference threshold, the sum of the k-1 observation information O(k-1) and the adjusted k-1 observation information O(k-1) obtained by processing the k-1 observation information O(k-1) with the phase adjustment amount is iteratively calculated to determine the k-th compensation coefficient h(k). The method also includes, if the difference between the absolute value of the new first observation information O(1) and the absolute value of the initial observation information O(0) is greater than the difference value threshold, updating the first compensation coefficient h(1) to the inverse of the adjusted first compensation coefficient h(1), iteratively calculating the difference between the k-1 compensation coefficient h(k-1) and the adjusted k-1 observation information O(k-1) obtained by processing the k-1 observation information O(k-1) with the phase adjustment amount, and determining the k-th compensation coefficient h(k).

[0052] For example, the first compensation coefficient h(1) is determined based on the initial observation information O(0).

[0053] For example, the value of the initial compensation coefficient h(0) is 0.

[0054] Exemplary, the signal transmission main path includes a baseband module and a radio frequency module, the baseband module being used to generate a baseband signal. The radio frequency module is used to perform frequency shifting and / or phase shifting on the baseband signal using a transmitting local oscillator signal to form a radio frequency transmission signal. The observation information is obtained from the radio frequency module.

[0055] Exemplary, the radio frequency module includes a transmit mixer and a power amplifier, the transmit mixer is used to frequency-shift and / or phase-shift the baseband signal using a transmit local oscillator signal, and the power amplifier is used to power-amplify the signal output from the transmit mixer, and the observation information is obtained from at least one of the signal output terminal of the transmit mixer, the signal output terminal of the power amplifier, and the signal output terminal of a voltage-current converter in the transmit mixer.

[0056] Embodiments of the present disclosure further provide a signal calibration link for a signal transmission main path, the signal transmission main path generates a radio frequency transmission signal after performing a compensation operation on a signal generated based on a compensation coefficient in order to achieve target detection and / or communication, the signal calibration link is configured to determine initial observation information O(0), first observation information O(1), and second observation information O(2) corresponding to the signal transmission main path under the conditions of initial compensation coefficients h(0), first compensation coefficient h(1), and second compensation coefficient h(2) with different numerical values, and to determine a third compensation coefficient h(3) as a compensation coefficient used in the compensation operation of the signal transmission link using the initial observation information O(0), first observation information O(1), and second observation information O(2).

[0057] Exemplary, the compensation coefficient includes at least one of the following: a harmonic distortion compensation parameter, a local oscillation leakage compensation parameter, and an orthogonal unbalance compensation parameter.

[0058] For example, the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

[0059] For example, the integrated circuit is a millimeter-wave radar chip, and / or the radio frequency transmission signal is an FMCW signal.

[0060] For example, determining the third compensation coefficient h(3) using the initial observation information O(0), the first observation information O(1), and the second observation information O(2) includes performing normalization on the first observation information O(1) and the second observation information O(2) using the initial observation information O(0), and determining the third compensation coefficient h(3) using the result of the normalization process, provided that the third observation information O(3) is 0.

[0061] For example, determining the third compensation coefficient h(3) using the result of the normalization process, provided that the third observation information O(3) is 0, is equivalent to determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2, provided that the third observation information O(3) is 0, where the first ratio d1 is the ratio of the first difference value to the initial observation information O(0), and the first difference value is the ratio of the initial observation information O(0) to the first The method includes: the difference value from the observation information O(1), the second ratio d2 being the ratio of the second difference value to the initial observation information O(0), the second difference value being the difference value between the initial observation information O(0) and the second observation information O(2); calculating the product of the first coefficient x1 and the second coefficient x2 to obtain the first multiplication result; calculating the product of the second coefficient x2 and the second compensation coefficient h(2) to obtain the second multiplication result; and determining the third compensation coefficient h(3) based on the first and second multiplication results.

[0062] Exemplary, determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2 includes constructing an observation equation and determining the first coefficient x1 and the second coefficient x2 in the observation equation. The formula for calculating the observation equation is a 1x2 second matrix obtained by calculating the product of the inverse of a 2x2 matrix and a 1x2 first matrix, where the first row of the 2x2 matrix records the real and imaginary parts of the first ratio d1, and the second row records the real and imaginary parts of the second ratio d2, where the value of the first row of the first matrix is ​​1 and the value of the second row is 0, where the first row of the second matrix is ​​the value of the first coefficient x1 and the second row is the value of the second coefficient x2.

[0063] For example, determining a third compensation coefficient h(3) based on the first and second multiplication results includes calculating the sum of the first and second multiplication results to obtain the third compensation coefficient h(3).

[0064] For example, the first compensation coefficient h(1) and the second compensation coefficient h(2) are determined based on the initial observation information O(0). For example, the value of the initial compensation coefficient h(0) is 0.

[0065] Exemplary, the signal transmission main path includes a baseband module and a radio frequency module, the baseband module being used to generate a baseband signal. The radio frequency module is used to perform frequency shifting and / or phase shifting on the baseband signal using a transmitting local oscillator signal to form a radio frequency transmission signal. The observation information is obtained from the radio frequency module.

[0066] Exemplary, the radio frequency module includes a transmit mixer and a power amplifier, the transmit mixer is used to frequency-shift and / or phase-shift the baseband signal using a transmit local oscillator signal, and the power amplifier is used to power-amplify the signal output from the transmit mixer, and the observation information is obtained from at least one of the signal output terminal of the transmit mixer, the signal output terminal of the power amplifier, and the signal output terminal of a voltage-current converter in the transmit mixer.

[0067] Embodiments of the present disclosure further provide a signal compensation link comprising a signal transmission link and a compensation unit as described in any embodiment of the present disclosure, wherein the compensation unit is configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion of the signal transmission link.

[0068] Exemplary, the compensation unit is configured to compensate the signal transmission link based on calibration data obtained by a signal calibration link described in any embodiment of the present disclosure.

[0069] For example, in order to perform real-time calibration operations between operations of the transmission link, the signal reception link in the signal calibration link is integrated as an auxiliary calibration module in the vicinity of the transmission link to be calibrated.

[0070] Embodiments of the present disclosure further provide a method for compensating for unequal lengths of transmission lines, which is applied to an antenna array of an electromagnetic wave sensor having at least two signal links, and which includes taking the signal link with the shortest transmission line among the at least two signal links as the reference link, obtaining the delay difference of each of the remaining transmission links with respect to the reference link, and performing unequal length compensation for transmission lines in the digital domain for the antenna array based on the delay difference.

[0071] Exemplary examples include unequal feedline lengths between different transmit links or between different receive links, between the local oscillator LO and the mixer in each transmit channel, and / or between the PA of each transmit channel and their respective transmit antennas.

[0072] Embodiments of the present disclosure further provide a signal calibration system applied to an electromagnetic wave sensor, the signal calibration system comprising a signal transmission link and an auxiliary link, the auxiliary link being integrated with the electromagnetic wave sensor adjacent to the signal transmission link, and the auxiliary link being configured to calibrate the signal transmission link in real time.

[0073] Exemplary, the signal transmission link includes a signal transmission link and / or a signal reception link, and the auxiliary link, corresponding to the signal transmission link, includes an auxiliary reception link and / or an auxiliary signal transmission link, wherein the auxiliary reception link is configured to calibrate the transmission link, and the auxiliary transmission link is configured to calibrate the reception link.

[0074] Exemplary, the signal receiving link includes a radio frequency portion and a baseband portion, and the auxiliary transmitting link includes, corresponding to the signal receiving link, a radio frequency auxiliary transmitting unit and a baseband auxiliary transmitting unit, wherein the radio frequency auxiliary transmitting unit is configured to calibrate the radio frequency portion, and the baseband auxiliary transmitting unit is configured to calibrate the baseband portion.

[0075] For example, the radio frequency auxiliary transmitting unit includes an IQ device, and when calibrating the signal receiving link using the auxiliary transmitting link, the baseband portion is calibrated using the baseband auxiliary transmitting unit, the radio frequency auxiliary transmitting unit is calibrated using the calibrated baseband portion, and then the radio frequency portion is calibrated using the calibrated radio frequency auxiliary transmitting unit.

[0076] Exemplary, the auxiliary receiving link includes the auxiliary receiving unit and a calibration receiving unit, the auxiliary receiving unit includes an IQ device, the calibration receiving unit is configured to calibrate the auxiliary receiving unit, and the calibrated auxiliary receiving unit is configured to calibrate the transmission link.

[0077] Embodiments of the present disclosure further provide an IQ mixer comprising an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit, wherein the I-branch mixing unit is configured to output an I-channel signal, the Q-branch mixing unit is configured to output a Q-channel signal, and the transformer unit is configured to magnetically couple the I-channel signal and the Q-channel signal to synthesize an IQ-mixed output signal.

[0078] Exemplary, the transformer unit has a three-winding transformer structure, which includes two branch inductances and one magnetically coupled inductance, with one branch inductance connected in series between the output terminals of the I branch mixing unit and one branch inductance connected in series between the output terminals of the Q branch mixing unit, and the magnetically coupled inductance is located between the two branch inductances.

[0079] Exemplary, the IQ mixer is applied to a digital phase shifter in a signal transmission link as described in any embodiment of the present disclosure, and / or to a signal calibration link as described in any embodiment of the present disclosure, and / or to a signal calibration system as described in any embodiment of the present disclosure.

[0080] Embodiments of the present disclosure further provide an integrated circuit comprising a sequentially connected radio frequency module, an analog signal processing module, and a digital signal processing module, wherein the radio frequency module is used to generate a radio frequency transmission signal and to receive a radio frequency reception signal; the analog signal processing module performs down-conversion processing on the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module performs analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; the radio frequency module comprises a signal transmission link, a signal transmission / reception link, a signal calibration link, a signal compensation link, a signal calibration system, and / or an IQ mixer as described in any embodiment of the present disclosure; and / or the digital signal processing module performs compensation in the digital domain based on a feed line inequality compensation method as described in any embodiment of the present disclosure.

[0081] Exemplary, the integrated circuit further includes a data processing module, which processes the digital signals to enable target detection and / or wireless communication. Exemplary, the integrated circuit is a chip such as millimeter wave or UWB.

[0082] For example, the radio frequency received signal is an echo signal formed when the radio frequency transmitted signal is transmitted and / or scattered by a target, and the integrated circuit is a sensor chip.

[0083] Embodiments of the present disclosure further provide an electromagnetic wave sensor comprising a carrier, an integrated circuit according to any embodiment of the present disclosure provided on the carrier, and an antenna, wherein the antenna is provided on the carrier, or the antenna is integrated with the integrated circuit as a single device provided on the carrier, and the integrated circuit is connected to the antenna and transmits the radio frequency transmission signal and / or receives the radio frequency reception signal.

[0084] Embodiments of the present disclosure further provide an apparatus comprising a main body of the apparatus and an electromagnetic wave sensor provided on the main body of the apparatus as described in any embodiment of the present disclosure, wherein the electromagnetic wave sensor is used for target detection and / or communication in order to provide reference information for the operation of the main body of the apparatus.

[0085] Embodiments of the present disclosure further provide a non-temporary computer-readable storage medium on which computer-readable instructions are stored, and which, when the computer-readable instructions are executed by a processor, causes the processor to execute a power line inequality compensation method described in any embodiment of the present disclosure.

[0086] The above-mentioned and other purposes, features, and advantages of this disclosure will become more apparent with reference to the following drawings and descriptions of embodiments of this disclosure. [Brief explanation of the drawing]

[0087] [Figure 1A] This is a simplified schematic diagram of the signal transmission link in an analog phase shifter architecture. [Figure 1B] Figure 1A is a simplified schematic diagram of an analog phase shifter in a signal transmission link. [Figure 2A] This is a simplified schematic diagram of the signal transmission link of the digital phase shifter architecture in an embodiment of the present disclosure. [Figure 2B] This is a simplified schematic diagram of another signal transmission link of the digital phase shifter architecture in an embodiment of the present disclosure. [Figure 2C] This is a schematic diagram of the waveforms of an FMCW transmit signal and echo signal modulated with a sawtooth wave. [Figure 3A] This is a schematic diagram of the signal model for transmitter orthogonal modulation under orthogonal unbalance. [Figure 3B] This is a schematic diagram of the signal generated by a real-mixer. [Figure 3C] This is a schematic diagram of third-harmonic distortion. [Figure 4A] This is a schematic diagram of the structure of a signal transmission link according to an embodiment of the present disclosure. [Figure 4B]This is a schematic diagram of orthogonal disequilibrium compensation according to an embodiment of the present disclosure. [Figure 4C] This is a first schematic diagram of harmonic distortion compensation according to an embodiment of the present disclosure. [Figure 4D] This is a second schematic diagram of harmonic distortion compensation according to an embodiment of the present disclosure. [Figure 5A] This is a schematic diagram of the digital phase shifter architecture in a signal transmission link in an embodiment of the present disclosure. [Figure 5B] This is a schematic diagram of a transmission link including a specific compensation unit in an embodiment of the present disclosure. [Figure 6A] This is a schematic diagram of the structure of a signal transmission and reception link according to an embodiment of the present disclosure. [Figure 6B] This is a schematic diagram of the structure of another signal transmission and reception link according to an exemplary embodiment of the present disclosure. [Figure 7A] This is a schematic diagram of a transmit / receive link including a TX IQ Mod, a BIST IQ Mod, and an RX Real Mixer in an embodiment of the present disclosure. [Figure 7B] This is a schematic diagram of a transmit / receive link including TX IQ Mod, RX IQ De-Mod, and LO Freq Diff in an embodiment of the present disclosure. [Figure 8] This is a schematic diagram of a transmit / receive link combining BIST based on the structure shown in Figure 7B in an embodiment of the present disclosure. [Figure 9] This is a schematic diagram of a transmit / receive link including TX IQ Mod, BIST IQ Mod, and RX IQ De-Mod in an embodiment of the present disclosure. [Figure 10] This is a schematic diagram of the auxiliary circuit and the transmit / receive link including the BIST IQ Mod in an embodiment of the present disclosure. [Figure 11] This is a schematic diagram of an auxiliary circuit and another transmit / receive link including a BIST IQ Mod in an embodiment of the present disclosure. [Figure 12A] This is a schematic diagram of the structure of the mixer in the embodiment of the present disclosure. [Figure 12B] This is a schematic diagram of the structure of the compensation unit in the transmitter according to an embodiment of the present disclosure. [Figure 13A]This is a schematic diagram of a digital pre-compensated HD3 architecture based on a cube module in an embodiment of the present disclosure. [Figure 13B] This is a schematic diagram of a digital pre-compensated HD3 architecture based on a frequency multiplier module in an embodiment of the present disclosure. [Figure 13C] This is a schematic diagram of the calibration compensation of a transmit link based on a digital phase shifter architecture in an embodiment of the present disclosure. [Figure 14A] This is a flowchart of a signal transmission method according to an embodiment of the present disclosure. [Figure 14B] This is a diagram illustrating the principle of the compensation coefficient determination method according to the embodiments of this disclosure. [Figure 15] This is a schematic diagram of a transmission link having at least two transmission channels in an embodiment of the present disclosure. [Figure 16] This is a schematic diagram of the structure of a digital LO signal generator in an embodiment of the present disclosure. [Figure 17] This is a schematic diagram of the structure of a transmission link including an unequal length compensation module for the power supply line in an embodiment of the present disclosure. [Figure 18] This is a schematic diagram of the calibration and compensation of the transmit / receive link using the auxiliary circuit in an embodiment of the present disclosure. [Figure 19] This is a schematic diagram of the calibration and compensation of the receiving link using the auxiliary transmitting circuit in an embodiment of the present disclosure. [Figure 20] This is a schematic diagram of the calibration and compensation of the transmission link using the auxiliary receiving circuit in an embodiment of the present disclosure. [Figure 21] This is a schematic diagram of the structure of an auxiliary circuit in an embodiment of the present disclosure. [Figure 22] This is a schematic diagram of the structure of another auxiliary circuit in an embodiment of the present disclosure. [Figure 23] This is a schematic diagram of the circuit module of the IQ Mixer in an embodiment of the present disclosure. [Figure 24] This is a schematic diagram of the structure of the IQ Mixer in an embodiment of the present disclosure. [Figure 25] This is a schematic diagram corresponding to the structure shown in Figure 24. [Figure 26]This is a schematic diagram of the physical structure of another IQ Mixer in an embodiment of the present disclosure. [Figure 27] This is a flowchart of another signal transmission method according to an embodiment of the present disclosure. [Figure 28] This is a flowchart of yet another signal transmission method according to an embodiment of the present disclosure. [Figure 29] This is a flowchart of yet another signal transmission method according to an embodiment of the present disclosure. [Modes for carrying out the invention]

[0088] Hereinafter, the disclosure will be described more comprehensively with reference to the relevant drawings to facilitate understanding of the disclosure. Preferred embodiments of the disclosure are shown in the drawings. However, the disclosure can be implemented in different forms and is not limited to the embodiments described herein. Rather, these embodiments are provided to provide a more thorough and comprehensive understanding of the disclosure.

[0089] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as those commonly understood by those skilled in the art. Terms used in this disclosure are for illustrative purposes only and are not intended to limit the disclosure.

[0090] The technical proposal of this disclosure will be described in detail below with reference to the drawings.

[0091] Figure 1A is a simplified schematic diagram of a signal transmission link in an analog phase shifter architecture, and Figure 1B is a simplified schematic diagram of the analog phase shifter in the signal transmission link shown in Figure 1A.

[0092] As shown in Figure 1A, when the sensor transmits a signal, it generates an LO signal (e.g., a frequency sweep signal in the 77 GHz band) for one transmission link using a signal generator 11 consisting of, for example, a phase-locked loop (PLL), which may also be, for example, an FMCW signal. The analog phase shifter (Analog PS) 12 performs a phase shift operation on the received LO signal and then radiates it into a predetermined spatial area via the transmitting antenna 13 to perform operations such as target detection and measurement.

[0093] Selectively, in the transmit link structure shown in Figure 1A, the corresponding analog phase shifter architecture may be as shown in Figure 1B, and its specific phase shift principle may be as shown by the following equation.

[0094]

number

[0095] Furthermore, the above analog phase shifter architecture can be implemented using a delay line unit (DRI) method, that is, by utilizing the narrow-band assumption of the signal and shifting the phase using a delay method, the principle of which is shown in the following equation.

[0096]

number

[0097] The above analog phase shifters have low phase modulation resolution and accuracy, making them unsuitable for meeting current sensor needs. While calibration can improve their phase modulation resolution and accuracy, it requires offline calibration for analog architecture phase shifters, significantly increasing the difficulty and complexity of process implementation and mass production. Furthermore, offline calibration results are not always accurate when working under different conditions (temperature). Additionally, analog phase shifters have relatively serious problems such as large area, high losses, and issues with stability and channel coupling. When multi-antenna phase shifters work together, the performance of the phase shifters affects each other, further degrading system performance.

[0098] Based on this, the inventors of this disclosure creatively propose a signal transmission link for a digital phase shifter architecture that effectively improves phase modulation accuracy and precision while avoiding non-online calibration operations for link and devices such as phase shifters in the transmission link, and further reducing the complexity and difficulty of process implementation. In addition, it can effectively reduce the transmission link area and loss of the phase shift architecture, improve system stability, and reduce channel coupling.

[0099] Figure 2A is a simplified schematic diagram of a signal transmission link of the digital phase shifter architecture in an embodiment of the present disclosure, Figure 2B is a simplified schematic diagram of another signal transmission link of the digital phase shifter architecture in an embodiment of the present disclosure, and Figure 2C is a schematic waveform diagram of a sawtooth-modulated FMCW transmission signal and echo signal.

[0100] Embodiments of the present disclosure provide a signal transmission link applicable to an electromagnetic wave sensor, the transmission link may include an analog signal source and a digital phase shifter, the analog signal source configured to supply an initial analog signal (e.g., an LO signal), the digital phase shifter configured to generate a phase shift signal in the digital domain, and the digital phase shifter can further phase shift the initial analog signal based on the generated phase shift signal in order to perform a preset phase shift operation on the initial analog signal.

[0101] As shown in Figure 2A, in some selectable embodiments, the signal transmission link of the digital phase shifter architecture may include an analog signal source 21, a digital phase shifter (Digital PS) 22, and a transmitting antenna 23, etc. That is, the analog signal source 21 is configured to supply an LO signal, and the digital phase shifter 22 is configured to perform a preset phase shift operation on the received LO signal so that the phase-shifted LO signal is radiated through the transmitting antenna 23 into a preset spatial region. The analog signal source 21 may have an architecture including a phase-locked loop PLL and can supply electromagnetic wave (e.g., laser, microwave, etc.) signals. The analog signal source 21, the digital phase shifter 22, and the transmitting antenna 23 may be integrated as a single device or as separate components. For example, the analog signal source 21 and the digital phase shifter 22 may be integrated in a package to form an SoC chip, and the transmitting antenna 23 may be connected via a peripheral port of the chip and formed on a carrier such as a PCB board. Simultaneously, in some selective embodiments, the transmitting antenna 23 may also be integrated into the chip package to form an AiP or AoP, and have a chip structure of the packaged antenna.

[0102] As shown in Figure 2A, the digital phase shifter 22 in the embodiment of this disclosure may include a mixer 221, a digital-to-analog converter (i.e., DAC) 222, and a phase shift signal source (e.g., a digital baseband signal source) 223. That is, the phase shift signal source 223 is configured to supply a digital phase shift signal. The digital-to-analog converter 222 is configured to convert the received digital phase shift signal to an analog phase shift signal. The mixer 221 is configured to perform a phase shift operation set for the electromagnetic wave signal using the digital phase shift signal by performing a mixing operation on the received analog phase shift signal and the electromagnetic wave signal from the received analog signal source 21. Selectively, when supplying a frequency sweep signal, for example, an FMCW laser signal or an FMCW microwave signal, the signal transmission link supplies a frequency sweep electromagnetic wave signal based on the analog signal source 21 and / or a frequency sweep digital phase shift signal based on the phase shift signal source 223, thereby enabling the output of a frequency sweep continuous wave signal after mixing by the mixer 221.

[0103] In some selectable embodiments, based on the structure shown in Figure 2A, the analog signal source 21 is configured to supply an FMCW signal in the centimeter-wave or millimeter-wave band in the microwave (e.g., bands such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.), the phase shift signal source 223 is configured to supply a digital phase shift signal at the MHz level (e.g., 3 MHz to 5 MHz, e.g., 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog converter 222 converts the MHz-level digital phase shift signal digital-to-analog to obtain an analog phase shift signal in the corresponding frequency range, and the mixer 221 is configured to perform up-mixing or down-mixing operations on the received millimeter-wave FMCW signal based on the received fixed-band analog phase shift signal to achieve a preset phase shift operation of the FMCW signal.

[0104] In some selectable embodiments, the centimeter-wave signals in the 3.1 GHz band may include frequencies from 3.1 GHz to 10.6 GHz, such as 3.1 GHz, 5 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc. The millimeter-wave signals in the 77 GHz band may include signals from 76 GHz to 81 GHz, such as frequency sweep signals from 76 GHz to 77 GHz, 77 GHz to 79 GHz, 79 GHz to 81 GHz, etc., or fixed-band signals such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0105] Based on the configuration shown in Figure 2A, the phase-shift signal source 223 supplies a digital signal, and to further adapt the signal characteristics, the mixer 221 may be installed as an IQ Mixer, and the digital-to-analog converter 222 is an IQ DAC. At the same time, the phase-shift signal source 223 is configured to supply a digital baseband signal source (DDFS) for phase shifting and / or to supply a corresponding source signal as a Waveform Control.

[0106] As shown in Figure 2B, an embodiment of the present disclosure provides a signal transmission link applied to a radar system and includes a transmit baseband digital module 201 (including the aforementioned phase-shift signal source 223), a digital-to-analog converter (DAC) module 202, a transmit local oscillator 203, and a transmit quadrature modulator 204, the digital-to-analog converter module 202 including two identical digital-to-analog converters. The transmit baseband digital module 201 is configured to generate two orthogonal transmit digital baseband signals (i.e., digital phase-shift signals) and to transmit each of the two orthogonal transmit digital baseband signals to one digital-to-analog converter module 202. The digital-to-analog converter module 202 is configured to convert the two orthogonal transmit digital baseband signals into two transmit analog baseband signals (i.e., analog phase-shift signals). The transmit local oscillator 203 is configured to supply the transmit local oscillator signal TX_LO (i.e., the initial analog signal). The transmitting quadrature modulator 204 is configured to perform a phase shift operation on the transmitting local oscillator signal TX_LO while frequency shifting it based on two channels of transmitting analog baseband signals, thereby forming an FMCW radio frequency transmitting signal after a predetermined phase shift.

[0107] In embodiments of this disclosure, the transmit digital baseband signal supplied by the transmit baseband digital module 201 includes preset phase information. The digital-to-analog conversion module 202 performs digital-to-analog conversion on the received transmit digital baseband signal, converting the transmit digital baseband signal into a transmit analog baseband signal (for example, without changing the phase information). The transmit quadrature modulator 204 mixes the received transmit analog baseband signal and the transmit local oscillator signal TX_LO generated by the transmit local oscillator 203, and performs a preset phase shift operation while frequency shifting the transmit local oscillator signal based on the transmit analog baseband signal to form a predetermined phase-shifted FMCW radio frequency transmit signal.

[0108] The signal transmission link in the embodiment of this disclosure comprises a digital phase shifter architecture using a transmitting baseband digital module 201, a digital-to-analog conversion module 202, and a transmitting quadrature modulator 204. The baseband signal of this architecture is generated in the digital domain and has better orthogonality and lower side lobes, so its phase shift can be generated with great accuracy, resulting in higher phase modulation accuracy. This enables a high-precision digital phase shift-enabled automotive radar system, reduces the demand for separation between antennas, and simultaneously offers the advantages of low link loss, low cost, and no need for offline calibration. It can support more flexible wave transmission solutions such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0109] In the embodiments of this disclosure, the transmitting baseband digital module 201 supplies a digital signal, and to further adapt the signal characteristics, the transmitting modulator is configured as an IQ modulator, and the digital-to-analog conversion module 202 is configured as an IQ DAC.

[0110] In embodiments of this disclosure, the transmitting local oscillator 203 may have an architecture including a phase-locked loop (PLL) and can supply electromagnetic wave (e.g., laser, microwave, etc.) signals.

[0111] In some exemplary embodiments, the signal transmission link further includes a power amplifier (PA) 205, which is configured to power amplified the phase-shifted radio frequency signal and output the amplified signal to the transmitting antenna.

[0112] In some exemplary embodiments, the signal transmission link further includes a transmitting antenna 206, which is configured to radiate the amplified signal into a predetermined spatial region.

[0113] In embodiments of this disclosure, the signal amplified by the power amplifier 205 may be radiated into a predetermined spatial region via a transmitting antenna 206 that is either integrally packaged or placed externally. That is, the transmitting local oscillator 203, digital phase shifter, and transmitting antenna 206 may be integrated as a single device or they may be separate components. For example, the transmitting local oscillator 203 and digital phase shifter may be integrated in a package to form an SoC chip, and the transmitting antenna 206 may be connected via peripheral ports of the chip and formed on a carrier such as a PCB board. At the same time, in some selective embodiments, the transmitting antenna 23 may also be integrated into the chip package to form an AiP or AoP, and have a chip structure of a packaged antenna.

[0114] In some exemplary embodiments, the frequency bandwidth of the frequency sweep signal is 2 GHz or greater. For example, the electromagnetic wave of the transmitted signal emitted by the transmitting antenna of a frequency-modulated continuous wave radar system is a high-frequency modulated continuous wave, and the echo signal received by the receiving antenna of a frequency-modulated continuous wave radar system is an electromagnetic wave reflected and scattered from an object. Figure 2C shows schematic waveforms of exemplary FMCW transmitted and echo signals. As shown in Figure 2C, the frequencies of the transmitted and echo signals change regularly over time. Frequency-modulated continuous waves are generally sawtooth, triangular, etc., and this disclosure describes the sawtooth shape as an example, where the electromagnetic wave within each frequency modulation period T is called a chirp, and the frequency of the signal in each chirp increases linearly over time. In embodiments of this disclosure, the bandwidth range B of a chirp is 2 GHz or greater.

[0115] In some exemplary embodiments, the transmitted digital baseband signal is a monotone signal, and the transmitted local oscillator signal is a frequency sweep signal. In embodiments of the present disclosure, the transmitting local oscillator 203 is configured to supply an FMCW signal in the centimeter-wave or millimeter-wave band in the microwave (e.g., bands such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.), the transmitting baseband digital module 201 is configured to supply a monotone transmitted digital baseband signal at the MHz level (e.g., 3 MHz to 5 MHz, e.g., 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 202 digital-to-analog converts the MHz-level monotone transmitted digital baseband signal to obtain a monotone transmitted analog baseband signal in the corresponding frequency range, and the transmitting quadrature modulator 204 is configured to upmix or downmix the received millimeter-wave band FMCW signal based on the received monotone transmitted analog baseband signal to achieve a preset phase shift operation of the FMCW signal.

[0116] For example, a 3.1 GHz band FMCW signal may include a frequency sweep signal between 3.1 GHz and 10.6 GHz, for example, 7.163 to 8.812 GHz. A 77 GHz band FMCW signal may include a frequency sweep signal between 76 GHz and 81 GHz, or frequency sweep signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, or 79 GHz to 81 GHz.

[0117] In other exemplary embodiments, the transmitted digital baseband signal may be a frequency sweep signal, and the transmitted local oscillator signal may be a monophonic signal.

[0118] In embodiments of this disclosure, the transmitting local oscillator 203 is configured to supply a monophonic transmitting local oscillator signal in the centimeter-wave or millimeter-wave band in the microwave (e.g., bands such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.), the transmitting baseband digital module 201 is configured to supply a transmitting digital baseband FMCW signal at the MHz level (e.g., 3 MHz to 5 MHz, e.g., 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 202 converts the MHz-level transmitting digital baseband FMCW signal from digital to analog to obtain a transmitting analog baseband FMCW signal in the corresponding frequency range, and the transmitting quadrature modulator 204 is configured to perform up-mixing or down-mixing operations on the received centimeter-wave or millimeter-wave monophonic transmitting local oscillator signal based on the received transmitting analog baseband FMCW signal to realize preset phase shift and frequency sweep operations on the monophonic transmitting local oscillator signal.

[0119] For example, a monophonic local oscillator signal in the 3.1 GHz band may be a monophonic analog signal with a fixed bandwidth such as 3.1 GHz, 5 GHz, 6 GHz, 8 GHz, or 10.6 GHz. A monophonic local oscillator signal in the 77 GHz band may be a monophonic analog signal with a fixed bandwidth such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, or 81 GHz.

[0120] In some exemplary embodiments, the signal transmission link further includes a low-pass filter (LPF) 207, which is installed between the digital-to-analog conversion module 202 and the transmit quadrature modulator 204, and which low-pass filters the transmit analog baseband signal output from the digital-to-analog conversion module 202 before outputting it to the transmit quadrature modulator 204.

[0121] As shown in Figure 2B, the transmitting baseband digital module 201 generates two orthogonal digital baseband signals, namely an I-channel digital baseband signal and a Q-channel digital baseband signal. The generated digital baseband signals are transmitted to the digital-to-analog conversion module 202 (which contains two identical DACs, i.e., IQ DACs) to obtain two-channel analog baseband signals. These two-channel analog baseband signals are then input to a low-pass filter 207 to filter out out-of-band noise signals, and then orthogonally modulated by a transmitting quadrature modulator 204 to obtain a modulated radio frequency signal. This modulated radio frequency signal is then radiated by a power amplifier 205 and a transmitting antenna 206.

[0122] In some exemplary embodiments, the signal transmission link may further include a Direct Digital Frequency Synthesizer (DDFS) (not shown in Figure 2B) installed between the transmit baseband digital module 201 and the digital-to-analog conversion module 202, the Direct Digital Frequency Synthesizer being configured to implement at least one of several signal waveforms and wave transmission methods, such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), and Digital IF (Digital Intermediate Frequency), based on the received source signal, in order to enable flexible configuration of the signal transmission form and transmitted waveform.

[0123] Figure 3A is a schematic diagram of the signal model of the transmitter quadrature modulation at the time of orthogonal imbalance. As shown in Figure 3A, DSP is a digital signal processor, DAC is a digital-to-analog converter, LPF is a low-pass filter, and LO is a local oscillation signal source. A I (f) is the frequency domain expression of the LPF output signal of the I channel, E I (f) is the baseband frequency response of the I channel, A‘ I (f) is the frequency domain expression of the baseband output signal of the I channel, S I (f) is the frequency domain expression of the modulator output signal of the I channel, G I (f) is the radio frequency response of the I channel, S’ I (f) is the frequency domain expression of the radio frequency output signal of the I channel, α I is the amplitude gain of the mixer of the I channel, φ I is the phase response of the mixer of the I channel. A Q (f) is the frequency domain expression of the LPF output signal of the Q channel, E Q (f) is the baseband frequency response of the Q channel, A‘ Q (f) is the frequency domain expression of the baseband output signal of the Q channel, S Q (f) is the frequency domain expression of the modulator output signal of the Q channel, G Q (f) is the radio frequency response of the Q channel, S’ Q (f) is the frequency domain expression of the radio frequency output signal of the Q channel, α Q is the amplitude gain of the mixer of the Q channel φ Q is the phase response of the mixer of the Q channel, U RF (f) is the frequency domain expression of the transmitted modulated signal. In the embodiments of the present disclosure, the radio frequency includes all circuits passed from the mixer to the radio frequency signal, and the baseband includes all circuits passed from the DAC to the mixer.

[0124] Based on the signal model shown in Figure 3A, the output U RF (f) of the quadrature demodulator can be expressed as follows.

[0125]

number

[0126] Therefore, the baseband equivalent model of the quadrature tuner is U RF (f) = A(f)V1(f) + A * It may also be (-f)V2(f), where A(f)V1(f) is of frequency f m This is a desired signal in the vicinity, A * (-f)V2(f) is the frequency -f m This is a mirror signal in the vicinity. The relative image ratio (RIR) is used to measure the ability of the quadrature mixer to suppress unwanted mirror frequency components. Therefore, the definition of RIR is the ratio of the power of the desired signal to the power of the mirror signal component.

[0127]

number

[0128] Figure 3B is a schematic diagram of the signal generated by the real-mixer. As shown in Figure 3B, F IN represents the input signal of the mixer, F OUT represents the output signal of the mixer, F LO This represents the local oscillator signal used in the mixer, and the input signal to the mixer has a frequency of F BB The baseband signal is a local oscillator signal used in a mixer, and its frequency is F LO It is a high-frequency signal, and the output signal of the mixer has a frequency of F LO ±F BB It is a signal.

[0129] An ideal mixer produces an output that is the product of two inputs. In terms of frequency, the output is F LO +F BB and F LO -F BB It should be, and does not include other terms. If any input is not in a driven state, there is no output. However, F BB and F LO Then, the real micsa generates some more energy. F BB The energy generated is far from the required output and is filtered by the RF device after being output by the mixer, so it can be ignored. BB Regardless of the energy generated, F LO The energy generated can be problematic because it is very close to, or within, the desired output signal, making it difficult or impossible to remove by filtering, as filtering also removes the desired signal. LO can also leak to the system output terminal through other means, such as through the power supply or across the silicon itself. How the local oscillator leaks can be called LO leakage (LOL).

[0130] Analog (wireless high-frequency) circuits can be approximated by linear models represented by small signals, but the nonlinearity of devices often gives rise to unpredictable, interesting, and important phenomena. The nonlinear input / output characteristics in memoryless systems can be approximated as follows:

[0131]

number

[0132] Figure 3C is a schematic diagram of third-harmonic distortion. As shown in Figure 3C, the baseband signal generated by the baseband processor is a baseband signal with frequency BB, which is processed by an IQ modulator to obtain the desired signal Wanted, the Miller signal IMG, and the third harmonic HD3. The signal output by the power amplifier PA includes the desired signal Wanted, the Miller signal IMG, and the third harmonic HD3.

[0133] The main source of HD3 is the third-order nonlinearity of the baseband analog device. As can be seen from the above derivation, the third-order harmonic component HD3 of the baseband BB signal is located at -3BB. After passing through the mixer, the baseband HD3 is upconverted to LO-3BB, i.e., C-IM3. After being amplified by the PA, it is difficult to distinguish the signal generated by modulation at high frequencies.

[0134] As can be seen from the above explanation, digital quadrature modulation signal transmission links face problems such as IQ imbalance, harmonic distortion, and LO leakage. When an ideal monophonic baseband signal passes through a quadrature upmixer, at high frequencies, multiple monophonic signals of different, non-ideal frequencies are generated. Therefore, this disclosure solves the problem of introduction due to the non-ideal characteristics of analog devices in digital quadrature modulation radar TX by utilizing the addition of a digital pre-compensation module at the transmitting end from the system architecture. This disclosure includes not only a pre-compensation module but also a calibration loop when designing the system architecture to achieve real-time calibration and compensation.

[0135] Figure 4A is a schematic diagram of the structure of a signal transmission link according to an embodiment of the present disclosure. As shown in Figure 4A, the signal transmission link includes a signal transmission main path and a signal calibration link integrated on the same integrated circuit, wherein the signal calibration link is configured to calibrate the signal transmission main path to acquire compensation information, and the signal transmission main path is configured to generate a radio frequency transmission signal after performing compensation operations based on the compensation information to achieve target detection and / or communication.

[0136] By integrating the signal calibration link into the integrated circuit including the transmitter path, the signal calibration link can perform calibration operations on the transmitter path in real time. Furthermore, the calibration operation of the calibration link does not change with changes in the operating environment of the transmitter path, allowing the transmitter path to obtain more accurate calibration information and improving the signal processing performance of the transmitter path.

[0137] Furthermore, the compensation information includes at least one of the following: a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal unbalance compensation parameter. The compensation information generated by the signal calibration link can solve the problems of orthogonal unbalance, LO leakage, and harmonic distortion, thereby effectively improving the signal quality in the transmitter path of the digital phase shifter architecture.

[0138] In one embodiment, the integrated circuit is a millimeter-wave radar chip or an ultra-wideband (UWB) chip, and / or the radio frequency transmission signal is an FMCW signal.

[0139] The signal transmission main path may include a first signal source and a phase shifter. The first signal source is configured to generate a first analog signal, and the phase shifter is configured to frequency shift and / or phase shift the first analog signal in order to form a radio frequency transmission signal.

[0140] If the phase shifter has a non-orthogonal architecture, the phase shifter includes a second signal source and a transmit mixer, wherein the second signal source is configured to generate a second analog signal, and the transmit mixer is configured to mix the first analog signal and the second analog signal to form the radio frequency transmit signal. If the phase shifter has an orthogonal architecture, the phase shifter includes a second signal source, a digital-to-analog conversion module and a transmit mixer, wherein the second signal source is configured to generate a first digital signal, the digital-to-analog conversion module is configured to convert the first digital signal to a second analog signal, and the transmit mixer is configured to frequency shift and / or phase shift the first analog signal based on the second analog signal to form the radio frequency transmit signal.

[0141] The transmitter path further includes a compensation circuit, the signal input terminal of the compensation circuit is connected to the second signal source, and the signal input terminal is connected to the phase shifter, and the compensation circuit merges the compensation signal and the signal output from the second signal source and outputs the result.

[0142] If the signal output from the signal transmission path is a two-channel orthogonal signal, the compensation signal used in the compensation circuit is a two-channel orthogonal signal. If the signal output from the signal transmission path is not a two-channel orthogonal signal, the compensation signal used in the compensation circuit is a one-channel signal of the same type as the signal output from the second signal source, and the signal type is either a digital or analog signal.

[0143] The compensation circuit includes a compensation signal generator and an adder, the compensation signal generator being configured to generate the compensation signal, and the adder being connected to the compensation signal generator and the second signal source, and superimposing the signal output from the second signal source and the compensation signal output from the compensation signal generator.

[0144] Specifically, if the signal output from the signal transmission main path is a two-channel orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal unbalanced compensation signal unit, and a local oscillator leak compensation signal unit. If the signal output from the signal transmission main path is not a two-channel orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leak compensation signal unit. The compensation signal generated by the harmonic compensation signal unit is used to cancel the harmonic signals of the main frequency signal in the signal transmission main path. The compensation signal from the local oscillator leak compensation signal unit is used to compensate for the leak signals generated by the transmitting local oscillator signal in the signal transmission main path. The compensation signal generated by the orthogonal unbalanced compensation signal unit is used to compensate for the mirror signals of the main frequency signal in the signal transmission main path.

[0145] The compensation signal used in the orthogonal unbalanced compensation circuit has the same frequency, the same amplitude, and opposite phase between the mirror signal corresponding to the signal transmitting main path and the mirror signal corresponding to the desired signal generated by the signal transmitting main path.

[0146] In one exemplary embodiment, the compensation signal used in the orthogonal unbalanced compensation circuit is determined by the signal output from the second signal source and the complex conjugate signal, which is the inverted frequency of the signal output from the second signal source. The method for obtaining the compensation signal generated by the orthogonal unbalanced compensation circuit includes obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal, and calculating the difference between the signal output from the second signal source and the adjustment signal to obtain the compensation signal generated by the orthogonal unbalanced compensation circuit. The pre-compensation coefficient is determined based on the ratio of the amplitude of the desired signal to the amplitude of the mirror signal corresponding to the desired signal.

[0147] Figure 4B is a schematic diagram of orthogonal unbalance compensation according to an embodiment of the present disclosure. As shown in Figure 4B, a pre-compensation method can be employed to realize compensation for the TX orthogonal modulator, that is, to suppress the presence of undesirable mirror signals. hIQ is used as the pre-compensation coefficient and is applied to the complex conjugate signal A*(-f) of the baseband signal to be transmitted. The frequency domain expression of the pre-compensated baseband signal can be written as follows.

[0148]

number

[0149]

number

[0150] In one exemplary embodiment, the compensation signal generated by the local oscillator leak compensation signal unit is generated based on a leak signal corresponding to a first analog signal used in a phase shifter. The compensation signal generated by the local oscillator leak compensation signal unit has the same frequency and amplitude as the leak signal, but inverse phase.

[0151] Specifically, LOL cancellation (LO Leakage Correction, LOC) is achieved by generating a signal with the same amplitude as LOL but in the opposite phase, thus canceling it out. After obtaining the correct amplitude and phase of LOL, the cancellation signal can be generated by applying a DC offset to the transmitter input. A quadrature mixer structure allows for efficient generation of the cancellation signal. Since the mixer contains quadrature signals of LO, it can generate signals of any phase and amplitude at the frequency of LO. Geometric interpretation can be performed in the complex plane. The first row is the result of combining the time-domain I and Q channels, and the second row is a geometric interpretation from the angles of the signal spectra of the I and Q channels. When a DC bias is applied to the transmit signal, the output terminal of the mixer contains the required transmit signal and the required LOL cancellation signal. The intentionally generated cancellation signal LOC cancels out the unwanted LOL, leaving only the required transmit signal.

[0152]

number

[0153] In one exemplary embodiment, the compensation signal generated by the harmonic compensation signal unit has the same frequency, the same amplitude, and the opposite phase to the harmonic signal in the signal transmission main path.

[0154] Specifically, the harmonic compensation signal unit includes an n-th power module or an n-frequency signal generator, wherein the n-th power module's signal input terminal is connected to the signal output terminal of the second signal source, and it is used to obtain the compensation signal by generating a signal with a frequency n times the frequency of the signal output from the second signal source using the signal output from the second signal source. The n-frequency signal generator is used to obtain the compensation signal by generating a signal with a frequency n times the frequency of the signal output from the second signal source. The value of n is a positive integer. The value of n is an odd number. Furthermore, the value of n is 3.

[0155] Figures 4C and 4D are schematic diagrams of harmonic distortion compensation according to embodiments of the present disclosure. As shown in Figures 4C and 4D, similar to the principle for canceling LOL, there are two types of compensation methods for third-harmonic distortion: a compensation architecture based on an n-th power module and a compensation architecture based on an n-frequency multiplier.

[0156]

number

[0157] Regarding the signal transmission link of the digital phase shifter architecture in the embodiments of this disclosure, the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, generate an analog baseband signal by a DAC, and then modulate the transmission signal to a high frequency via a quadrature mixer. That is, the baseband signal of the architecture is generated in the digital domain and has better orthogonality and lower side lobes, so its phase shift can be generated with great accuracy, resulting in higher phase modulation accuracy.

[0158] In some optional embodiments, when a digital phase shifter architecture's signal transmission link is implemented using an RF LO frequency sweep to ensure the transmitted signal is an FMCW signal, a compensation unit can be added to the signal transmission link to address potential problems such as TX IQ imbalance due to IQ mismatch, signal leakage (e.g., TX LO Leakage), and harmonic distortion (abbreviated as HD). As shown in Figure 5A, by providing a compensation unit (TX compensation) between the TX DDFS and the IQ DAC, calibration and compensation operations can be performed on the signal transmission link of the digital phase shifter architecture, thereby achieving a solution to at least one of the above problems. Of these, HD resulting from a third-order nonlinearity in the baseband can simply be called HD3.

[0159] In some optional embodiments, as shown in Figures 5A to 5B, the TX compensation unit (TX compensation) may include at least one of the following: an LO leakage compensation unit (TX LO leakage compensation), an IQ imbalance compensation unit (TX IQ imbalance compensation), and an HD3 compensation unit (TX HD3 compensation). The LO leakage compensation unit is configured to perform compensation for signal leakage, the IQ imbalance compensation unit is configured to perform compensation for IQ imbalance, and the HD3 compensation unit is configured to perform compensation for the above-mentioned HD3. The LO leak compensation unit may be configured to perform compensation for HD3. The IQ imbalance compensation unit is configured to perform compensation for at least one of the following: IQ modulator imbalance (TX IQ Modulator imbalance) and IQ channel imbalance (IQ channel imbalance). Furthermore, if the TX compensation unit includes at least two of the LO leak compensation unit, IQ imbalance compensation unit, and HD3 compensation unit, compensation may be performed synchronously (e.g., in parallel), sequentially (e.g., in series), or, as shown in Figure 5B, IQ imbalance compensation may be performed first, followed by LO leak compensation, and finally HD3 compensation.

[0160] In some optional embodiments, the signal transmission link of the digital phase shifter architecture may further include an error correction module for the DAC (TX DAC Board Error Correction) and an AWGN (additive white gaussian noise) module for white gaussian noise, etc., which are not shown in the figures and can be added or removed as required by actual needs. In the IQ referred to in the embodiments of this disclosure, I may be an abbreviation for In-Phase, Q may be an abbreviation for Quadrature, and RF may be an abbreviation for Radio Frequency.

[0161] Embodiments of the present disclosure further provide a signal transmission link including a signal transmission link and a signal reception link, the signal transmission link may include a transmit baseband digital module 201, a digital-to-analog converter module 202, a transmit local oscillator 203, and a transmit quadrature modulator 204, as shown in Figure 6A or Figure 6B. The transmit baseband digital module 201 is configured to generate two orthogonal transmit digital baseband signals and transmit the generated transmit digital baseband signals to the digital-to-analog converter module 202. The digital-to-analog converter module 202 is configured to convert the transmit digital baseband signals to transmit analog baseband signals. The transmit local oscillator 203 is configured to supply a transmit local oscillator signal TX_LO. The transmit quadrature modulator 204 is configured to phase-shift the transmit local oscillator signal TX_LO based on the transmit analog baseband signal to obtain a phase-shifted radio frequency signal.

[0162] The signal receiving link may include a receiving local oscillator 302, a receiving mixer 303, an analog-to-digital converter (ADC) 304, and a receiving baseband digital module 305. The receiving local oscillator 302 is configured to supply a received local oscillator signal. The receiving mixer 303 is configured to perform a mixing operation on the received echo signal based on the received local oscillator signal to obtain a received analog baseband signal. The analog-to-digital converter 304 is configured to convert the received analog baseband signal into a received digital baseband signal. The receiving baseband digital module 305 is configured to process the received digital baseband signal to enable target detection and / or wireless communication to obtain target parameter information such as distance, velocity, angle, height, and micromotion characteristics.

[0163] In the embodiments of this disclosure, the two ideal I-channel digital baseband signals and Q-channel digital baseband signals generated by the transmitting baseband digital module 201 can be converted through the digital-to-analog conversion module 202 to obtain a highly ideal complex signal, and the phase of the complex signal can be precisely controlled by the transmitting baseband digital module 201. The receiver structure in the signal transmission / reception link as shown in Figure 6A or Figure 6B allows for efficient acquisition of phase information of the radio frequency signal of the signal transmission link, thereby enabling multi-antenna phase modulation.

[0164] In some exemplary embodiments, the signal transmission link may further include a power amplifier 205 configured to power amplified the radio frequency signal after phase shifting and output the amplified signal to the transmitting antenna.

[0165] In some exemplary embodiments, the signal transmission link may further include a transmitting antenna 206 configured to radiate the amplified signal into a predetermined spatial region.

[0166] In some exemplary embodiments, the signal receiving link may further include a receiving antenna 301, which is configured to receive echo signals, which are signals formed when signals transmitted by the signal transmitting link are reflected and / or scattered by an object.

[0167] In some exemplary embodiments, the received local oscillator signal may be a frequency sweep signal, or the received local oscillator signal may be a monophonic signal.

[0168] In the embodiments of this disclosure, the frequencies of the TX-LO signal received by the transmitting quadrature modulator 204 in the signal transmission link and the RX-LO signal received by the receiving mixer 303 in the signal reception link may be the same. For example, assuming that the signal output from the transmitting baseband digital module 201 is a sine wave of x MHz, the TX-LO signal and the RX-LO signal may both be sine waves of z GHz, where x and z are both positive numbers and generally between 0 and 1000.

[0169] The principle of this disclosure will be explained below through the signal transmission link shown in Figure 6A. As mentioned above, for an FMCW radar system, there are two types of wave transmission methods for the signal transmission link: 1) The transmitting local oscillator signal is a frequency sweep signal, and the transmitting digital baseband signal is a monotone signal. 2) The transmitting local oscillator signal is a monotone signal, and the transmitting digital baseband signal is a frequency sweep signal. If we represent the transmitting local oscillator signal TX_LO, the transmitting digital baseband signal, and the modulated transmitted signal as TLO(t), BB(t), and TX(t), respectively, and the I-channel signal and Q-channel signal as subscripts I and q, and their complex signal form as a superscript a, then in the two types of wave transmission solutions, the signals at each stage of the signal transmission link can be represented as follows.

[0170] 1) The transmitted local oscillator signal is a frequency sweep signal, and the transmitted digital baseband signal is a monophonic signal.

[0171]

number

[0172]

number

[0173] If we assume that the received local oscillator signal RX_LO in the signal receiving link shown in Figure 6A is represented by RLO(t), and that in this embodiment the received local oscillator signal RX_LO is a frequency sweep signal, then RLO(t) can be expressed as follows.

[0174]

number

[0175]

number

[0176] The above equation explains the following: Both of the above two frequency sweep methods achieve high-precision and highly accurate phase shift by controlling the initial phase Φ0 of the transmitted digital baseband signal, thereby enabling precise phase modulation and avoiding direct phase shift at high frequencies.

[0177] In some optional embodiments, the receiving antenna 301 may be connected via peripheral ports of the chip and formed on a carrier such as a PCB board. At the same time, in other optional embodiments, the receiving antenna may be integrated into the chip package to form an AiP or AoP, and have a chip structure of the packaged antenna.

[0178] In some exemplary embodiments, the signal receiving link may further include a low-noise amplifier (LNA) 306, which is installed between the receiving antenna 301 and the receiving mixer 303, and which low-noise amplified the echo signal received by the receiving antenna 301 before transmitting it to the receiving mixer 303.

[0179] In some exemplary embodiments, the signal receiving link may further include a series-connected low-pass filter (LPF) 307 and a high-pass filter (HPF) 308 installed between the receiving mixer 303 and the analog-to-digital converter 304, wherein the low-pass filter 307 and the high-pass filter 308 constitute a band-pass filter for filtering out-of-band noise.

[0180] In some exemplary embodiments, as shown in Figure 6A, in the signal receiving link, the receiving mixer 303 may be a real number mixer, and the analog-to-digital converter 304 may be a real number analog-to-digital converter.

[0181] In the embodiments of this disclosure, the signal transmission link employs a digital phase-shift architecture, but the signal reception link may include receivers with quadrature or non-quadrature reception architectures, thus enabling effective compatibility with sensors of various reception link architectures and effectively reducing the overall development cost of the transmit / receive link system.

[0182] In another exemplary embodiment, as shown in Figure 6B, in the signal receiving link, the receiving mixer 303 may be a quadrature mixer, and the analog-to-digital converter 304 may be a quadrature analog-to-digital converter.

[0183] To match the signal transmission link of the digital phase shifter architecture, in the embodiments of this disclosure, the receiving mixer 303 in the signal receiving link is adjusted to an IQ demodulator, and at the same time, the analog-to-digital converter 304 is adjusted to an IQ ADC. The echo signal received by the receiving antenna is sequentially processed by the low-noise amplifier 306, receiving mixer 303, low-pass filter 307, high-pass filter 308, and analog-to-digital converter 304, and then converted into an IQ digital baseband signal. The subsequent receiving baseband digital module 305 processes the IQ digital baseband signal to obtain target parameter information such as distance, velocity, angle, height, and micro-motion characteristics (i.e., micro-Doppler).

[0184] If the receiving mixer 303 is a quadrature mixer and the received local oscillator signal is a frequency sweep signal, the received local oscillator signal RX_LO may be expressed as follows:

[0185]

number

[0186] If the transmitting mixer 303 is a real number mixer,

[0187]

number

[0188]

number

[0189] In summary, the embodiments of this disclosure can be extended to various system-level technical proposals based on different combinations of transmission and reception solutions (for example, the transmitting end employing a digital baseband signal monophony and a local oscillator frequency sweep, or a digital baseband signal frequency sweep and a local oscillator signal monophony; the receiving end employing a real-number mixer and a real-number analog-to-digital converter, or a quadrature mixer and a quadrature analog-to-digital converter; or the receiving end employing a monophonic local oscillator signal or a frequency sweep local oscillator signal).

[0190] Figure 7A is a schematic diagram of a transmit / receive link including TX IQ modulation (Mod) and RX real demodulation (Real De-Mod) in an embodiment of the present disclosure; Figure 7B is a schematic diagram of a transmit / receive link including TX IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure; Figure 8 is a schematic diagram of a transmit / receive link combining BIST based on the structure shown in Figure 7B in an embodiment of the present disclosure; and Figure 9 is a schematic diagram of a transmit / receive link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure.

[0191] The following describes the transmission and reception links that constitute the transmission link structure described in the embodiments of this disclosure.

[0192] As shown in Figure 7A, the transmit / receive link may include a transmit link and a receive link, etc. The transmit link may include sequentially connected digital baseband signal sources (Baseband), direct digital frequency combiners (TX DDFS), IQ digital-to-analog converters (IQ DAC), low-pass filters (LPF), IQ modulators (IQ modulators), power amplifiers (PA), etc., and simultaneously radiates the signal amplified by the power amplifier into a predetermined spatial area via the transmit antenna. The receiving link may include sequentially connected low-noise amplifiers (LNAs), real mixers, trans-impedance amplifiers (TIAs), low-pass filters (LPFs), high-pass filters (HPFs), real-digital-to-analog converters (Real ADCs), etc. The echo signal received by the receiving antenna is sequentially processed by the LNA, Real Mixer, TIA, LPF, HPF, and Real ADC, and then converted into a real-digital baseband signal. A subsequent digital signal processing module then processes this real-digital baseband signal to obtain target parameter information such as distance, velocity, angle, height, and micro-motion characteristics. In the above transmitting and receiving link, the frequencies of the TX-LO signal received by the IQ modulator in the transmitting link and the RX-LO signal received by the Real Mixer in the receiving link may be the same. For example, as shown in Figure 7A, if the signal output from the baseband is a sine wave of x MHz, then the TX-LO signal and RX-LO signal may also be sine waves of z GHz.

[0193] In the embodiment shown in Figure 7A, the transmit link employs a digital phase shift architecture, while the receive link can employ analog architecture components, i.e., it does not employ IQ components. Therefore, it is effectively compatible with the sensors of the analog architecture receive link, and the development cost of the entire transmit / receive link system can be effectively reduced.

[0194] Selectively, in embodiments of the present disclosure, the receiving link may include a receiving antenna, i.e., the receiving antenna may be connected via peripheral ports of the chip and formed on a carrier such as a PCB board. Simultaneously, in some selectable embodiments, the receiving antenna may be integrated into the chip package to form an AiP or AoP, and have a chip structure of the packaged antenna.

[0195] In some optional embodiments, corresponding adjustments may be made to the receive link to match the transmit link of the digital phase shifter architecture, and the transmit / receive link shown in Figure 7B may include a similar transmit link architecture and receive link based on Figure 7A (to avoid repetition, specific explanations of the same parts are not given here), that is, the Real Mixer in the receive link of Figure 7A is adjusted to an IQ Demodulator, and at the same time the Real ADC is adjusted to an IQ ADC, that is, the receive link may include sequentially connected low-noise amplifiers (LNAs), IQ Demodulators, transimpedance amplifiers (TIAs), low-pass filters (LPFs), high-pass filters (HPFs), IQ digital-to-analog converters (IQ ADCs), etc., that is, the echo signal received by the receiving antenna is sequentially processed by the above LNA, IQ Demodulator, TIA, LPF, HPF, IQ After processing by an ADC, the signal is converted to an IQ digital baseband signal, and a subsequent digital signal processing module processes the IQ digital baseband signal to obtain target parameter information such as distance, velocity, angle, height, and micromotion characteristics (i.e., micro-Doppler).

[0196] Simultaneously, when performing self-calibration based on the transmit / receive link shown in Figure 7B, the self-calibration operation of the receive and / or transmit link is achieved simply by directly connecting the signal output port of the transmit link and the signal input port of the receive link via a transmission line. That is, the transmit link directly transmits the transmit signal to the receive link via the transmission line, without going through the transmit and receive antennas. At this time, there is a certain frequency offset between the TX-LO signal received by the IQ modulator on the transmit link and the RX-LO signal received by the IQ Demodulator on the receive link. For example, as shown in Figure 7B, if the signal output from the baseband is a sine wave of x MHz, the TX-LO signal may be a sine wave of z GHz, and the RX-LO signal may be a sine wave of (z GHz - y MHz), that is, the frequency offset in this case is y MHz.

[0197] In some selectable embodiments, as shown in Figure 7B, the transmit link (e.g., the transmitter (TX) shown in the figure) can be calibrated by adding a receive link (e.g., the receiver (RX) shown in the figure), and compensation can be performed based on the data calibrated by the TX IQ unbalance compensation unit in the transmit link. Simultaneously, the transmit link (e.g., the transmitter (TX) shown in the figure) can be calibrated by multiplexing the receive link (e.g., the receiver (RX) shown in the figure) that is actually used to transmit and receive signals, and compensation can be performed based on the data calibrated by the TX IQ unbalance compensation unit in the transmit link and / or receive link. Similar implementations are possible in other embodiments, and for the sake of simplicity of explanation, they will not be described later.

[0198] In some selectable embodiments, based on the structure shown in Figure 7B, an internal self-test module (Built-in Self-Test, abbreviated as BIST) can be installed in the RX-LO port of the IQ Demodulator of the receiving link shown in Figure 7B to achieve precise calibration of the transmit and receive links. That is, as shown in Figure 8, based on the transmit and receive link structure shown in Figure 7B, an IQ BIST architecture can be installed in the RX-LO port of the IQ Demodulator of the receiving link to input an LO signal with a preset frequency offset to the RX-LO port of the IQ Demodulator of the receiving link. For example, an IQ BIST, consisting of a phase angle converter and an IQ modulator, uses the received signal, for example, the TX-LO signal, passes through the phase angle converter to the IQ modulator, forms a frequency-offset signal based on the frequency offset signal of the input signal BIST-LO of another channel of the IQ modulator, and then inputs it to the RX-LO port of the IQ Demodulator. For example, if the TX-LO signal is a zGHz sine wave and the BIST-LO signal is a yMHz sine wave, then the frequency-offset signal input to the IQ Demodulator's RX-LO port is (zGHz-yMHz). Note that x, y, and z are all schematic values ​​between different embodiments, and the specific numerical values ​​may be the same or different.

[0199] In some selectable embodiments, the transmit link of the digital phase shifter architecture may be calibrated by multiplexing the receive link in the transmit / receive link, based on the IQ BIST architecture of the transmit / receive link structure shown in Figure 8. In other embodiments, the calibration operation of the transmit link using the receive link, and the calibration operation of the receive link using the transmit link, may both be realized by multiplexing the corresponding receive or transmit link in the link that actually transmits and receives signals, or by adding a corresponding calibrated receive link or calibrated transmit link to realize the calibration operation of the corresponding transmit or receive link in the link that actually transmits and receives signals.

[0200] Selectively, the IQ BIST may include a phase-angle converter and an IQ modulator, the phase-angle converter being used to calibrate the I and Q channels in the digital architecture's transmit link, and the input signals BIST-LO for the other channels of the IQ modulator may be yMHz sine waves, used to simulate the characteristics of the echo signal formed when the transmitted signal is reflected off the target. In Figure 8, x, y, and z are all positive numbers, x≠y≠z, and generally may be between 0 and 1000.

[0201] Selectively, in the transmit / receive links shown in Figure 8, a TX IQ imbalance compensation unit may be provided in the transmit link (e.g., between TX DDFS and IQ DAC) and / or in the receive link (e.g., after IQ ADC), that is, in order to solve problems such as IQ imbalance, the transmit and / or receive signals may be supplemented based on calibration parameters (or coefficients) obtained by the self-calibration operation described above.

[0202] In some selectable embodiments, based on the structure shown in Figure 8, the IQ BIST module described above may be installed between the signal output port of the transmit link and the signal input port of the receive link, as shown in Figure 9, i.e., the transmit link transmits the transmit signal directly to the receive link via the IQ BIST module, thereby enabling self-calibration of the receive link and / or transmit link without going through the transmit antenna and the receive antenna.

[0203] Note that in the transmission link structure shown in Figures 7A and 8-9, only the IQ imbalance compensation unit (TX IQ Imbalance compensation) is shown as an example. In actual applications, an LO leakage compensation unit (TX LO propagation compensation) and an HD3 compensation unit (TX HD3 compensation) may be added to the transmission link according to actual needs. This constitutes a compensation unit (TX compensation) that includes units such as the LO leakage compensation unit (TX LO propagation compensation), the IQ imbalance compensation unit (TX IQ Imbalance compensation), and / or the HD3 compensation unit (TX HD3 compensation).

[0204] Figure 10 is a schematic diagram of a transmit / receive link including an auxiliary circuit and BIST IQ Mod in an embodiment of the present disclosure, and Figure 11 is a schematic diagram of another transmit / receive link including an auxiliary circuit and BIST IQ Mod in an embodiment of the present disclosure.

[0205] As shown in Figure 10, and in combination with the structures and related descriptions shown in Figures 5B and 9, the transmit / receive link may include a transmit link, a receive link, and a calibration link. The transmit link may include sequentially connected TX digital baseband signal source (TX Baseband), direct digital frequency combiner (TX DDFS), compensation unit (Compensation), IQ digital-to-analog converter (IQ DAC), low-pass filter (LPF), IQ modulator (IQ Modulator), and power amplifier (PA), and simultaneously radiates the signal amplified by the power amplifier into a predetermined spatial region via the transmit antenna. The receiving link may include sequentially connected components such as a low-noise amplifier (LNA), a real mixer, a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable gain amplifier (VGA), a real digital-to-analog converter (Real ADC), and an RX baseband for TX RF calibration. The echo signal received by the receiving antenna is sequentially processed by the LNA, Real Mixer, TIA, HPF, VGA, and Real ADC, and then converted into a real digital baseband signal. A subsequent digital signal processing module then processes this real digital baseband signal to obtain target parameter information such as distance, velocity, angle, height, and micromotion characteristics.For a transmit link, the compensation unit (TX compensation) provided between the TX DDFS and the IQ DAC may include units such as an LO leakage compensation unit (TX LO propagation compensation), an IQ imbalance compensation unit (TX IQ imbalance compensation), and / or an HD3 compensation unit (TX HD3 compensation), thereby enabling corresponding compensation operations for LO propagation, IQ imbalance, and HD3 in the transmit link of the digital phase shifter architecture.

[0206] In some optional embodiments, a calibration module may be further installed between the transmit link and the receive link, configured to perform operations such as multiplexing the receive link to calibrate the transmit link of the digital phase shifter architecture described above. Simultaneously, compensation modules (i.e., the various compensation units in the embodiments described above) can perform compensation operations on the transmit signal at the transmit link end based on parameters or coefficients obtained by the calibration operation of the calibration module. In other embodiments, a receive compensation module corresponding to the receive link may be provided simultaneously or separately, in which case the receive compensation module can perform compensation of the echo signal at the receive link end based on parameters or coefficients obtained by the calibration operation described above.

[0207] As shown in Figure 10, the calibration module described above may include a BIST unit and an auxiliary circuit unit, etc. That is, the output port of the transmit link is connected to one of the nodes between the Real Mixer and the Real ADC in the receive link via the BIST unit and the auxiliary circuit unit. For example, the IQ Modulator in the transmit link generates a radio frequency signal of (zGHz±xMHz) based on an xMHz digital phase-shifted baseband signal and a zGHz LO signal, and outputs it to the BIST unit via its output port. The BIST unit performs a frequency offset operation of yMHz on the received radio frequency signal to obtain an analog echo signal of (zGHz±xMHz±yMHz), down-converts it using the IQ De-Modulator in the auxiliary unit to obtain a preset intermediate frequency signal (zGHz±xMHz±yMHz-zGHz=±xMHz±yMHz), and then inputs this intermediate frequency signal to a preset node in the receive link to perform the calibration operation in the transmit link.

[0208] Selectively, the auxiliary circuit unit may be a quadrature demodulator circuit, and the output terminal of the auxiliary circuit unit may be connected to any of the following nodes in the receive link: the node between the TIA and HPF, the node between the HPF and VGA, or the node between the VGA and Real ADC. Furthermore, to maximize the multiplexing of the receive link structure, the output port of one channel transmit link can pass through the BIST unit and the auxiliary circuit unit, and then connect its two branches, I and Q, to different transmit links, i.e., one channel transmit link is calibrated by multiplexing two channels of receive links, as shown in Figure 10. After the calibration of the transmit link is complete, the LO leakage compensation unit (TX LO leakage compensation), the IQ imbalance compensation unit (TX IQ imbalance compensation), and / or the HD3 compensation unit (TX HD3 compensation) in the above compensation unit (TX compensation) are used to implement corresponding compensation operations for problems such as LO leakage, IQ imbalance, and HD3 in the transmit link of the digital phase shifter architecture based on the parameters obtained by calibration.

[0209] In some selectable embodiments, the BIST unit may include sequentially connected phase-angle converters and IQ modulators, and the auxiliary circuit unit may include sequentially connected LNAs, IQ de-modulators, and TIAs, namely the phase-angle converter receives the radio frequency signal output from the transmit link, one input terminal of the IQ modulator is connected to the output terminal of the phase-angle converter, and the other input terminal receives the yMHz BIST-LO signal and generates a preset echo signal. The LNA amplifies the received echo signal and transmits it to one input terminal of the IQ de-modulator, and the other input terminal of the IQ de-modulator receives the zGHz RX-LO signal, after which the two output branches of the IQ de-modulator (i.e., the I branch and the Q branch) are each connected via the TIA to the corresponding nodes in the corresponding receive links, outputting the generated preset intermediate frequency signal to the two receive links, thereby enabling calibration and more effectively multiplexing the receive link design.

[0210] In the calibration operation in the embodiments of this disclosure, when the transmission link transmits a frequency sweep signal, the actual calibration operation can be performed point by point using the TX LO signal as a monophonic signal. Simultaneously, the TX LO signal can be used as a frequency sweep signal to perform a high-bandwidth calibration operation, and furthermore, the calibration operation for frequency sweep signals across the entire frequency band can be achieved at once using frequency sweep bandwidth calibration.

[0211] Based on the structure shown in Figure 10, further suppression of defects such as HD3, LO Leakage, and IQ Imbalance in the transmit link of the digital phase shifter architecture can be achieved by cascading at least two BIST units to a predetermined level. As shown in Figure 11, two BIST units connected in series can reduce noise due to the above defects to about -50 dB, effectively reducing the difficulty of developing and designing the associated link analog devices.

[0212] In some optional embodiments, when performing calibration and compensation operations for IQ Imbalance based on the transmit link of the digital phase shifter architecture described in the embodiments of this disclosure, the compensation coefficient for IQ Imbalance can be obtained based on spectral analysis in the time domain, or based on spectral peak ratio in the frequency domain.

[0213] In some selectable embodiments, the precision of the IQ Imbalance compensation coefficient can be further improved by approximating it to an ideal compensation coefficient through a more iterative calibration and compensation method, or an ideal compensation coefficient can be obtained through a multi-observation calibration and compensation method.

[0214] For example, in the iterative calibration and compensation method, the decision to stop the iterative operation is made based on whether the magnitude relationship between the compensation coefficients of the preceding and succeeding calibration and compensation operations, or the difference value between the compensation coefficients of the two calibration and compensation operations, satisfies a predetermined iteration condition. The compensation coefficient obtained when the iterative operation is stopped can then be used as the final compensation coefficient for the current scene to perform subsequent operations. In the multi-observation calibration and compensation method, after performing calibration and compensation operations multiple times (e.g., three times), an FFT (Fast Fourier Transform) is performed on the measurement data obtained in each operation to obtain corresponding amplitude and phase information. Then, the measurements are normalized by subtracting the values ​​to obtain related data, and an observation matrix is ​​constructed. Subsequently, the corresponding compensation coefficients are determined in reverse based on the data obtained by inversely calculating the observation matrix.

[0215] In some optional embodiments, the compensation coefficients for LO leakage and / or HD3 can also be obtained based on an approach similar to that for obtaining the IQ Imbalance compensation coefficient described above, for example, using an iterative calibration and compensation method or a multi-observation calibration and compensation method.

[0216] Each of the above examples of transmitters with compensation units addresses the problem of resolving harmonic distortion in the transmitter. Research has shown that some harmonic distortions may be caused by devices within the transmitter that include nonlinear characteristics, such as mixers.

[0217] Taking Figure 12A as an example, Figure 12A is a schematic diagram of the structure of a mixer in an embodiment of the present disclosure. As shown in Figure 12A, the mixer includes a voltage-current converter (V / I Converter), a current switch, and a current-voltage converter (I / V Converter). The voltage-current converter converts a received voltage signal into a current signal. The current switch is connected to the voltage-current converter and the second signal generator and processes the current signal output from the voltage-current converter using a local oscillator signal. The current-voltage converter is connected to the current switch and converts the current signal output from the current switch into a voltage signal.

[0218] In the above structure, a transistor amplifier is provided in the voltage-current converter. Based on the nonlinear characteristics of the transistor amplifier and the low frequency characteristics of the baseband signal, the current signal output from the voltage-current converter contains harmonic signals corresponding to the baseband signal. For example, a harmonic (HD) caused by a third-order nonlinearity in the baseband can simply be called HD3. Similarly, a harmonic caused by a fifth-order nonlinearity is called HD5. When the current switch processes the current signal output from the voltage-current converter, it upconverts the harmonic frequencies before converting to the radio frequency band. The complexity of the operation to suppress harmonic signals in the radio frequency band is high, resulting in high hardware costs. Failure to remove harmonic signals in the radio frequency band will affect the signal quality of radar transmission and reception, and further affect the accuracy of radar measurements.

[0219] The compensation unit is used to input the generated cancellation signal to the signal transmission link in order to cancel the harmonic signals in the radio frequency signal. The compensation unit is independent of the first signal generator.

[0220] Therefore, the compensation unit may include a cancellation signal generator. By using the cancellation signal output from the compensation unit, harmonic signals in the radio frequency signal can be suppressed, harmonic components in the radio frequency signal can be reduced, and the signal quality of the radio frequency signal output from the transmitter can be improved.

[0221] In an embodiment of the present disclosure, for harmonic signals in a signal transmission link, the compensation unit inputs the generated cancellation signal into the signal transmission link using feedback or based on the characteristics of the transmitted wave, and cancels the harmonic signals in the radio frequency signal output from the signal transmission link. The cancellation signal has characteristics such as being opposite in phase and close in amplitude to the harmonic signal transmitted in the radio frequency transmission circuit, and achieves the purpose of suppressing the harmonic signal.

[0222] In some examples, the compensation unit generates a compensation signal including a cancellation effect based on parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, and thus the path length merged with the LO signal.

[0223] For example, FIG. 5A shows an example of a transmitter in which a compensation unit accesses a signal transmission link. In the structure shown in FIG. 5A, the compensation unit is a TX compensation unit. The TX compensation unit includes a cancellation signal generator (not shown) generated based on the characteristics of the transmitted wave. The cancellation signal generator may be, for example, a TX HD3 compensation unit as shown in FIG. 5B.

[0224] The baseband processor (simply called the baseband frame in FIG. 5A) controls the orthogonal digital baseband signal generated by TX DDFS. The TX compensation unit generates an orthogonal compensation signal based on the parameters of the orthogonal digital signal, merges the orthogonal compensation signal and the orthogonal digital signal, and transmits them to the IQ DAC for conversion into an analog baseband signal. After passing through the LPF filtering process, mixing is performed by a mixer (i.e., the IQ modulator in FIG. 5A) to obtain a radio frequency signal obtained by mixing the TX LO signal and the analog baseband signal. The PA amplifies the mixing signal and outputs it via a transmitting antenna. The compensation signal cancels at least some of the harmonic signals in the radio frequency transmission circuit, such as the HD3 harmonic signal. Therefore, the clutter in the transmitted radio frequency signal is significantly reduced. The radio frequency signal may be an FMCW signal.

[0225] In other examples, the compensation unit generates a compensation signal based on the harmonic information fed back by the radio frequency transmission circuit. Referring to FIG. 12B, FIG. 12B is a structural schematic diagram of the compensation unit in the transmitter shown in FIG. 5A. As shown in FIG. 12B, the compensation unit includes a collection circuit and a cancellation signal generator.

[0226] The collection circuit is coupled to the radio frequency transmission circuit to collect the signals in the radio frequency transmission circuit to obtain a collection signal. The collection signal (also called the sampling signal) can reflect waveform information (also called harmonic parameters) in harmonic signals, such as the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, and the power of the main frequency signal.

[0227] The harmonic parameters reflected in the acquired signal are related to the information contained in the signal that the acquisition circuit can acquire. For example, if the acquisition circuit is a type of power acquisition circuit, the corresponding acquired signal includes the power of the main frequency. Also, for example, if the acquisition circuit utilizes at least some of the circuits of a receiver, the acquired signal will reflect the phase of the main frequency signal, the phase of the harmonic signals, the frequencies of the harmonic signals, the frequencies of the main frequency signal, the power of the harmonic signals, the power of the main frequency signal, and so on.

[0228] At least one of the above harmonic parameters may be extracted by analog circuitry. For example, a coupler and a power detector may output the power of the dominant frequency signal. Alternatively, the harmonic parameters may be extracted by taking advantage of the frequency domain computational capabilities of digital circuitry within the radar chip. For instance, by coupling a radio frequency transmission circuit, the same signal transmitted at the coupling point is acquired as the acquired signal. This acquired signal includes both the dominant frequency signal and the harmonic signals. The acquired signal is converted to a digital signal via an ADC, transmitted to digital circuitry, and computed in the frequency domain to obtain more harmonic parameters.

[0229] In one implementation, the input terminal of the acquisition circuit is connected to the output terminal or signal detection terminal of the mixer. This system can detect harmonic signals generated by a voltage-current converter and has a simplified acquisition circuit.

[0230] In one exemplary embodiment, the signal calibration link has a first input terminal connected between a voltage-current converter and a current switch in the transmitting mixer in the signal main path, a second input terminal connected between the signal main path and the transmitting antenna, and a signal output terminal connected to a compensation circuit in the signal main path. It is used to acquire a signal in the signal main path from at least one of the first and second input terminals and to determine compensation information based on the acquired signal. For example, in one type of connection configuration shown in Figures 13A and 13B, the input terminal of the acquisition circuit is connected to a detection terminal between the voltage-current converter and the current switch and coupled to an ADC.

[0231] Referring to Figure 13C, the signal calibration link includes a calibration demodulator (IQ Demodulator), a multiplexer (MUX), and a calibration module (TX Cabration). The calibration demodulator is configured to acquire a signal from the second input terminal in the main signal transmission path and perform demodulation. The multiplexer has two signal input terminals and one signal input terminal, one of which is connected between a voltage-current converter and a current switch in the transmit mixer, and the other signal input terminal is connected to the signal output terminal of the calibration demodulator and is configured to output a signal corresponding to one of the two signal input terminals. The calibration module is configured to determine the compensation information based on the signal output from the multiplexer.

[0232] In another embodiment, the input terminal of the acquisition circuit is connected to the radio frequency output terminal or radio frequency detection terminal of the radio frequency transmission circuit. The radio frequency output terminal is, for example, the output terminal of the radio frequency transmission circuit. The radio frequency detection terminal is, for example, the input terminal or output terminal of at least one PA stage in the radio frequency transmission circuit. This method can acquire more accurate harmonic parameters in the radio frequency transmission circuit, but it has a complex circuit structure.

[0233] In some chips, including BIST modules, the acquisition circuit can acquire the acquired signal using part or all of the circuitry in the BIST module. For example, as shown in Figure 13B, the input terminal of the acquisition circuit is coupled to a radio frequency output terminal that sequentially includes a downconverter, filter, etc., and is connected to an IQ ADC to output a digital acquired signal. The downconverter, filter, etc., can be multiplexed with the BIST module or receiver.

[0234] The collected signal is input to a cancellation signal generator. The cancellation signal generator is at least one circuit in the compensation unit. By connecting the cancellation signal generator to the first signal generator, the signal received by the radio frequency transmission circuit includes both the baseband signal and the cancellation signal simultaneously.

[0235] For example, the cancellation signal generator includes the cancellation signal generator described above and a digital circuit for extracting harmonic information. The digital circuit for extracting harmonic information may be installed independently, or at least a portion of it may be shared with the digital circuit in the radar chip.

[0236] The digital circuit that extracts harmonic information uses, for example, a digital circuit that processes the difference frequency baseband signal within the radar chip to extract harmonic information such as harmonic frequencies, dominant frequencies, and dominant power, and supplies it to the cancellation signal generator. The cancellation signal generator generates a cancellation signal based on the received parameters.

[0237] For example, a digital circuit that extracts harmonic information extracts the principal frequency amplitude in the acquired signal and calculates the harmonic amplitude from the difference between the principal frequency amplitude and the harmonic amplitude, which are set in advance. A cancellation signal generator generates a harmonic signal compensation signal based on the calculated harmonic amplitude and other set harmonic parameters. Each of these set harmonic parameters can be calculated based on the frequency sweep range, phase, etc., of the principal frequency signal to be transmitted by the radar chip.

[0238] The cancellation signal generator may be installed independently of the first signal generator, or at least partially shared with it. For example, the cancellation signal generated by the cancellation signal generator is input to the first signal generator so that the baseband signal output by the first signal generator includes the cancellation signal. The cancellation signal generator may include a third harmonic generator and a fifth harmonic generator.

[0239] For example, the compensation unit further includes an adder that combines a cancellation signal generator and a first signal generator, and merges the baseband signal generated by the first signal generator with the cancellation signal generated by the cancellation signal generator. In this embodiment, the cancellation signal includes a cancellation signal Signal_HD3 that cancels the third harmonic generated by a third harmonic generator, and a cancellation signal Signal_HD5 that cancels the fifth harmonic generated by a fifth harmonic generator. The cancellation signals Signal_HD3, Signal_HD5, and the baseband signal generated by the first signal generator are merged by the adder and output to the radio frequency transmission circuit.

[0240] As can be seen from the above, each example of a transmitter circuit that uses the feedback method described herein to pre-input a cancellation signal to the radio frequency transmission circuit can ensure that the radio frequency signal transmitted by the chip contains sufficiently low harmonic signals under different environmental conditions.

[0241] In order to effectively suppress harmonic signals in the transmitter during use, depending on the actual operating environment of the chip, and taking into consideration, for example, the effect of ambient temperature on the semiconductor device, this disclosure further provides a method for signal-canceling harmonic signals in the transmitter using a feedback mechanism, comprising steps 10 and 20. In step 10, an acquisition operation is performed on the signal in the signal transmission link to obtain an acquired signal, the signal transmission link is used to generate a radio frequency signal for radar detection, the radio frequency signal includes harmonic signals. In step 20, the acquired signal is detected, a cancellation signal is generated to cancel the harmonic signals, and it is output to the signal transmission link.

[0242] The method according to the embodiments of this disclosure involves performing an acquisition operation on a signal in a signal transmission link, obtaining an acquired signal, generating a cancellation signal using the acquired signal, and outputting it to the signal transmission link. This suppresses harmonic signals in the radio frequency signal using the cancellation signal, reduces the harmonic components in the radio frequency signal, improves the signal quality of the radio frequency signal output from the transmitter, and further improves the reception performance of the radio frequency signal of the receiver.

[0243] The following shows examples of transmitters and their operating processes with reference to Figures 7B to 13C.

[0244] Figure 7B shows an example in which harmonic information is extracted from the transmitter using a feedback mechanism and a compensation unit generates a corresponding cancellation signal. In the structure shown in Figure 7B, the compensation unit is exemplified as including a TX HD3 compensation unit. The TX HD3 compensation unit generates a compensation signal according to the waveform characteristics of the received signal. To prevent a reduction in signal transmission power during normal detection by the radar chip, the feedback mechanism can be performed in the calibration mode of the radar chip.

[0245] The baseband processor in the transmitter (baseband frame in Figure 7B) controls the quadrature digital baseband signal generated by the TX DDFS, mates the quadrature digital cancellation signal generated by the TX compensation unit, and transmits it to the IQ DAC, converting it into an analog baseband signal. This analog baseband signal has an analog cancellation signal mixed in to cancel harmonic signals in the transmission link. After undergoing LPF filtering, this analog baseband signal enters the first mixer (i.e., the IQ modulator in Figure 7B). The first mixer mixes the filtered signal received using the TX LO to obtain a radio frequency signal. This radio frequency signal is coupled and output via the receiver to the TX HD3 calibration circuit in the compensation unit (TX HD3 calibration frame in Figure 7B). The TX HD3 calibration circuit can be considered a digital circuit that extracts harmonic information.

[0246] In a receiver, after amplifying the signal output from a transmitter by an LNA, the signal is output to a second mixer (i.e., the IQ demodulator in FIG. 7B) to obtain a demodulated signal, and the demodulated signal is transmitted to a transimpedance amplifier for amplification processing. After sequentially passing through an LPF and an HPF, an analog-digital conversion operation is performed via an IQ ADC, and then the signal is transmitted to a TX HD3 calibration circuit. The TX HD3 calibration circuit extracts harmonic information in the transmitter from the feedback signal, converts it into parameters necessary for generating a cancellation signal by an upper-layer controller, and supplies the parameters to a TX HD3 compensation unit. The harmonic information acquired by the TX HD3 calibration circuit includes, for example, one or more parameters such as the initial phase, start frequency, cut-off frequency, frequency change time length, center frequency, etc. of a harmonic signal (or a main frequency signal). The TX HD3 compensation unit or the upper-layer controller determines parameters for generating a cancellation signal, such as the initial phase, frequency of the cancellation signal, delay, etc. in the compensation unit based on the harmonic information.

[0247] Note that the cancellation operations for the third harmonic and / or fifth harmonic in each of the above examples may be set according to the requirements of the transmitter.

[0248] FIG. 13A is a schematic diagram of a digital pre-compensation HD3 architecture based on a cube module in an embodiment of the present disclosure, and FIG. 1B is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency multiplier module in an embodiment of the present disclosure.

[0249] In some selectable embodiments, when performing calibration and compensation operations for HD3 based on the transmission link of the digital phase shifter architecture described in the embodiments of the present disclosure, the non-linear third harmonic whose generation source is mainly a V / I Converter based on HD3 in an active mixer can be realized by a compensation architecture based on a cube module as shown in FIG. 13A, or a compensation structure based on a triple frequency multiplier as shown in FIG. 13B.

[0250] Figure 13C is a schematic diagram of the calibration compensation of a transmit link based on a digital phase shifter architecture in an embodiment of the present disclosure. As shown in Figure 13C, based on the relevant technical details of the calibration compensation operations for IQ Imbalance, LO leakage, and HD3 in an embodiment of the present disclosure, the compensation operation for IQ Imbalance can be implemented by compensating for the conjugate signal of the BB (baseband) signal to inversely cancel the Miller component, and this compensation operation method is not affected by the calibration method for IQ Imbalance. Compensation for LO Leakage can be implemented by adjusting the DC components (i.e., DC bias) of the I channel and Q channel, and similarly, the calibration method for LO Leakage is not affected by the compensation solution. In the case of HD3, the third harmonic distortion of the V / I Converter of the quadrature mixer is the main source of HD3, and since the harmonic distortion is affected by the DC bias, if calibration of both LO Leakage and HD3 of the transmit link is required, the HD3 calibration should be performed after the LO Leakage calibration to ensure the accurate performance of the HD3 calibration.

[0251] Furthermore, the HD3 compensation methods, which include a digital pre-compensation architecture based on a digital cube module and a digital pre-compensation architecture based on a frequency multiplier module, directly impact subsequent calibration solutions and compensation flows. Specifically, these are described below.

[0252] In selectable embodiments, for a digital pre-compensation architecture based on a digital cube module, after calibrating and compensating for LO Leakage, the root cause of the HD3 problem, i.e., the HD3 compensation coefficient, can be calibrated with a stable DC bias, then the IQ Imbalance can be calibrated, and after continuing to compensate for the IQ Imbalance, the third harmonic distortion can be compensated for the I channel and Q channel, respectively, based on the results of the pre-compensation for the IQ Imbalance.

[0253] In selectable embodiments, for a digital pre-compensation architecture based on a frequency multiplier module, LO Leakage is calibrated and compensated, the HD3 compensation coefficient is obtained through calibration, the IQ Imbalance is calibrated with a stable DC bias and compensated for, and then, from the compensated results, the actual waveforms of the I-channel and Q-channel signals and the HD3 compensation coefficient can be calculated, respectively, and the waveform information for the 3rd and 5th frequencies requiring pre-compensation can be calculated in reverse.

[0254] In another optional embodiment, for a digital pre-compensation architecture based on a frequency multiplier module, the LO Leakage is calibrated and compensated, then calibrated simultaneously by multiple observations (e.g., three times) to obtain pre-compensation coefficients for HD3 and IQ Imbalance, then calibrated again by observations (e.g., two times) to obtain pre-compensation coefficients for the mirrored position of HD3, and finally, the coefficients for the 3rd and 5th frequencies requiring pre-compensation are calculated inversely from HD3 and the pre-compensation coefficients for the mirrored position of HD3. Note that the observations in the embodiments of this disclosure are used to represent operations such as testing and comparative analysis of different test results.

[0255] As can be seen by referring to the structure shown in Figure 13C, the acquisition circuit has two acquisition branches, which can be dynamically switched. The input terminal of one acquisition branch is connected between a voltage-current converter and a current switch, and the input terminal of the other acquisition branch is connected to the output terminal of a power amplifier, and an IQ demodulator is provided in this acquisition branch. In addition, in the structure shown in Figure 13C, the acquisition circuit is further provided with multiplexers, the input terminals of which are connected to the output terminals of the two acquisition branches, and the output terminals of which output the acquired signal.

[0256] In the structures shown in Figures 13A to 13C, the acquired signal may be an analog signal, that is, the output terminal of the acquisition circuit may be connected to an IQ ADC. Alternatively, the acquired signal may be a digital signal, and the acquisition circuit may include at least an IQ ADC.

[0257] Furthermore, if the radio frequency signal transmitted from the signal transmission link is not an orthogonal signal, the acquisition circuit in Figures 13A to 13C can be completed using non-orthogonal components. For example, a single-ended down-converter mixer may be used instead of an IQ demodulator, and a single-ended analog-to-digital converter may be used instead of an IQ ADC.

[0258] As shown in Figure 14A, embodiments of the present disclosure further provide a signal transmission method applicable to an electromagnetic wave device having at least one signal transmission link, comprising steps 1401 to 1404. In step 1401, the phase of the radio frequency transmission signal for each signal transmission link is determined. In step 1402, the initial phase of the transmit digital baseband signal for each signal transmission link is determined based on the phase of the radio frequency transmission signal. In step 1403, the transmit digital baseband signal is generated based on the determined initial phase. In step 1404, the transmit digital baseband signal is converted to a transmit analog baseband signal, and a phase shift operation is performed on the transmit local oscillator signal based on the transmit analog baseband signal.

[0259] The signal transmission method of the embodiments of this disclosure generates a digital baseband signal in the digital domain using a digital phase shifter architecture, resulting in better orthogonality and lower side lobes. Its phase shift can be generated with great accuracy, leading to higher phase modulation accuracy. This enables high-precision digital phase-shifted automotive radar systems, reduces the demand for antenna isolation, and simultaneously offers advantages such as low link loss, low cost, and no need for offline calibration. It can support more flexible wave transmission solutions, including high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0260] In some exemplary embodiments, the transmitted digital baseband signal is a monophonic signal and the transmitted local oscillator signal is a frequency sweep signal, or the transmitted digital baseband signal is a frequency sweep signal and the transmitted local oscillator signal is a monophonic signal.

[0261] In some exemplary embodiments, the frequency bandwidth of the frequency sweep signal is 2 GHz or greater.

[0262] In some selectable embodiments, the precision of the IQ Imbalance compensation coefficient can be further improved by approximating iterative calibration and compensation methods to obtain an ideal compensation coefficient, or by obtaining a compensation coefficient used in actual operating conditions by multi-observation calibration and compensation methods. The determination of the compensation coefficient requires additional consideration of the influence of link characteristics on the calculation accuracy of the compensation coefficient.

[0263]

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[0264]

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[0265] Similarly, it can be seen that there are differences in the compensation coefficients for harmonic distortion and LO leakage between the actual operating state and the ideal state of the signal transmission link.

[0266] Figure 14B is a diagram illustrating the principle of the compensation coefficient determination method according to an embodiment of this disclosure. The contents of each coordinate system in Figure 14B will be explained one by one below. Figure 14B(a) shows the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link. Figure 14B(b) shows the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link, the phase and amplitude of the compensation coefficient h in the actual operating state, and the difference value D between the compensation coefficient h in the actual operating state and the compensation coefficient H in the ideal state. Figure 14B(c) shows that the three vectors in Figure 14B(b) do not change with changes in amplitude. Therefore, it is sufficient to control the compensation coefficient h to approach the compensation coefficient H in terms of phase, and specifically refer to Figure 14B(d). In Figure 14B(e), region 1 shows that the compensation coefficient h is approaching the compensation coefficient H, and region 2 shows that the compensation coefficient h is not approaching the compensation coefficient H. Figure 14B(f) shows the optimization of the region in Figure 14B(e), reducing the size of region 1 and increasing region 2. The increased region 2 is further divided to obtain region 3 and a new region 2. Region 3 shows that the compensation coefficient h is not approaching the compensation coefficient H, while the new region 2 shows that the compensation coefficient with the opposite phase of the compensation coefficient h is approaching the compensation coefficient H. For example, in Figures 14B(g) and 14B(h), the compensation coefficient h is not in region 1, but the compensation coefficient h in Figure 14B(h) is closer to the compensation coefficient H than the compensation coefficient h in Figure 14B(g), which is clearly evident from the length of the difference value D in Figures 14B(g) and 14B(h).

[0267] Based on the above analysis, the embodiments of this disclosure provide a method for iterative calibration and compensation.

[0268] Specifically, the system can determine whether to stop the repetitive operation based on the relative magnitudes of the compensation coefficients of two calibration compensation operations, or whether the difference between the compensation coefficients of the two calibration compensation operations satisfies a preset repetition condition. The compensation coefficient obtained when the repetitive operation is stopped can then be used as the final compensation coefficient for the current scene to perform subsequent operations.

[0269] In some optional embodiments, the compensation coefficients for LO leakage and / or HD3 can also be obtained based on an approach similar to that for obtaining the IQ Imbalance compensation coefficient described above, for example, using an iterative calibration and compensation method or a multi-observation calibration and compensation method.

[0270] Specifically, embodiments of the present disclosure provide a signal calibration link for a signal transmitter path, which generates a radio frequency transmission signal after performing a compensation operation on a signal generated based on a compensation coefficient in order to achieve target detection and / or communication, and the signal calibration link is configured to acquire current observation information of the signal transmitter path at the current compensation coefficient, and if the current observation information satisfies an iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link, otherwise the current compensation coefficient is repeated until the obtained observation information satisfies the iteration condition.

[0271] The above signal calibration link will be explained below using an example.

[0272] In some selectable embodiments, the signal calibration method may include the following steps A1 to A4. Step A1: The initial compensation coefficient h(0) is set to 0, the signal state of the signal transmission link is obtained, and initial observation information O(0) is obtained. Step A2: The first compensation is performed on the signal transmission link using the first compensation coefficient h(1) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and first observation information O(1) is obtained. Step A3: If the difference value between the initial observation information O(0) and the first observation information O(1) is greater than a preset difference threshold, the k-th compensation coefficient h(k) is determined based on the (k-1) compensation coefficient h(k-1) and the (k-1) observation information O(k-1), where k is an integer of 2 or greater. Determining the k-th compensation coefficient h(k) based on the k-1st compensation coefficient h(k-1) and the k-1st observation information O(k-1) involves iteratively calculating the sum of the k-th compensation coefficient h(k-1) and the k-1st observation information O(k-1) based on updating the first compensation coefficient h(1) to the initial observation amount O(0) if the difference between the absolute value of the initial observation information O(0) and the absolute value of the first observation information O(1) is greater than a preset difference threshold, thereby determining the k-th compensation coefficient h(k). If the difference between the absolute value of the first observation information O(1) and the absolute value of the initial observation information O(0) is greater than a preset difference threshold, the first compensation coefficient h(1) is updated to the inverse of the initial observation amount O(0), and the difference between the k-1st compensation coefficient h(k-1) and the k-1st observation information O(k-1) is iteratively calculated, thereby determining the k-th compensation coefficient h(k). Step A4: If the difference between the initial observation information O(0) and the first observation information O(1) is smaller than a preset difference threshold, the initial observation information O(0) is adjusted based on a preset phase adjustment amount to obtain the adjusted first compensation coefficient h(1), a new first observation information O(1) is obtained based on the adjusted first compensation coefficient h(1), and the k-th compensation coefficient h(k) is determined based on the new first observation information O(1), and the difference between the initial observation information O(0) and the new first observation information O(1) is larger than the aforementioned difference threshold.

[0273] Selectively, if the difference between the absolute value of the initial observation information O(0) and the absolute value of the new first observation information O(1) is greater than the difference threshold, the sum of the k-1 observation information O(k-1) and the adjusted k-1 observation information O(k-1) obtained by processing the k-1 observation information O(k-1) with the phase adjustment amount is iteratively calculated to determine the k-th compensation coefficient h(k).

[0274] Selectively, if the difference between the absolute value of the new first observation information O(1) and the absolute value of the initial observation information O(0) is greater than the difference threshold, the first compensation coefficient h(1) is updated to the inverse of the adjusted first compensation coefficient h(1), and the difference between the (k-1)th compensation coefficient h(k-1) and the adjusted (k-1)th observation information O(k-1) obtained by processing the (k-1)th observation information O(k-1) with the phase adjustment amount is iteratively calculated to determine the kth compensation coefficient h(k). In each iterative operation process of step A3 and step A4, it is determined whether the observation information corresponding to each compensation coefficient is smaller than a preset iteration threshold, and if the observation information is smaller than the iteration threshold, the iteration operation is stopped, and the compensation coefficient used for the observation information smaller than the iteration threshold is set as the compensation coefficient.

[0275]

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[0276]

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[0277] If the initial phase of the baseband (BB) signal is x degrees, then the initial phase of the subsequent 3BB signal must be 3x degrees, and O(n) represents the amplitude-phase value of the frequency point corresponding to the 3BB signal divided by the amplitude-phase value of the frequency point corresponding to the BB signal.

[0278]

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[0279] Furthermore, based on the above analysis, the embodiments of this disclosure further present a multi-observation calibration and compensation method.

[0280] Specifically, after performing calibration and compensation operations multiple times (for example, three times), an FFT (Fast Fourier Transform) is performed on the measurement data obtained in each operation to acquire corresponding amplitude and phase information. Then, the measurements are normalized by subtracting them to obtain related data, and an observation matrix is ​​constructed. Subsequently, the corresponding compensation coefficients are calculated in reverse based on the data obtained by inversely calculating the observation matrix.

[0281] In some optional embodiments, the compensation coefficients for LO leakage and / or HD3 can also be obtained based on an approach similar to that for obtaining the IQ Imbalance compensation coefficient described above, for example, using an iterative calibration and compensation method or a multi-observation calibration and compensation method.

[0282] Specifically, the signal calibration link is configured such that the signal transmission main path generates a radio frequency transmission signal after performing a compensation operation on a signal generated based on a compensation coefficient in order to achieve target detection and / or communication, and the signal calibration link determines initial observation information O(0), first observation information O(1), and second observation information O(2) corresponding to the signal transmission main path under the conditions of initial compensation coefficient h(0), first compensation coefficient h(1), and second compensation coefficient h(2) with different numerical values, and uses the initial observation information O(0), first observation information O(1), and second observation information O(2) to determine a third compensation coefficient h(3) as a compensation coefficient used in the compensation operation of the signal transmission link.

[0283] In another selectable embodiment, the orthogonal unbalance compensation method may include steps E1 to E5. Step E1: The initial compensation coefficient h(0) is set to 0, the signal state of the signal transmission link is obtained, and initial observation information O(0) is obtained. Step E2: The first compensation is performed on the signal transmission link using the first compensation coefficient h(1) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and first observation information O(1) is obtained. The second compensation is performed on the signal transmission link using the second compensation coefficient h(2) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and second observation information O(2) is obtained. The values ​​of the second compensation coefficient h(2) and the first compensation coefficient h(1) are different, and the values ​​of the second observation information O(2) and the first observation information O(1) are different. Step E3: Obtain the ratio of the first difference value to the initial observation information O(0) to obtain the first ratio d1, and obtain the ratio of the second difference value to the initial observation information O(0) to obtain the second ratio d2. The first difference value is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference value is the difference between the initial observation information O(0) and the second observation information O(2). Step E4: Condition that the third observation information O(3) is 0, determine the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2. Specifically, a 2x2 matrix is ​​constructed, with the first row of the 2x2 matrix recording the real and imaginary parts of the first ratio, and the second row recording the real and imaginary parts of the second ratio. The product of the inverse of the 2x2 matrix and the 1x2 first matrix is ​​calculated to obtain the 1x2 second matrix, where the value of the first row of the first matrix is ​​1 and the value of the second row is 0. In the second matrix, the first row is the value of the first coefficient x1 and the second row is the value of the second coefficient x2. In step E5, the product of the first coefficient x1 and the second coefficient x2 is calculated to obtain the first multiplication result, the product of the second coefficient x2 and the second compensation coefficient h(2) is calculated to obtain the second multiplication result, and the first and second multiplication results are taken as the third compensation coefficient h(3).

[0284]

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[0285] Specifically, a 2x2 matrix is ​​constructed, with the first row of the 2x2 matrix recording the real parts of the intermolecular ratio CI and orthogonal ratio CQ, and the second row recording the imaginary parts of the intermolecular ratio CI and orthogonal ratio CQ.

[0286] The product of the inverse of the 2x2 matrix and the 1x2 first matrix is ​​calculated to obtain a 1x2 second matrix. In the first matrix, the values ​​in the first row are the real part of the initial observation information O(0), and the values ​​in the second row are the imaginary part of the initial observation information O(0). In the second matrix, the first row is the value of the in-phase ratio CI, and the second row is the value of the orthogonal signal CQ.

[0287]

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[0288]

number

[0289] In this disclosure, electromagnetic waves include radio waves and light waves. Radio waves include short waves, medium waves, long waves, and microwaves. Microwaves include centimeter waves (i.e., electromagnetic waves from 3 GHz to 30 GHz, e.g., electromagnetic waves from 3.1 GHz to 10.6 GHz, electromagnetic waves in the 24 GHz band, etc.) and millimeter waves (i.e., electromagnetic waves from 30 GHz to 300 GHz, e.g., electromagnetic waves in the 60 GHz band, electromagnetic waves in the 77 GHz band (e.g., 77 GHz to 81 GHz, etc.)). Light waves may include ultraviolet light, visible light, infrared light, lasers, etc. The electromagnetic wave frequency band for lasers is (3.846 to 7.895)*10^5 GHz, meaning that lasers are included in the frequency bands of ultraviolet light and some of the visible light.

[0290] Figure 15 is a schematic diagram of a transmit link having at least two transmit channels in an embodiment of the present disclosure, Figure 16 is a schematic diagram of the structure of a digital LO signal generator in an embodiment of the present disclosure, and Figure 17 is a schematic diagram of the structure of a transmit link including a feed line unequal length compensation module in an embodiment of the present disclosure.

[0291] As shown in Figure 15, in the case of a transmit link using at least two transmit channels, each transmit channel (or transmit channel) is connected to the same local oscillator (LO) source. Therefore, the feed line lengths (TXLO) between the local oscillator signal and the mixer in each transmit channel differ, resulting in relative delay problems between different transmit channels. Furthermore, by employing unequal-length transmit antenna feed lines in antenna arrays, it is possible to achieve better link budgets, simpler antenna design and wiring, lower inter-antenna coupling, better target angle release performance, and smaller modules and costs. For this reason, antenna designs employing unequal-length transmit antenna feed lines are increasingly being applied, particularly in application scenarios such as short-range measurements, relatively enclosed spatial areas (such as indoors or shipboard lighting), or in various scenarios employing packaged antenna chip (AiP) applications, where frequency sweeps, for example, can cover a relatively wide range of application scenarios. However, because the length of the transmit antenna feed lines differs between different transmit channels, relative delay problems occur when radio frequency signals pass through different transmit antennas.

[0292] In the above-described transmit / receive link, to address the problem of relative signal delay due to differences in the length of feed lines between different transmit or receive channels, embodiments of this disclosure provide a compensation solution for relative delay due to unequal feed line lengths based on the above-described digital phase shift architecture, in order to effectively improve the quality of the transmit and receive signals.

[0293] Specifically, as shown in Figure 15, the transmitting antenna array of the digital phase shifter architecture includes four transmitting channels, each transmitting channel sharing a waveform control module and a local oscillator source (LO). Each transmitting channel is sequentially connected to a transmit direct digital frequency combiner (TX DDFS), an IQ digital-to-analog converter (DAC), an analog low-pass filter (Analog LPF), an IQ mixer (Mixer), a power amplifier (PA), and a transmitting antenna. That is, each waveform control module is connected to the TX DDFS in each transmitting channel, and simultaneously, each local oscillator source (LO) is connected to one input terminal of the Mixer in each transmitting channel. The phase-shift signals output from the LPF in each transmitting channel are up-mixed to form a radio frequency signal, which is then transmitted via each transmitting antenna. The problems of unequal feed line lengths between the local oscillator LO and the Mixer in each transmitting channel, and / or between the PA of each transmitting channel and its transmitting antenna, can be addressed by pre-compensating in the Transmit Direct Digital Frequency Synthesizer (TX DDFS), thereby reducing the transmission signal delay problem caused by the unequal feed line lengths.

[0294] For example, to address the issue of unequal feedline lengths between the local oscillator LO and the Mixer in each transmitting channel, the shortest feedline between the local oscillator LO and the Mixer in one transmitting channel can be used as a reference. The length difference between the shortest feedline and the length difference between the Mixer in each of the remaining transmitting channels and the local oscillator LO can then be obtained. Since the propagation speed c of the transmitted signal is relatively fixed, the corresponding delay value can be obtained by dividing the length difference between each transmitting channel by the speed c. Subsequently, when transmitting the signal, the corresponding compensation can be directly performed in TX DDFS to obtain the delay for each transmitting channel relative to the reference. At the same time, a similar method can be used to address the issue of unequal feedline lengths between the PA of each transmitting channel and its transmitting antenna, thereby obtaining and compensating for the delay problem caused by the differences in feedlines between each transmitting channel.

[0295]

number

[0296] Figures 16 and 17 show schematic diagrams of the structure of one transmit channel, and Figure 15 shows a transmit antenna array with four transmit channels. In actual applications, a transmit antenna array only needs to include at least two transmit channels (e.g., two, three, or five), and compensation can be performed during the process of receiving the signal and / or signal processing. At the same time, for structures with one transmit channel and / or one receive channel, the same approach can be used to perform compensation directly within the TX DFFS or during the process of receiving the signal and / or signal processing.

[0297] For related technical proposals that perform calibration and other operations on the receiving link and / or transmission link of an analog phase shifter architecture and the receiving link and / or transmission link of a digital phase shifter architecture, using the architecture of the transmitting link and / or receiving link of the digital phase shifter architecture described in the embodiments of this disclosure as an auxiliary link, please refer to Figures 18 to 21.

[0298] Figure 18 is a schematic diagram of the calibration and compensation of the transmit / receive link using an auxiliary circuit in an embodiment of the present disclosure, and Figure 19 is a schematic diagram of the calibration and compensation of the receive link using an auxiliary transmit circuit in an embodiment of the present disclosure.

[0299] Specifically, a signal transmission / reception link may generally include an antenna, radio frequency analog devices, baseband analog devices, a baseband digital processor, etc. The radio frequency analog devices may include a phase-locked loop (PLL), a power amplifier (PA), a phase shifter (PS), a low-noise amplifier (LNA), a mixer local oscillator (LO), and a power detector (PD), etc. The baseband analog devices may include a low-pass filter (LPF), a high-pass filter (HPF), an analog-to-digital converter (ADC), etc.

[0300] Analog devices or circuits are subject to various non-ideal conditions that can change with temperature. For example, unbalanced defects may exist in the frequency response of many parts, such as radio frequency analog devices and circuits in the transmit link (TX), baseband analog devices and circuits in the receive link (RX), and radio frequency analog devices and circuits in the receive link (RX). There may also be PD accuracy issues at the PA output port and LNA input port, and the receive link (RX) may have problems such as local oscillator leakage (LO leakage). Simultaneously, when analog devices operate at different frequencies, their amplitude-frequency responses also differ, which is equivalent to adding an unexpected window function to the received signal. This further adversely affects subsequent operations such as estimating target distance and velocity, and in multi-target scenes in particular, it can even lead to the detection of false targets or the leakage of true targets. Furthermore, in the case of at least two links, there are differences in the frequency response between different receive links, and additional noise information such as different phases and amplitudes is introduced into the receive link. In particular, the introduced phase directly affects the subsequent estimation of the target echo direction and further reduces the accuracy of target angle estimation.

[0301] In some embodiments, errors in analog devices and circuits can be measured by an external device, and then certain compensation operations can be performed to calibrate the performance of the analog devices and circuits of the receiving link, for example, using operations such as Bench calibration or ATE calibration.

[0302] The embodiments of this disclosure further provide a calibration solution for transmit and / or receive links based on an auxiliary link scheme, enabling accurate calibration, real-time calibration of analog devices and circuits, and effectively reducing the impact of fluctuations in the parameter indicators of radio frequency devices due to environmental changes without requiring external equipment.

[0303] As shown in Figure 18, the transmission and reception link of electromagnetic wave signals can be calibrated in real time by installing an auxiliary calibration circuit (Auxiliary). This auxiliary calibration circuit (i.e., auxiliary calibration link) can be integrated into the electromagnetic wave sensor, enabling accurate real-time calibration without the need for external devices, and effectively reducing the effects of fluctuations in the parameter indicators of radio frequency devices due to environmental changes.

[0304] For example, regarding the transmission path, an auxiliary receiver (ARX) can be installed adjacent to the transmission path in the sensor to enable real-time calibration of the transmission path. Similarly, regarding the reception path, an auxiliary transmission path (ATX) can be installed adjacent to the reception path (i.e., the signal reception link) in the sensor to enable real-time calibration of the reception path. Furthermore, an auxiliary path can be installed between two paths to be calibrated, and in this way, one auxiliary path can be multiplexed to perform calibration on at least two different paths. As shown in Figure 18, by providing one shared ARX between two transmission paths and one shared ATX between two reception paths, problems such as excessively long wiring or lines can be further avoided. In an optional embodiment, two ARXs and two ATXs can be installed for the 4-transmitter, 4-receive antenna array shown in Figure 18, that is, each ARX is installed between two transmit channels and each ATX is installed between two receive channels, effectively reducing the complexity of wiring and improving the real-time capability of calibration.

[0305] As shown in Figure 18, the receiving path can be divided into a radio frequency portion (i.e., RF Rx) and a baseband portion (i.e., Rx BB). To further improve the accuracy of calibration, auxiliary calibration paths can be installed corresponding to RF Rx and Rx BB, respectively. For example, a radio frequency auxiliary transmitter unit (which can simply be called an RF ATX) can be installed for RF Rx, and a baseband auxiliary transmitter unit (which can simply be called an IF ATX) can be installed for Rx BB. In addition, for the ARX path, a radio frequency monotone signal transmitter (RF Tone Generator) can be installed to calibrate the ARX, and then the transmit channel (TX) can be calibrated using the calibrated ARX.

[0306] In some selectable embodiments, as shown in Figure 18, calibration can be performed on the entire receiver path by first calibrating the Rx BB using the IF ATX, then calibrating the RF ATX using the calibrated Rx BB, and finally calibrating the Rx BB using the calibrated RF ATX.

[0307] Based on the structure shown in Figure 18 and combined with the contents shown in Figure 19, a receiver calibration solution will be described, which may include sequentially connected receiving antennas (or receiving ports), devices such as LNAs, mixers, TIAs, LPFs, HPFs, and Real ADCs, and PDs connected to the LNA input port. Based on the mixer, the receiver path can be divided into RF Rx and Rx BB, that is, along the signal transmission direction, the part before the mixer can be defined as RF Rx, and the remaining part can be defined as Rx BB. As shown in Figure 19, RF Rx may include receiving antennas (or receiving ports), LNAs, Real Mixers, and PDs.

[0308] The IF ATX installed in response to Rx BB of the above receiving path may include sequentially connected frequency dividers (e.g., Freq Divider 1 / N) and Real DACs (e.g., 1-bit Real DACs), i.e., the output terminal of the Real DAC is connected to the output terminal of the Real Mixer. As shown in Figure 19, the IF ATX may include sequentially connected frequency dividers (e.g., Freq Divider 1 / N) and 1-bit digital-to-analog converters (e.g., 1-bit Real DACs), i.e., the IF ATX can stably generate a single tone signal under various conditions using the frequency divider and digital-to-analog converter, and the single tone signal can be freely set to various frequencies.

[0309] In some optional embodiments, to improve the accuracy of the calibration, the IF ATX may be calibrated before the Rx BB is calibrated, for example, by calibrating the intermediate frequency (IF) frequency response or DC calibration of the IF ATX. Simultaneously, after the calibration of the IF ATX is completed, the calibrated IF ATX can be used to calibrate the Rx BB in the receiving path, for example, by transmitting monotone signals of different frequencies with the calibrated IF ATX to complete calibration, compensation, and other operations on the Rx BB section. Optionally, the frequency (e.g., tens of MHz) of the monotone signal transmitted by the IF ATX may be determined based on the sampling frequency of the Real ADC in the receiving path.

[0310] An RF ATX installed in correspondence with the RF Rx of the receiving path may include a sequentially connected TX DDFS, IQ unbalance compensation unit (TX IQ Comp), adder, IQ digital-to-analog converter (IQ DAC), low-pass filter (LPF), amplifier (e.g., PA), multiplier, local oscillator (LO, not shown), and squarer (x^2), wherein the output terminal of the multiplier is connected to the output terminal of the Mixer in the receiving path via the squarer to form a first calibration branch, and simultaneously the output terminal of the multiplier is connected to the link between the PD and the transmitting antenna in the receiving path to form a second calibration branch. The IQ unbalance compensation unit may be configured to compensate for IQ unbalance in the RF ATX, the adder may be configured to compensate for local oscillator leakage (LO Leakage) in the RF ATX, and the squarer may be configured to compensate for residual sideband effects resulting from IQ unbalance in the RF ATX.

[0311] In some optional embodiments, to improve the accuracy of the calibration, the RF ATX may be calibrated first, for example, to address issues such as local oscillator leakage, radio frequency response, and IQ imbalance in the RF ATX before the Rx ATX is calibrated. Simultaneously, after the RF ATX calibration is completed, the RF Rx of the receiving path is calibrated using the calibrated RF ATX, for example, by transmitting monophonic signals of different frequencies using the calibrated RF ATX, and then the baseband signal processing module of the receiving link calibrates and compensates for auxiliary calibrations such as the power detector (PD) at the input terminal of the LNA in the radio frequency portion of the receiving path RF Rx, the total gain from the receiving path LNA to the ADC, and the frequency response.

[0312] In some selectable embodiments, when calibrating the receiver using an IF ATX and an RF ATX, the IF ATX is first calibrated, the Rx BB in the receiver is calibrated using the calibrated IF ATX, then the RF ATX is calibrated using the calibrated Rx BB by the first calibration branch (i.e., squarer) of the RF ATX, and finally the RF Rx in the receiver is calibrated using the second calibration branch of the calibrated RF ATX, and further calibration and compensation operations can be performed for the entire receiver.

[0313] Specifically, as shown in Figure 19, first, the Rx BB and Real DAC are calibrated using a 1-bit Real DAC. Next, the LO leak and IQ imbalance in the RF ATX are calibrated using the calibrated Rx BB. Finally, the RF LO leak and RF frequency response of the RF Rx can be calibrated by inputting the calibrated RF ATX transmit signal from the LNA in the receiving path.

[0314] Figure 20 is a schematic diagram of the calibration and compensation of a transmit link using an auxiliary receiver circuit in an embodiment of the present disclosure. As shown in Figure 20, the transmit path (Transmitter) may include a phase shift module PS, an amplifier PA, a power detector PD, etc. For example, the transmit path can employ a digital phase shift architecture transmit link as described in any embodiment of the present disclosure, which can be specifically described in the relevant figures and textual descriptions and will not be described here. Because the transmit path employs a digital phase shift architecture, more precise phase shift operation can be achieved, and the transmit channel can simultaneously support multiple modes such as multi-antenna DDM and FDM (Frequency Division Multiplexing), eliminating the need for RF phase shifter calibration, reducing isolation and coupling in the phase shift system, and lowering link loss and production costs. Furthermore, to address potential issues such as TX IQ mismatch and LO leakage, the transmission path of this digital phase shifter architecture can also support RF frequency response compensation in the digital domain, as well as calibration operations for IQ imbalance and LO leakage.

[0315] To address issues in the transmission path such as TX IQ mismatch, LO leakage, and frequency response, an auxiliary receiver (ARX) can be installed to perform related calibration and compensation operations. As shown in Figure 20, the ARX may include a sequentially connected mixer, TIA, LPF, HPF, IQ ADC, adder, and RF calibration module (RF Calib). Specifically, one input terminal of the mixer receives the ARX IQ LO signal, and the other input terminal is connected along the signal transmission direction (i.e., the direction of the arrow shown in the figure) to a node before the transmission path PD or after the phase shifter (module), for example, to the output terminal of the PA (to synchronously calibrate the PA), or to the input terminal of the PA, thereby performing the calibration operation of the transmission path by the ARX. There is a set difference frequency between the LO signal frequency in the transmission path and the ARX IQ LO signal frequency, which creates a crossover frequency between the two signals and simulates the true transmit and receive signal loop.

[0316] In selectable embodiments, to further improve calibration accuracy, a corresponding calibration circuit (i.e., a calibration receiver unit) may be provided with respect to the ARX (i.e., the auxiliary receiver unit), such as the RF Tone Generator shown in Figure 20. The RF Tone Generator may include a sequentially connected TX DDFS, adder, Real DAC, LPF, amplifier, multiplier, and bandpass filter (BPF). The adder is configured to calibrate-compensate for TX LO leakage (TX LO leakage waveform). The multiplier is configured to compensate for RF Tone Gen LO leakage. The BPF is configured to filter the DC signal generated by the LO leakage of the RF Tone Generator. That is, the RF Tone Generator is configured to generate multiple stable tone signals of different frequencies to perform calibration operations on the ARX.

[0317] In some selectable embodiments, as shown in Figure 20, the ARX can first be calibrated using an RF Tone Generator, and then the calibrated ARX can be used to calibrate the transmitting channel (Transmitter), including the PA, and devices and circuits such as the PD at the PA output terminal, the phase shifter in the transmitting channel, and the total gain and frequency response output from the DAC to the PA.

[0318] Specifically, as shown in Figure 20, first, an RF Tone Generator is used to generate multiple stable monophonic signals of different frequencies to assist in the calibration of the ARX. Then, based on the calibrated ARX, it is possible to assist in the calibration of problems such as IQ imbalance, local oscillator leakage, and frequency response mismatch in the transmission path TX.

[0319] Figure 21 is a schematic diagram of the structure of an auxiliary circuit in an embodiment of the present disclosure, Figure 22 is a schematic diagram of the structure of another auxiliary circuit in an embodiment of the present disclosure, Figure 23 is a schematic diagram of the circuit module of the IQ Mixer in an embodiment of the present disclosure, Figure 24 is a schematic diagram of the structure of the IQ Mixer in an embodiment of the present disclosure, and Figure 25 is a schematic diagram corresponding to the configuration shown in Figure 24.

[0320] In some optional embodiments, based on the structure and related description shown in Figures 18-19, the ATX can further be implemented by a squarer combining multiple bit DACs. As shown in Figure 21, the receiving path includes sequentially connected LNA, mixer, TIA, HPF, ADC, BB processing module (BB Processor and local oscillator LO), i.e., the mixer uses the first LO signal supplied from the local oscillator to downconvert the echo signal supplied from the LNA to obtain an intermediate frequency signal (i.e., an analog baseband signal). The ATX may also include sequentially connected first DAC (i.e., DAC1), mixer, and squarer (x^2). The first DAC supplies an analog signal to the receiving end of the mixer of the ATX, and the other receiving end of the mixer may be connected to a local oscillator in the receiving path, and the analog output of the first DAC is used with the second LO signal supplied by the local oscillator. The echo signal is mixed and the resulting analog intermediate frequency signal is output to the input terminal of the squarer, which connects the processed analog intermediate frequency signal to the input terminal of the TIA, thereby calibrating the circuits and components such as the TIA, HPF, and ADC in the receiving path. In some other selectable embodiments, the ATX can further output the analog echo signal directly to the receiving terminal of the squarer via a DAC, which then transmits the processed analog echo signal to the input terminal of the LNA, thereby achieving calibration of the circuits and components such as the LNA, mixer, TIA, HPF, and ADC in the receiving path.

[0321] In some selectable embodiments, as shown in Figure 22, a second DAC is further installed in the receiving path, based on the structure shown in Figure 21, for the VGA and ADC including an SDM unit and a Decimation filter, and the output terminal of the second DAC can be connected to any node in the receiving path between the mixer and TIA, between the TIA and HPF, between the HPF and VGA, or between the VGA and ADC to calibrate the corresponding circuits and components. Selectively, the first DAC may be a multi-bit (e.g., 10-bit) DAC, and the second DAC may be a 1-bit DAC.

[0322] Those skilled in the art can make other reasonable modifications to the ATX based on the DAC and squarer described above, and specific embodiments are not limited to the Disclosure, as long as they can achieve the same or similar functions as the ATX in the embodiments of the Disclosure.

[0323] In some optional embodiments, when the ATX is installed near the receiving path, for example, by utilizing the air gap between receiving paths, it is possible to enable two or more receiving paths to share one ATX simultaneously. Similarly, when the ARX is installed near the transmitting path, for example, by utilizing the air gap between transmitting paths, it is possible to enable two or more transmitting paths to share one ARX. Selectively, the above ATX and / or ARX may operate intermittently. For example, the ATX can transmit a calibration monophonic signal between receiving path operations (e.g., between frames or between chirps) to perform operations such as effective calibration and compensation to the receiver in real time. Similarly, the ARX can perform calibration and compensation in a predetermined manner between receiver operations (e.g., between frames or between chirps) by first generating a calibration monophonic signal using a tone generator, first effectively calibrating the ARX, and then effectively calibrating the transmitter with the calibrated ARX.

[0324] The solution for calibrating the transmission and / or reception paths based on the auxiliary link method shown in Figures 18 to 22 effectively improves the performance of analog circuits and modules by adding auxiliary circuits to integrated circuits such as chips (or dies) based on the auxiliary link method, thereby assisting in the calibration of the main path circuit. At the same time, real-time online calibration can be performed for at least some radio frequency circuits and devices, thereby effectively improving calibration performance, further enhancing effective radio frequency performance, and reducing the difficulty of realizing radio frequencies.

[0325] Mixers are crucial and essential devices used for frequency conversion in transmit and receive links, and are therefore widely applied in radioelectric devices such as communications and radar, including, for example, single-sideband mixers that have a good suppression effect on mirror signals, and IQ mixers in transmit and receive links as in the embodiments of this disclosure. An IQ mixer (e.g., a single-sideband mixer) transmits a signal with two branches, an I branch and a Q branch, having a phase difference of 90°.

[0326] In application scenarios where the layout area is relatively compact, such as in high-frequency sensor applications in the millimeter-wave band, the physical distances between the I branch and the Q branch, and between the input branch and the output branch before mixing, are all short. As a result, electromagnetic signals can experience signal leakage problems between branches, between input / output ports, and between the IQ matching network and the mixer output, depending on the coupling method (magnetic coupling, substrate coupling, electrical coupling, etc.). Furthermore, this can lead to serious deterioration of the mixer's Miller suppression ratio and local oscillator leakage.

[0327] In selectable embodiments, the mixer-based output network can satisfy impedance matching requirements while ensuring power combining of the I-branch and Q-branch. This disclosure provides a novel mixer structure that effectively improves the isolation between the I-branch and Q-branch while effectively reducing local oscillator signal leakage by improving the mixer output passive network.

[0328] Figure 23 is a schematic diagram of the circuit module of an IQ Mixer in an embodiment of the present disclosure. As shown in Figure 23, the IQ mixer includes an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit. The I-branch mixing unit is configured to output a signal on the I channel, the Q-branch mixing unit is configured to output a signal on the Q channel, and the transformer unit is configured to magnetically couple the I-channel signal and the Q-channel signal to synthesize the IQ mixing output signal and transmit the IQ mixing output signal to the next block. The transformer unit may be installed between the I-branch mixing unit and the Q-branch mixing unit, resulting in a relatively short-circuit type IQ mixer that increases the physical distance between the I-branch mixing unit and the Q-branch mixing unit, further effectively reduces coupling between the I-branch mixing unit and the Q-branch mixing unit, thereby achieving the objective of improving the suppression ratio of the IQ mixer and facilitating layout design.

[0329] A short-connection type IQ mixer performs signal synthesis by short-connecting the outputs of the mixer's I-branch and Q-branch, and then matching them to the next circuit using a matching network, i.e., by electrical coupling. As shown in Figure 23, in an IQ mixer of a selectable embodiment of the present disclosure, one branch inductance is connected in series between the output terminals of the I-branch mixing unit and between the output terminals of the Q-branch mixing unit, and then another magnetically coupled inductance is provided between the two branch inductances to form a three-winding transformer structure. That is, the output terminals of the I-branch mixing unit and the output terminals of the Q-branch mixing unit are magnetically coupled and synthesized to obtain the above-mentioned IQ mixing output signal. Since the common modes of the I-branch mixing unit and the Q-branch mixing unit are not directly connected, they can be separated and adjusted, making the application scenarios of the device even more flexible. The signals are then transmitted directly to the next circuit (Next Block) with both ends of the above-mentioned magnetically coupled inductance as output terminals. In other words, the three-winding transformer structure has four input terminals and two output terminals.

[0330] Based on the three-winding transformer structure shown in Figure 23, when the LO signal and the signal to be mixed (e.g., an echo signal) enter the P and N terminals of the I branch and Q branch, respectively, the current direction and magnetic field direction formed in the three-winding transformer become as shown in Figure 24. At this time, the Isb magnetic field is superimposed, the usb magnetic field is canceled, and an even higher quality single-sideband signal can be output. In other words, based on this three-winding transformer structure, down-mixing or up-mixing of the signal to be mixed can be realized.

[0331] In some selectable embodiments, even if there is a phase difference between the LO signal and the signal to be mixed, the signal cancellation method in the three-winding transformer matches, so if there are no malfunctions in the LO signal and the signal to be mixed itself, the output from the three-winding transformer is still a single-sideband signal.

[0332] The magnetically coupled IQ mixer provided in the embodiments of this disclosure performs power combining using magnetic coupling, compared to a short-connection IQ mixer that performs power combining using electrical coupling. At the same time, the magnetically coupled IQ mixer can perfectly match the I branch and the Q branch, exhibit axial symmetry, and does not introduce extraneous phase errors due to mismatches in the wiring lengths of the I branch and the Q branch. Furthermore, it makes the associated circuitry (e.g., high-frequency circuits such as millimeter-wave) more robust in subsequent manufacturing processes and has a higher resistance to process instability. For example, in the magnetically coupled IQ mixer in the embodiment of this disclosure, as shown in Figure 25, the transformer tap can be placed on the midline between the P and N ports, and the structure of the mixer can be made axially symmetric based on the midline, making the common-mode path lengths of the P port and the N port equal. That is, the symmetry of PN in the magnetically coupled IQ mixer is superior to that of the short-circuited IQ mixer, and the leakage of local oscillator signals is further reduced.

[0333] In some selectable embodiments, the magnetically coupled IQ mixer according to the embodiments of this disclosure can be applied as an upconversion mixer in various electromagnetic circuits, such as in the transmit channel and on-chip self-test. For example, the IQ mixer in each transmit channel (or transmit link) and BIST in the embodiments of this disclosure can employ the magnetically coupled IQ mixer according to the embodiments of this disclosure.

[0334] Figure 26 is a schematic diagram of the physical structure of another IQ Mixer in an embodiment of the present disclosure. As shown in Figure 26, the embodiment of the present disclosure further provides a structure of another magnetically coupled IQ mixer, namely, the tap feed lines of the I-branch and Q-branch partially overlap, for example, in the case of the I-branch, the tap of the I-branch can be bypassed from the Q-branch side (e.g., via a via) and then connected to the common-mode bias voltage. Similarly, the tap of the Q-branch can be bypassed from the I-branch side (e.g., via a via) and then connected to the common-mode bias voltage. The above wiring arrangement makes the common-mode path more reliable and less susceptible to interference from other circuits. At the same time, the three-winding transformer in the IQ Mixer in the embodiment of the present disclosure may be square, octagonal, regular octagonal, etc., i.e., it is sufficient that it can be distributed axially symmetrically along the center between NPs in a top or bottom view.

[0335] As shown in Figure 27, embodiments of the present disclosure further provide a signal transmission method applicable to an electromagnetic wave device having at least one signal transmission link, comprising steps 2701 to 2705. Step 2701: Determine the phase of the radio frequency transmission signal for each signal transmission link. Step 2702: Determine the initial phase of the baseband signal for each signal transmission link based on the phase of the radio frequency transmission signal. Step 2703: Generate an initial baseband signal based on the determined initial phase. Step 2704: Obtain the baseband signal after compensating the initial baseband signal using previously acquired compensation information. Step 2705: Obtain the radio frequency transmission signal by performing a phase shift operation on the transmitting local oscillator signal based on the baseband signal.

[0336] The signal transmission method of the embodiments of this disclosure generates a digital baseband signal in the digital domain using a digital phase shifter architecture, resulting in better orthogonality and lower side lobes. Its phase shift can be generated with great accuracy, leading to higher phase modulation accuracy. This enables high-precision digital phase-shifted automotive radar systems, reduces the demand for antenna isolation, and simultaneously offers advantages such as low link loss, low cost, and no need for offline calibration. It can support more flexible wave transmission solutions, including high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0337] As shown in Figure 28, embodiments of the present disclosure further provide a signal transmission method applicable to an electromagnetic wave device having at least one signal transmission link, comprising steps 2801 to 2804. Step 2801: Observation information of the signal transmission main path is obtained with the current compensation coefficient, the signal transmission main path generates a radio frequency transmission signal after performing a compensation operation on the signal generated based on the compensation coefficient to achieve target detection and / or communication. Step 2802: If the current observation information satisfies the iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is repeated until the obtained observation information satisfies the iteration condition. Step 2803: The baseband signal is compensated using the compensation coefficient. Step 2804: Based on the compensated baseband signal, a phase shift operation is performed on the transmitting local oscillator signal to obtain a radio frequency transmission signal.

[0338] According to the signal transmission method, signal calibration link, transmission link and method, transceiver link, and integrated circuit of the embodiments of the present disclosure, the compensation information generated by the signal calibration link can compensate the generated signal, solve the problems of orthogonal imbalance, LO leakage, and harmonic distortion, and effectively improve the signal quality in the signal transmission main path. The compensation coefficient required for the signal transmission main path can be determined in an iterative manner, and the accuracy of the functional compensation coefficient can be effectively improved.

[0339] As shown in Figure 29, embodiments of the present disclosure further provide a signal transmission method applicable to an electromagnetic wave device having at least one signal transmission link, comprising steps 2901 to 2904: Step 2901, initial observation information O(0), first observation information O(1), and second observation information O(2) corresponding to the signal transmission main path under the conditions of initial compensation coefficients h(0), first compensation coefficient h(1), and second compensation coefficient h(2) with different numerical values, the signal transmission main path generates a radio frequency transmission signal after performing a compensation operation on the signal generated based on the compensation coefficients in order to achieve target detection and / or communication. Step 2902, a third compensation coefficient h(3) is determined using the initial observation information O(0), first observation information O(1), and second observation information O(2). Step 2903, the baseband signal is compensated using the compensation coefficients. Step 2904, a phase shift operation is performed on the transmitting local oscillator signal based on the compensated baseband signal to obtain a radio frequency transmission signal.

[0340] According to the signal transmission method, signal calibration link, transmission link and method, transceiver link, and integrated circuit of the embodiments of the present disclosure, the compensation information generated by the signal calibration link can compensate the generated signal, solve the problems of orthogonal imbalance, LO leakage, and harmonic distortion, and effectively improve the signal quality in the signal transmission main path. Observation information can be obtained through multiple calibration operations to determine the compensation coefficient required for the signal transmission main path and effectively improve the accuracy of the functional compensation coefficient.

[0341] In some exemplary embodiments, the transmitted digital baseband signal is a monophonic signal and the transmitted local oscillator signal is a frequency sweep signal, or the transmitted digital baseband signal is a frequency sweep signal and the transmitted local oscillator signal is a monophonic signal. In some exemplary embodiments, the frequency bandwidth of the frequency sweep signal is 2 GHz or greater.

[0342] Embodiments of this disclosure further provide an integrated circuit which may include sequentially connected high-frequency modules, analog signal processing modules, and digital signal processing modules. The high-frequency modules are used to generate high-frequency transmit signals and receive high-frequency receive signals. The analog signal processing modules are used to down-convert radio frequency received signals to obtain intermediate frequency signals. The digital signal processing modules are used to convert intermediate frequency signals from analog to digital to obtain digital signals. The radio frequency modules include any of the above-described signal transmit links, any of the above-described signal transmit / receive links, any of the above-described signal calibration links, any of the above-described signal compensation links, any of the above-described signal calibration systems, and / or the above-described IQ mixers. and / or the digital signal processing modules perform compensation in the digital domain based on the above-described unequal length compensation method for the feed lines. The radio frequency received signal is an echo signal formed when the radio frequency transmit signal is transmitted and / or scattered by a target. Selectively, the integrated circuit may further include a data processing module for processing digital signals to realize target detection and / or wireless communication, for example, the integrated circuit may be a millimeter-wave radar chip (chip or die).

[0343] In some selectable embodiments, the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, an AoC (Antenna-On-Chip) chip structure, or a RoP (Radiator on Package) structure, in which a radiating structure (Radiator) is placed on a package (Radiator), and a waveguide structure is formed by surrounding the Radiator with a sphere, so that the RF signal is transmitted through the radiating structure to the waveguide structure and then converted from the waveguide structure to an external antenna.

[0344] Other embodiments of the present disclosure further provide an electromagnetic wave sensor. The electromagnetic wave sensor may further include an antenna and an integrated circuit as described above. The integrated circuit is electrically connected to the antenna and transmits and receives electromagnetic wave signals. For example, the electromagnetic wave sensor may include a carrier, the integrated circuit described in any of the embodiments above, and an antenna, the integrated circuit may be installed on the carrier, and the antenna may be installed on the carrier or as an integrated device with the integrated circuit (i.e., in this case the antenna may be an antenna provided in an AiP, AoP, or AoC structure). The integrated circuit is connected to the antenna (i.e., in this case the antenna is not integrated into the sensor chip or integrated circuit, as in conventional SoCs, etc.) and transmits and receives electromagnetic wave signals. The carrier may be a printed circuit board (PCB), and the first transmission line may be PCB wiring.

[0345] Embodiments of this disclosure provide a device body and a device which may include an electromagnetic wave sensor as described above provided on the device body. The electromagnetic wave sensor is used for target detection and / or communication in order to provide reference information for the operation of the device body.

[0346] Embodiments of the present disclosure further provide electronic devices, which may appear in the form of general-purpose computing devices. Components of the electronic devices may include, but are not limited to, at least one processing unit, at least one storage unit, a bus connecting different system components (including the storage unit and the processing unit), a display unit, and the like. Program code is stored in the storage unit, and the program code can be executed by the processing unit, thereby enabling the processing unit to perform methods according to various exemplary embodiments of the present disclosure as described herein. The storage unit may include a readable medium in the form of a volatile storage unit, for example, a random access storage unit (RAM) and / or a cache storage unit, and may further include a read-only storage unit (ROM).

[0347] The storage unit may further include a program / utility having one set (at least one) of program modules, such program modules including, but not limited to, an operating system, one or more application programs, other program modules, and program data, and each of these examples, or any combination thereof, may include the implementation of a network environment.

[0348] The bus may represent one or more of several types of bus structures, including a storage unit bus or storage unit controller, a peripheral bus, a graphics accelerator port, a processing unit, or a local bus that uses one of the bus structures.

[0349] The electronic device may also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth® devices, etc.), and further with one or more devices that allow a user to interact with the electronic device, and / or with any devices (e.g., routers, modems, etc.) that allow the electronic device to communicate with one or more other computing devices. Such communication may be via input / output (I / O) interfaces. The electronic device may also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and / or public networks, e.g., the Internet) via a network adapter. The network adapter may communicate with other modules of the electronic device via a bus. For clarity, other hardware and / or software modules may be used in conjunction with the electronic device, although not shown in the diagram, including but not limited to microcode, device drivers, redundant processing units, external disk drive arrays, RAID systems, tape drives, and data backup storage systems.

[0350] For example, the electronic device in the embodiment of the present disclosure may further include a main body of the device and an electromagnetic wave sensor installed on the main body of the device, the electromagnetic wave sensor which can be used to realize functions such as target detection and / or wireless communication.

[0351] Specifically, based on the embodiments described above, in one optional embodiment of the present disclosure, the electromagnetic wave sensor may be installed outside or inside the device body, and in another optional embodiment of the present disclosure, the electromagnetic wave sensor may be further installed partially inside and partially outside the device body. The embodiments of the present disclosure are not limited thereto and may be specifically determined on a case-by-case basis.

[0352] In one selectable embodiment, the device body may be, for example, components and products applied to fields such as smart cities, smart houses, transportation, smart homes, consumer electronics, security monitoring, industrial automation, shipboard detection (e.g., smart cabins), medical devices, and hygiene and health. For example, the device body may be smart transportation devices (e.g., automobiles, bicycles, motorcycles, ships, subways, trains, etc.), security devices (e.g., cameras), liquid level / flow velocity detection devices, smart wearable devices (e.g., bracelets, glasses, etc.), smart home devices (e.g., cleaning robots, door locks, televisions, air conditioners, smart lamps, etc.), various communication devices (e.g., mobile phones, tablets, etc.), and, for example, barrier gates, smart traffic indicator lamps, smart signs, traffic cameras, and various industrial machine arms (or robots). It may also be various devices for detecting biological characteristic parameters and various devices that mount such devices, for example, biological characteristic detection in car cabins, indoor personnel monitoring, smart medical devices, consumer electronics devices, etc.

[0353] Embodiments of the present disclosure further provide a non-temporary computer-readable storage medium in which computer-readable instructions are stored, and when such instructions are executed by a processor, the processor performs the above-described method for compensating for unequal lengths of power lines.

[0354] As those skilled in the art will understand from the above description of embodiments, the embodiments described herein may be implemented by software or by a combination of software and necessary hardware. The technical solutions according to the embodiments of this disclosure may be embodied in the form of a software product, which may be stored on a non-volatile storage medium (which may be a CD-ROM, USB disk, removable hard disk, etc.) or on a network, and which includes some instructions for causing a computing device (which may be a personal computer, server, or network device, etc.) to perform the above method according to the embodiments of this disclosure.

[0355] A software product may employ any combination of one or more readable media. The readable media may be a readable signal medium or a readable storage medium. The readable storage medium may be, but is not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, or device, or any combination thereof. More specific examples of readable storage media (non-exclusive list) include electrical connections with one or more leads, portable disks, hard disks, random access memory (RAM), read-only memory (ROM), erasable programmable read-only memory (EPROM or flash memory), optical fibers, portable compact disk read-only memory (CD-ROM), optical storage devices, magnetic storage devices, or any suitable combination thereof.

[0356] A computer-readable storage medium may include data signals propagated in the baseband or as part of a carrier, thereby carrying readable program code. The propagated data signals may employ multiple forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. The readable storage medium may further be any other readable medium, which can transmit, propagate, or transmit programs for use by or in combination with instruction execution systems, apparatus, or devices. The program code contained in the readable storage medium may be transmitted by any suitable medium, including but not limited to wireless, wires, optical cables, RF, or any suitable combination thereof.

[0357] Program code for performing the operations of the Disclosure may be written in any combination of one or more program design languages, including object-oriented program design languages ​​such as Java and C++, and also including conventional process program design languages ​​such as the "C" language or similar programs. The program code may be executed entirely on the user's computer, partially on the user's computer, as a standalone software package, partially on the user's computer and partially on a remote computer, or entirely on a remote computer or server. Where a remote computer is involved, the remote computer may be connected to the user's computer via any type of network, including a local area network (LAN) or a wide area network (WAN), or it may be connected to an external computer (for example, via the Internet using an Internet service provider).

[0358] The computer-readable medium carries one or more programs, and when the one or more programs are executed by one of the devices, the computer-readable medium performs the functions described above.

[0359] Those skilled in the art will understand that each of the above-described modules may be arranged in the apparatus as described in the embodiment, or may be appropriately modified to be located in one or more apparatuses different from those in this embodiment. The modules of the above embodiment may be integrated as a single module, or they may be further divided into multiple submodules.

[0360] According to embodiments of the present disclosure, the above method can be performed by submitting a computer program, which includes a computer program or instructions, and when the computer program or instructions are executed by a processor. In one optional embodiment, the integrated circuit may be a millimeter-wave radar chip. The type of digital function module in the integrated circuit may be determined according to actual needs. For example, in a millimeter-wave radar chip, the data processing module may be used for, for example, distance-dimensional Doppler conversion, velocity-dimensional Doppler conversion, constant false alarm probability detection, wave arrival direction detection, point cloud processing, etc., and is used to acquire information such as the distance, horizontal angle, pitch angle, velocity, height, micro-Doppler motion characteristics, shape, dimensions, surface roughness, dielectric properties, etc.

[0361] Furthermore, by transmitting and receiving wireless signals, wireless devices can realize functions such as target detection and / or communication, thereby providing detected target information and / or communication information to the main device, and thereby supporting or controlling the operation of the main device.

[0362] For example, when the above-mentioned device is applied to an advanced driver-assistance system (i.e., ADAS), the wireless device as an in-vehicle sensor (e.g., millimeter-wave radar) can support the ADAS system and enable application scenarios such as adaptive cruise control, automatic emergency braking (i.e., AEB), blind spot monitoring (i.e., BSD), lane change assist (i.e., LCA), rear cross-traffic alert (i.e., RCTA), parking assist, rear vehicle warning, collision avoidance, and pedestrian detection.

[0363] The technical features of the above embodiments may be combined in any way, and for the sake of simplicity, not all possible combinations of the technical features in the above embodiments have been described. However, as long as there is no inconsistency among these combinations of technical features, they should all be considered to fall within the scope described herein.

[0364] The above embodiments represent only preferred embodiments and technical principles used in the present invention, and while their description is relatively specific and detailed, it should not be understood as a limitation on the scope of the patent. Those skilled in the art can make various obvious modifications, readjustments, and substitutions without departing from the scope of protection of the present invention. Therefore, although the present invention has been described in relatively detail by the above embodiments, the present invention is not limited to these embodiments and may include many other equivalent embodiments without departing from the concept of the present invention, and the scope of patent protection of the present invention is determined by the appended claims.

Claims

1. A signal transmission link, applied to an electromagnetic wave sensor, wherein the transmission link includes an analog signal source and a digital phase shifter, the analog signal source is configured to supply an initial analog signal, and the digital phase shifter is configured to generate a phase shift signal in the digital domain to perform a preset phase shift operation on the initial analog signal, and to phase shift the initial analog signal based on the phase shift signal.

2. The signal transmission link according to claim 1, further comprising a transmitting antenna, wherein the transmitting antenna is configured to radiate an initial analog signal after a phase shift into a predetermined spatial region.

3. The signal transmission link according to claim 1, wherein the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter, and a mixer, wherein the digital phase shifter is configured to generate a digital phase shift signal, the digital-to-analog converter is configured to convert the received digital phase shift signal into an analog phase shift signal, and the mixer is configured to perform a mixing operation on the received initial analog signal using the received analog phase shift signal in order to perform a preset phase shift operation on the initial analog signal.

4. The signal transmission link according to claim 3, characterized in that the digital phase shift signal source includes a direct digital frequency combiner, the digital-to-analog converter is an IQ digital-to-analog converter, and the mixer is an IQ mixer.

5. The aforementioned digital phase shift signal is a monophonic signal, and the aforementioned initial analog signal is a frequency sweep signal, or The signal transmission link according to claim 1, characterized in that the digital phase shift signal is a frequency sweep signal and the initial analog signal is a monophonic signal.

6. The signal transmission link according to any one of claims 1 to 5, characterized in that the signal transmission link transmits a frequency-modulated continuous wave signal.

7. A signal transmission link, which includes a signal transmission main path and a signal calibration link integrated on the same integrated circuit, A signal transmission link characterized in that the signal calibration link is configured to calibrate the signal transmission main path in order to acquire compensation information, and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation based on the compensation information in order to realize target detection and / or communication.

8. The signal transmission link according to claim 7, characterized in that the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillation leakage compensation parameter, and an orthogonal unbalance compensation parameter.

9. The signal transmission link according to claim 7, wherein the signal transmission main path includes a first signal source and a phase shifter, the first signal source is configured to generate a first analog signal, and the phase shifter is configured to frequency shift and / or phase shift the first analog signal in order to form a radio frequency transmission signal.

10. If the phase shifter has a non-orthogonal architecture, the phase shifter includes a second signal source and a transmit mixer, wherein the second signal source is configured to generate a second analog signal, and the transmit mixer is configured to perform mixing on the first analog signal and the second analog signal to form the radio frequency transmit signal. The signal transmission link according to claim 9, wherein, if the phase shifter has an orthogonal architecture, the phase shifter includes a second signal source, a digital-to-analog conversion module, and a transmit mixer, wherein the second signal source is configured to generate a first digital signal, the digital-to-analog conversion module is configured to convert the first digital signal to a second analog signal, and the transmit mixer is configured to frequency-shift and / or phase-shift the first analog signal based on the second analog signal in order to form the radio frequency transmission signal.

11. The signal transmission link according to claim 10, wherein the main transmitting path further includes a compensation circuit, the signal input terminal of the compensation circuit is connected to the second signal source, the signal input terminal is connected to the phase shifter, and the compensation circuit merges the compensation signal and the signal output from the second signal source and outputs the result.

12. A signal transmission and reception link, comprising a signal transmission link and a signal reception link according to any one of claims 1 to 11, The aforementioned signal receiving link includes a receiving mixer, an analog-to-digital converter, and a digital signal processing module. The receiving mixer is configured to downconvert the received echo signal based on the received received local oscillator signal in order to obtain an analog intermediate frequency signal; the analog-to-digital converter is configured to convert the received intermediate frequency signal from analog to digital in order to obtain a digital intermediate frequency signal; and the digital signal processing module is configured to process the digital intermediate frequency signal in order to obtain a target parameter. A signal transmission and reception link characterized in that the echo signal is a signal formed when a signal transmitted by the signal transmission link is reflected and / or scattered by an object.

13. The receiving mixer is a real-number mixer, and the analog-to-digital converter is a real-number analog-to-digital converter, or The signal transmission and reception link according to claim 12, characterized in that the receiving mixer is a quadrature mixer and the analog-to-digital converter is a quadrature analog-to-digital converter.

14. The signal transmission and reception link according to claim 12, characterized in that the received local oscillator signal is a frequency sweep signal, or the received local oscillator signal is a monophonic signal.

15. A signal calibration link, comprising the signal transmission / reception link described in claim 12, A signal calibration link characterized in that the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link, and the signal receiving link is configured to calibrate the signal transmitting link.

16. The signal calibration link according to claim 15, characterized in that there is a preset difference frequency between the local oscillator signal of the signal receiving link and the local oscillator signal of the signal transmitting link.

17. The system further includes a BIST module, which is installed between the local oscillator signal source and the receiving mixer. The signal calibration link according to claim 16, characterized in that the BIST template is configured to mix the received local oscillator signal based on a preset frequency offset signal such that there is a preset difference frequency between the local oscillator signal received by the receiving mixer and the local oscillator signal of the signal transmission link.

18. A signal calibration link comprising the signal transmission / reception link and BIST module described in claim 12, A signal calibration link characterized in that the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link via the BIST module, and the signal receiving link is configured to calibrate the signal transmitting link.

19. A signal calibration link comprising two signal receiving links, a BIST module, an auxiliary circuit unit, and a signal transmitting link according to any one of claims 1 to 11, Any of the signal receiving links includes a real number mixer, a real number analog-to-digital converter, and a digital signal processing module, wherein the real number mixer is configured to downconvert a received echo signal based on a received local oscillator signal to obtain an analog intermediate frequency signal, the real number analog-to-digital converter is configured to convert the received intermediate frequency signal from analog to digital to obtain a digital intermediate frequency signal, and the digital signal processing module is configured to process the digital intermediate frequency signal to obtain a target parameter, wherein the echo signal is a signal formed by the reflection and / or scattering of a signal transmitted by the signal transmitting link by an object. A signal calibration link characterized in that the receiving antenna connection ports of the two signal receiving links are sequentially connected to the transmitting antenna connection port of the signal transmitting link via the auxiliary circuit unit and the BIST module, respectively, and the signal receiving links are configured to calibrate the intermediate frequency portion of the signal transmitting link.

20. A signal calibration link of a signal transmission main path, wherein the signal transmission main path generates a radio frequency transmission signal after performing a compensation operation on a signal generated based on a compensation coefficient in order to achieve target detection and / or communication. The signal calibration link of the signal transmission path is configured to acquire current observation information of the signal transmission path at the current compensation coefficient, and if the current observation information satisfies the iteration condition, to use the current compensation coefficient as the compensation coefficient used in the compensation operation of the signal transmission link, otherwise to repeat the current compensation coefficient until the obtained observation information satisfies the iteration condition.

21. The signal calibration link according to claim 20, characterized in that the compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillation leakage compensation parameter, and an orthogonal unbalance compensation parameter.

22. The signal calibration link according to claim 20, characterized in that the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

23. A signal compensation link comprising a signal transmission link and a compensation unit according to any one of claims 1 to 11, wherein the compensation unit is configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion of the signal transmission link.

24. The signal compensation link according to claim 23, characterized in that the compensation unit is configured to compensate the signal transmission link based on calibration data obtained by the signal calibration link according to any one of claims 15 to 22.

25. The signal compensation link according to claim 24, characterized in that, in order to perform real-time calibration operations between operations of the transmission link, the signal receiving link in the signal calibration link is integrated as an auxiliary calibration module in the vicinity of the transmission link to be calibrated.

26. A method for compensating for unequal lengths of power transmission lines, applicable to an antenna array of an electromagnetic wave sensor having at least two signal links, The signal link with the shortest feed line among the at least two signal links is designated as the reference link, and the delay difference of each of the remaining transmission links relative to the reference link is obtained. A method for compensating for unequal lengths of a transmission line, characterized by comprising: performing unequal length compensation for the transmission line in the digital domain with respect to the antenna array based on the aforementioned delay difference.

27. An integrated circuit comprising a sequentially connected radio frequency module, an analog signal processing module, and a digital signal processing module, wherein the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal, the analog signal processing module performs down-conversion processing on the radio frequency reception signal to obtain an intermediate frequency signal, and the digital signal processing module performs analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal. The radio frequency module includes, and / or, a signal transmission link according to any one of claims 1 to 11, a signal transmission / reception link according to any one of claims 12 to 14, a signal calibration link according to any one of claims 15 to 22, a signal compensation link according to any one of claims 23 to 25. The digital signal processing module is an integrated circuit characterized by performing compensation in the digital domain based on the unequal length compensation method for power supply lines described in claim 26.

28. The integrated circuit according to claim 27, further comprising a data processing module, wherein the data processing module processes the digital signals to enable target detection and / or wireless communication.

29. The integrated circuit according to claim 27 or 28, characterized in that the integrated circuit is a millimeter-wave chip.

30. The integrated circuit according to claim 27 or 28, characterized in that the radio frequency received signal is an echo signal formed by the radio frequency transmitted signal being transmitted and / or scattered by a target, and the integrated circuit is a sensor chip.

31. It is an electromagnetic wave sensor, Career and, The carrier is provided with an integrated circuit according to any one of claims 27 to 30, Antenna and, The antenna is provided on the carrier, or the antenna is integrated as an integrated device with the integrated circuit and provided on the carrier. The electromagnetic wave sensor is characterized in that the integrated circuit is connected to the antenna, transmits the radio frequency transmission signal, and / or receives the radio frequency reception signal.

32. It is a device, The main body of the device, The device body includes the electromagnetic wave sensor described in claim 31, The apparatus is characterized in that the electromagnetic wave sensor is used for target detection and / or communication in order to provide reference information for the operation of the main body of the apparatus.

33. A non-temporary computer-readable storage medium that stores computer-readable instructions, and when the computer-readable instructions are executed by a processor, causes the processor to perform the method according to claim 26.

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