Magnetic resonance type wireless power supply device

Abstract

To provide a magnetic resonance type wireless power supply device capable of continuing to maintain constant transmission power and high transmission efficiency over a long transmission distance even when a distance (transmission distance) between a coil on a power supply side and a coil on a power reception side changes.SOLUTION: In a magnetic resonance type wireless power supply device configured to preserve PT symmetry of a negative resistance circuit 20, a resonance circuit 10 on a power supply side including a coil 11 on the power supply side and a resonance circuit 30 on a power reception side including a coil 31 on the power reception side, a Q value of the loop II is set to be higher than the Q value of the loop I so as to oscillate in the loop II where a leakage inductance component of the coil and a capacitance component of the capacitor resonate among the two resonance loop (loop I and loop II) in which the resonance current can circulate when the resonance circuit 10 on the power supply side and the resonance circuit 30 on the power reception side are regarded as multiple resonance circuits mutually coupled by mutual inductance.SELECTED DRAWING: Figure 14

Description

[Technical field]

[0001] The present invention relates to a magnetic resonance type wireless power feeder that feeds power in a non-contact manner by magnetically resonating between a power feeding coil and a power receiving coil. [Background technology]

[0002] Efforts are being made to commercialize wireless power feeders that supply power to electronic devices in a non-contact (wireless) manner without using power cords or the like. The power feed methods used in wireless power feeders are broadly divided into radiative types that transmit energy using radio waves (microwaves) or laser light, and non-radiative types that transmit energy using electric or magnetic fields. Of these, radiative wireless power feeders are currently in the research stage due to the large energy loss that occurs with current technology and the difficulty of improving the efficiency of power transmission. In contrast, non-radiative wireless power feeders have been commercialized in familiar devices such as smartphones and electric toothbrushes.

[0003] Non-radiative wireless power feeders that transmit energy using a magnetic field are further classified into those that use an electromagnetic induction method and those that use a magnetic resonance method. In a wireless power feeder using an electromagnetic induction method, an alternating current is passed through a power feed coil, and the magnetic flux passing through the power feed coil is changed, thereby changing the magnetic flux passing through the power receive coil disposed near the power feed coil, and an induced current is passed through the power receive coil, thereby transmitting power from the power feed coil to the power receive coil. In contrast, a wireless power feeder using a magnetic resonance method is composed of a power feed resonant circuit including a power feed coil and a power receive resonant circuit including a power receive coil having the same resonant frequency, and transmits power by resonating the power feed resonant circuit and the power receive resonant circuit with an alternating magnetic field generated when an alternating current is passed through the power feed coil.

[0004] Compared with wireless power feeders using the electromagnetic induction method, wireless power feeders using the magnetic resonance method have the advantage of being able to transmit power with high efficiency even when the transmission distance is long. In addition, they have excellent robustness against misalignment between the power feed coil and the power receiving coil. For this reason, wireless power feeders using the magnetic resonance method are expected to be used in applications where it is difficult to shorten the transmission distance and the relative positions of the power feed coil and the power receiving coil are not fixed, such as when feeding power to a moving object such as an automobile.

[0005] However, in a wireless power feeder using the magnetic resonance method, the transmission power is strongly dependent on the transmission distance, so in applications where the transmission distance varies, control must be performed to keep the transmission power constant, which not only requires a computer for control, but also increases power consumption and costs. Furthermore, in a conventional control method using a computer, the response speed depends on the processing power of the computer, so it may not be able to follow a sudden change in the transmission distance. In view of this situation, a wireless power feeder using the magnetic resonance method based on the principle of PT (Parity-Time) symmetry (hereinafter, sometimes referred to as a "PT symmetric wireless power feeder") has been proposed in recent years (see Non-Patent Document 1) and has attracted attention. Here, PT symmetry refers to a combination of two symmetries: space inversion symmetry, which means that the laws of physics do not change even if the spatial coordinates are inverted, and time inversion symmetry, which means that the laws of physics do not change even if the direction of time progresses is inverted.

[0006] In a PT symmetric type wireless power supply device, the oscillation frequency is automatically adjusted so that the transmission power is always kept constant even if the transmission distance changes. This eliminates the need for a computer for frequency tracking control, which can save power and reduce costs. Furthermore, it is expected to have a fast response performance that can track even sudden changes in the transmission distance.

[0007] A PT symmetric wireless power supply device is a device in which the AC power supply in a conventional magnetic resonance type wireless power supply device is replaced with a negative resistor. However, a resistor with a negative resistance value does not exist, and in reality, a negative resistance circuit that exhibits electrical behavior similar to that of a negative resistor is used. When an AC power supply is used as in a conventional magnetic resonance type wireless power supply device, the driving frequency is fixed to a specific value, and therefore PT symmetry is not preserved. However, when the AC power supply is replaced with a negative resistance circuit, the driving frequency (oscillation frequency) is no longer fixed to a specific value, and therefore one of the necessary conditions for preserving PT symmetry is satisfied.

[0008] It is already known that a negative resistance circuit can be realized by a circuit composed only of an operational amplifier and multiple resistors (see Non-Patent Document 1). However, the above circuit composed only of an operational amplifier and multiple resistors has the disadvantage of large power loss. As a practical negative resistance circuit with small power loss, a circuit composed of a current sensor, a zero-cross comparator, a half-bridge circuit composed of multiple transistors, and a gate signal generation circuit that generates a gate signal for the transistor based on the output voltage of the zero-cross comparator has already been proposed (see Non-Patent Document 2).

[0009] However, conventional PT symmetric wireless power feeders are driven at high frequencies of about 1 to 3 MHz, and therefore use air-core coils as the power feed coil and the power receiving coil, which requires the coils to be large in size, and has the drawback of limiting the applications in which they can be used (see Non-Patent Documents 1 and 2).In addition, because of the high frequency, when a metal object (conductive object) is placed around the power feed coil and the power receiving coil, the transmission efficiency is reduced due to eddy current loss generated in the conductive object. [Prior art documents] [Non-patent literature]

[0010] [Non-Patent Document 1] S.Assawawоrarit, X.Yu, and S.Fan, “Rоbust wireless power transfer using a nonlinear parity-time symmetric circuit” Nature, 546, 387(2017) [Non-Patent Document 2] J.Zhоu, B.Zhang, W.Xiaо, D.Qiu, and Y.Chen, IEEE Transactiónns оn Industrial Electronics, 66(5), 4097-4107(2019) Summary of the Invention [Problem to be solved by the invention]

[0011] The present invention has been made to solve the above problems, and provides a magnetic resonance type wireless power feeder that can maintain a constant transmission power and high transmission efficiency over a long transmission distance by operating at a low frequency. Another object of the present invention is to provide a magnetic resonance type wireless power feeder that is highly practical, such as being able to miniaturize the power feeder coil and the power receiver coil and being applicable to various applications. [Means for solving the problem]

[0012] The above issues are: A negative resistance circuit; a power supply side resonant circuit including a power supply side coil; a power receiving side resonant circuit including a power receiving side coil; A magnetic resonance type wireless power feeder configured so that PT symmetry is preserved, When the power supply side resonant circuit and the power receiving side resonant circuit are regarded as a multiple resonant circuit mutually coupled by mutual inductance, of the two resonant loops (hereinafter referred to as "loop I" and "loop II") through which the resonant current can circulate, the Q value of loop II is set to be higher than the Q value of loop I so that oscillation occurs in the loop where the leakage inductance component of the coil and the capacitance component of the capacitor resonate. A magnetic resonance type wireless power supply device characterized by This is solved by providing

[0013] The reason for this is explained below. In this section, ease of understanding is prioritized over rigor, and an explanation is given using an equivalent circuit. However, the oscillation frequencies in Mode I and Mode II cannot be expressed as rigorous formulas unless a theory called coupled mode theory (CMT) is used. Therefore, please note that Equation 1.3 and Equation 1.4 that appear below are also not rigorous. A more rigorous explanation using CMT is given in "Theoretical Analysis of PT Symmetric Wireless Power Transfer" in "Form for Carrying Out the Invention".

[0014] As shown in Fig. 1, the magnetic resonance type wireless power feeder of the present invention is composed of primary side (power feed side) circuits 10, 20 and secondary side (power feed side) circuit 30. The primary side circuits 10, 20 include a power feed side resonant circuit 10 having a power feed side coil 11, and a negative resistance circuit 20. The negative resistance circuit 20 is composed only of an operational amplifier and a plurality of resistors. On the other hand, the secondary side circuit 30 is a power receiving side resonant circuit 30 having a power receiving side coil 31.

[0015] In the magnetic resonance type wireless power feeder of FIG. 1, a capacitor is connected in parallel to the power feed coil 11, and a capacitor is also connected in parallel to the power receiving coil 31. This circuit configuration is sometimes called "Parallel-Parallel Topology (PP Topology)". As a circuit configuration other than the PP topology, there is also one in which a capacitor is connected in series to the power feed coil 11, and a capacitor is connected in parallel to the power receiving coil 31. This circuit configuration is sometimes called "Serial-Parallel Topology (SP Topology)". There is also one in which a capacitor is connected in series to the power feed coil 11, and a capacitor is also connected in series to the power receiving coil 31. This circuit configuration is sometimes called "Serial-Serial Topology (SS Topology)".

[0016] The input voltage V input from the negative resistance circuit 20 to the power supply side resonant circuit 10 in FIG. in and the input current I in From the above relationship, the apparent negative resistance R of the negative resistance circuit 20 is n can be defined as follows:

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[0017] As shown in the following formula 1.2, -R n The value of this is the equivalent resistance component R e Oscillation occurs when k is greater than 0. This phenomenon is widely known as negative resistance oscillation. m is the magnetic coupling coefficient, R L represents the load resistance.

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[0018] The power supply side resonant circuit 10 and the power receiving side resonant circuit 30 shown in FIG. 1 have a mutual inductance k m The mutual inductance can be expressed as a multi-resonant circuit in which the power supply coil 11 (FIG. 1) and the power receiving coil 31 (FIG. 1) are coupled by L (hereinafter, the mutual inductance may be expressed as M). L in FIG. 2 represents the self-inductance of each of the power supply coil 11 (FIG. 1) and the power receiving coil 31 (FIG. 1). L(1-k m ) represent the leakage inductance of the power supply side coil 11 (FIG. 1) and the power receiving side coil 31 (FIG. 1), respectively. r1' and r2' represent the winding resistance of the power supply side coil 11 (FIG. 1) and the power receiving side coil 31 (FIG. 1), respectively. r c represents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacity) of the capacitor. In the circuit of Fig. 2, there are two resonant loops (loop I and loop II) in which the resonant current can circulate. In the following, the mode in which the resonant current circulates in loop I may be referred to as "mode I", and the mode in which the resonant current circulates in loop II may be referred to as "mode II".

[0019] Resonant frequency f in mode I I and the resonant frequency f in mode II II is the magnetic coupling coefficient k m Using L and C, the following formulas 1.3 and 1.4 can be expressed, respectively. The following formulas 1.3 and 1.4 express the mode I as k m L and two series resonant circuits (L(1-k m ) and C are connected in series) in parallel resonance mode, and Mode II is L(1-k m 2, in order to simplify the explanation of the principle, L and C are treated as having the same values ​​in the power supply side resonant circuit 10 and the power receiving side resonant circuit 30, but in reality, they do not necessarily have to be the same values.

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[0020] Figure 3 shows the resonant frequency f of Loop I obtained by experiment and calculation. I and the resonant frequency f of Loop II. II The relationship between the transmission distance and the oscillation frequency is shown for the cases where the device is operated at 100 MHz and 150 MHz. From the experimental results shown in Fig. 3, it can be seen that the device oscillates in mode I for all transmission distances. However, the inventors of the present invention considered that there are also experimental conditions for resonating in mode II.

[0021] FIG. 4 shows the frequency characteristics of the resonant impedance in loop I and loop II obtained in the experiment. In this experiment, the impedance was measured at the input end of the power supply side resonant circuit 10. In addition, the load resistance R L The measurement was performed in a state where the resonant frequency f of the loop I was separated from the power receiving side resonant circuit 30. I and the resonant frequency f of loop IIII Bilateral resonance peaks appeared at the positions shown in Fig. 1. The sharpness of this resonance peak reflects the Q value of the resonance circuit, so the resonance loop with the sharper resonance peak can be said to have a higher Q value.

[0022] It has not been clear up until now which of mode I and mode II is selected. In this regard, the present inventor has found that, from the frequency characteristics of the resonance impedance as shown in FIG. 4, the resonance loop with the higher Q value between loop I and loop II is selected. In other words, in the frequency characteristics of the resonance impedance as shown in FIG. 4, the resonance peak of loop I is higher and sharper than the resonance peak of loop II at any transmission distance. Therefore, in this experimental example, the Q value of loop I is higher than that of loop II. In this case, as shown in the experimental results in FIG. 3, the oscillation frequency is always f I (Mode I) is selected. In addition, the iron loss equivalent resistance r c (Fig. 2) When the impedance was increased from 0.2Ω to 2.2Ω, (Mode I(f I ) to Mode II (f II ) was confirmed. This result does not contradict the above consideration. Also, in the experiment, the transition to mode II (f II ) can be selected (details of the experimental results will be described later with reference to FIG. 15).

[0023] The magnetic resonance type wireless power feeder of the present invention is designed to oscillate in Mode II. By using Mode II, the magnetic resonance type wireless power feeder can be suitably used even in applications where the transmission distance is long.

[0024] Because, k in the above formula 1.3 and formula 1.4 mSince is the magnetic coupling coefficient, in Mode I (Equation 1.3 above), as the transmission distance increases, the frequency shifts in the higher direction, whereas in Mode II (Equation 1.4 above), as the transmission distance increases, the frequency shifts in the lower direction. This can also be seen from Figure 3. There is an optimum value for load resistance that maximizes efficiency, known as the optimum load resistance value. In a wireless power transfer circuit with SP topology, the optimum load resistance value can be found using Equation 1.5 below.

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[0025] As can be seen from the above formula 1.5, as the frequency decreases, the optimum load resistance value increases. It is desirable to set the actual load resistance value larger than this optimum load resistance value in order to preserve PT symmetry. The reason is that if the self-inductance of the coil is set small and the load resistance value is set large, the transmission distance over which PT symmetry can be preserved is extended (the detailed reason is explained in "Theoretical Analysis of PT Symmetric Wireless Power Supply" in "Form for Carrying Out the Invention"). In mode II, as the transmission distance increases, the frequency decreases, so that as the transmission distance increases, the optimum load resistance value approaches the actual load resistance value and the efficiency increases. In mode I, the opposite is true, so as the transmission distance increases, the efficiency decreases. Therefore, in applications with long transmission distances, mode II is more advantageous.

[0026] In the magnetic resonance type wireless power feeder of the present invention, it is preferable that the coil driving frequency is set to 100 kHz or less. By operating at such a low frequency, it becomes possible to continuously select Mode II.

[0027] In conventional PT symmetrical wireless power transfer devices, the coils are driven at high frequencies, so air-core power transfer coils and power receiving coils are used. In air-core coils, the iron loss equivalent resistance r cis zero, mode I is always selected and mode II cannot be selected. This can also be confirmed from the experimental results shown in Non-Patent Document 1 and Non-Patent Document 2. By lowering the frequency, the power supply coil and the power receiving coil can be wound around a magnetic core. This allows r c becomes a finite value. On the other hand, the winding resistances r1' and r2' in Fig. 2 can be made smaller than that of an air-core coil because the coil diameter can be reduced by winding the coil around a magnetic core. In this state, the Q value of loop II is higher than the Q value of loop I, so mode II can be selected.

[0028] In addition, lowering the drive frequency has practical advantages since it is possible to reduce power loss in electronic components such as transistors that constitute the negative resistance circuit.

[0029] Furthermore, by winding the power supply coil and the power receiving coil around a magnetic core, the coils can be made smaller, thereby improving the practicality of the magnetic resonance type wireless power supply device.

[0030] In the magnetic resonance type wireless power feeder of the present invention, it is also preferable to connect a capacitor in parallel to the power receiving coil, and to configure the power receiving coil by summing multiple coils in parallel with each other. The power receiving coil can also be configured with a bundled wire (a wire material in which multiple wires (single wires) are bundled) (the power receiving coil is formed by winding the bundled wire). This can reduce the winding resistance of the power receiving coil, making it easier to select mode II. Furthermore, the self-inductance of the power receiving coil can be reduced, which has the effect of preserving PT symmetry over a long transmission distance. When the power receiving coil is configured with a bundled wire, it is more effective to use a bundle of wires (single wires) each having an insulating coating.

[0031] The negative resistance circuit in the magnetic resonance type wireless power feeder of the present invention comprises: A current sensor; a zero-cross comparator which receives the current waveform signal of the power supply coil detected by the current sensor; a gate signal generating circuit that generates a gate signal for a transistor based on an output voltage of the zero cross comparator; a switching circuit configured with a plurality of transistors for driving the power supply coil with an AC current in response to a gate signal output from the gate signal generating circuit; It is preferable to configure the This makes it possible to reduce power loss in the transistor in the low frequency band, and to increase power conversion efficiency.

[0032] It is also preferable to provide the negative resistance circuit with a trigger mechanism for starting up, which makes it possible to start up the magnetic resonance type wireless power feeder even when it is driven at a low frequency.

[0033] It is preferable to provide a phase-lead compensation circuit in the negative resistance circuit for leading and compensating for the delay of the gate signal. When the gate signal is delayed, a large phase difference occurs between the AC current flowing through the power supply coil and the gate signal, making it difficult to maintain PT symmetry. However, by providing the phase-lead compensation circuit, it becomes possible to maintain PT symmetry.

[0034] It is also preferable to provide a DC offset adjustment circuit in the negative resistance circuit. The zero cross comparator outputs a pulse-shaped voltage signal, and it is preferable that the duty ratio of this voltage signal is 1:1 (50%). This is because if the duty ratio deviates from 50%, the transmission power decreases. In this regard, by providing a DC offset adjustment circuit, it becomes possible to adjust the duty ratio of the voltage signal output from the zero cross comparator to 50%.

[0035] Furthermore, it is also preferable to provide the negative resistance circuit with a snubber circuit connected to the gate signal line of the transistor, which can eliminate chattering noise that occurs when the zero-crossing comparator detects a zero-crossing, and supply AC power with less noise to the load. Effect of the Invention

[0036] As described above, the present invention can preserve PT symmetry over a wide range of transmission distances at low frequencies, so that a constant transmission power and high transmission efficiency can be maintained even in applications where the transmission distance varies. In addition, it is possible to provide a magnetic resonance type wireless power supply device that is highly practical, such as being able to miniaturize the power supply coil and the power receiving coil and being applicable to various applications. [Brief description of the drawings]

[0037] [Figure 1] FIG. 1 is a diagram showing an example of an electric circuit in a magnetic resonance type wireless power feeder having PT symmetry and incorporating a negative resistance circuit. [Diagram 2] FIG. 1 shows an equivalent circuit of a PP topology. [Diagram 3] This is a graph showing the relationship between the transmission distance and the oscillation frequency obtained by experiments and calculations when operating at the resonant frequency fI of loop I in Figure 2 and when operating at the resonant frequency fII of loop II. [Figure 4] 3 is a graph showing frequency characteristics of resonant impedance in loop I and loop II of FIG. 2, obtained by an experiment. [Diagram 5] 1 is a diagram showing an electric circuit in a preferred embodiment of a magnetic resonance type wireless power feeder of the present invention. [Figure 6] FIG. 13 is a diagram showing a model of a PT symmetric type wireless power feeder based on the coupled mode theory. [Figure 7] FIG. 1 shows an SP topology. [Figure 8]7 is a diagram showing the relationship between the input voltage uin and the power supply side coil current iL1 in FIG. 6. FIG. [Figure 9] 1 shows a schematic diagram and photograph of the coil used in the experiment. [Figure 10] 1 is a graph showing the frequency characteristics of the Q value of the receiving coil (Rx coil) obtained by experiment and calculation. [Figure 11] 1 is a graph showing the relationship between the magnetic coupling coefficient km and the transmission distance dt obtained by experiment and calculation. [Figure 12] 13 is a graph showing the oscillation of the power supply coil current iL1 and the power receiving coil current iL2 observed with an oscilloscope immediately after the trigger button is pressed. [Figure 13] 13 is a graph showing waveforms in a steady state of a gate signal, an input voltage uin, a receiving-side capacitor voltage uC2, a power-transmitting-side coil current iL1, and a load current iRL obtained by an experiment. [Figure 14] FIG. 1 shows an equivalent circuit of an SP topology. [Figure 15] 1 is a graph showing the relationship between the transmission distance dt and the oscillation frequency obtained by experiment and calculation. [Figure 16] 1 is a graph showing the change in transmission power PL or transmission efficiency η versus transmission distance dt, obtained by experiment and calculation. [Figure 17] 13 is a graph showing changes in transmission power PL and transmission efficiency η when the power supply coil and the power receiving coil are misaligned in the lateral direction (x direction). [Figure 18] 13 is a graph showing changes in transmission power PL and transmission efficiency η when the power supply coil and the power receiving coil are misaligned in the vertical direction (y direction). [Figure 19] 13 is a graph showing changes in transmission power PL and transmission efficiency η when the power supply coil and the power receiving coil are twisted. [Figure 20] (a) A diagram showing the experimental setup used to investigate the tracking of transmission power and transmission efficiency when the transmission distance changes suddenly, (b) a graph showing the change in transmission distance over time, and (c) a graph showing the change in transmission power and transmission efficiency over time. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0038] 1. Embodiments of the magnetic resonance type wireless power supply device of the present invention A preferred embodiment of the magnetic resonance type wireless power feeder of the present invention will be described in more detail with reference to the drawings. FIG. 5 is a diagram showing an electric circuit in a preferred embodiment of the magnetic resonance type wireless power feeder of the present invention. As shown in FIG. 5, the magnetic resonance type wireless power feeder of this embodiment is composed of primary side (power feed side) circuits 10, 20 and secondary side (power feed side) circuit 30. The primary side circuits 10, 20 include a power feed side resonant circuit 10 having a power feed side coil 11, and a negative resistance circuit 20. On the other hand, the secondary side circuit 30 is a power receiver side resonant circuit 30 having a power receiver side coil 31. A load resistor 40 is connected to the power receiver side resonant circuit 30.

[0039] The negative resistance circuit 20 is configured by connecting a current sensor 21, an oscillator 22, a trigger mechanism 23, a zero-cross comparator 24, a gate signal generating circuit 25, a snubber circuit 26, a switching circuit 27, a phase lead compensation circuit 28, and a DC offset adjustment circuit 29 as shown in Fig. 5. Note that the oscillator 22 and the trigger mechanism 23 are not circuit elements required to achieve negative resistance, but are circuit elements required when starting up the negative resistance circuit 20.

[0040] The current sensor 21 detects the waveform of a current in the power supply side coil 11. This current sensor 21 may be any sensor that has a function of converting the current flowing in the power supply side coil 11 into a voltage signal and outputting the voltage signal.

[0041] The oscillator 22 functions as an AC power supply that outputs a low-frequency AC voltage. The frequency of the AC voltage output by the oscillator 22 is not particularly limited, but is preferably set to a low frequency of 100 kHz or less in consideration of making the above-mentioned mode II easier to select and reducing the power loss of the switching circuit 27. However, if the frequency of the oscillator 22 is set too low, there is a risk that the magnetic resonance type wireless power supply device will be difficult to start. For this reason, the frequency of the oscillator 22 is usually set to 20 kHz or more. In this embodiment, the oscillator 22 is set to output an AC voltage having a sine waveform with a frequency of 33 kHz and an amplitude of 100 mV.

[0042] The trigger mechanism 23 is for inducing oscillation of the negative resistance circuit 20 and activating the magnetic resonance type wireless power supply device. In this embodiment, a momentary switch (a switch with a mechanism that turns on only for a moment when the trigger button is pressed) is used for the trigger mechanism 23. Only at the moment when this momentary switch is turned on, an AC voltage from the oscillator 22 is input to the non-inverting input terminal of the zero-cross comparator 24, and a pulse-shaped voltage signal is output from the zero-cross comparator 24. This voltage signal is input to the gate signal generation circuit 25, and a pulse-shaped gate signal for driving the switching circuit 27 is output from the gate signal generation circuit 25 for a moment. This gate signal drives the switching circuit 27 for a moment, and an AC current flows through the power supply coil 11 for a moment. At the next moment, the trigger button turns OFF and the oscillator 22 and zero-cross comparator 24 are disconnected, but at the same time, the feedback line 20a is connected to the non-inverting input terminal of the zero-cross comparator 24, so that the AC voltage generated from the AC current flowing through the power supply coil 11 is input to the zero-cross comparator 24, and oscillation continues.

[0043] The switching circuit 27 supplies a current i L1In this embodiment, the switching circuit 27 is configured by arranging a plurality of transistors (FETI and FETII in FIG. 5) in a half-bridge configuration. The gate signal generating circuit 25 is for generating gate signals for the transistors (FETI and FETII) in the switching circuit 27 based on the pulse-shaped voltage signal output from the zero-crossing comparator 24. In addition to generating the gate signals, the gate signal generating circuit 25 is also responsible for generating the time (dead time) between ON / OFF switching of the plurality of transistors constituting the switching circuit 27.

[0044] The snubber circuits 26 are connected to the gate signal lines 20b of the transistors (FETI and FETII in FIG. 5) that constitute the switching circuit 27. In this embodiment, each snubber circuit 26 is composed of a capacitor, a resistor, and a diode. The snubber circuits 26 remove chattering noise that occurs when the zero-cross comparator 24 detects a zero-cross, making it possible to supply AC power with little noise superimposition to the load.

[0045] The phase lead compensation circuit 28 adjusts the current i L1 The gate signal and i of the transistors (FETI and FETII in FIG. 5) constituting the switching circuit 27 are connected to the rear stage of the current sensor 21 for detecting L1 It is desirable that the phase difference between the gate signal and i be zero. However, in an actual circuit, a delay always occurs. For this reason, the phase lead compensation circuit 28 compensates for this delay and adjusts the phase difference between the gate signal and i L1 The phase difference between the phase lead compensation circuit 28 and the phase lead compensation circuit 28 is set to be zero. In this embodiment, the phase lead compensation circuit 28 is configured by a circuit using a resistor and a capacitor, but is not limited to this configuration.

[0046] The DC offset adjustment circuit 29 is connected between the phase lead compensation circuit 28 and the zero crossing comparator 24. The DC offset adjustment circuit 29 has a role of adjusting the DC offset voltage of the current waveform of the power supply side coil 11 detected by the current sensor 21. If the duty ratio of the gate signal of the transistors (FETI and FETII in FIG. 5) constituting the switching circuit 27 deviates from 1:1 (50%), the transmission power decreases. Therefore, it is desirable that the duty ratio is 50%, and the DC offset adjustment circuit 29 makes it possible to adjust the duty ratio to 50% (the gate signal when the duty ratio was adjusted to 50% in the experiment is shown in FIG. 15(a) below).

[0047] Among the components of the negative resistance circuit 20, the current sensor 21, the zero-cross comparator 24, the gate signal generating circuit 25, the snubber circuit 26, the switching circuit 27, the phase lead compensation circuit 28, and the DC offset adjustment circuit 29 function as a current waveform feedback type half-bridge inverter, and constitute the main circuit of the negative resistance circuit. Therefore, the wireless power feeder circuit shown in FIG. 5 has PT symmetry, and the magnetic resonance type wireless power feeder of this embodiment corresponds to the above-mentioned PT symmetric type wireless power feeder. Therefore, in the magnetic resonance type wireless power feeder of this embodiment, even if the transmission distance (the distance from the power feeder coil 11 to the power receiver coil 31) changes, a frequency at which the transmission power is kept constant is automatically selected. Therefore, a computer or the like for frequency tracking control is not required.

[0048] In addition, in the magnetic resonance type wireless power feeder of this embodiment, as shown in Fig. 2 above, when the power supply side resonant circuit 10 and the power receiving side resonant circuit 30 are regarded as a mutually coupled multi-resonant circuit, in two resonant loops, loop I and loop II, in which the resonant current circulates, the Q value of loop II is set to be higher than the Q value of loop I. Therefore, even if the transmission distance changes, the operation continues in mode II. Therefore, the magnetic resonance type wireless power feeder of this embodiment can be suitably used in applications in which the transmission distance is long.

[0049] In the magnetic resonance type wireless power feeder of this embodiment, the Q value of loop II is higher than the Q value of loop I (mode II is selected instead of mode I) due to the following ingenuity.

[0050] That is, in the magnetic resonance type wireless power feeder of this embodiment, as shown in Fig. 9(a) described later, the power feed coil 11 and the power receiving coil 31 are wound around the magnetic cores 11a and 31a, respectively. c (see FIG. 2) to a finite value, the Q value of loop II can be made higher than the Q value of loop I, making it easier for mode II to be selected.

[0051] In addition, in the magnetic resonance type wireless power feeder of this embodiment, as shown in FIG. 5, a capacitor (a capacitor represented by capacitance C2 in the figure) is connected in parallel to the power receiving coil 31, and the power receiving coil 31 is configured by summing two coils in parallel (hereinafter, this configuration may be referred to as "parallel two-winding"). As a result, the winding resistance is lowered compared to when one coil is wound (parallel one-winding), and mode II is more likely to be selected. In addition, by making the parallel two-winding, the self-inductance is smaller compared to when the parallel one-winding is used, so that the effect of preserving PT symmetry over a long transmission distance is also achieved (the principle is explained in "Theoretical analysis of PT symmetric wireless power feed" in "Description of the embodiment of the invention"). Note that, although the parallel two-winding is adopted in this embodiment, the power receiving coil 31 can be made to have two or more windings. For example, the same effect can be confirmed when the power receiving coil 31 is made of three parallel windings, which is a summing connection of three coils in parallel.

[0052] The same effect (similar to that obtained by a two-winding parallel coil) can be obtained by forming the power-receiving coil 31 from a bundled wire (a wire material in which a plurality of wires (single wires) are bundled). However, if a typical Litz wire (a wire material in which each wire is twisted in a bare state) is used as the bundled wire, the space factor is poor and it is difficult to achieve an excellent effect. For this reason, when forming the power-receiving coil 31 from a bundled wire, it is preferable to use a bundled wire in which each wire has an insulating coating. For example, a three-layer insulated electric wire (reinforced insulated electric wire) "Litz wire type TEX-ELZ" manufactured by Furukawa Electric Co., Ltd. can be suitably used. However, the bundled wire forming the power-receiving coil 31 does not necessarily have to be a twisted wire.

[0053] 2. Theoretical analysis of the magnetic resonance type wireless power feeder of the present invention Next, a theoretical analysis of the magnetic resonance type wireless power feeder of the present invention will be performed. The magnetic resonance type wireless power feeder of the present invention corresponds to the above-mentioned PT symmetric type wireless power feeder. Figure 6 shows a model of the PT symmetric type wireless power feeder based on the coupled mode theory (CMT).

[0054] In Figure 6, energy mode a n =A n e j(ωt+θ n ) is expressed as a wave function of the resonant circuit, where n=1 indicates the power supply side resonant circuit and n=2 indicates the power receiving side resonant circuit. n | 2 corresponds to the amount of energy stored in each resonant circuit. The angular frequency ω represents the oscillation frequency when the two resonant circuits are coupled, and depends on the strength of the interaction between the two resonant circuits. θ n is a n and θ1-θ2 corresponds to the phase difference between a1 and a2. n Amplitude A n must be treated as a function of time because it grows gradually as the oscillations reach a steady state. 10 and load loss rate Γ L are supplied by the negative resistance and the load resistance, respectively. 10 and Γ 20 are the inherent loss rates of the power supply resonant circuit and the power receiving resonant circuit, respectively. κ is the coupling rate between the couplers. These rates g 10 ,Γ L ,Γ 10 ,Γ 20 , κ are all s -1 The gain rate and the sum of all the loss rates are balanced (g 10 = Γ 10 +Gamma 20 +Gamma L ) and the two resonant circuits are strongly coupled (κ>Γ 20 +Gamma L) is satisfied, PT symmetry is preserved and the transmission power is independent of the transmission distance. In a situation where the driving frequency is fixed, as in a conventional magnetic resonance type wireless power feeder, these two conditions cannot be satisfied, so a PT symmetric type wireless power feeder needs to use a negative resistance circuit, whose oscillation frequency is not fixed, as the power supply circuit.

[0055] J. Zhō et al. have studied the SS topology of a PT symmetric wireless power supply device (Non-Patent Document 2). They have shown through theoretical analysis and experimental results that the transmission power and efficiency are independent of the transmission distance. However, a similar theoretical analysis has not yet been performed for the SP topology. Since the magnetic resonance type wireless power supply device of the present invention adopts the SP topology (see FIG. 5), we present a theoretical analysis of the SP topology and show that the transmission power is independent of the transmission distance. We also theoretically explain a method for preserving PT symmetry over a long transmission distance at low frequencies.

[0056] The SP topology of a PT symmetric wireless power supply is shown in Fig. 7. The input voltage u in is the current flowing through the power supply coil i L1 It is expressed as the sign function of V in the following formula 2.1. DC is the DC power supply voltage of the negative resistance circuit 20. The power supply side resonant circuit 10 is configured by connecting a capacitor and a coil in series, so it can be regarded as a circuit (RLC series circuit) in which a resistor, a coil, and a capacitor are connected in series. In an RLC series circuit, when a pulse waveform voltage is input, a sinusoidal current flows. Therefore, i L1 and u in The waveform is as shown in FIG.

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[0057] Coil currents on the power supply and receiving sides i L1 ,i L2and the coil voltage u L1 ,u L2 Taking into account the mutual inductance M, the relationship is expressed by the following equation 2.2.

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[0058] u L1 ,u L2 and the capacitor voltage u on the power supply side and power receiving side C1 ,u C2 can be expressed by the following formula 2.3 from Kirchhoff's law: Here, r1 and r2 represent the resistance components of the power supply coil and the power receiving coil, respectively.

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[0059] u C1 ,u C2 and i L1 and the receiving capacitor current i C2 The relationship is expressed as the following formula 2.4. In the following formula 2.4, -i C2 is derived from Kirchhoff's law (i L2 -u C2 / R L ) is used.

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[0060] From the above formulas 2.1 to 2.4, the state equation of the following formula 2.5 is obtained. Here, ζ=L1L2-M 2 It is.

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[0061] The following Table 1 shows the correspondence between the CMT parameters in FIG. 6 and the circuit parameters in FIG. 7. m is the magnetic coupling coefficient. Also, ω1 and ω2 are the inherent resonant angular frequencies of the power supplying resonant circuit and the power receiving resonant circuit when there is no interaction between them. [Table 1]

[0062] Considering that the electrical energy stored in the power supply resonant circuit and the power receiving resonant circuit circulates between the coil and the capacitor, the coil current i Ln and the capacitor voltage u Cn The wave function of A n Using these, it can be expressed by the following formula 2.6a and formula 2.6b.

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[0063] A n should be treated as a function of time in the time domain (transient domain) from the start of oscillation to the steady state, so a n The derivative of can be expressed as follows:

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[0064] By substituting the above formula 2.6a and the above formula 2.6b into the above formula 2.5, dA n / dt and A n (ω+dθ n / dt) can be expressed by the following formulas 2.8a to 2.8d.

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[0065] A n The time scale of the change in is sufficiently longer than the period τ=2π / ω of the oscillation frequency. This is the A n This can also be seen from the fact that the time constant 2L1 / r1 of is sufficiently longer than τ. Therefore, the AC component of the last term in the above equations 2.8a to 2.8d is A n Since the effect on is small, it can be considered negligible. Therefore, if the AC component of the product of the sign function and the cosine function is ignored, it can be approximated as in the following equations 2.9a to 2.9d.

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[0066] The above formulas 2.8a to 2.8d can be rewritten as the following formulas 2.10a to 2.10d according to the correspondence in Table 1. Here, V DC / (π(2L1) 1 / 2 ) and the phase difference θ1-θ2 are defined as G0 and φ, respectively. Also, from these equations, k m =M / (L1L2) is used to eliminate L1 and L2.

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[0067] a n The derivative of can be expressed by the following formulas 2.11a and 2.11b by substituting the above formulas 2.10a to 2.10d into the above formula 2.7. Here, 1-k m 2 is approximated to 1. Furthermore, since ω1 = ω2 is a necessary condition for maintaining PT symmetry, the inherent resonant angular frequencies of the two resonant circuits are unified at ω0.

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[0068] k m is a dimensionless parameter, and s -1 Other CMT parameters (ω0, G0, Γ 10 ,Γ 20 ,Γ L ,κ) has a different dimension. Furthermore, k m The magnitude of is equal to 2κ / ω0, which is much smaller than the other parameters. Therefore, k m The term containing a n This has almost no effect on the differential value, so it was ignored. By this processing, the above equation 2.5, which is the state equation of the SP topology, can be converted into the coupled mode model of the following equation 2.12, which is a mathematical expression in CMT.

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[0069] Next, let us consider the state after a sufficient amount of time has passed since the start of oscillation (steady state). In the steady state, A n saturates, and the relationship between the gain rate and the loss rate is G0 / |a1|-Γ 10 = Γ 20 +Gamma L where G0 / |a1| is the gain rate g 10 (Table 1). The above formula 2.12 is G0 / |a1|-Γ 10 = Γ 20 +Gamma L Substituting this into the equation, we obtain the relationship shown in Equation 2.13 below.

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[0070] The characteristic equation of the above formula 2.12 is expressed as the following formula 2.14.

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[0071] From the condition for the imaginary part of the above formula 2.14 to be zero, the oscillation angular frequencies of the two oscillation modes (mode I and mode II) expressed by the following formulas 2.15a and 2.15b can be specified. l is the resonant frequency f I This is a more accurate expression of the angular frequency corresponding to (Equation 1.2 above), and is expressed by the following Equation 2.15b: h is the resonant frequency f in the above-mentioned loop II (mode II). II This is a more accurate expression of the angular frequency corresponding to (equation 1.3 above).

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[0072] From the relationship in equation 2.13 above and the condition for the real part of equation 2.14 above to be zero, the energy stored in the receiving-side resonant circuit is derived as shown in equation 2.16 below.

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[0073] By using the above formula 2.6b and the above formula 2.16, the transmission power P L is derived using the following equation 2.17.

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[0074] By going through the same process, the inherent power loss (power consumed by anything other than the load resistance) of the power supply resonant circuit and the power receiving resonant circuit is 2Γ 10 |a1| 2 and 2Γ 20 |a2| 2 Therefore, the transmission efficiency η can be expressed by the following formula 2.18 using the relationship in formula 2.13 above.

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[0075] From the above equations 2.17 and 2.18, it can be seen that the transmission power and efficiency do not depend on the coupling rate κ. m (See Table 1.) mSince the value of is determined by the transmission distance (see Figure 11 below), the fact that the above formulas 2.17 and 2.18 do not depend on κ means that they are also independent of the transmission distance. In other words, in the SP topology, under conditions where PT symmetry is preserved, the transmission power and efficiency are theoretically constant and independent of the transmission distance. Also, k m However, even in this case, the transmission power and efficiency are theoretically kept constant, so it is expected that a wireless power transfer device that is robust against positional misalignment can be realized.

[0076] Conditions under which PT symmetry is not preserved (κ<Γ 20 +Gamma L ), the oscillation angular frequency ω coincides with ω0. In this case, the transmission power and efficiency strongly depend on the coupling rate κ, as shown in the following formulas 2.19 and 2.20. Therefore, the transmission power and efficiency change significantly with changes in the transmission distance.

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[0077] Condition for PT symmetry to be preserved (κ>Γ 20 +Gamma L ), and the condition for which PT symmetry is not preserved (κ<Γ 20 +Gamma L ) the critical binding rate κ, which is the boundary between conservation and non-conservation, C is κ C = Γ 20 +Gamma L This critical coupling rate κ C is expressed as the critical magnetic coupling coefficient k mC Here, Q2 is the Q value of the receiving coil. k mCSince it can be converted into a transmission distance, the limit transmission distance (critical transmission distance) at which PT symmetry can be preserved can be found from the following equation 2.21.

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[0078] In an actual magnetic resonance type wireless power supply device, the second term in the above formula 2.21 has a larger value than the first term, so in order to maintain PT symmetry over a long transmission distance, ω0C2R L must be set as large as possible. However, the load resistance R L As shown in the following formula 2.22, there exists an optimum value for R that maximizes the transmission efficiency. L If the value of exceeds this optimum value and is increased without limit, the transmission efficiency will decrease significantly.

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[0079] Therefore, in order to maintain PT symmetry over a long transmission distance, it is important to set ω0C2 as large as possible. However, ω0 and C2 have a trade-off relationship, and increasing one of them decreases the other, so ω0C2 cannot be increased by adjusting only the values ​​of ω0 and C2.

[0080] ω0C2 is (C2 / L2) 1 / 2 Therefore, we can conclude that if the self-inductance L2 of the receiving coil is reduced, ω0C2 becomes larger and PT symmetry can be maintained over a long transmission distance even at low frequencies.

[0081] 3. Experiment In order to confirm the effectiveness of the magnetic resonance type wireless power feeder of the present invention, the following experiments and calculations were carried out.

[0082] 3.1 Power supply coil and power receiving coil We confirmed what kind of coils are suitable for use as the power supply coil and the power receiving coil. Figure 9 shows a schematic diagram and a photograph of the power supply coil 11 and the power receiving coil 31 used in the experiment, with Figure 9(a) showing a solenoid coil and Figure 9(b) showing a spiral coil. In the following, the power supply coil 11 is denoted with "Tx" and the power receiving coil 31 is denoted with "Rx". The specifications of the solenoid coil and the spiral coil are shown in Table 2 below. For the magnetic core (magnetic core), Mn-Zn ferrite with an initial magnetic permeability of 2300 x μ0 (μ0 is the magnetic permeability of a vacuum) and a saturation magnetic flux density of 0.39 T was used. For the winding, a bundled wire was used in which seven single-build copper wires with a diameter of 0.18 mm or 0.30 mm (the one with a diameter of 0.30 mm was used for the Tx solenoid coil) were twisted together. By configuring the receiving coil with two windings in parallel, the self-inductance was reduced to about one-quarter (484μH→119μH) compared to when a single coil was wound. At this time, there was no significant change in the Q value of the coil (253→242).

[0083] [Table 2]

[0084] Figure 10 shows the frequency response of the Q-factor of the Rx coil obtained by experiment and calculation. The calculation results are in good agreement with the experimental results. The Q-factor measured at a frequency of 40 kHz was 242 for the solenoid coil and 66 for the spiral coil. Figure 11 shows the k-factor obtained by experiment and calculation. m and d t The graph shows the relationship between the transmission efficiency and the solenoid coil. At transmission distances of 30 mm or less, the spiral coil outperforms the solenoid coil, but at transmission distances of 30 mm or more, there is almost no difference between the two types of coils. The transmission efficiency is k m It is already known that it is determined by the product of and Q. tThe theoretical maximum transmission efficiency at a frequency of 40 kHz and a diameter of 40 mm was 91% for the solenoid coil and 72% for the spiral coil. In addition, samples of the solenoid coil and spiral coil were also fabricated, and similar results were obtained with those samples.

[0085] From the above results, it was confirmed that a solenoid coil is more preferable than a spiral coil as the power supply coil and the power receiving coil in the magnetic resonance type wireless power supply device of the present invention.

[0086] 3.2 Operation and performance of magnetic resonance type wireless power transfer device Next, an experiment was carried out to confirm the oscillation and transmission performance (transmission efficiency, transmission power, robustness against coil position shift, etc.) of the magnetic resonance type wireless power feeder of the above embodiment. The magnetic resonance type wireless power feeder shown in FIG. 5 was used.

[0087] In FIG. 12, the current i in the power supply coil 11 (FIG. 5) is L1 and the current i of the power receiving coil 31 (FIG. 5) L2 The results of observing with an oscilloscope how the oscillator 22 (Fig. 5) begins to oscillate are shown. The oscillator 22 (Fig. 5) was set to output a sine wave with a frequency of 33 kHz and an amplitude of 100 mV. However, oscillation was confirmed in any case as long as the sine wave had an amplitude of 100 mV or more and was between 20 and 100 kHz. From the above results, the effectiveness of the trigger mechanism 13 was confirmed.

[0088] FIG. 13 shows the gate signal and the voltage (u in ,u C2 ) and current (i L1 ,i RL) with an oscilloscope. It was confirmed that the duty ratio of the gate signal can be adjusted to 1:1 (50%) by using the DC offset adjustment circuit 29 (Fig. 5). It was also confirmed that the chattering noise generated in the zero-cross comparator 24 (Fig. 5) can be removed by adding a snubber circuit 26 (Fig. 5) consisting of a resistor, a capacitor, and a diode to the gate signal line 20b (Fig. 5). It was also confirmed that the addition of a phase-lead compensation circuit 28 (Fig. 5) is effective in compensating for the time delay of the gate signal. The power efficiency of the negative resistance circuit 20 (Fig. 5) reached 97% by using a low on-resistance (2.3 mΩ) silicon field-effect transistor (Si-FET) used as the transistor in the switching circuit 27 and by setting the time between switching ON and OFF of the two Si-FETs (dead time) to a relatively long value of 400 ns (see the photograph in Fig. 13(a)).

[0089] FIG. 14 shows an equivalent circuit of the SP topology. The magnetic resonance type wireless power feeder of this embodiment employs the SP topology. The existence of two oscillation modes (mode I and mode II) has been explained using the equivalent circuit of the PP topology in FIG. 2, but the only difference between the SP topology and the PP topology is whether the capacitor of the power feed side resonant circuit 10 is connected in series or in parallel to the power feed side coil. Therefore, the SP topology also has two oscillation modes (mode I and mode II) like the PP topology. The mathematical formulas for the respective oscillation frequencies are also the same. Furthermore, the property that oscillation occurs in the loop with the higher Q value of the two resonant loops (loop I and loop II) is the same in the SP topology and the PP topology.

[0090] Figure 15 shows the transmission distance d t The relationship between the oscillation frequency and the transmission distance is shown in Fig. 15. The three curves in Fig. 15 are the results of calculations using CMT. Below the theoretical critical transmission distance of 40 mm, there are two oscillation modes. The mode shown as the PTS mode in Fig. 15 is mode I(ω lThe mode labeled as the anti-PTS mode corresponds to mode II (ω h ) In addition, when the transmission distance exceeds 40 mm, the PT symmetry is not preserved, and therefore, in FIG. 15, it is shown as an asymmetric mode. The magnetic resonance type wireless power feeder of this embodiment is designed to operate in the anti-PTS mode (mode II), and it was confirmed from the experimental results in FIG. 15 that the oscillation occurred in mode II at any transmission distance as intended.

[0091] Figure 16 shows the transmission distance d t Transmission power P L 16 shows the change in transmission efficiency η with respect to the PT symmetry. In FIG. 16, the data points shown as filled circles are the experimental results for the magnetic resonance type wireless power feeder of this embodiment. The solid and dashed lines in the figure are the calculation results using CMT. Furthermore, the data points shown as hollow circles in the figure are the experimental results when the feedback line 20a (FIG. 5) is cut and the operating frequency is fixed at 35 kHz. When the feedback line 20a is cut and the operating frequency is fixed, the PT symmetry is not preserved at any transmission distance. In other words, the data points shown as hollow circles correspond to the experimental results for the conventional magnetic resonance type wireless power feeder that does not have PT symmetry. The effectiveness of the present invention can be confirmed by comparing the results shown by filled circles and hollow circles.

[0092] First, P of the magnetic resonance type wireless power supply device of this embodiment L The values ​​were within 23±1W in the range of 12 to 40 mm. Compared with the experimental results of the conventional method shown as hollow circles, it is clear that the preservation of PT symmetry improves the robustness against the transmission distance. Since no solenoid coil with a transmission distance longer than the square root of the magnetic pole area has been found, we chose a solenoid coil with a magnetic pole area of ​​585 mm. 2 In the coil of d t The transmission distance of 40 mm is deemed sufficiently long. Lincreases sharply when it exceeds 40 mm. This coincides with the theoretical critical transmission distance. In addition, when the Rx coil was changed from two parallel turns to three parallel turns, the self-inductance decreased from 119 μH to 53 μH, and the critical transmission distance increased from 40 mm to 53 mm. On the other hand, when the Rx coil was changed from two parallel turns to one parallel turn, the self-inductance increased from 119 μH to 484 μH, and the critical transmission distance decreased from 40 mm to 31 mm. From this, it was confirmed that the parallel winding (two parallel turns or three parallel turns) adopted in the power receiving coil 31 of the magnetic resonance type wireless power feeder of this embodiment is an effective method for extending the transmission distance while preserving the PT symmetry.

[0093] In addition, the value of η of the magnetic resonance type wireless power feeder of this embodiment remains at 83±1% up to a transmission distance of 40 mm. This value of η is slightly lower than that of an air-core high-frequency coil that does not have iron loss. However, since there have been no reports of coils with an efficiency of 75% or more among coils with similar size and power, this value of 83% is deemed to be sufficiently high. When the winding of the Rx coil is changed from two parallel turns to three parallel turns, the efficiency decreased from 83%±1% to 73%±1% due to the decrease in the Q value (242→184). Future theoretical investigation is required for this decrease. In any case, these experiments confirmed that preserving PT symmetry improves robustness against transmission distance.

[0094] The coil position deviation is the transmission power P L It is important to understand the effect of the misalignment on the transmission efficiency η for practical use. L and the transmission efficiency η. L The change in the transmission power P and the transmission efficiency η are shown in Fig. 19. Land the change in transmission efficiency η. For comparison, the experimental results (hollow circles) for the conventional method (when the driving frequency is fixed at 35 kHz) are shown. As can be seen from Figs. 17 to 19, in all the graphs, the magnetic resonance type wireless power feeder of this embodiment has improved robustness against positional deviation compared to the conventional method. For example, the permissible range in the x direction that can hold a constant power (23±1 W) is ±25 mm, the permissible range in the y direction is ±40 mm, and the permissible range of the rotation angle is ±65°. In particular, considering that the permissible range of the rotation angle in the conventional method is about ±5°, it can be said that the preservation of PT symmetry has brought about a great advancement.

[0095] Furthermore, the transmission distance d t When the transmission power P changes suddenly L However, the transmission distance d t Since it was not possible to obtain an actuator capable of providing a large and fast displacement to the actuator, a small solenoid coil (dimensions: 50 mm x 23 mm x 6.5 mm, mass: 39.0 g) was fabricated. t A photograph of the experimental setup is shown in Fig. 20(a). A linear actuator was used to apply a step input to the position of the Rx coil. The change in the position of the Rx coil was measured using a laser displacement sensor. t When a step input of 7 mm displacement is applied to t The graph showing the time change of P around t=0 when the step occurs is shown in Fig. 20(c). L The measurement results of P and η are shown below. L From this, the magnetic resonance type wireless power feeder of this embodiment has a small change in d t It was confirmed that the sensor has a fast response speed capable of following even sudden changes in the sensor voltage. [Explanation of symbols]

[0096] 10 Power supply side resonant circuit 11 Power supply coil 11a Magnetic core (magnetic core) 20 Negative resistance circuit 20a Feedback line 20b Gate signal line 21 Current Sensor 22 Oscillator (AC power supply) 23 Trigger Mechanism 24 Zero-cross comparator 25 Gate signal generation circuit 26 Snubber Circuit 27 Switching Circuits 28 Phase lead compensation circuit 29 DC offset adjustment circuit 30 Receiving side resonant circuit 31 Receiving coil 31a Magnetic core (magnetic core) 40 Load Resistance

Claims

1. A negative resistance circuit; a power supply side resonant circuit including a power supply side coil; a power receiving side resonant circuit including a power receiving side coil; A magnetic resonance type wireless power feeder configured so that PT symmetry is preserved, When the power supply side resonant circuit and the power receiving side resonant circuit are regarded as a multiple resonant circuit mutually coupled by mutual inductance, of the two resonant loops (hereinafter referred to as "loop I" and "loop II") through which a resonant current can circulate, the Q value of loop II is set to be higher than the Q value of loop I so that oscillation occurs in the loop in which the leakage inductance component of the coil and the capacitance component of the capacitor resonate. A magnetic resonance type wireless power supply device.

2. 2. The magnetic resonance type wireless power feeder according to claim 1, wherein the driving frequency is set to 100 kHz or less.

3. 3. The magnetic resonance type wireless power feeder according to claim 1, wherein the power feed coil and the power receiving coil are wound around a magnetic core.

4. A capacitor is connected in parallel to the receiving coil, The receiving coil is composed of multiple coils connected in parallel or composed of bundled wires. The magnetic resonance type wireless power feeder according to any one of claims 1 to 3.

5. 5. The magnetic resonance type wireless power feeder according to claim 1, wherein the negative resistance circuit is provided with a trigger mechanism for starting the negative resistance circuit.

6. The negative resistance circuit is A current sensor; a zero-cross comparator which receives the current waveform signal of the power supply coil detected by the current sensor; a gate signal generating circuit that generates a gate signal for a transistor based on an output voltage of the zero cross comparator; a switching circuit configured with a plurality of transistors for driving the power supply coil with an AC current in response to a gate signal output from the gate signal generating circuit; 6. The magnetic resonance type wireless power feeder according to claim 1, wherein the magnetic resonance type wireless power feeder comprises:

7. The magnetic resonance type wireless power feeder according to claim 6 , wherein the negative resistance circuit further comprises a phase lead compensation circuit for leading-compensating for a delay of the gate signal.

8. 8. The magnetic resonance type wireless power feeder according to claim 6, wherein the negative resistance circuit further comprises a DC offset adjustment circuit for adjusting a duty ratio of the gate signal.

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