AC-DC Converter

Abstract

This paper describes an AC-DC conversion system. The conversion system consists of an electronic switch and control circuit used to provide controlled pulsed power to a storage element, which in turn provides power to a load at either a preselected voltage or a manually or automatically selectable voltage, while ensuring that the voltage drop across the switch is minimized, reducing the power dissipated through the switch itself, thereby significantly increasing efficiency and reducing heat losses. In one minimal version, the AC-DC converter consists of a pair of N-MOSFET transistors, a voltage divider, a storage element, and a pair of diodes. This design allows for high efficiency with a minimum of components that can be fully integrated on silicon.

Description

[Technical field]

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS This application claims priority to U.S. Provisional Application No. 62 / 987,045, entitled "Zero Voltage Switching AC Direct Power Regulator and Discriminator," filed March 9, 2020. Both applications include common inventors and are currently pending. <Statement regarding Federally Sponsored Research or Development> Not applicable. <Technical field> The present invention relates to a power management system and method for providing low voltage DC current from an AC mains power source with very high efficiency. [Background technology]

[0002] The traditional means for providing DC power from a mains AC source (mains power) was via an analog circuit including a step-down transformer, a diode rectifier, and a filter with electrolytic capacitors and resistors. The output voltage depended primarily on the turns ratio of the transformer, and the circuit was reasonably efficient. However, the size and weight of the magnetic structures required to implement the low-frequency transformer did not necessitate the use of this approach in small devices.

[0003] The latter approach, which does not use a transformer, involves direct rectification of the AC mains supply, and the rectified waveform is applied directly to a voltage regulation circuit that includes an active solid-state device, either in series or shunt connection. The shunt regulator configuration works by providing a current path across the rectified mains output through a variable resistance device, thereby diverting the current away from the load. In the simplest embodiment of a shunt regulator, a Zener diode is shunted across the load with a resistor in series with the shunt leg. Any rectifier output voltage that exceeds the Zener voltage drops across the resistor, resulting in excess power being dissipated as heat. This regulator configuration is therefore very inefficient. Because the Zener current must be greater than the load current to maintain regulation by the Zener effect, the efficiency of this regulator circuit is much less than the ratio of the output voltage to the rms value of the rectified supply voltage.

[0004] An improved approach uses series-connected solid-state devices, such as bipolar or field-effect transistors, to buffer the Zener voltage reference. The active devices are connected in a source-follower or emitter-follower configuration, with the load connected to the source or emitter, and the Zener reference connected to the gate or base. The Zener current can be much smaller than in the shunt configuration, and thus the total current is mostly the current supplied to the load. Thus, the efficiency of this circuit is generally no better than the ratio of input voltage to output voltage.

[0005] A further refinement of this circuit function is called a switched mode power supply. There are many such designs known in the art, but what they have in common is an input rectifier, a switching element that operates at high speed to switch a storage element (inductor or capacitor) in and out of the power supply. If input and output isolation is required, a high speed transformer can be introduced for both isolation and output voltage regulation. An RC filter is provided to reduce ripple at the output. Switch mode power supplies have the advantage of improved efficiency, since the power loss mechanisms of earlier linear systems are largely eliminated. However, when isolation is required, transformer losses can result in reduced efficiency. Fast switching is also a source of significant RF noise and introduces large losses in the conductors due to the skin effect. Theoretically high efficiencies can and have been obtained in specially designed systems. Efficiencies as high as 95% have been reported, but practical efficiencies are more typically 60%-70% in low-cost isolated systems.

[0006] A drawback of all systems known so far is that they cannot be easily integrated. Except for limited specialized applications, current AC-DC converter designs cannot be integrated on chip with other system functions. The power dissipated in the individual circuit elements is too large for the system-on-chip level of integration. Components such as the types of transformers required are not available for integration on silicon.

[0007] Ubiquitous electronic devices and subsystems typically operate at 3.3 or 5 volts. The requirement to convert 120 or 240 volt AC mains power to these lower operating voltages strains the efficiency of previously available power converters. For both linear and switched power supplies, the greater the difference between input and output voltages, the greater the inefficiency. Highly efficient, low voltage power supplies are needed to power a large number of low power, low voltage consumer devices. Electronic components are prevalent in "smart" cars and "smart" homes. Small, efficient power supplies that can support always-on sensors and networks are needed. An increasing number of homes, factories, and office buildings, both new and retrofitted, are incorporating electronic sensors to control all use of power to increase efficiency. Low voltage, integrated, highly efficient power supplies are needed to support both new and retrofitted power grids present in homes, factories, and office buildings. The power supplies must be integrable with the sensors and control electronics to allow such devices to physically fit within range of the plugs and outlets used to provide localized power. There is a need for high efficiency to avoid heat dissipation within the walls and power grids of homes, offices, and factories. There is a need for power converters with efficiencies in the range of 99-100%. There is a need for compact power converters that can fit within a wide range of devices rather than existing as a bulky box external to the device. There is a need for power converters that can be integrated. Summary of the Invention

[0008] An AC-DC power conversion system is described. The system and associated devices address the need for a highly efficient, compact, integrated, low-cost design that provides access to the low voltages used to power typical silicon-based electronic devices used in home sensors and networking, smart cars, and the like. In one embodiment, the system includes an efficient electronic switch that is used to decouple the input of a series voltage regulator circuit from the rectified AC mains power source to reduce the power dissipated in the series regulator. To optimize efficiency, the voltage across the open switch is minimized when the switch is closed and energy is stored in a shunt energy storage element. When the rectified AC mains waveform exceeds a threshold, the electronic switch is opened. While the switch is open, energy is supplied to the load by the energy storage element through the regulator circuit. In this way, the benefits of the regulator circuit are provided to the attached load circuit, while the power dissipated in the regulator circuit is significantly reduced compared to the prior art. In another embodiment, the rectifier is eliminated and the switch is synchronized with one half cycle of the AC mains waveform.

[0009] The comparator is used to control the electronic switch. In one embodiment, the comparator is comprised of an operational amplifier and a reference voltage source. In another embodiment, the comparator is comprised of a MOS field effect transistor. In one embodiment, the MOS field effect transistor is controlled via a voltage divider. In another embodiment, the voltage divider is replaced with a reference voltage source. In another embodiment, the reference voltage is adjustable.

[0010] The particular examples are not intended to limit the inventive concepts to illustrative applications.Other aspects and advantages of the present invention will become apparent from the accompanying drawings and detailed description. [Brief description of the drawings]

[0011] [Figure 1] FIG. 1 is a schematic diagram of a prior art AC-DC converter. [Diagram 2] It is a schematic diagram of a prior art AC-DC converter having current limiting. [Figure 3A] It is a schematic diagram showing the functions in an improved circuit. [Figure 3B] It is a schematic diagram showing the circuit of FIG. 3A without a full-wave rectifier [Figure 4] It is a schematic diagram showing an improved AC-DC converter. [Diagram 5] It is a schematic diagram of an embodiment of the improved circuit of FIG. 4 using a MOS field effect transistor. [Figure 6] It is a schematic diagram of a simplified version of FIG. 5. [Figure 7] It shows the drain-source voltage and drain current waveforms by a circuit simulation program of the MOSFET switch of FIG. 6. [Figure 8] It is a schematic diagram of an embodiment of a zero voltage switching circuit using a MOS field effect transistor. [Figure 9] It shows the drain-source voltage and drain current waveforms of the MOSFET switch of FIG. 8 generated by a circuit simulation program. [Figure 10] It is a schematic diagram of an alternative embodiment of a zero voltage switching circuit using a MOS field effect transistor. [Figure 11A] It shows the AC main voltage waveform and drain current waveform of the MOSFET switch of FIG. 10 during circuit startup generated by a circuit simulation program. [Figure 11B] It shows the drain-source voltage and drain current waveforms of the MOSFET switch of FIG. 10 in the steady state generated by a circuit simulation program. [Figure 12] It is a schematic diagram of a preferred embodiment of the zero voltage switching circuit of FIG. 8 including output current limiting and negative feedback stabilization of the output voltage. [Figure 13] It is a schematic diagram of a preferred embodiment of the zero voltage switching circuit of FIG. 10 including output current limiting, output voltage regulation, and negative feedback stabilization of the output voltage. [Figure 14] FIG. 14 is a schematic diagram of the zero voltage switching circuit of FIG. 13, in which the output voltage is manually adjustable. [Figure 15] FIG. 14 is a schematic diagram of the zero voltage switching circuit of FIG. 13, whose output voltage is electronically adjustable. [Figure 16] FIG. 11 is a block diagram of a third embodiment of a zero voltage switching circuit. [Figure 17] FIG. 17 is a schematic diagram of the embodiment of FIG. 16. [Figure 18] 1 shows a prior art AC-DC converter that includes isolation of the load from the power source. [Figure 19] 1 illustrates an embodiment of an AC-DC converter of the present invention that includes isolation of the load from the power supply. [Figure 20] 1 illustrates an embodiment of an AC-DC converter of the present invention that includes isolation of the load from the power supply and further includes feedback control from the load to the AC-DC converter. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0012] Figure 1 shows a schematic diagram of a prior art AC-DC converter circuit. The single-phase AC mains waveform 101 is sinusoidal and is full-wave rectified by a diode bridge 102 and the resulting time-varying DC voltage waveform is smoothed by a capacitor 103 (typically an electrolytic capacitor). Note that there is no control to charge this capacitor 103. The rectified line voltage is applied to this capacitor, so a capacitor with a large capacitance value and a rated voltage greater than the peak value of the rectified AC mains waveform is required. The smoothed voltage waveform is applied to the input of a series regulator circuit, which is connected to a bias resistor 104, a characteristic Zener voltage V Z and a pass transistor 106, which here has a characteristic threshold voltage V T The regulator output is applied to a load 107.

[0013] During operation, pass transistor 106 dynamically adjusts its drain-source voltage to regulate the load voltage to V Z -V T In other words, the pass transistor 106 keeps the Zener voltage V Z The pass transistor 106 forms a source follower circuit that buffers the load. Assuming that the Zener bias current is negligible since the entire load current passes through the pass transistor 106, the efficiency of this regulator circuit is simply the ratio of the load voltage to the rms value of the supply voltage. Thus, for a desired load voltage of nominally 3.3 V and a supply voltage of 120 V rms, the efficiency is less than 3%. Furthermore, if the load requires only a few tens of milliamps of current, the pass transistor 106 must continuously dissipate several watts of power as heat. This amount of heat dissipation typically results in an unacceptable temperature rise in small, enclosed equipment.

[0014] A further limitation of the circuit of FIG. 1 is that it does not provide protection against output current transients that may damage the pass transistor 106. Such transients may occur as a result of accidental shorting of the output terminals during operation or testing, or as a result of capacitive load impedance components. FIG. 2 shows a schematic diagram of a prior art AC-DC converter including additional components for limiting the output current and thereby protecting the pass transistor 106. In FIG. 2, a current-sensing resistor 201 having a small resistance value is placed in series with the load, and a current-limiting bipolar transistor 202 is connected between the gate of the pass transistor 106 and the load, forming a protective current-limiting circuit. Now, if the voltage drop across resistor 201 exceeds about 0.7V (for a silicon bipolar transistor), transistor 202 begins to conduct, reducing the gate-source bias on the pass transistor 106, thereby reducing the output current. However, the efficiency of this improved circuit is essentially unchanged compared to the efficiency of the circuit shown in FIG. 1.

[0015] To improve the efficiency of these prior art series regulator circuits, the power dissipated in the pass transistor must be significantly reduced. In one embodiment of the present invention, the pass transistor is disconnected from the rectified supply voltage when it is not needed. FIG. 3A shows a schematic diagram of an improved rectifier circuit including an AC mains supply 101, a diode bridge 102, and a filter capacitor 103, but with additional circuitry inserted between the diode bridge 102 output and the filter capacitor 103. The waveform at the diode bridge 102 output is a simple full-wave rectified sinusoidal waveform that conventionally ranges from 0V to a peak of about 170V when a conventional AC mains supply has an rms value of 120V. However, it should be noted that the method described below applies to any periodic power waveform, provided that the numerical specifications of the affected components are appropriately adjusted. Additionally, the power waveform may contain a DC offset if it is less than a reference voltage described below.

[0016] The additional circuitry includes a comparator circuit 302 having an inverting input connected to the diode bridge 102 output and a voltage reference 301 connected to the non-inverting input, the comparator 302 being configured to detect when the diode bridge output voltage is equal to a reference voltage V R If the reference voltage V exceeds the reference voltage V, the series switch 303 is controlled to disconnect the diode bridge output from the subsequent circuit (open the switch 303). RWhen V exceeds the diode bridge output voltage, switch 303 is closed and capacitor 103 is charged through series diode 304. Diode 304 prevents capacitor 103 from re-discharging through switch 303 when the diode bridge output voltage drops. The combination of diode 304 and capacitor 103 forms a "peak detector" circuit which stores energy each half of the AC mains cycle for supply to the subsequent regulator circuitry and load 305. Unlike other prior art examples, the voltage across capacitor 103 need only be large enough to meet the energy requirements of the subsequent regulator circuitry and load 305. The input voltage to the series regulator is significantly reduced compared to the effective value of the AC mains. The operation of the "peak detector" circuit is determined by the fact that when the AC mains voltage is greater than V R As long as the AC mains supply peak voltage remains greater than V, the peak voltage stored on capacitor 103 will always be V, regardless of fluctuations in the AC mains supply peak voltage. R This embodiment of the switching circuit operates as a voltage regulator circuit itself. Because the operation of switch 303 uses very little energy, the overall efficiency of the improved AC-DC converter circuit shown in FIG. 3A is much greater than that found in the prior art circuits of FIGS. 1 and 2. An additional advantage is a significant reduction in operating temperature rise. Although comparator 302 is a well-known analog circuit element, other analog or digital circuits can be used to achieve the desired thresholding functions necessary to operate switch 303.

[0017] In one embodiment, the reference voltage VR is fixed. In another embodiment, the reference voltage can be changed. In another embodiment, the reference voltage is selectable. In one embodiment, the circuit of FIG. 3A is connected to a load and the regulator function of the circuit is used to control the voltage supplied to the load. In another embodiment, an additional regulator is used in series with the circuit of FIG. 3A and the load.

[0018] As mentioned above, the operation of this circuit does not depend on the availability of a full-wave rectified AC mains waveform. In fact, assuming that the specifications of elements 301-305 are sufficient to handle the negative variations in the AC mains waveform, the diode bridge 102 can be eliminated and the switching and regulating components can be connected directly to the AC mains, resulting in the embodiment shown in FIG. 3B. Note that the peak detector diode 304 also blocks reverse current through the load during the negative half cycle of the AC mains when switch 303 is closed. The only drawback of this embodiment over that shown in FIG. 3A is that the total maximum power available to the load is halved.

[0019] FIG. 4 shows a schematic diagram of an improved rectifier circuit interconnected with series regulators 103, 104, 105, 106 and load 107, and provides a useful basis for establishing relationships between design variables in the new rectifier circuit described below. The voltage to the comparators is via a voltage divider network of resistors 401, 402. Comparators 301-304 have been previously described. To maintain regulation of the output voltage, the voltage across capacitor 103 is divided by the Zener 105 voltage V Z However, capacitor 103 discharges linearly in time over a half cycle of the AC mains supply due to the nominally constant current supplied to load 107. Thus, capacitor 103 initially discharges at the highest voltage V peak =V Z +I load *t MAINS / (2*C103) must be charged. MAINS is the period of the AC mains waveform. This means that V peak and V Z gives the value of capacitor 103 as a function of the difference between peak results in higher power dissipation in pass transistor 106, which is traded off against the maximum practical value of capacitor 103. The efficiency of a regulator is the ratio of the power delivered to the load divided by the total power dissipated in the circuit, and is given by 2*(V Z -V T ) / (VZ +V peak ) is given by V peak The minimum value of V Z is V, so the best efficiency is 1 - V T / V Z .

[0020] Figure 5 shows a schematic diagram of an improved rectifier circuit. Switch 303 is implemented using an enhancement-mode MOSFET 505, and the comparator circuit is realized as a single common-source amplifier stage using an enhancement-mode MOSFET 504 characterized by a threshold voltage V T1 and a load resistor 503. Thus, when the output of the voltage-divider network comprising resistors 501 and 502 exceeds the threshold voltage V T1 of MOSFET 504, the gate of MOSFET switch 505 is pulled up to ground, thereby opening switch 505. When the output of the voltage-divider network 501, 502 is less than V T1 , the gate of MOSFET 505 is connected to its drain, thereby closing the switch. However, MOSFET 505 is not an ideal switch and can experience significant power losses while in its conducting state. As a result, the efficiency of the circuit realized using MOSFETs is not as high as that obtained in the ideal case shown in FIG. 4. As the drain and gate voltages of MOSFET 505 increase during the positive half-cycle of source 101, when the gate voltage exceeds the threshold voltage V T2 of MOSFET 505 and the source voltage of MOSFET 505 rises within the threshold voltage V T2 of the gate voltage, an efficiency problem occurs due to the fact that the drain current Id(t) flows from the drain to the source. The drain current stops when MOSFET 505 is turned off by MOSFET 504. Thus, the drain-source voltage of MOSFET 505 during the charging current transient is also approximately its threshold voltage V T2, , which is typically about 4 - 6 volts for a power FET. As a result, the instantaneous power consumed within MOSFET 505 during the charging current transient is simply Id(t)*V T2which is significant compared to the power delivered to the load by the same current transient.

[0021] Additionally, as a result of its fabrication process, a power MOSFET typically includes a parasitic source-drain diode 506 associated with MOSFET 505 and a parasitic source-drain diode 507 associated with MOSFET 504. Note that diode 506 would allow capacitor 103 to discharge when MOSFET 505 is "off," but series track-and-hold diode 304 blocks this spurious discharge path. The presence of parasitic diodes 506 and 507 is assumed in subsequent figures. In one embodiment, all of the components of FIG. 5 are fabricated on a single semiconductor chip. In another embodiment, all of the components of FIG. 5, except for capacitor 103, are fabricated on a single semiconductor chip.

[0022] 6 is a schematic diagram of a simplified version of the improved rectifier circuit of FIG. 5 for circuit simulation to provide a basis for comparing the efficiency of the high efficiency circuit of the preferred embodiment described below. For simplicity, the voltage regulation circuitry has been omitted and the load is represented by resistor 107.

[0023] The results of a SPICE simulation of the circuit of Figure 6, assuming an AC mains frequency of 60Hz, are shown in Figure 7. Figure 7 shows the waveforms of the drain-source voltage 701, Vds, and drain current 702, Id, of the switch MOSFET 505 over one cycle of the AC mains 101. T2 Note that the step portion 703 having a magnitude of Id*Vds coincides with the drain current transient on the Vds waveform 701. Efficiency is the ratio of the average power delivered to the load 107: (Id*Vds) load ) divided by the average power supplied by the AC mains to the switching circuit (Id*VAC), which is 68%. 30% of the power supplied by the AC mains is dissipated in the switch MOSFET 505.

[0024] This V T2 To overcome the "overhead", the high efficiency circuit of Figure 8 includes a current limiting resistor 801 and a Zener diode 802 shunted by a capacitor 803 connected between the gate and source of MOSFET 505. During the negative half cycle of the source 101, current flows from ground, through the body diode 507 of MOSFET 504, through added components 801-803, and back to 101 through the body diode 506 of MOSFET 505, resulting in a Zener voltage V of diode 802. Z is stored on capacitor 803. Note that this stored potential applies a positive bias on the gate of 505 with respect to its source. This is V Z is equal to V T2 Therefore, when MOSFET 505 turns back on at the start of the positive half-cycle of 101, its drain-source voltage is limited only by the intrinsic channel resistance rds of MOSFET 505, V T2 , which is typically much less than 1 Ω. The results of a SPICE simulation of this "zero voltage" switching circuit are shown in Figure 9. Again, waveforms of the drain-source voltage 901, Vds, and drain current 902, Id, of the MOSFET 505 over one cycle of the AC mains are shown. Note that the Vds waveform is near zero during the drain current transient 903. The efficiency of this circuit is 89%, with less than 2% of the power provided by the AC mains being dissipated in the MOSFET 505.

[0025] The precharging of capacitor 803 to the Zener voltage of diode 802 in the negative half cycle of the AC mains provides the gate bias for MOSFET 505 so that resistor 503 no longer needs to provide it. Another embodiment of the zero voltage switching circuit shown in FIG. 10 eliminates the load resistor (item 503 in FIG. 8) in the drain circuit of MOSFET 504, which also provides a parasitic discharge path for capacitor 803 through MOSFET 505, limiting the duration of drain current transients in the circuit of FIG. 8. Furthermore, bias resistor 501 is disconnected from AC mains 101 and connected directly to the DC output node at the junction of diode 304, load resistor 107, and capacitor 103. Thus, MOSFET 504 conducts and turns off switch MOSFET 505 when the DC output node voltage reaches the threshold established by resistors 501 and 502 and the threshold voltage of MOSFET 504. The results of a SPICE simulation of this circuit are shown in FIG. 11A and FIG. 11B. FIG. 11A shows three initial cycles 1100 of the AC mains 101, Vac, and the drain current 1102, Id of the MOSFET 505 at circuit start-up, showing that the drain current transient 1102 reaches steady state after the first negative half cycle of the AC mains 1100. FIG. 11B again shows the waveforms of the drain-source voltage 1101, Vds, and the drain current 1102, Id of the MOSFET 505 over one cycle of the AC mains at steady state. Due to the larger drain current transient duration 1103, the total power delivered to the load increases by 25%. The efficiency of this zero voltage switching circuit increases to 90%, and less than 1% of the power supplied by the AC mains is consumed in the MOSFET 505.

[0026] FIG. 10 shows a first embodiment 1000 of the zero voltage switching circuit of FIG. 8, including a bias resistor 501 connected between a DC output node 1005 and a gate 1001 of a MOSFET 504 .

[0027] The first embodiment is an AC-DC conversion system (1000) for supplying direct current (DC) energy from an alternating current (AC) power source (101) to an electronic load (107) at an output node (1005), comprising: a. a voltage divider (501, 502) connected across a load (107); b. a first switch (504) having an input (1001) and an output (1002), the first switch (504) being connected to a voltage divider via its input (1001); c. a second switch (505) having an input (1003) and an output (1004), the input (1003) being connected to the output (1002) of the first switch (504); d. a storage capacitor (103) connected via a diode (304) to the output (1004) of the second switch (505); a Zener diode (802) having a Zener voltage ef, connected between the input (1003) and the output (1004) of the second switch (505), thereby clamping the voltage between the input (1003) and the output (1004) of the second switch (505) to the Zener voltage of the Zener diode (802); g. An electronic load (107) connected across the storage capacitor (103).

[0028] In a first embodiment, as shown in Figure 10, the switches 504, 505 are N-MOSFET transistors. In another embodiment, functionally equivalent to Figure 10 but not shown, the switches 504, 505 are bipolar transistors.

[0029] 12, the AC-DC converter includes an overcurrent protection circuit disposed between the output 1004 of the MOSFET switch 505 and the input of the diode 304. The protection circuit consists of a very small value series resistor 201 and a bipolar transistor 202 with a base terminal connected to the switch output 1004, an emitter terminal connected to the input of the diode 304 and a collector connected to the input 1003 of the MOSFET switch 505.

[0030] In a further embodiment shown in FIG. 13, the AC-DC converter includes a regulation circuit having elements 104 , 105 , 106 arranged between the front output terminal 1005 and the load 107 at a new DC output terminal 1301 .

[0031] 14 is a schematic diagram of an embodiment of a zero voltage switching circuit using a MOSFET in which the output voltage at output terminal 1005 is manually adjustable. Resistor 502 of FIG. 12 is replaced with a potentiometer 1401 that can be manually adjusted to change the voltage waveform applied to gate 1001 of MOSFET 504 and thereby change the voltage stored on capacitor 103.

[0032] 15 is a schematic diagram of an embodiment of a zero voltage switching circuit using MOSFETs in which the output voltage of an AC-DC converter at output 1005 can be electronically adjusted. An additional control MOSFET 1501 is connected in place of resistor 502 in FIG. 12 and is connected to an external DC control voltage V C is applied to the gate of MOSFET 1501 , thereby changing the voltage applied to gate 1001 of MOSFET 504 , which changes the voltage stored on capacitor 103 .

[0033] In another embodiment, AC-DC converter 1600 is generally comprised of the elements shown in FIG. 16 and the methodology illustrated by these elements. A non-limiting example of the circuit elements is shown in FIG. 17. Referring to FIG. 16, AC power source 1601 is connected to inrush protection element 1602. In one embodiment, the inrush element is comprised of a resistive element on the line and neutral of the AC power source. In another embodiment where higher power and efficiency are required, the inrush protection element includes a switch element that provides a high resistance at start-up and switches the resistive element out of the circuit during steady state operation. After inrush protection, a sampling element 1603 is used to create a scaled copy of the AC source 1601 waveform for control purposes. In one embodiment, sampling element 1603 includes resistors configured into a voltage divider network. One embodiment is the voltage divider shown and described in FIG. 5. In another embodiment, the sampling element includes a reference voltage source and a comparator as shown in FIG. 4. In another embodiment, sampling element 1603 can be manually adjusted as shown in FIG. 14. In another embodiment, the sampling element can be electronically adjusted as shown in FIG. 15. The sampled voltage is used as an input to the switch driver element 1604. In a preferred embodiment, the switch driver element 1604 receives a feedback voltage signal 1609 from the storage element 1606 and controls the voltage applied to the gate of a switching element in the control switch and clamp element 1605 based on the voltage signal, thereby opening and closing the control switch 1605 to provide power to the storage element 1606 and ultimately the load 1608. In an embodiment in which the feedback 1609 is eliminated, the AC-DC converter is a feedforward converter and the charging of the storage element 1606 is controlled from the forward sides 1603, 1604, and 1605. The addition of the feedback control 1609 provides a means for both feedforward and feedback control. In an embodiment, the balance of feedforward and feedback control is determined by the selection of components in the voltage sampling element 1603 and the feedback line 1609.In one embodiment, the balance of the feedforward and feedback control is determined by resistive elements in the sampling element 1603 and the feedback 1609. In another embodiment, variable elements are used to allow the feedforward and feedback control to be adjusted. In a preferred embodiment, the switch driver 1604 is comprised of a voltage divider and a switch. The switch, current limit and clamp element 1605 controlled by the switch driver 1604 provides pulsed power at a fixed maximum current to the storage element 1606. In a preferred embodiment, the switch, current limit and clamp element 1605 comprises an N-MOSFET, a current sensing resistor and a bipolar peak current limit transistor, as well as a source-gate connected Zener diode that clamps the peak gate-source voltage at the negative half cycle of the AC source 1601, thereby providing the zero voltage switching characteristic of the circuit. Power from the switch and clamp element consisting of a preselected peak current pulse is provided to the storage element 1606. The voltage on the storage element 1606, which in one embodiment consists of a capacitor used as an energy storage element and a diode, is fed back 1609 to the switch driver 1604 via a voltage divider circuit, thereby maintaining a constant charge on the capacitor. The output from the storage element is provided to a load 1608 via a voltage regulator 1607. In another embodiment, the AC-DC converter further includes a galvanic isolation element 1610. In another embodiment, the AC-DC converter further includes an element 1611 that enables feedback from the load 1608. In a preferred embodiment, the feedback circuit 1611 also includes galvanic isolation between the control element 1604 and the load 1608.

[0034] FIG. 17 shows a preferred embodiment of a zero voltage switching AC-DC converter. The individual components of the circuit function in the same way as those of the circuits already described in FIGS. 5-15. Elements 1701-1708 correspond to elements 1601-1608 of FIG. 16, respectively. An AC power source 1701 is connected to an inrush protection circuit 1702, which in this preferred embodiment includes resistors R1 and R2. In another embodiment (not shown), the inrush protection element comprises a switch, which causes current to flow through resistors R1 and R2 at start-up and bypass the resistors once steady state operation is reached. In another embodiment, the inrush control uses an inductor; i.e., elements R1 and R2 are replaced by inductors L1 and L2. The output from the inrush protection goes to switch Q2 of a switch, current limit and clamp circuit 1705 and to a voltage sampling element 1703. The voltage sampling element 1703 consists of resistors R3, R4, which sample the voltage from the storage capacitor C1. The values ​​of R3, R4 are selected such that the voltage to the gate of switch Q1 in switch driver element 1704 turns switch Q1 on and off, which in turn turns switch Q2 off and on in a synchronous manner, thereby providing a preselected timing output pulse from switch Q2 to charge storage element C1. Resistor R4 provides a feedback path for the charge on capacitor C1, and thus an output voltage to voltage sampling circuit 1703, and thus to control circuit 1704. Switch, current limit and clamp element 1705 is comprised of switch Q2, current sense resistor R10 and bipolar transistor Q4, Zener diode D1, capacitor C3 and resistor R7. Switch Q2 is controlled by switch driver circuit 1704. The peak output current of switch Q2 is limited to a preselected maximum value based on the selected value of current sense resistor R10. Capacitor C3 charges to the Zener voltage of diode D1 during the negative half-cycle of AC source 1701 and provides a gate-source bias for Q2 that minimizes the drain-source voltage of Q2 during the charging current transient.This pulse output from switch Q2 is connected to a voltage regulator 1706, which maintains a constant charge on storage capacitor C1 through voltage sampling 1703 and feedback of R4 to switch driver 1704. The control element switch Q1, and therefore the supply switch Q2, are operated either open or closed in synchronism with the AC input 1701. The AC-DC converter provides a low voltage output with pulse modulation at the frequency of the incoming AC source. The switches are operated either open or closed at a voltage near the zero crossing of the AC source, within the thresholds of Q1 and Q2. The output then goes to voltage regulator 1707 and then to load 1708. Voltage regulator 1707 includes switch Q3, Zener diode D3, resistor R9, and capacitor C2. The circuit components D3, Q3, and R9 function as a voltage regulator equivalent to those already described for circuit elements 105, 104, and 106 of FIG. 1, respectively. Capacitor C2 provides storage capacitance to buffer, and therefore smooth, the output from the AC-DC converter to the load 1708.

[0035] The AC-DC converter in the preferred embodiment of Figures 16 and 17 is composed of the following elements: Inrush Protection 1602, Voltage Sampling 1603, Switch Driver 1604, Switch and Clamp 1605, Storage Element 1606, and Voltage Regulator 1607. The selection of components in Voltage Sampling 1603 determines the timing of Switch Driver 1604. The selection of elements in the Switch and Clamp determines the peak voltage and current of the output pulse. The power output is controlled by selecting both the peak current and the pulse timing. Feedback from the storage element via the voltage sampling is used to select the pulse timing. The AC-DC converter operates synchronously with the AC source.

[0036] The preferred embodiment of Figures 16 and 17 generally includes a voltage divider 1603 connected to a power supply 1601, a first switch 1604 connected via an input to the voltage divider, a second switch 1605 having an input connected to the output of the first switch, a capacitor C1 connected via a diode to the output of the second switch, and a sense resistor 1609 connected between the capacitor and the voltage divider, thereby providing feedback control of the zero voltage switching AC-DC extraction conversion system, and includes a Zener diode D1 connected between the input and output of the second switch, and an electronic load 1608 connected to the capacitor C1. The switches 1604, 1605 may be any electronically actuated switch. In one embodiment, the switches are N-MOSFETs. In another embodiment, the switches are bipolar transistors, and in another embodiment, the switches are micro-electromechanical switches.

[0037] FIG. 18 shows a prior art AC-DC converter system that includes galvanic isolation of the AC source 1808 from the load. A typical prior art AC-DC converter has a full wave rectifier 1802 providing a grounded (1809) DC source that is filtered (1803), a controller 1804, which is typically a pulse controller that uses a switch 1805 to control the output through a transformer 1806, thereby providing a DC voltage to the load 1808. A diode 1807 prevents the flow of current back through the transformer 1806 from the load. Typically, the transformer also acts as a step-down transformer to control the voltage required by the load 1808. Note that the high side of the transformer 1806 operates on the rectified voltage of the AC source 1801. The transformer provides the galvanic isolation, but the high voltage connected to the transformer thereby requires a transformer that can operate at this high voltage.

[0038] In contrast, a power supply of the present invention is shown in a first embodiment in FIG. 19. An AC power supply 1901 is connected to a load 1905 through an isolation device 1904 and further through an AC-DC converter 1902. The ground 1906 on the AC-DC converter 1902 is not necessarily at the same level as the ground 1907 on the load 1905. In a preferred embodiment, the AC-DC converter 1902 is as described in FIG. 12 and FIG. 13. The AC-DC converter 1902 includes a clamp of the output voltage, which is fed to a storage capacitor (C1 in FIG. 13) so that the isolation device 1904 obtains a clamped voltage at a maximum. In the preferred embodiment and in comparison in FIG. 16 and FIG. 19, the isolation device is placed between the blocking element 1607 and the blocking element 1608.

[0039] Another embodiment shown in Figure 20 further includes feedback from the load 1905 to the AC-DC converter 1902. Feedback is provided to the AC-DC converter through sense lines 2002 which pass through an isolator 2001 and through an isolated sense line 2003. In one embodiment, one of the sense lines 2003 is grounded and the other feeds the voltage sampling circuit 1303, similar to how the sense line shown in Figure 17 feeds from a capacitor C1 through a resistor R8. In one embodiment, the isolator 2001 is an optical isolator as shown. In another embodiment (not shown), instead of an optical isolator, a transformer is used as the isolator.

[0040] Figures 16, 17, 19, and 20 show an AC-DC converter that can be fully integrated on silicon. Not all of the components shown in the figures are required for a fully functional device. In one embodiment, the AC-DC converter consists of a voltage divider (1703) connected to an AC power source 1701 for sampling and further connected to the base of a first switch transistor Q1. The value of the resistors in the voltage divider controls the voltage seen by Q1, thereby providing forward control of Q1 and thus the output of the AC-DC converter. The drain of Q1 is then connected to the base of a second switch transistor Q2 that supplies a pulse current to a storage element C1. Diode D2 prevents discharge of capacitor C1 through switch Q2. A sense line is connected from storage element C1 to voltage divider 1703 through resistor R8, providing feedback control to prevent complete discharge of storage capacitor C1. A Zener diode D1 connected between the gate and source of Q2 clamps the gate-source voltage seen by Q2 to the Zener voltage of diode D1, and that bias voltage is stored in capacitor C3. An inductor or a low-resistance winding resistor R1 in series with the AC line filters the transient current seen by Q2 and limits the current. Thus, a fully functional AC-DC converter consists of a voltage divider, two switches, a storage device, a Zener diode, a normal diode, and an inductor, where the switches are N-MOSFETs and the storage device is a capacitor. In another embodiment, the AC-DC converter further includes inrush control 1702. In one embodiment, the inrush control consists of resistors connected in series to the line and neutral of the AC power source. In another embodiment, the AC-DC converter further includes a voltage regulator. In one embodiment, the voltage regulator consists of a switch Q3 connected to the output line from storage element C1. The switch is controlled through a Zener diode D3 connected to the base from capacitor C2. The output of the voltage regulator is connected to a load 1708. In another embodiment, the AC-DC converter further includes galvanic insulation, which is an isolation transformer connected to the output of the voltage regulator.In another embodiment, there is no voltage regulator and an isolation transformer is connected between the storage capacitor C2 and the load 1708. Another embodiment further includes feedback from the load to the voltage divider 1703. The feedback from the load is provided to the voltage divider 1703 through a second isolation device 2001.

[0041] <Summary> This application describes an AC-DC conversion system. The conversion system consists of an electronic switch and control circuit used to provide controlled pulsed power to a storage element, which provides power to a load at either a preselected voltage or a manually or automatically selectable voltage, while ensuring that the voltage drop across the switch is minimized, reducing the power dissipated through the switch itself, thereby greatly increasing efficiency and reducing heat losses. The AC-DC converter in one minimal version consists of a pair of N-MOSFET transistors, a voltage divider, a storage element, and a pair of diodes. This design allows for high efficiency with a minimum of components that can be fully integrated on silicon.

Claims

1. An AC-DC conversion system that supplies energy from an alternating current (AC) power source (101) as direct current (DC) to an electronic load (107) at an output node (1005), comprising: a. a first N-MOSFET transistor (504) having an input (1001) and an output (1002), the first N-MOSFET transistor (504) being connected to a voltage divider via said input (1001); b. a second N-MOSFET transistor (505) having an input (1003) and an output (1004), said input (1003) being connected to the output (1002) of the first N-MOSFET transistor (504) through a current limiting resistor (801); c. a storage capacitor (103) connected through a diode (304) to the output (1004) of the second N-MOSFET transistor (505); d. a voltage divider (501, 502) connected across said storage capacitor (103) and thereby across said load (107); e. a Zener diode (802) having a Zener voltage, connected between the input (1003) and the output (1004) of the second N-MOSFET transistor (505), and having a shunt capacitor (803) connected in parallel with the Zener diode (802), thereby clamping the voltage between the input (1003) and the output (1004) of the second N-MOSFET transistor (505) to the Zener voltage of the Zener diode (802); f. an electronic load (107) connected to the storage capacitor (103); An AC-DC conversion system comprising:

2. The AC-DC conversion system further comprises a series voltage regulator circuit disposed between the storage capacitor (103) and the electronic load (107); The series voltage regulator circuit provides a characteristic threshold voltage (V Ｔ ), a bias resistor (104) connected across said pass transistor, and an output voltage to said load being V Ｚ -V Ｔ a Zener diode having a Zener voltage (Vz) connected to the bias resistor so as to be maintained at 2. The AC-DC conversion system according to claim 1.

3. The AC-DC conversion system further comprises a current limiting circuit (201, 202) disposed between an output of the second N-MOSFET transistor and the storage capacitor, the current limiting circuit limiting a current through the second N-MOSFET transistor; the current limiting circuit includes a sense resistor (201) connected to the output of the second N-MOSFET transistor and to the load (107), and a bipolar transistor (202) connected between the load and the input of the second N-MOSFET transistor; 2. The AC-DC conversion system according to claim 1.

4. 2. The AC-DC conversion system of claim 1, wherein all semiconductor devices are fabricated on a single integrated circuit chip.

5. 2. The AC-DC conversion system of claim 1, wherein the voltage divider further comprises a potentiometer (1401) configured to allow an input voltage to the first N-MOSFET transistor to be manually adjusted.

6. the voltage divider further comprises a control MOSFET (1501) connected in place of the resistors of the voltage divider and an external DC control voltage applied to the input of the control MOSFET (1501) to vary the voltage applied to the input of the first N-MOSFET transistor and thus vary the voltage stored on the storage capacitor; 2. The AC-DC conversion system according to claim 1.

7. 7. The AC-DC conversion system of claim 6, wherein all semiconductor devices are fabricated on a single integrated circuit chip.

8. 2. The AC-DC conversion system of claim 1, further comprising an isolation transformer between said storage capacitor and said electronic load.

9. 2. The AC-DC conversion system of claim 1, further comprising: a sense line from the load through an isolator to the voltage divider, thereby providing feedback control from the load.

10. 4. The AC-DC conversion system of claim 3, further comprising an isolation transformer between the storage capacitor and the electronic load.

11. 4. The AC-DC conversion system of claim 3, further comprising a sense line from the load through an isolator to the voltage divider, thereby providing feedback control from the load.

12. An AC-DC conversion system that supplies energy from an alternating current (AC) power source (101) as direct current (DC) to an electronic load (107) at an output node (1005), comprising: a. a first N-MOSFET transistor (504) having an input (1001) and an output (1002), the first N-MOSFET transistor (504) being connected to a voltage divider via said input (1001); b. a second N-MOSFET transistor (505) having an input (1003) and an output (1004), said input (1003) being connected to the output (1002) of the first N-MOSFET transistor (504) through a current limiting resistor (801); c. a storage capacitor (103) connected through a diode (304) to the output (1004) of the second N-MOSFET transistor (505); d. a voltage divider (501, 502) connected across said storage capacitor (103) and thereby across said load (107); e. a Zener diode (802) having a Zener voltage, connected between the input (1003) and the output (1004) of the second N-MOSFET transistor (505), and having a shunt capacitor (803) connected in parallel with the Zener diode (802), thereby clamping the voltage between the input (1003) and the output (1004) of the second N-MOSFET transistor (505) to the Zener voltage of the Zener diode (802); f. an electronic load (107) connected to the storage capacitor (103); An AC-DC conversion system comprising:

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