Receiving device

Abstract

To provide a semiconductor receiving device having excellent power consumption characteristic and noise resistance.SOLUTION: A receiving device 400A includes an input part 401, a first high-pass filter 410, a gate-grounded amplifier 450, a second high-pass filter 430, and a demodulator 440. The input part 401 removes low-frequency noise in received digital modulation signal and reduces influence of parasitic element in an isolation element 300. The high-pass filter 410 removes noise in a signal outputted from the input part 401 and operates so as to determine DC operation point of the gate-grounded amplifier 450. The gate-grounded amplifier 450 also serves as a band-pass filter. The second high-pass filter 430 removes low-frequency noise outputted from the gate-grounded amplifier 450. The demodulator 440 demodulates the digital modulation signal.SELECTED DRAWING: Figure 2

Description

Technical Field

[0001] The present invention relates to a receiving device using an isolator.

Background Art

[0002] Generally, a communication system used in high-voltage equipment is often composed of a semiconductor device with a multi-chip configuration that transmits signals in an electrically insulated state using an insulating element such as a capacitor or a transformer between the input side and the output side. This type of communication system is used, for example, in high-voltage equipment with a lot of noise.

[0003] Here, an example of the operation of a communication system using an isolator will be introduced. The transmitter digitally modulates an input signal and outputs it to a communication path through an insulating element. The receiving device inputs a digitally modulated signal through the communication path. The digitally modulated signal is attenuated by the influence of the insulating element and the parasitic components of the communication path, but the receiving device amplifies the digitally modulated signal and transmits it to a filter connected in cascade at the subsequent stage.

[0004] When the filter removes unnecessary components such as noise components and transmits them to the demodulator at the final stage, the demodulator demodulates this signal. Various proposals have been made for receiving devices used in such communication systems. As an example, Patent Documents 1 to 3 can be cited.

[0005] According to the receiving circuit of Patent Document 1, in addition to a high-pass filter composed of passive elements, a circuit that biases using a resistor is provided. Although a band-pass filter is mentioned, its configuration is not disclosed. Furthermore, if the filter is composed only of passive elements, the signal will be attenuated, but the problem of attenuation has not been recognized, and the solution to the problem of signal attenuation has not been solved.

[0006] Patent Document 2 discloses a circuit in which an RC filter is configured in the input section of a receiver. According to Patent Document 2, noise of a received signal is removed using a high-pass filter composed of passive elements, and a reference potential is supplied using a divided voltage of a resistor. Patent Document 2 is a technology presented mainly for the purpose of countermeasures against common-mode transient noise, but details of the receiving device connected to the subsequent stage are not disclosed, and it is not clear to what kind of receiving device it can be applied.

[0007] According to the configuration of the receiving circuit of Patent Document 3, after a high-pass filter composed of an operational amplifier circuit removes noise of a received signal, demodulation is performed using a Schmitt trigger circuit. However, according to the technology described in Patent Document 3, since it is directly connected from an insulating element to an amplifier, there is a risk of signal attenuation.

[0008] As a configuration common to the technologies described in Patent Documents 1 to 3 above, there is a high-pass filter or a band-pass filter at the connection part with the insulating element. Thereby, an attempt is made to ensure noise resistance against external disturbance noise. However, simply installing a filter also attenuates the frequency components of the necessary signals, deteriorating the reception characteristics.

Prior Art Documents

Patent Documents

[0009]

Patent Document 1

Patent Document 2

Patent Document 3

Summary of the Invention

Problems to be Solved by the Invention

[0010] The present disclosure has been made in view of the above circumstances, and an object thereof is to provide a receiving device with excellent noise resistance.

Means for Solving the Problems

[0011] The invention according to claim 1 is directed to a receiving device that receives a signal from a transmitter through an insulating element. The receiving device includes an input unit, a first high-pass filter, a gate-grounded amplifier circuit, a second high-pass filter, and a demodulator. The input unit inputs a pair of signals through the insulating element and includes an impedance conversion unit coupled to the insulating element. The first high-pass filter is coupled to the input unit, and the gate-grounded amplifier circuit is coupled to the first high-pass filter. The second high-pass filter is configured in the subsequent stage of the gate-grounded amplifier circuit.

[0012] The demodulator is configured in the subsequent stage of the second high-pass filter. Since it is not directly connected from the insulating element to the gate-grounded amplifier circuit, the signal including common-mode noise can be filtered in the first high-pass filter without being affected by the frequency amplification characteristics of the gate-grounded amplifier circuit, and can be processed with minimal influence on the frequency characteristics of the gate-grounded amplifier circuit. Thereby, a configuration excellent in noise resistance against common-mode noise can be achieved. According to the invention described in claim 1, the second high-pass filter is configured by commonly connecting a capacitor and a resistor, and a common connection node of the capacitor and the resistor is connected to a subsequent circuit connected to the subsequent stage of the second high-pass filter, and a bias voltage of the subsequent circuit is generated from an intermediate node of a voltage divider constituted by a plurality of voltage dividing resistors through the resistor. According to the invention described in claim 2, the second high-pass filter is configured by connecting a capacitor and a resistor, and includes a reference voltage source that outputs a reference voltage, and the reference voltage source outputs the reference voltage as a bias voltage to the subsequent circuit through the resistor of the second high-pass filter.

Brief Description of the Drawings

[0013]

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Modes for Carrying Out the Invention

[0014] Hereinafter, several embodiments of the receiver will be described with reference to the drawings. In each of the embodiments described below, components that perform the same or similar operations are denoted by the same or similar reference numerals, and descriptions thereof may be omitted as necessary.

[0015] (First Embodiment) The first embodiment will be described with reference to FIGS. 1 to 5. First, the basic configuration will be described with reference to FIGS. 1 and 2. The communication system 100 shown in FIG. 1 is composed of an input terminal 110, a transmitter 200, an insulating element 300, a receiver 400, and an output terminal 120.

[0016] The transmitter 200 is connected to the input terminal 110. When the transmitter 200 receives a signal from the input terminal 110, it digitally modulates the signal and transmits the digitally modulated signal to the receiver 400 through the insulating element 300. The receiver 400 receives the digitally modulated signal through the insulating element 300.

[0017] FIG. 2 shows a specific example of the differential configuration. As shown in FIG. 2, the receiver 400 includes an input unit 401 that receives a signal through the insulating element 300, a first high-pass filter 410 coupled to the input unit 401, a gate-grounded amplifier circuit 450 coupled to the first high-pass filter 410, a second high-pass filter 430 configured at the subsequent stage of the gate-grounded amplifier circuit 450, and a demodulator 440 configured at the subsequent stage of the second high-pass filter 430.

[0018] In FIG. 2, the configurations corresponding to the differential circuits are labeled with subscripts A and B for each identification code. Hereinafter, the subscripts A and B will be omitted as necessary for explanation. The insulating element 300 is composed of a pair of capacitors 300A and 300B. The receiver 400A receives the transmission signal of the transmitter 200 through the insulating element 300.

[0019] The input unit 401 includes low-impedance resistance elements 402A and 402B connected between the output node of the insulating element 300 and the ground node, and is configured as an impedance conversion unit that converts the input / output impedance. The resistance elements 402A and 402B are configured to remove noise components included in the received signal, and act as a high-pass filter together with the insulating element 300.

[0020] The first high-pass filter 410 is configured by connecting a capacitor 411A, 411B, and resistor elements 412A, 412B as shown in the figure, and is used to remove noise included in the received signal. A gate-grounded amplifier circuit 450 is connected to the common connection point of the capacitor 411A, 411B and the resistor elements 412A, 412B.

[0021] The gate-grounded amplifier circuit 450 includes a pair of bias current sources 453A, 453B and an output transistor pair 451A, 451B to which the current of the bias current sources 453A, 453B is supplied. The sources of the output transistor pair 451A, 451B are configured as a differential input section that differentially inputs the output of the first high-pass filter 410. The output transistor pair 451A, 451B is each composed of an N-channel MOSFET. Resistors 452A, 452B are respectively connected between the gates and drains of the output transistor pair 451A, 451B, and gate biases are respectively applied from the bias current sources 453A, 453B through the resistors 452A, 452B.

[0022] In the gate-grounded amplifier circuit 450, the sources of the output transistor pair 451A, 451B serving as the differential input section and their gates are capacitively cross-coupled by capacitor pairs 454A, 454B. Thereby, the gate-grounded amplifier circuit 450 is configured to have band-pass filter characteristics.

[0023] The drains of the output transistor pair 451A, 451B are the output nodes of the gate-grounded amplifier circuit 450 and are connected to the second high-pass filter 430. The second high-pass filter 430 is configured by connecting capacitors 431A, 431B and resistors 435A, 435B, and the common connection node of the capacitors 431A, 431B and the resistors 435A, 435B is connected to a demodulator 440 which is a subsequent circuit connected to the subsequent stage of the second high-pass filter 430.

[0024] The voltage divider is configured by connecting voltage dividing resistors 432 and 433 in series between the power supply node and the ground node, respectively, and outputs a bias voltage from the intermediate node of the voltage dividing resistors 432 and 433 to the input transistor pair 441A and 441B of the demodulator 440 through resistors 435A and 435B. Thereby, it is configured to generate the bias voltage of the input transistor pair 441A and 441B of the demodulator 440 described later. The demodulator 440 is configured by a shaping process of the input signal and further an envelope detector that extracts the waveform of the peak voltage. The envelope detector is composed of a rectifier circuit 447, a low-pass filter 448, and a comparator 446.

[0025] The demodulator 440 includes a differential input section formed by an input transistor pair 441A and 441B whose gates are driven by the output of the second high-pass filter 430, and a constant current source 442 that draws current between the drains / sources of these input transistor pairs 441A and 441B, and these constitute a rectifier circuit 447 that rectifies a signal, for example, full-wave rectification. The input transistor pair 441A and 441B are each composed of an N-channel MOSFET.

[0026] The low-pass filter 448 is composed of a resistor element 444 and a capacitor 445 and cuts the high frequency of the output of the rectifier circuit 447. The comparator 446 compares the output of the low-pass filter 448 with a reference signal Vt input to the terminal 443. The comparator 446 compares the output of the low-pass filter 448 with the reference voltage and outputs a signal through the output terminal 120.

[0027] Next, the transmission and reception operations of the data signal in the above configuration and the influence of the parasitic capacitances 403A and 403B will be described. As shown in A of FIG. 4, a data signal is input to the input terminal 110. As shown in B of FIG. 4, the transmitter 200 digitally modulates the data signal. Here, on-off modulation (on-off keying; OOK) is used as the modulation and demodulation method of the data signal. In on-off modulation, when the transmitter 200 outputs 1 as a data value, it outputs a high-frequency carrier pulse signal, and when it outputs 0, it does not output the carrier pulse signal. When the receiver 400 receives the carrier pulse signal, it returns 1 as a data value. Conversely, when the receiver 400 does not receive the carrier pulse signal, that is, when a DC signal is input, it returns 0 as a data value. The signal after being modulated by the transmitter 200 is transmitted to the receiver 400 via the insulating element 300.

[0028] FIG. 3 shows a diagram of the insulating element 300 taking into account the influence of the parasitic capacitances 403A and 403B. At this time, the signal is affected such as being attenuated by the filter formed between the insulating element 300 and its parasitic capacitance 403. Refer to the output waveform of the insulating element shown in C of FIG. 4. Here, the impedance Z of the parasitic capacitance 403 depends on the frequency and is |Z| = 1 / jωC. Note that ω is 2×π×f. The resistance elements 402A and 402B can reduce the influence of the signal attenuation of the high-pass filter formed by the capacitors 300A and 300B and the parasitic capacitances 403A and 403B. Therefore, it is desirable to set the resistance elements 402A and 402B to appropriate impedances based on the carrier pulse frequency of the digital modulation signal. By adopting this configuration, signal transmission can be achieved without attenuating the signal components even when the carrier pulse frequency is low.

[0029] The transfer function is as follows. fc = 1 / (2·π·RC) Here, C is the capacitance value of the insulating element 300, R is the resistance value of the resistance element 402, and the influence of the parasitic capacitance 403 occurs here.

[0030] After passing through the input unit 401, the signal is subsequently input to the first high-pass filter 410. This first high-pass filter 410 allows the desired signal components to pass through while removing unnecessary noise frequency components.

[0031] The signal that has passed through the first high-pass filter 410 is amplified by the gate-grounded amplifier circuit 450. Here, the gate-grounded amplifier circuit 450 removes relatively low-frequency signal components by the capacitor 454 capacitively cross-connected to the output transistor pair 451A, 451B. As a result, the gate-grounded amplifier circuit 450 acts as a high-pass filter. The gate-grounded amplifier circuit 450 includes a differential output transistor pair 451A, 451B, and the output transistor pair 451A, 451B has characteristics equivalent to those of the input low-pass filter of a normal operational amplifier. Therefore, in addition to acting as the aforementioned high-pass filter, the gate-grounded amplifier circuit 450 is configured as a band-pass filter in combination with the input low-pass filter characteristics. It is still desirable to set the frequency characteristics of this band-pass filter based on the carrier pulse frequency of the on-off modulation.

[0032] Then, as shown in D of FIG. 4, the gate-grounded amplifier circuit 450 can output a sine-wave-like signal waveform. Subsequently, as shown in E of FIG. 4, after full-wave rectification by the rectifier circuit 447 of the demodulator 440, the high-frequency signal components are cut by the action of the low-pass filter 448. Thereafter, as shown in F of FIG. 4, the comparator 446 can restore the input signal by shaping this signal. In this way, the receiver 400 with enhanced noise tolerance can be provided.

[0033] The point that the configuration of this embodiment is excellent from the viewpoints of noise reduction and power consumption reduction in particular compared with the conventional configuration will be described. The input unit 401 that inputs a signal through the insulating element 300 has a function of reducing common-mode transient noise N having a period several times that of the carrier pulse period shown in FIG. 5, particularly in a wide frequency range, and differential noise caused by the common-mode transient noise.

[0034] Therefore, as shown at C in FIG. 5, even if common-mode transient noise N is mixed in at the output of the insulating element 300, the components of the noise N can be cut off by the action of the band-pass filter of the first high-pass filter 410 and the gate-grounded amplifier circuit 450. Refer to D in FIG. 5.

[0035] When the resistance value of the resistance element 402 is relatively high, if a common-mode transient current flows through the insulating element 300, the voltage based on this transient current increases. For this reason, there is a concern about breakdown of the withstand voltage of the element connected to the subsequent stage. On the other hand, when the resistance value of the resistance element 402 is relatively low, since the cut-off frequency of the high-pass filter formed by the insulating element 300 and the resistance element 402 becomes high, signal components are likely to be attenuated, raising a concern about deterioration of reception characteristics.

[0036] Generally, the wiring capacitance has a higher withstand voltage than the capacitance of a transistor that can operate at high speed. Therefore, by installing the first high-pass filter 410 at the subsequent stage of the input section 401 by the resistance element 402, a high resistance value can be selected within a range where the withstand voltage of the resistance element 402 can be ensured, and attenuation of signal components can be avoided.

[0037] In addition, the resistance element 412 that constitutes the fully differential first high-pass filter 410 has two types of roles. One is the role of reducing the common-mode transient noise N. Furthermore, by allowing a constant current to flow from the current source 453 into the resistance element 412, it is the role of determining the DC operating point of the input signal. It is desirable to appropriately set the resistance value of the resistance element 412 in consideration of these elements.

[0038] The gate-grounded amplifier circuit 450 acts as a high-pass filter according to the characteristics of the capacitor pairs 454A and 454B for the input signal, and acts as a band-pass filter according to the characteristics of the output transistor pairs 451A and 451B, removing the noise N2 that has passed through the first high-pass filter 410. As a result, the signal after noise removal is output to the subsequent second high-pass filter 430. Further, the second high-pass filter 430 reduces the residual component of the noise N2, and then the demodulator 440 restores the original input signal before modulation by envelope detection using the rectifier circuit 447 and the low-pass filter 448 and shaping the waveform into a rectangular wave by the comparator 446. Thus, a total of four-stage configured filters can be used to remove unnecessary frequency noise. Also, since the form of a fully differential operational amplifier is adopted, common-mode transient noise N can be removed.

[0039] By adopting such a configuration, compared with a circuit that configures a filter by connecting four operational amplifiers in cascade, the number of differential input transistor pairs and constant current sources, which are components of the operational amplifier, can be reduced, and thereby, a receiver 400 with reduced power consumption and manufacturing cost can be provided.

[0040] Although a method of increasing the carrier pulse frequency to reduce the influence of the filter can be considered, the power consumption increases in proportion to the frequency and the capacitor capacitance and also increases in proportion to the square of the voltage. Therefore, it is not preferable that the power consumption increases as the frequency increases. According to the present embodiment, since attenuation of the signal amplitude can be suppressed, it becomes possible to design without increasing the carrier pulse frequency, and a configuration with reduced power consumption can be realized.

[0041] (Second Embodiment) The second embodiment will be described with reference to FIG. 6. The receiver 400B of this embodiment differs from the receiver 400A of the first embodiment in that it includes a band-pass filter 460 having the same configuration as the gate-grounded amplifier circuit 450 instead of the gate-grounded amplifier circuit 450. The band-pass filter 460 is configured based on the configuration of the gate-grounded amplifier circuit 450.

[0042] The band-pass filter 460 includes a pair of bias current sources 463A, 463B and 467A, 467B, and an output transistor pair 465A, 465B to which the currents of the bias current sources 463A, 463B and 467A, 467B are supplied, and is configured as a differential input section for differentially inputting a signal to the sources of the output transistor pair 465A, 465A, 465B.

[0043] The band-pass filter 460 includes differential pair transistors 461A, 461B that are connected in a folded cascode manner between the differential input section and the output of the first high-pass filter 410. The drains of the differential pair transistors 461A, 461B are commonly connected, and a constant current based on the power supply is supplied to the differential pair transistors 461A, 461B from a current source 462.

[0044] The differential pair transistors 461A, 461B are transistors that fold and invert the output of the first high-pass filter 410 when inputting a signal. By adopting this configuration, a signal can be input with a voltage while keeping the gate input of the differential pair transistors 461A, 461B at a high impedance. For this reason, signals with a wide input range can be input, and the selection range of the configuration of the first high-pass filter 410 can be widened.

[0045] (Third Embodiment) The third embodiment will be described with reference to FIGS. 7 to 10. The receiver 400C of the third embodiment differs from the receiver 400A of the first embodiment in that a variable-gain band-pass filter 470 is added between the second high-pass filter 430 and the demodulator 440. The band-pass filter 470 includes a current source 471, differential input transistor pairs 472A and 472B, a variable resistor 473, resistor pairs 474A and 474B, transistor pairs 475A and 475B, and variable capacitor pairs 476A and 476B.

[0046] The differential input transistor pairs 472 are composed of N-channel MOSFETs, and their sources are commonly connected. The commonly connected sources are connected to the ground node through the current source 471. The transistor pairs 475 constitute an active load with P-channel MOSFETs, and a bias is applied to their gates by the variable resistor 473, the resistor 474, and the capacitor 476. The drains of the differential input transistor pairs 472 are connected to the power supply through the drain-source of the transistor pairs 475, and thus power is supplied. It is connected to the demodulator 440 at the output stage.

[0047] The gain-frequency characteristics of the band-pass filter 470 can be changed according to the specifications of the input noise. The gain adjustment can be performed by adjusting the variable resistor 473, or the variable capacitor pairs 476A and 476B, or both of them. By adjusting the gain, the output signal of the demodulator 440 can be adjusted so as not to saturate, and unnecessary noise components can be removed. Also, according to the specifications of the input noise, a band-pass filter having the same configuration as the band-pass filter 470 may be further connected in cascade.

[0048] Examples of signal waveforms during normal operation are shown in FIGS. 8 and 9. FIGS. 8 to 12 illustrate the signal waveforms of the input terminal 110 and nodes 201A, 201B, 405A, 405B, 413A, 413B, 455A, 455B, 434A, 434B, 477A, 477B shown in FIG. 7. As shown in A of FIG. 8, when the digital signal 501 is input, the transmitter 200 transmits the on-off modulated signal 502 to the receiver 400 through the insulating element 300 as shown in B of FIG. 8. The receiver 400 inputs the output signal 503 of the insulating element 300 as shown in C of FIG. 8. At this time, the receiver 400 inputs the signal 503 that has been subjected to high-pass filter processing in the insulating element 300 and the input section 401.

[0049] Also, as shown in D0 of FIG. 8, the first high-pass filter 410 further inputs the signal 504 that has been subjected to high-pass filter processing to the gate-grounded amplifier circuit 450. When the gate-grounded amplifier circuit 450 inputs the signal 504, as shown in D of FIG. 9, the signal 504 is amplified and subjected to band-pass filter processing to output a signal 505.

[0050] As shown in D2 of FIG. 9, the second high-pass filter 430 further inputs the signal 506 that has been subjected to high-pass filter processing to the variable-gain band-pass filter 470. As shown in D3 of FIG. 9, the band-pass filter 470 outputs the signal 507 obtained by cutting the low-frequency and high-frequency components of the input signal 506 to the demodulator 440. As shown in E of FIG. 9, the demodulator 440 generates the signal 508 by envelope detection, and then, as shown in F of FIG. 9, the comparator 446 reshapes the waveform into a rectangular wave to demodulate it into the original signal 509.

[0051] Next, as an example for reference, signal waveform examples when common-mode noise is input are shown in FIGS. 10 and 11, and a single-ended signal at the output of the insulating element 300 when common-mode transient noise N occurs is shown in FIG. 12. When common-mode transient noise occurs, as shown in the lower diagram of FIG. 12, noise N of the same potential occurs in phase on both the plus side and the minus side at the outputs of capacitors 300A and 300B. For this reason, as shown in C of FIG. 10, the waveform is distorted by the influence of noise N based on common-mode noise. Refer to signal 603.

[0052] As shown in D0 of FIG. 11 and D of FIG. 12, the influence of noise N also occurs in signals 604 and 605 at the input and output of the gate-grounded amplifier circuit 450. However, as shown in D2 of FIG. 11, the distortion of signal 606 is gradually removed by the action of the second high-pass filter 430. Further, as shown in signal 607 of D3 of FIG. 11, the influence of the harmonic noise generated in signal 606 is further removed by the action of the band-pass filter 470.

[0053] In particular, as shown in D3 of FIG. 11, the band-pass filter 470 can blunt the waveform of its output signal 607. As shown in E of FIG. 11, the demodulator 440 generates signal 608 by performing full-wave rectification and envelope detection through the actions of the rectifier circuit 447 and the low-pass filter 448. Here, although the influence of noise N is largely absorbed, the influence of noise N appears slightly as shown in the voltage envelope N3. However, since this signal 608 is compared with the reference signal Vt input to terminal 443 by the comparator 446 and exceeds the lower-end voltage of the envelope N3, as shown in F of FIG. 11, the comparator 446 can demodulate it to the original signal 609 by shaping the waveform into a rectangular wave. As a result, the same operational effects as those of the foregoing embodiment can be obtained. However, since the band-pass filter 670 is added, the influence of noise N based on common-mode excessive noise can be significantly reduced.

[0054] (Fourth Embodiment) The fourth embodiment will be described with reference to FIG. 13. The receiver 400D of the fourth embodiment is different from the 400A of the first embodiment in that a gate-grounded amplifier circuit 420 without capacitive cross-connection is used instead of the gate-grounded amplifier circuit 450. The gate-grounded amplifier circuit 420 includes a differential input transistor pair 421A, 421B, a resistor pair 422A, 422B, and a pair of current sources 423A, 423B, but does not include the capacitor pair 454A, 454B described in the first embodiment. Such a configuration may be adopted. According to the fourth embodiment, the receiver 400D can be provided at low cost.

[0055] (Fifth Embodiment) The fifth embodiment will be described with reference to FIG. 14. The receiver 400E of the fifth embodiment is different from the receiver 400D of the fourth embodiment in that the output bias voltage by the second high-pass filter 480 is generated from a reference voltage source 486 instead of a voltage divider.

[0056] The reference voltage source 486 is configured at the subsequent stage of capacitors 481A, 481B. The reference voltage source 486 includes a voltage divider composed of resistors 484 and 485 that divide the power supply voltage, and a voltage buffer 483 that buffers and outputs the divided voltage of the voltage divider using an operational amplifier, and outputs the output of the voltage buffer 483 to the subsequent stage of the capacitors 481A, 481B through resistors 482A, 482B.

[0057] The output voltage of the reference voltage source 486 is applied as the bias voltage of the input transistor pair 441A, 441B of the input of the demodulator 440. Even when the reference voltage source 486 is used as in the fifth embodiment like this, the bias voltage of the input transistor pair 441A, 441B of the demodulator 440 in the subsequent stage circuit can be generated.

[0058] <Circuit Modification Example> Regarding the band - pass filter 470 shown in FIG. 7 described in the third embodiment, even if a reference voltage source 486 is provided when generating the gate - bias voltages of the differential - input transistor pairs 472A and 472B, it can be configured in the same way. That is, also in the configuration shown in FIG. 7, instead of the voltage divider composed of the resistors 432A, 432B, resistors 433A, 433B, it may be configured using the reference voltage source 486.

[0059] (Modification example of the structure) Hereinafter, modification examples will be described. It is applicable regardless of the structure in which the insulating element 300 is used. The following shows variations in the applicable combination configurations of the semiconductor chips 101, 102 and the insulating chip 103.

[0060] In the communication system 100A shown in FIG. 15, an example is shown in which the transmitter 200 is configured on the semiconductor chip 101, the insulating element 300 is configured on the insulating chip 103, and the receiver 400 is configured on the semiconductor chip 102. The semiconductor device 130 shown in FIG. 16 is configured to include these first semiconductor chip 101, second semiconductor chip 102, and insulating chip 103.

[0061] Also, in the communication system 100B shown in FIG. 16, an example is shown in which the transmitter 200 and the insulating element 300 are configured on the first semiconductor chip 101, and the insulating element 300 and the receiver 400 are configured on the second semiconductor chip 102. The semiconductor device 130 shown in FIG. 16 is configured to include the first semiconductor chip 101 and the second semiconductor chip 102.

[0062] Also, in the communication system 100C shown in FIG. 17, an example is shown in which the insulating element 300 is configured on the insulating chip 103 by a capacitor. In the communication system 100D shown in FIG. 18, an example is shown in which the insulating element 300 is configured on the insulating chip 103 by a transformer.

[0063] In addition, the communication system 100E shown in FIG. 19 is composed of two semiconductor chips 101 and 102, and shows an example in which the insulating element 300 is included only in one of the semiconductor chips 102. The communication system 100F shown in FIG. 20 is composed of two semiconductor chips 101 and 102, and shows an example in which the insulating element 300 is included in both of the semiconductor chips 101 and 102.

[0064] (Other embodiments) The present invention is not limited to the above-described embodiments, and can be implemented in various modifications, and is applicable to various embodiments without departing from the gist thereof.

[0065] In the above-described embodiments, an example in which one transmitter 200 and one receiver 400 are included in each of the semiconductor chips 101 and 102 has been shown. However, as in the communication system 100G shown in FIG. 21, each of the semiconductor chips 101 and 102 may be provided with a transmitter 200 and a receiver 400, and / or the insulating chip 103 may be configured to include a plurality of insulating elements 300.

[0066] The capacitors 411A, 411B, 431A, 431B, 454A, 454B, 445, 476A, 476B, 481A, 481B shown in the above-described embodiments may have any structure such as a MIM structure, a MOM structure, or a MOS structure.

[0067] Although the present invention has been described in accordance with the above-described embodiments, it is understood that the present invention is not limited to such embodiments or structures. The present invention includes various modifications and modifications within the equivalent range. In addition, various combinations and forms, and further other combinations and forms including one element, more, or less thereof, are within the scope and spirit of the present invention.

Description of reference numerals

[0068] In the drawings, 100, 100A to 100G represent communication systems, 200 represents a transmitter, 300 represents an insulating element, 400 represents a receiver (receiving device), 401 represents an input section, 403, 403A, 403B represent parasitic capacitances, 410 represents a first high-pass filter, 420 represents a gate-grounded amplifier circuit, 430 represents a second high-pass filter, 440 represents a demodulator, 450 represents a gate-grounded amplifier circuit (band-pass filter), and 460 represents a band-pass filter.

Claims

1. A receiving device (400) that receives a pair of signals from a transmitter (200) through an insulating element (300), comprising: An input section (401) that inputs a pair of signals through the insulating element and includes an impedance conversion section (402) coupled to the insulating element; A first high-pass filter (410) coupled to the input section; Gate-grounded amplifier circuits (450, 460) coupled to the first high-pass filter; A second high-pass filter (430) configured at a subsequent stage of the gate-grounded amplifier circuits; A demodulator (440) configured at a subsequent stage of the second high-pass filter; Comprising: The second high-pass filter is configured by commonly connecting capacitors (431A, 431B) and resistors (435A, 435B), and connecting a common connection node of the capacitors and the resistors to a subsequent-stage circuit connected to a subsequent stage of the second high-pass filter; A receiving device that generates a bias voltage for the subsequent-stage circuit through the resistors (435A, 435B) from an intermediate node of a voltage divider constituted by a plurality of voltage-dividing resistors (432, 433).

2. A receiving device (400) that receives a pair of signals from a transmitter (200) through an insulating element (300), comprising: An input section (401) that inputs a pair of signals through the insulating element and includes an impedance conversion section (402) coupled to the insulating element; A first high-pass filter (410) coupled to the input section; Gate-grounded amplifier circuits (450, 460) coupled to the first high-pass filter; A second high-pass filter (430) configured at a subsequent stage of the gate-grounded amplifier circuits; A demodulator (440) configured at a subsequent stage of the second high-pass filter; Comprising: The second high-pass filter is configured by connecting capacitors (431A, 431B) and resistors (482A, 482B), and Includes a reference voltage source (486) that outputs a reference voltage, A receiving device in which the reference voltage source outputs the reference voltage as a bias voltage to a subsequent-stage circuit through the resistors of the second high-pass filter.

3. The receiving device according to claim 1 or 2, wherein the subsequent-stage circuit is constituted by the demodulator.

4. The gate-grounded amplifier circuit includes a pair of bias current sources (453A, 453B) and an output transistor pair (451A, 451B) to which the currents of the bias current sources are supplied, and the sources of the output transistor pair are configured as a differential input section for differentially inputting the output of the first high-pass filter. The receiving apparatus according to any one of claims 1 to 3, wherein a differential input section of the gate-grounded amplifier circuit and the gates of the output transistor pair are capacitively cross-coupled to provide characteristics as a band-pass filter (450).

5. The gate-grounded amplifier circuit includes a pair of bias current sources (453A, 453B) and an output transistor pair (451A, 451B) to which the currents of the bias current sources are supplied, and is configured as a differential input section for differentially inputting a signal to the sources of the output transistor pair. The receiving apparatus according to any one of claims 1 to 3, wherein the gate-grounded amplifier circuit includes differential pair transistors (461A, 461B) connected in a folded cascode between a differential input section of the gate-grounded amplifier circuit and the output of the first high-pass filter, and has characteristics as a band-pass filter (450).

6. The receiving apparatus according to any one of claims 1 to 3, wherein the first high-pass filter includes capacitors (411A, 411B) coupled in series to the input section and resistor elements (412A, 412B) coupled in parallel to the capacitors.

7. The transmitter is configured to transmit an on-off modulated digital modulation signal. The receiving apparatus according to any one of claims 1 to 3, wherein the demodulator demodulates the digital modulation signal.

8. The receiving apparatus according to any one of claims 1 to 3, wherein the demodulator includes an envelope detector that shapes an input signal and extracts a waveform of a peak voltage.

9. The envelope detector includes a rectifying circuit (447) that rectifies an input signal, a low-pass filter (448) that cuts high frequencies of an output of the rectifying circuit, and further includes a comparator (446) that compares an output of the low-pass filter with a reference signal to shape the waveform into a rectangular wave. The receiving apparatus according to claim 8 is configured.

10. The insulating element includes at least one or more inputs and at least one or more outputs and is connected to the input section. The receiving apparatus according to any one of claims 1 to 9.

11. The insulating element includes at least one or more inputs and at least one or more outputs and is connected to the input section. The receiving device according to any one of claims 1 to 10, wherein the transmitter that on-off modulates an input signal is connected to a preceding stage of the insulating element.

12. The receiving device according to claim 10 or 11, wherein the insulating element is constituted by a capacitor.

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