Motor control method and motor control device

Abstract

To achieve sensorless control in which torque fluctuation caused by transition of powering operation and regenerative operation is suppressed.SOLUTION: A motor control method includes: setting power command values Vd1*, Vq1*, Vd2*, and Vq2* for regulating power to be supplied to a motor 200 on the basis of a predetermined torque command value T*; making high frequency voltages Vdh* and Vqh* overlap the power command values; estimating a magnetic pole position θ' of the motor on the basis of response power values idh and iqh to the input of the high frequency voltages; and controlling the power supplied to the motor on the basis of the power command value in which the high frequency voltage is overlapped and the estimated magnetic pole position. Especially, the motor control method includes: determining whether the driving state of the motor is power operation or regenerative operation; setting a retardation φdq1 for powering as a retardation φdq of a q-axis component to a d-axis component of the high frequency voltage at the time of the powering operation; and setting a retardation φdq2 for regeneration different from the retardation φdq1 for powering as the retardation φdq at the time of the regenerative operation.SELECTED DRAWING: Figure 3

Description

[Technical Field]

[0001] The present invention relates to a motor control method and a motor control device. [Background technology]

[0002] Patent Document 1 discloses a control method (sensorless control) in which a high-frequency voltage with a higher frequency than the frequency of the drive power is superimposed on the drive power (power command value) applied to drive the motor and supplied to the motor, and the high-frequency current, which is the response, is extracted and processed to estimate the rotor phase (magnetic pole position) of the motor. [Prior art documents] [Patent Documents]

[0003] [Patent Document 1] Japanese Patent Publication No. 2007-185080 [Overview of the project] [Problems that the invention aims to solve]

[0004] In the control method described in Patent Document 1, the high-frequency voltage is set without considering the motor's operating state (powering or regenerative operation). As a result, the response of the motor's output torque to the driving power superimposed with the high-frequency voltage varies during both powering and regenerative operation, causing torque fluctuations.

[0005] Therefore, the object of the present invention is to provide a motor control method and a motor control device that realize sensorless control in which the occurrence of torque fluctuations is suppressed during both power operation and regenerative operation. [Means for solving the problem]

[0006] According to one aspect of the present invention, a motor control method is provided which sets a power command value that defines the power to be supplied to the motor based on a predetermined torque command value, superimposes a high-frequency voltage on the power command value, estimates the magnetic pole position of the motor based on the response power value to the input of the high-frequency voltage, and controls the power supplied to the motor based on the power command value with the superimposed high-frequency voltage and the estimated magnetic pole position. In this motor control method, it is determined whether the motor is in a powered operation or a regenerative operation, and in a powered operation, the phase difference between the q-axis component and the d-axis component of the high-frequency voltage is set to a powered phase difference, and in a regenerative operation, the phase difference is set to a regenerative phase difference different from the powered phase difference. [Effects of the Invention]

[0007] According to the present invention, sensorless control can be achieved in which torque fluctuations are suppressed during both power operation and regenerative operation. [Brief explanation of the drawing]

[0008] [Figure 1] Figure 1 is a block diagram showing the configuration of a motor control device according to each embodiment of the present invention. [Figure 2] Figure 2 shows an example of the relationship between torque, elliptic coefficient, and flatness. [Figure 3] Figure 3 shows the first torque-elliptic coefficient table. [Figure 4] Figure 4 shows the second torque-elliptic coefficient table. [Figure 5] Figure 5 shows a table of minimum values ​​for the salient pole ratio-coefficient. [Figure 6] Figure 6 is a flowchart showing the processing flow in the high-frequency voltage command generation unit. [Figure 7] Figure 7 is a timing chart showing an example of the control result obtained by the motor control method of this embodiment. [Modes for carrying out the invention]

[0009] Embodiments of the present invention will be described below with reference to the drawings.

[0010] [First Embodiment] Figure 1 is a block diagram showing the configuration of the motor control device 100. As shown in Figure 1, the motor control device 100 is assumed to be a device that, instead of measuring the magnetic pole position (rotor position) and rotational speed (hereinafter referred to as "motor rotational speed N") of the motor 200 with sensors such as resolvers and encoders, calculates these based on an estimation algorithm described later, and uses the calculated estimated value to operate the inverter 18 and control the power supplied to the motor 200. In other words, the motor control system of this embodiment, consisting of the motor control device 100, inverter 18, and motor 200, can be configured as a position sensorless system. In particular, the motor 200 to be controlled is assumed to be an on-board motor for driving or a power generation motor. Furthermore, the motor 200 is, for example, a three-phase AC IPM (Interior Permanent Magnet) motor, with a q-axis inductance L q and d-axis inductance L d It is configured as a sali-polarity motor, where each element exhibits a different value.

[0011] The motor control device 100 comprises a current command generation unit 11, a first voltage command generation unit 12, a second voltage command generation unit 13, a final voltage command generation unit 14, a control mode signal generation unit 15, a coordinate transformation unit 16, a PWM conversion unit 17, a phase / rotation speed estimation unit 19, a high-frequency voltage command generation unit 20, and a coordinate transformation unit 23. The motor control device 100 is composed of a computer (controller) with a program to realize the functions of each unit. The hardware constituting the computer may consist of one or more units.

[0012] The current command generation unit 11 generates a torque command value T * DC voltage V dc Based on the rotational speed estimate N′, the d-axis current command value i d * and q-axis current command value i q * Generate (calculate) and output it.

[0013] Here, the torque command value T * is the target value of the output torque of the motor 200 defined by the upper control device. In particular, when the motor 200 is used as the driving source of the vehicle, it is determined to an appropriate value according to the required driving force (such as the operation amount of the accelerator pedal). Also, the DC voltage V dc is the output voltage of the battery 21 for supplying the driving power of the motor 200. The DC voltage V dc is detected by the voltage sensor 22. Note that the DC voltage V dc may be obtained as an estimated value obtained by a battery controller (not shown) or the like instead of the detected value by the voltage sensor 22. Further, the estimated rotational speed N′ is an estimated value of the rotational speed of the motor 200 (hereinafter, also referred to as "motor rotational speed N") calculated according to the control state of the motor 200. The estimated rotational speed N′ is calculated (estimated) by the phase / rotational speed estimation unit 19 according to an estimation algorithm (high-frequency voltage application method) described later.

[0014] The d-axis current command value i<00000l3> * is the command value for the d-axis current i d of the motor 200. The q-axis current command value i q * is the command value for the q-axis current i q of the motor 200. Note that hereinafter, for the sake of notation simplification, the d-axis current i d and the q-axis current i q are collectively referred to as the dq-axis current i d ,i q and the d-axis current command value i d * and the q-axis current command value i q * are collectively referred to as the dq-axis current command value i d * ,i q * .

[0015] The first voltage command generation unit 12 is the torque command value T * , the DC voltage V dc, rotational speed estimated value N′, dq axis current command value i d * ,i q * , and dq axis current i d ,i q Based on the detected value, the first d-axis voltage command value V d1 * and the first q-axis voltage command value V q1 * (Hereafter, "First voltage command value V d1 * ,V q1 * It generates (calculates) and outputs (also known as ).

[0016] First voltage command value V d1 * ,V q1 * This is a voltage command value for controlling the motor 200 by so-called current vector control. That is, the first voltage command generation unit 12 generates the dq axis current i according to, for example, the following equation (1). d ,i q and dq axis current command value i d * ,i q * Deviation (i d -i d * ,i q -i q * ) PI control based on and dq axis decoupling, the first voltage command value V d1 * ,V q1 * Perform the calculation.

[0017]

number

[0018] Note that "s" in equation (1) is a differential operator. Also, "K p1 " is a proportional gain, and "K i1 " is the integral gain. Furthermore, "V d-dcpl * " is the d-axis interference voltage, and "Vq-dcpl * " is the q-axis interference voltage. In the present embodiment, the first voltage command generation unit 12 determines the torque command value T * , the DC voltage V dc , and the estimated rotational speed N', and associates them with the d-axis interference voltage V d-dcpl * and the q-axis interference voltage V q-dcpl * in an interference voltage table (not shown). Therefore, the first voltage command generation unit 12 refers to this interference voltage table to calculate the d-axis interference voltage V * , the DC voltage V dc , and the estimated rotational speed N' corresponding to the q-axis interference voltage V d-dcpl * and the q-axis interference voltage V q-dcpl * . The interference voltage table is preset through experiments or simulations, etc.

[0019] Note that the dq-axis currents i d1 * ,V q1 * used by the first voltage command generation unit 12 for calculating the first voltage command values V d ,i q are the detected values of the currents flowing through the motor 200 and are obtained from the coordinate conversion unit 23. The coordinate conversion unit 23 performs coordinate conversion using, for example, the phase estimation value θ' output by the phase / rotational speed estimation unit 19 to calculate the dq-axis currents i u ,i v ,i w from the currents i d ,i q of each phase of the motor 200 detected by the current sensor 24. Specifically, the coordinate conversion unit 23 calculates the dq-axis currents i d ,i q according to the following equation (2).

[0020]

Equation

[0022]

number

[0023] In the following, the dq-axis current i is calculated based on equations (2) and (3) above. d ,i q The value of the dq axis current i d ,i q The detected value is considered to be the value, and when it is specifically indicated that it is a detected value, it is referred to as "dq axis current detected value i ddet ,i qdet "d-axis current detection value i" ddet ", or "q-axis current detection value i qdet It is written as "".

[0024] The second voltage command generation unit 13 generates a torque command value T * DC voltage V dc , rotational speed estimate N′, and dq axis current i d ,i q Based on this, the 2nd d-axis voltage command value V d2 * and the 2q-axis voltage command value V q2 * This generates (calculates) the second d-axis voltage command value V. d2 * and the 2q-axis voltage command value V q2 * The appropriate value is "Second voltage command value V d2 * ,V q2 * It is also called "..."

[0025] Second voltage command value V d2 * ,V q2 * This is a voltage command value for controlling the motor 200 by so-called voltage phase control. That is, the second voltage command generation unit 13 generates a voltage norm Va The command value for the voltage norm is V. a * And the command value for voltage phase α is the voltage phase command value α. * The second voltage command value V is calculated using the following: d2 * ,V q2 * Perform the calculation.

[0026] Specifically, the second voltage command generation unit 13 generates a DC voltage V according to, for example, the following equation (4): dc And the modulation rate command value MF is the command value for the modulation rate. * Based on this, the voltage norm command value V a * Perform the calculation.

[0027]

number

[0028] Furthermore, the second voltage command generation unit 13 generates a torque command value T * DC voltage V dc Based on the estimated rotational speed N′, the voltage phase target value α ff * The following is calculated: Voltage phase target value α ff * This is the target value of the voltage phase α due to feedforward control. In this embodiment, the second voltage command generation unit 13 generates the torque command value T * DC voltage V dc , and the rotational speed estimate N′, and the voltage phase target value α ff * It has a voltage phase target value table (not shown) that associates these values. Therefore, the second voltage command generation unit 13 refers to this voltage phase target value table to generate the torque command value T * DC voltage V dc , and the voltage phase target value α corresponding to the rotational speed estimate N′ ff * The calculation is performed. The voltage phase target value table is predetermined by experimentation or simulation.

[0029] Furthermore, the second voltage command generation unit 13 detects the dq axis current i ddet ,i qdet Based on the rotational speed estimate N', the estimated torque T is the torque output by the motor 200 (hereinafter also simply referred to as "torque T"). est The calculation is performed. In this embodiment, the second voltage command generation unit 13 calculates the dq axis current i d ,i q And the rotational speed estimate N′ and the torque estimate T est It has a torque estimation table (not shown) that associates and . Therefore, the second voltage command generation unit 13 refers to this torque estimation table and determines the dq axis current detection value i ddet ,i qdet and the torque estimate T corresponding to the rotational speed estimate N′ est The calculation is performed. The torque estimation table is pre-set by experimentation or simulation, etc.

[0030] Furthermore, the second voltage command generation unit 13 generates a torque command value T * and torque estimate T est Based on this, the voltage phase correction value α fb * The following is calculated: Voltage phase correction value α fb * The voltage phase target value α ff * This is a correction value for the torque command value T, and is calculated by feedback control. For example, the second voltage command generation unit 13 calculates the torque command value T according to the following equation (5). * and torque estimate T est The deviation from (T fin * -T est ) is controlled by PI control, which provides a voltage phase correction value α fb * Perform the calculation.

[0031]

number

[0032] In equation (5), "Kp2 " is a proportional gain, and "K i2 This represents the integral gain.

[0033] Then, the second voltage command generation unit 13 calculates the voltage phase target value α as described above. ff * Voltage phase correction value α fb * By adding this, the voltage phase command value α * Perform the calculation.

[0034] Furthermore, the second voltage command generation unit 13 generates a voltage norm command value V as shown in equation (6) below. a * and voltage phase command value α * Based on the vector transformation, the second voltage command value V d2 * ,V q2 * Perform the calculation.

[0035]

number

[0036] The final voltage command generation unit 14 generates a first voltage command value V based on the control mode signal Msw input from the control mode signal generation unit 15. d1 * ,V q1 * and the second voltage command value V d2 * ,V q2 * Select one of the following, and the final voltage command value is the final voltage command value V d * ,V q * It outputs as follows: That is, the final voltage command generation unit 14 sets the control mode according to the control mode signal Msw, and the first voltage command value V d1 * ,V q1 * A current vector control mode using and a second voltage command value V d2 * ,V q2* It selectively switches between a voltage phase control mode using and another mode.

[0037] In this embodiment, the final voltage command generation unit 14 selects a first voltage command value V based on the control mode signal Msw. d1 * ,V q1 * Or the second voltage command value V d2 * ,V q2 * The high-frequency voltage V, described later, is input from the high-frequency voltage command generation unit 20. dh * ,V qh * The output is obtained by superimposing (adding) the two values. Note that this is done by the phase / rotation speed estimation unit 19, which calculates the high-frequency voltage V dh * ,V qh * The final voltage command value V is obtained by superimposing the two values. d * ,V q * Response power value (response high-frequency current i) dh ,i qh This is to calculate the rotational speed estimate N′ and the phase estimate θ′ from ).

[0038] The control mode signal generation unit 15 generates a DC voltage V dc and final voltage command value V d * ,V q * Based on this, the control mode signal Msw is generated.

[0039] Specifically, the control mode signal generation unit 15 generates a DC voltage V according to the following equation (7). dc and final voltage command value V d * ,V q * Based on this, the modulation index (MF) is calculated.

[0040]

number

[0041] Furthermore, the control mode signal generation unit 15 sets the calculated modulation rate MF to a predetermined threshold (modulation rate threshold TH). MF ) is compared with the modulation rate threshold TH. Then, the control mode signal generation unit 15 generates a control mode signal such that, for example, the modulation rate MF is compared with the modulation rate threshold TH. MF When it is less than the first voltage command value V d1 * ,V q1 * Control mode signal M that selects (current vector control mode) sw It generates and outputs the following. Meanwhile, the control mode signal generation unit 15 generates the following signal, for example, when the modulation rate MF is the modulation rate threshold TH. MF When the above is true, the second voltage command value V d2 * ,V q2 * A control mode signal Msw is generated and output to select (voltage phase control mode).

[0042] The modulation rate threshold TH used to determine the switching from current vector control mode to voltage phase control mode is also used. MF The modulation rate threshold TH is used to determine the switching from voltage phase control mode to current vector control mode. MF And can be set to different values. In this case, hysteresis can be introduced into the switching between the current vector control mode and the voltage phase control mode, thereby suppressing frequent switching between each mode (so-called chattering).

[0043] The coordinate transformation unit 16 performs a coordinate transformation using, for example, the phase estimated value θ′ output by the phase / rotation speed estimation unit 19 to obtain the final voltage command value V d * ,V q * From there, the three-phase voltage command value V u * ,V v * ,V w * The coordinate transformation unit 16 calculates the three-phase voltage command value V according to the following equation (8). u * ,Vv * ,V w * Perform the calculation.

[0044]

number

[0045] The PWM conversion unit 17 outputs a DC voltage V dc and three-phase voltage command value V u * ,V v * ,V w * Based on this, a PWM (Pulse Width Modulation) signal is generated to drive the power elements of the inverter 18. Specifically, the PWM conversion unit 17 generates a three-phase voltage command value V u * ,V v * ,V w * Power element drive signal D corresponding to this signal uu * ,D ul * ,D vu * ,D vl * ,D wu * ,D wl * This is generated and input to the inverter 18. The PWM conversion unit 17 can perform so-called dead time compensation processing and voltage utilization rate improvement processing when generating the PWM signal.

[0046] The inverter 18 switches the power elements according to the PWM signal, thereby controlling the DC voltage V of the battery 21. dc This is a pseudo-AC voltage V u ,V v ,V w This is converted and input to each of the UVW phases of motor 200. As a result, motor 200 receives the torque command value T. * It is controlled to output a torque T corresponding to the value.

[0047] The phase / rotation speed estimation unit 19 detects the dq axis current i ddet ,i qdet and final voltage command value V d * ,V q * Based on this, the rotational speed estimate N′ and the phase estimate θ′ are calculated and output. The phase estimate θ′ is an estimate of the phase θ that defines the rotor position of the motor 200. In particular, the phase / rotational speed estimation unit 19 calculates the q-axis inductance L of the motor 200. q and d-axis inductance L d The rotational speed estimate N′ and phase estimate θ′ are calculated using the high-frequency voltage application method (mirror phase estimation method), which is a magnetic position estimation algorithm that utilizes the differences in these factors.

[0048] Specifically, first, the phase / rotation speed estimation unit 19 calculates the phase error estimate θ. γ The value ' is calculated and output. Here, the estimated phase error value θ is calculated. γ ′ represents the phase error θ between the dq axis coordinate system and a predetermined two-axis control coordinate system (γδ axis coordinate system). γ This is an estimated value.

[0049] First, as a premise, the first voltage command value V is as described above. d1 * ,V q1 * Or the second voltage command value V d2 * ,V q2 * A high-frequency voltage V is superimposed on it. dh * ,V qh * It traces an elliptical orbit with major and minor axes in directions coinciding with the γ-axis and δ-axis, respectively. Then, the high-frequency voltage V dh * ,V qh * The response power value for the response is the response high-frequency current i. dh ,i qh It also traces an elliptical orbit on the γδ axis coordinate system. Here, the response high-frequency current i dh ,i qh The major axis has a predetermined phase difference with respect to the d-axis (hereinafter referred to as "major axis phase θ").re This results in a phase difference of θ (referred to as "). On the other hand, the phase error θ between the dq axis coordinate system and the γδ axis coordinate system mentioned above occurs. γ and the phase θ of the major axis γe When both are infinitesimally small, they can be considered to coincide with each other. Therefore, the major axis phase θ γe If we find this, then this will be the phase error θ γ The estimated value of (hereinafter referred to as "phase error estimate θ") γ It can also be called ′」).

[0050] Therefore, the phase / rotation speed estimation unit 19 determines the dq axis current detection value i ddet ,i qdet and final voltage command value V d * ,V q * From the major axis phase θ γe The following calculation logic is executed to determine the estimated phase error θ γ Perform the calculation '.

[0051] Specifically, the phase / rotation speed estimation unit 19 uses filtering processing, such as a bandpass filter, to determine the dq axis current detection value i ddet ,i qdet From response high-frequency current i dh ,i qh Extracts the high-frequency voltage V. A bandpass filter is used, for example, for high-frequency voltage V. dh * ,V qh * The frequency ω h Depending on the current detected on the dq axis, i ddet ,i qdet The DC component is removed or reduced from it. The phase / rotation speed estimation unit 19 then calculates the response high-frequency current i according to equations (9) and (10) below. dh ,i qh The positive phase component (in-phase component) of [c p ,s p ] and the reversed phase component (mirror phase component) [c n ,s n Perform the operation on ] and .

[0052] [Number]

[0053] Then, the phase / rotation speed estimation unit 19 further uses a low-pass filter or the like to perform filtering processing for removing or reducing harmonic components such as frequency 2ω from the positive-phase component [c p ,s p and the negative-phase component [c n ,s n . The positive-phase component [c h etc.] and the negative-phase component [c p ,s p and the negative-phase component [c n ,s n are symmetric (mirror correlation) with respect to the major-axis phase θ re (phase error estimated value θ γ ′) in the γδ-axis coordinate system. Therefore, the phase / rotation speed estimation unit 19 calculates the phase error estimated value θ p ,s p and the negative-phase component [c n ,s n according to the following equation (11). γ ′.

[0054] [Number]

[0055] Also, the phase / rotation speed estimation unit 19 calculates an estimated value of the electrical angular velocity ω′, which is an estimated value of the electrical angular velocity of the motor 200, by PI control of the following equation (12) based on the phase error estimated value θ γ ′ determined by equation (11).

[0056] [Number]

[0057] Furthermore, the phase / rotation speed estimation unit 19 calculates a phase estimated value θ′ by integrating the estimated value of the electrical angular velocity ω′ according to the following equation (13).

[0058]

number

[0059] Furthermore, the phase / rotational speed estimation unit 19 calculates the rotational speed estimate N' [rad] by converting the electrical angular velocity estimate ω' [rad / sec] into units.

[0060] The high-frequency voltage command generation unit 20 generates a torque command value T * With the above high-frequency voltage V as input, dh * ,V qh * Generate and output the following.

[0061] More specifically, the high-frequency voltage command generation unit 20 generates a high-frequency voltage V, for example, represented by the following equation (14). dh * ,V qh * Generates.

[0062]

number

[0063] In equation (14), "V h " is a high-frequency voltage V dh * ,V qh * The fundamental amplitude is "ω h " is the frequency. Note that the fundamental amplitude V h , and frequency ω h This is pre-set by compatibility.

[0064] Also, the coefficient "K h (-1≦K h ≤1) is a predetermined phase difference (phase error θ described later) with respect to the dq axis coordinate system. γ High-frequency voltage V in a two-axis control coordinate system (γδ axis coordinate system) with ) dh * ,V qh * These are coefficients that determine the trajectory. In particular, Kh If = 0, then the high-frequency voltage V dh * ,V qh * The trajectory is a straight line along the γ axis, -1 <K h <0,0 <K h When <1, the locus is an ellipse centered at the origin of the γδ axis (the origin of the dq axis). Furthermore, K h When = -1 or 1, the locus is a perfect circle centered at the origin of the γδ axis. Also, as can be seen from equation (14), "K h " is the d-axis high-frequency voltage V dh * Q-axis high-frequency voltage V qh * This defines the ratio of the amplitudes. In the following, this will be referred to as the "elliptic coefficient K". h It is called "".

[0065] Here, the high-frequency voltage V shown in equation (14) dh * ,V qh * In this case, the d-axis high-frequency voltage V dh * Q-axis high-frequency voltage V qh * phase difference (hereinafter referred to as "dq axis phase difference φ dq The elliptic coefficient K (also known as "...") is h It changes depending on the sign.

[0066] Specifically, K h When it is positive, the phase difference between the dq axes is φ. dq This corresponds to a 90° phase lead of the sine function relative to the cosine function. On the other hand, K h When it is negative, the phase difference between the dq axes is φ. dq This corresponds to the phase lag of the -sin function relative to the cosine function, which is -90°. Note that K h When = 0, the q-axis high-frequency voltage V qh * It becomes 0. That is, the high-frequency voltage V dh * ,V qh * Since it only has a d-axis component, the phase difference between the d and q axes is φ. dqThe result is 0.

[0067] Therefore, the elliptic coefficient K h By appropriately changing the sign of φ, the phase difference between the dq axes can be changed. dq The torque command value T can be adjusted. In particular, the high-frequency voltage command generation unit 20 of this embodiment can adjust the torque command value T * The system refers to the current drive state of motor 200 to determine whether it is in a powered or regenerative state, and the elliptic coefficient K is determined according to the result of that determination. h Change the sign of the symbol.

[0068] Here, the inventors apply a high-frequency voltage V to the motor 200. dh * ,V qh * When applying the same torque T, the elliptic coefficient K h The sign of (i.e., the phase difference between the dq axes φ) dq ) If the response high-frequency current i is different dh ,i qh Experiments have shown that the degree of flattening f (=[major axis - minor axis] / major axis) of the elliptical trajectory traced on the γδ axis differs.

[0069] Figure 2 shows torque T and elliptic coefficient K. h This figure shows an example of the relationship between the elliptic coefficient K and the flattening degree f. In particular, Figure 2 shows the elliptic coefficient K. h Assuming that the absolute values ​​of the two values ​​are the same, the relationship between torque T and flatness f when the sign is positive is shown in a solid line graph, and the relationship between torque T and flatness f when the sign is negative is shown in a dashed line graph.

[0070] As shown in the figure, the elliptic coefficient K h When is positive, the flattening f decreases sharply in the high torque region in the powering region (T>0). On the other hand, the elliptic coefficient K h When is negative, the flatness f is maintained at a relatively large value even in the high torque range. Here, the response high-frequency current i dh ,i qhIt has been confirmed that the larger the flatness f of the elliptical trajectory drawn, the easier it is to ensure the accuracy of the estimation of the magnetic pole position of the motor 200, even in high torque ranges. In other words, the larger the flatness f, the wider the torque range in which motor drive based on estimated magnetic pole position (sensorless drive) can be made. Therefore, during powering operation, the elliptic coefficient K h Setting the sign of this parameter to a negative value allows for a higher maximum torque that can be driven without a sensor compared to setting it to a positive value (it expands the torque range in which sensorless driving is possible).

[0071] On the other hand, the elliptic coefficient K in the regenerative region (T<0) h The relationship between the sign and the magnitude of the flattening f shows the opposite trend to that in the powering region. Therefore, during regenerative operation, the elliptic coefficient K h Setting the sign of this parameter to positive allows for a higher maximum torque that can be driven without a sensor compared to setting it to negative (it expands the torque range in which sensorless driving is possible).

[0072] Furthermore, the inventors have identified a specific elliptic coefficient K h Regardless of the absolute value of, the elliptic coefficient K in the above-mentioned powering region and regenerative region h We have confirmed that a general trend holds true regarding the relationship between the sign of and the magnitude of the flattening f.

[0073] Therefore, in this embodiment, the high-frequency voltage command generation unit 20 sets the elliptic coefficient K to be negative in the motoring region (T>0) and positive in the regenerative region of the motor 200. h Using equation (14), the high-frequency voltage V dh * ,V qh * This defines the torque range in which sensorless driving is possible, both during power operation and regenerative operation.

[0074] The following describes the process in more detail. First, the high-frequency voltage command generation unit 20 generates a torque command value T *By applying a filter determined in consideration of the response delay of the control system to, the theoretical value (torque specification value) of the torque T actually output by the motor 200 is calculated, and the elliptic coefficient K is obtained by referring to the first torque-elliptic coefficient table from the torque specification value. h is determined.

[0075] FIG. 3 is a diagram showing the first torque-elliptic coefficient table. As shown in the figure, in the first torque-elliptic coefficient table, the sign of the elliptic coefficient K h is set negative in the power running region and positive in the regeneration region. Also, in the first torque-elliptic coefficient table, for the same absolute value of the torque T in both the power running region and the regeneration region, the elliptic coefficient K h is defined. Therefore, by using the elliptic coefficient K h determined by referring to the above first torque-elliptic coefficient table, during the power running operation and the regeneration operation of the motor 200, for the same absolute value of the torque |T|, the ratio of the amplitudes of the d-axis high-frequency voltage V dh * and the q-axis high-frequency voltage V qh * does not change, and their phase difference is reversed. In other words, the phase difference φ dq between the d and q axes during the power running operation (hereinafter, also referred to as "phase difference φ dq1 for power running") and the phase difference φ dq between the d and q axes during the regeneration operation (hereinafter, also referred to as "phase difference φ dq2 for regeneration") differ by 180°.

[0076] Furthermore, in the first torque-elliptic coefficient table, in the region where the absolute value of the torque T is less than or equal to a predetermined reference value A in both the power running region and the regeneration region (hereinafter, also referred to as the "transition region"), the elliptic coefficient K h is defined to continuously change as the absolute value of the torque |T| decreases. That is, the elliptic coefficient K h is determined to gently switch between positive and negative in the vicinity of the transition point (T = 0) between power running and regeneration.

[0077] Thereby, the elliptic coefficient K hThis method can suppress the abrupt change in the phase estimate θ′ and the resulting abrupt change in torque T caused by abruptly changing the sign of the function.

[0078] In particular, in the first torque-elliptic coefficient table, the elliptic coefficient K is set such that in the transition region, the absolute torque value |T| approaches 0 as it approaches 0. h It stipulates that...

[0079] Here, the inventors have found that the smaller the absolute torque value |T|, the smaller the estimated phase error value θ. γ The ' becomes smaller, and the elliptic coefficient K h The larger the phase error estimate θ, the greater the phase error estimate. γ Stability region for ′ (major axis phase θ) re and phase error θ γ The phase error estimate θ shows a positive correlation. γ We have confirmed that the range of ′ becomes wider. In contrast, in the first torque-elliptic coefficient table, the elliptic coefficient K h The magnitude (absolute value) is the phase error estimate θ. γ In the low-torque region where a certain degree of narrowing of the stable region relative to ' is acceptable, the elliptic coefficient K can be gradually reduced, thereby preventing abrupt changes in torque T while ensuring estimation accuracy. More specifically, in the region where phase errors are likely to occur (the region where the absolute torque value |T| is relatively large), the elliptic coefficient K h Maintaining a certain magnitude, the phase error estimate θ γ While ensuring a wide range of stability relative to ', the elliptic coefficient K is used in the region where phase errors are less likely to occur (the region where the absolute torque value |T| is relatively small). h By reducing the value of the elliptic coefficient K so that its sign switches at the transition point, estimation accuracy is ensured while maintaining the elliptic coefficient K. h By gradually switching between positive and negative polarity, sudden changes in torque T can be avoided.

[0080] Furthermore, from the viewpoint of suppressing the abrupt changes in torque T and the decrease in estimation accuracy mentioned above, a second torque-elliptic coefficient table different from the first torque-elliptic coefficient table may be used.

[0081] Figure 4 shows the second torque-elliptic coefficient table. As shown in the figure, in the second torque-elliptic coefficient table, the elliptic coefficient K h The absolute value of the coefficient K gradually decreases in the transition region (-A ≤ T ≤ A) until it reaches a predetermined minimum coefficient value B, but once it reaches the minimum coefficient value B, it is specified to maintain that value until the transition point. At the transition point, the elliptic coefficient K h This is defined such that the sign is reversed while its absolute value is maintained at the minimum coefficient value B.

[0082] Here, the minimum coefficient value B is the estimated phase error value θ. γ The elliptic coefficient K is determined from the perspective of ensuring that the width of the stable region for ' is above a certain level. h This is the value of . The minimum coefficient value B can be determined in advance by experiment or simulation to an appropriate value. Furthermore, the inventors have determined the phase error estimate value θ γ The width of the stable region relative to ' is the salient pole ratio L of motor 200 q / L d (d-axis inductance L) d Q-axis inductance L q It has been found that this correlates with the magnitude of the ratio of . Therefore, when using the second torque-elliptic coefficient table, refer to the pre-prepared salient pole ratio-minimum coefficient value table to determine the salient pole ratio L of motor 200. q / L d The minimum coefficient value B is determined from this.

[0083] Figure 5 shows a salient ratio-minimum coefficient value table. As shown in the figure, in the salient ratio-minimum coefficient value table, the minimum coefficient value B is equal to the salient ratio L q / L d The larger the value, the smaller the value it takes. This results in the salient pole ratio L q / L d The larger the phase error estimate θ, the greater the phase error estimate. γ The minimum coefficient value B can be appropriately determined in accordance with the tendency for the stable region to widen with respect to ′. Therefore, using the second torque-elliptic coefficient table and the salient pole ratio-minimum coefficient table, the torque T (torque reference value) and salient pole ratio L q / L d From the elliptic coefficient Kh By defining this, it is possible to suppress abrupt changes in the phase estimate θ′ while ensuring its estimation accuracy.

[0084] When using the second torque-elliptic coefficient table and the salient pole ratio-minimum coefficient value table, the high-frequency voltage command generation unit 20 performs the following processing.

[0085] The high-frequency voltage command generation unit 20 first refers to the dq-axis inductance table, which is determined in advance by experiment or analysis, and sets the detected dq-axis current i ddet ,i qdet From the salient pole ratio L q / L d And this salient pole ratio L is determined. q / L d and the torque command value T mentioned above * Based on the torque reference value determined according to the second torque-elliptic coefficient table and the salient pole ratio-minimum coefficient value table, the elliptic coefficient K is calculated by referring to the second torque-elliptic coefficient table. h Perform the calculation.

[0086] Next, a series of processes in the high-frequency voltage command generation unit 20 will be explained in summary.

[0087] Figure 6 is a flowchart showing the processing flow in the high-frequency voltage command generation unit 20. Each of the following processes is repeatedly executed at predetermined calculation cycles.

[0088] As shown in the figure, first in step S10, the torque command value T * , and the detected current value i on the dq axis ddet ,i qdet Obtain it.

[0089] In step S20, the salient pole ratio L of the motor 200 q / L d The following calculation is performed. Specifically, the dq-axis current detection value i is calculated by referring to a dq-axis inductance table that has been obtained in advance by experiment or analysis. ddet ,i qdet From the salient pole ratio L q / L d To define.

[0090] In step S30, the elliptic coefficient K h The first method involves calculating the torque command value T by referring to the first torque-elliptic coefficient table. * The elliptic coefficient K is obtained from the torque reference value obtained by filtering. h The following is calculated. Alternatively, as a second method, the torque reference value and the salient pole ratio L obtained in step S20 are calculated by referring to the second torque-elliptic coefficient table and the salient pole ratio-minimum coefficient value table. q / L d From the elliptic coefficient K h Perform the calculation.

[0091] In step S40, the high-frequency voltage V dh * ,V qh * This generates the elliptic coefficient K obtained in step S30. h The high-frequency voltage V obtained by applying this to equation (14) dh * ,V qh * Generates.

[0092] [Example of control results] Figure 7 is a timing chart showing an example of the control result by the motor control method of this embodiment. In particular, Figure 7(a) shows the elliptic coefficient K during power operation. h Figure 7(b) shows the control result (comparative example) when the sign of is set to positive, and the elliptic coefficient K during power operation. h The control result (example) when the sign of is set to negative is shown.

[0093] In the comparative example shown in Figure 7(a), the estimated phase error θ is obtained around time t1. γ The value ' is diverging, and the actual output torque of motor 200 is equal to the torque command value T * This is significantly different. In contrast, in the embodiment shown in Figure 7(b), the estimated phase error value θ γ No divergence was observed in ', and the actual output torque was equal to the torque command value T * It is maintained at the elliptic coefficient K during power operation. hBy setting the sign of the parameter to negative, it can be seen that the torque range in which sensorless driving is possible during power operation is expanded.

[0094] The effects and advantages of the motor control method of this embodiment, as described above, will be summarized below.

[0095] According to this embodiment, a predetermined torque command value T * Based on this, the power command value (first voltage command value V) specifies the power to be supplied to the motor 200. d1 * ,V q1 * Or the second voltage command value V d2 * ,V q2 * ) set, and the power command value is set to a high-frequency voltage V dh * ,V qh * By superimposing this, the high-frequency voltage V dh * ,V qh * Response power value (response high-frequency current i) to the input dh ,i qh Based on this, the magnetic pole position (phase estimated value θ′) of motor 200 is estimated, and the high-frequency voltage V dh * ,V qh * The superimposed power command value (final voltage command value V d * ,V q * A motor control method is provided that controls the power supplied to the motor 200 based on the estimated magnetic pole position.

[0096] In this motor control method, it is determined whether the motor 200 is in power operation or regenerative operation, and in power operation, a high-frequency voltage V dh * ,V qh * The d-axis component (d-axis high-frequency voltage V dh * ) for the q-axis component (q-axis high-frequency voltage V qh *) phase difference (dq axis phase difference φ dq ) as the phase difference for power running (phase difference for power running φ dq1 ) is set, and during regenerative operation, the phase difference between the dq axes φ dq Phase difference φ during powering dq1 A different phase difference for regeneration (phase difference φ for regeneration) dq2 Set ).

[0097] This allows a high-frequency voltage V to be superimposed on the power command value depending on whether the motor 200 is in a regenerative operation or a power operation state. dh * ,V qh * By adjusting this, the magnetic pole position of the motor 200 can be appropriately estimated. Therefore, sensorless control can be achieved in which torque fluctuations are suppressed during both powered and regenerative braking.

[0098] In particular, in this embodiment, the phase difference φ during powering is used. dq1 and the phase difference φ for regeneration dq2 Set the difference to 180°.

[0099] As a result, in both the powering and regenerative braking scenarios, the high-frequency voltage V dh * ,V qh * Response high-frequency current i dh ,i qh By ensuring that the degree of flattening f of the elliptical trajectory is above a certain level, the torque range in which sensorless driving is possible can be expanded.

[0100] More specifically, the q-axis high-frequency voltage V qh * The coefficient (elliptic coefficient K) that defines the amplitude h Set the elliptic coefficient K h The elliptic coefficient K is set to a variable value corresponding to the torque T of motor 200, and is set for both powered operation and regenerative operation. h By reversing the sign, the phase difference φ during powering is obtained. dq1 and the phase difference φ for regeneration dq2 Generate the difference.

[0101] This results in a phase difference φ during powering. dq1 and the phase difference φ for regeneration dq2 A specific calculation logic is implemented to make the difference 180°.

[0102] Furthermore, in this embodiment, a transition region is set in which the absolute torque value |T| of the motor 200 is less than or equal to a predetermined reference value A, and the elliptic coefficient K h This is defined such that it changes continuously in the transition region in response to the change in the absolute value of the torque, while its sign reverses.

[0103] This expands the torque range in which sensorless driving is possible, while also increasing the elliptic coefficient K. h To suppress the abrupt change in the phase estimate θ′ and the resulting abrupt change in torque due to the sudden change in the positive and negative values ​​of the high-frequency voltage V dh * ,V qh * It can generate [this].

[0104] In particular, the elliptic coefficient K h This is set such that, in the transition region described above, it gradually approaches 0 as the absolute torque value |T| approaches 0 (see Figure 3).

[0105] This allows for scenes where the absolute torque |T| is relatively small, i.e., the estimated phase error θ. γ The smaller the value of ′, the smaller the estimated phase error θ. γ The elliptic coefficient K is adjusted to a situation where a certain degree of narrowing of the stable region of ' is acceptable. h While minimizing the elliptic coefficient K to ensure estimation accuracy, h This enables switching between modes, which can suppress sudden changes in torque.

[0106] Furthermore, in this embodiment, a transition region is set in which the absolute torque value |T| of the motor 200 is less than or equal to a predetermined reference value A, and the elliptic coefficient K h The absolute value of is determined such that it is greater than or equal to a predetermined minimum coefficient value B in the transition region. Then, the elliptic coefficient K h The sign of is defined such that it reverses at the transition point where the torque T becomes 0 (see Figure 4).

[0107] This results in the elliptic coefficient K h Maintain the phase error estimate θ above a certain value γ While ensuring a stable region for ′, the elliptic coefficient K during the transition between powering and regeneration h The sign can be switched.

[0108] In particular, in this embodiment, the minimum coefficient value B is the salient pole ratio L of the motor 200. q / L d The larger the value, the smaller it should be (see Figure 5).

[0109] This results in the estimated phase error θ γ The minimum coefficient value B (elliptic coefficient K when switching between positive and negative) required to secure a stable region for ′ h The size of the salient pole is L. q / L d This can be determined more appropriately by also considering the correlation with the relevant stable region.

[0110] Furthermore, this embodiment provides a motor control device 100 suitable for executing the above motor control method.

[0111] This motor control device 100 includes a determination unit that determines whether the motor 200 is in power operation or regenerative operation, and a high-frequency voltage V during power operation. dh * ,V qh * The d-axis component (d-axis high-frequency voltage V dh * ) for the q-axis component (q-axis high-frequency voltage V qh * ) phase difference (dq axis phase difference φ dq ) as the phase difference for power running (phase difference for power running φ dq1 A phase difference setting unit for powering sets the phase difference φ between the dq axes during regenerative operation. dq Phase difference φ during powering dq1 A different phase difference for regeneration (phase difference φ for regeneration) dq2 It has a regenerative phase difference setting unit that sets ) and

[0112] Although embodiments of the present invention have been described above, the configurations described in the above embodiments and each of the modifications represent only a part of the application examples of the present invention and are not intended to limit the technical scope of the present invention. [Explanation of symbols]

[0113] 11 Current command generation section 12 First Voltage Command Generation Unit 13. Second Voltage Command Generation Unit 14. Final Voltage Command Generation Unit 18 Inverters 19 Phase / Rotation Speed ​​Estimation Unit 20 High-frequency voltage command generation unit 100 Motor control device 200 motor

Claims

1. A motor control method comprising: setting a power command value that defines the power to be supplied to a motor based on a predetermined torque command value; superimposing a high-frequency voltage on the power command value; estimating the magnetic pole position of the motor based on a response power value to the input of the high-frequency voltage; and controlling the power supplied to the motor based on the power command value with the high-frequency voltage superimposed and the estimated magnetic pole position, Determine whether the motor is in a powered operation or a regenerative operation. During power operation, a phase difference for power operation is set as the phase difference between the q-axis component and the d-axis component of the high-frequency voltage. During regenerative braking, a phase difference different from the phase difference used during power application is set for regenerative braking. Motor control method.

2. A motor control method according to claim 1, The difference between the phase difference for powering and the phase difference for regeneration is set to 180°. Motor control method.

3. A motor control method according to claim 2, A coefficient is set that defines the amplitude of the q-axis component, The coefficient is set to a variable value corresponding to the torque of the motor. The sign of the coefficient is reversed between the powering operation and the regenerative operation to generate the difference between the phase difference for powering and the phase difference for regenerative operation. Motor control method.

4. A motor control method according to claim 3, A transition region is set in which the absolute value of the torque of the motor is less than or equal to a predetermined reference value. The aforementioned coefficient is In the transition region, the sign is set to change continuously in response to the change in the absolute value of the torque, and to reverse. Motor control method.

5. A motor control method according to claim 4, The aforementioned coefficient is In the transition region, the torque is set to gradually approach 0 as the absolute value of the torque approaches 0. Motor control method.

6. A motor control method according to claim 3, A transition region is set in which the absolute value of the torque of the motor is less than or equal to a predetermined reference value. The absolute value of the coefficient is determined to be greater than or equal to a predetermined minimum coefficient value in the transition region. The sign of the coefficient is determined to reverse at the transition point where the torque becomes zero. Motor control method.

7. A motor control method according to claim 6, The minimum coefficient value is made smaller as the salient pole ratio of the motor increases. Motor control method.

8. A motor control device that sets a power command value that defines the power to be supplied to a motor based on a predetermined torque command value, superimposes a high-frequency voltage on the power command value, estimates the magnetic pole position of the motor based on a response power value to the input of the high-frequency voltage, and controls the power supplied to the motor based on the power command value with the high-frequency voltage superimposed and the estimated magnetic pole position, A determination unit that determines whether the motor is in a powered operation or a regenerative operation, During power operation, a power phase difference setting unit sets a power phase difference as the phase difference between the q-axis component and the d-axis component of the high-frequency voltage, During regenerative operation, the system includes a regenerative phase difference setting unit that sets a regenerative phase difference different from the powered phase difference as the phase difference. Motor control device.

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