Power converter and motor control system

Abstract

To provide a power conversion device and a motor control system capable of realizing a stable speed control even when a response speed is increased.SOLUTION: A power conversion device has a power converter 2 converting DC power into AC power based on AC voltage command values vu*, vv*, and vw* in stationary coordinates and outputting the power to an induction motor, and a controller CT calculating a voltage command value in the stationary coordinates by a vector control. The controller CT calculates a slip frequency command value ωs** by adding a transient value determined by a differential operation using d-axis current (id) and q-axis current (iq) to a stationary value of the slip frequency determined by the d-axis current (id), the q-axis current (iq), and a secondary frequency (T2).SELECTED DRAWING: Figure 10

Description

[Technical Field]

[0001] The present invention relates to power conversion devices and motor control systems, and more particularly to control technology for induction motors. [Background technology]

[0002] Patent Document 1 (Japanese Patent Publication No. 2018-182989) describes a method for driving an induction motor using speed sensorless vector control, in which the command value or detected value is coordinate-transformed to a value on a control axis (mt axis) where the direction of the primary current and the direction 90 degrees behind it are used as a rotating coordinate system, and the speed estimate of the induction motor is calculated based on the transformed value. Furthermore, in equation (6) of Patent Document 1, the current command values ​​of the d axis and q axis are I d * ,I q * Using this, the slip frequency command value ω is calculated according to the following equation (1). s * This is calculated. In equation (1), T2 is a second-order time constant, and T ACR This is the time constant corresponding to the delay in current control.

[0003]

number

[0004] [Patent Document 1] Japanese Patent Publication No. 2018-182989 [Overview of the project] [Problems that the invention aims to solve]

[0005] In the method of Patent Document 1, the speed of the induction motor is estimated using m-t coordinates different from d-q coordinates. When such a method is used, it is possible to suppress torque loss caused by the change in the primary resistance value R1 on the stator side in the induction motor due to the winding temperature, and to realize motor control with low sensitivity to the error (R1 * -R1) of the primary resistance value R1. Also, in Patent Document 1, as shown in Equation (2) of that document, the speed control operation unit calculates the current command value (I ASR ) of the q-axis using the proportional gain (Kp ASR ) and the integral gain (Ki q * ) of speed control. The proportional gain and the integral gain determine the response performance of the speed control system.

[0006] On the other hand, when controlling an induction motor, it is required to quickly adjust the speed of the induction motor to the speed command value without error. That is, high-precision and high-response speed control is required. However, in the method of Patent Document 1, if the response of the speed control system is increased beyond the value corresponding to the reciprocal of the secondary time constant in the induction motor, the speed control becomes unstable due to the change in the magnetic flux in the m-t coordinates, and there is a risk that vibrations may occur in the motor speed and torque.

[0007] The present invention has been made in view of such circumstances, and one of its objects is to provide a power conversion device and a motor control system that can achieve stable speed control even when the speed response is increased.

[0008] The above and other objects and novel features of the present invention will become apparent from the description of this specification and the accompanying drawings.

Means for Solving the Problems

[0009] The outline of a representative embodiment of the invention disclosed in the present application will be briefly described as follows.

[0010] A power conversion device according to one embodiment comprises a power converter that converts DC power to AC power and outputs it to an induction motor based on a voltage command value in stationary coordinates, and a controller that calculates the voltage command value in stationary coordinates by vector control. The controller calculates the slip frequency command value by adding a transient value determined by differential operation using the d-axis current and q-axis current to a steady value of the slip frequency determined by the d-axis current, q-axis current and second-order time constant. [Effects of the Invention]

[0011] To briefly explain the effects obtained by a representative embodiment of the invention disclosed in this application, it becomes possible to achieve stable speed control even with a high speed response. [Brief explanation of the drawing]

[0012] [Figure 1] This is a block diagram showing an example configuration of a motor control system according to Embodiment 1. [Figure 2] Figure 1 is a vector diagram illustrating the dq coordinate, which is one of the control coordinates. [Figure 3] Figure 1 is a vector diagram illustrating the mt coordinate, which is another control coordinate. [Figure 4] This block diagram shows a detailed configuration example of the frequency estimation and phase calculation unit in Figure 1. [Figure 5] This figure shows an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1. [Figure 6] This figure shows an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1. [Figure 7] This figure shows an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula according to Embodiment 1. [Figure 8]This figure shows an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula according to Embodiment 1. [Figure 9] This diagram illustrates an example of a method for observing the slip frequency in an induction motor. [Figure 10] This block diagram shows an example configuration of the power conversion device shown in Figures 1 and 4, with the main components extracted. [Figure 11] This is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in the power conversion device according to Embodiment 2, as shown in Figure 1. [Figure 12] This block diagram shows a detailed configuration example of the frequency estimation and phase calculation unit in the power conversion device according to Embodiment 3, as shown in Figure 1. [Figure 13] This is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in the power conversion device according to Embodiment 4, as shown in Figure 1. [Figure 14] This is a block diagram showing an example configuration of a motor control system according to Embodiment 5. [Figure 15] Figure 14 is a block diagram showing a detailed configuration example of the frequency detection and phase calculation unit. [Figure 16] This block diagram shows an example configuration of a motor control system according to Embodiment 6. [Figure 17] This is a block diagram showing an example configuration of a motor control system according to Embodiment 7. [Modes for carrying out the invention]

[0013] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In all drawings used to illustrate the embodiments, the same reference numerals are generally used for identical components, and repeated descriptions of such components will be omitted.

[0014] (Embodiment 1) <Outline of Motor Control System> Figure 1 is a block diagram showing an example configuration of a motor control system according to Embodiment 1. The motor control system shown in Figure 1 comprises an induction motor 1 and a power converter 20 that controls the rotation of the induction motor 1 by outputting AC power to the induction motor 1. The power converter 20 controls the rotation of the induction motor 1 using vector control, in this case, speed sensorless vector control. First, the control axes used in the vector control in Figure 1 will be explained using Figures 2 and 3.

[0015] Figure 2 is a vector diagram illustrating the dq coordinate system, one of the control coordinates in Figure 1. In Figure 2, the d-axis represents the magnetic flux direction of the induction motor 1, and the q-axis represents the direction perpendicular to the d-axis, specifically the direction 90° (π / 2) ahead of the d-axis. The power converter 20 supplies the d-axis current i to the d-axis. d And the q-axis current i that flows along the q-axis q By appropriately controlling these factors, the primary current i1 flowing through the stator of the induction motor 1 is controlled.

[0016] The primary current i1 is equal to the d-axis current i d and q-axis current i q It is determined by vector addition with the primary current i1. That is, the value of the primary current i1 is determined by the d-axis current i d The value and q-axis current i q It is determined by equation (2) using the value of . Also, the d-axis current i d The phase angle θ between the phase of the primary current i1 and the phase of the primary current i1. φ This is determined by equation (3).

[0017]

number

[0018]

number

[0019] Figure 3 is a vector diagram illustrating the mt coordinate, another control coordinate in Figure 1. The motor control system according to Embodiment 1 performs vector control using the mt coordinate in addition to the general dq coordinate, similar to the case of Patent Document 1. In Figure 3, the t-axis is the axis representing the direction of the primary current i1, and the m-axis is the axis representing the direction perpendicular to the t-axis, more specifically, the direction 90° (π / 2) behind the t-axis. The value of the primary current i1 is the m-axis current i m Value and t-axis current i t The value of is used and is determined by equation (4). In equation (4), the m-axis current i m The value of becomes zero, and the t-axis current i t The value of is equal to the value of the primary current i1. Also, the phase angle θ between the m axis and the d axis is equal to the value of the primary current i1. mt is “θ mt =(π / 2)-θ φ "That is the case."

[0020]

number

[0021] Returning to Figure 1, the induction motor 1 has a d-axis current i, which is the current of the magnetic flux axis component. d The magnetic flux generated by (φ 2d ) and the q-axis current i, which is the torque axis component current perpendicular to the magnetic flux axis. q Torque is generated accordingly. The mt coordinate is mainly used when estimating the rotational speed of the induction motor 1. The power conversion device 20 consists of a power converter 2 connected to a DC power supply 3, a current detector 4, and a voltage command value v to the power converter 2. u * , v v * , v w * It includes a controller CT that calculates by vector control.

[0022] The power converter 2 converts the DC voltage from the DC power supply 3 to a three-phase AC voltage command value v u * , v v * , v w *Based on this, it is converted to a three-phase AC voltage and output to induction motor 1. The output amplitude and output frequency of the three-phase AC voltage are determined by the voltage command value v u * , v v * , v w * It is determined based on the above. Power converter 2 is implemented, for example, by a three-phase inverter circuit that includes switching elements such as IGBTs (Insulated Gate Bipolar Transistors) and MOSFETs (Metal Oxide Semiconductor Field Effect Transistors).

[0023] The current detector 4 detects the three-phase alternating current i flowing through the induction motor 1. u i v i w It detects the three-phase current detection value i uc i vc i wc The output is given by the current detector 4, which detects the phase currents of two of the three phases in the induction motor 1, for example, the U phase and the W phase, and the AC condition, i.e., "i u +i v +i w From the relationship "=0", the line current of the V phase is given by "i v =-(i u +i w It may also be calculated using the formula "). The details of the controller CT are explained below.

[0024] The coordinate transformation unit 5 detects the three-phase current i in stationary coordinates. uc i vc i wc The phase estimate θ dc Using this, the current detection values ​​i of the two phases in the rotating coordinate system, i.e., the d-axis and q-axis. dc i qc Convert and output the phase estimate θ. dc This is an estimated value of the phase angle between the stationary coordinates and the magnetic flux axis (d-axis) that rotates relative to the stationary coordinates. The coordinate transformation unit 6 uses the current detection values ​​i of the d-axis and q-axis. dc i qc The phase estimation angle θ mtcUsing it, the current detection values i of the m-axis and the t-axis mc , i tc are converted and output. The phase estimation angle θ mtc is an estimated value of the phase angle θ mt between the m-axis and the d-axis.

[0025] The frequency estimation and phase calculation unit 7 uses the current command values i d * , i q * of the d-axis and the q-axis, the current detection values i mc , i tc of the m-axis and the t-axis, the voltage command value v mc ** of the m-axis, the secondary magnetic flux command value φ 2d * of the d-axis, and the electric circuit parameters (R1, R2’, Lσ, M, L2) of the induction motor 1 to perform calculations. Then, as the calculation result, the frequency estimation and phase calculation unit 7 outputs the speed estimated value ω r ^ of the induction motor 1, the slip frequency command value ω s ** , the output frequency command value ω1 * and the phase estimated value θ dc .

[0026] The reference phase calculation unit 8 uses the slip frequency command value ω s ** to calculate and output the estimated angle θ mt of the phase angle θ mtc between the m-axis and the d-axis according to Equation (5). In Equation (5), T2 is the secondary time constant of the induction motor 1 and is a value determined by the rotor resistance component (R2) and the self-inductance component (L2).

[0027]

Equation

[0028] The excitation current and magnetic flux setting unit 9 uses the excitation current command value with a positive polarity, that is, the current command value i d * of the d-axis, and the secondary magnetic flux command value φ 2d * of the d-axis with a positive polarity,2d * and the secondary flux command value φ of the q-axis, which is "0" 2q * are set and output. The speed control calculation unit 10 is the speed command value ω for the induction motor 1 r * and the speed estimated value ω of the induction motor 1 calculated by the frequency estimation and phase calculation unit 7 r ^ The deviation "ω r * - ω r ^ ", based on which, using proportional-integral (PI) control, the current command value i of the q-axis is calculated and output q * .

[0029] The coordinate conversion unit 11 is the current command values i of the d-axis and q-axis d * , i q * , and using the phase estimated angle θ mtc to convert and output the current command values i of the m-axis and t-axis m * , i t * . The coordinate conversion unit 12 is the secondary flux command values φ of the d-axis and q-axis 2d * , φ 2q * , and using the phase estimated angle θ mtc to convert and output the flux command values φ of the m-axis and t-axis m * , φ t * .

[0030] The mt-axis vector control calculation unit 13 is the electrical circuit parameters (R1, R2’, Lσ, M, L2) of the induction motor 1, and the flux command values φ of the m-axis and t-axis m <000013(0>, φ t * , the current command values i m * , i t * , the current detection values i mc , i tc , and the speed estimated value ω of the rotor r^ And the output frequency command value ω1 for the stator * Based on this, calculations are performed. Then, the mt-axis vector control calculation unit 13 calculates the voltage command values ​​v for the m-axis and t-axis as the calculation result. mc ** , v tc ** Outputs.

[0031] The coordinate transformation unit 14 sets the voltage command values ​​v for the m axis and t axis. mc ** , v tc ** The phase estimation angle θ mtc Using the voltage command values ​​v for the d axis and q axis dc ** , v qc ** The coordinate transformation unit 15 converts and outputs the voltage command values ​​v for the d axis and q axis in the rotation coordinate system. dc ** , v qc ** The phase estimate θ dc Using this, the voltage command value v of the three-phase AC in stationary coordinates u * , v v * , v w * Convert and output.

[0032] Next, we will explain the basic operation using speed sensorless vector control with the motor control system shown in Figure 1. First, the excitation current / magnetic flux setting unit 9 sets the secondary magnetic flux value φ of the d axis within the induction motor 1. 2d The d-axis current command value i required to generate this d * It outputs the speed command value ω. r * Velocity estimate ω r ^ The current command value i of the q axis follows according to equation (6). q * The following is calculated. In equation (6), Kp_ASR is the proportional gain of the speed control, Ki_ASR is the integral gain of the speed control, and s is a complex number.

[0033]

Number

[0034] The mt-axis vector control operation unit 13 uses the current command values i d * 、i q * obtained by coordinate transformation of the d-axis and q-axis, the current command values i m * 、i t * of the m-axis and t-axis, the electrical circuit parameters (R1, R2’, Lσ, M, L2) of the induction motor 1, the magnetic flux command values φ m * 、φ t * of the m-axis and t-axis, the speed estimated value ω r ^ and the output frequency command value ω1 * [[ID=第33]]to calculate Equation (7). As a result, the mt-axis vector control operation unit 13 calculates the voltage reference values v mc * 、v mt · *

[0035]

Number

[0036] In Equation (7), R1 * is the primary resistance value, that is, the resistance value of the stator. R2’ * is the secondary resistance value converted to the primary side, that is, the resistance value of the rotor converted to the stator side. Lσ * is the leakage inductance value. M * is the mutual inductance value. L2 * is the self-inductance value of the secondary side, that is, the rotor. T ACR is the time constant corresponding to the delay of current control, and s is a complex number.

[0037] ​Furthermore, the mt-axis vector control calculation unit 13 calculates the current detection value i of the m-axis. mc The current command value i m * By performing PI control according to equation (8) to track the m-axis voltage correction value Δv m * Similarly, the mt-axis vector control calculation unit 13 calculates the current detection value i of the t-axis. tc The current command value i t * By performing PI control according to equation (8) to track the t-axis voltage correction value Δv t * The following is calculated. In equation (8), Kp_m is the proportional gain of the current control along the m axis, and Ki_m is the integral gain of the current control along the m axis. Kp_t is the proportional gain of the current control along the t axis, and Ki_t is the integral gain of the current control along the t axis.

[0038]

number

[0039] Then, the mt-axis vector control calculation unit 13 calculates the m-axis and t-axis voltage reference values ​​v calculated in equation (7), as shown in equation (9). mc * , v mt * Furthermore, the voltage correction values ​​Δv for the m-axis and t-axis calculated using equation (8) are used. m * Δv t * By adding this, the voltage command values ​​v for the m axis and t axis are obtained. mc ** , v tc ** Calculate the voltage command values ​​v for the m axis and t axis. mc ** , v tc ** This involves coordinate transformations to the d and q axes, and coordinate transformations to the u, v, and w axes, resulting in the voltage command value v of the three-phase AC voltage. u * , v v * , v w * It will be converted.

[0040]

number

[0041] <Details of the frequency estimation and phase calculation unit> Figure 4 is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit 7 in Figure 1. The frequency estimation and phase calculation unit 7a shown in Figure 4 comprises a low-pass filter (LPF) 7a1, a slip command calculation unit 7a2, a frequency estimation calculation unit 7a3, an adder 7a4, and a phase estimation calculation unit 7a5.

[0042] The low-pass filter (LPF) 7a1 controls the current command value i in the q-axis, as shown in equation (10). q * The time constant T ACR The transfer function of the first-order lag based on "1 / (1+T)" is "1 / (1+T)". ACR By delaying it with "·s"), the delayed current command value i of the q axis q * td The time constant T is calculated. ACR As mentioned in equation (7), this is set to a value corresponding to the delay in current control. The delayed current command value i of the q-axis q * td The current command value i of the q axis q * The time constant T ACR By delaying by this amount, the current detection value i of the q axis qc This represents a pseudo-explanation.

[0043]

number

[0044] The slip command calculation unit 7a2 calculates the delayed current command value i of the q axis, as shown in equation (11). q * td And the current command value i on the d axis d *Using the second-order time constant T2 of the induction motor 1, the slip frequency command value ω s ** The slip frequency command value ω shown in equation (11) is calculated. s ** This is the slip frequency command value ω shown in equation (1). s * For this, “d / dt(tan -1 (i q * td / i d * The term "))" has been added.

[0045]

number

[0046] Here, the first term in equation (11) “i q * td / (T2i d * )” that is, the slip frequency command value ω shown in equation (1) s * This represents the steady-state value of the slip frequency command. On the other hand, the second term in equation (11) "d / dt(tan -1 (i q * td / i d * ))” represents the transient value of the slip frequency command, and the d-axis current I shown in Figures 2 and 3 d The phase angle θ between the primary current i1 and the primary current i1 φ This represents the time derivative of [the function].

[0047] The frequency estimation calculation unit 7a3 calculates the secondary magnetic flux command value φ of the d axis, as shown in equation (12). 2d * And the voltage command value v on the m axis mc ** and current detection value i mc And the current detection value i on the t-axis tc And the slip frequency command value ω s ** And, the output frequency command value ω1 *and the phase estimation angle θ mtc Using the electrical circuit parameters of the induction motor 1 and the phase estimation angle θ, the speed estimation value ω of the induction motor 1 r ^ is calculated. The electrical circuit parameters of the induction motor 1 include R1, which was also described in Equation (7). * R2’ * Lσ * M * L2 * is included. Also, T obs is the time constant of the filter included in the disturbance observer that constitutes the frequency estimation calculation unit 7a3.

[0048]

Equation

[0049] Here, in Equation (12), the frequency estimation calculation unit 7a3 calculates the speed estimation value ω based on the command value and detection value in the m-t coordinates. r ^ In this case, in “(R1 * +R2’ * )i mc ” in Equation (12), the current detection value i mc becomes approximately zero as can be seen from Figure 3. On the other hand, if the speed estimation value ω r ^ is calculated based on the command value and detection value in the d-q coordinates, the current detection value (i mc ) used instead of the current detection value i dc does not become approximately zero. Therefore, by using the m-t coordinates instead of the d-q coordinates, it becomes less sensitive to the error of the primary resistance value R1 * , and it becomes possible to calculate the speed estimation value ω r ^ with high precision.

[0050] The addition unit 7a4 adds the slip frequency command value ω r ^ to the speed estimation value ω<A0003A1> ** to obtain the output frequency command value ω1 *The phase estimation calculation unit 7a5 calculates the output frequency command value ω1 as shown in equation (14). * By integrating this, we can estimate the phase θ of the magnetic flux axis of induction motor 1. dc The phase estimate θ of this magnetic flux axis is calculated. dc Sensorless control is performed using this as the reference phase.

[0051]

number

[0052]

number

[0053] <Verification results of speed control characteristics> Figures 5 and 6 show an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1. Figures 5 and 6 show the characteristics when formula (1) is used to calculate the slip frequency command value. Also, Figure 5 shows the response angular frequency ω of the velocity control system. ASR This is the reciprocal of the second-order time constant T2 of induction motor 1, which is "1 / T2 ” The characteristics when set to (rad / s) are shown. The response angular frequency ω of the speed control system. ASR This is determined by the values ​​of the proportional gain Kp_ASR and integral gain Ki_ASR of the speed control shown in equation (6).

[0054] As shown in Figure 5, at time A, the velocity command value ω r * When the value is changed in a stepwise manner at 3Hz, the corresponding speed command value ω r * In response to the change in ω, the speed of induction motor 1 r It also changes. Velocity ω r It can be seen that the speed ω is maximum at time B, decreases at time C, but increases again up to time D. In other words, the speed ω of induction motor 1 r It can be seen that vibrations are occurring.

[0055] On the other hand, Figure 6 shows the response angular frequency ω of the speed control system. ASR This is higher than in the case of Figure 5, “5 / T2 ” The characteristics when set to (rad / s) are shown. Response angular frequency ω ASR By increasing the speed command value ω r * The speed ω of induction motor 1 in response to the change r This can improve the response. 。 However, the velocity ω in Figure 6 r In the case of Figure 5, the oscillation is significantly greater from time E, where the velocity is maximum. Furthermore, this oscillation continues for a longer period, up to time F, compared to the case of Figure 5.

[0056] Thus, when using the method described in Patent Document 1, it was difficult to improve the response of the speed control system in terms of vibration, etc. One of the reasons for this is that the magnetic flux in the dq coordinate is related to the slip frequency ω s Unlike the constant value of the slip frequency ω, the magnetic flux in mt coordinates is constant regardless of the slip frequency ω s One example is that it changes depending on the magnetic flux φ on the d-axis. Specifically, 2d It is “M×i d " and the magnetic flux φ on the q-axis 2q It is zero. On the other hand, the magnetic flux φ on the m-axis m is “ω s ×T2×φ t " and the magnetic flux φ on the t-axis t is “M×i t -ω s ×T2×φ m "That is the case."

[0057] In other words, the magnetic flux φm and φt in mt coordinates, and more specifically the secondary flux, are determined by the slip frequency ω s The response angular frequency ω of the speed control system changes over a period of time equivalent to the second-order time constant T2, and then reaches a steady state. ASR to “1 / T2 ” If the level is increased beyond a certain point, the velocity control becomes unstable during the transient state in which the secondary magnetic flux changes over a period of time equivalent to the secondary time constant T2, resulting in oscillations as shown in Figure 6.

[0058] Therefore, the slip command calculation unit 7a2 calculates the steady-state value "i q * td / (T2i d * Equation (1) consisting of )" is not the steady-state value and transient value "d / dt(tan -1 (i q * td / i d * Using equation (11) consisting of ))” the slip frequency command value ω s ** The slip command calculation unit 7a2 calculates the delayed current command value i of the q axis in the transient state. q * td and the current command value i of the d axis d * A transient value that follows the change, in this case the phase angle θ. φ Using the derivative of , the slip frequency command value ω s ** Correct it.

[0059] Figures 7 and 8 show an example of the step response characteristics of velocity when the slip frequency command value is calculated using the calculation formula according to Embodiment 1. Figures 7 and 8 show the characteristics when formula (11) is used to calculate the slip frequency command value. Also, in Figure 7, as in Figure 6, the response angular frequency ω ASR to “5 / T2 ” The characteristics when set to (rad / s) are shown, and Figure 8 shows the response angular frequency ω ASR Further enhancing the "10 / T2 ” The characteristics when set to (rad / s) are shown.

[0060] As can be seen from the comparison of the characteristics shown in Figure 6 and the characteristics shown in Figure 7, by using the method of Embodiment 1, the speed ω of the induction motor 1 r The vibration can be suppressed or prevented. Response angular frequency ω ASR Even when the speed is increased further, as shown in Figure 8, the speed ω of the induction motor 1 rThe vibration of the induction motor 1 can be suppressed or prevented. Thus, by using the method of Embodiment 1, even if the speed response is high, the speed ω r This enables stable speed control without generating vibrations.

[0061] <Method for observing slip frequency> Figure 9 illustrates an example of a method for observing the slip frequency in an induction motor. In Figure 9, a voltage detector 21 and a current detector 22 are attached to the power converter 20 that drives the induction motor 1, and an encoder 23 is attached to the shaft of the induction motor 1. A computer equipped with a voltage / current calculation unit 24 and a slip calculation unit 25 for the dq axis is also provided.

[0062] In the voltage / current calculation unit 24 of the dq axis, as the first step, the voltage detection value v of the three-phase AC detected by the voltage detector 21 is calculated. uc , v vc , v wc And the current detection value i of the three-phase AC detected by the current detector 22 uc i vc i wc The position θ detected by the encoder 23 is input. The dq axis voltage / current calculation unit 24 uses these inputs to calculate the vector voltage components v of the d axis and q axis. dc ^ , v qc ^ and vector current component i dc ^ i qc ^ The velocity detection value ω is calculated by first calculating the position θ and then differentiating the position θ. rc Calculate.

[0063] Then, the voltage / current calculation unit 24 for the dq axis performs the second step by calculating the vector voltage component v dc ^ , v qc ^ and vector current component i dc ^ i qc ^Using the electrical circuit parameters of induction motor 1, the output frequency command value ω1 is calculated using equation (15) or equation (16). ^ The following calculations are performed. The electrical circuit parameters of induction motor 1 may be, for example, the results of offline auto-tuning installed in a general-purpose inverter or design values.

[0064]

number

[0065]

number

[0066] The slip calculation unit 25 outputs the frequency command value ω1 ^ and velocity detection value ω rc Using and , the first slip frequency value ω is given by equation (17). s ^ The slip calculation unit 25 calculates the second slip frequency value ω using equation (18), which corresponds to equation (11). s ^^ The following is calculated: Here, when the power converter 20 of Embodiment 1 is used, the first slip frequency value ω s ^ This is the second slip frequency value ω s ^^ This will be consistent with that.

[0067]

number

[0068]

number

[0069] <Outline configuration of the main components of a power converter> Figure 10 is a block diagram showing an example configuration with the main parts extracted from the power converter 20 shown in Figures 1 and 4. In Figure 10, the power converter 2 receives the voltage command value v in stationary coordinates. u * , v v * , v w * Based on this, DC power is converted to AC power and output to induction motor 1. The controller CT controls the voltage command value v in stationary coordinates by vector control. u * , v v * , v w * The controller CT uses the slip command calculation unit 7a2 to calculate the d-axis current i according to equation (11). d and q-axis current i q The steady-state value of the slip frequency is determined by the second-order time constant T2, and the d-axis current i d and q-axis current i q The slip frequency command value ω is obtained by adding the transient values ​​determined by differential operations using ω. s ** Calculate.

[0070] Here, in equation (11), the d-axis current i d and q-axis current i q The current command value i on the d axis d * and the delayed current command value i of the q axis q * td While this is used, it is not limited to this; for example, the current detection value i of the d axis dc and the current detection value i of the q axis qc The following may be used. That is, ideally, the current detection value i of the d axis dc and the current detection value i of the q axis qc It is preferable to use the following for calculations. However, since the detected value may be less stable than the command value, the command value is used here. Also, the current command value i for the q axis q * The current detection value i on the q axis qc To get closer to the q-axis delayed current command value i q* td This is used.

[0071] Then, the controller CT uses the summing unit 7a4 to set the slip frequency command value ω s ** The speed value of induction motor 1, for example, the estimated speed ω r ^ By adding this, the output frequency command value ω1 * The velocity value is obtained by detection or calculation. That is, when performing velocity sensorless vector control as shown in Figure 1, the velocity value is the velocity estimate ω obtained by calculation. r ^ However, when performing vector control with a speed sensor, the speed is the speed detection value obtained through detection.

[0072] On the other hand, the controller CT uses the coordinate transformation unit 11 to determine the current command value i of the d axis. d * and the current command value i of the q axis q * By performing a coordinate transformation, the current command value i on the m axis is obtained. m * and the current command value i on the t-axis t * The controller CT uses the coordinate transformation unit 6 to calculate the current detection value i of the d axis. dc and the current detection value i of the q axis qc By performing a coordinate transformation, the current detection value i on the m axis can be obtained. mc and the current detection value i on the t-axis tc Calculate.

[0073] Then, the controller CT uses the mt-axis vector control calculation unit 13 to calculate the current command value i of the m axis. m * and the current command value i on the t-axis t * And the current detection value i on the m axis mc and the current detection value i on the t-axis tc And, the output frequency command value ω1 * Based on this, the voltage command value v on the m axis mc ** and the voltage command value v on the t-axistc ** The mt-axis vector control calculation unit 13 calculates equation (7). Then the controller CT calculates the voltage command value v of the m axis. mc ** and the voltage command value v on the t-axis tc ** The voltage command value v in stationary coordinates is obtained by sequentially transforming the coordinates using the coordinate transformation units 14 and 15. u * , v v * , v w * Calculate.

[0074] <Main effects of Embodiment 1> In the method of Embodiment 1, the slip frequency command value ω s ** By including transient values ​​in the calculation, it becomes possible to achieve stable speed control without vibrations, even with a high speed response. In other words, high-precision and highly responsive speed control becomes possible. Furthermore, by estimating the speed of the induction motor 1 using the mt coordinate, it is possible to suppress torque loss and torque pulsation caused by changes in the primary resistance value within the induction motor 1 due to winding temperature, thereby achieving high-precision and stable speed control.

[0075] (Embodiment 2) <Details of the frequency estimation and phase calculation unit> Figure 11 is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit 7 in Figure 1 in the power conversion device according to Embodiment 2. The frequency estimation and phase calculation unit 7b shown in Figure 11 differs from the configuration example shown in Figure 4 in the processing content of the slip command calculation unit 7b2. In Figure 4, the slip frequency command value ω s ** The calculation uses the current command value of the dq axis to determine the phase angle θ. φ The inverse tangent to "tan" -1 The calculation of “tan” was required, but in Figure 11, the arctangent “tan” -1 The operation " is not required.

[0076] Specifically, the slip command calculation unit 7b2 calculates the delayed current command value i of the q axis from the low-pass filter (LPF) 7a1. q * td And the current command value i on the d axis d * Using the second-order time constant T2, the slip frequency command value ω is obtained according to equation (19). s ** Calculate the arctangent “tan” in equation (11). Equation (19) is the arctangent “tan” in equation (11). -1 This is an expansion of the differential operation of " and is equivalent to equation (11).

[0077]

number

[0078] <Main effects of Embodiment 2> As described above, by using the method of Embodiment 2, the same effects as those described in Embodiment 1 can be obtained. Furthermore, without providing a calculation table or the like in the controller CT, the slip frequency command value ω s ** This allows us to calculate the inverse tangent “tan -1 In order to perform the calculation, it is usually necessary to refer to a predetermined calculation table, but in the method of Embodiment 2, it is not necessary to refer to such a calculation table.

[0079] (Embodiment 3) <Details of the frequency estimation and phase calculation unit> Figure 12 is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit 7 in the power conversion device according to Embodiment 3, as shown in Figure 1. The frequency estimation and phase calculation unit 7c shown in Figure 12 differs from the configuration example shown in Figure 4 in the processing content of the slip command calculation unit 7c2. In Figure 12, as in the case of Figure 11, the slip frequency command value ω s ** For the operation of the inverse tangent “tan -1 The operation of " is not required.

[0080] The slip command calculation unit 7c2 calculates the delayed current command value i of the q axis from the low-pass filter (LPF) 7a1. q * td And the current command value i on the d axis d * The second-order time constant T2 and the t-axis current command value i t * Using and , the slip frequency command value ω according to equation (20) s ** Calculate the following. Equation (20) is the same as “i in Equation (19) d *2 +i q * td 2 " to "i t *2 This is a replacement of " and is equivalent to equation (19).

[0081]

number

[0082] <Main effects of Embodiment 3> As described above, by using the method of Embodiment 3, the same effects as those described in Embodiments 1 and 2 can be obtained. d *2 +i q * td 2 " to "i t *2 By replacing it with ", the calculation can be simplified compared to the case of Embodiment 2.

[0083] (Embodiment 4) <Details of the frequency estimation and phase calculation unit> Figure 13 is a block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit 7 in the power conversion device according to Embodiment 4, as shown in Figure 1. The frequency estimation and phase calculation unit 7d shown in Figure 13 differs from the configuration example shown in Figure 4 in the processing content of the slip command calculation unit 7d2. In Figure 13, as in the case of Figure 11, the slip frequency command value ω s **For the operation of the inverse tangent “tan -1 The operation of " is not required.

[0084] The slip command calculation unit 7d2 calculates the delayed current command value i of the q axis from the low-pass filter (LPF) 7a1. q * td And the current command value i on the d axis d * Using the second-order time constant T2, the slip frequency command value ω is obtained according to equation (21). s ** Calculate the following. Equation (21) is the same as “tan in Equation (11)”. -1 (i q * td / i d * )” to “i q * td / i d * It is an approximation of ".

[0085]

number

[0086] <Main effects of Embodiment 4> As described above, by using the method of Embodiment 4, the same effects as those described in Embodiments 1 and 2 can be obtained. Furthermore, the calculations can be simplified compared to the case of Embodiment 3. However, from the viewpoint of calculation accuracy, it is preferable to use the methods of Embodiments 1, 2, and 3.

[0087] (Embodiment 5) <Outline of Motor Control System> Figure 14 is a block diagram showing an example configuration of a motor control system according to Embodiment 5. In Figure 1, sensorless speed control was used, but in Figure 14, control with a speed sensor is used. Therefore, in Figure 14, an encoder 1a is installed on the induction motor 1. The encoder 1a detects the phase of the induction motor 1 and outputs an encoder signal θ representing the detection result. encIt outputs the following. Also, in Figure 14, a frequency detection and phase calculation unit 7e is provided instead of the frequency estimation and phase calculation unit 7 in Figure 1. The frequency detection and phase calculation unit 7e outputs the encoder signal θ enc The speed value of the induction motor 1 is detected based on this.

[0088] <Details of the frequency estimation and phase calculation unit> Figure 15 is a block diagram showing a detailed configuration example of the frequency detection and phase calculation unit 7e shown in Figure 14. Compared to the configuration example shown in Figure 4, the frequency detection unit 7e3 is provided instead of the frequency estimation unit 7a3 in the frequency detection and phase calculation unit 7ee shown in Figure 15, and a hold circuit 7e6 is also provided. The hold circuit 7e6 receives the encoder signal θ at the current detection period. enc By holding [n], the encoder signal θ from the previous detection period is obtained. enc Output [n-1].

[0089] The frequency detection unit 7e3 detects the encoder signal θ at the current detection period. enc [n] and encoder signal θ at the previous detection period enc Using [n-1], the speed of induction motor 1 is determined according to equation (22) by the speed detection value ω rc It is detected as such. In equation (22), t smp This is the value of the velocity detection period. Velocity detection value ω rc This is the velocity estimate ω shown in Figures 1, 4, etc. r ^ It is used instead.

[0090]

number

[0091] <Main effects of Embodiment 5> As described above, by using the method of Embodiment 5, the same effects as those described in Embodiment 1 can be obtained. In other words, the same effects can be obtained not only when using speed sensorless control but also when using control with a speed sensor. It should be noted that the method of Embodiment 5 can, of course, be combined with the methods of Embodiments 2, 3, and 4.

[0092] (Embodiment 6) <Outline of Motor Control System> Figure 16 is a block diagram showing an example configuration of a motor control system according to Embodiment 6. For example, in the configuration example shown in Figure 1, a method was used in which the electrical circuit parameters of the induction motor 1 were pre-set to a fixed value in the controller CT, such as a microcontroller. On the other hand, in Figure 16, a higher-level control device, such as an IoT controller 16 implemented by a cloud computer, is provided.

[0093] The IoT controller 16 is connected to the power converter, which includes the controller CT, via a communication line and controls the power converter. Specifically, the IoT controller 16 receives control state quantities calculated within the controller CT via the communication line, calculates electrical circuit parameters using machine learning, and resets the calculated electrical circuit parameters to the controller CT via the communication line.

[0094] Figure 16 shows a simplified representation of the controller CT shown in Figure 10. The IoT controller 16 uses the voltage command values ​​v for the m axis and t axis as control state variables calculated within the controller CT. mc ** , v tc ** and current detection value i mc i tc And, the estimated velocity ω r ^ And the slip frequency command value ω s ** The inputs are used to perform machine learning on the IoT controller 1, and for example, to satisfy the state equation of the induction motor 1 corresponding to equation (7), the electrical circuit parameters (R1* , R2' * , Lσ * M * , L2 * The parameters are corrected sequentially. The IoT controller 16 then feedback controls the controller CT by resetting the corrected electrical circuit parameters to the mt axis vector control calculation unit 13, etc.

[0095] <Main effects of Embodiment 6> As described above, the same effects as those described in Embodiment 1 can be obtained by using the method of Embodiment 6. Furthermore, since machine learning makes it possible to more accurately determine the electrical circuit parameters of the induction motor 1, it becomes possible to further improve the accuracy of speed control of the induction motor 1.

[0096] (Embodiment 7) <Outline of Motor Control System> Figure 17 is a block diagram showing an example configuration of a motor control system according to Embodiment 7. In Figure 17, the induction motor 1 drives industrial equipment such as a fan, pump, or crane. The power converter 20 consists of a single housing containing a controller CT, a power converter 2, and a digital operator 20b, which is one of the user interfaces. Here, for simplicity, a configuration is shown in which the speed control calculation unit 10 shown in Figure 10 is added to the controller CT shown in Figure 10. The controller CT can be implemented by, for example, a microcontroller or a programmable logic controller.

[0097] Specifically, a controller CT is primarily implemented by a processor built into a microcontroller or programmable logic controller executing a program stored in its internal memory. However, a controller CT may also be implemented using an FPGA (Field Programmable Gate Array) or an ASIC (Application Specific Integrated Circuit).

[0098] The power converter 2 is implemented, for example, by a three-phase inverter circuit including switching elements such as MOSFETs and IGBTs. The switching elements may be Si (silicon) semiconductor elements or wide-bandgap semiconductor elements such as SiC (silicon carbide) or GaN (gallium nitride). The current detector 4 shown in Figure 1 is implemented, for example, by a combination of a resistor element or current transformer for current detection and an analog-to-digital converter that converts the detected value into a digital value.

[0099] Figure 17 also shows user terminals such as a personal computer 26, a tablet 27, and a smartphone 28. Users can set various parameters on the controller CT using such user terminals or using the digital operator 20b of the power converter 20. For example, the power converter 20 is connected directly to such user terminals via a local area network or the like, or indirectly via an IoT controller 16 as shown in Figure 16.

[0100] The user can set control parameters and electrical circuit parameters of the induction motor 1 to the controller CT via the user terminal and local area network. In this example, the user can set the response angular frequency ω of the speed control system to the speed control calculation unit 10 in the controller CT via the user terminal. ASR That is, the proportional gain Kp_ASR and integral gain Ki_ASR of the speed control shown in equation (6), and the speed command value ω for the induction motor 1. r * These are some of the instructions given.

[0101] Note that Figure 17 shows a configuration based on the method of Embodiment 1, but a configuration based on any of the methods of Embodiments 2 to 6 may also be used. Furthermore, in Embodiments 1 to 6, as shown in equations (11) and (19) to (21), the slip frequency command value ω s ** To calculate this, the current command value i on the d axis is used. d* and the delayed current command value i of the q axis q * td The current detection values ​​i for the d axis and q axis were used, but dc i qc This may also be used.

[0102] Furthermore, in embodiments 1 to 6, as shown in equation (8), the current command values ​​i for the m axis and t axis are m * i t * And, current detection value i mc i tc Using and the voltage correction value Δv m * Δv t * The following was calculated. Then, as shown in equation (9), the voltage reference value v mc * , v tc * Voltage correction value Δv m * Δv t * By adding this, the voltage command values ​​v for the m axis and t axis are obtained. mc ** , v tc ** The result was calculated.

[0103] Instead, the current command values ​​i for the m axis and t axis m * i t * And, current detection value i mc i tc Using and according to equation (23), the voltage command values ​​v for the m axis and t axis are calculated. mc *** , v tc *** The following can also be calculated. In equation (23), Kp_m1 is the proportional gain of the current control along the m axis, and Ki_m1 is the integral gain of the current control along the m axis. Kp_t1 is the proportional gain of the current control along the t axis, and Ki_t1 is the integral gain of the current control along the t axis.

[0104]

number

[0105] Alternatively, the current command values ​​i for the m axis and t axis m * i t * And, current detection value i mc i tc Using and according to equation (24), the intermediate current command value i for the m axis and t axis is obtained. m ** i t ** You may also calculate the following. In equation (24), Kp_m2 is the proportional gain of the current control along the m axis, and Ki_m2 is the integral gain of the current control along the m axis. Kp_t2 is the proportional gain of the current control along the t axis, and Ki_t2 is the integral gain of the current control along the t axis.

[0106]

number

[0107] Using equation (24), we can further estimate the velocity ω r ^ And, the output frequency command value ω1 * And the magnetic flux command values ​​φ for the m axis and t axis. m * , φ t * And the electrical circuit parameters of induction motor 1 (R * , Lσ * M * , L2 * Using ), the voltage command values ​​v for the m axis and t axis are calculated according to equation (25). mc **** , v tc **** This is calculated. In equation (25), R * Refer to equation (7) for “R1 * +R2' * " and T ACR This is the time constant corresponding to the delay in current control.

[0108]

number

[0109] Alternatively, the current command values ​​i for the m axis and t axis m * i t * And, current detection value i mc i tc Using and according to equation (26), the voltage correction value Δv of the proportional component of the m axis is obtained. mp * and the voltage correction value Δv of the integral calculation component. mi * And the voltage correction value Δv of the proportional calculation component on the t-axis. tp * and the voltage correction value Δv of the integral calculation component. ti * You may also calculate the following. In equation (26), Kp_m3 is the proportional gain of the current control along the m axis, and Ki_m3 is the integral gain of the current control along the m axis. Kp_t3 is the proportional gain of the current control along the t axis, and Ki_t3 is the integral gain of the current control along the t axis.

[0110]

number

[0111] Using equation (26), we can further estimate the velocity ω r ^ And, the output frequency command value ω1 * And the magnetic flux command values ​​φ for the m axis and t axis. m * , φ t * Using the electrical circuit parameters of induction motor 1, the voltage command values ​​v for the m axis and t axis are calculated according to equation (27). mc ***** , v tc ***** The following is calculated. In equation (27), the electrical circuit parameters are the same as in the case of equation (25).

[0112]

number

[0113] <Main effects of Embodiment 7> As described above, the same effects as those described in Embodiment 1, etc., can be obtained by using the method of Embodiment 7. In particular, the response angular frequency ω of the speed control system can be changed by the user depending on the application of the induction motor 1. ASR Even if the parameters are changed, it becomes possible to achieve stable speed control without vibrations or other issues.

[0114] The present invention has been described in detail above based on embodiments, but the present invention is not limited to the embodiments described above and can be modified in various ways without departing from its essence. For example, the embodiments described above are described in detail in order to explain the present invention in an easy-to-understand manner and are not necessarily limited to those having all the described configurations. Furthermore, it is possible to replace a part of the configuration of one embodiment with the configuration of another embodiment, and it is also possible to add a configuration from another embodiment to the configuration of one embodiment. In addition, it is possible to add, delete, or replace a part of the configuration of each embodiment with a configuration from another embodiment. [Explanation of Symbols]

[0115] 1...Induction motor, 2...Power converter, 3...DC power supply, 4...Current detector, 5...Coordinate transformation unit, 6...Coordinate transformation unit, 7, 7a~7d...Frequency estimation and phase calculation unit, 7a1...Low-pass filter, 7a2~7d2...Slip command calculation unit, 7e...Frequency detection and phase calculation unit, 8...Reference phase calculation unit, 9...Excitation current / magnetic flux setting unit, 10...Speed ​​control calculation unit, 11...Coordinate transformation unit, 12...Coordinate transformation unit, 13...MT axis vector control calculation unit, 14...Coordinate transformation unit, 15...Coordinate transformation unit, 16...IoT controller, 20...Power converter, 23...Encoder, CT...Controller, i d * ...d-axis current command value, i q * ...current command value for the q axis, i q * td ...q-axis delayed current command value, i dc ...d-axis current detection value, iqc ...q-axis current detection value, v dc ** ...d-axis voltage command value, v qc ** ...q-axis voltage command value, i m * ...m-axis current command value, i t * ...current command value on the t-axis, i mc ...m-axis current detection value, i tc ...current detection value on the t-axis, v mc ** ...m-axis voltage command value, v tc ** ...voltage command value on the t-axis, ω r * ...Speed ​​command value, ω r ^ ...estimated velocity, ω rc ...speed detection value, ω s ** ...Slip frequency command value, ω1 * ...output frequency command value, ω ASR ...Response angular frequency of the speed control system, v u * , v v * , v w * ...Three-phase AC voltage command value, i uc i vc i wc ...Three-phase current detection value

Claims

1. A power converter that converts DC power to AC power based on the voltage command value in stationary coordinates and outputs it to an induction motor, A controller that calculates the voltage command value at the stationary coordinates by vector control, Equipped with, The aforementioned controller, Using a d-q coordinate system, which is a rotational coordinate system where the d-axis represents the magnetic flux direction of the induction motor and the q-axis represents the direction perpendicular to the d-axis, the slip frequency command value is calculated by adding a transient value determined by differential calculation using the d-axis current and the q-axis current to the steady-state value of the slip frequency determined by the d-axis current, the q-axis current and the second-order time constant. The output frequency command value is calculated by adding the detected or calculated speed value of the induction motor to the slip frequency command value. Using an m-t coordinate system where the t-axis represents the direction of the primary current and the m-axis represents the direction perpendicular to the t-axis, the current command value of the m-axis and the current command value of the t-axis are calculated by performing a coordinate transformation on the current command value of the d-axis and the current command value of the q-axis. The current detection value of the m-axis and the current detection value of the t-axis are calculated by performing a coordinate transformation on the current detection value of the d-axis and the current detection value of the q-axis. Based on the current command value of the m axis and the current command value of the t axis, the detected current value of the m axis and the detected current value of the t axis, and the output frequency command value, the voltage command value of the m axis and the voltage command value of the t axis are calculated, and the voltage command value in the stationary coordinates is calculated by performing a coordinate transformation on the voltage command value of the m axis and the voltage command value of the t axis. Power converter.

2. In the power conversion device according to claim 1, The controller calculates the delayed current command value of the q-axis by delaying the current command value of the q-axis using a low-pass filter, and sets the current command value of the d-axis to "i ｄ ＊ ", the delayed current command value of the q-axis is "i ｑ ＊ ｔｄ "The transient value in the slip frequency command value is "d / dt(tan －１ (i ｑ ＊ ｔｄ / i ｄ ＊ ))” Calculated by, Power converter.

3. In the power conversion device according to claim 1, The controller calculates the delayed current command value of the q-axis by delaying the current command value of the q-axis using a low-pass filter, and sets the current command value of the d-axis to "i ｄ ＊ ", the delayed current command value of the q-axis is "i ｑ ＊ ｔｄ "The transient value in the slip frequency command value is "(i ｄ ＊２ / (i ｄ ＊２ +i ｑ ＊ ｔｄ ２ ))×d / dt(i ｑ ＊ ｔｄ / i ｄ ＊ )” Calculated by, Power converter.

4. In the power conversion device according to claim 1, The controller calculates the delayed current command value of the q-axis by delaying the current command value of the q-axis using a low-pass filter, and sets the current command value of the d-axis to "i ｄ ＊ ", the delayed current command value of the q-axis is "i ｑ ＊ ｔｄ ", the current command value of the t-axis is "i ｔ ＊ "The transient value in the slip frequency command value is "(i ｄ ＊２ / i ｔ ＊２ )×d / dt(i ｑ ＊ ｔｄ / i ｄ ＊ )” Calculated by, Power converter.

5. In the power conversion device according to claim 1, The controller calculates the delayed current command value of the q-axis by delaying the current command value of the q-axis using a low-pass filter, and sets the current command value of the d-axis to "i ｄ ＊ ", the delayed current command value of the q-axis is "i ｑ ＊ ｔｄ "The transient value in the slip frequency command value is "d / dt(i ｑ ＊ ｔｄ / i ｄ ＊ )” Calculated by, Power converter.

6. In the power conversion device according to claim 1, The controller calculates the speed value of the induction motor using the current detection value of the m axis and the current detection value of the t axis. Power converter.

7. In the power conversion device according to claim 1, The induction motor is equipped with an encoder for detecting the phase of the induction motor. The controller detects the speed value of the induction motor based on the output signal of the encoder. Power converter.

8. In the power conversion device according to any one of claims 1 to 7, The controller is capable of variably setting the response frequency of the speed control system according to instructions from the user. Power converter.

9. A power conversion device comprising: a power converter that converts DC power to AC power and outputs it to an induction motor based on a voltage command value in stationary coordinates; and a controller that calculates the voltage command value in stationary coordinates by vector control. A higher-level control device that controls the power conversion device, Equipped with, The aforementioned controller, Using a d-q coordinate system, which is a rotational coordinate system where the d-axis represents the magnetic flux direction of the induction motor and the q-axis represents the direction perpendicular to the d-axis, the slip frequency command value is calculated by adding a transient value determined by differential calculation using the d-axis current and the q-axis current to the steady-state value of the slip frequency determined by the d-axis current, the q-axis current and the second-order time constant. The output frequency command value is calculated by adding the detected or calculated speed value of the induction motor to the slip frequency command value. Using an m-t coordinate system where the t-axis represents the direction of the primary current and the m-axis represents the direction perpendicular to the t-axis, the current command value of the m-axis and the current command value of the t-axis are calculated by performing a coordinate transformation on the current command value of the d-axis and the current command value of the q-axis. The current detection value of the m-axis and the current detection value of the t-axis are calculated by performing a coordinate transformation on the current detection value of the d-axis and the current detection value of the q-axis. Based on the current command value of the m axis and the current command value of the t axis, the detected current value of the m axis and the detected current value of the t axis, and the output frequency command value, the voltage command value of the m axis and the voltage command value of the t axis are calculated, and the voltage command value in the stationary coordinates is calculated by performing a coordinate transformation on the voltage command value of the m axis and the voltage command value of the t axis. The higher-level control device receives at least the voltage command value of the m-axis and the voltage command value of the t-axis calculated by the controller, and the current detection value of the m-axis and the current detection value of the t-axis, and modifies the electrical circuit parameters of the induction motor used in calculations within the controller by machine learning using the input values, and resets the modified electrical circuit parameters to the controller. Motor control system.

10. In the motor control system according to claim 9, The controller calculates the delayed current command value of the q-axis by delaying the current command value of the q-axis using a low-pass filter, and sets the current command value of the d-axis to "i ｄ ＊ ", the delayed current command value of the q-axis is "i ｑ ＊ ｔｄ "The transient value in the slip frequency command value is "d / dt(tan －１ (i ｑ ＊ ｔｄ / i ｄ ＊ ))” Calculated by, Motor control system.

11. In the motor control system according to claim 9 or 10, The controller is capable of variably setting the response frequency of the speed control system according to instructions from the user. Motor control system.

Images