Signal processing method and link, and target detection method and electronic device

Abstract

Disclosed in the present application are a signal processing method and link, and a target detection method and an electronic device. For an FMCW radar having at least two receiving channels and unequal-length lines, the unequal-length receiving of a digital baseband signal is compensated for in a digital domain according to a length difference between the receiving channels, such that the problem of the generation of a relative time delay between different receiving channels which is inevitably caused by a line length difference, is eliminated, thereby improving the signal receiving quality, and further improving the receiving performance.

Description

Signal processing method and link, target detection method, electronic equipment

[0001] This application claims the priority of the Chinese patent application No. 202311830908.0 filed with the China Patent Office on December 27, 2023, entitled “Signal Transmission Link and Method, Integrated Circuit, Electromagnetic Wave Device and Equipment”, the priority of the Chinese patent application No. 202311828622.9 filed with the China Patent Office on December 27, 2023, entitled “Signal Transceiver Link and Method, Integrated Circuit, Electromagnetic Wave Device and Equipment”, the priority of the Chinese patent application No. 202311791286.5 filed with the China Patent Office on December 22, 2023, entitled “Received Signal Compensation Method, Device, Integrated Circuit and Radio Device”, and the priority of the Chinese patent application No. 202311791286.5 filed with the China Patent Office on June 14, 2023. The present application claims priority from the Chinese patent application No. 202310702586.5 filed with the Patent Office of China on December 29, 2023, with application number 202311870043.0, with invention name “Target detection method, electronic device and storage medium”; and the Chinese patent application No. 202410586691.1 filed with the Patent Office of China on May 11, 2024, with invention number 202410586691.1, with invention name “FMCW reception leakage processing method, user terminal equipment and storage medium”, the contents of which should be understood as incorporated into this application by reference. Technical Field

[0002] The present application relates to, but is not limited to, radar technology, and in particular to a signal processing method and link, a target detection method, and electronic equipment. Background Art

[0003] To improve radar measurement accuracy, Frequency Modulated Continuous Wave (FMCW) radar systems often use multi-antenna transceiver technology. However, in order to achieve better isolation between the transmit and receive antennas, designing identical antenna feeder lengths for different antennas is becoming increasingly difficult.

[0004] Unequal-length antenna feeders can address the aforementioned antenna design challenges. They can achieve a better link budget, simpler antenna design and routing, lower inter-antenna coupling, better target resolution, and a smaller module size and cost. However, differences in antenna feeder length between transmit and receive channels inevitably lead to relative latency in different receive channels. This inconsistency in transmit and receive antenna feeder lengths reduces transmit and receive signal quality, leading to poor transmit and receive performance.

[0005] SUMMARY OF THE INVENTION

[0006] To solve the above technical problems, an embodiment of the present application provides a signal processing method that can be applied to an FMCW radar having at least two receiving channels with unequal line lengths, wherein the at least two receiving channels include a reference receiving channel and at least one other receiving channel that has a length difference compared to the reference receiving channel. For any of the receiving channels, the method includes: processing the echo signal received by the receiving channel to obtain a digital baseband signal; and compensating the digital baseband signal obtained after processing based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitted signal corresponding to the received echo signal.

[0007] An embodiment of the present application provides a computer-readable storage medium storing computer-executable instructions, wherein the computer-executable instructions are used to execute a signal processing method described in any embodiment of the present application.

[0008] An embodiment of the present application provides a device for implementing signal processing, which may include a memory and a processor, wherein the memory stores the following instructions that can be executed by the processor: for executing the steps of a signal processing method described in any embodiment of the present application.

[0009] An embodiment of the present application also provides a signal processing link that can be applied to an FMCW radar having at least two receiving channels and unequal line lengths, including: a processing module and a compensation module; the processing module is used to process the echo signal received by any receiving channel to obtain a digital baseband signal; the at least two receiving channels include a reference receiving channel and at least one other receiving channel that has a length difference compared to the reference receiving channel; the compensation module is used to compensate the digital baseband signal obtained after processing based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitted signal corresponding to the received echo signal.

[0010] An embodiment of the present application provides a signal processing link, comprising: a waveform generator and a plurality of transmission channels connected to the waveform generator, wherein at least two of the transmission channels have different signal transmission delays, and at least one of the transmission channels includes a direct digital frequency synthesizer;

[0011] The direct digital frequency synthesizer is configured to generate a compensation signal according to a delay difference of a signal transmission delay of the corresponding transmission channel relative to a reference transmission delay, and use the compensation signal to compensate for the transmission signal of the corresponding transmission channel.

[0012] An embodiment of the present application provides another signal processing method, which is applied to an antenna array of an electromagnetic wave device having at least two signal transmission chains, wherein the signal transmission chain is a phase shifter including a digital phase shift architecture. The signal transmission method includes: determining a reference transmission delay and a delay difference between the signal transmission delay of each transmission channel and the reference transmission delay; generating a compensation signal corresponding to each transmission channel based on the delay difference; and using the compensation signal to compensate the transmission signal of the corresponding transmission channel.

[0013] An embodiment of the present application provides a signal processing link, including: a signal transmission link and a signal receiving link, wherein: the signal transmission link includes multiple transmission channels implemented based on analog circuits, each transmission channel includes its own transmitting antenna, at least two transmission channels have different signal transmission delays, and the multiple transmitting antennas transmit electromagnetic wave signals in a time-division multiplexing manner; the signal receiving link includes at least one receiving channel, and the receiving channel includes a signal compensator implemented using a digital phase-shifting architecture; the signal compensator is configured to determine the transmission channel corresponding to the currently received echo signal, generate a compensation signal based on the delay difference of the signal transmission delay of the corresponding transmission channel relative to a reference transmission delay, and use the compensation signal to compensate for the currently received echo signal.

[0014] An embodiment of the present application provides another signal processing method, which is applied to an antenna array of an electromagnetic wave device having at least two signal transmission links. The signal transmission and reception method includes: determining a reference transmission delay and a delay difference between the signal transmission delay of each transmission channel and the reference transmission delay; generating a compensation signal corresponding to each transmission channel based on the delay difference; determining the transmission channel corresponding to the currently received echo signal, and using the corresponding compensation signal to compensate for the intermediate frequency phase difference and / or frequency difference of the currently received echo signal.

[0015] An embodiment of the present application provides a target detection method, which is applied to a DDM radar sensor having at least one transceiver channel. The method includes: performing a range-dimensional Fourier transform on an echo signal received by a receiving channel to obtain range-dimensional FFT data, and using a phase shift error coefficient of a transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain range-dimensional FFT compensated data; performing a Doppler-dimensional Fourier transform on the range-dimensional FFT compensated data to obtain Doppler-dimensional FFT data; and determining velocity information of a target object based on the Doppler-dimensional FFT data.

[0016] The embodiment of the present application provides another signal processing link, including a transmitting link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal; the transmitting link includes an analog signal source and a digital phase shifter, the analog signal source can be configured to provide an initial analog signal, the digital phase shifter can be configured to generate a digital phase-shifted signal, and phase-shift the initial analog signal based on the digital phase-shifted signal to perform a preset phase-shift operation on the initial analog signal; the receiving link includes an analog-to-digital converter and a digital baseband processing module, the analog-to-digital converter can be configured to use The received echo signal is subjected to analog-to-digital conversion to obtain a digital baseband signal, and the digital baseband signal processing module can be configured to sequentially perform distance-dimensional Fourier transform and velocity-dimensional Fourier transform on the digital baseband signal; wherein the digital baseband processing module includes a phase shift compensation unit, and the phase shift compensation unit can be configured to adopt the phase shift error coefficient of the digital phase shifter in the transmission link to obtain the distance-dimensional FFT data phase shift compensation by performing the distance-dimensional Fourier transform; and the digital baseband processing module can be configured to perform the velocity-dimensional Fourier transform based on the compensated distance-dimensional FFT data.

[0017] An embodiment of the present application provides a method for processing FMCW reception leakage, comprising: in a target-free scenario, performing a range-dimensional Fourier transform on a received signal to obtain range-dimensional FFT data; and extracting, for each chirp signal in the range-dimensional FFT data, the first k data of the chirp signal sorted from low to high frequency; obtaining a leakage value for each data item of the same rank in each group of the first k data items extracted from different chirp signals; wherein, the data items of the same rank in each group of the first k data items also have the same corresponding frequency; and, during target detection, using the leakage value for each rank, performing subtraction processing on the data items of the corresponding rank of each chirp signal in the range-dimensional FFT data generated by the currently received signal, thereby eliminating leakage from the received signal.

[0018] An embodiment of the present application provides another method for processing FMCW reception leakage, including: in a targetless scenario, performing a range-dimensional Fourier transform on a received signal to obtain range-dimensional FFT data; and extracting the first k data of each chirp signal in the range-dimensional FFT data; and obtaining leakage values ​​for each data item of the same sequence in each group of the first k data items extracted from different chirp signals; wherein, the data items of the same sequence in each group of the first k data items also have the same corresponding frequency.

[0019] An embodiment of the present application provides another method for processing FMCW reception leakage, including: in a target-free scenario, performing a range-dimensional Fourier transform on a received signal to obtain range-dimensional FFT data; based on a frame of the range-dimensional FFT data, for multiple predetermined range bins, taking the average value between different chirp signals as the first leakage value for each of the predetermined range bins.

[0020] An embodiment of the present application provides another method for processing FMCW reception leakage, comprising: retrieving leakage values ​​for each rank of multiple data items in a chirp signal sorted from low to high frequency from pre-stored data obtained based on a range-dimensional transform; and, during target detection, performing subtraction processing on multiple data items in each chirp signal sorted from low to high frequency in the range-dimensional FFT data generated by the currently received signal based on the retrieved leakage values ​​for each rank, thereby eliminating leakage from the received signal.

[0021] An embodiment of the present application provides a signal processing link for use in an electromagnetic wave sensor; the transmission link includes: an analog signal source and a digital phase shifter; wherein the analog signal source is configured to provide an initial analog signal; the digital phase shifter is configured to generate a phase-shifted signal in the digital domain and phase-shift the initial analog signal based on the phase-shifted signal to perform a preset phase-shift operation on the initial analog signal.

[0022] An embodiment of the present application provides another signal processing link, including any of the signal transmission links described above, and a signal receiving link; the signal receiving link includes a real mixer, a real analog-to-digital converter and a digital signal processing module; wherein the real mixer is configured to down-convert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal; the echo signal is a signal formed by the signal transmitted by the signal transmission link being reflected and / or scattered by the target object; the real analog-to-digital converter is configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal; the digital signal processing module is configured to process the digital intermediate frequency signal to obtain target parameters.

[0023] An embodiment of the present application provides a signal calibration link, comprising the above-mentioned another signal processing link; the receiving antenna connection port of the signal receiving link in the signal processing link is connected to the transmitting antenna connection port of the signal transmitting link in the signal processing link; the signal receiving link is also configured to calibrate the signal transmitting link.

[0024] An embodiment of the present application provides another signal calibration link, including the above-mentioned another signal processing link and a BIST module; the receiving antenna connection port of the signal receiving link in the signal processing link is connected to the transmitting antenna connection port of the signal transmitting link in the signal processing link through the BIST module; the signal receiving link in the signal processing link is also configured to calibrate the signal transmitting link.

[0025] An embodiment of the present application provides another signal calibration link, comprising two signal receiving links, a BIST module, an auxiliary circuit unit and any one of the above-mentioned signal processing links, and; any one of the signal receiving links comprises a real mixer, a real analog-to-digital converter and a digital signal processing module; the real mixer is configured to down-convert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal, and the echo signal is a signal formed by the signal transmitted by the signal transmission link being reflected and / or scattered by the target object; the real analog-to-digital converter is configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal; the digital signal processing module is configured to process the digital intermediate frequency signal to obtain target parameters; the receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection port of the signal processing link through the auxiliary circuit unit and the BIST module in sequence, and the signal receiving link is configured to calibrate the intermediate frequency part of the signal transmission link.

[0026] An embodiment of the present application provides a signal compensation link, comprising any one of the above-mentioned signal processing links and a compensation module; the compensation module is configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion defects of the signal processing link.

[0027] An embodiment of the present application provides a method for compensating for unequal feeder lengths, which is applied to an antenna array of an electromagnetic wave sensor having at least two signal links. The method includes: using the one with the shortest feeder among the at least two signal links as a reference link, obtaining the delay difference of each remaining transmission link relative to the reference link; and compensating for the unequal feeder lengths of the antenna array in the digital domain based on the delay difference.

[0028] An embodiment of the present application provides a signal calibration system, which is applied to an electromagnetic wave sensor. The signal calibration system includes a signal transmission link and an auxiliary link. The auxiliary link is integrated into the electromagnetic wave sensor adjacent to the signal transmission link. The auxiliary link is configured to perform real-time calibration on the signal transmission link.

[0029] An embodiment of the present application provides an IQ mixer, comprising an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit; the I-branch mixing unit is configured to output an I-branch signal; the Q-branch mixing unit is configured to output a Q-branch signal; and the transformer unit is configured to magnetically couple the I-branch signal and the Q-branch signal to synthesize the IQ mixing output signal.

[0030] An embodiment of the present application provides an integrated circuit, which may include: a signal transceiver channel, for transmitting a radio signal, and receiving an echo signal formed by the radio signal being reflected by a target object; and a signal processing link as described in any embodiment of the present application, for compensating the digital baseband signal obtained after processing; and / or a signal calibration and / or compensation link as described in any embodiment of the present application, for performing calibration and / or compensation operations on the signal transceiver / transmitter link.

[0031] An embodiment of the present application also provides a radio device, which may include: a carrier; an integrated circuit described in any embodiment of the present application, disposed on the carrier; and an antenna, disposed on the carrier, for transmitting and receiving radio signals.

[0032] An embodiment of the present application provides a terminal device, which may include: a device body; a radio device as described in any embodiment of the present application arranged on the device body, wherein the radio device is used for target detection and / or communication.

[0033] Other features and advantages of the present application will be described in the following description, and in part will become apparent from the description, or will be understood by practicing the present application. The purposes and other advantages of the present application can be achieved and obtained through the structures particularly pointed out in the description, claims and drawings.

[0034] Summary of the Figures

[0035] The accompanying drawings are used to provide a further understanding of the technical solution of the present application and constitute a part of the specification. Together with the embodiments of the present application, they are used to explain the technical solution of the present application and do not constitute a limitation on the technical solution of the present application.

[0036] FIG1 is a schematic diagram of a transceiver link in an embodiment of the present application;

[0037] FIG2 is a schematic diagram of antenna feed lines in a multi-receiving antenna according to an embodiment of the present application;

[0038] FIG3 is a flow chart of a signal processing method according to an embodiment of the present application;

[0039] FIG4 is a schematic flow chart of an embodiment of a received signal compensation method according to an embodiment of the present application;

[0040] FIG5 is a schematic diagram of the structure of a signal processing link in an embodiment of the present application;

[0041] FIG6 is a schematic diagram of the structure of an embodiment of a received signal compensation device according to an embodiment of the present application;

[0042] FIG7 is a schematic diagram of a first circuit embodiment of a compensation module in an embodiment of the present application;

[0043] FIG8 is a schematic diagram of a second circuit embodiment of a compensation module in an embodiment of the present application;

[0044] FIG9 is a schematic structural diagram of a signal transmission link provided in an embodiment of the present application;

[0045] FIG10 is a schematic diagram of an FMCW transmission signal using sawtooth wave modulation;

[0046] 11A and 11B are schematic diagrams of FMCW transmission signals and their echo signals from two different transmitting antennas;

[0047] FIG12A is a schematic structural diagram of a transmission channel according to an embodiment of the present application;

[0048] FIG12B is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture according to an embodiment of the present application;

[0049] FIG12C is a schematic diagram of a digital phase shifter architecture in the signal transmission link shown in FIG4B ;

[0050] FIG12D is a schematic diagram of a transmission link including a specific compensation module according to an embodiment of the present application;

[0051] FIG12E is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present application;

[0052] FIG13 is a flow chart of another signal processing method according to an embodiment of the present application;

[0053] FIG14A is a schematic structural diagram of another signal transmission link according to an embodiment of the present application;

[0054] FIG14B is a schematic diagram of the structure of another signal receiving link according to an embodiment of the present application;

[0055] FIG15A is a schematic diagram of an FMCW transmission signal using sawtooth wave modulation;

[0056] FIG15B is a schematic diagram of signals from two transmitting antennas and four receiving antennas;

[0057] FIG16 is a schematic structural diagram of a receiving channel in an embodiment of the present application;

[0058] FIG17 is a flow chart of another signal processing method according to an embodiment of the present application;

[0059] FIG18 is a schematic diagram of a flow chart of a target detection method in an embodiment of the present application;

[0060] FIG19a is a schematic diagram of chirp stepped phase modulation of a transmitting antenna in an embodiment of the present application;

[0061] FIG19b is a 2D FFT plane diagram corresponding to FIG19a;

[0062] FIG20 a is a schematic diagram of a process of performing range-dimensional FFT processing on a single receiving channel in an embodiment of the present application;

[0063] FIG20 b is a 1D FFT data plane diagram obtained by the processing of FIG20 a;

[0064] FIG21 a is a first schematic diagram of a process of performing phase shift compensation on 1D FFT data of the Mth range gate in an embodiment of the present application;

[0065] FIG21 b is a second schematic diagram of a process of performing phase shift compensation on 1D FFT data of the Mth range gate in an embodiment of the present application;

[0066] FIG22 is a schematic diagram of the process of multi-receiving channel combined detection and virtual array composite angle solution in an embodiment of the present application;

[0067] FIG23 is a schematic diagram of the structure of an integrated circuit according to an embodiment of the present application;

[0068] FIG24 is a schematic diagram of a model signal of LO leakage in an embodiment of the present application;

[0069] FIG25 is a flowchart of a method for processing FMCW reception leakage according to an embodiment of the present application.

[0070] FIG26 is a second flowchart of a method for processing FMCW reception leakage according to an embodiment of the present application:

[0071] FIG27 is a flowchart of a third method for processing FMCW reception leakage according to an embodiment of the present application:

[0072] FIG28 is a fourth flowchart of a method for processing FMCW reception leakage according to an embodiment of the present application;

[0073] FIG29 is a simplified schematic diagram of a signal processing link for an analog phase shifter architecture;

[0074] FIG30 is a simplified schematic diagram of an analog phase shifter in the signal processing chain shown in FIG29;

[0075] FIG31 is a schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod, and LO Freq Diff according to an embodiment of the present application;

[0076] FIG32 is a schematic diagram of a transceiver link based on the structure shown in FIG7 in combination with BIST according to an embodiment of the present application;

[0077] FIG33 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod, and RX IQ De-Mod according to an embodiment of the present application.

[0078] FIG34 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod according to an embodiment of the present application;

[0079] FIG35 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod according to an embodiment of the present application;

[0080] FIG36 is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module according to an embodiment of the present application;

[0081] FIG37 is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module according to an embodiment of the present application;

[0082] FIG38 is a schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture according to an embodiment of the present application;

[0083] FIG39 is a schematic diagram of a transmission link having at least two transmission channels according to an embodiment of the present application;

[0084] FIG40 is a schematic diagram of the structure of a digital LO signal generator according to an embodiment of the present application;

[0085] FIG41 is a schematic structural diagram of a feeder unequal length compensation module according to an embodiment of the present application;

[0086] FIG42 is a schematic diagram of calibrating and compensating a transceiver link using an auxiliary circuit in an embodiment of the present application;

[0087] FIG43 is a schematic diagram of calibrating and compensating a receiving link using an auxiliary transmitting circuit in an embodiment of the present application;

[0088] FIG44 is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present application;

[0089] FIG45 is a schematic structural diagram of an auxiliary circuit according to an embodiment of the present application;

[0090] FIG46 is a schematic structural diagram of another auxiliary circuit in an embodiment of the present application;

[0091] FIG47 is a schematic diagram of a circuit module of an IQ Mixer according to an embodiment of the present application;

[0092] FIG48 is a schematic diagram of the structure of an IQ Mixer according to an embodiment of the present application;

[0093] FIG49 is a schematic diagram corresponding to the structure shown in FIG48;

[0094] FIG50 is a schematic diagram of the physical structure of another IQ Mixer in an embodiment of the present application. DETAILED DESCRIPTION

[0095] To make the purpose, technical solutions and advantages of the present application clearer, the embodiments of the present application will be described in detail below with reference to the accompanying drawings. It should be noted that, unless there is a conflict, the embodiments in the present application and the features in the embodiments can be combined with each other in any manner. To facilitate understanding of the present application, the present application will be described more comprehensively below with reference to the relevant drawings. The accompanying drawings provide embodiments of the present application. However, the present application can be implemented in many different forms and is not limited to the embodiments described herein. On the contrary, the purpose of providing these embodiments is to make the disclosure of the present application more thorough and comprehensive.

[0096] The electromagnetic wave emitted by the radar's transmitting antenna is a high-frequency continuous wave, whose frequency varies regularly over time. The waveform of a high-frequency continuous wave can be sawtooth or triangular. For example, each sawtooth wave is called a chirp. The duration of each chirp is T, called a cycle, and the frequency of each chirp increases linearly with time. When the transmitted signal encounters a target, it is reflected back by the target. The reflected electromagnetic wave is called the echo signal. The radar system's receiving antenna receives the echo signal, identifies it based on the transmitted signal, and, using a mixer, mixes the echo signal with the transmitted signal to obtain the difference frequency signal between the transmitted and echo signals.

[0097] FIG1 is a schematic diagram of a transceiver link in an embodiment of the present application. As shown in FIG1 , it may include a transmitting link (TX, Transmitter) and a receiving link (RX, Receiver), etc. Among them, the transmitting link may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA, Power Amplifier), etc. connected in sequence. At the same time, the signal amplified by the PA is radiated to a preset spatial area through a transmitting antenna. The receiving chain may include a low-noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), a real analog-to-digital converter (Real ADC), and the like, connected in sequence. That is, the echo signal received by the receiving antenna is processed in sequence by the above-mentioned LNA, Real Mixer, TIA, LPF, HPF, and Real ADC before being converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information such as the target's distance, speed, angle, altitude, and micro-motion characteristics.

[0098] Optionally, in an embodiment of the present application, the receiving link may include a receiving antenna, which may be connected via a peripheral port of the chip and formed on a carrier such as a PCB. In some optional embodiments, the receiving antenna may also be integrated into the chip package to form an AiP or AoP, i.e., a chip structure with a packaged antenna.

[0099] In order to improve the radar measurement accuracy, multi-antenna receiving technology is usually used. The length from the receiving antenna to the LNA (referred to as the receiving antenna feeder length in this application) is different, which will cause the RF signal to pass through different receiving antennas to have a relative delay problem. Moreover, since each receiving antenna channel uses the same local oscillator (LO), the length from the LO to the mixer (Mixer) of each receiving channel (referred to as the RXLO length in this application) is different, which will cause different receiving channels to have a relative delay problem. Figure 2 is a schematic diagram of the antenna feeder in the multiple receiving antennas in an embodiment of the present application. In the receiving end shown in Figure 2, the dotted lines shown by numbers 10, 11, 12, and 13 respectively represent the lengths from different receiving antennas to the LNA; the dotted lines shown by numbers 20, 21, 22, and 23 respectively represent the lengths from the LO to the different mixers of each receiving channel.

[0100] In order to eliminate the problem of relative delay between different receiving channels caused by differences in line lengths, an embodiment of the present application provides a signal processing method that can improve the quality of the received signal by compensating the received signal, thereby improving the receiving performance.

[0101] FIG3 is a flow chart of a signal processing method according to an embodiment of the present application. The signal processing method provided in the embodiment of the present application is applied to an FMCW radar having at least two receiving channels with unequal lengths. The at least two receiving channels include a reference receiving channel and at least one other receiving channel having a length difference with the reference receiving channel. As shown in FIG3 , the method includes:

[0102] Step 300: For any receiving channel, process the echo signal received by the receiving channel to obtain a digital baseband signal.

[0103] In an exemplary embodiment, the processing of the echo signal received by the receiving channel in this step may include but is not limited to low-pass filtering, analog-to-digital conversion, etc.

[0104] Step 301: Compensate the processed digital baseband signal based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitted signal corresponding to the received echo signal.

[0105] In an exemplary embodiment, the frequency information of the transmitted signal corresponding to the received echo signal may include, but is not limited to, a sweep bandwidth, a sweep period, a sweep center frequency, and the like.

[0106] In an exemplary embodiment, the unequal line lengths may include unequal receive antenna feeder line lengths and / or unequal RXLO lengths.

[0107] The embodiments of the present application provide a signal processing method for receiving signal compensation. For an FMCW radar having at least two receiving channels with unequal line lengths, the method compensates for the unequal reception lengths of digital baseband signals in the digital domain based on the length difference between the receiving channels. This method eliminates the relative delays caused by the differences in line lengths between the receiving channels, improves the quality of the received signal, and thus enhances the receiving performance.

[0108] In an exemplary embodiment, for a case where the line lengths are unequal, including the lengths of the receiving antenna feeders, step 301 may include: obtaining an echo signal received by the reference receiving channel based on a time delay caused by a length of a reference receiving antenna feeder corresponding to the reference receiving channel; processing the echo signal to obtain a digital baseband signal of the reference receiving antenna at the receiving end; generating an echo signal received by receiving channel i based on a time delay caused by a length of a receiving antenna i feeder corresponding to receiving channel i and the obtained echo signal received by the reference receiving channel; processing the echo signal to obtain a digital baseband signal of receiving antenna i at the receiving end; and compensating for the digital baseband signal of receiving antenna i at the receiving end using a difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0109] In an exemplary embodiment, for the case where the line lengths are unequal including the RXLO lengths, step 301 includes: obtaining an echo signal received by the reference receiving channel according to the time delay generated by the reference RXLO length corresponding to the reference receiving antenna corresponding to the reference receiving channel; processing the echo signal to obtain a digital baseband signal of the reference receiving antenna at the receiving end; and calculating the time delay generated by the reference RXLO length corresponding to the reference receiving antenna corresponding to the reference receiving channel according to ... i The echo signal received by receiving channel i is generated by taking into account the time delay caused by the length and the echo signal received by the obtained reference receiving channel; the digital baseband signal of receiving antenna i at the receiving end is obtained after processing the echo signal; and the digital baseband signal of receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0110] In an exemplary embodiment, for a case where the line lengths are unequal, including the lengths of the receiving antenna feeders and the lengths of the RXLOs, step 301 includes: obtaining an echo signal received by the reference receiving channel according to the time delay caused by the length of the reference receiving antenna feeder corresponding to the reference receiving channel and the time delay caused by the length of the reference RXLO corresponding to the reference receiving antenna; processing the echo signal to obtain a digital baseband signal of the reference receiving antenna at the receiving end; and processing the echo signal according to the time delay caused by the length of the feeder of the receiving antenna i corresponding to the receiving channel i and the time delay caused by the length of the RXLO corresponding to the receiving antenna i. i The time delay caused by the length and the echo signal received by the obtained reference receiving channel are used to generate the echo signal received by the receiving channel i; the digital baseband signal of the receiving antenna i at the receiving end is obtained after processing the echo signal; and the digital baseband signal of the receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0111] Figure 4 is a flow chart of an embodiment of a received signal compensation method in an embodiment of the present application, which is applied to an FMCW radar having at least two receiving channels and unequal line lengths, wherein the at least two receiving channels include a reference receiving channel and at least one remaining receiving channel i having a length difference compared to the reference receiving channel, where i is an integer greater than or equal to 1; the antenna corresponding to the reference receiving channel is the reference receiving antenna, and the antenna corresponding to receiving channel i is the receiving antenna i; in this embodiment, taking the unequal line lengths including unequal receiving antenna feeder lengths and unequal RXLO lengths as an example, the method may include: Step 400: Acquire an echo signal received by the reference receiving channel based on a time delay caused by a reference receiving antenna feeder length corresponding to the reference receiving channel and a time delay caused by a reference RXLO length corresponding to the reference receiving antenna.

[0112] In an exemplary embodiment, obtaining an echo signal received by a reference receiving channel includes: obtaining a first receiving signal x1(t-τ) input from a reference receiving antenna via an LNA to a mixer on a receiving channel where the reference receiving antenna is located. 11 ), and the second receiving signal x1(t-τ 12 ); for the first received signal x1(t-τ 11 ) and the second received signal x1(t-τ 12 ) is correlated to obtain the echo signal x1(t-τ received by the reference receiving channel 11 )x1*(t-τ 12 ); where τ 11 represents the time delay caused by the feeder length of the reference receiving antenna, τ 12 Indicates the time delay caused by the reference RXLO length corresponding to the reference receiving antenna.

[0113] If we ignore the relative delay of the RF signal passing through different receiving antennas due to the inconsistent length of the receiving antenna feeder, and the relative delay of different receiving channels due to the different RXLO lengths, then the echo signal x1(t) received by each receiving channel at the receiving end is as shown in formula (1):

[0114] In formula (1), β represents the sweep bandwidth, T represents the time it takes for the frequency of a single chirp signal of the FMCW radar to rise, θ represents the initial phase of the signal, and f c is the center frequency of the swept signal.

[0115] For the reference receiving antenna corresponding to the reference receiving channel, assuming that the delay caused by the reference receiving antenna feeder length F is τ 11Then, according to formula (1), the first receiving signal x1(t-τ) from the reference receiving antenna through the LNA to the mixer on the receiving channel where the reference receiving antenna is located is 11 ) is shown in formula (2):

[0116] Assume that the delay caused by the reference RXLO length corresponding to the reference receiving antenna is τ 12 Then, according to formula (1), the second receiving signal x1(t-τ 12 ) is shown in formula (3):

[0117] According to the time delay caused by the reference receiving antenna feeder length and the time delay caused by the reference RXLO length corresponding to the reference receiving antenna, after passing through the mixer, the echo signal received by the reference receiving channel is x1(t-τ 11 )x1*(t-τ 12 ).

[0118] Step 401: Based on the delay caused by the length of the receiving antenna i feeder corresponding to the receiving channel i and the RXLO corresponding to the receiving antenna i i The time delay generated by the length and the obtained echo signal received by the reference receiving channel are used to generate the echo signal received by the receiving channel i; where i is an integer greater than or equal to 1.

[0119] In an exemplary embodiment, generating an echo signal received by a receiving channel i includes: obtaining a third receiving signal x1(t-τ) from a receiving antenna i via an LNA and inputting it into a mixer on the receiving channel where the receiving antenna i is located. 11 -μ i11 ), and the fourth receiving signal x1(t-τ 12 -μ i12 ); the third received signal x1(t-τ 11 -μ i11 ) and the fourth received signal x1(t-τ 12 -μ i12 ) is correlated to obtain the echo signal x1(t-τ received by receiving channel i 11 -μ i11 )x1*(t-τ 12 -μ i12 ); where μ i11 represents the time delay caused by the feeder length of the receiving antenna i relative to the feeder length of the reference antenna, μ i12 Indicates the RXLO corresponding to the receiving antenna i iThe delay caused by the length relative to the reference RXLO length.

[0120] For a receiving antenna i among multiple receiving antennas, assuming that the feeder length of receiving antenna i is F i The time delay relative to the reference antenna feeder length F is μ i11 , RXLO corresponding to receiving antenna i i The delay caused by the length relative to the reference RXLO length is μ i12 , antenna feed line length F1 and RXLO i After the length passes through the mixer on the receiving channel where the receiving antenna i is located, the echo signal x1(t-τ 11 -μ i11 )x1*(t-τ 12 -μ i12 ) is shown in formula (4):

[0121] In formula (4), represents the initial digital mixer phase within each chirp; represents the beat frequency compensation of the digital mixer; Represents the delay μ of different receiving antennas i11 compensate.

[0122] From formula (4), we can deduce the relationship between the echo signal received by the reference receiving channel and the echo signal received by receiving channel i due to time delay, as shown in formula (5):

[0123] Step 402: Process the echo signal received by the receiving channel i and the echo signal received by the reference receiving channel to obtain a digital baseband signal of the receiving antenna i at the receiving end and a digital baseband signal of the reference receiving antenna at the receiving end.

[0124] In an exemplary embodiment, the processing of the received echo signal in this step may include but is not limited to low-pass filtering, analog-to-digital conversion, etc.

[0125] Step 403: Compensate the digital baseband signal of the receiving antenna i at the receiving end by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital received signal of the reference receiving antenna at the receiving end.

[0126] In an exemplary embodiment, the digital baseband signal of receiving antenna i at the receiving end can be compensated based on the sweep bandwidth, the chirp effective edge period, and the delay of the current receiving antenna compared to the reference antenna. For the sake of convenience, analog signals are still used for description. According to formula (5), it can be used Perform frequency compensation on the digital baseband signal of receiving antenna i at the receiving end.

[0127] The present invention provides a signal processing method for receiving signal compensation, which eliminates the delay caused by the length of the feeder of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the frequency of the received signal of the receiving antenna i at the receiving end. i The time delay caused by the length partially affects the received signal of the receiving antenna i at the receiving end, thereby improving the quality of the received signal and thus improving the receiving performance.

[0128] In an exemplary instance, a signal processing method provided by an embodiment of the present application may also include: performing a first phase compensation on the digital baseband signal of receiving antenna i at the receiving end using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0129] In one embodiment, a first phase compensation can be performed on the digital baseband signal of receiving antenna i at the receiving end based on the sweep bandwidth, the chirp effective edge period, the center frequency of the sweep signal, and the time delay of the current receiving antenna compared to the reference antenna. For the sake of convenience, an analog signal is still used for description. According to formula (5), it can be used A first phase compensation is performed on the received signal of the receiving antenna i at the receiving end.

[0130] The embodiment of the present application provides a signal processing method for receiving signal compensation, which further eliminates the delay caused by the feeder length of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the first phase of the received signal of the receiving antenna i at the receiving end. i The influence of the time delay caused by the length on the received signal of the receiving antenna i at the receiving end further improves the quality of the received signal, thereby improving the receiving performance.

[0131] In an exemplary instance, a signal processing method provided in an embodiment of the present application, on the basis of frequency compensation, or on the basis of frequency compensation and first phase compensation, may also include: using the difference between the received signal of the receiving antenna i at the receiving end and the received signal of the reference receiving antenna at the receiving end, to perform a second phase compensation on the received signal of the receiving antenna i at the receiving end.

[0132] In one embodiment, a second phase compensation can be performed on the digital baseband signal of receiving antenna i at the receiving end based on the sweep bandwidth, the chirp effective edge period, the delay generated by the reference antenna, and the delay of the current receiving antenna compared to the reference antenna. For the sake of convenience, an analog signal is still used for description. According to formula (5), it can be used A second phase compensation is performed on the received signal of the receiving antenna i at the receiving end.

[0133] The signal processing method provided by the embodiment of the present application eliminates the delay caused by the feeder length of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the second phase of the received signal of the receiving antenna i at the receiving end. i The influence of the time delay caused by the length on the received signal of the receiving antenna i at the receiving end better improves the quality of the received signal, thereby improving the receiving performance.

[0134] In an exemplary embodiment, the received signal of receiving antenna i at the receiving end is subjected to frequency compensation, first phase compensation and second phase compensation by using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0135] In one embodiment, for the sake of convenience, analog signals are still used for description. According to formula (5), The signal received by the receiving antenna i at the receiving end is compensated. This embodiment provides a signal processing method that effectively eliminates the delay caused by the feeder length of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the signal received by the receiving antenna i at the receiving end. i The influence of the time delay caused by the length on the received signal of the receiving antenna i at the receiving end further improves the quality of the received signal, thereby improving the receiving performance.

[0136] In one embodiment, in actual use, the information used for compensation can be Simplified to:

[0137] The present application also provides a computer-readable storage medium storing computer-executable instructions, wherein the computer-executable instructions are used to execute any one of the signal processing methods described above.

[0138] The present application further provides a device for implementing received signal compensation, comprising a memory and a processor, wherein the memory stores the following instructions executable by the processor: for executing the steps of any one of the signal processing methods described above.

[0139] FIG5 is a schematic diagram of the structure of a signal processing link according to an embodiment of the present application, which is applied to an FMCW radar having at least two receiving channels and unequal line lengths. As shown in FIG5 , a processing module and a compensation module are provided. The processing module is configured to process the echo signal received by any receiving channel to obtain a digital baseband signal. The at least two receiving channels include a reference receiving channel and at least one other receiving channel having a length difference compared to the reference receiving channel. The compensation module is configured to compensate the processed digital baseband signal based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitted signal corresponding to the received echo signal. In one exemplary embodiment, the unequal line lengths may include unequal lengths of receiving antenna feeders and / or unequal lengths of RXLOs.

[0140] The signal processing link provided in the embodiments of the present application compensates for the unequal reception lengths of digital baseband signals in the digital domain based on the length difference between the receiving channels for an FMCW radar having at least two receiving channels with unequal line lengths. This eliminates the relative delays inherent in different receiving channels due to the line length differences between the receiving channels, improves the quality of the received signal, and thus enhances reception performance.

[0141] In an exemplary instance, the unequal line lengths include unequal lengths of receiving antenna feeders; the compensation module can be used to: obtain an echo signal received by a reference receiving channel based on a time delay caused by a reference receiving antenna feeder length corresponding to the reference receiving channel; obtain a digital baseband signal of the reference receiving antenna at the receiving end after processing the echo signal; generate an echo signal received by receiving channel i based on a time delay caused by a feeder length of receiving antenna i corresponding to receiving channel i and the obtained echo signal received by the reference receiving channel; obtain a digital baseband signal of receiving antenna i at the receiving end after processing the echo signal; and compensate for the digital baseband signal of receiving antenna i at the receiving end by using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0142] In an exemplary embodiment, the unequal line lengths include unequal RXLO lengths; the compensation module can be used to: obtain an echo signal received by the reference receiving channel according to the time delay generated by the reference RXLO length corresponding to the reference receiving antenna corresponding to the reference receiving channel; obtain a digital baseband signal of the reference receiving antenna at the receiving end after processing the echo signal; and obtain a digital baseband signal of the reference receiving antenna at the receiving end according to the RXLO length corresponding to the receiving antenna i corresponding to the receiving channel i. iThe echo signal received by receiving channel i is generated by taking into account the time delay caused by the length and the echo signal received by the obtained reference receiving channel; the digital baseband signal of receiving antenna i at the receiving end is obtained after processing the echo signal; and the digital baseband signal of receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

[0143] In an exemplary embodiment, the unequal line lengths include unequal lengths of receiving antenna feeders and unequal lengths of RXLOs; the compensation module can be used to: obtain an echo signal received by a reference receiving channel according to the time delay caused by the reference receiving antenna feeder length corresponding to the reference receiving channel and the time delay caused by the reference RXLO length corresponding to the reference receiving antenna; obtain a digital baseband signal of the reference receiving antenna at the receiving end after processing the echo signal; and compensate the time delay caused by the feeder length of receiving antenna i corresponding to receiving channel i and the time delay caused by the RXLO length corresponding to receiving antenna i. i The processing module is further configured to generate an echo signal received by receiving channel i based on the time delay caused by the length of the received signal and the echo signal received by the obtained reference receiving channel; the echo signal is processed to obtain a digital baseband signal of receiving antenna i at the receiving end; and the digital baseband signal of receiving antenna i at the receiving end is compensated using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end. In one exemplary embodiment, the processing module is further configured to perform low-pass filtering on the echo signal received by the receiving channel to convert the low-pass filtered signal into a digital baseband signal.

[0144] FIG6 is a schematic diagram of the structure of an embodiment of a signal processing link in an embodiment of the present application. As shown in FIG6 , the signal processing link may include: a first processing module, a second processing module, a third processing module, and a compensation module; wherein the first processing module is configured to obtain the echo signal received by the reference receiving channel according to the delay caused by the reference receiving antenna feeder length corresponding to the reference receiving channel and the delay caused by the reference RXLO length corresponding to the reference receiving antenna; the second processing module is configured to obtain the echo signal received by the reference receiving channel according to the delay caused by the feeder length of the receiving antenna i corresponding to the receiving channel i and the delay caused by the RXLO length corresponding to the receiving antenna i. i The time delay generated by the length and the echo signal received by the obtained reference receiving channel are used to generate the echo signal received by the receiving channel i; the third processing module is used to process the echo signal received by the receiving channel i and the echo signal received by the reference receiving channel to obtain the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end; the compensation module is used to compensate the digital baseband signal of the receiving antenna i at the receiving end by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital receiving signal of the reference receiving antenna at the receiving end.

[0145] In an exemplary embodiment, for the sake of convenience, analog signals are still used for description. The compensation module can be used to: Perform frequency compensation on the digital baseband signal of receiving antenna i at the receiving end; where β represents the frequency sweep bandwidth, T represents the time for the frequency rise of a single chirp signal of the FMCW radar, and μ i11 represents the time delay caused by the feed line length of the receiving antenna i relative to the reference antenna feed line length, μ i12 Indicates the RXLO corresponding to the receiving antenna i i The delay caused by the length relative to the reference RXLO length.

[0146] In an exemplary embodiment, the compensation module can employ a signal processing method described in any of the embodiments of the present application to compensate the processed digital baseband signal. In an exemplary embodiment, the compensation module can also be configured to perform a first phase compensation on the digital baseband signal of receiving antenna i at the receiving end using the difference between the digital baseband signal of receiving antenna i at the receiving end and the digital received signal of the reference receiving antenna at the receiving end.

[0147] In one embodiment, for ease of description, analog signals are still used for description. The compensation module may adopt: Perform the first phase compensation on the received signal of receiving antenna i at the receiving end; wherein, f c is the center frequency of the sweep signal, τ 12 Indicates the time delay caused by the reference RXLO length corresponding to the reference receiving antenna.

[0148] The signal processing link provided by the embodiment of the present application further eliminates the delay caused by the feeder length of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the first phase of the received signal of the receiving antenna i at the receiving end. i The influence of the time delay caused by the length on the received signal of the receiving antenna i at the receiving end further improves the quality of the received signal, thereby improving the receiving performance.

[0149] In an exemplary embodiment, the compensation module can also be used to: use the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital receiving signal of the reference receiving antenna at the receiving end to perform a second phase compensation on the digital baseband signal of the receiving antenna i at the receiving end.

[0150] In one embodiment, for the sake of convenience, analog signals are still used for description. The compensation module can be used Perform the second phase compensation on the received signal of receiving antenna i at the receiving end; where τ 11 Indicates the delay caused by the feeder length of the reference receiving antenna.

[0151] The embodiment of the present application provides a signal processing link, which eliminates the delay caused by the length of the feeder of the receiving antenna i and the RXLO corresponding to the receiving antenna i by compensating the second phase of the received signal of the receiving antenna i at the receiving end. i The influence of the time delay caused by the length on the received signal of the receiving antenna i at the receiving end better improves the quality of the received signal, thereby improving the receiving performance.

[0152] In an exemplary embodiment, the compensation module is used to compensate the digital baseband signal of the receiving antenna i at the receiving end based on the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital received signal of the reference receiving antenna at the receiving end. In one embodiment, the digital baseband signal of the receiving antenna i at the receiving end can be compensated based on the sweep bandwidth, the chirp effective edge period, the center frequency of the sweep signal, and the delay of the current receiving antenna compared to the reference antenna. For the sake of convenience, the analog signal is still used for description here, and the compensation module can adopt Compensate the received signal of receiving antenna i at the receiving end.

[0153] FIG7 is a schematic diagram of a first circuit embodiment of the compensation module in an embodiment of the present application. As shown in FIG7 , before compensating the received signal of the receiving antenna i at the receiving end, the received signal RX of the receiving antenna i at the receiving end can be processed by the third processing module. i Perform one-way preprocessing, for example: after processing through a low-pass filter (LPF) and a digital-to-analog converter (ADC), the preprocessed digital baseband signal of the receiving antenna i at the receiving end is obtained, which is expressed as Will Input compensation module.

[0154] In the first embodiment shown in FIG. 7 , the information used for compensation middle, is θ0, f c μ 11 is Δθ. Information used for compensation It can be expressed as In a first embodiment, as shown in FIG7 , the compensation module may include: a first multiplier M1, a second multiplier M2, and a first digital local oscillator (Digital LO generator); wherein the input of the first digital local oscillator includes θ0 and Δθ, which are information for compensation, and the output of the first digital local oscillator is The output of the first digital local oscillator One input of the first multiplier M1 is the pre-processed digital baseband signal of the receiving antenna i at the receiving end. The other input of the first multiplier M1 is the output LO_I of the first digital local oscillator, and the output of the first multiplier M1 is the real part Y of the signal received by the compensated receiving antenna i. i _I; One input of the second multiplier M2 is the preprocessed receiving signal of the receiving antenna i at the receiving end The other input of the second multiplier M2 is the output LO_Q of the first digital local oscillator, and the output of the second multiplier M2 is the imaginary part Y of the signal received by the compensated receiving antenna i. i _Q. The signal received by receiving antenna i after compensation can be expressed as

[0155] FIG8 is a schematic diagram of a second circuit embodiment of the compensation module in an embodiment of the present application. As shown in FIG8 , before compensating the received signal of the receiving antenna i at the receiving end, the received signal RX of the receiving antenna i at the receiving end can be processed by the third processing module. i The real and imaginary parts of the signal are preprocessed separately. For example, the real part of the digital baseband signal RXi at the receiving end of the preprocessed receiving antenna i is obtained by processing the signal through the first LPF and the first ADC. It is expressed as After processing by the second LPF and the second ADC, the digital baseband signal RX of the receiving antenna i at the receiving end after preprocessing is obtained. i The imaginary part of Will Enter the compensation module separately.

[0156] In the second embodiment shown in FIG8 , the information used for compensation middle, is θ0, f c μ 11 is Δθ. Information used for compensation It can be expressed as In the second embodiment, as shown in FIG6 , the compensation module may include: a third multiplier M3, a fourth multiplier M4, a fifth multiplier M5, a sixth multiplier M6, a first adder S1, a second adder S2, and a second digital local oscillator; wherein the input of the second digital local oscillator includes θ0 and Δθ of the information used for compensation, and the output of the second digital local oscillator includes The output of the second digital local oscillator One input of the third multiplier M3 is the real part of the received signal of the receiving antenna i at the receiving end after preprocessing. Another input of the third multiplier M3 is the output LO_I of the second digital local oscillator, and the output of the third multiplier M3 is connected to an input of the second adder; the input of the fourth multiplier M4 is the output LO_Q of the second digital local oscillator, and the output of the fourth multiplier M4 is connected to an input of the first adder; one input of the fifth multiplier M5 is the imaginary part of the received signal of the receiving antenna i at the receiving end after preprocessing The other input of the fifth multiplier M5 is the output LO_I of the second digital local oscillator, and the output of the fifth multiplier M5 is connected to the other input of the first adder; the input of the sixth multiplier M6 is the output LO_Q of the second digital local oscillator, and the output of the sixth multiplier M6 is connected to the other input of the first adder; the two inputs of the first adder S1 are the output of the fourth multiplier M4 and the output of the fifth multiplier M5, respectively, and the output of the first adder S1 is the real part Y of the received signal of the receiving antenna i after compensation i _I; the two inputs of the second adder S2 are the output of the third multiplier M3 and the output of the sixth multiplier M6, the output of the first adder S1 is the compensated imaginary part Y of the receiving signal received by the receiving antenna i i _Q. The signal received by receiving antenna i after compensation can be expressed as

[0157] In an exemplary embodiment, the first processing module may be configured to: obtain a first receiving signal x1(t-τ) which is input from the reference receiving antenna via the LNA to the mixer on the receiving channel where the reference receiving antenna is located. 11 ), and the first receiving signal x1(t-τ 12 ); for the first received signal x1(t-τ 11 ) and the second received signal x1(t-τ 12 ) is correlated to obtain the echo signal x1(t-τ received by the reference receiving channel 11 )x1*(t-τ 12 );in,

[0158] in,

[0159] In an exemplary embodiment, the second processing module may be configured to obtain a third receiving signal x1(t-τ) input from receiving antenna i via LNA to a mixer on a receiving channel where receiving antenna i is located. 11 -μ i11 ), and the fourth receiving signal x1(t-τ 12 -μ i12); the third received signal x1(t-τ 11 -μ i11 ) and the fourth received signal x1(t-τ 12 -μ i12 ) is correlated to obtain the echo signal x1(t-τ received by receiving channel i 11 -μ i11 )x1*(t-τ 12 -μ i12 );

[0160] in,

[0161] In some optional embodiments, radar is an electronic device that uses electromagnetic waves to detect targets. That is, it transmits a beam through a signal transmission link. When the transmitted beam encounters an obstacle, the echo reflected by the obstacle is received by the receiving antenna and transmitted to the radar chip. The radar chip then determines the position, distance, speed, and other information of the target relative to the electromagnetic wave emission point. With the development of technologies such as microelectronics, radar has gradually been widely used, especially millimeter-wave radar (such as automotive radar). Due to its small antenna size, it has been widely used in autonomous driving, smart home devices, and industrial automation devices. Currently, the miniaturization and integration of radar have become the current development trend.

[0162] Antenna feeders are devices used in radars to transmit, control, and distribute radio frequency (RF) electromagnetic signal energy. Due to the need for miniaturization and integration of radars, the placement of antenna feeders significantly impacts the quality of the radar chip's perceived signal. Currently, the placement of antenna feeders depends on the positional relationship between the transceiver unit and the antenna in the radar chip. In a Multiple Input Multiple Output (MIMO) system, multiple groups of transceiver antennas exist. To ensure accurate target detection, the relative delays generated by each transmitting antenna must be the same. However, when designing unequal-length transmitting antenna feeders, the varying lengths of the transmitting antenna feeders result in varying relative delays for RF signals after passing through different transmitting antennas, impacting both the quality of the received signal and the accuracy of the radar system's detection results.

[0163] As shown in Figure 9, an embodiment of the present application provides a signal processing link for signal transmission, including a waveform generator 1101, multiple transmission channels 1100 connected to the waveform generator 1101, and multiple transmission antennas 1107 connected one-to-one to the transmission channels 1100. At least two transmission channels 1100 have different signal transmission delays from the waveform generator 1101 to the transmission antenna 1107. At least one transmission channel 1100 includes: a direct digital frequency synthesizer 1102, which can be configured to generate a compensation signal based on the delay difference of the signal transmission delay of the corresponding transmission channel 1100 relative to the reference transmission delay, and use the compensation signal to compensate for the transmission signal of the corresponding transmission channel 1100.

[0164] In an exemplary embodiment, the signal transmission link shown in FIG9 includes four transmission channels. It should be noted that the embodiment of the present application does not limit this, and the number of transmission channels can be set as needed. As shown in FIG9, in the four transmission channels, the lengths of the transmission antenna feeder 110, the transmission antenna feeder 111, the transmission antenna feeder 112, and the transmission antenna feeder 113 are different, and the relative delays generated by the transmission signal passing through the transmission antenna feeder 110, the transmission antenna feeder 111, the transmission antenna feeder 112, and the transmission antenna feeder 113 are also different. In this embodiment, a compensation signal is generated by the direct digital frequency synthesizer 1102 based on the delay difference of the signal transmission delay of the corresponding transmission channel 1100 relative to the reference transmission delay, and the compensation signal is used to compensate for the transmission signal of the corresponding transmission channel 1100, thereby solving the problem that the relative delay of the RF signal passing through different transmission antennas due to inconsistent transmission antenna feeder lengths is different, thereby improving the quality of the transmission signal, thereby improving the quality of the received signal, and ultimately improving the detection accuracy and detection range of the radar system.

[0165] In some exemplary embodiments, the transmission channel further includes a digital-to-analog converter 1103, a low-pass filter 1104, a mixer 1105, and a power amplifier 1106. The signal transmission chain further includes a local oscillator 1108. The waveform generator 1101, the direct digital frequency synthesizer 3102, the digital-to-analog converter 1103, the low-pass filter 1104, the mixer 1105, the power amplifier 1106, and the transmitting antenna 1107 are connected in sequence. The local oscillator 1108 is respectively connected to the waveform generator 1101 and the mixer 1105 of each transmission channel, wherein: the waveform generator 1101 is configured to generate an initial transmission signal and output it to the direct digital frequency synthesizer 1102 and the local oscillator 1108; the direct digital frequency synthesizer 1102 is configured to perform phase compensation and / or frequency compensation on the transmission signal; the digital-to-analog converter 1103 is configured The received signal is converted from digital to analog and output to a low-pass filter 1104. The low-pass filter 1104 is configured to perform low-pass filtering on the received signal and output the signal to a mixer 1105. The mixer 1105 is configured to multiply the eigensignal generated by the local oscillator 1108 by the signal output by the low-pass filter 1104 to generate a radio frequency signal and output the radio frequency signal to a power amplifier 1106. The power amplifier 1106 is configured to amplify the radio frequency signal and output the amplified signal to a transmitting antenna 1107. The local oscillator 1108 is configured to generate an eigensignal based on the initial transmit signal and output the eigensignal to the mixer 1105. In an exemplary embodiment, the initial transmit signal may be, but is not limited to, FMCW.

[0166] In some exemplary embodiments, a direct digital frequency synthesizer 1102 is provided for each transmission channel, wherein each direct digital frequency synthesizer 1102 is configured to generate a compensation signal based on the delay difference of the signal transmission delay of the transmission channel in which it is located relative to the reference transmission delay, and use the compensation signal to compensate for the transmission signal of the transmission channel in which it is located.

[0167] It should be noted that although each transmission channel 1100 in the signal transmission link shown in Figure 9 is provided with a direct digital frequency synthesizer 1102, in actual applications, some transmission channels 1100 may not be provided with a direct digital frequency synthesizer 1102. For example, when the signal transmission delay of a certain transmission channel 1100 is a reference transmission delay and the signal transmission link is a fixed-frequency signal transmission link, the transmission channel 1100 may not be provided with a direct digital frequency synthesizer 1102. For a transmission channel without a direct digital frequency synthesizer 1102, it includes a digital-to-analog converter 1103, a low-pass filter 1104, a mixer 1105 and a power amplifier 1106 connected in sequence. The waveform generator 1101 generates an initial transmission signal and outputs it to the digital-to-analog converter 1103 and the local oscillator 1108. The digital-to-analog converter 1103 performs digital-to-analog conversion on the received signal and outputs it to the low-pass filter 1104. The low-pass filter 1104 performs low-pass filtering on the received signal and outputs it to the mixer 1105. The mixer 1105 multiplies the eigensignal generated by the local oscillator 1108 by the signal output by the low-pass filter 1104 to generate a radio frequency signal and output the radio frequency signal to the power amplifier 1106. The power amplifier 1106 amplifies the radio frequency signal and outputs the amplified signal to the transmitting antenna 1107. The local oscillator 1108 generates an eigensignal based on the initial transmission signal and outputs it to the mixer 1105.

[0168] As shown in FIG9 , since each transmit channel uses the same local oscillator 1108, the lengths of the signal lines from the local oscillator 1108 to the mixer 1105 of each transmit channel (labeled 120, 121, 122, and 123 in FIG9 ) also vary. This difference will also cause different transmit channels to have different relative delays.

[0169] In some exemplary embodiments, the signal transmission delays of at least two transmitting channels are different, including different signal transmission delays generated by the transmitting antenna feed lines of the at least two transmitting channels, and / or different signal transmission delays generated by the signal lines from the local oscillator to the mixers of the at least two transmitting channels.

[0170] In one exemplary embodiment, the direct digital frequency synthesizer 1102 generates a compensation signal based on a delay difference between the signal transmission delay of its corresponding transmit channel 1100 and a reference transmission delay, including: the direct digital frequency synthesizer 1102 generates a compensation signal based on a delay difference between the signal transmission delay generated by the transmit antenna feed line of its corresponding transmit channel 1100 and a first reference transmission delay, and / or a delay difference between the signal transmission delay generated by the signal line between the local oscillator 1108 and the mixer 1105 of its corresponding transmit channel 1100 and a second reference transmission delay. However, the present embodiment is not limited to this, and the direct digital frequency synthesizer 1102 may also generate a compensation signal based on a delay difference between the signal transmission delay generated at other locations in its corresponding transmit channel 1100 and the reference transmission delay, thereby resolving the problem of different signal transmission delays caused by length differences between the signal line from the local oscillator 1108 to the mixers 1105 of at least two transmit channels 1100, or by differences at any other location.

[0171] In some exemplary embodiments, when the delay difference between the signal transmission delay of a transmission channel and a reference transmission delay includes the delay difference between the signal transmission delay of a transmission antenna feeder of the transmission channel and a first reference transmission delay, the first reference transmission delay may be the signal transmission delay generated by the transmission antenna feeder of any one of the multiple transmission channels. When the delay difference between the signal transmission delay of a transmission channel and the reference transmission delay includes the delay difference between the signal transmission delay from a local oscillator to a mixer of the transmission channel and a second reference transmission delay, the second reference transmission delay may be the signal transmission delay generated by a signal line from the local oscillator to the mixer of any one of the multiple transmission channels.

[0172] In some exemplary embodiments, when the delay difference between the signal transmission delay of a transmit channel and a reference transmission delay includes the delay difference between the signal transmission delay of a transmit antenna feeder of the transmit channel and a first reference transmission delay, the first reference transmission delay may be the signal transmission delay generated by the shortest transmit antenna feeder among the multiple transmit channels. When the delay difference between the signal transmission delay of a transmit channel and the reference transmission delay includes the delay difference between the signal transmission delay from a local oscillator to a mixer of the transmit channel and a second reference transmission delay, the second reference transmission delay may be the signal transmission delay generated by the shortest signal line from the local oscillator to the mixers among the multiple transmit channels.

[0173] FIG10 is a schematic diagram of an FMCW transmission signal using sawtooth wave modulation. As shown in FIG10 , a sawtooth is generally referred to as a chirp. In some exemplary embodiments, the mathematical expression corresponding to the initial transmission signal x2(t) generated by the waveform generator is as follows:

[0174] Where t is time, f c is the starting frequency of a chirp during the frequency sweep process, β is the frequency variation range of a chirp during the frequency sweep process, T is the scanning duration of a chirp during the frequency sweep process, and θ is the initial phase.

[0175] Figures 11A and 11B are schematic diagrams of the FMCW transmission signals and their echo signals of two different transmitting antennas. As shown in Figures 11A and 11B, due to the different relative delays generated by the transmission channels corresponding to the two transmitting antennas, Δt1>Δt0, that is, the relative delay of the transmitting antenna shown in Figure 11B is greater than the relative delay of the transmitting antenna shown in Figure 11A, resulting in a difference in the range frequency of the target detected by the two transmitting antennas, Δf1>Δf0, which in turn affects the final detection result of the radar chip.

[0176] Assume that the signal transmission delay generated by the transmitting antenna feeder of any transmitting channel in the multiple transmitting channels is defined as the first reference delay τ 21 , and define the signal transmission delay generated by the signal line from the local oscillator to the mixer of any one of the multiple transmission channels as the second reference delay τ22, then after the first reference delay τ 21 and the second reference delay τ 22 The FMCW signal can be expressed as follows:

[0177] where t≥τ=τ 21 +τ 22 .

[0178] In some exemplary instances, the first reference delay τ may be 21 Defined as the signal transmission delay generated by the shortest transmitting antenna feeder, the second reference delay τ 22 It is defined as the signal transmission delay generated by the shortest signal line between the local oscillator and the mixer. However, this embodiment of the present application is not limited to this. In other exemplary embodiments, the signal transmission delay generated by the transmitting antenna feed line of any transmitting channel can also be defined as the first reference delay τ 21 The signal transmission delay generated by the signal line from the local oscillator to the mixer of the transmission channel is the second reference delay τ 22 .

[0179] Assume that the signal transmission delay generated by the transmitting antenna feeder of any transmitting channel is relative to the first reference delay τ 21 The additional delay is μ 21 , and the signal transmission delay generated by the signal line from the local oscillator to the mixer of the transmitting channel relative to the second reference delay τ22 is μ 22, then the FMCW signal passing through the transmission channel can be expressed as follows:

[0180] Where t≥τ+μ, μ=μ 21 +μ 22 .

[0181] In some exemplary embodiments, the corresponding compensation signal may be generated by a direct digital frequency synthesizer. Compensate the transmit signal of the transmit channel.

[0182] In some exemplary embodiments, as shown in FIG12A , each transmit channel includes an I transmit channel and a Q transmit channel, the compensation signal includes a cosine compensation signal and a sine compensation signal, the cosine compensation signal and the sine compensation signal have the same frequency, the cosine compensation signal and the sine compensation signal have the same initial phase, and one of the cosine compensation signal and the sine compensation signal is used to compensate for the current transmit signal of the I transmit channel, and the other is used to compensate for the current transmit signal of the Q transmit channel.

[0183] As shown in Figure 12A, the transmit channel includes an IQ digital-to-analog converter (i.e., digital-to-analog converter 11031 and digital-to-analog converter 11032), an IQ low-pass filter (i.e., low-pass filter 11041 and low-pass filter 11042), an IQ mixer (i.e., mixer 11051 and mixer 11052), a power amplifier 1106, and a transmit antenna 1107. The signal transmission chain includes a local oscillator 1108. The waveform generator 1101, direct digital frequency synthesizer 1102, IQ digital-to-analog converter, IQ low-pass filter, IQ mixer, power amplifier 1106, and transmit antenna 1107 are connected in sequence. The local oscillator 1108 is connected to the waveform generator 1101 and the IQ mixer of each transmit channel. The waveform generator 1101 is configured to generate a frequency-modulated continuous wave signal; the direct digital frequency synthesizer 1102 is configured to perform phase compensation and / or frequency compensation on the transmit signal.

[0184] In some exemplary embodiments, the cosine compensation signal may be The sinusoidal compensation signal can be Where n is the digital signal sampling point, f c is the starting frequency of the initial transmission signal generated by the waveform generator, β is the frequency variation range of the initial transmission signal generated by the waveform generator, T is the scanning duration of one chirp in the initial transmission signal generated by the waveform generator, and μ is the delay difference of the signal transmission delay of the transmission channel relative to the reference transmission delay.

[0185] The signal processing link of the embodiment of the present application generates a compensation signal based on the delay difference μ of the signal transmission delay of the transmission channel relative to the reference transmission delay, and uses the compensation signal to compensate the transmission signal of the corresponding transmission channel. This solves the problem of different relative delays of RF signals passing through different transmission antennas due to inconsistent transmission antenna feeder lengths and / or inconsistent signal line lengths from the local oscillator to the mixers of multiple transmission channels, thereby improving the quality of the transmission signal, thereby improving the quality of the received signal, and ultimately improving the detection accuracy and detection range of the radar system.

[0186] As shown in FIG13 , an embodiment of the present application further provides a signal processing method, which is applied to an antenna array of an electromagnetic wave device having at least two signal transmission chains, wherein the signal transmission chain is a phase shifter including a digital phase shift architecture, and may include:

[0187] Step 1301: Determine a reference transmission delay and a delay difference between the signal transmission delay of each transmission channel and the reference transmission delay.

[0188] In some exemplary embodiments, the reference transmission delay includes a first reference transmission delay and / or a second reference transmission delay, wherein the first reference transmission delay may be the signal transmission delay of the transmitting antenna feed line of any transmitting channel, and the second reference transmission delay may be the signal transmission delay of the signal line from the local oscillator to the mixer of any transmitting channel.

[0189] In other exemplary embodiments, the reference transmission delay includes a first reference transmission delay and / or a second reference transmission delay, wherein the first reference transmission delay may be the signal transmission delay of the shortest transmitting antenna feed line in the multiple transmission channels, and the second reference transmission delay may be the signal transmission delay of the shortest signal line from the local oscillator to the mixers in the multiple transmission channels.

[0190] In an exemplary embodiment, the distance of the transmitting antenna feed line and / or the signal line from the local oscillator to the mixer in each transmitting channel may be measured. The distance measurement may include model simulation measurement or actual measurement.

[0191] Step 1302: Generate a compensation signal corresponding to each transmission channel according to the delay difference.

[0192] In one exemplary embodiment, the frequency control parameter of the compensation signal Phase control parameters Among them, f cis the starting frequency of the initial transmission signal generated by the waveform generator, β is the frequency variation range of the initial transmission signal generated by the waveform generator, T is the scanning duration of one chirp in the initial transmission signal generated by the waveform generator, and μ is the delay difference of the signal transmission delay of the transmission channel relative to the reference transmission delay.

[0193] Step 1303: Use the compensation signal to compensate for the transmission signal of the corresponding transmission channel.

[0194] In an exemplary embodiment, the transmission channel includes an I transmission channel and a Q transmission channel, and the compensation signal includes a cosine compensation signal and sinusoidal compensation signal One of the cosine compensation signal and the sine compensation signal is used to compensate for the transmission signal of the I transmission channel, and the other is used to compensate for the transmission signal of the Q transmission channel.

[0195] Another signal processing method provided in an embodiment of the present application generates a compensation signal based on the delay difference of the signal transmission delay of each transmission channel relative to a reference transmission delay, and uses the compensation signal to compensate the transmission signal of the corresponding transmission channel. This solves the problem of different relative delays of RF signals passing through different transmission antennas due to inconsistent transmission antenna feed line lengths and / or inconsistent signal line lengths from a local oscillator to mixers of multiple transmission channels, thereby improving the quality of the transmission signal, thereby improving the quality of the received signal, and ultimately improving the detection accuracy and detection range of the radar system.

[0196] An embodiment of the present application provides another signal processing link for signal transmission and reception, including: a signal transmission link as shown in FIG14A, and a signal receiving link as shown in FIG14B. The signal transmission link includes multiple transmission channels 2100 implemented based on analog circuits, each transmission channel 2100 includes its own transmission antenna 2105, at least two transmission channels 2100 have different signal transmission delays, and the multiple transmission antennas 2105 transmit electromagnetic wave signals in a time-division multiplexing manner; the signal receiving link includes at least one receiving channel 2200, and the receiving channel 2200 includes a signal compensator 2201 implemented using a digital phase-shifting architecture; the signal compensator 2201 is configured to determine the transmission channel 2100 corresponding to the currently received echo signal, generate a compensation signal based on the delay difference between the signal transmission delay of the corresponding transmission channel 2100 and a reference transmission delay, and use the compensation signal to compensate the currently received echo signal.

[0197] In an exemplary embodiment, the signal transmission link shown in FIG14A includes four transmission channels, and the signal reception link shown in FIG14B includes one reception channel. However, the embodiment of the present application does not limit this, and the number of transmission channels and reception channels can be set as needed. In the embodiment of the present application, as long as the number of transmission channels is greater than or equal to 2 and the number of reception channels is greater than or equal to 1, it is sufficient. As shown in FIG14A, among the four transmission channels, the lengths of the transmission antenna feeder 210, the transmission antenna feeder 211, the transmission antenna feeder 212, and the transmission antenna feeder 213 are different, and the relative delays generated by the transmission signal passing through the transmission antenna feeder 210, the transmission antenna feeder 211, the transmission antenna feeder 212, and the transmission antenna feeder 213 will also be different. In one embodiment, a compensation signal is generated by the signal compensator 2201 based on the delay difference of the signal transmission delay of the transmission channel corresponding to the currently received echo signal relative to the reference transmission delay, and the compensation signal is used to compensate for the currently received echo signal. For example, the compensation signal is used to compensate for the phase difference and / or intermediate frequency signal frequency difference of the currently received echo signal to improve the quality of the received signal, thereby achieving the purpose of improving the detection accuracy and detection range of the radar system.

[0198] In some exemplary embodiments, as shown in FIG14A , the signal transmission chain further includes a waveform generator 2101 and a local oscillator 2102 connected in sequence, each transmission channel includes a phase shifter 2103, a power amplifier 2104, and a transmission antenna 2105 connected in sequence, the local oscillator 2102 is respectively connected to the phase shifter 2103 of each transmission channel 2100, and the waveform generator 2101 is respectively connected to the phase shifter 2103 of each transmission channel, wherein: the waveform generator 2101 is configured to generate an initial transmission signal and output it to the local oscillator 2102 and the phase shifter 2103; the local oscillator 2102 is configured to generate an eigenvalue signal based on the initial transmission signal and output it to the phase shifter 2103; the phase shifter 2103 is configured to phase shift the received initial transmission signal and output it to the power amplifier 2104; and the power amplifier 2104 is configured to power amplify the received signal and output the amplified signal to the transmission antenna 2105.

[0199] In some exemplary embodiments, as shown in FIG14B , each receiving channel 2200 further includes a receiving antenna 2205, a first mixer 2204, a low-pass filter 2203, and an analog-to-digital converter 2202, etc., which are connected in sequence. The receiving antenna 2205, the first mixer 2204, the low-pass filter 2203, the analog-to-digital converter 2202, and the signal compensator 2201 are connected in sequence, wherein: the first mixer 2204 is configured to multiply the signal received by the receiving antenna with the chirp signal transmitted by the transmitting antenna to obtain a mixed signal and output it to the low-pass filter 2203; the low-pass filter 2203 is configured to perform low-pass filtering on the received mixed signal to obtain a difference frequency signal and output it to the analog-to-digital converter 2202; the analog-to-digital converter 2202 is configured to perform analog-to-digital conversion on the received difference frequency signal and output it to the signal compensator 2201; and the signal compensator 2201 is configured to compensate for phase difference and / or frequency difference in the intermediate frequency signal output by the IQ analog-to-digital converter.

[0200] As shown in FIG14A , since each transmission channel uses the same local oscillator 2102, the lengths of the signal lines from the local oscillator 2102 to the phase shifter 2103 of each transmission channel (labeled 220, 221, 222, and 223 in FIG14A ) also vary. This difference will also cause different transmission channels to have different relative delays.

[0201] In some exemplary embodiments, the signal transmission delays of at least two transmitting channels are different, including: the signal transmission delays generated by the transmitting antenna feed lines of the at least two transmitting channels are different, and / or the signal transmission delays generated by the signal lines from the local oscillator to the phase shifters of the at least two transmitting channels are different.

[0202] In an exemplary embodiment, the signal compensator 2201 generates a compensation signal based on the delay difference of the signal transmission delay of the corresponding transmission channel 2100 relative to the reference transmission delay, including: the signal compensator 2201 generates a compensation signal based on the delay difference of the signal transmission delay generated by the transmission antenna feed line of the corresponding transmission channel 2100 relative to the first reference transmission delay, and / or, the delay difference of the signal transmission delay generated by the signal line between the local oscillator 2102 and the phase shifter 2103 of the corresponding transmission channel 2100 relative to the second reference transmission delay. However, the embodiment of the present application is not limited to this. The signal compensator 2201 can also generate a compensation signal based on the signal transmission delay generated at other locations in the corresponding transmission channel 2100, thereby solving the problem of different signal transmission delays caused by the length difference of the signal line from the local oscillator 2102 to the phase shifter 2103 of each transmission channel or the difference at any other location.

[0203] In some exemplary embodiments, when the delay difference of the signal transmission delay of a transmission channel relative to a reference transmission delay includes the delay difference of the signal transmission delay of a transmission antenna feeder of the transmission channel relative to a first reference transmission delay, the first reference transmission delay may be the signal transmission delay generated by the transmission antenna feeder of any one of the multiple transmission channels; when the delay difference of the signal transmission delay of a transmission channel relative to the reference transmission delay includes the delay difference of the signal transmission delay from a local oscillator to a phase shifter relative to a second reference transmission delay, the second reference transmission delay may be the signal transmission delay generated by the signal line from the local oscillator to the phase shifter of any one of the multiple transmission channels. In the disclosed embodiments, the transmission channels corresponding to the first reference transmission delay and the second reference transmission delay may be the same or different.

[0204] In some exemplary embodiments, when the delay difference of the signal transmission delay of a transmitting channel relative to a reference transmission delay includes the delay difference of the signal transmission delay of a transmitting antenna feeder of the transmitting channel relative to a first reference transmission delay, the first reference transmission delay may be the signal transmission delay generated by the shortest transmitting antenna feeder in the multiple transmitting channels; when the delay difference of the signal transmission delay of the transmitting channel relative to the reference transmission delay includes the delay difference of the signal transmission delay from the local oscillator to the phase shifter relative to the second reference transmission delay, the second reference transmission delay may be the signal transmission delay generated by the shortest signal line from the local oscillator to the phase shifters in the multiple transmitting channels.

[0205] FIG15A is a schematic diagram of an FMCW transmission signal using sawtooth wave modulation. As shown in FIG15A , a sawtooth is generally referred to as a chirp. The FMCW transmission signal shown in FIG15A is formed by two transmitting antennas transmitting in a TDM mode, where TX0 represents the first transmitting antenna, TX1 represents the second transmitting antenna, and so on. It should be noted that the first transmitting antenna TX0 and the second transmitting antenna TX1 described in this application only represent the logical order of the antennas and are not required to correspond one-to-one to the physical order of the antennas. The starting frequency, frequency range, chirp duration, etc. of the chirp signals transmitted by multiple transmitting antennas can be the same or different, as long as the chirp signals transmitted by the multiple transmitting antennas do not overlap in time.

[0206] In some exemplary embodiments, the mathematical expression corresponding to the transmitted signal x2(t) is as follows: Where t is time, f c is the starting frequency of a chirp during the frequency sweep process, β is the frequency variation range of a chirp during the frequency sweep process, T is the scanning duration of a chirp during the frequency sweep process, and θ is the initial phase.

[0207] Figure 15B is a schematic diagram of signals for two transmitting antennas and four receiving antennas, where RX0 represents the first receiving antenna, RX1 represents the second receiving antenna, RX2 represents the third receiving antenna, and RX3 represents the fourth receiving antenna. As shown in Figure 15B, each receiving antenna can receive chirp signals transmitted by all transmitting antennas. Figure 15A illustrates a signal transmission chain comprising two transmitting antennas, while Figure 15B illustrates a signal transmission chain comprising two transmitting antennas and a signal receiving chain comprising four receiving antennas. However, this embodiment of the present application is not limited to this.

[0208] In some exemplary embodiments, a corresponding compensation signal may be generated by a signal compensator in a receiving channel. Compensate for the additional delay of the signal sent by the transmit channel.

[0209] In some exemplary embodiments, as shown in FIG14B , the receiving channel includes an I-channel receiving channel Y_I and a Q-channel receiving channel Y_Q, the compensation signal includes a cosine compensation signal and a sine compensation signal, the cosine compensation signal and the sine compensation signal have the same frequency, the cosine compensation signal and the sine compensation signal have the same initial phase, one of the cosine compensation signal and the sine compensation signal is used to compensate for the intermediate frequency signal of the currently received echo signal and then output to the I-channel receiving channel Y_I, and the other of the cosine compensation signal and the sine compensation signal is used to compensate for the intermediate frequency signal of the currently received echo signal and then output to the Q-channel receiving channel Y_Q.

[0210] In some exemplary embodiments, the cosine compensation signal may be The sinusoidal compensation signal can be Where n is the digital signal sampling point, f c is the starting frequency of the chirp signal, β is the frequency variation range of the chirp signal, T is the sweep duration of the chirp signal, and μ is the delay difference between the signal transmission delay of the transmitting channel and the reference transmission delay.

[0211] In some exemplary embodiments, the delay difference μ of the signal transmission delay of the transmission channel relative to the reference transmission delay includes the delay difference of the signal transmission delay of the transmission antenna feeder of the transmission channel relative to the first reference transmission delay, and / or the delay difference of the signal transmission delay from the local oscillator to the phase shifter relative to the second reference transmission delay. The method for setting the first reference transmission delay and the second reference transmission delay is described above and is not further described here.

[0212] In other exemplary embodiments, as shown in FIG16A , when the signal after analog-to-digital conversion of each receiving channel is a complex signal, the signal output end of the receiving antenna 2205 is connected to the first mixer 22041 and the first mixer 22042 (the first mixer 22041 and the first mixer 22042 are collectively referred to as a first IQ mixer), and the signal output ends of the first mixer 22041 and the first mixer 22042 are connected to the low-pass filter 22031 and the low-pass filter 22032 (the low-pass filter 22031 and the low-pass filter 22032 are collectively referred to as an IQ low-pass filter). The signal output by the coupled transmission channel is also connected to the first mixer 22041 and the 90° (πt / 2) phase shifter, respectively. The other end of the 90° phase shifter is connected to the first mixer 22042. The signal output end of the low-pass filter 22031 is connected to the analog-to-digital converter 22021, and the signal output end of the low-pass filter 22032 is connected to the analog-to-digital converter 22022. The signals output from the signal output ends of the analog-to-digital converter 22021 and the analog-to-digital converter 22022 (the analog-to-digital converter 22021 and the analog-to-digital converter 22022 are collectively referred to as IQ analog-to-digital converters) are respectively connected to the cosine compensation signal. The signal obtained by multiplying and summing is output to the Q receiving channel Y_Q, and the signals output from the signal output terminals of the analog-to-digital converter 22021 and the analog-to-digital converter 22022 are respectively combined with the sinusoidal compensation signal The signal obtained after multiplication and summation is output to the I receiving channel Y_I.

[0213] The signal processing link in this embodiment generates a compensation signal through a signal compensator based on the signal transmission delay corresponding to the currently received echo signal, and uses the compensation signal to compensate for the intermediate frequency signal of the corresponding received echo signal, thereby improving the quality of the received signal and further enhancing the detection accuracy and detection range of the radar system.

[0214] In some optional embodiments, the phase shifter 2103 in the transmit chain in FIG14A may be an analog architecture phase shifter or a digital phase shifter architecture phase shifter. For example, in the field of FMCW millimeter wave radar, the structure of the phase shifter in the traditional transmit chain is generally composed of analog circuits. At the same time, the function of the analog phase shifter can be replaced by a digital architecture phase shifter, thereby achieving precise phase shifting operations in each transmit chain. The following is an overview of the transmit and receive chain of the digital phase shift architecture:

[0215] Because mixers are crucial components for frequency conversion in transceiver links, they are widely used in radio devices such as communications and radar. Examples include single-sideband mixers, which effectively suppress image signals, and IQ mixers in transceiver links in the disclosed embodiments. An IQ mixer (such as a single-sideband mixer) transmits signals through two branches: an I branch and a Q branch, each with a 90° phase difference.

[0216] For application scenarios with relatively compact layout areas, such as high-frequency sensor applications in the millimeter wave band, the physical distance between the I branch and the Q branch, and between the input branch and the output branch before mixing are short. As a result, the transmitted signal may leak between branches, between input and output ports, and between the IQ matching network and the mixer output through methods such as magnetic coupling, substrate coupling, and electrical coupling, thereby causing serious deterioration of the mixer's image rejection ratio and local oscillator leakage.

[0217] Figure 12B is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present disclosure; Figure 12C is a schematic diagram of a digital phase shifter architecture in a signal transmission link based on the signal transmission link shown in Figure 12B; and Figure 4D is a schematic diagram of a transmission link including a specific compensation module in an embodiment of the present disclosure.

[0218] A signal processing link provided in an embodiment of the present application can be applied to electromagnetic wave devices. The transmission link may include an analog signal source and a digital phase shifter. The analog signal source can be configured to provide an initial analog signal (such as an LO signal), and the digital phase shifter can be configured to provide a phase-shifted signal generated in the digital domain. The digital phase shifter can also phase-shift the initial analog signal based on the generated phase-shifted signal to perform a preset phase shift operation on the initial analog signal.

[0219] As shown in FIG12B , in some optional embodiments, a signal transmission link with a digital phase shifter architecture may include an analog signal source 141, a digital phase shifter (Digital PS) 142, and a transmitting antenna 143. Specifically, the analog signal source 141 may be configured to provide an LO signal, and the digital phase shifter 142 may be configured to perform a predetermined phase shift on the received LO signal, so that the phase-shifted LO signal is radiated into a predetermined spatial region via the transmitting antenna 143. The analog signal source 141 may also include a phase-locked loop (PLL) architecture that provides electromagnetic wave (e.g., laser, microwave, etc.) signals. The analog signal source 141, digital phase shifter 142, and transmitting antenna 143 may be integrated into a single device or may be separate components. For example, the analog signal source 141 and digital phase shifter 142 may be integrated into a package, such as a SoC chip, while the transmitting antenna 143 may be connected via a peripheral port of the chip and formed on a substrate, such as a PCB. At the same time, in some optional embodiments, the transmitting antenna 143 can also be integrated on the chip package to form AiP or AoP, with a chip structure having a packaged antenna.

[0220] As shown in Figure 12B, in an exemplary instance, the digital phase shifter 142 may include a mixer (Mixer) 1421, a digital-to-analog converter (i.e., DAC) 1422 and a phase-shifted signal source (such as a digital baseband signal source Baseband) 1423, etc., that is, the phase-shifted signal source 1423 may be configured to provide a digital phase-shifted signal; the digital-to-analog converter 1422 may be configured to perform analog-to-digital conversion on the received digital phase-shifted signal to convert the digital phase-shifted signal into an analog phase-shifted signal; the mixer 1421 may be configured to perform a mixing operation on the received analog phase-shifted signal with the received transmission signal from the analog signal source 141 to achieve a phase-shifting operation of setting the above-mentioned transmission signal using the digital phase-shifted signal. Optionally, when the above-mentioned signal transmission link provides a swept frequency signal, such as providing an FMCW laser signal or an FMCW microwave signal, a swept frequency transmission signal can be provided based on the analog signal source 141, and / or a swept frequency digital phase-shifted signal can be provided based on the phase-shifted signal source 1423, so that after mixing by the mixer 1421, a swept frequency continuous wave signal is output.

[0221] In some optional embodiments, based on the structure shown in Figure 12B, the analog signal source 141 can be configured to provide an FMCW signal in the centimeter wave band or millimeter wave band (such as 3.1GHz, 24GHz, 60GHz, 77GHz, 120GHz, 250GHz, etc.) in the microwave, and the phase-shifted signal source 1423 can be configured to provide a digital phase-shifted signal at the MHz level (for example, 3MHz to 5MHz, such as 3MHz, 4MHz, 5MHz, etc.), that is, the digital-to-analog converter 1422 performs digital-to-analog conversion on the MHz-level digital phase-shifted signal to obtain an analog phase-shifted signal in the corresponding frequency range, and the mixer 1421 can be configured to perform up-mixing or down-mixing operations on the received FMCW signal in the millimeter wave band based on the received analog phase-shifted signal of the fixed frequency band, so as to realize a preset phase shift operation on the FMCW signal.

[0222] In some optional embodiments, the centimeter wave signal in the 3.1GHz frequency band may include 3.1GHz to 10.6GHz, such as 3.1GHz, 5GHz, 5GHz, 6GHz, 8GHz, 10.6GHz, etc.; the millimeter wave signal in the 77GHz frequency band may include 76GHz to 81GHz signals, such as 76GHz to 77GHz, 77GHz to 79GHz, 79GHz to 81GHz and other swept frequency signals, or 76GHz, 77GHz, 78GHz, 79GHz, 80GHz, 81GHz and other fixed frequency band signals.

[0223] Based on the structure shown in FIG12B , since phase-shift signal source 1423 provides a digital signal, to further adapt the signal characteristics, mixer 1421 can be configured as an IQ Mixer and digital-to-analog converter 1422 as an IQ DAC. Furthermore, phase-shift signal source 1423 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and / or to provide a corresponding source signal as a waveform controller.

[0224] In one embodiment, as shown in FIG12C , a signal transmission link (TX digital phase shifter architecture) of a digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (DDFS), an IQ digital to analog converter (DAC), a low-pass filter (LPF), an IQ modulator (IQ modulator / IQ mixer), a power amplifier (PA), etc., that is, the baseband signal source is configured to provide a digital phase-shifted source signal, and the direct digital frequency synthesizer may be configured to implement code division multiplexing (CDM), Doppler division multiplexing (DDM), time division multiplexing (TDM), space division multiplexing (SDM), circuit switch data (CSD), digital intermediate frequency (DIF), etc. based on the received source signal. Frequency) and other signal waveforms and transmission modes to achieve flexible configuration of signal transmission form and transmission waveform. The signal amplified by the power amplifier can be radiated to a predetermined spatial area via an integrated or external transmitting antenna.

[0225] Regarding the signal transmission link of the digital phase shifter architecture in the embodiment of the present application, since the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, and can generate an analog baseband signal through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower sidelobes, so its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.

[0226] In some optional embodiments, when using RF LO frequency sweeping to achieve an FMCW signal, a compensation module can be added to the signal transmission link of a digital phase shifter architecture to address issues such as TX IQ imbalance, signal leakage (e.g., TX LO leakage), and harmonic distortion (HD) caused by IQ mismatch. As shown in Figure 12C , a compensation module (TX compensation) can be provided between the TX DDFS and the IQDAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture, thereby addressing at least one of the aforementioned issues. The HD caused by baseband third-order nonlinearity can be referred to as HD3.

[0227] In some optional embodiments, as shown in Figures 12C and 12D, the compensation module (TX compensation) may include at least one of an LO (LO) compensation unit, an IQ (IQ) compensation unit, and an HD3 (HD3) compensation unit. Specifically, the LO compensation unit may be configured to compensate for signal leakage, the IQ compensation unit may be configured to compensate for IQ imbalance, and the HD3 compensation unit may be configured to compensate for HD3. The LO compensation unit may be configured to compensate for at least one of IQ modulator imbalance and IQ channel imbalance. Furthermore, when the compensation module includes at least two of the LO, IQ, and HD3 compensation units, compensation may be performed simultaneously (e.g., in parallel) or sequentially (e.g., in series) based on actual needs and signal characteristics. For example, as shown in Figure 4D, IQ compensation may be performed first, followed by LO compensation, and finally HD3 compensation.

[0228] In some optional embodiments, the signal transmission link of the digital phase shifter architecture may also include a TX DAC Board Error Correction module for the DAC and an AWGN (additive white gaussian noise) module for Gaussian white noise, etc., which are not shown in the figure and may be added or deleted according to actual needs. In the embodiments of the present application, I in IQ can be represented by the abbreviation of In-Phase (i.e., in-phase), Q can be represented by the abbreviation of Quadrature (i.e., orthogonal), and RF can be represented by the abbreviation of Radio Frequency (i.e., radio frequency).

[0229] In an optional embodiment, IQ Imbalance compensation can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation method is not affected by the IQ Imbalance calibration method. LO Leakage compensation can be achieved by adjusting the DC component (i.e., DC offset) of both the IQ and Q channels. Similarly, this LO Leakage calibration method has no impact on the compensation scheme. Regarding HD3, since the third-order harmonic distortion of the orthogonal mixer V / IConverter is the primary source of HD3, and harmonic distortion is affected by DC offset, when both LO Leakage and HD3 of the transmit chain require calibration, HD3 calibration should be performed after LO Leakage calibration to ensure accurate HD3 calibration performance.

[0230] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency multiplication waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process.

[0231] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO leakage can be calibrated and compensated first, and then the root cause of the HD3 problem, namely the HD3 compensation coefficient, can be calibrated under a stable DC bias. Subsequently, IQ imbalance is calibrated and further compensated for, and then the third harmonic distortion is compensated for both the IQ and Q channels based on the results of IQ imbalance pre-compensation.

[0232] In an optional embodiment, for a digital pre-compensation architecture based on a frequency-doubling waveform generator module, after calibrating and compensating for LO leakage, the HD3 compensation coefficient can be calibrated to obtain the HD3 compensation coefficient, and the IQ imbalance can be calibrated and compensated under a stable DC bias. Subsequently, based on the compensation results, the actual waveforms of the IQ signals and the HD3 compensation coefficient are calculated, and the waveform information of the 3x and 5x frequencies that require pre-compensation is calculated in reverse.

[0233] In another optional embodiment, for a digital precompensation architecture based on a frequency-doubling waveform generator module, LO leakage can be calibrated and compensated first. Then, multiple (e.g., three) observations are performed to simultaneously calibrate the precompensation coefficients for HD3 and IQ Imbalance. Then, further observations are performed (e.g., two) to calibrate the precompensation coefficients at the HD3 mirror position. Finally, the 3rd and 5th frequency coefficients requiring precompensation are calculated from the precompensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in this disclosed embodiment represent operations such as testing and comparative analysis of different test results.

[0234] FIG12E is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present application. As shown in FIG12E , the transmission path (Transmitter) may include a phase shift module PS, an amplifier PA, a power detector PD, etc. For example, the transmission path may adopt the transmission link of the digital phase shifter architecture (Digital Phase Shifter) described in any embodiment of the present application. For details, please refer to the relevant figures and text descriptions, which will not be repeated here. Because the transmission path adopts a digital phase shifter architecture, while achieving more precise phase shifting operations, the transmission channel can simultaneously support multiple modes such as DDM and frequency division multiplexing (FDM) of multiple antennas. It can also eliminate the calibration operation of the RF phase shifter (Phase Shifter), reduce the isolation and coupling degree in the phase shift system, and reduce link loss and production costs. In addition, to address possible TX IQ mismatch and LO leakage issues, the transmit path of this digital phase shifter architecture can also support RF frequency response compensation, IQ imbalance, and LO leakage calibration in the digital domain.

[0235] To address issues such as TX IQ mismatch, LO leakage, and frequency response in the transmit path, an auxiliary receiver (ARX) can be configured to perform relevant calibration and compensation operations. As shown in Figure 12E , the ARX can include a mixer, TIA, LPF, HPF, IQ ADC, adder, and RF calibration module (RF Calib) connected in sequence. One input of the mixer receives the ARX IQ LO signal, while the other input is connected along the signal transmission direction (i.e., the direction of the arrow in the figure) to a node before the transmit path PD or to any node after the phase shifter (module). For example, it can be connected to the output of the PA (synchronously calibrating the PA) or the input of the PA, thereby calibrating the transmit path through the ARX. The LO signal frequency in the transmit path and the ARX IQ LO signal frequency have a set difference frequency, so that there is a frequency offset between the two signals, simulating a real transmit / receive signal loop.

[0236] In an optional embodiment, to further improve calibration accuracy, a corresponding calibration circuit (i.e., a calibration receiving unit) may be provided for the ARX (i.e., the auxiliary receiving unit). For example, the RF tone signal generation circuit (RF tone generator) shown in FIG12E may include a TX DDFS, an adder, a real DAC, an LPF, an amplifier, a multiplier, and a bandpass filter (BPF) connected in sequence. The adder may be configured to calibrate and compensate for TX LO leakage, the multiplier may be configured to compensate for RF tone generator LO leakage, and the BPF may be configured to filter out DC signals generated by LO leakage from the RF tone generator. In other words, the RF tone generator may be configured to generate multiple stable tone signals of different frequencies to implement calibration operations for the ARX.

[0237] In some optional embodiments, as shown in FIG12E , the ARX may be calibrated using an RF Tone Generator first, and then the calibrated ARX may be used to calibrate a transmitter channel including a PA, for example, calibrating the PD at the PA output, the phase shifter in the transmit channel, the total gain from the DAC to the PA output, and the frequency response and other devices and circuits.

[0238] In one embodiment, as shown in FIG12E , a plurality of stable single-tone signals of different frequencies may be generated by using an RF Tone Generator to assist in calibrating the ARX. Then, based on the calibrated ARX, the IQ imbalance, local oscillator leakage, inconsistent frequency response, and other issues of the transmit path TX may be calibrated.

[0239] As shown in FIG16 , an embodiment of the present disclosure further provides another signal processing method, which is applied to an antenna array of an electromagnetic wave device having at least two signal transmission links, including:

[0240] Step 1601: Determine a reference transmission delay and a delay difference between the signal transmission delay of each transmission channel and the reference transmission delay.

[0241] In some exemplary embodiments, the reference transmission delay includes a first reference transmission delay and / or a second reference transmission delay, wherein the first reference transmission delay can be the signal transmission delay of the transmitting antenna feed line of any transmitting channel, and the second reference transmission delay can be the signal transmission delay of the signal line from the local oscillator to the phase shifter of any transmitting channel.

[0242] In other exemplary embodiments, the reference transmission delay includes a first reference transmission delay and / or a second reference transmission delay, wherein the first reference transmission delay may be the signal transmission delay of the shortest transmitting antenna feed line in multiple transmission channels, and the second reference transmission delay may be the signal transmission delay of the shortest signal line from the local oscillator to the phase shifters in the multiple transmission channels.

[0243] In an exemplary embodiment, the distance of the transmitting antenna feed line and / or the signal line from the local oscillator to the phase shifter in each transmitting channel may be measured. The distance measurement may include model simulation measurement or actual measurement.

[0244] Step 1302: Generate a compensation signal corresponding to each transmit channel according to the delay difference;

[0245] In one exemplary embodiment, the frequency control parameter of the compensation signal Phase control parameters Among them, f c is the starting frequency of the chirp signal, β is the frequency variation range of the chirp signal, T is the sweep duration of the chirp signal, and μ is the delay difference between the signal transmission delay of the transmitting channel and the reference transmission delay.

[0246] Step 1303: Determine the transmission channel corresponding to the currently received echo signal, and use the corresponding compensation signal to compensate the currently received echo signal.

[0247] In an exemplary embodiment, the receiving channel includes an I-channel transmitting channel and a Q-channel transmitting channel, and the compensation signal includes a cosine compensation signal and sinusoidal compensation signal The currently received echo signal is compensated using one of the cosine compensation signal and the sine compensation signal and then output to the I-channel receiving channel Y_I. The currently received echo signal is compensated using the other of the cosine compensation signal and the sine compensation signal and then output to the Q-channel receiving channel Y_Q.

[0248] Another signal receiving and transmitting method provided in an embodiment of the present application generates a compensation signal through a signal compensator according to the signal transmission delay corresponding to the currently received echo signal, and uses the compensation signal to compensate for the currently received echo signal, thereby improving the quality of the received signal and thereby improving the detection accuracy and detection range of the radar system.

[0249] In a radar system, a series of continuous linear frequency modulated pulse signals are emitted through the transmitting antenna, and the echo of the pulse signal is received by the receiving antenna. After processing, the detected target information can be output. This process is generally called a frame. The radar system detects surrounding moving or stationary objects in real time through continuous frame processing.

[0250] In millimeter-wave radar systems, to synthesize a large-aperture receive antenna array and improve the radar system's angular resolution, multiple transmit antennas are typically used to create a larger number of virtual receive arrays, increasing the effective receive aperture. This is known as a Multiple-Input, Multiple-Output (MIMO) radar system. Doppler Division Multiplexing (DDM) technology can separate the signals transmitted by different transmit antennas. However, DDM requires modulating an initial phase for each transmit antenna that steps with the chirp (a linear frequency-modulated continuous wave). This is typically achieved using phase shifters in the RF circuit. Phase shifters typically have phase shift errors and also affect the amplitude of the pulse signal. Therefore, when different phase shift values ​​are set for the transmit antennas, the pulse signal amplitude will fluctuate slightly. When the phase shift error and amplitude fluctuation are large, accurate target detection cannot be achieved.

[0251] Therefore, phase shifter calibration plays an important role in the performance of DDM MIMO radar systems. The current calibration method is to find the appropriate phase shifter settings under different operating parameters and temperatures so that the phase shifter output is closest to the ideal phase shift value, thereby achieving the purpose of phase shifter calibration. However, there is still a residual error between the calibrated phase shifter value and the ideal phase shift value, which cannot effectively improve the performance of the radar system.

[0252] The present invention provides a target detection method for a DDM radar sensor having at least one transceiver channel. The method can compensate for phase shifter errors, improve the accuracy of the radar system's angle calculation, and thus improve the accuracy of detected targets. As shown in Figure 18, the target detection method provided by the present invention can include:

[0253] Step 1801: Perform range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and use the phase shift error coefficient of the transmitting channel to perform phase shift compensation on the range-dimensional FFT data to obtain range-dimensional FFT compensated data.

[0254] In DDM MIMO radar systems, pre-calibration is typically used. This involves finding the appropriate phase shifter, adapted to different operating parameters and temperatures, to achieve an output close to the ideal phase shift value for the chirp signal transmitted by the transmit channel. This reduces the error introduced by the phase shifter. However, in reality, there is still a residual error between the calibrated phase shift value and the ideal phase shift value.

[0255] Each transmitting antenna has a phase shifter, and the phase shift errors of different transmitting antennas are different. This error has no effect on the range-dimensional FFT data, but only affects the accuracy of the Doppler-dimensional FFT data. Therefore, an embodiment of the present application proposes that after performing a range-dimensional Fourier transform (also called "1D FFT") on the echo signals received by each receiving channel to obtain the range-dimensional FFT data (also called "1D FFT data"), the compensation coefficient of each transmitting channel is used to perform phase shift compensation on the range-dimensional FFT data to effectively eliminate the influence of the phase shifter error, thereby obtaining accurate range-dimensional FFT data.

[0256] The target detection method provided in the embodiments of this application compensates for the phase shift error of the phase shifter of the transmitting antenna by performing phase compensation processing on the range-dimensional FFT data. The specific operation process for performing phase compensation using the phase error coefficient, the specific coefficients included in the phase error coefficient, and the method and steps for obtaining the phase error coefficient are not limited. These details will be explained through multiple embodiments in subsequent embodiments and are not further elaborated here.

[0257] Step 1802: Perform Doppler-dimension Fourier transform processing on the range-dimension FFT compensation data to obtain Doppler-dimension FFT data.

[0258] After obtaining the distance-dimensional FFT compensation data, Doppler-dimensional Fourier transform (also called "2D FFT") processing may be performed on the data to obtain Doppler-dimensional FFT data (also called "2D FFT data" or "velocity-dimensional FFT data").

[0259] Step 1803: Determine the velocity information of the target object based on the Doppler FFT data.

[0260] By performing constant false alarm rate (CFAR) on the Doppler FFT data, false targets are removed from the energy dimension. At this time, the target information contains information in the distance and velocity dimensions.

[0261] The target detection method provided in the embodiment of the present application can obtain accurate range-dimensional FFT data by performing phase shift compensation on the range-dimensional FFT data using the compensation coefficients of each transmission channel after obtaining the range-dimensional FFT data; then, the Doppler-dimensional FFT data is obtained through the compensated range-dimensional FFT data, and the speed information of the target object determined based on the Doppler-dimensional FFT data is accurate speed information, thereby achieving accurate detection of the target object.

[0262] The phase shift compensation process in the target detection method in the above embodiment will be described below through multiple embodiments.

[0263] In some embodiments, the phase shift error coefficient of the transmission channel can be used to perform phase shift compensation on the range-dimensional FFT data, including: for data corresponding to each chirp sequence in the range-dimensional FFT data, using the phase shift error coefficient of the transmission channel corresponding to each chirp sequence to perform phase shift compensation.

[0264] Taking a DDM radar sensor with two transmit channels (corresponding to two transmit antennas TX0 and TX1) as an example, Figure 19a shows a schematic diagram of chirp stepped phase modulation for two transmit antennas. In the chirp sequence transmitted by transmit antenna TX0, each chirp is not phase-modulated by a phase shifter (this can also be considered as a phase sequence modulated with a step of 0 degrees), so its echo signal has no velocity offset. In the chirp sequence transmitted by transmit antenna TX1, the modulation phase step is an initial phase of θ = π / 2. When the modulated phase error is zero, the receiver performs a two-dimensional Fourier transform (2D FFT) on the received echo signal, resulting in a 2D FFT plane as shown in Figure 19b. The two peaks on the 2D FFT plane are the 2D FFT peaks formed by the echo signals transmitted by TX0 and TX1 and reflected by targets at a certain distance and velocity. According to the pre-modulated stepped phases of TX0 and TX1, the distance between the two peaks in the Doppler dimension is known, and thus can correspond to TX0 and TX1, achieving the purpose of multiplexing and smooth separation in the Doppler dimension.

[0265] From this, it can be seen that, in principle, as long as the echo signals of TX0 and TX1 are obtained, the phase shift error coefficient of the transmission channel corresponding to TX0 can be used to perform phase shift compensation for the data corresponding to the chirp sequence sent by TX0 in the range gate FFT data obtained by performing 1D FFT processing on the echo signals, and the phase shift error coefficient of the transmission channel corresponding to TX1 can be used to perform phase shift compensation for the data corresponding to the chirp sequence sent by TX0.

[0266] However, in the range-gate FFT data, the data corresponding to the chirp sequences belonging to TX0 and TX1 can only be separated through the Doppler dimension obtained after 2D FFT processing. Therefore, when compensating the range-gate FFT data, it is possible not to distinguish the data corresponding to the two chirp sequences. That is, the phase shift error coefficients corresponding to the two transmission channels (TX0 and TX1) are used to simultaneously perform phase shift compensation on the entire obtained range-gate FFT data. This covers the operation of performing phase shift compensation on the data corresponding to each chirp sequence in the range-dimension FFT data using the phase shift error coefficients of the transmission channels corresponding to each chirp sequence. Then, from the final Doppler FFT data, the data corresponding to the chirp sequences corresponding to the two transmission channels (TX0 and TX1) are separated. This allows the phase-compensated range-gate FFT data corresponding to each Doppler FFT data (range-gate FFT compensation data) to be distinguished as the data obtained by phase-compensating the range-gate FFT data corresponding to the chirp sequence sent by which transmission channel using the phase shift error coefficient of which transmission channel.

[0267] In an exemplary embodiment, the range-dimensional FFT data is composed of multiple range-gate FFT data. Accordingly, for data corresponding to each chirp sequence in the range-dimensional FFT data, phase shift compensation is performed using the phase shift error coefficients of the transmission channels corresponding to each chirp sequence. This may include: performing phase shift compensation on each range-gate FFT data using the phase shift error coefficients of the transmission channels corresponding to each chirp sequence, thereby obtaining range-gate FFT compensation data corresponding to each range-gate FFT data. Correspondingly, performing Doppler-dimensional Fourier transform processing on the range-dimensional FFT compensation data to obtain Doppler-dimensional FFT data may include: performing Doppler-dimensional Fourier transform processing on each range-gate FFT compensation data to obtain Doppler-dimensional FFT data corresponding to each range-gate FFT compensation data.

[0268] In one embodiment, FIG20a is a schematic diagram of a process of performing distance dimension FFT processing on a single receiving channel. In FIG20a, the echo signal received by the receiving channel, i.e., the chirp sequence (C0, C1, ... C N) contains the chirp sequences of the two transmission channels corresponding to TX0 and TX1. By performing distance-dimensional FFT processing on the echo signal, the distance-dimensional FFT data (P0, P1, ...P N-1 ), forming a 1D FFT data plane diagram as shown in Figure 20b. The 1D FFT data plane contains two dimensions: the range dimension and the Doppler dimension. The range dimension includes M range gates, each of which contains the range-dimensional FFT data of chirps with the same distance but different numbers. The Doppler dimension includes N Doppler gates, each of which contains the range-dimensional FFT data of a chirp pulse.

[0269] Combining FIG20a and FIG20b, it can be seen that the distance dimension FFT data P0, P1, ...P corresponding to each receiving channel N-1 With chirp sequence C1, C2, ...C N is corresponding, and the chirp sequence C1, C2, ...C N The chirp sequences corresponding to the two transmit channels, TX0 and TX1, are included. The range-dimensional FFT data is composed of multiple range-gate FFT data. Therefore, when performing phase compensation on the range-dimensional FFT data, phase compensation can be performed individually for each range gate as shown in Figure 20b.

[0270] In one embodiment, as shown in FIG21a , taking the Mth range gate as an example, phase shift compensation is performed on the 1D FFT data of the Mth range gate (also referred to as the "Mth range gate data slice") along the Doppler dimension using the phase shift error coefficients of the transmit channels of transmit antennas TX0 and TX1, respectively, to obtain two corresponding range gate FFT compensated data slices. Because phase shift compensation is performed on each range gate FFT data slice using each transmit channel separately, multiple range gate FFT compensated data slices, equal in number to the number of transmit channels, can be obtained for each range gate FFT data slice.

[0271] Thus, after subsequent 2D FFT processing of the range gate FFT compensated data slices, multiple 2D FFT data slices equal in number to the number of transmit channels can be obtained, such as 2D FFT data slice 0 and 2D FFT data slice 1 in FIG. 21a . Since 2D FFT data slice 0 and 2D FFT data slice 1 each contain data corresponding to the chirp sequences of two transmit channels (TX0, TX1), after obtaining the 2D FFT data slices, the data corresponding to each transmit channel can be separated, thereby separating the Doppler-dimensional FFT data obtained after phase shift compensation of the range-dimensional FFT data of a particular transmit channel using the phase shift compensation coefficient of that transmit channel.

[0272] In one embodiment, as shown in FIG21b, still taking the Mth range gate as an example, the Mth range gate data slices P0, P1, ...P N-1 Using phase error coefficients W0, W1, ...W N-1 Perform phase shift compensation. For example, the compensation algorithm is to multiply the two bits in a phase-wise manner to obtain 1D FFT data slices Q0, Q1, ...Q after phase shift compensation. N-1 Then, a 2D FFT is performed on the compensated data to obtain a 2D FFT data slice. This data slice contains two peaks, one corresponding to each of the two transmit channels (the 2D FFT peaks corresponding to TX0 and TX1). By determining the transmit channel to which the phase error coefficient used in this phase compensation process belongs, it is possible to determine which transmit channel's phase error coefficient was used to perform phase compensation on the range gate FFT data of which transmit channel, and the corresponding 2D FFT data.

[0273] In DDM radar sensors, the process of determining target object velocity information based on Doppler FFT data varies depending on the number of transmitting and receiving channels in the transceiver channel. This is because when there are two or more receiving channels, it is necessary to consider merging the 2D FFT data from different receiving channels.

[0274] Taking a receiving channel as an example, for example, when the transceiver channel is composed of at least two transmitting channels and one receiving channel, determining the speed information of the target object based on Doppler dimension FFT data may include: determining the speed information of the target object based on all Doppler gate FFT data.

[0275] In the foregoing embodiment, the processing process of multiple Doppler gate FFT data obtained from one range gate FFT data in the phase-shift-compensated 2D FFT data formed by the transmitting channels corresponding to two transmitting channels TX0 and TX1 and one receiving channel (a single receiving channel) has been described. Since it is a single receiving channel, there is no need to consider merging the 2D FFT data for a particular transmitting channel between channels. Therefore, after obtaining the multiple Doppler gate FFT data corresponding to each range gate FFT data, CFAR processing can be performed based on all the Doppler gate FFT data to remove false targets from the energy dimension, thereby obtaining information on the distance and velocity of the target object.

[0276] When the transceiver channel consists of two or more onboard receiving channels, it is necessary to consider merging the 2D FFT data from different receiving channels. For example, when the transceiver channel consists of at least two transmit channels and at least two receive channels, determining the target object's velocity information based on Doppler FFT data can be accomplished through the following steps a through d.

[0277] Step a: extracting Doppler gate FFT data from each receiving channel respectively, wherein the Doppler gate FFT data from each receiving channel includes Doppler gate FFT data compensated by phase shift error coefficients of different transmitting channels.

[0278] Step b: combining the Doppler gate FFT data in all receiving channels that have been compensated by the phase shift error coefficient of the same transmitting channel to obtain multiple groups of Doppler gate FFT data corresponding to each transmitting channel.

[0279] Step c: for each set of Doppler gate FFT data, extract target Doppler gate FFT data therefrom, where the target Doppler gate FFT data is the echo signal of the transmission channel corresponding to the set of Doppler gate FFT data.

[0280] Step d: synthesizing the target Doppler gate FFT data extracted from each set of Doppler gate FFT data, and determining the velocity information of the target object based on the target Doppler gate FFT data extracted from each set of Doppler gate FFT data.

[0281] In one embodiment, as shown in FIG22 , in the 2D FFT data slices (Doppler gate FFT data) of multiple receiving channels, each slice covers data from multiple receiving channels and multiple transmitting channels. Therefore, it is necessary to perform multi-channel merging detection on each 2D FFT data slice to determine the 2D FFT data of each transceiver channel (one receiving channel + one transmitting channel constitutes one transceiver channel), and then extract the FFT vectors of the target echo of the transmitted signal of each transmitting channel. Then, the FFT vectors of different transmitting channels are synthesized into a MIMO virtual array to form the FFT vector of the receiving channel. Then, after the target angle is solved based on the synthesized FFT vector, the velocity information of the target object is determined.

[0282] In step a, for example, the processing shown in Figures 21a and 21b can be employed. For each range gate data slice of each receiving channel, for example, for the Mth range gate FFT data slice, two Doppler gate FFT data slices can be obtained. One Doppler gate FFT data slice is obtained by performing phase shift compensation based on the phase error coefficient of TX0, denoted as 2D FFT data slice 0, and the other Doppler gate FFT data slice is obtained by performing phase shift compensation based on the phase error coefficient of TX1, denoted as 2D FFT data slice 1. In other words, for the Doppler gate FFT data of an entire receiving channel, two types of Doppler gate FFT data can be obtained by extracting the phase error coefficients of different transmitting channels used in the phase shift compensation process: one type of Doppler gate FFT data is obtained by performing phase shift compensation based on the phase error coefficient of TX0, and the other type of Doppler gate FFT data is obtained by performing phase shift compensation based on the phase error coefficient of TX1.

[0283] In step b, after obtaining Doppler gate FFT data for each receiving channel and classifying and extracting it according to the phase error coefficients of the different transmitting channels used in the phase compensation process, the extracted Doppler gate FFT data, which has been compensated for the phase error coefficients of the same transmitting channel, can be merged to obtain multiple sets of Doppler gate FFT data corresponding to the corresponding transmitting channels. For example, for each M-th range gate shown in FIG21a, all 2D FFT data slices 0 obtained by performing phase shift compensation based on the TX0 phase error coefficients of each receiving channel are merged to obtain 2D FFT data slices 0 for the multiple receiving channels (shown in FIG22). Similarly, all 2D FFT data slices 1 obtained by performing phase shift compensation based on the TX1 phase error coefficients of each receiving channel are merged to obtain 2D FFT data slices 1 for the multiple receiving channels (shown in FIG22).

[0284] It should be noted that in 2D FFT data slice 0 of multiple receiving channels, although all Doppler gate FFT data is obtained by phase-shift compensation based on the TX0 phase error coefficient, since there is 2D FFT data corresponding to the echo signals of two transmitting channels, there are two phase-shift compensation scenarios when performing phase-shift compensation based on the TX0 phase error coefficient: one is phase-shift compensation performed on the echo signal of the TX0 transmitting channel based on the TX0 phase error coefficient, and the other is phase-shift compensation performed on the echo signal of the TX1 transmitting channel based on the TX0 phase error coefficient. Obviously, only the first scenario, that is, phase-shift compensation performed on the echo signal of the TX0 transmitting channel based on the TX0 phase error coefficient, is effective phase-shift compensation. The corresponding 2D FFT data slice 0 is the valid Doppler gate FFT data.

[0285] Similarly, in 2D FFT data slice 1 of the multi-receive channel, although all Doppler gate FFT data is obtained by phase-shift compensation based on the TX1 phase error coefficient, since there is 2D FFT data corresponding to the echo signals of two transmit channels, there are two phase-shift compensation scenarios when performing phase-shift compensation based on the TX1 phase error coefficient: one is phase-shift compensation performed on the echo signal of the TX0 transmit channel based on the TX1 phase error coefficient, and the other is phase-shift compensation performed on the echo signal of the TX1 transmit channel based on the TX1 phase error coefficient. Obviously, only the second scenario, that is, phase-shift compensation performed on the echo signal of the TX1 transmit channel based on the TX1 phase error coefficient, is effective phase-shift compensation. The corresponding 2D FFT data slice 1 is the valid Doppler gate FFT data.

[0286] Therefore, in step c, valid Doppler gate FFT data must be extracted from each set of Doppler gate FFT data corresponding to each transmit channel. Specifically, for each set of Doppler gate FFT data, it is necessary to determine which Doppler gate FFT data are echo signals from the corresponding transmit channel (this can be achieved through multi-channel combined detection as shown in FIG. 22 ). These Doppler gate FFT data are then used as target Doppler gate FFT data and extracted from the set of Doppler gate FFT data.

[0287] As shown in FIG. 22 , in one embodiment, for each set of Doppler gate FFT data corresponding to the transmit channel of TX0, the portion of the Doppler gate FFT data that serves as the echo signal of the transmit channel of TX0 needs to be extracted as the target Doppler gate FFT data (the corresponding operation is to extract the FFT vector of the target echo of TX0). For each set of Doppler gate FFT data corresponding to the transmit channel of TX1, the portion of the Doppler gate FFT data that serves as the echo signal of the transmit channel of TX1 needs to be extracted as the target Doppler gate FFT data (the corresponding operation is to extract the FFT vector of the target echo of TX1).

[0288] In step d, all target Doppler gate FFT data are combined. This means that the Doppler gate FFT data obtained from all receive channels based on effective phase shift compensation are combined to obtain a MIMO virtual array, which serves as the 2D FFT data vector for multiple receive channels (the corresponding operation is to synthesize the MIMO virtual array and obtain the receive FFT vector in Figure 22). The combined 2D FFT data vector is then subjected to CFAR processing to remove false targets from the energy dimension. At this point, the target information contains information in the range and velocity dimensions.

[0289] The steps for obtaining the phase shift error coefficient will be described below through at least one embodiment.

[0290] In some embodiments, the phase shift error coefficient of the transmission channel can be obtained by comparing the chirp sequence corresponding to the transmission channel with a standard chirp sequence generated by a standard phase shifter to obtain the phase shift error coefficient of the transmission channel.

[0291] In one exemplary embodiment, the chirp sequence transmitted by each transmit channel of a DDM radar sensor under modulation by a built-in phase shifter can be pre-detected. A standard chirp sequence can then be generated using a standard phase shifter. The chirp sequence corresponding to each transmit channel is then compared with the standard chirp sequence to determine a compensation relationship between the two and obtain the phase shift error coefficient for the transmit channel. The compensation relationship can be, but is not limited to, a multiplication or addition relationship.

[0292] In some embodiments, using the phase shift error coefficient of the transmission channel to perform phase shift compensation on the range-dimensional FFT data to obtain the range-dimensional FFT compensation data may include: multiplying or adding the phase shift error coefficient of the transmission channel with the range-dimensional FFT data to obtain the range-dimensional FFT compensation data.

[0293] For example, in Figure 21b, for the Mth range gate slice in the 1D FFT data plane, it can be multiplied by the phase shift error coefficient to obtain the compensated 1D FFT data slice. In some embodiments, the phase shift error coefficient may include a phase shift error and / or a phase shift amplitude error.

[0294] The above method will be supplemented by at least one embodiment below, adding a processing procedure for determining the angle information of the target object.

[0295] In some embodiments, the target detection method may further include determining the target object's angle information based on the Doppler FFT data. In one embodiment, when radar detects a target object, in addition to detecting the target object's position and velocity, another important detection aspect is determining the target object's angle information. Specifically, this is the angle information of the target object relative to the monitoring point. In some embodiments, determining the target object's angle information based on the Doppler FFT data may include extracting and compensating the complex data at the peak of the Doppler FFT data generated by the corresponding transmit channel from the Doppler FFT data, and synthesizing the received vector to determine the target object's angle information.

[0296] For example, in Figure 22, after obtaining the target 2D FFT data slices corresponding to TX0 and TX1, the complex values ​​at the 2D FFT peaks corresponding to TX0 and TX1 are extracted from these two sets of 2D FFT data slices and combined into a single receive vector. This expands the receive array and improves angular resolution. Because these 2D FFT complex values ​​are accurate after appropriate compensation, accurate angle information of the target object can be obtained through angle calculation.

[0297] In some embodiments, the transmission link corresponding to the transmission channel in the embodiments of the present application may include an analog signal source and a phase shifter. The analog signal source may be configured to provide an initial analog signal, and the phase shifter may be configured to generate a phase-shifted signal and perform phase shifting on the initial analog signal based on the phase-shifted signal, so as to perform a preset phase shifting operation on the initial analog signal, thereby realizing the DDM wave transmission mode; wherein, the phase shifter may be a delay line phase shifter or a digital phase shifter.

[0298] In some embodiments, when the phase shifter is a digital phase shifter, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter, and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal, the digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal, and the mixer can be configured to use the received analog phase shift signal to perform a mixing operation on the received initial analog signal to perform a preset phase shift operation on the initial analog signal.

[0299] An embodiment of the present application provides a signal processing link, including a transmitting link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal; the transmitting link includes an analog signal source and a digital phase shifter, the analog signal source can be configured to provide an initial analog signal, the digital phase shifter can be configured to generate a digital phase-shifted signal, and based on the digital phase-shifted signal, phase-shift the initial analog signal to perform a preset phase-shift operation on the initial analog signal; the receiving link includes an analog-to-digital converter and a digital baseband processing module, the analog-to-digital converter can be configured to perform analog-to-digital conversion on the received echo signal to obtain a digital baseband signal, and the digital baseband signal processing module can be configured to perform distance-dimensional Fourier transform and velocity-dimensional Fourier transform on the digital baseband signal in sequence; wherein the digital baseband processing module includes a phase shift compensation unit, the phase shift compensation unit can be configured to adopt the phase shift error coefficient of the digital phase shifter in the transmitting link, and obtain phase shift compensation for distance-dimensional FFT data by performing distance-dimensional Fourier transform; and the digital baseband processing module can be configured to perform velocity-dimensional Fourier transform based on the compensated distance-dimensional FFT data.

[0300] In the transceiver links of the present embodiment, the TX-LO signal received by the IQ modulator in the transmit link and the RX-LO signal received by the real mixer in the receive link can have the same frequency. For example, as shown in FIG12B , if the baseband output signal is a sine wave at x MHz, then both the TX-LO signal and the RX-LO signal can be sine waves at z GHz.

[0301] In some embodiments, the transmitting link may adopt a digital phase-shifting architecture, while the receiving link may adopt analog architecture components, that is, IQ components are not required, so it can effectively be compatible with the sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system. Optionally, in an embodiment of the present application, the receiving link may include a receiving antenna, that is, the receiving antenna can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the receiving antenna can also be integrated into the chip package to form AiP, AoP, RoP and other structures, that is, a chip structure with a packaged antenna.

[0302] In some optional embodiments, in order to match the transmitting chain of the digital phase shifter architecture, corresponding adjustments may be made to the receiving chain. For example, the transceiver chain shown in FIG12C may include a transmitting chain architecture and a receiving chain similar to those in FIG12B (to avoid redundancy, the same parts will not be described in detail here), that is, the Real Mixer in the receiving chain in FIG12B is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. In this case, the receiving chain may include a low noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), an IQ digital-to-analog converter (IQ ADC), etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is sequentially passed through the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ After ADC processing, it is converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information such as distance, speed, angle, height and micro-motion characteristics (i.e. micro-Doppler) of the target.

[0303] In some optional embodiments, in a signal transmission and reception link of a digital phase shifter architecture, the digital baseband processing module in the receiving link (which may be located after the IQADC) may further include a phase compensation unit for compensating for phase shifter errors. The phase compensation unit may obtain range-dimensional FFT data by Fourier transforming the range dimension, and perform phase compensation on the range-dimensional FFT data using the phase error coefficient of the transmitting channel (or transmitting link) to obtain range-dimensional FFT compensated data. Subsequently, operations such as velocity-dimensional FFT and wave arrival estimation are performed based on the range-dimensional FFT compensated data to achieve target data detection. Specifically, this may be implemented according to the method described in any embodiment of the present application.

[0304] Another embodiment of the present application relates to an integrated circuit, as shown in FIG23 , comprising: a signal transceiver channel 331 for transmitting radio signals and receiving echo signals generated by the radio signals reflected by target objects; a target detection module 332 for detecting target objects based on the echo signals according to the method described in any of the above embodiments. Optionally, the integrated circuit may be a chip structure, such as a millimeter-wave radar chip. Of course, the integrated circuit may also be implemented using other hardware. In some embodiments, the target detection module 332 may include: a first processing unit for performing a range-dimensional Fourier transform on the echo signals received by the receiving channel to obtain range-dimensional FFT data, and performing phase shift compensation on the range-dimensional FFT data using the phase shift error coefficient of the transmitting channel to obtain range-dimensional FFT-compensated data; a second processing unit for performing a Doppler-dimensional Fourier transform on the range-dimensional FFT-compensated data to obtain Doppler-dimensional FFT data; and a detection unit for determining the velocity information of the target object based on the Doppler-dimensional FFT data.

[0305] The receiver of a linear frequency modulated continuous wave (LFMCW) radar mixes the local oscillator (LO) with the received echo to generate an intermediate frequency (IF) signal. This signal is then sampled by an analog-to-digital converter (ADC) and further digitally processed to determine the target. The LO can couple to the mixer's other input port through multiple paths, mixing with the LO signal itself to produce a low-frequency IF signal. This phenomenon is known as LO leakage. Furthermore, the signal from the transmitting antenna can also couple to the receiver, a phenomenon known as transmit-receive leakage. Both of these phenomena can result in false targets near direct current (DC) in the digital processing output.

[0306] For FMCW, send the signal: Among them, z(t) is the transmitted signal, f c is the operating frequency, β is the scanning bandwidth of the transmission signal, T is the transmission signal bandwidth, and θ is the initial phase of the transmission signal.

[0307] Figure 24 is a schematic diagram of a model signal for LO leakage in an embodiment of the present application. As shown in Figure 24 , LPF represents a low-pass filter, and the circle represents a mixer. Under normal circumstances, a portion of the local oscillator signal x(t) leaks through other circuit components to the other end of the mixer. This leakage can occur along multiple paths. Therefore, for K leakage paths, the following expression is used:

[0308] Among them, A k is the amplitude of the leakage signal on the kth leakage path, τ k is the delay of the leakage signal on the kth leakage path.

[0309] Then, the mixer output at the receiving end can be obtained after passing through the LPF:

[0310] It is equivalent to having K Therefore, for LO leakage at the receiving end, it is necessary to estimate and filter out K low-frequency components in the echo signal at the receiving end.

[0311] In order to filter out several low-frequency interference frequency components in the received signal to suppress low-frequency leakage, thereby avoiding the generation of false targets in the vicinity of DC in the digital processing results, an embodiment of the present application provides an FMCW receive leakage processing method for estimating and filtering K low-frequency frequency components in the signal received by the receiving end, so as to effectively eliminate the impact of LO leakage on the received signal, thereby avoiding the generation of false targets in the vicinity of DC in the digital processing results.

[0312] One embodiment of the present application relates to a method for processing FMCW receive leakage. The method may be performed by an FMCW radar system. As shown in FIG25 , the method for processing FMCW receive leakage includes the following steps:

[0313] Step 2501: In a targetless scenario, perform range-dimensional Fourier transform on the received signal to obtain range-dimensional FFT data, and for each chirp signal in the range-dimensional FFT data, extract the first k data of the chirp signal sorted from low to high in frequency.

[0314] In one exemplary embodiment, in a target-free scenario, a radar system performs a range-dimensional Fourier transform (also known as a "1D FFT") on the signal received by the receive channel. After obtaining range-dimensional FFT data (also known as "1D FFT data"), the system sorts the data within each chirp in the range-dimensional FFT data from low to high frequency. Specifically, the data within each chirp is sorted from small to large range gates, with smaller frequencies corresponding to smaller range gates. When LO leakage signals are present in the received signal, their frequencies are primarily distributed in the low-frequency band, making it easy for false targets near DC to appear in radar detection results. Therefore, it is necessary to evaluate the low-frequency LO leakage signals in target-free scenarios and eliminate them from each chirp in the received signal during target detection.

[0315] In one embodiment, for each chirp in the range-dimensional FFT data generated by the received signal in a target-free scenario, the first k data in the chirp, sorted from low to high frequency, are extracted and used as the LO leakage evaluation target. Here, k is an integer greater than 1.

[0316] Step 2502: For each data of the same rank among the first k data of each group extracted from different chirp signals, obtain the leakage value of the rank; wherein, among the first k data of each group, the data of the same rank have the same corresponding frequency.

[0317] In an exemplary embodiment, for each chirp in the range-dimensional FFT data generated by the received signal in a target-free scenario, a set of the top k data sorted from low to high frequency can be extracted. The order of each data in the top k data is called the rank, which can be expressed as index(i), where i is an integer in [1, k]. When the sampling frequency used to receive each chirp is the same, the data at the same rank in each group have the same corresponding frequency and can represent the same range gate. By performing energy evaluation on each data at the same rank in each chirp, the signal leakage corresponding to the rank can be obtained, and the signal leakage can be quantified by a leakage value. In this way, a leakage value can be obtained for each rank data in each chirp. By sorting the leakage values ​​corresponding to each rank according to the same rank, k leakage values ​​with a rank relationship can be obtained.

[0318] Step 2503: When performing target detection, the leakage values ​​of each sequence are used to perform subtraction processing on the data of the corresponding sequence of each chirp signal in the range-dimensional FFT data generated by the currently received signal to achieve leakage elimination of the received signal.

[0319] In an exemplary embodiment, after obtaining k leakage values ​​with a sequence relationship in a target-free scenario, the radar system can use the same sequence relationship to process the received signal to obtain each chirp in the distance-dimensional FFT data, and use the leakage values ​​of each sequence to perform subtraction processing on the data of the corresponding sequence in each chirp to eliminate leakage of the received signal, that is, to obtain the distance-dimensional FFT data after leakage elimination.

[0320] An embodiment of the present application provides a method for processing FMCW reception leakage. When multiple leakage paths exist at the receiving end of an LFMCW radar, the leakage signal is estimated to obtain the low-frequency frequency of the leakage signal. Based on this low-frequency frequency, several low-frequency interference frequency components in the signal received during target detection are filtered out to suppress leakage, thereby avoiding the generation of false targets in the digital processing results near the digital conversion (DC).

[0321] Another embodiment of the present application relates to a method for processing FMCW receive leakage. This method is based on the embodiment shown in FIG25 and refines each execution step in units of frames. As shown in FIG26 , the method for processing FMCW receive leakage includes the following steps:

[0322] In some embodiments, step 2501 in Figure 25 may include step 26011.

[0323] Step 26011: In a target-free scenario, perform distance-dimensional Fourier transform on the received signal of each frame to obtain distance-dimensional FFT data of each frame, and for each chirp signal in the distance-dimensional FFT data of each frame, extract the first k data of the chirp signal sorted from low to high in frequency.

[0324] As shown in Figure 20b, this is a one-dimensional FFT plane ("1D FFT plane") obtained after performing range-dimensional FFT processing on a frame of signals. The horizontal direction is the Doppler dimension (v), and one frame contains M Doppler gates, each of which corresponds to one chirp. The vertical direction is the range dimension (r), and one frame contains N range gates, each of which corresponds to range-dimensional FFT data at the same frequency. Each chirp is divided into multiple range gates according to frequency from low to high (range gates from small to large).

[0325] In this embodiment, for each chirp in the range-dimensional FFT data corresponding to each frame, the first k data in the chirp sorted from small to large by frequency are extracted, i.e., the data of the first k range gates sorted from small to large by range gate in FIG20b.

[0326] In some embodiments, step 2502 in FIG. 25 may include steps 26021 and 26022 .

[0327] Step 26021: Taking a frame as a unit, for each data of the same sequence in each chirp signal in the extracted range-dimensional FFT data within each frame, obtain the first leakage value of the sequence in the current frame.

[0328] In one embodiment, the signal received in each frame includes M chirps, and each chirp contains N data sampling points. In ascending order of frequency, s1[0] represents the first data sampling point corresponding to the first chirp, and Fs1[0] represents the distance dimension FFT data obtained after the first data sampling point corresponding to the first chirp is processed by the distance dimension FFT transform. By analogy, the distance dimension data corresponding to the N data sampling points in the M chirps is expressed as follows:

[0329] s1[0], s1[1],...s1[N-1], (FFT)=>Fs1[0], Fs1[1], ..Fs1[k],...Fsi[N-1],

[0330] s2[0], s2[1],...s2[N-1], (FFT)=>Fs2[0], Fs2[1], ..Fs2[k],...Fs2[N-1],

[0331] …

[0332] sM[0], sM[1],...sM[N-1], (FFT)=>FsM[0], FsM[1], ..FsM[k],...FsM[N-1].

[0333] For the data at the same order in each chirp in the distance-dimensional FFT data (such as Fs1[0], Fs2[0], ... FsM[0] are all distance-dimensional FFT data at the first order), calculate the leakage value corresponding to the order (called "first leakage value" in the frame).

[0334] In some embodiments, the arithmetic mean of each data at the same sequence may be used as the first leakage value.

[0335] In one embodiment, if the signal energy difference between each chirp in the distance dimension FFT data within the frame is small, the leakage value corresponding to the same sequence in the frame can be calculated based on the average value of each data at that sequence in each chirp, which can reflect the signal changes with small differences between each chirp in the frame.

[0336] For example, q[0], q[1]...q[k-1] represent the first leakage values ​​of the first k ranges of each chirp in the range-dimensional FFT data within the frame, where q[0] represents the first sampling data of each chirp (a total of M chirps) in the signal received within the frame, and the distance-dimensional FFT transform is performed on the first sampling data to obtain the distance-dimensional FFT data Fs1[0], Fs2[0], ...FsM[0]. The average value obtained by averaging these M distance-dimensional FFT data is used as the first leakage value q[0] corresponding to the range 0. Similarly, the first leakage value (q[0]~q[k]) corresponding to each range in the first k ranges is obtained as follows:

[0337] q[0]=(Fs1[0]+Fs2[0]+Fs3[0]+...+FsM[0]) / M,

[0338] q[1]=(Fs1[1]+Fs2[1]+Fs3[0]+...+FsM[1]) / M,

[0339] …

[0340] q[k-1]=(Fs1[k-1]+Fs2[k-1]+Fs3[k-1]+...+FsM[k-1]) / M;

[0341] Step 26022: For each first leakage value of the same sequence in each frame, obtain the second leakage value of the same sequence between frames.

[0342] In an exemplary embodiment, after obtaining the first leakage values ​​of the same order in each frame, the leakage situation of the same order between frames can be further obtained. This embodiment records the leakage value of the leakage situation of the same order between quantized frames as the second leakage value.

[0343] In some embodiments, the median value of the first leakage values ​​may be used as the second leakage value.

[0344] If the signal energy difference between each chirp in the inter-frame distance dimension FFT data is large, the leakage value corresponding to the same sequence between frames can be calculated based on the median value of the first leakage value at that sequence in each frame, which can reflect the large difference in signal changes between chirps between different frames.

[0345] After obtaining the first leakage value q[0] to q[k-1] corresponding to each range in the first k ranges of each chirp signal of each frame's distance-dimensional FFT data, the leakage of this frame can be evaluated, and the evaluation result is Q. For example, Q[0] represents the first leakage value of the first k ranges in the first frame signal, that is, the first leakage values ​​q[0] to q[k-1] corresponding to each range in the first frame are arranged in sequence to obtain Q[0] = {q[0], q[1]...q[k-1]}. Similarly, the first leakage value of each range in the first k ranges of the L-frame distance-dimensional FFT signal is obtained as follows:

[0346] Q[0]={q[0],q[1]...q[k-1]},q[0],q[1]...q[k-1]∈the first frame;

[0347] Q[1]={q[0],q[1]...q[k-1]},q[0],q[1]...q[k-1]∈the second frame;

[0348] …

[0349] Q[L-1]={q[0],q[1]...q[k-1]},q[0],q[1]...q[k-1]∈Lth frame;

[0350] L is an integer greater than 1.

[0351] After obtaining the first leakage of each range in the first k ranges of the range-dimensional FFT signal of each frame, the first leakage values ​​of each range in the first k ranges of each frame are reordered to obtain the sorted (OS) first leakage values ​​q*[0] to q*[k-1], where:

[0352] q*[0]=OS{Q[0].q[0],Q[1].q[0],...Q[L-1].q[0]}(q*[0] represents the first leakage value of all 0 ranges in the first k ranges in L frames, Q[0].q[0] represents the first leakage value q[0] of the 0 range in the first k ranges in the first frame Q[0]), and the first leakage values ​​of the same sequence in L frames are ordered statistically to obtain the following sequences, and then the median value is selected from each sequence as the second leakage value.

[0353] q*[1]=OS{Q[0].q[1], Q[1].q[1],...Q[L-1].q[1]};

[0354] …

[0355] q*[k-1]=OS{Q[0].q[k-1], Q[1].q[k-1],...Q[L-1].q[k-1]};

[0356] Select a median value from q*[0], q*[1], ...q*[k-1] as the second leakage value of the sequence.

[0357] Based on this, the leakage value in step 2603 can be specifically the second leakage value.

[0358] In addition, when performing leakage cancellation processing on the range-dimensional FFT data generated during target detection based on the leakage values ​​corresponding to each rank in a target-free scenario, the leakage values ​​can be directly used for cancellation, or the leakage values ​​can be processed and then used for leakage cancellation processing. In one embodiment, step 2603 may include: when performing target detection, subtracting the leakage values ​​of the same rank from the first k data of each chirp signal in the range-dimensional FFT data generated by the currently received signal to achieve leakage cancellation for the received signal; or, when performing target detection, subtracting the fine-tuned leakage values ​​of the same rank from the first k data of each chirp signal in the range-dimensional FFT data generated by the currently received signal to achieve leakage cancellation for the received signal.

[0359] The fine-tuning of the leakage value may include, but is not limited to, performing a uniform linear processing on the leakage values ​​at different ranks, or performing a weighted processing with different weights on the leakage values ​​at different ranks.

[0360] In this embodiment, regarding the leakage cancellation of the received signal, the specific signal leakage path and the application scenario of the radar system are not limited. The k leakage values ​​obtained here can be used to cancel intrinsic leakage, to cancel transmission channel-receiving channel leakage, or to be used in scenarios that include both types of leakage, as long as it is used to cancel low-frequency leakage signals.

[0361] Furthermore, after leakage cancellation is performed on the received signal to obtain the post-leakage cancellation range-dimensional FFT data, further processing may be performed as follows: Specifically, the post-leakage cancellation range-dimensional FFT data may be subjected to a Doppler-dimensional Fourier transform to obtain a range-Doppler spectrum; and based on the range-Doppler spectrum, information such as the distance and / or velocity of the target object may be determined. In some optional embodiments, operations such as direction of arrival estimation may be performed based on the range-Doppler spectrum to determine information such as the target's angle (e.g., horizontal angle and / or pitch angle).

[0362] In one exemplary embodiment, after obtaining the leakage-eliminated range-dimensional FFT data, it can be subjected to a Doppler-dimensional Fourier transform (also known as a "2D FFT") to obtain Doppler-dimensional FFT data (also known as "2D FFT data" or "velocity-dimensional FFT data"), thereby forming a range-Doppler spectrum. Constant False Alarm Rate (CFAR) detection is performed on the range-Doppler spectrum to remove false targets in the energy dimension. At this point, the target information contains information in the range and / or velocity dimensions.

[0363] In addition, the angle information of the target object can also be determined based on the range-Doppler spectrum.

[0364] In one embodiment, complex data at the peak of the Doppler dimension FFT data generated by the transmit channel may be extracted from the Doppler dimension FFT data in the range-Doppler spectrum, and a received vector may be synthesized to determine the angle information of the target object.

[0365] Another embodiment of the present application relates to a method for processing FMCW receive leakage, comprising the following steps:

[0366] Step a: In a target-free scenario, perform range-dimensional Fourier transform on the received signal to obtain range-dimensional FFT data, and extract the first k data of each chirp signal in the range-dimensional FFT data;

[0367] Step b: For each data of the same sequence among the first k data extracted, obtain the leakage value of the sequence.

[0368] The method described in this embodiment is intended to provide a method for calculating leakage values. This embodiment does not limit the application scenarios of leakage values. For the description of steps a-b, please refer to the aforementioned steps 2501-2502. Different from this, in this embodiment, the order of the first k data is determined according to the order in which the data in each chirp signal in the distance dimension FFT data is acquired. For example, when the scanning frequency changes from low frequency to high frequency, the corresponding frequencies of the first k data in each chirp signal acquired are relatively low; when the scanning frequency changes from high frequency to low frequency, the corresponding frequencies of the first k data in each chirp signal acquired in the order are relatively high. In the method steps in the embodiment shown in Figure 26, the specific implementation processes of the corresponding steps 2501-2502 are applicable to the corresponding steps a and b in this embodiment, and this embodiment will not elaborate on this.

[0369] As shown in FIG. 27 , another embodiment of the present invention relates to a method for processing FMCW receive leakage, comprising the following steps:

[0370] Step 2701: In a target-free scenario, perform range-dimensional Fourier transform on the received signal to obtain range-dimensional FFT data. The description of this step can be found in the aforementioned step 2501, and will not be repeated in this embodiment.

[0371] Step 2702: Based on a frame of range-dimensional FFT data, for a plurality of predetermined range bins, the average value between different chirp signals is used as the first leakage value of each predetermined range bin.

[0372] As shown in FIG20b , a frame of range-dimensional FFT data can be divided into multiple range intervals along the range dimension, each of which has the same length. Each such range interval can be referred to as a range bin, and the data contained in each chirp signal is distributed in each range bin. For these range bins, the range bin for which leakage cancellation is desired can be selected based on actual needs to generate the leakage value corresponding to that range bin. The selected range bins become the predetermined range bins described above, and the method for generating the corresponding leakage value for each predetermined range bin is as follows: among the chirp signals contained in the current predetermined range bin, the average value between the chirp signals is calculated as the leakage value for that predetermined range bin, also referred to as the first leakage value.

[0373] As can be seen from Figure 20b, the distance bins are arranged from small to large along the distance dimension, and the frequency of the data in each chirp signal is also arranged from small to large along the distance dimension. Calculating the first leakage value of each predetermined distance bin is essentially calculating the first leakage value of the frequency segment corresponding to each predetermined distance bin.

[0374] This embodiment aims to obtain corresponding leakage values ​​for the range bin of interest, so as to perform leakage cancellation processing on the chirp signal in the range bin of interest in an actual measurement scenario.

[0375] Furthermore, step 2703 may be included after step 2702 .

[0376] Step 2703: Based on the multi-frame distance dimension FFT data, for a plurality of predetermined distance bins, the median of the first leakage values ​​between different frames is used as the second leakage value of each predetermined distance bin.

[0377] In one exemplary embodiment, after obtaining the first leakage values ​​for the same predetermined distance bin within each frame, the leakage situation for the same predetermined distance bin between frames can be further obtained. In this embodiment, the leakage value that quantifies the leakage situation for the same predetermined distance bin between frames is recorded as the second leakage value. For example, the median value of the first leakage values ​​for the same predetermined distance bin between different frames can be used as the second leakage value.

[0378] As shown in FIG. 28 , another embodiment of the present invention relates to a method for processing FMCW receive leakage, comprising the following steps:

[0379] Step 2801: retrieve the leakage value of each sequence in a plurality of data sorted from low to high frequency in the chirp signal from the pre-stored data obtained based on the distance dimension transformation.

[0380] In one exemplary embodiment, multiple pre-stored leakage values ​​are established based on data at different frequencies in the range-dimensional FFT data obtained by performing a range-dimensional transformation of the signal received by the FMCW radar. Typically, multiple chirp signals are present in the range-dimensional FFT data, and the multiple leakage values ​​in this embodiment correspond to leakage values ​​at corresponding positions in the multiple data points sorted from low to high frequency within each chirp signal. For example, a pre-stored leakage value may correspond to the first-ranked leakage value in the multiple chirp signals sorted from low to high frequency.

[0381] In one embodiment, when storing leakage values, one leakage value may be stored for each sampling frequency point in each chirp signal in the range-dimensional FFT data, or only for each low-frequency sampling frequency point. The specific number of leakage values ​​stored, the granularity of the corresponding frequency points, and whether the frequency points are evenly spaced and continuous can be customized based on actual needs.

[0382] In addition, the process of obtaining the leakage value may include but is not limited to the method steps in the previous embodiment. This embodiment does not limit the process of obtaining the leakage value of each sequence.

[0383] Step 2802: When performing target detection, based on the retrieved leakage values ​​of each sequence, multiple data sorted from low to high in frequency in each chirp signal in the distance-dimensional FFT data generated by the currently received signal are subtracted to eliminate leakage of the received signal.

[0384] In an exemplary instance, based on the specific order of the chirp signal corresponding to the leakage value of each order retrieved, which is sorted from low to high by frequency, and the frequency of the distance-dimensional FFT data corresponding to the order, the data at the same or similar frequency and the same or similar order (the frequency corresponding to the same order is the same or similar) in each chirp signal in the distance-dimensional FFT data generated by the currently received signal can be subtracted to eliminate leakage of the received signal and obtain the distance-dimensional FFT data after leakage cancellation.

[0385] In some embodiments, if the storage space for storing leakage values ​​is sufficiently large, the leakage values ​​for each order of each data point in the chirp signal sorted from low to high frequency based on the data obtained by the distance-dimensional transformation can be stored, or the leakage values ​​for each order of each data point at a low frequency can be stored. This allows the frequency points of the order corresponding to the stored leakage values ​​to be the same as or corresponding to the frequency points of the corresponding order of each chirp signal in the distance-dimensional FFT data to be eliminated. Accordingly, step 2802 can include:

[0386] When performing target detection, the retrieved leakage values ​​of each sequence can be interpolated;

[0387] During target detection, based on the interpolated leakage values ​​of each sequence, the data of the corresponding sequence in each chirp signal in the range-dimensional FFT data generated by the currently received signal is subtracted to eliminate leakage of the received signal.

[0388] In this embodiment, based on the retrieved leakage values ​​of each sequence, subtraction processing is performed on the data of the corresponding sequence in each chirp signal in the range-dimensional FFT data generated by the currently received echo signal, so as to achieve leakage elimination of the received signal.

[0389] In this embodiment, since the order of the retrieved leakage value is aligned with the frequency of the corresponding order in each chirp signal in the range-dimensional FFT data to be processed, the corresponding order can be directly subtracted based on the same order relationship to complete the leakage elimination of the received signal.

[0390] In other embodiments, if the storage space for storing leakage values ​​is limited, it is not possible to store all leakage values ​​for each order in the data obtained based on the distance-dimensional transformation, which is sorted from low to high frequency in the chirp signal. Or, even if all leakage values ​​are stored, they are different from or not close to the frequency points of the corresponding order in each chirp signal in the distance-dimensional FFT data to be processed. In this case, it is necessary to interpolate the extracted leakage values ​​so that the frequency points of the order corresponding to the interpolated leakage values ​​are the same as or correspond to the frequency points of the corresponding order in each chirp signal in the distance-dimensional FFT data to be eliminated. Accordingly, step 2802 may include:

[0391] Interpolate the retrieved leakage values ​​of each sequence;

[0392] During target detection, based on the interpolated leakage values ​​of each sequence, the data of the corresponding sequence in each chirp signal in the range-dimensional FFT data generated by the currently received signal is subtracted to eliminate leakage of the received signal.

[0393] In this embodiment, since the order of the leakage value after interpolation is aligned with the frequency of the corresponding order in each chirp signal in the range-dimensional FFT data to be processed, it is possible to directly perform positional subtraction based on the same order relationship to complete leakage elimination of the received signal.

[0394] In an exemplary embodiment, the interpolation method may be a linear interpolation method.

[0395] Since the signal transmission link adopts an analog phase shifter architecture, it has problems such as low phase modulation precision (resolution) and phase modulation accuracy (accuracy), which makes it unable to meet various requirements of high precision and accuracy.

[0396] FIG29 is a simplified schematic diagram of a signal processing chain of an analog phase shifter architecture, and FIG30 is a simplified schematic diagram of an analog phase shifter in the signal processing chain shown in FIG29 .

[0397] As shown in FIG29 , when the sensor transmits a signal, for a transmission link, a signal generator 311, such as one composed of a phase-locked loop (PLL), generates an LO signal (e.g., a swept frequency signal in the 77 GHz band), which can be, for example, an FMCW signal. An analog phase shifter (Analog PS) 312 performs a phase shift on the received LO signal, which is then radiated into a predetermined spatial region via a transmitting antenna 313 for operations such as target detection and measurement.

[0398] Optionally, in the transmission chain structure shown in FIG29 , the corresponding analog phase shifter architecture may be shown in FIG30 , and the specific phase shifting principle may be shown in the following formula:

[0399] in,

[0400] It can be seen that by performing 0° and 90° phase shifts on the received LO signal (i.e., LOIN), two signals (i.e., LOI and LOQ) are obtained. After amplification by a power amplifier (PA), they are fitted into an output signal LOOUT. That is, the output signal LOOUT is phase-shifted by θ compared to the LOIN input signal. The value of the phase shift θ is determined by the amplitude A of the two signals LOI and the amplitude B of the LOQ.

[0401] In an exemplary embodiment, the analog phase shifter architecture shown in FIG30 can also be implemented by a delay line unit. That is, the phase shift is performed by delay using the narrow-band assumption of the signal. The principle is shown in the following equation:

[0402] Where τ is the delay time of the delay line.

[0403] Because the aforementioned analog phase shifters have low resolution and accuracy, they cannot meet the requirements of current sensors. Although calibration can improve resolution and accuracy, this requires offline calibration of analog phase shifters, significantly increasing the difficulty and complexity of engineering implementation and mass production. Furthermore, analog phase shifters also present significant challenges, such as large area, high loss, stability, and channel coupling.

[0404] Based on this, embodiments of the present application propose a signal transmission link with a digital phase shifter architecture to effectively improve phase modulation precision and accuracy while also avoiding the need for off-line calibration of the phase shifter and other components in the transmission link, thereby reducing the complexity and difficulty of engineering implementation. Furthermore, this approach effectively reduces the area and loss of the phase shifter architecture's transmission link, improving system stability and reducing channel coupling.

[0405] Figure 12B is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present application; Figure 12C is a schematic diagram of a digital phase shifter architecture in a signal transmission link based on the signal transmission link shown in Figure 12B; Figure 12D is a schematic diagram of a transmission link including a specific compensation module in an embodiment of the present application.

[0406] An embodiment of the present application provides a signal transmission link that can be applied to an electromagnetic wave sensor. The transmission link may include an analog signal source and a digital phase shifter. The analog signal source can be configured to provide an initial analog signal (such as an LO signal), and the digital phase shifter can be configured to provide a phase-shifted signal generated in the digital domain. The digital phase shifter can also phase-shift the initial analog signal based on the generated phase-shifted signal to perform a preset phase shift operation on the initial analog signal.

[0407] Figure 1 is a schematic diagram of a transceiver link in an embodiment of the present application, Figure 31 is a schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present application, Figure 32 is a schematic diagram of a transceiver link based on the structure shown in Figure 31 combined with BIST in an embodiment of the present application, and Figure 33 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present application.

[0408] The following describes the transmit and receive links formed by the transmit link structure described in the embodiments of the present application:

[0409] In the signal processing chain shown in Figure 1, the TX-LO signal received by the IQ modulator in the transmit chain and the RX-LO signal received by the real mixer in the receive chain can have the same frequency. For example, as shown in Figure 6, if the baseband output signal is a sine wave at x MHz, then both the TX-LO and RX-LO signals can be sine waves at z GHz.

[0410] In the embodiment shown in FIG1 , the transmitting link adopts a digital phase-shifting architecture, while the receiving link can adopt analog architecture components, i.e., IQ components are not required. Therefore, it is effectively compatible with sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system.

[0411] In this embodiment, the receiving link may include a receiving antenna, which can be connected via a peripheral port of the chip and formed on a carrier such as a PCB. Furthermore, in some optional embodiments, the receiving antenna can also be integrated into the chip package to form an AiP or AoP, i.e., a chip structure with a packaged antenna.

[0412] In some optional embodiments, in order to match the transmitting chain of the digital phase shifter architecture, corresponding adjustments may be made to the receiving chain. For example, the transceiver chain shown in FIG31 may include a transmitting chain architecture and a receiving chain similar to that in FIG1 (in order to avoid redundancy, the same parts will not be described in detail here), that is, the Real Mixer in the receiving chain in FIG1 is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. In this case, the receiving chain may include a low-noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), an IQ digital-to-analog converter (IQ ADC), etc. connected in sequence. That is, the echo signal received by the receiving antenna is sequentially passed through the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ After ADC processing, it is converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information such as distance, speed, angle, height and micro-motion characteristics (i.e. micro-Doppler) of the target.

[0413] In one embodiment, when performing self-calibration based on the transceiver link shown in FIG31 , the signal output port of the transmitting link and the signal input port of the receiving link are directly connected via a transmission line, that is, the transmitting link directly sends the transmitting signal to the receiving link via the transmission line, so as to achieve self-calibration of the receiving and / or transmitting link without passing through the transmitting antenna and the receiving antenna. In this case, there is a certain frequency offset between the TX-LO signal received by the IQ modulator in the transmitting link and the RX-LO signal received by the IQ Demodulator in the receiving link. For example, as shown in FIG12C , if the signal output by the Baseband is a sine wave of x MHz, the TX-LO signal can be a sine wave of z GHz, and the RX-LO signal can be a sine wave of (z GHz-y MHz), that is, the frequency offset is y MHz.

[0414] In some optional embodiments, as shown in FIG31 , a transmitting link (such as a transmitter, TX, shown in the figure) can be calibrated by adding a receiving link (such as a receiver, RX, shown in the figure), and the TX IQ imbalance compensation module in the transmitting link performs compensation based on the calibrated data. Simultaneously, the transmitting link (such as a transmitter, TX, shown in the figure) can also be calibrated by reusing the receiving link (such as a receiver, RX, shown in the figure) actually used for signal transmission and reception, and the TX IQ imbalance compensation module in the transmitting link and / or receiving link performs compensation based on the calibrated data. Similar implementations can also be performed in other embodiments, which will not be described in detail for simplicity.

[0415] In some optional embodiments, based on the structure shown in FIG31 , to achieve precise calibration of the transceiver link, a built-in self-test (BIST) module may be provided at the RX-LO port of the IQ Demodulator of the receive link shown in FIG31 . Specifically, as shown in FIG32 , based on the transceiver link structure shown in FIG31 , an IQ BIST architecture is provided at the RX-LO port of the IQ Demodulator of the receive link to input an LO signal with a preset frequency offset to the RX-LO port of the IQ Demodulator of the receive link. For example, the IQ BIST, which is composed of a phase angle converter and an IQ modulator, utilizes a received TX-LO signal, such as a TX-LO signal, to pass through the phase angle converter and the IQ modulator. A frequency-offset signal is generated based on the frequency offset signal of another input signal, the BIST-LO, of the IQ modulator and is then input to the RX-LO port of the IQ Demodulator. For example, if the TX-LO signal is a z GHz sine wave and the BIST-LO signal is a y MHz sine wave, the frequency-shifted signal input to the RX-LO port of the IQ Demodulator is (z GHz - y MHz). It should be noted that x, y, and z are schematic values ​​in different embodiments and their specific values ​​may be the same or different.

[0416] In some optional embodiments, based on the IQ BIST architecture of the transceiver link structure shown in FIG32 , the transmit link of the digital phase shifter architecture can be calibrated by multiplexing the receive link within the transceiver link. In other embodiments, calibration operations involving the transmit link using the receive link, and calibration operations involving the receive link using the transmit link, can be performed by multiplexing the corresponding receive link or transmit link within the link actually performing signal transmission and reception. Alternatively, calibration operations can be performed on the corresponding transmit link or receive link within the link actually performing signal transmission and reception by adding a corresponding calibration receive link or calibration transmit link.

[0417] Optionally, the IQ BIST may include a phase angle converter and an IQ modulator. The phase angle converter is used to implement separate calibration of the I and Q paths in the digital architecture's transmission link, while the other input signal BIST-LO of the IQ modulator may be a y MHz sine wave, which is used to simulate the characteristics related to the echo signal formed by the transmission signal being reflected by the target. In Figure 32, x, y, and z are all positive numbers, and x≠y≠z, and can generally be between 0 and 1000.

[0418] Optionally, in the transceiver link shown in FIG32 , a TX IQ imbalance compensation module (TX IQ Imbalance Compensation) may be provided in the transmitting link (e.g., between the TX DDFS and the IQ DAC), and / or a TX IQ imbalance compensation module (TX IQ Imbalance Compensation) may be provided in the receiving link (e.g., after the Real ADC), so as to supplement the transmitted and / or received signals based on the calibration parameters (or coefficients) obtained by the self-calibration operation to resolve problems such as IQ imbalance.

[0419] In some optional embodiments, based on the structure shown in FIG32 , as shown in FIG33 , the above-mentioned IQ BIST module can be set between the signal output port of the transmitting link and the signal input port of the receiving link, that is, the transmitting link directly sends the transmitting signal to the receiving link through the IQ BIST module, so as to realize the self-calibration operation of the receiving link and / or the transmitting link without passing through the transmitting antenna and the receiving antenna.

[0420] It should be noted that the transmit chain structures shown in Figures 1, 32, and 33 only illustrate the TX IQ imbalance compensation unit. In actual applications, based on actual needs, an LO compensation unit (TX LO leakage compensation) and an HD3 compensation unit (TX HD3 compensation) may be added to the transmit chain to form a compensation module (TX compensation) that includes the LO compensation unit (TX LO leakage compensation), the TX IQ imbalance compensation unit, and / or the HD3 compensation unit (TX HD3 compensation).

[0421] Figure 34 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application, and Figure 35 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application.

[0422] As shown in FIG34, a transceiver link, combined with the structures and related descriptions shown in FIG12D and FIG33, may include a transmitting link, a receiving link, and a calibration link. The transmitting link may include a TX digital baseband signal source (TX Baseband), a direct digital frequency synthesizer (TX DDFS), a compensation module (Compensation), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), and a power amplifier (PA), etc., which are connected in sequence. At the same time, the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving chain may include an LNA, a real mixer, a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable gain amplifier (VGA), a real digital-to-analog converter (ADC), and an RX baseband for TX RF calibration, all connected in sequence. Specifically, the echo signal received by the receiving antenna is processed by the LNA, real mixer, TIA, HPF, VGA, and ADC before being converted into a real digital baseband signal. Subsequent digital signal processing by the digital signal processing module can obtain target parameter information such as range, speed, angle, altitude, and micro-motion characteristics.

[0423] For the transmit chain, a compensation module (TX compensation) located between the TX DDFS and the IQ DAC can include units such as an LO compensation unit (TX LO leakage compensation), an IQ compensation unit (TX IQ Imbalance compensation), and / or an HD3 compensation unit (TX HD3 compensation). This implements corresponding compensation operations for LO leakage, IQ Imbalance, and HD3 in the transmit chain of the digital phase shifter architecture.

[0424] In some optional embodiments, a calibration module may be provided between the transmit and receive links. This calibration and compensation module may be configured to multiplex the receive link to perform operations such as calibration on the transmit link of the aforementioned digital phase shifter architecture. Simultaneously, the compensation module may utilize the parameters or coefficients obtained through the calibration of this calibration module to compensate for the transmitted signal at the transmit link end. In other embodiments, a corresponding receive compensation module may also be provided simultaneously or separately in the receive link. In this case, the receive compensation module may utilize the parameters or coefficients obtained through the calibration to compensate for the echo signal at the receive link end.

[0425] As shown in FIG34 , the calibration module may include a BIST unit and an auxiliary circuit unit, among others. Specifically, the output port of the transmit link is connected to any node between the real mixer and the real ADC in the receive link via the BIST unit and the auxiliary circuit unit. For example, the IQ Modulator in the transmit link generates a (z GHz±x MHz) RF signal based on an x ​​MHz digital phase-shifted baseband signal and a z GHz LO signal, and then outputs the signal to the BIST unit via the output port. The BIST unit then performs a y MHz frequency offset operation on the received RF signal to obtain an analog echo signal of (z GHz±x MHz±y MHz). The signal is then down-converted using the IQ De-Modulator in the auxiliary unit to obtain a preset intermediate frequency (IF) signal (z GHz±x MHz±y MHz-z GHz=±x MHz±y MHz). This IF signal is then input to a preset node in the receive link to implement calibration operations in the transmit link.

[0426] Optionally, the auxiliary circuit unit can be a quadrature demodulator circuit, and the output of the auxiliary circuit unit can be connected to any of the nodes between the TIA and HPF, between the HPF and VGA, or between the VGA and the real ADC in the receive chain. Furthermore, to maximize the reuse of the receive chain structure, after the output port of a transmit chain passes through the BIST unit and the auxiliary circuit unit, the I and Q branches can be connected to different transmit chains, respectively. This is shown in Figure 10 , calibrating one transmit chain by multiplexing two receive chains. After calibration of the transmit chain, compensation units such as the TX LO leakage compensation unit, the TX IQ imbalance compensation unit, and / or the TX HD3 compensation unit in the aforementioned compensation module (TX compensation) can be used to compensate for LO leakage, IQ imbalance, and HD3 issues in the transmit chain of the digital phase shifter architecture based on the parameters obtained from the calibration.

[0427] In some optional embodiments, the BIST unit may include a phase angle converter and an IQ modulator (IQ Modulator) connected in sequence, and the auxiliary circuit unit may include an LNA, an IQ De-Modulator, and a TIA connected in sequence, wherein the phase angle converter receives the RF signal output from the transmit link, one input of the IQ Modulator is connected to the output of the phase angle converter, and the other input receives a y MHz BIST-LO signal to generate a preset echo signal. The LNA amplifies the received echo signal and sends it to one input of the IQ De-Modulator. The other input of the IQ De-Modulator is used to receive a z GHz RX-LO signal. The two output branches of the IQ De-Modulator (i.e., the I branch and the Q branch) are respectively connected to corresponding nodes in their respective receive links after passing through the TIA, so as to output the generated preset intermediate frequency signal to the two receive links, thereby achieving a more efficient multiplexing of the receive link design while implementing the calibration operation.

[0428] It should be noted that for the calibration operation in the embodiment of the present application, if the transmit link transmits a swept frequency signal, during the actual calibration operation, the TX LO signal can be used as a single-tone signal for point-by-point calibration. At the same time, the TX LO signal can also be used as a swept frequency signal for large-bandwidth calibration operations, and even the swept frequency bandwidth calibration can be used to implement the calibration operation for the swept frequency signal of the entire frequency band in one go.

[0429] Based on the structure shown in Figure 34 , in order to further suppress the preset levels of HD3, LO Leakage, IQ Imbalance, etc. in the transmit link of the digital phase shifter architecture, this can be achieved by cascading at least two BIST units. As shown in Figure 35 , by using two BIST units in series, the noise caused by the above defects can be suppressed to a level of -50dB, thereby effectively reducing the difficulty of developing and designing related link analog devices.

[0430] In some optional embodiments, based on the transmission link of the digital phase shifter architecture described in the embodiments of the present application, when calibrating and compensating for IQ Imbalance, the IQ Imbalance compensation coefficient can be obtained in the time domain (Time-Domain) based on spectrum analysis, and the IQ Imbalance compensation coefficient can also be obtained in the frequency domain (Frequency-Domain) based on the spectrum peak ratio.

[0431] In some optional embodiments, in order to further improve the accuracy of the IQ Imbalance compensation coefficient, the ideal compensation coefficient can be approximated by iterative calibration and compensation, or the ideal compensation coefficient can be obtained by multi-observation calibration and compensation.

[0432] In one embodiment, for the iterative calibration and compensation method, whether to stop the iterative operation can be determined based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iterative conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations. For the multi-observation calibration and compensation method, after multiple (for example, three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (Fast Fourier Transform) and the corresponding amplitude and phase information can be obtained. The measurements can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed; subsequently, the corresponding compensation coefficient can be reversely solved based on the data obtained by inverting the observation matrix.

[0433] In some optional embodiments, based on a similar concept to the above-mentioned IQ Imbalance compensation coefficient, the LO leakage and / or HD3 compensation coefficients may be obtained by adopting an iterative calibration and compensation method or a multiple observation calibration and compensation method.

[0434] FIG36 is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module in an embodiment of the present application, and FIG37 is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module in an embodiment of the present application.

[0435] In some optional embodiments, based on the transmission chain of the digital phase shifter architecture described in the embodiments of the present application, when calibrating and compensating for HD3, since the main source of HD3 in the active mixer is the third harmonic in the nonlinearity of the V / I Converter, it can be achieved through a compensation architecture based on a cubic module as shown in Figure 36, or a compensation structure based on a tripled frequency waveform generator as shown in Figure 37.

[0436] Figure 38 is a schematic diagram of calibration and compensation for a transmit link based on a digital phase shifter architecture, according to an embodiment of the present application. As shown in Figure 38 , based on the technical details of the calibration and compensation operations for IQ Imbalance, LO leakage, and HD3 in the embodiment of the present application, IQ Imbalance compensation is achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation is unaffected by the IQ Imbalance calibration method. LO Leakage compensation is achieved by adjusting the DC component (i.e., DC offset) of both the IQ and Q channels, and similarly, this LO Leakage calibration method has no impact on the compensation scheme. Regarding HD3, since the third-order harmonic distortion of the quadrature mixer V / I converter is the primary source of HD3, and harmonic distortion is affected by DC offset, when both LO Leakage and HD3 calibration are required in the transmit link, HD3 calibration should be performed after LO Leakage calibration to ensure accurate HD3 calibration.

[0437] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency multiplication waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process.

[0438] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO leakage can be calibrated and compensated first, and then the root cause of the HD3 problem, namely the HD3 compensation coefficient, can be calibrated under a stable DC bias. Subsequently, IQ imbalance is calibrated and further compensated for, and then the third harmonic distortion is compensated for both the IQ and Q channels based on the results of IQ imbalance pre-compensation.

[0439] In an optional embodiment, for a digital pre-compensation architecture based on a frequency-doubling waveform generator module, after calibrating and compensating for LO leakage, the HD3 compensation coefficient can be calibrated to obtain the HD3 compensation coefficient, and the IQ imbalance can be calibrated and compensated under a stable DC bias. Subsequently, based on the compensation results, the actual waveforms of the IQ signals and the HD3 compensation coefficient are calculated, and the waveform information of the 3x and 5x frequencies that require pre-compensation is calculated in reverse.

[0440] In another optional embodiment, for a digital precompensation architecture based on a frequency-multiplying waveform generator module, LO leakage can be calibrated and compensated first. Then, multiple (e.g., three) observations are performed to simultaneously calibrate the precompensation coefficients for HD3 and IQ Imbalance. Then, further observations are performed (e.g., two) to calibrate the precompensation coefficients at the HD3 mirror position. Finally, the 3rd and 5th frequency coefficients that require precompensation are calculated from the precompensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in this embodiment of the present application are used to represent operations such as testing and comparative analysis of different test results.

[0441] The architecture of the transmitting link and / or receiving link of the digital phase shifter architecture described in the embodiments of the present application, in addition to being used to transmit and / or receive signals in electromagnetic wave sensors, can also address the problems of unequal lengths of antenna feed lines between different transmitting / receiving links, unequal lengths of TXLO feed lines (i.e., between LO and the mixers in each transmitting channel), and use the architecture of the transmitting link and / or receiving link of the digital phase shifter architecture described in the embodiments of the present application as an auxiliary link to calibrate the receiving link and / or transmitting link of the analog phase shifter architecture.

[0442] Figure 39 is a schematic diagram of a transmission link with at least two transmission channels in an embodiment of the present application, Figure 40 is a structural schematic diagram of a digital LO signal generator in an embodiment of the present application, and Figure 41 is a structural schematic diagram of a feeder unequal length compensation module in an embodiment of the present application.

[0443] As shown in Figure 39, for a transmit link using at least two transmit channels, since each transmit channel (or transmit channel) uses the same local oscillator (LO) source, the feeder length (referred to as TXLO) between the LO signal and the mixer in each transmit channel varies, leading to relative latency between different transmit channels. Furthermore, since using unequal transmit antenna feeders in an antenna array can achieve a better link budget, simpler antenna design and routing, lower inter-antenna coupling, better target angle resolution performance, smaller modules, and lower costs, antenna designs using unequal transmit antenna feeders are becoming increasingly popular, especially for applications such as close-range measurements, relatively enclosed spaces (such as indoor or cabin lighting), and various scenarios using packaged antenna chips (AiP), such as those with wide frequency sweeps. However, the varying transmit antenna feeder lengths between different transmit channels can cause relative latency for RF signals passing through different transmit antennas.

[0444] In response to the above-mentioned problem of relative signal delay in the transceiver link caused by different feeder lengths between different transmitting channels or between receiving channels, the embodiment of the present application provides a compensation solution for the relative delay caused by unequal feeder lengths based on the above-mentioned digital phase shift architecture, so as to effectively improve the quality of the transmitted and received signals.

[0445] In one exemplary embodiment, as shown in FIG39 , a digital phase shifter-based transmit antenna array includes four transmit channels. Each transmit channel shares a waveform control module and a local oscillator (LO). Each transmit channel includes a transmit direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (DAC), an analog low-pass filter (Analog LPF), an IQ mixer (Mixer), a power amplifier (PA), and a transmit antenna, all connected in sequence. Specifically, the waveform control module is connected to the TX DDFS in each transmit channel, while the LO is connected to an input of the mixer in each transmit channel to up-mix the phase-shifted signal output by the LPF in each transmit channel to form a radio frequency signal for transmission via each transmit antenna. To address the issue of unequal feeder lengths between the LO and the mixers in each transmit channel, and / or between the PAs in each transmit channel and their transmit antennas, pre-set compensation can be implemented in the transmit direct digital frequency synthesizer (TX DDFS) to reduce the transmit signal delay caused by these unequal feeder lengths.

[0446] In one embodiment, to address the issue of unequal feeder lengths between the local oscillator (LO) and the mixers in each transmit channel, the shortest feeder from the local oscillator (LO) to the mixer in one transmit channel can be used as a benchmark. The length differences between the mixers in the remaining transmit channels and the local oscillator (LO) compared to this benchmark are then calculated. Because the propagation speed c of the transmitted signal is relatively fixed, the corresponding delay values ​​can be obtained by dividing the length differences between the transmit channels by the speed c. Subsequently, when transmitting the signal, corresponding compensation can be performed directly in the TX DDFS to determine the delay of each transmit channel compared to the benchmark. At the same time, a similar approach can be used to address the issue of unequal feeder lengths between the PAs of each transmit channel and their transmit antennas, thereby obtaining and compensating for the delay caused by feeder differences between the transmit channels.

[0447] When specifically compensating for the transmission link of the FMCW wave, a digital LO generator can be used to compensate for the delay μ caused by the feeder length compared to the reference, the sweep period T, the sweep bandwidth β, and the sweep start frequency f. c Parameters such as Δθ and the subsequent DAC clock signal are adapted to compensate for the aforementioned delay. As shown in Figure 40, a digital LO generator is used in the DDFS for compensation, i.e., one input is Δθ = μ*β / T, and the other input is θ0 = fcμ-μ2*β / 2T; where μ is the delay difference between the feeder corresponding to each transmit channel and the reference feeder; at the same time, as shown in Figures 40-41, the digital LO generator serves as the output LO-I (i.e., cos(ω)) of the signal compensator. n +ψ)) serves as the input of the I path in the DAC, and the output LO-Q (i.e. sin(ωn+ψ)) of the digital LO generator serves as the input of the Q path in the DAC.

[0448] It should be noted that Figures 40-41 illustrate the structural diagrams of a transmission channel, while Figure 15 shows a transmission antenna array with four transmission channels. In actual applications, the transmission antenna array only needs to include at least two (such as two, three or five, etc.) transmission channels, and compensation operations can also be performed during the reception of signals and / or signal processing. At the same time, for a structure with one transmission channel and / or one reception channel, a similar idea can also be used to perform compensation directly in the TX DFFS or in the reception of signals and / or signal processing.

[0449] For the relevant technical solutions for using the idea of ​​the architecture of the transmitting link and / or receiving link of the digital phase shifter architecture recorded in the embodiments of the present application as an auxiliary link to perform calibration and other operations on the receiving link and / or transmitting link of the analog phase shifter architecture and the receiving link and / or transmitting link of the digital phase shifter architecture, please refer to Figures 42-45.

[0450] Figure 42 is a schematic diagram of using an auxiliary circuit to calibrate and compensate the transceiver link in an embodiment of the present application, and Figure 43 is a schematic diagram of using an auxiliary transmitting circuit to calibrate and compensate the receiving link in an embodiment of the present application.

[0451] In an exemplary embodiment, the signal transceiver link may generally include an antenna, RF analog devices, baseband analog devices, and a baseband digital processor, etc. The RF analog devices may include PLL, PA, PS, LNA, LO, and power detector (PD), etc., while the baseband analog devices may include LPF, HPF, ADC, etc.

[0452] Analog components and circuits can exhibit a variety of non-ideal conditions, which can also vary with temperature. For example, the frequency responses of the RF analog components and circuits in the transmit chain (TX), the baseband analog components and circuits in the receive chain (RX), and the RF analog components and circuits in the receive chain (RX) may all be unbalanced. PD accuracy issues may exist at both the PA output and LNA input ports, and local oscillator (LO) leakage may also exist in the receive chain (RX). Furthermore, when analog components operate at different frequencies, their amplitude-frequency responses vary, effectively adding an unexpected window function to the received signal. This can adversely affect subsequent operations such as target range and velocity estimation. This can even lead to false target detection or missed targets in multi-target scenarios. Furthermore, in scenarios with at least two links, the frequency responses of different receive chains can vary, introducing additional noise, such as phase and amplitude variations. This introduced phase, in particular, can directly affect the subsequent estimation of the target's echo direction, reducing the accuracy of target angle estimation.

[0453] In some embodiments, the errors of analog devices and circuits can be measured by external equipment, and then certain compensation operations can be performed on them, such as bench calibration or ATE calibration, to calibrate the performance of analog devices and circuits of the receiving link.

[0454] The embodiments of the present application also provide a solution for calibrating the transmit link and / or receive link based on an auxiliary link method, so as to achieve real-time calibration of analog devices and circuits while achieving precise calibration without the need for external equipment, thereby effectively reducing the impact of changes in RF device parameters due to environmental changes.

[0455] As shown in Figure 42, the transmission and reception link of the electromagnetic wave signal can be calibrated in real time by setting up an auxiliary calibration circuit (Auxiliary). The auxiliary calibration circuit (i.e., auxiliary calibration link) can be integrated into the electromagnetic wave sensor, thereby achieving real-time and accurate calibration without the involvement of external equipment, and effectively reducing the impact of changes in RF device parameters due to environmental changes.

[0456] In one embodiment, for the transmit path, an auxiliary receive path (ARX) can be positioned adjacent to the transmit path in the sensor to enable real-time calibration of the transmit path. Similarly, for the receive path, an auxiliary transmit path (ATX) can be positioned adjacent to the receive path (i.e., the signal receiving link) in the sensor to enable real-time calibration of the receive path. Furthermore, an auxiliary path can be positioned between two paths to be calibrated, allowing one auxiliary path to be reused for calibration of at least two different paths. As shown in FIG42 , a shared ARX is positioned between the two transmit paths, while a shared ATX is positioned between the two receive paths. This further avoids issues such as excessive wiring or excessively long lines. In an alternative embodiment, for the four-transmit, four-receive antenna array shown in FIG42 , two ARXs and two ATXs can be positioned, with each ARX positioned between the two transmit paths and each ATX positioned between the two receive paths. This effectively reduces wiring complexity and improves the real-time nature of calibration.

[0457] As shown in Figure 42, since the receive path can be divided into the RF portion (i.e., RF Rx) and the baseband portion (i.e., Rx BB), auxiliary calibration paths can be set up for each of the RF Rx and Rx BB to further improve calibration accuracy. For example, an RF auxiliary transmitter unit (RF ATX) can be set up for the RF Rx, and an IF ATX can be set up for the Rx BB. Furthermore, for the ARX path, an RF tone generator can be set up to calibrate the ARX, and then the calibrated ARX can be used to calibrate the transmit path (TX).

[0458] In some optional embodiments, as shown in FIG42 , for the RF Rx and Rx BB in the receiving path (Receiver), the Rx BB may be calibrated using the IF ATX first, and then the RF ATX may be calibrated using the calibrated Rx BB, and finally the Rx BB may be calibrated using the calibrated RF ATX, thereby achieving calibration of the entire receiving path.

[0459] Based on the structure shown in Figure 42 and combined with the content shown in Figure 43, the receiver calibration scheme is described. That is, the receiving channel may include a receiving antenna (or receiving port), LNA, Mixer, TIA, LPF, HPF, Real ADC and other devices connected in sequence, as well as a PD connected to the LNA input port. Based on the Mixer, the receiving channel can be divided into RF Rx and Rx BB. That is, along the direction of signal transmission, the Mixer and the part before it are defined as RF Rx, and the rest can be defined as Rx BB. As shown in Figure 43, the RF Rx can include a receiving antenna (or receiving port), LNA, Real Mixer and PD.

[0460] The IF ATX corresponding to the Rx BB of the aforementioned receive path may include a frequency divider (e.g., Freq Divider 1 / N) and a Real DAC (e.g., a 1-bit Real DAC) connected in sequence, with the output of the Real DAC connected to the output of the Real Mixer. As shown in Figure 43 , the IF ATX may include a frequency divider (e.g., Freq Divider 1 / N) and a 1-bit digital-to-analog converter (e.g., a 1-bit Real DAC) connected in sequence. This allows the IF ATX to stably generate a single-tone signal (single tone) under various conditions, and this single-tone signal can be freely configured to a variety of frequencies.

[0461] In some optional embodiments, to improve calibration accuracy, the IF ATX may be calibrated before calibrating the Rx BB. For example, this may involve calibrating the intermediate frequency (IF) frequency response and DC calibration within the IF ATX. Furthermore, after the IF ATX is calibrated, the calibrated IF ATX can be used to calibrate the Rx BB of the receive path. For example, the calibrated IF ATX can be used to transmit single-tone signals of different frequencies to perform calibration and compensation operations on the Rx BB. Alternatively, the frequency (e.g., tens of MHz) of the single-tone signal transmitted by the IF ATX can be determined based on the sampling frequency of the real ADC in the receive path.

[0462] The RF ATX configured for the RF Rx of the receive channel may include a TX DDFS, an IQ compensation module (TX IQ Comp), an adder, an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an amplifier (e.g., a PA), a multiplier, a local oscillator (LO, not shown), and a squarer (x^2), connected in sequence. The output of the multiplier is connected to the output of the mixer in the receive path via the squarer to form a first calibration branch. The output of the multiplier is also connected to the link between the PD and the transmit antenna in the receive path to form a second calibration branch. The IQ compensation module can be configured to compensate for IQ imbalance in the RF ATX, the adder can be configured to compensate for local oscillator leakage (LO leakage) in the RF ATX, and the squarer can be configured to compensate for residual sideband effects caused by IQ imbalance in the RF ATX.

[0463] In some optional embodiments, to improve calibration accuracy, the RF ATX may be calibrated before the RF ATX calibrates the Rx ATX. For example, calibration operations may be performed to address issues such as local oscillator leakage, RF frequency response, and IQ imbalance within the RF ATX. Furthermore, after the RF ATX is calibrated, the calibrated RF ATX may be used to calibrate the RF Rx in the receive path. For example, single-tone signals of different frequencies may be transmitted through the calibrated RF ATX. The baseband signal processing module in the receive chain then calibrates and compensates for the power detector (PD) at the input of the LNA in the RF portion of the receive path, the total gain from the LNA to the ADC in the receive path, and auxiliary calibration of the frequency response.

[0464] In some optional embodiments, when calibrating a receiver path using an IF ATX and an RF ATX, the IF ATX may be calibrated first, and the calibrated IF ATX may be used to calibrate the Rx BB in the receiver path. Then, the calibrated Rx BB may be used to calibrate the RF ATX via a first calibration branch of the RF ATX (i.e., through a squarer). Finally, the second calibration branch of the calibrated RF ATX may be used to calibrate the RF Rx in the receiver path, thereby achieving calibration and compensation operations for the entire receiver path.

[0465] Specifically, as shown in Figure 43, the Rx BB and Real DAC can be calibrated using a 1-bit Real DAC. The LO leakage and IQ imbalance in the RF ATX can then be calibrated using the calibrated RxBB. Finally, the calibrated RF ATX transmit signal can be input to the LNA in the receive path to correct the RF LO leakage and RF frequency response of the RF Rx.

[0466] FIG44 is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present application. As shown in FIG44 , the transmission path (Transmitter) may include a phase shift module PS, an amplifier PA, a power detector PD, etc. For example, the transmission path may adopt the transmission link of the digital phase shifter architecture (Digital Phase Shifter) described in any embodiment of the present application. For details, please refer to the relevant figures and text descriptions, which will not be described in detail here. Because the transmission path adopts a digital phase shifter architecture, while achieving more accurate phase shifting operations, the transmission channel can simultaneously support multiple modes such as DDM and FDM (Frequency Division Multiplexing) of multiple antennas. It can also eliminate the calibration operation of the RF phase shifter (Phase Shifter), reduce the isolation and coupling degree in the phase shift system, and reduce link loss and production costs. In addition, to address possible TX IQ mismatch and LO leakage issues, the transmit path of this digital phase shifter architecture can also support RF frequency response compensation, IQ imbalance, and LO leakage calibration in the digital domain.

[0467] To address issues such as TX IQ mismatch, LO leakage, and frequency response in the transmit path, an auxiliary receiver (ARX) can be configured to perform relevant calibration and compensation operations. As shown in Figure 44, the ARX can include a mixer, TIA, LPF, HPF, IQ ADC, adder, and RF calibration module (RF Calib) connected in sequence. One input of the mixer receives the ARX IQ LO signal, while the other input is connected along the signal transmission direction (i.e., the direction of the arrow in the figure) to a node before the transmit path PD or to any node after the phase shifter (module). For example, it can be connected to the output of the PA (synchronously calibrating the PA) or the input of the PA, thereby calibrating the transmit path through the ARX. The LO signal frequency in the transmit path and the ARX IQ LO signal frequency have a set offset frequency, so that there is a frequency offset between the two signals, simulating a real transmit / receive signal loop.

[0468] In an optional embodiment, to further improve calibration accuracy, a corresponding calibration circuit (i.e., a calibration receiving unit) may be provided for the ARX (i.e., the auxiliary receiving unit). For example, the RF tone signal generation circuit (RF tone generator) shown in FIG44 may include a TX DDFS, an adder, a real DAC, an LPF, an amplifier, a multiplier, and a bandpass filter (BPF) connected in sequence. The adder can be configured to calibrate and compensate for TX LO leakage, the multiplier can be configured to compensate for RF tone generator LO leakage, and the BPF can be configured to filter out DC signals generated by LO leakage from the RF tone generator. In other words, the RF tone generator can be configured to generate multiple stable tone signals of different frequencies to implement calibration operations for the ARX.

[0469] In some optional embodiments, as shown in FIG44 , the ARX may be calibrated using an RF Tone Generator first, and then the calibrated ARX may be used to calibrate a transmitter channel including a PA, for example, calibrating the PD at the PA output, the phase shifter in the transmit channel, the total gain from the DAC to the PA output, and the frequency response and other devices and circuits.

[0470] Specifically, as shown in FIG44 , the RF Tone Generator can be used to generate multiple stable single-tone signals of different frequencies to assist in calibrating the ARX, and then the calibrated ARX can be used to assist in calibrating the IQ imbalance, local oscillator leakage, inconsistent frequency response, and other issues of the transmit path TX.

[0471] Figure 45 is a structural schematic diagram of an auxiliary circuit in an embodiment of the present application, Figure 46 is a structural schematic diagram of another auxiliary circuit in an embodiment of the present application, Figure 48 is a structural schematic diagram of an IQ Mixer in an embodiment of the present application, and Figure 49 is a schematic diagram corresponding to the structure shown in Figure 48.

[0472] In some optional embodiments, based on the structures shown in Figures 42-43 and the related descriptions, the ATX can also be implemented using a squarer combined with a multi-bit DAC. As shown in Figure 45, the receive path includes an LNA, a mixer, a TIA, a HPF, an ADC, a BB processor, and a local oscillator (LO) connected in sequence. Specifically, the mixer uses a first LO signal provided by the LO to downconvert the echo signal provided by the LNA to produce an intermediate frequency (IF) signal (i.e., an analog baseband signal). The ATX may include a first DAC (i.e., DAC1), a mixer, and a squarer (x^2) connected in sequence. The first DAC provides an analog signal to the mixer's receiving end in the ATX. The mixer's other receiving end can be connected to the local oscillator in the receive path, where it mixes the analog echo signal output by the first DAC with a second LO signal provided by the local oscillator, thereby outputting an analog IF signal to the input of the squarer. The squarer then connects the processed analog IF signal to the input of the TIA to calibrate circuits and components in the receive path, such as the TIA, HPF, and ADC. In some other optional embodiments, the ATX can also output the analog echo signal directly to the receiving end of the squarer through the DAC, and the squarer sends the processed analog echo signal to the input end of the LNA to calibrate the LNA, mixer, TIA, HPF, ADC and other circuits and components in the receiving path.

[0473] In some optional embodiments, as shown in FIG46 , based on the structure shown in FIG45 , a second DAC may be provided for a receive path including a VGA, an SDM unit, and an ADC with a decimation filter. The output of the second DAC may be connected to any node in the receive path between the mixer and the TIA, between the TIA and the HPF, between the HPF and the VGA, or between the VGA and the ADC to calibrate the corresponding circuits and components. Optionally, the first DAC may be a multi-bit (e.g., 10-bit) DAC, and the second DAC may be a 1-bit DAC.

[0474] The above-mentioned ATX composed of DAC and squarer can be reasonably modified by those skilled in the art based on the above-mentioned contents. The specific implementation method is not limited in this application, as long as it can achieve the same or similar functions as the ATX in the embodiment of this application.

[0475] In some optional embodiments, when setting up an ATX near a receiving path, the ATX can be set up by utilizing the gaps between the receiving paths, and at the same time, two or more receiving paths can share one ATX; similarly, when setting up an ARX near a transmitting path, the ARX can be set up by utilizing the gaps between the transmitting paths, and at the same time, two or more transmitting paths can share one ARX. Optionally, the above-mentioned ATX and / or ARX can be operated intermittently. For example, the ATX can transmit a calibration tone signal in the gaps between the working of the receiving path (such as between frames or chirps) to effectively calibrate and compensate the receiver in real time. Similarly, the ARX can also use a tone generator to generate a calibration tone signal in the gaps between the working of the receiver (such as between frames or chirps) to effectively calibrate the ARX first. The transmitter is then effectively calibrated using the calibrated ARX to ensure that calibration and compensation operations can be performed in a preset manner.

[0476] With regard to the scheme for implementing transmit path and / or receive path calibration based on an auxiliary link as shown in Figures 42-46, based on the use of an auxiliary link, auxiliary circuits are added to integrated circuits such as chips (chips or dies) to assist in calibrating the main path circuits, thereby effectively improving the performance of analog circuits and modules. At the same time, at least some RF circuits and devices can be calibrated in real time online (on the fly), thereby effectively improving the calibration performance, and thus effectively improving the RF performance while reducing the difficulty of RF implementation.

[0477] Because mixers are very important and critical components for frequency conversion in transceiver links, they are widely used in radio devices such as communications and radar. For example, single-sideband mixers have good suppression effects on image signals, as well as IQ mixers in transceiver links in embodiments of the present application. An IQ mixer (such as a single-sideband mixer) includes two branches, an I branch and a Q branch, with a phase difference of 90° to transmit signals.

[0478] For application scenarios with relatively compact layout areas, such as high-frequency sensor applications in the millimeter wave band, the physical distance between the I branch and the Q branch, and between the input branch and the output branch before mixing are short. As a result, electromagnetic wave signals may leak between branches, between input and output ports, and between the IQ matching network and the mixer output through methods such as magnetic coupling, substrate coupling, and electrical coupling, thereby causing serious deterioration in the mixer's image rejection ratio and local oscillator leakage.

[0479] In an optional embodiment, based on the mixer output network ensuring power synthesis of the IQ branches while also meeting the impedance matching requirements, the present application provides a new mixer structure, that is, by improving the mixer output passive network, the isolation between the IQ branches is effectively improved while effectively reducing the leakage of the local oscillator signal.

[0480] FIG47 is a schematic diagram of the circuit module of the IQ Mixer in an embodiment of the present application. As shown in FIG47 , an IQ mixer may include an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit, wherein the I-branch mixing unit may be configured to output an I-branch signal, the Q-branch mixing unit may be configured to output a Q-branch signal, and the transformer unit may be configured to magnetically couple the I-branch signal and the Q-branch signal to synthesize an IQ mixing output signal, and transmit the IQ mixing output signal to a subsequent circuit (Next Block). The transformer unit may be arranged between the I-branch mixing unit and the Q-branch mixing unit, thereby increasing the physical distance between the I-branch mixing unit and the Q-branch mixing unit relative to a short-circuited IQ mixer, thereby effectively reducing the coupling between the I-branch mixing unit and the Q-branch mixing unit, thereby achieving the purpose of improving the IQ mixer rejection ratio, while also facilitating the layout design of the circuit board.

[0481] A short-circuit IQ mixer is constructed by short-circuiting the outputs of the two IQ branches of the mixer, and then matching them to the subsequent circuit through a matching network, that is, performing signal synthesis through electrical coupling. As shown in Figure 47, in an IQ mixer of an optional embodiment of the present application, a branch inductor can be connected in series between the output ends of the I branch mixing unit and the output ends of the Q branch mixing unit, and then a magnetic coupling inductor is set between the two branch inductors to form a three-turn transformer structure. That is, the output of the I branch mixing unit and the output of the Q branch mixing unit are magnetically coupled and synthesized by magnetic coupling to obtain the above-mentioned IQ mixing output signal; wherein the common mode of the I branch mixing unit and the O branch mixing unit are not directly connected, so they can be adjusted separately, making the application scenario of the device more flexible; then, the two ends of the magnetic coupling inductor are used as output ends to directly transmit to the subsequent circuit (next block). That is, the three-turn transformer structure has four input ends and two output ends.

[0482] Based on the three-turn transformer structure shown in Figure 47, if the LO signal and the signal to be mixed (such as an echo signal) enter the P and N terminals of the IQ branches of the three-turn transformer, the current and magnetic field directions formed on the three-turn transformer are shown in Figure 48. At this time, the Isb magnetic field is superimposed and the USB magnetic field is canceled, thereby outputting a high-quality single-sideband signal. This three-turn transformer structure can achieve down-mixing or up-mixing of the signal to be mixed.

[0483] In some optional embodiments, even in a scenario where there is a phase difference between the LO signal and the signal to be mixed, since the signals are canceled in the same manner on the three-turn transformer, when there is no imbalance between the LO signal and the signal to be mixed, the output of the three-turn transformer is still a single-sideband signal.

[0484] The magnetically coupled IQ mixer provided in the embodiment of the present application performs power synthesis through magnetic coupling, compared to the short-circuited IQ mixer that uses electrical coupling to perform power synthesis. At the same time, for the I and Q branches, the magnetically coupled IQ mixer can be completely consistent and can be axially symmetrical, without introducing additional phase errors due to the inconsistent wiring lengths of the I and Q branches, thereby making the related circuits (such as millimeter wave and other high-frequency circuits) more robust in subsequent manufacturing processes and more resistant to process instabilities. For example, the magnetically coupled IQ mixer in the embodiment of the present application can be set to the transformer tap on the midline between the P and N ports as shown in Figure 49, so that the structure of the mixer can be axially symmetrical based on the midline, thereby making the common-mode path lengths of the P port and the N port equal, that is, the PN symmetry in the magnetically coupled IQ mixer is better than the PN symmetry in the short-circuited IQ mixer, thereby making the leakage of the local oscillator signal smaller.

[0485] In some optional embodiments, the magnetically coupled IQ mixers provided in the embodiments of the present application can be applied to various circuits such as transmit channels and on-chip self-test (OST), serving as up-conversion mixers in various electromagnetic wave circuits. For example, the IQ mixers in various transmit channels (or transmit links) and BIST in the embodiments of the present application can employ the magnetically coupled IQ mixers in the embodiments of the present application.

[0486] Figure 50 is a schematic diagram of the physical structure of another IQ mixer in an embodiment of the present application. As shown in Figure 50, the embodiment of the present application also provides another structure of a magnetically coupled IQ mixer, that is, the tap feed lines of the I and Q branches partially overlap. For example, for the I branch, the tap of the I branch can bypass the side of the Q branch (such as through a via) and extend to the side of the I branch before being connected to the common-mode bias voltage. Similarly, the tap of the Q branch can bypass the side of the I branch (such as through a via) and extend to the side of the Q branch before being connected to the common-mode bias voltage. The above-mentioned routing method can make the path of the common-mode path more certain and less susceptible to interference from other circuits. At the same time, the three-turn transformer in the IQ mixer in the embodiment of the present application can be square, octagonal, regular octagonal, etc., that is, as long as it can be axially symmetrically distributed along the center between the NPs in the top view or bottom view.

[0487] An embodiment of the present application also provides an integrated circuit, which may include: a signal transceiver channel, used to transmit radio signals, and receive echo signals formed by the radio signals being reflected by a target object; and a signal processing link as described in any embodiment of the present application, which compensates the digital baseband signal obtained after processing; and / or a signal calibration and / or compensation link as described in any embodiment of the present application, which is used to calibrate and / or compensate the signal transceiver / transmitter link.

[0488] In some optional embodiments, the integrated circuit may be a millimeter-wave radar chip or a lidar chip (such as an FMCW lidar chip), etc., for obtaining target distance, angle, speed, shape, size, surface roughness, and dielectric properties. Optionally, the integrated circuit may be an antenna-in-package (AiP) chip structure, an antenna-on-package (AoP) chip structure, or an antenna-on-chip (AoC) chip structure.

[0489] In an optional embodiment, different integrated circuits (such as chips) can be combined with each other to form a cascade structure. For the sake of simplicity, they are not described in detail here, but it should be understood that the technologies that people in this field should know based on the contents recorded in this application should be included in the scope of this application.

[0490] The embodiments of the present application also provide a radio device, which may include: a carrier; the integrated circuit described in any embodiment of the present application, which is arranged on the carrier; an antenna, which is arranged on the carrier and is used to transmit and receive radio signals. The antenna can be integrated with the integrated circuit as an integrated device arranged on the carrier (that is, the antenna can be an antenna arranged in an AiP or AoC structure in this case), and the integrated circuit and the antenna can also be two separate components, which are connected to form a system-on-chip (SoC) structure. The carrier can be a printed circuit board (PCB), such as a development board, a data acquisition board or a mainboard of a device, and the first transmission line can be a PCB trace.

[0491] An embodiment of the present application also provides a terminal device, which may include: a device body; a radio device as described in any embodiment of the present application arranged on the device body, and the radio device is used for target detection and / or communication.

[0492] Specifically, based on the above embodiments, in some optional embodiments of the present application, the radio device may be disposed outside the device body or inside the device body, while in other optional embodiments of the present application, the radio device may be disposed partially inside the device body and partially outside the device body. This embodiment of the present application is not limited to this, and the specific configuration may vary depending on the circumstances.

[0493] In some optional embodiments, the above-mentioned device body may be a component or product used in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, in-cabin detection (such as smart cockpits), medical devices, and health care. For example, the device body may be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level / flow rate detection equipment, smart wearable devices (such as bracelets, glasses, etc.), smart home devices (such as sweeping robots, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablets, etc.), as well as gates, smart traffic lights, smart signs, traffic cameras, and various industrial robotic arms (or robots), etc. It can also be various instruments for detecting life characteristic parameters and various devices equipped with the instruments, such as life characteristic detection in car cabins, indoor personnel monitoring, smart medical equipment, consumer electronic devices, etc.

[0494] The radio device may be the radio device described in any embodiment of the present application. The structure and working principle of the radio device have been described in detail in the above embodiments and will not be repeated here.

[0495] It should be noted that radio devices can achieve functions such as target detection and / or communication by transmitting and receiving radio signals to provide detection target information and / or communication information to the device body, thereby assisting or even controlling the operation of the device body.

[0496] For example, when the above-mentioned device body is applied to an advanced driving assistance system (ADAS), the radio device (such as millimeter-wave radar, lidar, etc.) as an on-board sensor can assist the ADAS system to achieve application scenarios such as adaptive cruise control, automatic emergency braking (AEB), blind spot detection warning (BSD), lane change assist (LCA), rear cross traffic alert (RCTA), parking assistance, rear vehicle warning, collision avoidance, pedestrian detection, etc., and can also be used in application scenarios such as car door opening collision avoidance.

[0497] The above-described embodiments merely express preferred embodiments of the present invention and the technical principles employed. While the descriptions are relatively specific and detailed, they should not be construed as limiting the scope of the invention. It is apparent to those skilled in the art that various changes, readjustments, and substitutions can be made without departing from the scope of protection of the present invention. Therefore, although the present invention has been described in detail through the above embodiments, the present invention is not limited to the above embodiments. Without departing from the concept of the present invention, the present invention may also include more other equivalent embodiments. The scope of protection of the present invention patent is determined by the scope of the appended claims.

Claims

1. A signal processing method, applied to a FMCW radar having at least two receiving channels and lines of unequal lengths, wherein the at least two receiving channels include a reference receiving channel and at least one other receiving channel having a length difference with the reference receiving channel; For any of the receiving channels, the method includes: Processing the echo signal received by the receiving channel to obtain a digital baseband signal; The digital baseband signal obtained after the processing is compensated based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitting signal corresponding to the received echo signal.

2. The signal processing method according to claim 1, wherein: The unequal lengths of the lines include: unequal lengths of receiving antenna feed lines and / or unequal lengths of RXLOs.

3. The signal processing method according to claim 1, wherein: The frequency information of the transmission signal corresponding to the received echo signal includes: a frequency sweep bandwidth, a frequency sweep period or a frequency sweep center frequency.

4. The signal processing method according to claim 1, wherein: The unequal lengths of the lines include unequal lengths of receiving antenna feed lines; The compensating the processed digital baseband signal comprises: Acquire the echo signal received by the reference receiving channel according to the time delay caused by the feeder length of the reference receiving antenna corresponding to the reference receiving channel; obtain the digital baseband signal of the reference receiving antenna at the receiving end after processing the echo signal; Generate an echo signal received by the receiving channel i according to the time delay caused by the feeder length of the receiving antenna i corresponding to the receiving channel i and the echo signal received by the reference receiving channel; and obtain a digital baseband signal of the receiving antenna i at the receiving end after processing the echo signal; wherein the receiving channel is any one of the remaining at least one receiving channel; The digital baseband signal of the receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

5. The signal processing method according to claim 1, wherein: The unequal line lengths include unequal RXLO lengths; The compensating the processed digital baseband signal comprises: Acquire an echo signal received by the reference receiving channel according to a time delay generated by a reference RXLO length corresponding to a reference receiving antenna corresponding to the reference receiving channel; and obtain a digital baseband signal of the reference receiving antenna at a receiving end after processing the echo signal; According to the RXLO corresponding to the receiving antenna i corresponding to the receiving channel i i The time delay generated by the length and the echo signal received by the reference receiving channel are used to generate the echo signal received by the receiving channel i; after processing the echo signal, a digital baseband signal of the receiving antenna i at the receiving end is obtained; wherein the receiving channel is any receiving channel of the remaining at least one receiving channel; The digital baseband signal of the receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

6. The signal processing method according to claim 1, wherein: The unequal line lengths include unequal lengths of receiving antenna feed lines and unequal lengths of RXLOs; The compensating the processed digital baseband signal comprises: Acquire an echo signal received by the reference receiving channel according to a time delay generated by a reference receiving antenna feeder length corresponding to the reference receiving channel and a time delay generated by a reference RXLO length corresponding to the reference receiving antenna; and obtain a digital baseband signal of the reference receiving antenna at a receiving end after processing the echo signal; According to the delay caused by the length of the feeder line of the receiving antenna i corresponding to the receiving channel i and the RXLO i The time delay generated by the length and the echo signal received by the reference receiving channel are used to generate the echo signal received by the receiving channel i; after processing the echo signal, a digital baseband signal of the receiving antenna i at the receiving end is obtained; wherein the receiving channel is any receiving channel of the remaining at least one receiving channel; The digital baseband signal of the receiving antenna i at the receiving end is compensated by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

7. The signal processing method according to claim 4, 5 or 6, wherein: The echo signal received by the receiving channel is processed to obtain a digital baseband signal, including: Acquire an echo signal received by the reference channel according to a time delay caused by a reference receiving antenna feeder length corresponding to the reference receiving channel and a time delay caused by a reference RXLO length corresponding to the reference receiving antenna; According to the delay caused by the length of the receiving antenna i feeder corresponding to the receiving channel i and the RXLO corresponding to the receiving antenna i i The time delay caused by the length and the echo signal received by the reference receiving channel are used to generate the echo signal received by the receiving channel i; The echo signal received by the receiving channel i and the echo signal received by the reference receiving channel are processed to obtain a digital baseband signal of the receiving antenna i at the receiving end and a digital baseband signal of the reference receiving antenna at the receiving end.

8. The signal processing method according to claim 7, wherein: Based on the frequency sweep bandwidth, the chirp effective edge period and the delay of the current receiving antenna compared with the reference antenna, the digital baseband signal of the receiving antenna i at the receiving end is compensated.

9. The signal processing method according to claim 8, further comprising: A first phase compensation is performed on the digital baseband signal of the receiving antenna i at the receiving end by using the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

10. The signal processing method according to claim 9, wherein: Based on the sweep bandwidth, the chirp effective edge period, the center frequency of the sweep signal, and the delay of the current receiving antenna compared to the reference antenna, the first phase compensation is performed on the digital baseband signal of the receiving antenna i at the receiving end.

11. The signal processing method according to claim 8 or 10, further comprising: The digital baseband signal of the receiving antenna i at the receiving end is subjected to second phase compensation by utilizing the difference between the digital baseband signal of the receiving antenna i at the receiving end and the digital baseband signal of the reference receiving antenna at the receiving end.

12. The signal processing method according to claim 11, wherein: Based on the frequency sweep bandwidth, the chirp effective edge period, the delay generated by the reference antenna, and the delay of the current receiving antenna compared to the reference antenna, the second phase compensation is performed on the digital baseband signal of the receiving antenna i at the receiving end.

13. The signal processing method according to claim 7, wherein the compensation comprises: Based on the frequency sweep bandwidth, the chirp effective edge period, the center frequency of the frequency sweep signal, and the delay of the current receiving antenna compared to the reference antenna, the digital baseband signal of the receiving antenna i at the receiving end is compensated.

14. The signal processing method according to claim 7, wherein: The acquiring of the echo signal received by the reference receiving channel comprises: Obtain a first receiving signal x1(t-τ) input from the reference receiving antenna to the mixer on the receiving channel where the reference receiving antenna is located via a low noise amplifier LNA. 11 ), and the second receiving signal x1(t-τ) inputted into the mixer on the receiving channel where the reference receiving antenna is located through the local oscillator LO 12 ); The first received signal x1(t-τ 11 ) and the second received signal x1(t-τ 12 ) is correlated to obtain the echo signal x1(t-τ received by the reference receiving channel 11 )x1*(t-τ 12 );in, in, Where, β represents the frequency sweep bandwidth, T represents the time for the frequency of a single chirp signal of the FMCW radar to rise, θ represents the initial phase of the signal, and f c is the center frequency of the sweep signal, τ 11 represents the delay caused by the feeder length of the reference receiving antenna, τ 12 Indicates the time delay caused by the reference RXLO length corresponding to the reference receiving antenna.

15. The signal processing method according to claim 14, wherein: The generating the echo signal received by the receiving channel i includes: Obtain the third receiving signal x1(t-τ) input from the receiving antenna i through the LNA to the mixer on the receiving channel where the receiving antenna i is located 11 -μ i11 ), and the fourth receiving signal x1(t-τ) inputted into the mixer on the receiving channel where the receiving antenna i is located through the local oscillator LO 12 -μ i12 ); The third received signal x1(t-τ 11 -μ i11 ) and the fourth received signal x1(t-τ 12 -μ i12 ) is correlated to obtain the echo x1(t-τ received by the receiving channel i 11 -μ i11 )x1*(t-τ 12 -μ i12 ); in, μ i11 represents the delay caused by the feeder length of the receiving antenna i relative to the feeder length of the reference antenna, μ i12 Indicates the RXLO corresponding to the receiving antenna i i The delay caused by the length relative to the reference RXLO length. 16 . A computer-readable storage medium storing computer-executable instructions, wherein the computer-executable instructions are used to execute the signal processing method according to claim 1 .

17. A signal processing link, applied to a FMCW radar having at least two receiving channels and lines of unequal lengths, comprising: Processing module, compensation module; wherein, A processing module, for processing the echo signal received by any receiving channel to obtain a digital baseband signal; wherein the at least two receiving channels include a reference receiving channel and at least one other receiving channel having a length difference with the reference receiving channel; The compensation module is used to compensate the processed digital baseband signal based on the length difference between the receiving channel and the reference receiving channel and the frequency information of the transmitting signal corresponding to the received echo signal.

18. The signal processing chain of claim 17, wherein: The compensation module uses the received signal compensation method according to any one of claims 1 to 15 to compensate the processed digital baseband signal.

19. A signal processing chain according to claim 17 or 18, wherein: The compensation module includes: a first multiplier M1, a second multiplier M2, and a first digital local oscillator; wherein, The input of the first digital local oscillator includes θ0 and Δθ containing information for compensation, and the output of the first digital local oscillator The output of the first digital local oscillator One input of the first multiplier M1 is the preprocessed received signal of the receiving antenna i at the receiving end. Another input of the first multiplier M1 is the output LO_I of the first digital local oscillator. The output of the first multiplier M1 is the real part Y of the signal received by the compensated receiving antenna i. i _I; One input of the second multiplier M2 is the preprocessed digital baseband signal of the receiving antenna i at the receiving end. Another input of the second multiplier M2 is the output LO_Q of the first digital local oscillator, and the output of the second multiplier M2 is the imaginary part Y of the compensated receiving signal received by the receiving antenna i i _Q.

20. The signal processing chain of claim 19, wherein: The compensation module includes: a third multiplier M3, a fourth multiplier M4, a fifth multiplier M5, a sixth multiplier M6, a first adder S1, a second adder S2, and a second digital local oscillator; wherein, The input of the second digital local oscillator includes θ0 and Δθ containing information for compensation, and the output of the second digital local oscillator The output of the second digital local oscillator One input of the third multiplier M3 is the real part of the digital baseband signal of the receiving antenna i at the receiving end after preprocessing. Another input of the third multiplier M3 is the output LO_I of the second digital local oscillator, and the output of the third multiplier M3 is an input of the second adder; The input of the fourth multiplier M4 is the output LO_Q of the second digital local oscillator, and the output of the fourth multiplier M4 is an input of the first adder; One input of the fifth multiplier M5 is the imaginary part of the digital baseband signal of the receiving antenna i at the receiving end after preprocessing. Another input of the fifth multiplier M5 is the output LO_I of the second digital local oscillator, and the output of the fifth multiplier M5 is another input of the first adder; The input of the sixth multiplier M6 is the output LO_Q of the second digital local oscillator, and the output of the sixth multiplier M6 is connected to another input of the first adder; The two inputs of the first adder S1 are the output of the fourth multiplier M4 and the output of the fifth multiplier M5. The output of the first adder S1 is the real part Y of the signal received by the receiving antenna i after compensation. i -I; The two inputs of the second adder S2 are the output of the third multiplier M3 and the output of the sixth multiplier M6. The output of the first adder S1 is the compensated imaginary part Y of the receiving signal received by the receiving antenna i. i _Q.

21. A signal processing link, comprising: A waveform generator and a plurality of transmission channels connected to the waveform generator, at least two of the transmission channels having different signal transmission delays, and at least one of the transmission channels comprising a direct digital frequency synthesizer; The direct digital frequency synthesizer is configured to generate a compensation signal according to a delay difference of a signal transmission delay of the corresponding transmission channel relative to a reference transmission delay, and use the compensation signal to compensate for a transmission signal of the corresponding transmission channel.

22. A signal processing method, applied to an antenna array of an electromagnetic wave device having at least two signal transmission links, wherein the signal transmission link is a phase shifter including a digital phase shift architecture, and the signal transmission method comprises: Determine a reference transmission delay and a delay difference of a signal transmission delay of each of the transmission channels relative to the reference transmission delay; Generate a compensation signal corresponding to each of the transmitting channels according to the time delay difference; The compensation signal is used to compensate the transmission signal of the corresponding transmission channel.

23. A signal processing link, comprising: Signal transmission link and signal receiving link, where: The signal transmission link includes multiple transmission channels implemented based on analog circuits, each of the transmission channels includes a respective transmission antenna, at least two of the transmission channels have different signal transmission delays, and the multiple transmission antennas transmit electromagnetic wave signals in a time division multiplexing manner; The signal receiving link includes at least one receiving channel, and the receiving channel includes a signal compensator implemented using a digital phase shift architecture; The signal compensator is configured to determine the transmission channel corresponding to the currently received echo signal, generate a compensation signal according to the delay difference of the signal transmission delay of the corresponding transmission channel relative to the reference transmission delay, and use the compensation signal to compensate the currently received echo signal.

24. A signal processing method, applied to an antenna array of an electromagnetic wave device having at least two signal transmission links, the signal transceiving method comprising: Determine a reference transmission delay and a delay difference of a signal transmission delay of each of the transmission channels relative to the reference transmission delay; Generate a compensation signal corresponding to each of the transmitting channels according to the time delay difference; The transmission channel corresponding to the currently received echo signal is determined, and the corresponding compensation signal is used to compensate for the intermediate frequency phase difference and / or frequency difference of the currently received echo signal.

25. A target detection method, applied to a DDM radar sensor having at least one transceiver channel, the method comprising: Performing range-dimensional Fourier transform processing on the echo signal received by the receiving channel to obtain range-dimensional FFT data, and performing phase shift compensation on the range-dimensional FFT data using the phase shift error coefficient of the transmitting channel to obtain range-dimensional FFT compensation data; Performing Doppler dimension Fourier transform processing on the distance dimension FFT compensation data to obtain Doppler dimension FFT data; Based on the Doppler FFT data, velocity information of the target object is determined.

26. A signal processing link, comprising a transmitting link for transmitting an electromagnetic wave signal and a receiving link for receiving an echo signal formed based on the electromagnetic wave signal; The transmission link includes an analog signal source and a digital phase shifter, wherein the analog signal source may be configured to provide an initial analog signal, and the digital phase shifter may be configured to generate a digital phase-shifted signal, and perform a phase shift on the initial analog signal based on the digital phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal; The receiving link includes an analog-to-digital converter and a digital baseband processing module, wherein the analog-to-digital converter may be configured to perform analog-to-digital conversion on the received echo signal to obtain a digital baseband signal, and the digital baseband signal processing module may be configured to perform distance-dimensional Fourier transform and velocity-dimensional Fourier transform on the digital baseband signal in sequence; in, The digital baseband processing module includes a phase compensation unit, which can be configured to use a phase error coefficient of a digital phase shifter in the transmission link to obtain a phase compensation for the distance-dimensional FFT data by Fourier transforming the distance-dimensional FFT data; and The digital baseband processing module may be configured to perform the speed dimension Fourier transform according to the compensated distance dimension FFT data.

27. A method for processing FMCW reception leakage, comprising: In a target-free scenario, performing a range-dimensional Fourier transform on the received signal to obtain range-dimensional FFT data, and extracting first k data of each chirp signal in the range-dimensional FFT data, sorted from low to high in frequency; For each data belonging to the same sequence in the first k data of each group extracted from different chirp signals, a leakage value of the sequence is obtained; wherein, in the first k data of each group, the data belonging to the same sequence have the same corresponding frequency; When performing target detection, the leakage values ​​of each sequence are used to perform subtraction processing on the data of the corresponding sequence of each chirp signal in the range dimension FFT data generated by the currently received signal, so as to achieve leakage elimination of the received signal.

28. A method for processing FMCW reception leakage, comprising: In a target-free scenario, the received signal is processed by range-dimensional Fourier transform to obtain range-dimensional FFT data; Based on one frame of the distance-dimensional FFT data, for a plurality of predetermined distance bins, an average value between different chirp signals is used as a first leakage value of each of the predetermined distance bins.

29. The processing method according to claim 28, further comprising: Based on the distance dimension FFT data of multiple frames, for multiple predetermined distance bins, the median of the first leakage values ​​between different frames is used as the second leakage value of each predetermined distance bin.

30. A signal processing link is used in an electromagnetic wave sensor; The signal processing link is a transmission link; the transmission link includes: an analog signal source and a digital phase shifter; wherein, The analog signal source is configured to provide an initial analog signal; The digital phase shifter is configured to generate a phase shift signal in a digital domain, and perform a phase shift on the initial analog signal based on the phase shift signal, so as to perform a preset phase shift operation on the initial analog signal.

31. A signal calibration link, comprising the signal processing link of claim 30; The receiving antenna connection port of the signal receiving link in the signal processing link is connected to the transmitting antenna connection port of the signal transmitting link in the signal processing link; the signal receiving link is also configured to calibrate the signal transmitting link.

32. A signal calibration link, comprising the signal processing link of claim 30, and a BIST module; The receiving antenna connection port of the signal receiving link in the signal processing link is connected to the transmitting antenna connection port of the signal transmitting link in the signal processing link through the BIST module; The signal receiving link in the signal processing link is further configured to calibrate the signal transmitting link.

33. A signal calibration link, comprising two signal receiving links, a BIST module, an auxiliary circuit unit and the signal processing link according to any one of claims 30, and; Any of the signal receiving links includes a real mixer, a real analog-to-digital converter and a digital signal processing module; the real mixer is configured to down-convert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal, and the echo signal is a signal formed by the signal transmitted by the signal transmitting link being reflected and / or scattered by the target object; the real analog-to-digital converter is configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal; the digital signal processing module is configured to process the digital intermediate frequency signal to obtain the target parameter; The receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection ports of the signal processing link through the auxiliary circuit unit and the BIST module in turn, and the signal receiving link is configured to calibrate the intermediate frequency part of the signal transmitting link.

34. A signal compensation link, comprising the signal processing link according to claim 30, and a compensation module; The compensation module is configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion defects of the signal processing link.

35. An integrated circuit comprising: A signal transceiver channel, used for transmitting a radio signal and receiving an echo signal formed by the radio signal being reflected by a target object; and a signal processing link as claimed in any one of claims 17 to 21, 23 and 26, compensating the digital baseband signal obtained after the processing; And / or, a signal calibration link as described in any one of claims 30-33; And / or, the signal compensation link described in claim 34 is used to calibrate and / or compensate the signal receiving / transmitting link.

36. A radio device comprising: Carrier; The integrated circuit of claim 35, disposed on the carrier; The antenna is arranged on the carrier and is used for sending and receiving radio signals.

37. A terminal device may include: Equipment body; The radio device as claimed in claim 36, which is arranged on the device body, is used for target detection and / or communication.

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