Electric current sensor, and measuring device

Abstract

This electric current sensor includes: a magnetic core that surrounds a measurement target through which an electric current flows; a detection winding that is wound around the magnetic core; a feedback winding that is wound around the magnetic core and that accepts input of a current output from an integrating circuit that integrates a voltage across the detection winding; and a capacitive element. Furthermore, the current sensor adds a current that is output from the detection winding via the capacitive element and a current that is output from the feedback winding, and outputs a converted voltage from an arithmetic circuit that converts the added current into said voltage.

Description

Current sensors and measuring devices

[0001] The present invention relates to a current sensor and a measurement device.

[0002] WO2008-519269A discloses a circuit comprising an AC pickup winding capable of detecting AC currents with frequencies above 1 Hz, and an auxiliary winding which generates an auxiliary current which tends to counteract the electromotive force induced by the main current flowing in a conductor.

[0003] In a current sensor having the circuit described above, in the low frequency band of approximately 1 kHz or less, zero flux operation causes the magnetic flux generated in the magnetic core wound around both the AC detection winding corresponding to the AC pickup winding and the feedback winding corresponding to the auxiliary winding to converge to approximately zero.

[0004] However, as the frequency of the measurement current flowing through the measurement object becomes higher than the low frequency band, the magnetic permeability of the magnetic core decreases and the influence of the parasitic capacitance of each winding and the line-to-line capacitance between the windings increases. As a result, the magnetic flux cancellation action of the feedback winding weakens, magnetic flux remains in the magnetic core, and an induced current flows in the AC detection winding.

[0005] Furthermore, an induced current flows in the AC detection winding in the high frequency band of approximately 10 kHz or more, causing the AC detection winding and feedback winding wound around the magnetic core to function as a current transformer, resulting in a problem of increased detection error in the measured current.

[0006] The present invention has been made in view of these problems, and has as its object to suppress detection errors caused by windings in the high frequency band.

[0007] According to one aspect of the present invention, the current sensor includes a magnetic core that surrounds a measurement object through which a current flows, an AC detection winding wound around the magnetic core, a feedback winding that is wound around the magnetic core and receives a current output from an integration circuit that integrates the voltage between both ends of the AC detection winding, and a capacitance element. The current output from the AC detection winding via the capacitance element and the current output from the feedback winding are added together, and the added current is converted into a voltage (converted into a voltage whose magnitude is proportional to the magnitude of the added current) and the converted voltage is output from an arithmetic circuit.

[0008] According to this aspect, in the high frequency band, the impedance of the capacitance element is low, and the current output from the AC detection winding via the capacitance element and the current output from the feedback winding are added together, so that the AC detection winding operates as a second feedback winding. As a result, the induced current in the AC detection winding becomes the current in the feedback winding and strengthens the current in the feedback winding, thereby suppressing detection errors caused by the windings in the high frequency band.

[0009] FIG. 1 is a diagram showing the configuration of a measurement device including a current sensor according to a first embodiment. FIG. 2 is a schematic diagram showing an example configuration of the current sensor. FIG. 3 is a circuit diagram showing an example detailed configuration of the current sensor. FIG. 4 is a diagram for explaining the operation of the current sensor. FIG. 5 is a diagram showing simulation results of the gain frequency characteristics of the current sensor. FIG. 6 is a circuit diagram showing an example detailed configuration of the current sensor according to a second embodiment. FIG. 7 is a schematic diagram showing an example configuration of the current sensor according to a third embodiment.

[0010] DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, various embodiments of the present invention will be described with reference to the accompanying drawings. In this specification, the same or equivalent elements are designated by the same reference numerals throughout.

[0011] First Embodiment FIG. 1 is a diagram showing the configuration of a measurement device in the first embodiment.

[0012] The measuring device 100 is a device for measuring a physical quantity of a measurement object Lm. The measurement object Lm is, for example, an electric circuit through which a DC or AC measurement current Im flows, and examples of this electric circuit include a single-phase or three-phase power cable. Furthermore, examples of the physical quantity of the measurement object Lm include a current value, a magnetic flux value, and a power value.

[0013] The measurement device 100 in the first embodiment measures the magnitude of a measurement current Im flowing through a measurement object Lm. The measurement device 100 includes a current sensor 1 and a measurement unit 2.

[0014] The current sensor 1 is a sensor that detects magnetic flux generated by an AC measurement current Im. The current sensor 1 detects the magnetic flux generated by the measurement current Im and outputs a voltage as a detected quantity. The current sensor 1 is configured with a magnetic core having multiple windings.

[0015] For example, the current sensor 1 may be realized by a current sensor that employs a winding detection type AC zero flux system (magnetic balance system), or a current sensor that employs this system and also includes a magnetic detection element such as a flux gate or a Hall element. The current sensor 1 may have a through-type structure that surrounds the measurement object Lm, or may have an openable and closable clamp-type structure.

[0016] In the first embodiment, the current sensor 1 detects the amount of detection applied to the measurement current Im by the AC zero-flux method when the frequency of the measurement current Im is within a first frequency band that is equal to or lower than the cutoff frequency of the feedback winding 13 (see FIG. 2 ). On the other hand, when the measurement current Im is within a second frequency band that is higher than the first frequency band, the current sensor 1 detects the amount of detection applied to the measurement current Im by a CT (current transformer) operation of the measurement object Lm wound around the magnetic core 11 and the feedback winding 13.

[0017] Hereinafter, the first frequency band will be referred to as the “negative feedback operating region,” and the second frequency band will be referred to as the “CT operating region.” The current sensor 1 outputs a detected amount of the detected measurement current Im to the measurement unit 2.

[0018] The measurement unit 2 calculates the current value of the measurement current Im for the measurement object Lm based on the detected amount indicating the magnitude of the voltage obtained by the current sensor 1. The measurement unit 2 is realized by, for example, a power analyzer, a current measuring device, and a magnetic field measuring device.

[0019] Next, the circuit configuration of the current sensor 1 will be described with reference to FIG.

[0020] FIG. 2 is a circuit diagram that schematically shows the circuit configuration of the current sensor 1 according to the first embodiment.

[0021] The current sensor 1 comprises a sensor unit 10 for detecting the measurement magnetic flux generated by the measurement current Im, and a circuit unit 20 for converting the output signal of the sensor unit 10 into a voltage whose magnitude is proportional to the magnitude of the measurement magnetic flux.

[0022] The sensor unit 10 includes a magnetic core 11, a detection winding 12, a feedback winding 13, and a capacitance element 14. The circuit unit 20 includes an integral amplifier circuit 21 and an arithmetic circuit 22.

[0023] The magnetic core 11 is formed to surround the measurement object Lm. The overall shape of the magnetic core 11 is formed into an annular shape so that the measurement object Lm can be inserted therethrough. The annular shape here includes a circular shape, an elliptical shape, a rectangular shape, a polygonal shape, and the like.

[0024] The magnetic core 11 in the first embodiment is made of an iron material such as permalloy, but may be realized by a hollow structure instead of an iron material.

[0025] The detection winding 12 is an AC detection winding wound around the magnetic core 11. The detection winding 12 functions as a magnetic flux detection winding in the negative feedback operating region. For example, the detection winding 12 converts the differential magnetic flux (φm-φr) between the measurement magnetic flux φm based on the measurement current Im in the magnetic core 11 and the anti-magnetic flux φr generated by the feedback winding 13 into a voltage V12 whose magnitude is proportional to the time derivative of the differential magnetic flux. Hereinafter, the differential magnetic flux will be referred to as the residual magnetic flux.

[0026] One end of the detection winding 12, which is an output end, is connected to an input end of the integrating amplifier circuit 21, and the other end of the detection winding 12 is connected to a reference potential G. The detection winding 12 is wound around all or part of the magnetic core 11.

[0027] The feedback winding 13 is wound around the magnetic core 11 and is supplied with current by the integral amplifier circuit 21. A current I2 flows through the feedback winding 13 due to electromagnetic induction so as to cancel out the measurement magnetic flux φm generated in the magnetic core 11 by the measurement current Im. At this time, a demagnetizing magnetic flux φr having a direction opposite to that of the measurement magnetic flux φm is generated in the magnetic core 11.

[0028] The feedback winding 13 functions as a current transformer in the CT operating region, and converts the measurement magnetic flux φm generated in the magnetic core 11 only by the feedback winding 13 into a current I2 whose magnitude is proportional to the magnitude of the measurement magnetic flux φm.

[0029] One end of the feedback winding 13 is connected to the output terminal of the integrating amplifier circuit 21, and the other end, which is the output terminal, of the feedback winding 13 is connected to a first input terminal of the arithmetic circuit 22. Like the detection winding 12, the feedback winding 13 is wound around all or part of the magnetic core 11.

[0030] The capacitive element 14 is a capacitive element that bypasses the current flowing through the detection winding 12 to the arithmetic circuit 22 by AC coupling, without passing through the integral amplifier circuit 21. As the capacitive element 14, a capacitor is used, for example.

[0031] One end of the capacitance element 14 is connected between one end of the detection winding 12 and the input end of the integrating amplifier circuit 21, and the other end of the capacitance element 14 is connected to a second input end of the arithmetic circuit 22. The capacitance value of the capacitance element 14 is determined so that the current I1 flowing through the detection winding 12 in the CT operating region passes through the capacitance element 14.

[0032] When the frequency of the measurement current Im is within the negative feedback operating region, the capacitive element 14 prevents the current I1 flowing through the detection winding 12 from passing directly to the arithmetic circuit 22 via the capacitive element 14 itself.

[0033] On the other hand, when the frequency of the measurement current Im is within the CT operating region, the capacitance element 14 directly outputs the current I1 flowing through the detection winding 12 to the arithmetic circuit 22. In this case, the higher the frequency of the measurement current Im, the greater the proportion of the current I1 flowing through the detection winding 12 that passes through the capacitance element 14.

[0034] The circuit unit 20 is, for example, a circuit board on which an integral amplifier circuit 21 and an arithmetic circuit 22 are mounted.

[0035] The integrating amplifier circuit 21 functions as an integrating circuit that integrates the voltage V12 across the detection winding 12. In the first embodiment, the integrating amplifier circuit 21 amplifies and integrates the voltage V12 across the detection winding 12. The integrating amplifier circuit 21 supplies the amplified and integrated current to the feedback winding 13.

[0036] The arithmetic circuit 22 functions as an adder circuit that adds together the current I1 that flows from the detection winding 12 via the capacitance element 14 and the current I2 that flows through the feedback winding 13. The arithmetic circuit 22 also functions as an IV conversion circuit that converts the current obtained by the addition into a voltage whose magnitude is proportional to the magnitude of the current.

[0037] The calculation circuit 22 outputs to the measurement unit 2 a detection amount whose amplitude changes in proportion to the current value of the measurement current Im based on the current I2 flowing through the feedback winding 13 and the current I1 from the detection winding 12 passing through the capacitance element 14.

[0038] For example, when the winding directions of the detection winding 12 and the feedback winding 13 are the same, the calculation circuit 22 adds the current I2 flowing through the feedback winding 13 to the current I1 flowing through the detection winding 12. On the other hand, when the winding directions of the detection winding 12 and the feedback winding 13 are opposite to each other, the calculation circuit 22 subtracts the current I2 flowing through the feedback winding 13 from the current I1 flowing through the detection winding 12.

[0039] Next, the detailed configuration of the current sensor 1 will be described with reference to FIG.

[0040] 3 is a circuit diagram showing a detailed configuration example of the current sensor 1 according to the first embodiment. In the first embodiment, the detection winding 12 and the feedback winding 13 are both wound in the same direction around the circular magnetic core 11, and the detection winding 12 and the feedback winding 13 have the same number of turns.

[0041] In this example, for convenience, the detection winding 12 and the feedback winding 13 are wound around different parts of the magnetic core 11, but the detection winding 12 and the feedback winding 13 may be wound around the same part of the magnetic core 11.

[0042] In the feedback winding 13, an anti-magnetic flux φr is generated by the electromagnetic induction phenomenon to cancel out the measurement magnetic flux φm generated in the magnetic core 11 by the measurement current Im, and an induced current flows.

[0043] The detection winding 12 converts the residual magnetic flux (φm-φr), which is the difference between the measurement magnetic flux φm and the anti-magnetic flux φr generated in the magnetic core 11, into a current I1 whose magnitude is proportional to its magnitude so that the residual magnetic flux (φm-φr) converges to zero through zero flux operation.

[0044] In the first embodiment, one end of the capacitance element 14 is connected to one end 12a of the detection winding 12 which is connected to the input terminal of the integration amplifier circuit 21, and the other end of the capacitance element 14 is connected to the other end 13b of the feedback winding 13 which is connected to the input terminal of the arithmetic circuit 22.

[0045] The integrating amplifier circuit 21 includes a resistive element 211 , a capacitive element 212 , a resistive element 213 , and an operational amplifier 214 .

[0046] In the integrating amplifier circuit 21, one end 12a of the detection winding 12 is connected to one end of a resistive element 211, and the other end of the resistive element 211 is connected to the inverting input terminal (-) of an operational amplifier 214. A capacitive element 212 is connected between the output terminal and the inverting input terminal (-) of the operational amplifier 214, and a resistive element 213 is connected in parallel to the capacitive element 212.

[0047] The other end 12b of the detection winding 12 is connected to the non-inverting input terminal (+) of the operational amplifier 214 and the reference potential G.

[0048] The arithmetic circuit 22 includes a resistive element 221 for converting the flowing current into a voltage proportional to its magnitude, and outputs to the measuring unit 2 the voltage generated across the resistive element 211 by the current flowing into it.

[0049] In the arithmetic circuit 22, the other end 13b of the feedback winding 13 is connected to one end of a resistive element 221, and the other end of the resistive element 221 is connected to a reference potential G. One end of the resistive element 221 is connected to the input end of the measuring unit 2 as an output end that outputs a voltage proportional to the inflow current.

[0050] In the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns, so that the detection sensitivity of both the detection winding 12 and the feedback winding 13 to changes in the magnetic flux generated in the magnetic core 11 is equal.

[0051] Therefore, in the CT operating region of the current sensor 1, it is possible to directly add the current I1 flowing through the detection winding 12 to the current I2 flowing through the feedback winding 13 without adjusting the output levels of the detection winding 12 and the feedback winding 13. Therefore, the arithmetic circuit 22 can be configured with a single resistance element 221.

[0052] Next, the operation of the current sensor 1 will be described with reference to FIGS.

[0053] 4 is a diagram showing the gain frequency characteristics of the current sensor 1 in an ideal state, where the vertical axis represents the gain of the current sensor 1, and the horizontal axis represents the frequency f of the measurement current Im.

[0054] In Figure 4, the gain frequency characteristic of the detection winding 12 is shown by a solid line, the gain frequency characteristic of the feedback winding 13 is shown by a dotted line, and the gain frequency characteristic of the entire current sensor 1 combining these is shown by a dashed line.

[0055] The maximum gain Gmax shown on the vertical axis is the maximum value of the gain, and means the maximum conversion efficiency of converting the magnetic flux interlinking with the windings into current. Ideally, the maximum gain Gmax is a value obtained by dividing the resistance value R of the load resistor by the number of turns N of the detection winding 12 and the feedback winding 13.

[0056] Furthermore, the negative feedback operating region defined by the horizontal axis is the frequency band in which negative feedback operation is performed to converge the magnetic flux generated in the magnetic core 11 by the detection winding 12 to zero, and the CT operating region is the frequency band in which the feedback winding 13 operates as a current transformer.

[0057] In the first embodiment, the negative feedback operating region is a frequency band from the cutoff frequency fc of the current sensor 1 including the detection winding 12 and the feedback winding 13 to the cutoff frequency fc_ct of the feedback winding 13. The CT operating region is a frequency band higher than the cutoff frequency fc_ct of the feedback winding 13.

[0058] In the negative feedback operating region, a measurement magnetic flux φm, which is part of the magnetic flux produced by the measurement current Im and interlinks with the feedback winding 13, generates an anti-magnetic flux φr in the magnetic core 11 in the opposite direction to the measurement magnetic flux φm. As a result, a current is generated in the feedback winding 13 so as to cancel out the measurement magnetic flux φm.

[0059] At this time, a differential magnetic flux between the measurement magnetic flux φm and the anti-magnetic flux φr remains in the magnetic core 11, and this residual magnetic flux generates a voltage V12 between one end 12a and the other end 12b of the detection winding 12. At this time, the current passing through the capacitance element 14 is substantially zero, and the voltage V12 between one end 12a and the other end 12b of the detection winding 12 is input to the integral amplifier circuit 21.

[0060] A voltage V12 outputted between one end 12a and the other end 12b of the detection winding 12 is inputted to an integrating amplifier circuit 21, where the voltage V12 outputted from the detection winding 12 is integrated and amplified.

[0061] The current output from the integrating amplifier circuit 21 is input to one end 13a of the feedback winding 13, and the output current of the integrating amplifier circuit 21 is added to the current generated in the feedback winding 13. As a result, the current I2 having a magnitude equal to the sum of the output current of the integrating amplifier circuit 21 and the induced current of the feedback winding 13 is output from the other end 13b of the feedback winding 13 to the arithmetic circuit 22.

[0062] As shown in Figure 4, as the frequency of the measurement current Im approaches the cutoff frequency fc of the current sensor 1 from a frequency lower than fc, the gain of the current sensor 1 increases monotonically, and the gain of the current sensor 1 reaches a maximum gain Gmax near the cutoff frequency fc.

[0063] In the negative feedback operating region, the gain of the current sensor 1 is maintained at the maximum gain Gmax, and as the frequency of the measurement current Im increases, the gain of the current transformer composed of the measurement object Lm and the feedback winding 13 monotonically increases toward the maximum gain Gmax.

[0064] In the CT operating region, the measurement magnetic flux φm is detected only by the feedback winding 13. At this time, the feedback winding 13 converts the measurement magnetic flux φm into a current I2 whose magnitude is proportional to the magnitude of the measurement magnetic flux φm with higher accuracy than in the negative feedback operating region.

[0065] That is, a current flows through the feedback winding 13 so as to prevent any magnetic flux from remaining in the magnetic core 11, in other words so as to cancel out all of the measurement magnetic flux φm. Then, only the induced current generated in the feedback winding 13 is output as a current I2 from the other end 13b, and this current I2 is input to one end of the resistor element 221 in the arithmetic circuit 22.

[0066] In the CT operating region, as the frequency of the measurement current Im increases, the magnetic permeability of the magnetic core 11 decreases, and the influence of the parasitic capacitance of each of the detection winding 12 and the feedback winding 13 and the line capacitance generated between the detection winding 12 and the feedback winding 13 increases. As a result, the conversion efficiency of the feedback winding 13 decreases, and magnetic flux remains in the magnetic core 11. This causes an induced current to flow in the detection winding 12.

[0067] The current I2 flowing through the detection winding 12 causes the gain to drop significantly from the maximum gain Max near the upper limit of the CT operating region. At this time, the frequency of the measurement current Im is higher than the negative feedback operating region, so the integrating amplifier circuit 21 shown in Fig. 3 functions as a low-pass filter, and therefore the integrating amplifier circuit 21 does not operate, and the operational amplifier 214 does not integrate or amplify the current output from one end 12a of the detection winding 12. Therefore, the current flowing through the detection winding 12 is not superimposed on the feedback winding 13 through the integrating amplifier circuit 21.

[0068] In the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns, so the current I1 flowing through the detection winding 12 is directly added to the current I2 flowing through the feedback winding 13 via the capacitance element 14. This compensates for the decrease in the current I2 that accompanies a decrease in the conversion efficiency of the feedback winding 13.

[0069] Then, the current I1 flowing through the detection winding 12 is added to the current I2 flowing through the feedback winding 13 as an added current, that is, a current (I1+I2), which is input to one end of the resistance element 221 in the arithmetic circuit 22.

[0070] Here, the capacitance value of the capacitive element 14 will be briefly described.

[0071] It is desirable to set the capacitance value of the capacitance element 14 so that the current I1 flowing through the detection winding 12 passes through the capacitance element 14 in the CT operating region.

[0072] The capacitance value of the capacitance element 14 is determined based on the cutoff frequency of a high-pass filter configured by the series-connected capacitance element 14 and resistance element 221. The cutoff frequency fc_hpf of this high-pass filter is calculated using the capacitance value C of the capacitance element 14 and the resistance value R of the resistance element 221, as shown in the following equation (1).

[0073] fc_hpf = 1 / (2πCR) ... (1)

[0074] Specifically, the capacitance value C of the capacitive element 14 is determined so that the cutoff frequency fc_hpf of the high-pass filter is a predetermined frequency that is higher than the cutoff frequency fc_ct of the feedback winding 13 and lower than the upper limit of the CT operating region.

[0075] The cutoff frequency fc_ct of the feedback winding 13 is calculated by the following equation (2) using the inductance value L of the feedback winding 13 and the resistance value R of the resistive element 221, without taking the winding resistance value into consideration: fc_ct=R / (2πL) (2)

[0076] FIG. 5 is a diagram showing a simulation result of the gain frequency characteristic of the current sensor 1 according to the first embodiment.

[0077] Here, the vertical axis represents the gain of the current sensor 1 on a linear scale, and the horizontal axis represents the frequency of the measurement current Im on a logarithmic scale. The gain-frequency characteristics obtained by the simulation results of the current sensor 1 in the first embodiment are shown by a solid line, and the gain-frequency characteristics of a current sensor not including the capacitive element 14 as a comparative example are shown by a dashed line.

[0078] As shown in Figure 5, in the current sensor of the comparative example, as the frequency of the measurement current Im increases from around 1 MHz, the gain gradually decreases from the maximum gain Gmax of approximately -34 dB, and then decreases sharply from around 20 MHz.

[0079] In contrast, in the current sensor 1 of the first embodiment, even in the high frequency band where the gain of the current sensor of the comparative example is reduced, the current flowing in the detection winding 12 is supplied through the capacitance element 14 to the calculation circuit 22 which calculates the current flowing in the feedback winding 13.

[0080] As a result, the arithmetic circuit 22 outputs a voltage value corresponding to the magnitude of the current obtained by adding the current I1 flowing through the detection winding 12 to the current I2 flowing through the feedback winding 13. In other words, the detection amount of the arithmetic circuit 22 is corrected by the current I1 of the detection winding 12.

[0081] In this way, in the current sensor 1 of the first embodiment, the decrease in the current I2 flowing through the feedback winding 13 is compensated for by using the current I1 that occurs in the detection winding 12 as the conversion efficiency of the feedback winding 13 decreases. Therefore, the decrease in the gain of the current sensor 1 in the high frequency band can be suppressed more effectively than in the current sensor of the comparative example.

[0082] Next, the effects of the current sensor 1 of the first embodiment will be described.

[0083] The current sensor 1 in the first embodiment includes a magnetic core 11 surrounding a measurement object Lm through which a measurement current Im flows, a detection winding 12 functioning as an AC detection winding wound around the magnetic core 11, and a feedback winding 13 functioning as a feedback winding wound around the magnetic core 11. A current is supplied to the feedback winding 13 by an integrating amplifier circuit 21 functioning as an integrating circuit that integrates the current flowing through the detection winding 12.

[0084] Furthermore, the current sensor 1 includes a capacitance element 14, and outputs a voltage output from an arithmetic circuit 22. The arithmetic circuit 22 adds the current output from the detection winding 12 via the capacitance element 14 and the current output from the feedback winding 13, and converts the sum into a voltage proportional to the magnitude of the current obtained by the addition.

[0085] According to this configuration, in the CT operating region where the feedback winding 13 operates as a current transformer, the higher the frequency of the measurement current Im, the greater the proportion of the current flowing through the detection winding 12 that is input to the arithmetic circuit 22 via the capacitance element 14.

[0086] At this time, as the frequency of the measurement current Im increases, the magnetic permeability of the magnetic core 11 decreases and the influence of the parasitic capacitance of each of the detection winding 12 and feedback winding 13 and the line-to-line capacitance between the windings increases. As a result, the magnetic flux cancellation action by the feedback winding 13 weakens, magnetic flux remains in the magnetic core 11, and an induced current flows in the detection winding 12. Then, the induced current flows in the detection winding 12, causing the detection winding 12 and feedback winding 13 wound around the magnetic core 11 to function as a current transformer, resulting in a large detection error in the measurement current.

[0087] Therefore, in the CT operating region, the detection winding 12 operates as a second feedback winding by using the current passing through the capacitive element 14, which makes it possible to compensate for the decrease in induced current in the feedback winding 13 that accompanies a decrease in the conversion efficiency of the feedback winding 13. Therefore, it is possible to suppress detection errors in the measurement current Im in the high frequency region.

[0088] In the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns.

[0089] According to this configuration, by making the number of turns of the detection winding 12 and the number of turns of the feedback winding 13 equal, the degree of influence on the induced current due to a decrease in the conversion efficiency of the detection winding 12 and the feedback winding 13 is equalized. Therefore, there is no need to adjust the output level of the current I2 flowing through the feedback winding 13 and the output level of the current I1 of the detection winding 12 passing through the capacitive element 14, and the current I2 and the current I1 can be directly added together.

[0090] Therefore, since the added current can be detected by one resistor element 221 in the arithmetic circuit 22, the arithmetic circuit 22 can have a simple configuration.

[0091] In the first embodiment, the capacitance element 14 is connected between the output terminal of the detection winding 12 and the output terminal of the feedback winding 13 .

[0092] According to this configuration, the capacitance element 14 is connected between the output terminal of the detection winding 12 and the output terminal of the feedback winding 13, so it is possible to arrange the capacitance element 14 close to the detection winding 12 and the feedback winding 13. This makes it possible to shorten the wiring distance from the output terminals of the detection winding 12 and the feedback winding 13 to the capacitance element 14, compared to a configuration in which the wiring is connected on the board on which the integrating amplifier circuit 21 and the arithmetic circuit 22 are mounted. This makes it possible to suppress noise from being mixed into the capacitance element 14.

[0093] In the first embodiment, the arithmetic circuit 22 includes a resistive element 221, and the series-connected capacitive element 14 and resistive element 221 form a filter. The capacitance value C of the capacitive element 14 is determined so that the cutoff frequency fc_hpf of the filter expressed by equation (1) is a predetermined frequency higher than the cutoff frequency fc_ct of the feedback winding 13.

[0094] According to this configuration, the capacitive element 14 can supply the current I1 flowing through the detection winding 12 to the arithmetic circuit 22 in the high frequency band within the CT operating region.

[0095] The current sensor 1 in the first embodiment also includes an integral amplifier circuit 21 and an arithmetic circuit 22 .

[0096] According to this configuration, the arithmetic circuit 22 can output a detection amount in which detection errors caused by a decrease in the conversion efficiency of the feedback winding 13 in the high frequency band are suppressed.

[0097] The measurement device 100 in the first embodiment also includes a current sensor 1 and a measurement unit 2 that calculates a measurement quantity for the measurement object Lm based on the detection quantity detected by the current sensor 1.

[0098] According to this configuration, it is possible to suppress detection errors caused by a decrease in the conversion efficiency of the feedback winding 13 in the high frequency band, thereby improving the accuracy of the measured quantity in the high frequency band.

[0099] In the current sensor 1 of the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns, but the numbers of turns may be different. Therefore, a detailed configuration of a current sensor 1 in which the detection winding 12 and the feedback winding 13 have different numbers of turns will be described below as a second embodiment.

[0100] Second Embodiment FIG. 6 is a circuit diagram showing a detailed configuration of a current sensor 1A according to a second embodiment.

[0101] In the second embodiment, the number of turns N1 of the detection winding 12 is different from the number of turns N2 of the feedback winding 13. Accordingly, the current sensor 1A differs from the first embodiment in that it includes an arithmetic circuit 22A instead of the arithmetic circuit 22 of the current sensor 1 shown in Fig. 3, and the connection location of the capacitance element 14 is changed. Therefore, only the differences in configuration from the first embodiment will be described, and the same reference numerals will be used to denote the other configuration elements, and description thereof will be omitted.

[0102] The arithmetic circuit 22A adjusts the output level of the current I2 flowing through the feedback winding 13 and the output level of the current I1 passing from the detection winding 12 to the capacitive element 14, and outputs the sum of the adjusted currents I2 and I1. The arithmetic circuit 22A includes a resistive element 221, a resistive element 222, and an adder 223.

[0103] The resistive element 221 is a second resistor that functions as a second detection element that detects the current I2 flowing through the feedback winding 13. The resistive element 221 outputs a voltage to the adder 223, the voltage being proportional to the magnitude of the current I2 flowing through the feedback winding 13.

[0104] The resistance value of the resistive element 221 is determined based on the ratio between the number of turns N1 of the detection winding 12 and the number of turns N2 of the feedback winding 13 so that the output levels of both the detection winding 12 and the feedback winding 13 are equal.

[0105] Specifically, the resistance value R2 of the resistive element 221 is determined using the number of turns N1 of the detection winding 12, the number of turns N2 of the feedback winding 13, and the resistance value R1 of the resistive element 222 so that the relationship of the following equation (3) holds.

[0106] R2 / R1=N2 / N1...(3)

[0107] The resistive element 222 is a first resistor that functions as a first detection element for detecting the current I2 that passes through the capacitive element 14, out of the current I2 that flows through the detection winding 12. The resistive element 222 outputs a voltage to the adder 223, the voltage being proportional to the magnitude of the current that flows through the detection winding 12. The resistance value R1 of the resistive element 222 is determined, similar to the resistance value R2 of the resistive element 221, so that the relationship in the above formula (3) is established.

[0108] The adder 223 adds the voltage generated across the resistor element 221 and the voltage generated across the resistor element 222. The adder 223 outputs the sum of the voltage values ​​obtained by the addition to the measurement unit 2.

[0109] The voltage across the resistor element 221 and the voltage across the resistor element 222 indicate voltage values ​​adjusted for the difference in output level between the detection winding 12 and the feedback winding 13, which is caused by the difference between the number of turns N2 of the feedback winding 13 and the number of turns N1 of the detection winding 12.

[0110] Therefore, even if the detection winding 12 and the feedback winding 13 have different numbers of turns, it is possible to compensate for the decrease in current I2 due to a decrease in the conversion efficiency of the feedback winding 13 by using the current I2 of the detection winding 12 passing through the capacitance element 14.

[0111] In the arithmetic circuit 22A, the other end 13b of the feedback winding 13 is connected to one end of a resistive element 221, and the other end of the resistive element 221 is connected to a reference potential G. In addition, one end of the capacitive element 14 is connected to one end 12a of the detection winding 12, and the other end of the capacitive element 14 is connected to one end of a resistive element 222, and the other end of the resistive element 222 is connected to the reference potential G.

[0112] One end of the resistive element 222 is connected to a first input terminal of the adder 223, and one end of the resistive element 221 is connected to a second input terminal of the adder 223. The output terminal of the adder 223 is connected to the input terminal of the measurement unit 2.

[0113] In this way, even if the number of turns of the detection winding 12 and the feedback winding 13 are different from each other, by determining the resistance values ​​of the resistive elements 221 and 222 based on the ratio of the number of turns of each winding, the amount of decrease in current I2 can be appropriately compensated for using the current passing through the capacitive element 14.

[0114] The capacitance value of the capacitance element 14 is determined using the above formula (1), as in the first embodiment. Specifically, the capacitance value C of the capacitance element 14 is set so that the cutoff frequency fc_hpf of the high-pass filter having the capacitance element 14 and the resistance element 222 connected in series is higher than the cutoff frequency fc_ct of the feedback winding 13.

[0115] Next, the effects of the current sensor 1A according to the second embodiment will be described.

[0116] In the second embodiment, the arithmetic circuit 22A includes a resistive element 222 that functions as a first resistor for detecting the current I1 flowing through the detection winding 12, and a resistive element 221 that functions as a second resistor for detecting the current I2 flowing through the feedback winding 13.

[0117] The arithmetic circuit 22A further includes an adder 223 that functions as an adder that adds the detection amounts detected by the resistance element 222 and the resistance element 221. The capacitance element 14 is connected between the input terminal of the integrating amplifier circuit 21 and one input terminal of the resistance element 222.

[0118] According to this configuration, the resistance values ​​of the resistive elements 222 and 221 are determined based on the ratio of the number of turns N1 and N2 of the detection winding 12 and the feedback winding 13, making it possible to adjust the output levels of the detection winding 12 and the feedback winding 13.

[0119] For example, one end of the resistive element 222 is connected to the output end of the detection winding 12, one end of the resistive element 221 is connected to the output end of the feedback winding 13, and the other end of the resistive element 222 and the other end of the resistive element 221 are each connected to the reference potential G. Then, as in the above equation (3), the ratio of the number of turns N1 of the detection winding 12 to the resistance value R1 of the resistive element 222 is equal to the ratio of the number of turns N2 of the feedback winding 13 to the resistance value R2 of the resistive element 221.

[0120] With this configuration, the output levels of the detection winding 12 and the feedback winding 13, which have different numbers of turns, can be adjusted so that the residual magnetic flux (φm-φr) converges to zero. Therefore, even if the detection winding 12 and the feedback winding 13 have different numbers of turns, the amount of decrease in the current I2 that occurs due to a decrease in the conversion efficiency of the feedback winding 13 can be appropriately compensated for by using the current I2 that passes from the detection winding 12 through the capacitive element 14 in the high-frequency band.

[0121] Third Embodiment FIG. 7 is a diagram schematically illustrating the configuration of a current sensor 1B according to a third embodiment.

[0122] The current sensor 1B includes a magnetic detection element in addition to the configuration of the sensor unit 10 shown in Fig. 2. The current sensor 1B according to the third embodiment includes a sensor unit 10B and a circuit unit 20B.

[0123] The sensor unit 10B includes, in addition to the configuration of the sensor unit 10 shown in FIG. 2, a flux gate 110 as a magnetic detection element disposed in all or part of the annular magnetic core 11.

[0124] The fluxgate 110 in the third embodiment includes annular magnetic cores 111 and 112 , an excitation winding 15 wound around the magnetic core 111 , and an excitation winding 16 wound around the magnetic core 112 .

[0125] The magnetic cores 111 and 112 are made of, for example, permalloy material and are arranged side by side within the magnetic core 11 .

[0126] The excitation windings 15 and 16 are wound around the magnetic cores 111 and 112, respectively, so that the magnetic fluxes generated in the magnetic cores 111 and 112 cancel each other out.

[0127] The circuit section 20B includes an oscillator circuit 31, an excitation circuit 32, a detection circuit 33, an adder circuit 34, and a feedback circuit 35 in addition to the configuration of the circuit section 20 shown in FIG.

[0128] The oscillator circuit 31 generates an AC signal having a specific frequency f, and generates a synchronization signal having a double frequency 2f, which is twice the specific frequency. The oscillator circuit 31 then outputs the generated AC signal having the specific frequency f to the excitation circuit 32, and outputs the generated synchronization signal having the double frequency 2f to the detection circuit 33.

[0129] The excitation circuit 32 generates an AC excitation current of frequency f based on the AC signal output from the oscillation circuit 31. The excitation circuit 32 outputs the generated AC excitation current to both the excitation windings 15 and 16.

[0130] As a result, a magnetic flux is generated in the magnetic core 111 which is the difference between a first magnetic flux generated by the excitation winding 15 and a second magnetic flux generated by the excitation winding 16, which is in the opposite direction to the first magnetic flux.

[0131] The detection circuit 33 synchronously detects the difference signal between the voltage signal of the excitation winding 15 and the voltage signal of the excitation winding 16 based on the synchronization signal of the oscillation circuit 31. The detection circuit 33 outputs a signal proportional to the current to be measured generated in the magnetic cores 111 and 112.

[0132] The adder circuit 34 adds the output signal of the integral amplifier circuit 21 and the output signal of the detection circuit 33 together, and outputs the resulting signal to the feedback circuit 35 .

[0133] The feedback circuit 35 converts the sum signal obtained from the addition circuit 34 into a current and outputs it to the feedback winding 13 .

[0134] In this way, by including a flux gate 110 in addition to the configuration of current sensor 1 or current sensor 1A, current sensor 1B is able to obtain a detection quantity whose amplitude changes in proportion to the value of the measured current Im in a frequency band lower than the negative feedback operating region.

[0135] In the third embodiment, the fluxgate 110 has a configuration in which the two excitation windings 15 and 16 are wound around one magnetic core 111, but the present invention is not limited to this.

[0136] For example, the fluxgate 110 may have two annular magnetic cores, with the excitation winding 15 wound around one magnetic core and the excitation winding 16 wound around the other magnetic core. In this case, one magnetic core around which the excitation winding 15 is wound and the other magnetic core around which the excitation winding 16 is wound are arranged side by side within the magnetic core 11.

[0137] In addition, in the third embodiment, the two excitation windings 15 and 16 are wound around the magnetic core 111 , but either one of them may be wound around the magnetic core 11 .

[0138] Although the flux gate 110 is used as the magnetic detection element in the third embodiment, a Hall element may be used instead. In this case, the Hall element is disposed in a part of the magnetic core 11, and an output signal from the Hall element is input to the adder circuit 34, where the output signal from the integral amplifier circuit 21 and the output signal from the Hall element are added together.

[0139] Next, the effects of the current sensor 1B according to the third embodiment will be described.

[0140] A current sensor 1B in the third embodiment has the configuration of the first and second embodiments.

[0141] According to this configuration, similar to the effects of the first and second embodiments, in the CT operating region, it is possible to compensate for the decrease in induced current in the feedback winding 13 that accompanies a decrease in the conversion efficiency of the feedback winding 13 by using the current passing through the capacitive element 14. Therefore, it is possible to suppress detection errors in the measurement current Im in the high frequency region.

[0142] Moreover, the current sensor 1B in the third embodiment further includes a magnetic detection element that detects the magnetic flux generated in the magnetic core 11. This makes it possible to improve the detection accuracy in a frequency band lower than the CT operating range.

[0143] Although the embodiments of the present invention have been described above, the above embodiments merely illustrate some of the application examples of the present invention, and it is not intended that the technical scope of the present invention be limited to the specific configurations of the above embodiments.

[0144] This application claims priority based on Japanese Patent Application No. 2023-149392, filed with the Japan Patent Office on September 14, 2023, the entire contents of which are incorporated herein by reference.

[0145] REFERENCE SIGNS LIST 1, 1A, 1B Current sensor 2 Measuring unit 11 Magnetic core 12 Detection winding (AC detection winding) 13 Feedback winding 21 Integrating amplifier circuit (integrating circuit) 22 Arithmetic circuit 100 Measuring device 110 Fluxgate (magnetic detection element) Lm Measurement object Im Measurement current

Claims

1. A current sensor comprising: a magnetic core surrounding a measurement object through which a current flows; an AC detection winding wound around said magnetic core; a feedback winding wound around said magnetic core and receiving as input a current output from an integrating circuit that integrates the voltage between both ends of said AC detection winding; and a capacitive element, which adds together the current output from said AC detection winding via said capacitive element and the current output from said feedback winding, and outputs the converted voltage from an arithmetic circuit that converts the added current into a voltage.

2. A current sensor according to claim 1, wherein the AC detection winding and the feedback winding have an equal number of turns.

3. A current sensor according to claim 1 or 2, wherein the capacitance element is connected between an output terminal of the AC detection winding and an output terminal of the feedback winding.

4. A current sensor as described in claim 1, wherein the AC detection winding and the feedback winding have different numbers of turns, the arithmetic circuit has a first resistor and a second resistor, one end of the first resistor is connected to the output end of the AC detection winding, one end of the second resistor is connected to the output end of the feedback winding, the other end of the first resistor and the other end of the second resistor are each connected to a reference potential, and the ratio of the number of turns of the AC detection winding to the resistance value of the first resistor is equal to the ratio of the number of turns of the feedback winding to the resistance value of the second resistor.

5. A current sensor according to any one of claims 1 to 4, further comprising a magnetic detection element for detecting a magnetic flux generated in the magnetic core.

6. A current sensor as claimed in any one of claims 1 to 5, wherein the arithmetic circuit has one or more resistive elements, and the capacitance value of the capacitive element is determined so that the cutoff frequency of a filter formed from the capacitive element and at least one of the resistive elements connected in series becomes a predetermined frequency higher than the cutoff frequency of a current transformer formed from the feedback winding and the resistive element.

7. A current sensor according to any one of claims 1 to 6, comprising the integrating circuit and the arithmetic circuit.

8. A measuring device comprising: a current sensor according to any one of claims 1 to 7; and a measuring section which calculates a measurement quantity for the measurement object based on a detection quantity detected by the current sensor.

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