Antenna module and communication device equipped with same

Abstract

This antenna module comprises a plurality of radiation elements (600) and a plurality of transmission circuits (200A). Each radiation element has power supply points (V1, V2). Each transmission circuit includes a phase shift circuit (110) and an amplifier circuit (CT1A). The amplifier circuit amplifies a signal from the phase shift circuit and transmits the amplified signal to each power supply point. The amplifier circuit includes a carrier amplifier (141A), a hybrid coupler (160A), switches (S1A to S3A), and phase shift lines (171A, 172A). The hybrid coupler includes lines (161A to 164A). The switch (S1A) is connected between a ground potential (GND) and a connection node (N1A) between the line (161A) and the line (162A). The switch (S2A) is connected between the ground potential and a connection node (N2A) between the line (162A) and the line (164A). The switch (S3A) is connected between the ground potential and a connection node (N3A) between the line (163A) and the line (164A). The phase shift line (171A) is connected between the connection node (N3A) and the power supply point (V1). The phase shift line (172A) is connected between the connection node (N2A) and the power supply point (V2). A signal from the carrier amplifier (141A) is supplied to a connection node (N4A) between the line (161A) and the line (163A).

Description

Antenna module and communication device equipped with same

[0001] The present disclosure relates to an antenna module and a communication device equipped with the same, and more particularly to a phase adjustment circuit for beamforming in an array antenna.

[0002] Array antennas in which multiple radiating elements are arranged linearly or planarly are known, such as that disclosed in International Publication No. 2023 / 210118 (Patent Document 1). In such array antennas, it is possible to change the directionality of radiated radio waves by adjusting the phase of a high-frequency signal supplied to each radiating element, thereby performing so-called beamforming.

[0003] WO 2023 / 210118

[0004] When performing beamforming, a phase shifter that adjusts the phase of the high-frequency signal supplied to each radiating element may be configured, for example, to appropriately bypass multiple LC circuits for phase adjustment that are connected in series using semiconductor switches.

[0005] Meanwhile, in recent years, developments in wireless communication devices have been progressing toward communications in the so-called sub-terahertz frequency band, which exceeds 100 GHz, with the aim of achieving high-capacity, high-speed communications. Signals in such sub-terahertz frequency bands tend to have greater losses in their signal paths than signals in millimeter waves and lower frequency bands. Therefore, when a phase shifter having a switch arranged in series on the signal path is used, as described above, the loss in the switch may prevent the desired characteristics from being achieved.

[0006] The present disclosure has been made to solve such problems, and its purpose is to provide a phase adjustment circuit that can reduce losses during beamforming in an array antenna.

[0007] An antenna module according to a first aspect of the present disclosure includes a plurality of radiating elements each having a flat plate shape and a plurality of transmitting circuits connected to the radiating elements, respectively. Radio waves are radiated from the radiating elements. Each of the radiating elements has a first feed point and a second feed point offset from the center of the radiating element along a first direction in opposite directions. Each of the transmitting circuits includes a phase-shift circuit configured to change the phase of a high-frequency signal and an amplifier circuit that amplifies the signal from the phase-shift circuit. The amplifier circuit amplifies the signal from the phase-shift circuit and transmits it to the first feed point and second feed point of the corresponding radiating element. The amplifier circuit includes an amplifier stage, a first hybrid coupler, first to third switches, and first and second phase-shift lines. The first hybrid coupler includes first to fourth lines connected in a ring shape. The first switch is connected between a first connection node between the first and second lines and ground potential. The second switch is connected between a second connection node between the second line and the fourth line and ground potential. The third switch is connected between a third connection node between the third line and the fourth line and ground potential. The first phase shift line is connected between the third connection node and the first feed point. The second phase shift line is connected between the second connection node and the second feed point. A signal from the amplifier stage is supplied to a fourth connection node between the first line and the third line.

[0008] An antenna module according to a second aspect of the present disclosure includes a plurality of radiating elements each having a flat plate shape and a plurality of transmitting circuits connected to the radiating elements, respectively. Radio waves are radiated from the radiating elements. Each of the radiating elements has a first feed point to a fourth feed point. In each radiating element, the first feed point and the second feed point are disposed at positions offset in opposite directions from the center of the radiating element along a first direction. In each radiating element, the third feed point and the fourth feed point are disposed at positions offset in opposite directions from the center of the radiating element along a second direction intersecting the first direction. Each of the transmitting circuits includes a phase shift circuit, a branch circuit, a first amplifier circuit, and a second amplifier circuit. The phase shift circuit is configured to change the phase of a high-frequency signal. The branch circuit branches a signal from the phase shift circuit into two paths. The first amplifier circuit and the second amplifier circuit amplify the signal from the branch circuit and transmit it to the corresponding radiating element. Each of the first amplifier circuit and the second amplifier circuit includes an amplifier stage that amplifies a signal from the phase shift circuit, a first hybrid coupler, first to third switches, and a first and second phase shift line. The first hybrid coupler includes first to fourth lines connected in a ring. The first switch is connected between a first connection node between the first and second lines and ground potential. The second switch is connected between a second connection node between the second and fourth lines and ground potential. The third switch is connected between a third connection node between the third and fourth lines and ground potential. One end of the first phase shift line is connected to the third connection node. One end of the second phase shift line is connected to the second connection node. In each amplifier circuit, the signal from the amplifier stage is supplied to a fourth connection node between the first and third lines. The other end of the first phase shift line of the first amplifier circuit is connected to a first feed point. The other end of the second phase shift line of the first amplifier circuit is connected to a second feed point. The other end of the first phase shift line of the second amplifier circuit is connected to a third feed point. The other end of the second phase shift line of the second amplifier circuit is connected to a fourth feed point.

[0009] In the antenna module according to the present disclosure, a hybrid coupler is disposed after the amplification stage, and radio-frequency signals are supplied from two output terminals (second connection node, third connection node) of the hybrid coupler to two feed points on the radiating element, each of which is positioned symmetrically about the center. A signal from the amplification stage is supplied to one input terminal (first connection node) of the hybrid coupler. The other input terminal (fourth connection node) and the two output terminals of the hybrid coupler are provided with switches between ground potential and the other input terminal (fourth connection node). In this configuration, the phase of the radio-frequency signal supplied to the radiating element can be changed by switching the switch to select the output terminal that outputs the radio-frequency signal. Since the switches are not connected in series to the signal path, no loss occurs due to the switching of the switches. Therefore, a phase adjustment circuit capable of reducing loss during beamforming in an array antenna can be provided.

[0010] 1 is a schematic configuration diagram of a communication device to which an antenna module according to a first embodiment is applied. FIG. 2 is a diagram showing a detailed configuration of the transmission circuit in FIG. 1. FIG. 3 is a diagram for explaining variations in signal combining in an amplifier stage. FIG. 4 is a diagram for explaining the relationship between output power and efficiency accompanying amplifier switching in an amplifier stage in the case of series combining. FIG. 5 is a diagram showing a first form of an output stage when radiating V polarized waves. FIG. 6 is a diagram showing a second form of an output stage when radiating V polarized waves. FIG. 7 is a diagram showing a first form of an output stage when radiating H polarized waves. FIG. 8 is a diagram showing a second form of an output stage when radiating H polarized waves. FIG. 9 is a diagram showing a first form of an output stage when radiating circular polarized waves. FIG. 10 is a diagram showing a second form of an output stage when radiating circular polarized waves. FIG. 11 is a diagram showing the influence of load fluctuations at maximum power output in the case of parallel combining. FIG. 12 is a diagram showing the influence of load fluctuations at back-off in the case of parallel combining. FIG. 13 is a diagram showing the influence of load fluctuations at maximum power output in the case of parallel combining. FIG. 14 is a diagram showing the influence of load fluctuations at back-off in the case of parallel combining. FIG. 15 is a diagram showing the influence of load fluctuations at maximum power output in the case of series combining. Fig. 2 is a second diagram for explaining the influence of load fluctuations at the time of maximum power output in the case of series combining. Fig. 2 is a second diagram for explaining the influence of load fluctuations at the time of back-off in the case of series combining. Fig. 3 is a diagram showing a detailed configuration of a transmission circuit in embodiment 2. Fig. 4 is a diagram showing a detailed configuration of a transmission circuit in embodiment 3.

[0011] Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. In the drawings, the same or corresponding parts are designated by the same reference numerals, and description thereof will not be repeated.

[0012] 1 is a schematic diagram of a communication device 1 to which an antenna module 5 according to an embodiment is applied. The communication device 1 is, for example, a mobile terminal such as a mobile phone, a smartphone, or a tablet, or a personal computer with a communication function.

[0013] 1 , the communication device 1 includes a transmission module 10, a BBIC (Baseband Integrated Circuit) 20 constituting a baseband signal processing circuit, an RFIC (Radio Frequency Integrated Circuit) 30, a power supply circuit 40, and an antenna 60. The transmission module 10 includes a power amplifier circuit 50 and a bias control circuit 55. In general, the communication device 1 upconverts an intermediate frequency (IF) signal transmitted from the BBIC 20 to a high frequency (radio frequency: RF) signal in the RFIC 30, amplifies the high frequency signal in the power amplifier circuit 50, and radiates the signal from the antenna 60. The transmission module 10 and the antenna 60 constitute an antenna module 5 of the present disclosure.

[0014] The antenna 60 is an array antenna in which a plurality of flat-plate-shaped radiating elements are arranged. In the example of FIG. 1, four radiating elements 600A to 600D, each having a substantially rectangular shape, are arranged in a line on a substrate. The shape of the radiating elements is not limited to a rectangular shape, and may be a circle, an ellipse, or another polygon. The number of radiating elements is also not limited to four. Furthermore, the radiating elements may be arranged two-dimensionally.

[0015] The RFIC 30 is an example of a signal processing circuit that processes a high-frequency signal. The RFIC 30 up-converts the intermediate frequency signal transmitted from the BBIC 20 into a high-frequency signal and outputs the generated high-frequency signal to the transmission module 10.

[0016] The bias control circuit 55 receives a control signal CON from the RFIC 30. Based on the control signal CON, the bias control circuit 55 generates a bias signal BS for controlling the magnitude and supply timing of the bias current of the amplifier included in the power amplifier circuit 50, and outputs the bias signal BS to the power amplifier circuit 50.

[0017] The power amplifier circuit 50 amplifies the input signal Pin received from the RFIC 30 and generates output signals Pout_A to Pout_D to be output to the antenna 60. The power amplifier circuit 50 includes transmission circuits 100A to 100D corresponding to the radiating elements 600A to 600D. Each transmission circuit amplifies the input signal Pin, changes the phase of the input signal Pin, and transmits the output signal to the corresponding radiating element in the antenna 60. By transmitting the input signal Pin with a predetermined phase difference relative to adjacent radiating elements, the directivity of the radio waves radiated from the array antenna can be adjusted.

[0018] In the following description, the radiating elements 600A to 600D will also be collectively referred to as "radiating elements 600," and the transmitting circuits 100A to 100D will also be collectively referred to as "transmitting circuits 100."

[0019] The power supply circuit 40 is an example of a so-called digital tracker, and can supply power supply voltages Vcc of a plurality of different voltage levels to the power amplifier circuit 50. The power supply circuit 40 includes a multilevel power converter (MPC) 410, a power supply selection circuit 420, and a digital envelope tracker (digital ET) 430.

[0020] The MPC 410 includes a plurality of DC / DC converters, which are not shown in Fig. 1. The MPC 410 converts the battery voltage VB supplied from an external battery into a plurality of different voltage levels and supplies the voltages to the power supply selection circuit 420.

[0021] The digital ET 430 receives the I and Q waveform signals of the transmit signal from the BBIC 20 and tracks the envelope of the transmit signal in the digital ET mode. The digital ET 430 generates a selection signal SEL corresponding to the voltage level of the envelope of the transmit signal and outputs it to the power supply selection circuit 420.

[0022] The power supply selection circuit 420 selects a voltage corresponding to the selection signal SEL from the plurality of voltage levels supplied from the MPC 410 and supplies it to the power amplifier circuit 50 as the power supply voltage Vcc.

[0023] (Detailed Configuration of Transmitter Circuit) Next, the detailed configuration of the transmitter circuit 100 in the power amplifier circuit 50 will be described with reference to FIG.

[0024] The transmitter circuit 100 includes a phase-shift circuit 110 and amplifier circuits CT1 and CT2. The phase-shift circuit 110 includes a phase shifter 111 and a branching circuit 112. The amplifier circuit CT1 includes phase-shift lines 113A, 145A, 171A, and 172A, a hybrid coupler 114A, an amplifier stage 120A, and an output stage 150A. Similarly, the amplifier circuit CT2 includes phase-shift lines 113B, 145B, 171B, and 172B, a hybrid coupler 114B, an amplifier stage 120B, and an output stage 150B.

[0025] The phase shifter 111 advances the input signal Pin from the RFIC 30 by a predetermined phase shift amount θ and outputs the signal to the branching circuit 112. The phase shift amount θ set by the phase shifter 111 is set to a value between 0° and 90°. The phase shift amount θ corresponds to the deflection angle of beamforming in the antenna 60, as will be described later.

[0026] The branch circuit 112 branches the signal that has passed through the phase shifter 111 into two paths, and supplies the branched signals to the amplifier circuits CT1 and CT2, respectively.

[0027] Next, the detailed configurations of the amplifier circuits CT1 and CT2 will be described. Note that since the amplifier circuits CT1 and CT2 basically have the same configuration, the configuration of the amplifier circuit CT1 will be described in detail, and as for the amplifier circuit CT2, only the correspondence between the amplifier circuit CT1 and each element and the differences from the amplifier circuit CT1 will be described.

[0028] The amplifier stage 120A in the amplifier circuit CT1 includes a carrier amplifier 141A, a peak amplifier 142A, a hybrid coupler 130A, and a switch S4 A. The output stage 150A in the amplifier circuit CT1 includes a hybrid coupler 160A and switches S1A, S2A, and S3A.

[0029] The signal supplied from the branch circuit 112 to the amplifier circuit CT1 is supplied to a hybrid coupler 114A via a phase-shift line 113A. The hybrid coupler 114A has two input terminals and two output terminals. One input terminal is connected to the phase-shift line 113A, and the other input terminal is connected to the ground potential GND. One output terminal of the hybrid coupler 114A is connected to a carrier amplifier 141A, and the other output terminal is connected to a peak amplifier 142A.

[0030] The hybrid coupler 130A includes lines 131A to 134A connected in a ring shape. An output terminal of a carrier amplifier 141A is connected to a connection node N5A between the line 131A and the line 132A. An output terminal of a peak amplifier 142A is connected to a connection node N6A between the line 131A and the line 133A. A switch S4A is connected between a connection node N7A between the line 132A and the line 134A and the ground potential GND. A connection node N8A between the line 133A and the line 134A is connected to a hybrid coupler 160A of the output stage 150A via a phase-shift line 145A.

[0031] The hybrid coupler 160A includes lines 161A to 164A connected in a ring shape. A switch S1A is connected between a connection node N1A between the lines 161A and 162A and the ground potential GND. A phase-shift line 145A is connected to a connection node N4A between the lines 161A and 163A.

[0032] A switch S2A is connected between a connection node N2A between the lines 162A and 164A and the ground potential GND. A switch S3A is connected between a connection node N3A between the lines 163A and 164A and the ground potential GND.

[0033] The radiating element 600 has feed points V1 and V2 at positions offset in opposite directions from the center of the radiating element 600 along the direction of arrow DR1 (first direction). The radiating element 600 also has feed points H1 and H2 at positions offset in opposite directions from the center of the radiating element 600 along the direction of arrow DR2 (second direction) that intersects with the first direction. In the example of FIG. 2 , the first direction and the second direction are orthogonal. In the following description, for convenience, the polarization of the radio waves radiated from feed points V1 and V2 will be referred to as "vertically polarized waves (V polarization)," and the polarization of the radio waves radiated from feed points H1 and H2 will be referred to as "horizontally polarized waves (H polarization)."

[0034] The connection node N3A in the hybrid coupler 160A is connected to the feed point V1 of the radiating element 600 via a phase shift line 171A. The connection node N2A in the hybrid coupler 160A is connected to the feed point V2 of the radiating element 600 via a phase shift line 172A.

[0035] Each of the lines 131A to 134A, 161A to 164A and the phase-shift lines 113A, 145A, 171A, and 172A has a line length of approximately 1 / 4 wavelength of the center frequency of the high-frequency signal to be transmitted by the amplifier circuit CT1. Therefore, when the high-frequency signal passes through each of these lines, the phase of the signal is delayed by approximately 90°.

[0036] 5 and 6, in the amplifier circuit CT1, the three hybrid couplers 114A, 130A, and 150A cause a phase difference of 90° between the signal supplied to the feed point V1 of the radiating element 600 and the signal supplied to the feed point V2. Furthermore, the feed point V1 and the feed point V2 are arranged at positions offset from each other on opposite sides of the center of the radiating element 600.

[0037] In this configuration, by switching the power feed position to radiating element 600 using switches S1A to S3A in output stage 150A, it is possible to change the phase of the radio waves radiated from radiating element 600 in the positive and negative directions within the range of phase shift amount θ in phase shifter 111. Therefore, by appropriately setting the phase shift amount of phase shifter 111 for each radiating element in the array antenna, it is possible to adjust the directivity (beamforming) of the V-polarized radio waves radiated from the array antenna.

[0038] In recent years, developments in wireless communication devices have been progressing toward communications in the so-called sub-terahertz frequency band, which exceeds 100 GHz, with the aim of achieving high-capacity and high-speed communications. Signals in this sub-terahertz frequency band tend to have greater losses in their signal paths than signals in millimeter waves and lower frequency bands. Therefore, when a phase shifter having a switch arranged in series in the signal path is used, the loss in the switch may prevent the desired characteristics from being achieved.

[0039] However, in the transmitting circuit of embodiment 1, the signal transmitted to radiating element 600 does not pass through switches S1A to S3A used to switch the feed position of each radiating element, and therefore no loss occurs due to switching of each switch. Therefore, even when radiating radio waves using a signal in the sub-terahertz frequency band, it is possible to suppress a decrease in the gain of the radio waves radiated from radiating element 600.

[0040] The amplifier circuit CT2 is a circuit for transmitting a high-frequency signal for radiating a horizontally polarized wave (H polarization) to the radiating element 600. The amplifier stage 120B in the amplifier circuit CT2 includes a carrier amplifier 141B, a peak amplifier 142B, a hybrid coupler 130B, and a switch S4B. The output stage 150B in the amplifier circuit CT2 includes a hybrid coupler 160B and switches S1B, S2B, and S3B. Furthermore, the hybrid coupler 130B includes lines 131B to 134B connected in a ring, and the hybrid coupler 160B includes lines 161B to 164B connected in a ring.

[0041] The hybrid couplers 114A, 130A, and 160A in the amplifier circuit CT1 correspond to the hybrid couplers 114B, 130B, and 160B in the amplifier circuit CT2, respectively. The hybrid coupler 114A and the peak amplifier 142A in the amplifier circuit CT1 correspond to the carrier amplifier 141B and the peak amplifier 142B in the amplifier circuit CT2, respectively. The phase shift lines 113A, 145A, 171A, and 172A in the amplifier circuit CT1 correspond to the phase shift lines 113B, 145B, 171B, and 172B in the amplifier circuit CT2, respectively.

[0042] The connection node N3B in the hybrid coupler 160B is connected via a phase shift line 171B to a feed point H1 of the radiating element 600. The connection node N2B in the hybrid coupler 160B is connected to a feed point H2 of the radiating element 600 via a phase shift line 172B.

[0043] As with the amplifier circuit CT1, in the beamforming of H-polarized waves by the amplifier circuit CT2, the loss caused by switching the switches can be suppressed, and a decrease in gain can be prevented.

[0044] (Signal Combining in Amplification Stages) The technique and efficiency of signal combining in the amplification stages 120A and 120B will be described using Figures 3 and 4. Note that in Figures 3 and 4, the suffixes of the reference numerals in the drawings are omitted in order to comprehensively describe both amplification stages 120A and 120B.

[0045] 3 is a diagram for explaining variations in signal synthesis in the amplifier stage 120. Signal synthesis in the amplifier stage can be switched between two modes depending on the state of the switch S4.

[0046] Specifically, when switch S4 connected to connection node N7 is in a non-conducting state (OFF), as shown in the upper part, the output signal from carrier amplifier 141 is combined with the output signal from peak amplifier 142 at connection node N8 via lines 132 and 134, and is also combined with the output signal from peak amplifier 142 at connection node N6 via line 131. In the following description, this signal combining technique will be referred to as "parallel combining."

[0047] On the other hand, when switch S1 is in a conductive state (ON), connection node N7 is shorted, as shown in the lower diagram. Therefore, due to impedance conversion by line 132 and line 134, the impedance when line 132 is viewed from connection node N5 and the impedance when line 134 is viewed from connection node N8 become open. As a result, the output signal from carrier amplifier 141 does not pass through lines 132 and 134, but passes through line 131 and is combined with the output signal of peak amplifier 142 at connection node N6. In the following description, this signal combining technique is referred to as "series combining." In the case of series combining, amplifier stage 120 forms a so-called Doherty amplifier.

[0048] 4 is a diagram for explaining the relationship between output power and efficiency in the case of series combining when switching amplifiers in the amplifier stage 120. In the amplifier stage 120, the amplifier to be used is switched depending on the power level of the input signal Pin.

[0049] Specifically, when the power level of the input signal Pin is greater than a predetermined power, both the carrier amplifier 141 and the peak amplifier 142 are driven (upper left in FIG. 4 ). On the other hand, when the power level of the input signal Pin is less than the predetermined power, only the carrier amplifier 141 is driven, and the peak amplifier 142 is deactivated (lower left in FIG. 4 ).

[0050] 4 shows a graph illustrating the relationship between output power and efficiency in amplifier stage 120. In this graph, the horizontal axis represents the power level of the output signal from amplifier stage 120, and the vertical axis represents the efficiency of amplifier stage 120. Note that the solid line LN1 represents the efficiency in the case of amplifier stage 120 of embodiment 1, and the dashed line LN2 represents the efficiency in the case of using a class AB amplifier alone that can output the same maximum power as amplifier stage 120.

[0051] In the amplifier stage 120, the power supply voltage Vcc from the power supply circuit 40 is switched between two stages depending on the power level of the input signal Pin. In the high output region RG1, the power supply voltage Vcc is set to VA1. In the low output region RG2, the power supply voltage Vcc is set to VA2, which is lower than VA1 (VA1>VA2).

[0052] In region RG1, a large amount of power is required, so that both the carrier amplifier 141 and the peak amplifier 142 of the amplification stage 120 are driven. In region RG2, the peak amplifier 142 of the amplification stage 120 is deactivated, and only the carrier amplifier 141 is driven.

[0053] When a class AB amplifier is used, the efficiency is maximized when the output power is at its maximum, and the efficiency gradually decreases as the output power decreases (dashed line LN2). On the other hand, in the amplifier stage 120 of the first embodiment, in region RG1, the parallel operation of the carrier amplifier 141 and the peak amplifier 142 improves the efficiency compared to the class AB amplifier.

[0054] On the other hand, in region RG2, the peak amplifier 142 is stopped, which increases the load impedance of the carrier amplifier 141. This increases the efficiency of the carrier amplifier 141, making it possible to achieve a back-off amount of 6 dB (solid line LN1).

[0055] Furthermore, even when the amplifier stage 120 is configured as a parallel combination, by setting the peak amplifier 142 to a non-driven state in the low output region RG2, the number of amplifiers is halved and the load impedance is increased compared to the operating state in region RG1, so that a back-off amount of 6 dB can be obtained.

[0056] (Output stage settings and polarization direction) (1) Linear polarization Next, using Figures 5 to 8, we will explain the configuration of output stages 150A and 150B in the case of a transmitting circuit that is compatible with a so-called dual-band type antenna, which emits radio waves of two different linear polarizations from the radiating element 600.

[0057] 5 and 6 are diagrams showing the configuration of the output stage when radiating V-polarized waves. In this case, a high-frequency signal is supplied to the radiating element 600 from the amplifier circuit CT1, but not from the amplifier circuit CT2.

[0058] 5 shows a case where a high-frequency signal is supplied from amplifier circuit CT1 to feed point V1 of radiating element 600. In this case, switches S1A and S2A of output stage 150A of amplifier circuit CT1 are turned on, and switch S3A is turned off. As a result, connection nodes N1A and N2A are shorted, and the impedance when looking at line 161A from connection node N4A and the impedance when looking at line 164A from connection node N3A are open. As a result, the high-frequency signal supplied from amplifier stage 120A to output stage 150A is supplied to feed point V1 of radiating element 600 via line 163A and phase-shift line 171A (arrow AR1).

[0059] In this case, if the phase of input signal Pin is 0° and the amount of phase shift by phase shifter 111 is θ, the phase of the signal at the input end of hybrid coupler 114A is delayed by 90° due to phase shift line 113A. Then, hybrid coupler 114A inputs signals with a phase difference of 90° to carrier amplifier 141A and peak amplifier 142A, respectively. In this specification, the phase of the signal input to carrier amplifier 141A is set to 90°+θ, and the phase of the signal input to peak amplifier 142A is set to 0°+θ.

[0060] In the hybrid coupler 130A, the path length from the carrier amplifier 141A to the output terminal, the connection node N8A, is twice the path length from the peak amplifier 142A to the connection node N8A, so the phase of the signals at the connection node N8A is 270°+θ, and the signals from the two amplifiers are combined in phase at the connection node N8A.

[0061] The phase of the signal at connection node N4A, which is the input end of hybrid coupler 160A, is set to 180°+θ by phase-shift line 145A. Then, because the phase is delayed by 90° by each of line 163A and phase-shift line 171A, the phase of the signal supplied to feed point V1 becomes 0°+θ.

[0062] In this case, all of the switches S1B to S3B in the output stage 150B of the amplifier circuit CT2 are turned on, which causes the impedances of the lines 161B and 163B when viewed from the connection node N4B to be open, and therefore no high-frequency signal is supplied from the amplifier circuit CT2 to the radiating element 600.

[0063] By setting the switches of output stages 150A and 150B as described above and setting the phase shift amount θ relative to adjacent radiating elements in phase shift circuit 110 in the range of 0° to 90°, the deflection angle of the V polarization direction of the radio waves radiated from the array antenna can be adjusted in the range of 0° to 90°.

[0064] 6 shows a case where a high-frequency signal is supplied from amplifier circuit CT1 to feed point V2 of radiating element 600. In this case, switches S1A and S2A of output stage 150A of amplifier circuit CT1 are set to a non-conductive state, and switch S3A is set to a conductive state. As a result, connection node N3A is shorted, and the impedance when line 163A is viewed from connection node N4A, and the impedance when line 164A is viewed from connection node N2A, become open. As a result, the high-frequency signal supplied from amplifier stage 120A to output stage 150A is supplied to feed point V2 of radiating element 600 via lines 161A and 162A and phase-shift line 172A (arrow AR2).

[0065] In this case as well, if the phase of input signal Pin is 0° and the amount of phase shift by phase shifter 111 is θ, the phase of the signal at connection node N4A, which is the input end of hybrid coupler 160A, is 180°+θ, as in Fig. 5. In hybrid coupler 160A, the phase is delayed by 90° by each of lines 161A, 162A and phase shift line 172A, so the phase of the signal supplied to feed point V2 is -90°+θ.

[0066] In this case, too, the switches S1B to S3B in the output stage 150B of the amplifier circuit CT2 are all in a conductive state.

[0067] By setting the switches of output stages 150A and 150B as described above and setting the phase shift amount θ relative to adjacent radiating elements in phase shift circuit 110 in the range of 0° to 90°, the deflection angle of the V polarization direction of the radio waves radiated from the array antenna can be adjusted in the range of -90° to 0°.

[0068] As described above, by using the phase-shift circuit 110 to set the phase shift amount θ in the range of 0° to 90° relative to adjacent radiating elements and switching the switches S1A to S3A of the output stage 150A, the deflection angle of the radio waves in the V-polarized direction radiated from the array antenna can be switched between the −θ direction and the +θ direction.

[0069] (b) H-polarized wave Figures 7 and 8 show the configuration of the output stage when radiating H-polarized wave. In this case, a high-frequency signal is supplied to the radiating element 600 from the amplifier circuit CT2, but not from the amplifier circuit CT1. Therefore, in Figures 7 and 8, switches S1A to S3A in the output stage 150A of the amplifier circuit CT1 are all in a conductive state.

[0070] 7 shows a case where a high-frequency signal is supplied from amplifier circuit CT2 to feed point H1 of radiating element 600. In this case, switches S1B and S2B of output stage 150B of amplifier circuit CT2 are turned on, and switch S3B is turned off. As a result, connection nodes N1B and N2B are shorted, and the impedance when looking at line 161B from connection node N4B and the impedance when looking at line 164B from connection node N3B are open. As a result, the high-frequency signal supplied from amplifier stage 120B to output stage 150B is supplied to feed point H1 of radiating element 600 via line 163B and phase-shift line 171B (arrow AR3).

[0071] In this case, if the phase of input signal Pin is 0° and the amount of phase shift by phase shifter 111 is θ, the phase of the signal supplied to feed point H1 will be 0°+θ, as in the case of Fig. 5. Therefore, by setting the amount of phase shift θ relative to adjacent radiating elements in phase shift circuit 110 in the range of 0° to 90°, the deflection angle of the H polarization direction of the radio waves radiated from the array antenna can be adjusted in the range of 0° to 90°.

[0072] 8 shows a case where a high-frequency signal is supplied from amplifier circuit CT2 to feed point H2 of radiating element 600. In this case, switches S1B and S2B of output stage 150B of amplifier circuit CT2 are made non-conductive, and switch S3B is made conductive. As a result, connection node N3B is shorted, and the impedance when line 163B is viewed from connection node N4B, and the impedance when line 164B is viewed from connection node N2B, become open. As a result, the high-frequency signal supplied from amplifier stage 120B to output stage 150B is supplied to feed point H2 of radiating element 600 via lines 161B, 162B and phase-shift line 172B (arrow AR4).

[0073] In this case as well, if the phase of input signal Pin is 0° and the amount of phase shift by phase shifter 111 is θ, the phase of the signal supplied to feed point H2 will be −90°+θ, as in the case of Fig. 6. Therefore, by setting the amount of phase shift θ relative to adjacent radiating elements in phase shift circuit 110 in the range of 0° to 90°, the deflection angle in the H polarization direction of the radio waves radiated from the array antenna can be adjusted in the range of −90° to 0°.

[0074] As described above, by using the phase-shift circuit 110 to set the phase shift amount θ in the range of 0° to 90° relative to adjacent radiating elements and switching the switches S1B to S3B of the output stage 150B, the deflection angle of the radio waves in the V-polarized direction radiated from the array antenna can be switched between the −θ direction and the +θ direction.

[0075] (2) Circularly Polarized Waves Next, a case where circularly polarized radio waves are emitted from the radiating element 600 will be described.

[0076] (a) Setting the Output Stage When a flat patch antenna is used to radiate circularly polarized radio waves, high-frequency signals with a phase difference of 90° are supplied to two feed points that are offset from the center of the radiating element in directions perpendicular to each other.

[0077] 9 shows the configuration of the output stage when radiating a right-handed circularly polarized wave. In this case, a high-frequency signal is supplied to the radiating element 600 from the amplifier circuit CT1 at the feed point V1, and a high-frequency signal is supplied to the radiating element 600 from the amplifier circuit CT2 at the feed point H2.

[0078] If the phase shift amount in phase shift circuit 110 is θ, as described above, the phase of the signal supplied to feed point V1 is 0°+θ, and the phase of the signal supplied to feed point H2 is −90°+θ. In other words, high-frequency signals having a phase difference of 90° are supplied. Because the phase of the signal supplied to feed point H2 is delayed relative to the signal supplied to feed point V1, circularly polarized waves that rotate to the right (clockwise) in the radiation direction are radiated from radiating element 600. Furthermore, by setting the phase shift amount θ between 0° and 90° relative to adjacent radiating elements in phase shift circuit 110, the oscillation angle of the radio waves radiated from the array antenna can be switched between the −θ direction and the +θ direction.

[0079] 10 is a diagram showing the configuration of the output stage when radiating a counterclockwise circularly polarized wave. In this case, a high-frequency signal is supplied from amplifier circuit CT1 to feed point V2 of radiating element 600, and a high-frequency signal is supplied from amplifier circuit CT2 to feed point H1.

[0080] If the phase shift amount in phase shift circuit 110 is θ, as described above, the phase of the signal supplied to feed point V2 is −90°+θ, and the phase of the signal supplied to feed point H1 is 0°+θ. In other words, high-frequency signals with a phase difference of 90° are supplied. Because the phase of the signal supplied to feed point V2 lags behind the phase of the signal supplied to feed point H1, circularly polarized waves that rotate to the left (counterclockwise) in the radiation direction are radiated from radiating element 600. Furthermore, by setting the phase shift amount θ between 0° and 90° relative to adjacent radiating elements in phase shift circuit 110, the oscillation angle of the radio waves radiated from the array antenna can be switched between the −θ direction and the +θ direction.

[0081] (b) Robustness Against Load Fluctuations In the antenna module 5 of the first embodiment, when a circularly polarized radio wave is radiated, the antenna has the effect of improving its robustness against load fluctuations, as will be described below.

[0082] 11 to 14 show the influence of load fluctuations when the amplifier stages 120A and 120B are combined in parallel. More specifically, Fig. 11 and Fig. 12 show the influence of load fluctuations when the load impedance R ＡＮＴ is the characteristic impedance R Ｌ When it becomes larger than (R Ｌ <R ＡＮＴ 11 illustrates the output state of each amplifier in the power amplifier circuit shown in FIG. 11. FIG. 12 illustrates the case of back-off in which the peak amplifier is not driven.

[0083] 13 and 14 show the load impedance R ＡＮＴ is the characteristic impedance R Ｌ When it becomes smaller than (R ＡＮＴ <R Ｌ 13 shows the output state of each amplifier in the case of maximum power output, and FIG. 14 shows the case of back-off.

[0084] 11 to 14, as well as FIGS. 15 to 18, which will be described later, the direction of load impedance fluctuation at each point in the amplifier circuits CT1 and CT2 is indicated on the left side by "Hi" or "Lo." The upper right-hand column shows the fluctuation in the output power of each amplifier in the amplifier circuit CT1, and the lower right-hand column shows the fluctuation in the output power of each amplifier in the amplifier circuit CT2. In the following explanation, the case of radiating right-handed circularly polarized waves shown in FIG. 9 will be used as an example.

[0085] Referring to FIG. 11, the load impedance R of the radiating element 600 ＡＮＴ is the characteristic impedance R Ｌ When the load impedance at the connection node N3A becomes Hi by being larger than the load impedance at the connection node N4A, the load impedance at the connection node N4A becomes Lo by the phase shift line 171A in the amplifier circuit CT1. Then, the load impedance at the connection node N4A becomes Hi by the phase shift line 163A, and the load impedance at the output end (connection node N8A) of the hybrid coupler 130A becomes Lo by the phase shift line 145A.

[0086] The signal path from connection node N8A to carrier amplifier 141A passes through two lines (line 132A + line 134A, or line 131A + line 133A), so the load impedance as seen from carrier amplifier 141A is Lo.

[0087] On the other hand, the signal path from the connection node N8A to the peak amplifier 142A passes through only one line (line 133A), and therefore the load impedance seen from the peak amplifier 142A is Hi.

[0088] In each amplifier, an increase in load impedance reduces the output power, and a decrease in load impedance increases the output power. Therefore, in the amplifier circuit CT1, as shown in the graph in the upper right diagram, when the phase is between 0° and 90° and between 270° and 360°, a decrease in the load increases the output power of the carrier amplifier 141A (solid line LN10). On the other hand, an increase in the load decreases the output power of the peak amplifier 142A (dashed line LN11). As a result, the power fluctuations are offset by the increase in output power of the carrier amplifier 141A and the decrease in output power of the peak amplifier 142A, so the combined output power of the amplifier circuit CT1 exhibits a flat characteristic, as shown by the solid line LN12.

[0089] In the amplifier circuit CT2, the load impedance at the connection node N2B becomes Lo due to the phase shift line 172B. In the hybrid coupler 160B, the load impedance at the connection node N1B becomes Hi due to the line 162B, and the load impedance at the connection node N4B becomes Lo due to the line 161B. Furthermore, the load impedance at the output end (connection node N8B) of the hybrid coupler 130B becomes Hi due to the phase shift line 145B.

[0090] The signal path from connection node N8B to carrier amplifier 141B passes through two lines (line 132B+line 134B, or line 131B+line 133B), so the load impedance as seen from carrier amplifier 141B is Hi.

[0091] On the other hand, the signal path from the connection node N8B to the peak amplifier 142B passes through only one line (line 133B), and therefore the load impedance seen from the peak amplifier 142B is Lo.

[0092] Therefore, in the amplifier circuit CT2, as shown in the graph in the lower right diagram, an increase in the load reduces the output power of the carrier amplifier 141B (solid line LN15), and a decrease in the load increases the output power of the peak amplifier 142B (dashed line LN16). As a result, power fluctuations are offset by the decrease in output power of the carrier amplifier 141B and the increase in output power of the peak amplifier 142B, and the output power combined in the amplifier circuit CT2 exhibits flat characteristics as shown by the solid line LN17.

[0093] That is, as a result, fluctuations in output power due to load impedance fluctuations are canceled out in each of the amplifier circuits CT1 and CT2, and fluctuations in output power of the entire transmitter circuit are also suppressed, thereby improving robustness against antenna load fluctuations.

[0094] 12 , during backoff, the peak amplifier 142A in the amplifier circuit CT1 is deactivated, and the peak amplifier 142B in the amplifier circuit CT2 is deactivated. In this case, the load impedance of the amplifier circuit CT1 as seen from the carrier amplifier 141A becomes low, so the output power from the amplifier circuit CT1 increases (solid line LN10). On the other hand, the load impedance of the amplifier circuit CT2 as seen from the carrier amplifier 141B becomes high, so the output power from the amplifier circuit CT2 decreases (solid line LN15).

[0095] In this way, during backoff, the output power of each amplifier circuit increases or decreases in response to load fluctuations of the radiating element 600, but because the direction of fluctuation in the output power of amplifier circuit CT1 and the direction of fluctuation in the output power of amplifier circuit CT2 are opposite to each other, fluctuations in the output power of each transmitting circuit as a whole are suppressed. Therefore, even during backoff, robustness to antenna load fluctuations can be improved.

[0096] Next, referring to FIG. 13, the load impedance R of the radiating element 600 at the time of maximum power output is ＡＮＴ is the characteristic impedance R ＬWhen the load impedance of the amplifier circuit CT1 is smaller than the load impedance of the carrier amplifier 141A and becomes Lo, the load impedance of the amplifier circuit CT1 when viewed from the carrier amplifier 141A becomes Hi and the load impedance of the amplifier circuit CT2 when viewed from the peak amplifier 142A becomes Lo.

[0097] As a result, the output power of the carrier amplifier and the peak amplifier fluctuate in opposite directions in each amplifier circuit, as in the case of a phase of 90° to 270° in the right diagram of Figure 13. This cancels out the fluctuations in output power in each amplifier, thereby suppressing fluctuations in the entire transmitter circuit. This improves robustness in response to antenna load fluctuations.

[0098] 14 , during back-off, the peak amplifier 142A in the amplifier circuit CT1 is deactivated, and the peak amplifier 142B in the amplifier circuit CT2 is deactivated. In this case, the load impedance of the amplifier circuit CT1 as seen from the carrier amplifier 141A becomes high, so the output power from the amplifier circuit CT1 decreases. On the other hand, the load impedance of the amplifier circuit CT2 as seen from the carrier amplifier 141B becomes low, so the output power from the amplifier circuit CT2 increases.

[0099] Therefore, even during back-off, if a load fluctuation occurs in the antenna, the fluctuation in output power of the entire transmitting circuit is suppressed, thereby improving robustness against the load fluctuation in the antenna.

[0100] 15 to 18 show the influence of load fluctuations when the amplifier stages 120A and 120B are series-combined. More specifically, Fig. 15 and Fig. 16 show the influence of load fluctuations when the load impedance R ＡＮＴ is the characteristic impedance R Ｌ When it becomes larger than (R Ｌ <R ＡＮＴ15 shows the output state of each amplifier in the case of maximum power output, and FIG. 16 shows the case of back-off.

[0101] 17 and 18 show the load impedance R ＡＮＴ is the characteristic impedance R Ｌ When it becomes smaller than (R ＡＮＴ <R Ｌ 17 shows the output state of each amplifier in the case of maximum power output, and FIG. 18 shows the case of back-off.

[0102] In the case of series synthesis, as in the case of parallel synthesis, the load impedance R ＡＮＴ is the characteristic impedance R Ｌ When the load impedance of the carrier amplifier 141B in the amplifier circuit CT1 becomes low, the load impedance of the peak amplifier 142A becomes high. On the other hand, when the load impedance of the carrier amplifier 141B in the amplifier circuit CT2 becomes high, the load impedance of the peak amplifier 142A becomes low.

[0103] Therefore, at maximum power in Fig. 15, the output power fluctuations of the carrier amplifier and the peak amplifier cancel each other out in each amplifier circuit. Also, at back-off in Fig. 16, the increase in output power in the carrier amplifier 141A of the amplifier circuit CT1 and the decrease in output power in the peak amplifier 142A cancel out the output power fluctuations in the entire transmitter circuit.

[0104] Furthermore, the load impedance R of the radiating element 600 ＡＮＴ is the characteristic impedance R Ｌ When it becomes smaller than (R ＡＮＴ <R Ｌ17 ), the load impedance of the carrier amplifier 141A in the amplifier circuit CT1 becomes Hi, and the load impedance of the peak amplifier 142A becomes Lo. On the other hand, in the amplifier circuit CT2, the load impedance of the carrier amplifier 141B becomes Lo, and the load impedance of the peak amplifier 142A becomes Hi. As a result, fluctuations in output power are canceled out across the entire transmission circuit at the maximum power in FIG. 17 and during back-off in FIG.

[0105] As described above, in the antenna module 5 of embodiment 1, when radiating circularly polarized radio waves, even if the antenna load fluctuates, the fluctuations in output power in each amplifier circuit are canceled out at maximum power, and the fluctuations in output power between amplifier circuits are canceled out at back-off. Therefore, it is possible to improve the robustness of the antenna against load fluctuations.

[0106] The "feed point V1," "feed point V2," "feed point H1," and "feed point H2" in the first embodiment correspond to the "first feed point," "second feed point," "third feed point," and "fourth feed point," respectively, in the present disclosure. The "amplifier circuit CT1" and "amplifier circuit CT2" in the first embodiment correspond to the "first amplifier circuit" and "second amplifier circuit," respectively, in the present disclosure. The "hybrid coupler 160A" and "hybrid coupler 160B" in the first embodiment correspond to the "first hybrid coupler" in the present disclosure. The "hybrid coupler 130A" and "hybrid coupler 130B" in the first embodiment correspond to the "second hybrid coupler" in the present disclosure.

[0107] Each of "line 161A" and "line 161B" in the first embodiment corresponds to a "first line" in the present disclosure. Each of "line 162A" and "line 162B" in the first embodiment corresponds to a "second line" in the present disclosure. Each of "line 163A" and "line 163B" in the first embodiment corresponds to a "third line" in the present disclosure. Each of "line 164A" and "line 164B" in the first embodiment corresponds to a "fourth line" in the present disclosure.

[0108] Each of "line 131A" and "line 131B" in the first embodiment corresponds to the "fifth line" in the present disclosure. Each of "line 132A" and "line 132B" in the first embodiment corresponds to the "sixth line" in the present disclosure. Each of "line 133A" and "line 133B" in the first embodiment corresponds to the "seventh line" in the present disclosure. Each of "line 134A" and "line 134B" in the first embodiment corresponds to the "eighth line" in the present disclosure.

[0109] Each of the "switch S1A" and "switch S1B" in the first embodiment corresponds to a "first switch" in the present disclosure. Each of the "switch S2A" and "switch S2B" in the first embodiment corresponds to a "second switch" in the present disclosure. Each of the "switch S3A" and "switch S3B" in the first embodiment corresponds to a "third switch" in the present disclosure. Each of the "switch S4A" and "switch S4B" in the first embodiment corresponds to a "fourth switch" in the present disclosure.

[0110] Each of the "phase shift line 171A" and the "phase shift line 171B" in the first embodiment corresponds to a "first phase shift line" in the present disclosure. Each of the "phase shift line 172A" and the "phase shift line 172B" in the first embodiment corresponds to a "second phase shift line" in the present disclosure. Each of the "phase shift line 145A" and the "phase shift line 145B" in the first embodiment corresponds to a "third phase shift line" in the present disclosure.

[0111] Each of the "connection node N1A" and "connection node N1B" in the first embodiment corresponds to a "first connection node" in the present disclosure. Each of the "connection node N2A" and "connection node N2B" in the first embodiment corresponds to a "second connection node" in the present disclosure. Each of the "connection node N3A" and "connection node N3B" in the first embodiment corresponds to a "third connection node" in the present disclosure. Each of the "connection node N4A" and "connection node N4B" in the first embodiment corresponds to a "fourth connection node" in the present disclosure. Each of the "connection node N5A" and "connection node N5B" in the first embodiment corresponds to a "fifth connection node" in the present disclosure. Each of the "connection node N6A" and "connection node N6B" in the first embodiment corresponds to a "sixth connection node" in the present disclosure. Each of the "connection node N7A" and "connection node N7B" in the first embodiment corresponds to a "seventh connection node" in the present disclosure. Each of the "connection node N8A" and the "connection node N8B" in the first embodiment corresponds to the "eighth connection node" in the present disclosure.

[0112] Second Embodiment In the first embodiment, a transmission circuit compatible with a dual-polarized or circularly polarized radiating element has been described.

[0113] In the second embodiment, a single-polarized wave type transmission circuit configured to radiate radio waves from a radiating element in a single polarization direction will be described.

[0114] 19 is a diagram showing the configuration of a transmission circuit 200 according to the second embodiment. The transmission circuit 200 has a configuration similar to that of the transmission circuit 100 according to the first embodiment, except that the amplifier circuit CT2 and the branch circuit 112 are removed. The transmission circuit 200 transmits a high-frequency signal to feed points V1 and V2, which are located at positions offset in opposite directions from the center of the radiating element 600 along the direction of arrow DR1 from the center of the radiating element 600. The method for switching the power supply to the feed points V1 and V2 is the same as that described in the first embodiment.

[0115] In the transmitting circuit 200 of the second embodiment, by switching the power feeding position to the radiating element 600 using the switches S1A to S3A in the output stage 150A, it is possible to change the phase of the radio waves radiated from the radiating element 600 in the positive and negative directions within the range of the phase shift amount θ in the phase shifter 111. Therefore, in the array antenna, by appropriately setting the phase shift amount θ of the phase shifter 111 for each radiating element, it is possible to adjust the directivity of the radio waves radiated from the array antenna.

[0116] Furthermore, since the signal transmitted to radiating element 600 does not pass through switches S1A to S3A used to switch the power supply position of each radiating element, it is possible to suppress a decrease in the gain of the radio waves radiated from radiating element 600 even when radio waves are radiated using a signal in the sub-terahertz frequency band.

[0117] Third Embodiment In the first and second embodiments, the amplifier stage includes two amplifiers, a carrier amplifier and a peak amplifier. However, the amplifier stage does not have to include a plurality of amplifiers.

[0118] 20 is a diagram showing the configuration of a transmission circuit 200A according to embodiment 3. The transmission circuit 200A is a single-polarized transmission circuit similar to the transmission circuit 200 according to embodiment 2, and includes one amplifier circuit CT1A.

[0119] The amplifier circuit CT1A is provided with only a carrier amplifier 141A as an amplification stage. In this case, it is not possible to improve efficiency by switching the number of amplifiers, but it is possible to perform beamforming by switching the power supply position by switching the switches S1A to S3A in the output stage 150A. Therefore, even when radiating radio waves using a signal in the sub-terahertz frequency band, it is possible to suppress a decrease in the gain of the radio waves radiated from the radiating element 600.

[0120] In the configuration of the third embodiment, the "carrier amplifier 141A" corresponds to the "amplification stage" of the present disclosure.

[0121] The embodiments disclosed herein should be considered to be illustrative in all respects and not restrictive. The scope of the present invention is defined by the claims, not by the description of the above embodiments, and is intended to include all modifications within the meaning and scope of the claims.

[0122] 1 Communication device, 5 Antenna module, 10 Transmission module, 20 BBIC, 30 RFIC, 40 Power supply circuit, 50 Power amplifier circuit, 55 Bias control circuit, 60 Antenna, 100, 100A to 100D, 200, 200A Transmission circuit, 110 Phase shift circuit, 111 Phase shifter, 112 Branch circuit, 113, 113A, 113B, 145, 145A, 145B, 171A, 172A, 171B, 172B Phase shift line, 114, 114A, 114B, 130A, 130B, 160A, 160B Hybrid coupler, 120, 120A, 120B Amplification stage, 131 to 134, 131A to 134A, 131B to 134B, 161A to 164A, 161B to 164B; lines, 141, 141A, 141B; carrier amplifier, 142, 142A, 142B; peak amplifier, 150A, 150B; output stage, 410; MPC, 420; power supply selection circuit, 430; digital ET, 600, 600A to 600D; radiating element, BS; bias signal, CON; control signal, CT1, CT2, CT1A; amplifier circuit, GND; ground potential, H1, H2, V1, V2; feeding point, N1A to N8A, N1B to N8B, N5 to N8; connection node, Pin; input signal, Pout_A to Pout_D Output signal, S4, S1A to S4A, S1B to S4B switches, VB battery voltage, Vcc power supply voltage.

Claims

1. An antenna module for radiating radio waves, comprising: a plurality of radiating elements each having a flat plate shape; and a plurality of transmitting circuits respectively connected to the plurality of radiating elements, wherein each of the plurality of radiating elements has a first feed point and a second feed point disposed at positions offset in opposite directions from the center of the radiating element along a first direction, and each of the plurality of transmitting circuits includes: a phase shift circuit configured to be able to change the phase of a high frequency signal; and an amplifier circuit that amplifies the signal from the phase shift circuit and transmits it to the first feed point and the second feed point of the corresponding radiating element, wherein the amplifier circuit includes: an amplifier stage that amplifies the signal from the phase shift circuit; a first hybrid coupler including first, second, third and fourth lines connected in a ring; a first switch connected between a first connection node between the first line and the second line and ground potential; and a second switch connected between a second connection node between the second line and the fourth line and ground potential. an antenna module including: a third connection node between the third line and the fourth line and a third switch connected between the third connection node and the ground potential; a first phase shift line connected between the third connection node and the first feed point; and a second phase shift line connected between the second connection node and the second feed point, wherein a signal from the amplification stage is supplied to a fourth connection node between the first line and the third line.

2. The antenna module according to claim 1, wherein when a high frequency signal is supplied to the first feed point, the first switch and the second switch are brought into a conductive state and the third switch is brought into a non-conductive state, and when a high frequency signal is supplied to the second feed point, the first switch and the second switch are brought into a non-conductive state and the third switch is brought into a conductive state.

3. The antenna module according to claim 1 or 2, wherein the amplification stage includes: a carrier amplifier; a peak amplifier; a second hybrid coupler including fifth, sixth, seventh, and eighth lines connected in a ring; and a fourth switch; an output terminal of the carrier amplifier is connected to a fifth connection node between the fifth and sixth lines; an output terminal of the peak amplifier is connected to a sixth connection node between the fifth and seventh lines; the fourth switch is connected between a seventh connection node between the sixth and eighth lines and the ground potential; the amplification circuit further includes a third phase-shift line connected between an eighth connection node between the seventh and eighth lines and the fourth connection node; and the carrier amplifier is supplied with a signal whose phase is 90° ahead of the high-frequency signal supplied to the peak amplifier.

4. The antenna module according to claim 3, wherein when the power of the high frequency signal from the phase shift circuit is greater than a predetermined power, the carrier amplifier and the peak amplifier are driven, and when the power of the high frequency signal from the phase shift circuit is less than the predetermined power, the carrier amplifier is driven and the peak amplifier is deactivated.

5. An antenna module for radiating radio waves, comprising: a plurality of radiating elements each having a flat plate shape; and a plurality of transmitting circuits respectively connected to the plurality of radiating elements, wherein each of the plurality of radiating elements has a first feed point and a second feed point disposed at positions offset in opposite directions from the center of the radiating element along a first direction, and a third feed point and a fourth feed point disposed at positions offset in opposite directions from the center of the radiating element along a second direction intersecting the first direction, wherein each of the plurality of transmitting circuits includes: a phase shift circuit configured to be able to change the phase of a high-frequency signal; a branch circuit that branches a signal from the phase shift circuit into two paths; and first and second amplifier circuits that amplify the signal from the branch circuit and transmit it to the corresponding radiating element, wherein each of the first and second amplifier circuits includes: an amplifier stage that amplifies the signal from the phase shift circuit; and a first hybrid coupler including a first line, a second line, a third line, and a fourth line connected in a ring shape. the amplifier circuit includes a first switch connected between a first connection node between the first line and the second line and a ground potential, a second switch connected between a second connection node between the second line and the fourth line and the ground potential, a third switch connected between a third connection node between the third line and the fourth line and the ground potential, a first phase shift line having one end connected to the third connection node, and a second phase shift line having one end connected to the second connection node, wherein in each amplifier circuit, a signal from the amplifier stage is supplied to a fourth connection node between the first line and the third line, the other end of the first phase shift line of the first amplifier circuit is connected to the first feed point, the other end of the second phase shift line of the first amplifier circuit is connected to the second feed point, and the other end of the first phase shift line of the second amplifier circuit is connected to the third feed point, The other end of the second phase-shift line of the second amplifier circuit is connected to the fourth feeding point.

6. The antenna module according to claim 5, wherein the first direction and the second direction are orthogonal to each other, and the phase difference between the signal supplied from the first amplifier circuit to the corresponding radiating element and the signal supplied from the second amplifier circuit to that radiating element is 90°.

7. The antenna module according to claim 5, wherein when radiating radio waves polarized in the first direction, the first amplifier circuit is driven and the second amplifier circuit is deactivated, and when radiating radio waves polarized in the second direction, the first amplifier circuit is deactivated and the second amplifier circuit is driven.

8. The antenna module according to any one of claims 5 to 7, wherein the amplification stage in each amplification circuit includes: a carrier amplifier; a peak amplifier; a second hybrid coupler including fifth, sixth, seventh, and eighth lines connected in a ring; and a fourth switch; and in each amplification circuit, an output terminal of the carrier amplifier is connected to a fifth connection node between the fifth line and the sixth line; an output terminal of the peak amplifier is connected to a sixth connection node between the fifth line and the seventh line; and the fourth switch is connected between a seventh connection node between the sixth line and the eighth line and the ground potential; and each amplification circuit further includes a third phase-shift line connected between an eighth connection node between the seventh line and the eighth line and the fourth connection node; and in each amplification circuit, a signal whose phase is 90° ahead of the high-frequency signal supplied to the peak amplifier is supplied to the carrier amplifier.

9. The antenna module according to claim 8, wherein the amplifier stage in each amplifier circuit operates as a Doherty amplifier by bringing the fourth switch into a conductive state.

10. A communication device comprising: an antenna module according to any one of claims 1 to 9; and a signal processing circuit for processing a high-frequency signal supplied to the antenna module.

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