Method for processing audio signals

Abstract

The invention relates to a method (100) for processing audio signals, comprising the following method steps: providing (110) an audio signal x(ռ); providing (120) a filter function; applying (130) the filter function to the audio signal x(ռ) in the time domain so as to provide a filtered audio signal x(ռ); and outputting (140) the filtered audio signal x(ռ); wherein the filter function is provided on the basis of K individual filter functions, wherein K ≥ 2; and wherein the individual filter functions comprise at least two high-pass filter functions or at least two low-pass filter functions from which a bandpass filter function is determined.

Description

[0001] March 14, 2025 Method for processing audio signals The present invention relates to a method for processing audio signals and a corresponding device. Various methods for audio signal processing are known from the prior art for different application scenarios. For example, DE 10 2022111300 A1 and US Pat. No. 6,141,672 A describe methods for (audio) signal processing. The methods known from the prior art are used in particular for suppressing background noise in the field of communications technology and in hearing aids. For this purpose, unfiltered audio signals are often recorded and then converted to the frequency domain before the audio signal is filtered in the frequency domain.To convert the recorded time signal into the frequency domain, the short-time Fourier transform (STFT) is typically used. Signal processing in the frequency domain has the advantage that specific frequencies can be selectively suppressed or amplified. For example, an audio signal can be attenuated in the frequency ranges where noise is expected, while the signal in the frequency ranges where speech information is expected can be amplified or at least left unattenuated. A disadvantage of audio signal processing in the frequency domain is that audio data must be processed frame by frame, resulting in higher latencies.In addition, the audio signal (recorded as a time-dependent signal) must first be transformed into the frequency domain, filtered in the frequency domain, and then transformed back into the time domain, which increases the computational effort. The resulting latency can lead to an unnatural perception or even disorientation of a user, as the perceived acoustic signals are delayed compared to the visual perception (for example, in the case of a March 14, 2025 conversation between two people, one of whom wears a hearing aid). The superposition of passive and delayed, actively reproduced sound can create comb filter effects that distort the frequency spectrum. If the processing latency is also too high, the system cannot react quickly enough to impulse noises, for example, to protect hearing from damage.Many of the available audio codecs have dedicated processors for recording, processing, and playing back audio signals with low latency. Signal conversion and processing are often performed per sampling point, so that the latency can be in the low microsecond range at correspondingly high sampling rates. However, such processors generally have only rudimentary components and limited resources from which to assemble an audio signal processing topology. Therefore, for more complex audio applications, such as noise suppression, the available methods typically rely on proven frameworks based on the aforementioned short-time Fourier transform. An alternative to this is filter bank approaches. The filter banks are designed, for example, based on linear-phase FIR filters.However, the approaches known from the prior art lead to a significant latency, which can often amount to several milliseconds, which is usually undesirable in practice. Based on the disadvantages of the methods known from the prior art described above, the object of the present invention is to provide a method for processing audio signals that enables efficient audio signal processing with low latency. To achieve the above-mentioned object, the present invention proposes a method for processing audio signals that comprises the following method steps: - Providing an audio signal ^(^); - Providing a filter function; 14.March 2025 - Applying the filter function to the audio signal ^(^) in the time domain to provide a filtered audio signal ^^(^); and - Outputting the filtered audio signal ^^(^); wherein - the provision of the filter function is dependent on ^ individual filter functions, where ^ ≥ 2; and - wherein the individual filter functions comprise at least two high-pass filter functions or at least two low-pass filter functions, from which a bandpass filter function is determined. Unlike most methods known from the prior art, the method according to the invention filtering the audio signal is performed in the time domain (or in the time domain). Furthermore, the method according to the invention enables efficient provision of the filter function using at least two high-pass filters or at least two low-pass filters.As will be explained below, different implementation variants can be used within the scope of the present invention to provide the filter function, depending on the individual filter functions. Even if the present invention can be described in its general form using two individual filter functions, it is clear that more than two individual filter functions can also be used. In particular, it can be provided that four, eight, or 16 individual filter functions (high-pass filters or low-pass filters) are used to provide several bandpass filter functions. Based on the bandpass filter functions, the filter function (also referred to as the overall filter function) can then be calculated.As explained below, the individual bandpass filter functions can be used to provide a bandpass-filtered audio signal, which is then multiplied by individual weighting functions, depending on the specific application scenario (for example, to suppress specific noises, such as street noise, cafeteria noise, or channel noise). In the method according to the invention, it can preferably be provided that the high-pass filter functions are provided online, i.e., in real time. This is made possible by the particularly efficient implementation of the method according to the invention and, in particular, by the computationally efficient provision of the bandpass filter function from at least two high-pass filter functions or at least two low-pass filter functions.Preferably, the method according to the invention can provide for the filter function to be provided by subtracting a first individual filter function from a second individual filter function, wherein the first individual filter function and the second individual filter function are each embodied as a high-pass filter function or a low-pass filter function. The individual filter function can be subtracted either directly or implicitly by introducing modified weighting factors, as will be explained in more detail below. As a result, a bandpass filter function can be provided with relatively little computational effort by subtracting two high-pass filter functions or two low-pass filter functions from one another.In some embodiments of the method according to the invention, it can be provided that two high-pass filters (in particular two adjacent high-pass filters) are used, the high-pass filter transfer function ^^(^) of which is designed such that its magnitude response above a first cut-off frequency ^. ^,^ a deviation of a maximum of 10% from each other, preferably a deviation of a maximum of 5% and particularly preferably a deviation of a maximum of 3% or a maximum of 1%, or- two low-pass filters (in particular two adjacent low-pass filters) are used, whose low-pass filter transfer function ^^(^) is designed such that their magnitude response below a second cut-off frequency ^ ^,^a deviation of a maximum of 10% from each other, preferably a deviation of a maximum of 5%, and particularly preferably a deviation of 3% or a maximum of 1%. March 14, 2025 In other words, two high-pass filters can be used that are designed such that their magnitude responses approach each other at high frequencies, or two low-pass filters can be used that are designed such that their magnitude responses approach each other at low frequencies. The first cutoff frequency and the second cutoff frequency are also referred to as stopband frequencies in the context of the present invention. The deviation of the magnitude responses (in percent) for two high-pass filters or two low-pass filters can be defined as follows: where ^^^^^^^^^^ is the deviation of the magnitude responses, |^^| is the magnitude response of a first individual filter (high-pass filter or low-pass filter), and |^^| is the magnitude response of a second individual filter (high-pass filter or low-pass filter) above or below the corresponding stopband frequency. The deviation ^ ^^^^^^^^^The magnitude responses above or below the corresponding stopband frequency are always lower than the aforementioned 10%, 5%, 3%, or 1% in the preferred embodiments. By using individual filter functions whose magnitude responses approach one another at least on one side, destructive interference is achieved, whereby the bandpass filters can be provided in an efficient manner. This will be explained in more detail below in connection with the figures. The high-pass filter function and the low-pass filter function are generally specified in the z-range. In some embodiments of the method according to the invention, it can be provided that two high-pass filters are used, whose high-pass filter transfer function is each designed such that their phase response above a first cutoff frequency deviates by a maximum of 10% from one another.March 2025, preferably has a deviation of maximum 5% and particularly preferably a deviation of maximum 3% or maximum 1%, or- two low-pass filters are used whose low-pass filter transfer function^^(^) is designed such that their phase response below a second cut-off frequency ^. ^,^ a deviation of a maximum of 10% from each other, preferably a deviation of a maximum of 5%, and particularly preferably a deviation of 3% or a maximum of 1%. In other words, two high-pass filters can be used, which are designed such that their phase responses approach each other at high frequencies, or two low-pass filters can be used, which are designed such that their phase responses approach each other at low frequencies. The deviation of the phase responses (in percent) for two high-pass filters or two low-pass filters can be defined analogously to the deviation of the magnitude responses described above: 100,where ^^^^^^ is the deviation of the phase responses, the phase response of a first individual filter (high-pass filter or low-pass filter), and ^ ^ the phase response of a second individual filter (high-pass filter or low-pass filter) above or below the corresponding stopband frequency. The deviation ^ ^^^^^The phase responses above or below the corresponding stopband frequency are correspondingly always lower than the aforementioned 10%, 5%, 3%, or 1% in the preferred embodiments. By using individual filter functions whose phase responses approach each other at least on one side, as well as subtraction, destructive interference is achieved, whereby the desired bandpass filters can be provided. Destructive interference through a modified phase function, in particular through a phase shift of 180°, and addition is also possible and, within the meaning of the present invention, is also considered subtraction. March 14, 2025 Furthermore, the method according to the invention can provide the individual filters as finite impulse response (FIR) filters or as infinite impulse response (IIR) filters. The use of FIR filters has the advantage that they can be easily designed and implemented.In addition, FIR filters are always stable because they do not use feedback loops. The use of IIR filters has the advantage of being highly efficient. Furthermore, sharper cutoff frequencies can be achieved by using IIR filters compared to FIR filters. Preferably, the method according to the invention can provide for the individual filters to have a non-linear phase response. Initial studies have shown that a further reduction in latency can be achieved by using individual filters with a non-linear phase response. In some preferred embodiments of the method according to the invention, the filtered audio signal ^^(^) can be determined as follows: where ^^(^) denotes a weighting factor and ^^(^) is determined as follows: ^(^) ^^^(^) − ^^^^(^), 1 ≤ ^ < ^^ ^ = ^^(^), ^ = ^,where ^^(^) describes an audio signal^(^) filtered by a high-pass filter ^^, and the high-pass filters ^^ each have a cutoff frequency ^^,^ at which ^^,^ < ^^,^^^ applies. The implementation of the filter function described above on March 14, 2025 allows efficient processing of the audio signals using high-pass filters, thereby enabling low latency. According to further preferred embodiments of the method according to the invention, it can be provided that the filtered audio signal ^^(^) is determined as follows: where ^^(^) denotes a weighting factor and ^^(^) is determined as follows: where ^^(^) each describes an audio signal^(^) filtered by a low-pass filter ^^, and where the low-pass filters ^^ each have a cutoff frequency^^,^ at which ^^,^ < ^^,^^^. The implementation of the filter function described above enables efficient processing of the audio signals using low-pass filters, thereby enabling low latency. As already explained above, the subtraction of the individual filter functions can also be achieved implicitly using modified weighting factors. For this purpose, it can preferably be provided that the filtered audio signal ^^(^) is determined as follows: where ^^(^) each describes an audio signal^(^) filtered by means of a high-pass filter ^^, where the high-pass filters ^^ each have a cutoff frequency March 14, 2025^^,^ for which ^^,^ < ^^,^^^ applies, and ^^^(^) each denotes a modified gain factor which is determined as follows: . This allows a particularly efficient implementation of the filter function using high-pass filters, thereby achieving low latency. It can also preferably be provided that the filtered audio signal ^^(^) is determined as follows: where ^^(^) each describes an audio signal^(^) filtered by a low-pass filter ^^, and where the low-pass filters ^^ each have a cutoff frequency^^,^ at which ^^,^ < ^^,^^^ applies, and ^^^(^) each denotes a modified gain factor, which is determined as follows: This makes it possible to provide a particularly efficient implementation of the filter function using low-pass filters, thereby enabling low latency. In some embodiments of the method according to the invention, it can be provided that at least one individual filter function is designed as a delta function (also referred to as a Dirac function or unit impulse function). In this case, preferably ^ ≥ 3, ^ ≥ 4, ^ ≥ 8 or ^ ≥ 16 individual filter functions can be provided, of which at least one individual filter function is designed as a delta function. This corresponds to a short circuit of a high-pass filter or a low-pass filter. As a result, the overall latency caused by the high-pass filters or the low-pass filters can be reduced if necessary. At the same time, the complexity of the filter is reduced if necessary.If high-pass filters are used, the first individual filter function (^ = 1) can preferably be designed as a delta function. If low-pass filters are used, the last filter function (^ = ^) can preferably be designed as a delta function. According to some embodiments of the method according to the invention, it can be provided that - the multiplication of the weighting factors ^^(^) with the signals ^^(^) calculated from the audio signals ^^(^) filtered by means of the high-pass filters or the low-pass filters, or the multiplication of the modified weighting factors ^^^(^) with the audio signals ^^(^) filtered by means of the high-pass filters or the low-pass filters, with a first frequency. erfolgt;- the summation of the products of the weighting factors ^^(^) with the signals ^^(^) calculated from the filtered audio signals ^^(^) or the summation of the products of the modified weighting factors ^^^(^) with the audio signals filtered by means of high-pass filters or low-pass filters ^ ^ ( ^ ) with the first frequency is carried out; and- the calculation of the weighting factors ^^(^) or the modified weighting factors ^^^(^) is carried out with a second frequency ^^; where- the second frequency ^^ is lower than the first frequency ^^.In initial studies it was shown that the second frequency (^ ^ ) or frequency with which the weighting factors or the modified weighting factors are calculated can be chosen to be lower than the first frequency (^ ^) or the frequency with which the multiplication of the (modified) weighting factors with the filtered audio signals ^^(^) is carried out as well as the summation of the products of the (modified) weighting factors with the filtered audio signals ^ ^ ( ^ ) , without significantly impairing the quality of the filtered audio signals. This allows for more efficient processing of the audio signals without noticeably impairing the quality of the filtered audio signals. As a result, a further reduction in latency can be achieved. For example, it can be provided that the first frequency = 192 ^^^ while the second frequency is ^^ = 16 ^^^. It can also be provided that the first frequency = 96 ^^^ and the second frequency^^ = 8 ^^^. Depending on the application scenario, the first frequency and the second frequency can be modified. Advantageously, for efficient sampling rate conversion, the ratio ^ ^ between and ^ ^ integer. The ratio can be selected according to requirements, advantageously with a value between 2 and 64, preferably with a value between 4 and 32, particularly preferably with a value between 8 and 16. In particular, as shown in the above examples, the ratio ^^ = 12 can be selected. The second frequency ^ ^preferably ≤ 48 kHz, ≤ 24 kHz or ≤ 16 kHz. Initial investigations have shown that by selecting the above-mentioned parameter values, efficient processing of the audio signals can be achieved, while at the same time ensuring good quality of the filtered audio signals. Furthermore, the method according to the invention can be provided such that the application of the filter function to the audio signal ^(^) in the time domain comprises the application of a convolution operation to the audio signal ^(^) and the impulse response ℎ(^) of a high-pass filter or a low-pass filter.Furthermore, to achieve the object described above, a device for processing audio signals is proposed, comprising: - a recording unit for recording an audio signal ^(^); - a computing unit for calculating a filter function and for processing the audio signal ^(^); wherein the computing unit is designed to apply the filter function to the audio signal ^(^) in the time domain and to calculate a filtered audio signal ^^(^); - an output unit for outputting the filtered audio signal ^^(^); characterized in that March 14, 2025 - the computing unit is designed to determine the filter function as a function of ^ individual filter functions, where ^ ≥ 2; and - to determine a bandpass filter function from at least two individual filter functions, which comprise two high-pass filter functions or two low-pass filter functions. The device according to the invention allows particularly efficient processing of audio signals with low latency.Preferably, the device according to the invention can be provided with the computing unit being designed to calculate a bandpass filter function by subtracting a first individual filter function from a second individual filter function, wherein the first individual filter function and the second individual filter function are each embodied as a filter function of a high-pass filter or a filter function of a low-pass filter. Furthermore, the device according to the invention can be provided with the properties described above in connection with the method according to the invention and with the computing unit of the device according to the invention being designed to carry out the method steps described in connection with the method. In the method according to the invention, it can be provided, in particular, that the high-pass filters or the low-pass filters used are embodied as non-linear phase filters.In the present invention, it can preferably be provided that the high-pass filters or the low-pass filters are designed as recursive, minimum-phase filters. It can also preferably be provided that the first high-pass filter (^ = 1) or the last low-pass filter (^ = ^) is replaced by a short circuit. In this way, the computational complexity can be reduced, whereby a lower latency can be achieved. March 14, 2025 Furthermore, it can preferably be provided that the weighting factors or the modified weighting factors are calculated as a function of at least one sensor signal, in particular as a function of a microphone signal. This makes it possible to react to a dynamic interference signal and the filter function can be adapted to the recorded interference signal. The present invention is explained in more detail below with reference to the figures. In this case, Fig.1 shows a flow diagram for an embodiment of the present invention, Fig. 2 shows a signal flow diagram for an embodiment of the present invention, Fig. 3 shows an exemplary magnitude response of high-pass filters according to an embodiment of the present invention, Fig. 4 shows an exemplary phase response of high-pass filters according to an embodiment of the present invention, Fig. 5 shows an exemplary magnitude response of resulting band-pass filters according to an embodiment of the present invention, Fig. 6 shows a signal flow diagram for an embodiment of the present invention based on low-pass filters, Fig. 7 shows a signal flow diagram for an embodiment of the present invention with modified weighting factors, Fig. 8 shows a signal flow diagram for an embodiment of the present invention, wherein the first high-pass filter is replaced by a short circuit, Fig.9 shows a schematic representation of an embodiment of the device according to the invention, Fig. 10 shows a schematic representation of a further embodiment of the device according to the invention comprising a plurality of sensors, preprocessing units and a loudspeaker, and March 14, 2025 Fig. 11 shows a schematic representation of a further embodiment of the device according to the invention in the form of in-ear headphones comprising a plurality of sensors, a loudspeaker and a computing unit for calculating and applying the filter function. Fig. 1 shows a flowchart for an embodiment of the method 100 according to the invention. In a first method step 110, an audio signal is provided. This can be done using a microphone or another sensor. Alternatively, it can be provided that the audio signal is provided by an external source that has previously recorded the audio signal.In a second method step 120, a filter function is provided. The filter function is provided as a function of ^ individual filter functions, where ^ ≥ 2. The individual filter functions can be implemented as high-pass filter functions or low-pass filter functions. A bandpass filter function is calculated from the individual filter functions. This can be done in particular by subtracting two high-pass filter functions or two low-pass filter functions. In a third method step 130, the provided filter function is applied to the audio signal in the time domain. This provides a filtered audio signal. In a fourth method step 140, the filtered audio signal is output. The filtered audio signal can be output either via a loudspeaker or another output unit, or via signal transmission to an external device. Fig.Figure 2 shows an exemplary embodiment of the method according to the invention. In this embodiment, an input signal ^(^) is transmitted to a plurality of high-pass filters 10 via a parallel circuit, wherein Figure 2 depicts a first high-pass filter 10a, a second high-pass filter 10b, and a ^-th high-pass filter 10c. ^^ with ^ ∈ [1 .. ^] has the cutoff frequencies ^^,^, where ^^,^ < ^^,^^^. The specific design of the cutoff frequencies can be configured differently for different application scenarios. The high-pass-filtered signals ^^(^) are output at the output of the high-pass filters 10. Adjacent high-pass-filtered signals are then subtracted from one another to generate bandpass signals. March 14, 2025. The bandpass signals are then weighted with time-varying weighting factors ^^(^) and then combined to form the output signal summed up. The weighting factors ^^(^) can preferably assume a value of 0 to 1. In some embodiments of the present invention, it can also be provided that the weighting factors can also assume values ​​that are greater than 1 or less than 0. The weighting factors are also calculated depending on the specific application. When calculating the weighting factors, the approaches known from the prior art can be used. In order to generate corresponding bandpass signals from the individual high-pass filters by subtraction, it is preferred that the individual high-pass filters have certain magnitude and phase properties. These preferred properties are shown as examples in Figs. 3 and 4. In particular, it can be seen in Fig. 3 that the magnitude responses 20 for the individual high-pass filters approach each other at high frequencies. In Fig.Figure 3 shows a first magnitude response 20a, a second magnitude response 20b, a third magnitude response 20c, and a fourth magnitude response 20d. As can be seen in Figure 3, the magnitude responses 20 are closely spaced above a certain frequency (in particular above an upper stopband frequency ^^,^). Furthermore, Figure 4 shows that the phase responses 20 also converge at high frequencies. Figure 4 shows a first phase response 21a, a second phase response 21b, a third phase response 21c, and a fourth phase response 21d. The bandpass transfer function can be defined as the difference between two high-pass filter transfer functions in the ^ range as follows: March 14, 2025 . Due to the high-pass characteristic of the high-pass filters, the band-pass filter also has a high-pass characteristic with a corresponding lower stopband frequency ^^,^ = ^^,^. The neighboring high-pass filters can, in particular, be designed such that their magnitude and / or phase response approaches an upper stopband frequency ^^,^, so that subtraction leads to destructive interference. In summary, the ^-th band-pass filter has a passband of ^^,^ ≤ ^ ≤ ^^,^. Due to the definition of the band-pass filters by the high-pass filters, the lower stop frequency of the following band-pass filter corresponds to the upper stop frequency of the band-pass filter, correspondingly ^^,^^^ = ^^,^. The cutoff frequencies of the high-pass filters, and accordingly the passbands of the band-pass filters, can advantageously be distributed evenly on a psychoacoustically motivated frequency scale, such as the Bark scale.This allows the input signal to be processed in frequency bands that mimic the human ear. However, the choice of cutoff frequencies depends on the specific application scenario. The present invention is not limited to a specific choice of cutoff frequencies. A key advantage of the inventive topology is that the overall transfer function for unity gain ^^ = 1 simplifies to March 14, 2025 The overall transfer function is therefore defined exclusively by the first high-pass filter. This makes the overall transfer function smooth in the passband and has a short group delay. The high-pass filters can be designed, for example, using an optimization method with a cost function based on the magnitude response of the bandpass filters in the passband and stopband, as well as the sum of all bandpass filters. The filters can be implemented as FIR or IIR filters. They can advantageously be minimum-phase, exhibiting a nonlinear phase response. A filter can be described as minimum-phase if its zeros, i.e., the zeros of the numerator polynomial of its filter transfer function, lie within the unit circle or have an amplitude ≤ 1.This definition applies to FIR filters as well as to IIR filters that are not implemented as all-pole filters, i.e., filters whose transfer function exclusively comprises denominator coefficients and, if applicable, a gain factor. All-pole filters are, by definition, minimal-phase. The high-pass filters can advantageously also be optimized so that the overall transfer function ^^(^) follows a desired magnitude and phase response, so that the filter bank, for example, implicitly performs frequency weighting or equalization. Fig. 5 shows the resulting magnitude responses 22 of the exemplary bandpass filters based on the high-pass filters shown in Figs. 3 and 4. Overall, Fig. 5 shows a first magnitude response 22a, a second magnitude response 22b, a third magnitude response 22c, and a fourth magnitude response 22d for the corresponding bandpass filters.The overall transfer function ^^(^) then corresponds to the high-pass filter with a solid line from Fig. 3 and Fig. 4. Instead of high-pass filters, low-pass filters 11 (also designated as ^^(^)) can also be used to implement the inventive method, as shown by way of example in Fig. 6. Fig. 6 shows, by way of example, a first low-pass filter 11a, a second low-pass filter 11b, and a ^-th low-pass filter 11c. The basic principle of bandpass behavior described above, based on the characteristics of the filters and on destructive interference by subtraction, still applies; however, some adjustments must be made compared to the embodiment shown in Fig. 2. In particular, the subtraction is adjusted so that. For unity gain ^^ = 1, the overall transfer function in this case simplifies to ^^(^) = ^^(^). Fig. 7 shows a further embodiment of the method according to the invention, wherein, compared to the embodiment shown in Fig. 2, the explicit subtraction of the high-pass signals ^^(^) has been removed. Instead, modified weighting factors ^^^(^) ∈ [−1, 1] are used, so that the difference is formed implicitly during weighting. For the arrangement based on high-pass filters, the modified weighting factors can be calculated, in particular, as follows: This rule can be determined by rearranging the equation for the overall transfer function, here without loss of generality in the ^-range and neglecting the time variance of the weighting factors: Fig. 8 shows a further embodiment of the method according to the invention, wherein the first high-pass filter 10a shown in Fig. 2 has been replaced by a short circuit. Accordingly, the last low-pass filter 11c in the arrangement from Fig. 6 can also be replaced by a short circuit. As a result, the structure for unity gain has a transfer function ^^(^) = 1 and therefore no inherent latency. It should be noted here that the overall transfer function for deviating weighting factors ^^ ≠ 1 is not simplified accordingly, which can increase the latency. However, even in these cases, the input-to-output latency is very low due to the filter structure. Fig. 9 shows an embodiment of the device 50 according to the invention comprising a recording unit 30 designed as a microphone, a filter bank 31, an output filter 32, and an output unit 33 designed as a loudspeaker. Fig.Figure 10 shows a further embodiment of the device 50 according to the invention, which has two sensors 30, two input filters 34, a filter bank 31, and a loudspeaker 33. These arrangements are found, for example, in modern headphones, where the microphones record ambient sound, the pre- / post-processing units (also referred to as input and output filters) filter the microphone signals so that, for example, active noise cancellation, for reducing the volume of ambient sound, or an ambient mode, for natural communication with the environment, is implemented, and a filtered audio signal is then reproduced via the loudspeaker of the headphones.The method according to the invention can be used in such an application, for example, to remove microphone or wind noise from the microphone signal, to perform speech enhancement, or to compress the microphone signal so that loud signals are reduced in level before playback. Finally, Fig. 11 shows an example of an in-ear headphone 60 equipped with several external microphones 40, an internal microphone 41, a vibration sensor 42, a loudspeaker 33, and a computing unit 43. Fig. 12 also shows an ear insert 44, an ear canal 45, and an eardrum 46. The computing unit 43 is designed to carry out the method steps according to the method according to the invention. In particular, the computing unit 43 is designed to provide the filter function and to apply it to an audio signal in the time domain.Various pre- and post-processing steps can optionally be performed on the sensor signals, such as filtering, amplification, compression, or limiting. These can be implemented according to the approaches described in the prior art. A single-channel signal based on the processing by the inventive method is then reproduced via the loudspeaker 33. Multiple instances of the inventive method can also be used to process various sensor signals, which are subsequently combined and fed to an output unit. Any sensor signals can be used to calculate the weighting factors ^^(^). For the purposes of the present invention, bandpass signals can preferably be included in the calculation of the respective weighting factors. In particular, a weighting factor ^^(^) can be a function of the bandpass signal ^^(^).The respective weighting factor can also depend on additional bandpass signals. While the filter bank, the application of the weighting factors, and the summation can be processed at a first frequency, it can preferably be provided that the weighting factors are preferably calculated at a second frequency. The second frequency can preferably be lower than the first frequency. For this purpose, either the bandpass signals ^ ^ (^), the high / low pass signals ^ ^(^), or even the input signal ^(^) can be passed to a second process via a sampling rate converter. The second process can then emulate parts of the filter bank or even the entire filter bank accordingly, so that the bandpass signals are available at the second frequency. Since the weighting factors usually change only slowly, the second sampling rate is advantageously smaller than the first sampling rate to reduce computational complexity. However, when designing the sampling rates, the Nyquist frequency should be taken into account (March 14, 2025) so that the signals are transmitted with the appropriate bandwidth and no information is lost. The weighting factors calculated at the second sampling rate do not necessarily have to be adapted to the first rate by a sampling rate conversion.The weighting factors can, for example, be calculated to allow speech components per band to pass through accordingly and attenuate all background noise, such as wind noise, ambient noise, or microphone noise, thereby improving speech quality or intelligibility. Furthermore, the weighting factors can be calculated to attenuate loud signal components per band, for example, to protect the hearing of users of an ambient mode in headphones or hearing protection with a communication function. Likewise, the inventive method can also be applied to other audio sources, such as music, telephone calls, computer games, and films. The inventive method can be used to align or personalize a music playback system, an ambient mode, and active noise cancellation in headphones.Furthermore, the method can be used, for example, in hearing aid applications to compensate for hearing loss. The method can also be used for the (also frequency-dependent) calibration of audio devices.

[0002] March 14, 2025 REFERENCE SYMBOL LIST H ochpassfilter a first high-pass filter b second high-pass filter c ^-th high-pass filter low-pass filter a first low-pass filter b second low-pass filter c ^-th low-pass filter Magnitude response of the high-pass filters a first magnitude response b second magnitude response c third magnitude response d fourth magnitude response Phase response of the high-pass filters a first phase response b second phase response c third phase response d fourth phase response Magnitude response of the determined band-pass filters a first magnitude response b second magnitude response c third magnitude response d fourth magnitude response Recording unit Filter bank Output filter Output unit Input filter Outer microphone Inner microphone Vibration sensor March 14, 2025 Computing unit Ear insert GehörgangEardrum device according to the invention In-ear headphones Method according to the invention First method step Second method step Third method step Fourth method step

Claims

March 14, 2025 CLAIMS 1. Method (100) for processing audio signals, comprising the following method steps: - providing (110) an audio signal ^(^); - providing (120) a filter function; - applying (130) the filter function to the audio signal ^(^) in the time domain to provide a filtered audio signal ^^ ( ^ ); and - outputting (140) the filtered audio signal ^^(^); wherein - the provision of the filter function is dependent on ^ individual filter functions, where ^ ≥ 2; and - the individual filter functions comprise at least two high-pass filter functions or at least two low-pass filter functions, from which a band-pass filter function is determined.

2. Method (100) according to claim 1, characterized in that the provision (120) of the filter function comprises a subtraction of a first individual filter function from a second individual filter function, wherein the first individual filter function and the second individual filter function are each implemented as a filter function of a high-pass filter or a filter function of a low-pass filter.

3. Method (100) according to claim 2, characterized in that - two high-pass filters are used, whose high-pass filter transfer function ^^(^) is designed such that their magnitude response above a first cutoff frequency ^ ^,^a deviation of maximum 10% from each other, preferably a deviation of maximum 5% and particularly preferably a deviation of maximum 3% or maximum 1%, or -two low-pass filters are used whose low-pass filter transfer function ^^(^) is designed such that their magnitude response below a second cut-off frequency ^ ^,^ a deviation of maximum 10% from each other, preferably a deviation of maximum 5% March 14, 2025 and particularly preferably has a deviation of 3% or a maximum of 1%.

4. Method (100) according to one of claims 2 or 3, characterized in that - two high-pass filters are used, whose high-pass filter transfer function ^^(^) is designed such that their phase response above a first cutoff frequency ^ ^,^a deviation of maximum 10% from each other, preferably a deviation of maximum 5% and particularly preferably a deviation of maximum 3% or maximum 1%, or - two low-pass filters are used whose low-pass filter transfer function ^^(^) is designed such that their phase response below a second cut-off frequency ^ ^,^a deviation of a maximum of 10% from each other, preferably a deviation of a maximum of 5%, and particularly preferably a deviation of 3% or a maximum of 1%.

5. Method (100) according to one of claims 1 to 4, characterized in that the individual filters are designed as finite impulse response, FIR, filters or as infinite impulse response, IIR, filters.

6. Method (100) according to one of claims 1 to 5, characterized in that the individual filters have a non-linear phase response.

7. Method (100) according to one of claims 1 to 6, characterized in that the filtered audio signal ^^(^) is determined as follows: March 14, 2025 where ^^(^) denotes a weighting factor and ^^(^) is determined as follows: ^(^) = ^^^(^) − ^^^^(^), 1 ≤ ^ < ^^ ^^(^), ^ = ^, where ^^(^) describes an audio signal ^(^) filtered by means of a high-pass filter ^^^, and the high-pass filters ^^^ each have a cutoff frequency ^^,^ at which ^^,^ < ^^,^^^ applies.

8. Method (100) according to one of claims 1 to 6, characterized in that the filtered audio signal ^^(^) is determined as follows: where ^^(^) denotes a weighting factor and ^^(^) is determined as follows: where ^^(^) each describes an audio signal ^(^) filtered by a low-pass filter ^^, wherein the low-pass filters ^^ each have a cutoff frequency^^,^ for which ^^,^ < ^^,^^^ applies.

9. Method (100) according to one of claims 1 to 6, characterized in that the filtered audio signal ^^(^) is determined as follows: where ^^(^) describes an audio signal ^(^) filtered by a high-pass filter ^^, where the high-pass filters ^^ each have a March 14, 2025 cutoff frequency ^^,^, for which ^^,^ < ^^,^^^ applies, and ^^^(^) denotes a modified gain factor which is determined as follows: ^^ (^) ^ ^^(^), ^ = 1^ = ^^(^) − ^^^^(^), 1 < ^ ≤ ^.

10. Method (100) according to one of claims 1 to 6, characterized in that the filtered audio signal ^^(^) is determined as follows: where ^^(^) describes an audio signal ^(^) filtered by a low-pass filter ^^, wherein the low-pass filters ^^ each have a cutoff frequency^^,^ at which ^^,^ < ^^,^^^ applies, and ^^^(^) each denotes a modified gain factor, which is determined as follows:

11. Method (100) according to one of claims 1 to 10, characterized in that at least one individual filter function is designed as a delta function.

12. Method (100) according to one of claims 8 to 11, characterized in that - the multiplication of the weighting factors ^^(^) with the signals ^^(^) calculated from the audio signals ^^(^) filtered by means of the high-pass filters or the low-pass filters or the multiplication of the modified weighting factors ^^^(^) with the audio signals ^^(^) filtered by means of the high-pass filters or low-pass filters with a first frequency occurs; 14 March 2025 - the summation of the products of the weighting factors ^^(^) with the signals calculated from the filtered audio signals ^^(^) or the summation of the products of the modified weighting factors ^^^(^) with the audio signals filtered by means of high-pass filters or low-pass filters ^^(^) with the first frequency and - the calculation of the weighting factors ^^(^) or the modified weighting factors ^^^(^) is carried out with a second frequency ^^; where - the second frequency ^^ is lower than the first frequency ^^.

13. Method (100) according to one of claims 1 to 12, characterized in that the application of the filter function to the audio signal ^(^) in the time domain comprises the application of a convolution operation to the audio signal ^(^) and the impulse response ℎ(^) of a high-pass filter or a low-pass filter.14.Device (50) for processing audio signals, comprising: - a recording unit (30) for recording an audio signal ^(^); - a computing unit (43) for calculating a filter function and for processing the audio signal ^(^); wherein the computing unit (43) is designed to apply the filter function to the audio signal ^(^) in the time domain and to calculate a filtered audio signal ^^(^); - an output unit (33) for outputting the filtered audio signal^^(^); characterized in that - the computing unit (43) is designed to determine the filter function as a function of ^ individual filter functions, where ^ ≥ 2; and - to determine a bandpass filter function from at least two individual filter functions, which comprise two high-pass filter functions or two low-pass filter functions.

15. Device (50) according to claim 14, characterized in that the computing unit (43) is designed to perform a bandpass filter function. March 14, 2025 Subtraction of a first individual filter function from a second individual filter function, wherein the first individual filter function and the second individual filter function are each implemented as a filter function of a high-pass filter or a filter function of a low-pass filter.

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