Methods and apparatuses for enabling suppression of spurious tones in a transceiver

Abstract

A method (1200) for enabling suppression of spurious tones in a transceiver (300), the method comprising: injecting (1210) a first test signal (X Φ1) and a second test signal (X Φ2) into the transmitter (310) with the transceiver (300) being in a first and a second loopback configurations, respectively; obtaining (1220) linear frequency characteristics (H Φ1[κ], H Φ1[−κ]) and conjugate-linear frequency characteristics (Formula I) of the transceiver (300) in the first loopback configuration and linear frequency characteristics (H Φ2[κ], H Φ2[−κ]) and conjugate-linear frequency characteristics (Formula II) of the transceiver (300) in the second loopback configuration; and estimating (1230), based on the first and second test signals (X Φ1, X Φ2) as injected into the transmitter (310) and as received from the receiver (340) and on the obtained linear and conjugate-linear frequency characteristics for the first and the second loopback configurations (Formula III) a first compensation signal (CTX[κ], CTX[-κ]) for suppression of a spurious tone during a transmission by the transmitter (310) and / or a second compensation signal (CRX[κ], CRX[-κ]) for suppression of a spurious tone during a reception by the receiver (340).

Description

[0001]METHODS AND APPARATUSES FOR ENABLING SUPPRESSION OF SPURIOUS TONES IN A TRANSCEIVER TECHNICAL FIELD The present disclosure relates generally to the field of wireless communications. More particularly, it relates to a method for enabling suppression of spurious tones in a transceiver, an apparatus performing the disclosed method, a computer program for carrying out the disclosed method, and a computer program carrier carrying said computer program. BACKGROUND Bitrate demands in wireless communications networks continue to increase. Moreover, low-frequency spectra fill up and higher-frequency spectra is taken into use. In fifth generation (5G) wireless communications networks, new frequency ranges have been introduced, such as Frequency Range 2 (FR2), which is 24350 MHz – 53600 MHz. To overcome higher path loss for higher-frequency spectra such as FR2, network nodes (such as base stations) and wireless devices (such as user equipments, UEs) may be built with an Advanced Antenna System (AAS). An AAS radio is built up using multiple transmit and receive units, where the amplitude and phase of each unit may be controlled so that antenna diagram may be optimized for a radio link between a wireless device and a network node. For transceiver functionality in wireless devices and network nodes built with AAS and operating in FR2, a trend is that the transceiver architecture is moving from heterodyne to homodyne (which is also called direct conversion). A transceiver for a wireless device or network node may be called a radio frequency (RF) transceiver. A homodyne transceiver architecture enables integration of anti-aliasing filters and has superior spurious response performance, where the latter is needed due to ever-increasing requirements on co-locate / co-existence interference levels that come from street level deployment and general densification of the deployment scenario. Due to the large signal bandwidths used in FR2 products (e.g., 1600MHz), it is power efficient to build the converters as In-phase-Quadrature (IQ) converters using first Nyquist zone. The wanted signal bandwidth and digital-to-analog converter (DAC) / analog-to-digital converter (ADC) clock rate needed creates a requirement for the converters to be integrated into the latest circuit technology / process (i.e., semiconductor technology). In contrast, RF integrated circuits (RFICs) benefit of staying in a mature circuit technology optimized for RF performance specifically. This leads to the converts (ADC / DAC) being integrated with the digital front end (DFE), which is in a different circuit technology from the RFICs. Thus, the converters are physically separated from the RFICs. Consequently, there will be an analog interface between the RFIC and the converters (DAC / ADC) that is sensitive to interference. Any spurious tone could interfere with wanted signal information in the transceiver, which is undesired. A spurious tone may, e.g., be DC offset from the DAC / ADC, local oscillator (LO) leakage, and a leakage from a reference clock (e.g., phase-locked loop, PLL, reference). As an example, the required isolation between a PLL reference clock and the analog interface could be above 100 dB. In addition, the reference clock needs to be distributed with high amplitude to fulfill the PLL phase noise requirements for high-bandwidth system, which makes it challenging to meet isolation requirements without digital compensation. The processing of the baseband signals from the DAC to a modulator of the transmitter from a demodulator of the receiver to the ADC also generates an unwanted DC offset caused by mismatch. Furthermore, LO leakage causes LO leakage in the transmitter and a DC offset in the receiver. US 7646704 B2 discloses a method for spur cancellation in an orthogonal frequency division multiplexing communication system. WO 2011 / 089184 A1 discloses an apparatus for estimating a DC offset in a transmitter. There is a need for improved ways of enabling suppression of spurious tones in a transceiver for wireless communications. In particular, there is need for improved ways of enabling suppression of spurious tones that are computationally efficient and that minimizes disruption of the transceiver operation. SUMMARY It is an object of the present disclosure to mitigate, alleviate or eliminate one or more of the above-identified deficiencies and disadvantages in the prior art and solve at least the above-mentioned problem. In particular, an object is to provide improved ways of enabling suppression of spurious tones in a transceiver for wireless communications. This object is attained at least in part by a method for enabling suppression of spurious tones in a transceiver. The transceiver comprises a transmitter having a modulator and a receiver having a demodulator. The transceiver is adapted for being put in a first and a second loopback configuration, respectively, in which respective signals injected to the transmitter are modulated by the modulator, looped back to the receiver with a difference in phase shift, and thereafter demodulated by the demodulator. The method comprises injecting a first test signal and a second test signal into the transmitter with the transceiver being in the first and the second loopback configurations, respectively. The method further comprises obtaining linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the first loopback configuration and linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the second loopback configuration. The method also comprises estimating, based on the first and second test signals as injected into the transmitter and as received from the receiver and on the obtained linear and conjugate-linear frequency characteristics for the first and the second loopback configurations, a first compensation signal for suppression of a spurious tone during a transmission by the transmitter and / or a second compensation signal for suppression of a spurious tone during a reception by the receiver. Known ways of estimating compensation signals for suppressing spurious tones often contain many steps and often take significant time. Furthermore, in some known ways of estimation compensation signals, the transceiver needs to be put in calibration mode (e.g., powering down a low noise amplifier) during the estimation such that traffic cannot be upheld during the estimation. In general, it is desired to repeat the estimation concurrently during operation of the transceiver (since spurious tones may vary with, e.g., temperature). The method disclosed herein enables estimation of compensation signals for suppressing spurious tones quickly and with relatively few calculations. Furthermore, the disclosed method can be performed with low or no disruption of traffic. In the disclosed method, compensation signals for multiple frequencies (i.e., for compensating for spurious tones at multiple frequencies) can be estimated simultaneously. Consequently, the disclosed method saves time and resources, which may further reduce the impact on traffic. The estimation of the compensation signals in the disclosed method can be based on the same type of test signals / experiments used for estimation frequency-dependent IQ (FDIQ) imbalance. Consequently, the disclosed method saves time and resources, which may further reduce the impact on traffic. In transceivers with an analog interface between RFICs and converters, the compensation of spurious tones relaxes isolation requirements between the reference clock and the analogue interface, and thereby enable integration of RF-PLL into the RFIC without extreme strict isolation requirements between the PLL reference clock (or other reference clocks in the system) and the analog interface. In other words, the disclosed method enables the use of an analog interface between RFIC and ADC / DAC, which consequently enables an architecture that integrates ADC / DAC and DFE in respective circuit technologies, which in turn leads to cost and energy savings. According to some embodiments, the method further comprises transmitting a third signal with the transmitter and / or receiving a fourth signal with the receiver, where the third and the fourth signals may be adapted for carrying traffic or user data. In that case, the method further comprises adding the first compensation signal to the third signal at the input of the transmitter to suppress a spurious tone during the transmission of the third signal, and / or adding the second compensation signal to the fourth at the output of the receiver to suppress a spurious tone during the reception of the fourth signal. In this way, an improved transceiver is provided, which in turn enables an improved wireless communications system. There is also disclosed herein an apparatus for enabling suppression of spurious tones in a transceiver. The apparatus is associated with the above-discussed advantages. The transceiver comprises a transmitter having a modulator and a receiver having a demodulator. The transceiver is adapted for being put in a first and a second loopback configuration, respectively, in which respective signals injected to the transmitter are modulated by the modulator, looped back to the receiver with a difference in phase shift, and thereafter demodulated by the demodulator. The apparatus comprises a processing circuitry configured to initiate injection of a first test signal and a second test signal into the transmitter with the transceiver being in the first and the second loopback configurations, respectively. The processing circuitry is further configured to obtain linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the first loopback configuration and linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the second loopback configuration. The processing circuitry is also configured to estimate, based on the first and second test signals as injected into the transmitter and as received from the receiver and on the obtained linear and conjugate-linear frequency characteristics for the first and the second loopback configurations, a first compensation signal for suppression of a spurious tone during a transmission by the transmitter and / or a second compensation signal for suppression of a spurious tone during a reception by the receiver. There is also disclosed herein a computer program comprising instructions which, when executed on at least one processing circuitry, cause the at least one processing circuitry to carry out the method according to the discussions above. The computer program is associated with the above-discussed advantages. There is also disclosed herein a computer program carrier carrying a computer program according to the discussion above, wherein the computer program carrier is one of an electronic signal, optical signal, radio signal, or computer- readable storage medium. The computer program carrier is associated with the above-discussed advantages. BRIEF DESCRIPTION OF THE DRAWINGS With reference to the appended drawings, below follows a more detailed description of embodiments of the present disclosure cited as examples. In the drawings: Figure 1 is a schematic illustration of a wireless communications network; Figure 2 is an example of a widely linear system represented by tuples (G, G̃) of two linear systems; Figure 3 is a schematic illustration of a transceiver; Figure 4 is a schematic illustration of a transceiver with a transmitter compensation filter at an input of a transmitter of the transceiver and a receiver compensation filter at an output of a receiver of the transceiver; Figure 5 shows a schematic of an example implementation of a compensation filter; Figure 6 shows a model of a transceiver with spurious tones and compensation for the spurious tones; Figure 7 shows a re-arrangement of the model of Figure 6; Figures 8A-8D illustrate vector representations of various test signals; Figures 9A-9B show test signals in the frequency domain; Figures 10A-10D show example implementations of loopback paths for example transceivers; Figure 11 is a schematic illustration of a transceiver; Figure 12 is a flow chart illustrating a method; and Figure 13 schematically illustrates an apparatus. DETAILED DESCRIPTION The present disclosure is described below with reference to the accompanying drawings, in which certain aspects of the present disclosure are shown. The present disclosure may, however, be embodied in many different forms and should not be construed as limited to the embodiments and aspects set forth herein; rather, these embodiments are provided by way of example so that this disclosure will be thorough and complete, and will fully convey the scope of the present disclosure to those skilled in the art. The same features are denoted by the same reference signs throughout the description. It is to be understood that the present disclosure is not limited to the embodiments described herein and illustrated in the drawings; rather, the skilled person will recognize that many changes and modifications may be made within the scope of the appended claims. Figure 1 – Wireless communications network Figure 1 depicts a wireless communications network 100 in which embodiments herein may operate. In some embodiments, the wireless communications network 100 may be a radio communications network, such as, sixth generation (6G), New Radio (NR), or NR+ telecommunications network. However, the wireless communications network 100 may also employ technology of any one of third / fourth / fifth generation, Long Term Evolution (LTE), LTE-Advanced, Wideband Code Division Multiple Access (WCDMA), Global System for Mobile Communications (GSM), Enhanced Data Rates for GSM Evolution (EDGE), Ultra Mobile Broadband (UMB), or any other similar network or system. The wireless communications network 100 may also employ technology transmitting on millimeter-waves (mmW), such as, e.g., an ultra-dense network (UDN). In some embodiments, the wireless communications network 100 may also employ transmissions supporting WiFi transmissions, e.g., the wireless communications standard IEEE 802.11ad or similar, or other non-cellular wireless transmissions. The wireless communications network 100 comprises a network node 110. The network node 110 may serve wireless devices in at least one cell 115, or coverage area. The network node 110 may correspond to any type of network node or radio network node capable of communicating with a wireless device and / or with another network node, such as, a base station (BS), a radio base station, gNB, eNB, eNodeB, a Home NodeB, a Home eNodeB, a femto BS, or a pico BS in the wireless communications network 100. Further examples of the network node 110 may be a repeater, multi-standard radio (MSR) radio node such as MSR BS, network controller, radio network controller (RNC), base station controller (BSC), relay, donor node controlling relay, base transceiver station (BTS), access point (AP), transmission points, transmission nodes, a remote radio unit (RRU), a remote radio head (RRH), nodes in distributed antenna system (DAS), or core network node. The network node 110 may be arranged to communicate with a remote data processing unit 140, e.g., via a core network 150 of the wireless communications network 100. The remote data processing unit 140 may, for example, be a remote standalone server, a cloud- implemented server, a distributed server, dedicated data processing resources in a server farm, or similar. As is also shown in Figure 1, a wireless device 121 is located within the cell 115. The wireless device 121 is configured to communicate within the wireless communications network 100 via the network node 110 over a radio link served by the network node 110. The wireless device 121 may transmit data over an air or radio interface to the network node 110 in uplink (UL), transmissions 132 and the radio base station may transmit data over an air or radio interface to the wireless device 121 in downlink (DL) transmissions 131. The wireless device 121 may refer to any type of wireless devices or user equipment (UE) communicating with a network node and / or with another wireless device in a cellular, mobile or radio communication network or system. Examples of such wireless devices are mobile phones, cellular phones, personal digital assistants (PDAs), smart phones, tablets, sensors equipped with a UE, laptop mounted equipment (LME) (e.g. universal serial bus, USB), laptop embedded equipment (LEE), machine type communication (MTC) devices, or machine to machine (M2M) device, customer premises equipment (CPE), target device, device-to-device (D2D) wireless device, wireless device capable of machine to machine (M2M) communication. As part of the developing of the embodiments described herein, it has been realized that spurious tones in a transceiver (e.g., of the wireless device 121 or the network node 110) may be estimated by using linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver. In embodiments disclosed herein, a test signal is looped through a transmitter and a receiver of the transceiver, where the received signal from the receiver (i.e., the test signal as received from the receiver) is subsequently analyzed. The test signal is injected into the transmitter such that it is modulated by a modulator of the transmitter, looped back to the receiver, and thereafter demodulated by a demodulator of the receiver. When the test signal is injected, the transceiver may, e.g., provide a signal path from the transmitter to the receiver by means of a switch. In particular, two test signals are injected, namely a first test signal and a second test signal. The two test signals are injected with the transceiver in a first and a second loopback configuration, respectively, in which the first and the second test signals looped back with a difference in phase shift. In other words, the respective loopback configurations are configured such that respective signals are looped from the transmitter to the receiver with a difference in phase relative to each other. Furthermore, linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the first loopback configuration, as well as linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver in the second loopback configuration are obtained. These frequency characteristics may be obtained in various ways, which are discussed in more detail further below. With the obtained frequency characteristics, and the first and the second test signals as injected and received, it is possible to estimate a first compensation signal for suppressing a spurious tone during a transmission by the transmitter and a second compensation signal for suppressing a spurious tone during a reception by the receiver. The embodiments disclosed herein enable estimation of compensation signals for suppressing spurious tones quickly and with relatively few calculations. Furthermore, the embodiments disclosed herein can be performed with low disruption of served traffic (i.e., low capacity loss). Below follows a presentation of how a transceiver may be modelled mathematically using linear and conjugate- linear frequency characteristics (Figure 2). Thereafter, a model of a transceiver, using linear and conjugate-linear frequency characteristics, is presented (Figure 3). After that, compensation of IQ imbalance in the transceiver is discussed (Figure 4). Thereafter, estimation and compensation of spurious tones in the transceiver are presented (Figures 5-13). Figure 2 – Widely linear systems To analyze “almost-rotationally-invariant” systems, e.g., frequency dependent IQ imbalance in analog baseband, it is helpful to use so-called widely linear systems. First, correspondence between complex-coefficient single-input- single-output (SISO) systems and rotationally invariant real-coefficient two-input two-output (TITO) systems are presented.A linear time-invariant (LTI) SISO system acting on complex signal u(t) = ure(t) + i ⋅ uim(t) may be describedin the Laplace domain with a transfer function G(s) = Gre(s) + i ⋅ Gim(s) where the transfer functions Gre(s)and Gim(s) have real coefficients. Such linear system may equivalently be described by a two-input two-output system acting on vector-valued signals ^^(t) = [ure(t)ut]. im( )With a transfer function matrix (s) −Gim(s)G] ( r(s)1) e Note the “rotationally invariant” structure of ^^TITO(s). Even if a TITO system does not quite have the structure as in Equation (1), but is only approximately of that form, e.g., ss s s where G̃re(s) and G̃im(s) are small relative to Gre(s) or Gim(s), it might still be helpful with a complex signal perspective. In this case, the complex-signal system may take on the form in Figure 2, where G(s) = Gre(s) + i ⋅ Gim(s), andG̃(s) = G̃re(s) + i ⋅ G̃im(s).Figure 2 shows a representation of a widely linear system, where G and G̃ are linear systems representing a linear frequency characteristic and a conjugate-linear frequency characteristic, respectively. The double lines indicate complex signals. Note that in addition to the linear dependence on the input signal, ^^(t), via G(s), there is also a linear dependence on the conjugate of the input signal, ^^∗(t), via G̃(s). Systems of the form in Figure 2 are referred to as widely linear systems and are represented by tuples (G, G̃) of two complex-coefficient systems. The term widely linear was introduced in Bernard Picinbono and Pascal Chevalier. “Widely linear estimation with complex data”. In: IEEE transactions on Signal Processing 43.8 (1995), pp.2030–2033. Here, G represents a first linear system and G̃ represents a second linear system. Furthermore, G and G̃ may be said to be the dynamics of the widely linear system represented by (G, G̃). G may be referred to as the linear frequency characteristic and G̃ may be referred to as the conjugate-linear frequency characteristic. Alternatively, G may be referred to as the linear dynamics or same frequency dynamics, and G̃ may be referred to as the antilinear dynamics or conjugate-linear dynamics. From Equation (2), it may be seen that any real-coefficient TITO system ^^(s) = [G11(s) G12(s)G (s) ( )] 21 G22 smay be modeled as a widely linear system (G, G̃) = (Gre + i ⋅ Gim, G̃re + i ⋅ G̃im)where Gre = (G11 + G22) / 2Gim = (−G12 + G21) / 2G̃re = (G11 − G22) / 2G̃im = (G12 + G21) / 2The dynamics described herein are typically frequency dependent. However, frequency dependencies are initially omitted from the equations and expressions to keep the notation manageable. While this widely linear representation is possible for any real-coefficient TITO system, it is typically only meaningful for systems that, in some sense, are almost rotationally invariant.From Figure 2 it follows that the output of a widely linear system (G, G̃) subjected to a complex-sinusoidal inputx(t) = eiω0t is, after transients, given by= G eiω0t + G̃ e−iω0t. This means that if the conjugate-linear frequency characteristic G̃ is nonzero (e.g., if there are IQ imbalances) there will be a mirror image G̃(−iω0)e−iω0t, in addition to the frequency image G(iω0)eiω0tthat is to be expected from a linear system (in contrast to a widely linear system). Correspondingly, in the frequency domain, the response to a signal X(iω) is given by Y(iω) = G(iω)X(iω) + G̃(iω)X∗(−iω) where X∗denotes the complex conjugate of X. Thus, IQ imbalance of the widely linear system may be quantified by a relative image rejection ratio (IRR) expressed as 2 Note that there also other ways of defining IRR. Below, a type of test signal that may be useful for analyzing widely linear systems is presented.To analyze and / or compensate IQ imbalance of a widely linear system (G, G̃), a first step may be to identify thelinear and the conjugate-linear frequency characteristics of the linear systems G and G̃, over some finite frequency grid Ω.From Equation (3) it can be seen that by injecting a test signal consisting of a single tone at frequency ω0 ∈ Ωand analyzing the components at ±ω0yields estimates of G(iω0)and G̃(−iω0). Repeating this experiment forall ω0 ∈ Ω (and ω0 ∈ −Ω if Ω is not symmetric about 0) gives the frequency characteristics for G and G̃ at allfrequencies Ω. To reduce the number of test signals to use, it is possible to use a test signal that has energy at more than one frequency. However, to make it easy to resolve the contributions from the same-frequency and mirror-frequency dynamics it is helpful if the signals have the following property. Definition: A finite discrete-time signal X may be referred to as a “vampire signal” if the set ΩXof digital Fourier transform (DFT) frequencies where it has non-zero energy is disjoint from the mirror of itself, i.e., if ΩX ∩ −ΩX = ∅.Two examples of sets of DFT frequencies that correspond to vampire signals are ΩX = {... , −5Δω, −3Δω, −Δω, 2Δω, 4Δω, 6Δω, .. . },and ΩX = {... , −4Δω, −3Δω, −2Δω, −Δω}.where Δω is some positive frequency. If a so-called vampire signal is injected into a widely linear system, the spectral content of the output signal at frequencies ΩXis determined only by the linear frequency characteristic and the spectral content at the mirror frequencies −ΩXis determined only by the conjugate-linear frequency characteristic. To identify the dynamics (i.e., linear frequency characteristic and conjugate-linear frequency characteristic) at (almost) all frequencies, it is possible to use two test signals XAand XB, which are respective vampire signalswhere XA has spectral content at frequencies ΩA and where XB has spectral content at frequencies ΩB = −ΩA. 3 – Model of a transceiver Figure 3 shows a schematic representation of a transceiver 300. The transceiver 300 may, e.g., be part of a wireless device 121 or a network node 110, which in turn may be part of the wireless communications network 100. The transceiver 300 comprises a transmitter 310 provided with a transmitter antenna element 330. A signal X comprising an in-phase (I) component and a quadrature (Q) component is injected into the transmitter 310. The signal X is modulated, i.e., upconverted to RF or IF (intermediate frequency), by a modulator 320. The transceiver 300 further comprises a receiver 340 provided with a receiver antenna element 360. A signal Y comprising an I- component and a Q-component is outputted by the receiver 340. The signal Y has been demodulated, i.e., downconverted from RF or IF, by a demodulator 350. The transceiver 300 is provided with a with a loopback path that enables the signal X, as injected, to be looped back from the transmitter 310, after being modulated by the modulator 320, into the receiver 340 such that the signal is demodulated by the demodulator 350. Thus, the signal Y is the signal X as received from the receiver when signal X is injected using the loopback path. In Figure 3, the loopback path is illustrated by arrow 301. The loopback path may be implemented in different ways, such as by a transmission line providing a signal path from point along the transmitter 310 after the modulator 320 modulates the injected signal to a point along the receiver 340 before the demodulator 350 demodulates the injected signal. Such transmission line may be connected and disconnected by means of a switch. When such switch is connecting the transmitter 310 to the receiver 340, the transceiver 300 is arranged in a loopback configuration. Different ways of looping back the signal injected into the transmitter 310 to the receiver 340, i.e., providing a loopback configuration of the transceiver 300, are discussed in more detail below.The dynamics of the transceiver 300 in the loopback configuration is represented by a widely linear system (G, G̃).The dynamics of the transmitter 310 is represented by a widely linear system (FTX, FT̃X) and the dynamics of thereceiver 340 is represented by a widely linear system (FRX, FR̃X). The widely linear system of the transceiver 300may thus be expressed as =+ ∗ + Here, it is assumed that the loopback path of the loopback configuration is an ideal direct connection represented by the widely linear system (1,0). Figures 4-5 – Compensation of IQ imbalance To compensate for IQ imbalance, compensation filters may be used. In Figure 4, a transmitter compensation filter411 with dynamics represented by the widely linear system (1, WT̃X) has been provided at the input of thetransmitter 310, and a receiver compensation filter 441 with dynamics represented by the widely linear system(1, WR̃X) has been provided at the output of the receiver 340. Figure 5 shows an example implementation of acompensation filter, where W(z) in the figure may be any of WT̃Xand WR̃X. To start with, it is assumed that transmitter and receiver compensation filters 411, 441 are respective, possibly non- causal, complex-coefficient filters. Furthermore, it is assumed that transmitter and receiver compensation filters 411, 441 are respective continuous time compensators. In practice, however, the compensation filters would normally be implemented in discrete time. For the transmitter 310, the compensation is applied before the IQ imbalance generated by the transmitter 310, i.e., at the input of the transmitter 310. The combined dynamics for the transceiver 300 and the transmitter compensation filter 411 (not including the receiver compensation filter 441) is (G + G̃WT̃∗X , G̃ + GWT̃X).For the receiver 340, the compensation is applied after the IQ imbalance is generated by the receiver 340, i.e., after the output of the receiver 340. The combined dynamics for the transceiver 300 and the receiver compensation filter 441 (not including the transmitter compensation filter 411) is +W ∗ G̃ +. These expressions seem conceptually the same. However, G ≠ although, typically G ≈ G∗. The two equations above may be re-arranged to ̃ G G∗G∗(+ W̃̃ ,Q̃ + W̃)G∗RXG∗RX RXwhere Q̃TX = G̃ / G and Q̃RX = G̃ / G∗. Note that the first elements of the widely linear representations above, i.e., the respective linear frequency characteristics, are close to unity since the magnitude of the respective IQ imbalances is relatively small. To zero out the mirror-frequency dynamics (i.e., maximize the IRR), ideal (hypothetical) continuous-time compensators 411 and 441 should be chosen as (iω) = (iω) = (iω).Note that such compensators give a change to the respective linear dynamics. Such change, however, is typically rather small. There are several ways of implementing a filter structure in practice that approximates the continuous-time compensators 411 and 441 discussed above. Such ways are generally known and will not be discussed further herein. Since the transmitter 310 and the receiver 340 operate individually during online transmission / reception, it is desired to obtain a transmitter compensation filter 411 that only compensates for the IQ-imbalance introduced by the transmitter 310, and a receiver compensation filter 441 that only compensates for the IQ-imbalance introduced bythe receiver 340. Consequently, it is desired to obtain (FTX, FT̃X) of the transmitter 310 and (FRX, FR̃X) of thereceiver 340 separated from (G, G̃) of the transceiver 300. One way of achieving this is to inject the signal X duringtwo different loopback configurations of the transceiver 300. Each of the two different loopback configurations cause an injected signal to be modulated by the modulator 320, be looped back to the receiver 340, and thereafter be demodulated by the demodulator 350. The signal injected during the first loopback configuration may be referred to as XΦ1and signal injected during the second loopback configuration may be referred to as XΦ2. Furthermore, the two different loopback configurations are configured such that XΦ1and XΦ2are looped back to the receiver 340 with a difference in phase shift ∆^^. The two different loopback configurations may, e.g., be provided by adding a controllable phase shifter to a transmission line providing a signal path between the transmitter 310 and the receiver 340, as discussed above. Alternatively, the difference in phase shift ∆Φmay be achieved by means of using different phases of a local oscillator (LO) signal driving the modulator and the demodulator for the two different loopback configurations. In another alternative, the transmitter 310 and the receiver 340 are part of different transmit and receive chains using respective phase locked loops (PLLs). In that case, the difference in phase shift ∆Φmay be achieved by changing the phase of the PLL itself, and not a phase shift of the signal from the PLL. In another alternative, two different transmission lines with different delays connects the transmitter 310 to the receiver 340 for the two different loopback configurations, respectively. These different ways of providing the two different loopback configurations, and other ways, are discussed in more detail below. When the transceiver 300 is configured in the first loopback configuration, the combined dynamics of the transceiver300, denoted may be represented by a combination of the dynamics of the transmitter 310, dynamicsof a widely of a first loopback path, and the dynamics of the receiver 340. The dynamicsof the first loopback path may be expressed as (J1, 0). When the transceiver 300 is in the second loopbackconfiguration, the combined dynamics of the transceiver 300, denoted (GΦ2, may be represented by acombination of the dynamics of the transmitter 310, the dynamics of the and dynamics of a widely liner system representation of a second loopback path. The dynamics of the second loopback path may beexpressed as (J2, 0).Note the first and the second loopback configurations may use the same (physical) signal path between the transmitter 310 and the receiver 340 (such as a single transmission line). Alternatively, the first and the second loopback configurations may separate the same (physical) signal paths between the transmitter 310 and the receiver 340 (such as two different transmission lines). In any case, the loopback path during the first loopback configuration is referred to as the first loopback path and the loopback path during the second loopback configuration is referred to as the second loopback path. The combined widely linear system of the transceiver 300 for the two loopback paths are then given by =+ ∗ + Preferably, the two loopback paths provide a frequency-independent difference in shift for XΦ1and XΦ2that is 90°, i.e., ∆ Φ= 90°. In that case, J1 = c and J2 = ic, where c is a widelylinear system of the transceiver 300 for the two loopback paths in that case then given by =+ ∗ + From these expressions, the following steps estimates the IQ imbalance ratios of the transmitter 310 and the receiver 340, respectively, and estimates the transmitter and the receiver compensator filters 411, 441, respectively:Step A. Identify and (GΦ2, G̃Φ2). In practice, these parameters are estimated, and not exacttheoretical computed IQ impairment ratios are also estimations, and not exacttheoretical representations. Thus, and (GΦ2, G̃Φ2) correspond to vectors of transfer functionsestimated at a finite set of below should be interpreted in an element-wise fashion, according to the corresponding transfer functions. Step B. Form the following linear combinations L= GΦ1 − iGΦ2 = cFRXFTXL̃TX = G̃Φ1 − iG̃Φ2 = cFRXFT̃XL̃RX = G̃Φ1 + iG̃Φ2 = c∗FR̃XFTXStep C. Compute estimates of the IQ imbalance ratios, i.e., L̃ F̃ Step D. Fit a transmitter compensation filter 411, such as a FIR filter, to Q̃TX, and a receiver compensation filter 441, such as a FIR filter, to Q̃RX. As mentioned, a so-called vampire signal may advantageously be used as the test signal during the two loopback configurations. If XA,Φ1is a vampire signal injected into the transmitter 310 in the first loopback configuration, the dynamics obtained for the loopback configuration will only correspond to the frequency content of XA,Φ1. For example, XA,Φ1may be a discrete signal with spectral content at ΩX^^ = {... , −5Δω, −3Δω, −Δω, 2Δω, 4Δω, 6Δω, ... }. With such signal, XA,Φ1 as received from the receiver 340 enables estimation of where GA,Φ1 isthe linear frequency characteristic mapping to the frequency content of XA,Φ1, linear frequency characteristic mapping to the frequency content of XA,Φ1. In other words, GA,Φ1maps the frequency content ΩX^^of the injected signal to the same frequencies of the received signal from the receiver, and G̃A,Φ1maps the frequency content ΩX^^of the injected signal to the corresponding mirror frequencies of ΩX^^of the received signal from the receiver.To obtain the dynamics of for all discrete frequencies within a bandwidth, another vampire signalXB,Φ1may be injected into 310 during the first loopback configuration, where XB,Φ1and X^^,Φ1have a disjoint support in the frequency domain. Following the example XA,Φ1as defined above, XB,Φ1may be a discrete signal with spectral content at ΩX^^ = {... , −6Δω, −4Δω, −2Δω, Δω, 3Δω, 5Δω, . .. }. With such signal, XB,Φ1 as received from the receiver 340 enables estimation of where GB,Φ1 is the linear frequency characteristic mapping to the frequency content of XB,Φ1, linear frequency characteristic mapping to the frequency content of XB,Φ1. In other words, GB,Φ1maps the frequency content ΩX^^of the injected signal to the same frequencies of the received signal receiver, and G̃B,Φ1maps the frequency content ΩX^^of the injected signal to the corresponding mirror frequencies of ΩX^^of the received signal from the receiver.Thereafter, (GA,Φ1, G̃A,Φ1) may be combined with (GB,Φ1, G̃B,Φ1) to obtain (GΦ1, G̃Φ1).Figures 6-7 – Spurious tones in a transceiver As mentioned, as part of the developing of the embodiments described herein, it has been realized that spurious tones in a transceiver may be estimated by using linear frequency characteristics and conjugate-linear frequency characteristics of the transceiver. The linear and the conjugate-linear frequency characteristics may be obtained in different ways (e.g., using the procedures discussed in connection to Figures 4-5). The obtained characteristics may thereafter be used to estimate compensation signals to suppress spurious tones in the transceiver. The spurious tones may represent any of undesired DC-offset, LO-leakage, and leakage (e.g., from a clock in the transceiver). Magnitudes and phases of spurious tones in the receiver and the transmitter (of one or more different baseband frequencies including DC) may, in some embodiments, be estimated from the same experiments as for calibration of frequency-dependent IQ (FDIQ) imbalance described above in connection to Figures 4-5, without additional calibration time. Furthermore, in some embodiments, the spurious tones may be estimated during runtime of the transceiver, which is advantageous. In other words, it is not required to abrupt operation of the transceiver by, e.g., turning of a low noise amplifier (LNA), to estimate the spurious tones. Figure 6 shows a mathematical model of the transceiver 300, in a loopback configuration, represented by a widelylinear system (G, G̃). Spurious tones may, e.g., enter in the analog baseband or in the RF components. In Figure6, the spurious tones are modelled as ^^^^^^and ^^^^^^. ^^^^^^is signal combined at an input of the transmitter 310 and represents spurious tones entering into the transmitter 310 that result in undesired spurious content from the transmitter 310. ^^^^^^is signal combined at an output of the receiver 340 and represents spurious tones entering into the receiver 340 that result in undesired spurious content from the receiver 340. ^^^^^^and ^^^^^^may include LO- leakage, DC-offset and other tones such as a reference clock leaking into the analog interface or anywhere else in the analog signal path. Figure 6 also shows ^^^^^^and ^^^^^^, which represent compensation signals to be applied at the input of the transmitter and the output of the receiver, respectfully, to cancel, or at least suppress, ^^^^^^and ^^^^^^. As an example, ^^^^^^may be selected as −^^^^^^and ^^^^^^may be selected as −^^^^^^. Figure 6 also shows a transmitter compensation filter 411 with dynamics represented by the widely linear system(1, WT̃X) provided at the input of the transmitter 310, and a receiver compensation filter 441 with dynamicsrepresented by the widely linear system (1, WR̃X) provided at the output of the receiver 340.Figure 7 shows a rearranged version of the mathematical model of Figure 6. In Figure 7, the widely linear system(^^, ^̃^) represent the transceiver 300 with the transmitter compensation filter 411 cascaded at the input of thetransmitter 310 and with the receiver compensation filter 441 cascaded at the output of the receiver 340. Furthermore, ^^^^^^, ^^^^^^, ^^^^^^, and ^^^^^^have been shifted and combined into ^^^^^^and ^^^^^^. More specifically, ^^^^^^ = (1, ^^^^^^)−1(^^^^^^ + ^^^^^^) and ^^^^^^ = (1, ^^^^^^)−1(^^^^^^ + ^^^^^^).The widely linear system of the transmitter and of the receiver, i.e., (^^^^^^ , ^̃^^^^^) and (^^^^^^ , ^̃^^^^^), as well thecompensation filters (1, ^^^^^^) and (1, ^^^^^^), may be obtained according to the procedures discussed above inconnection to Figures 4-5. With known vales of (^^, ^̃^) at a frequency of interest, it is possible to estimated desired^^^^^^, ^^^^^^(or equivalently desired ^^^^^^, ^^^^^^) at that frequency of interest. Note that for the purpose of estimating^^^^^^, ^^^^^^, it is not necessary to have compensation filters (1, ^^^^^^) and (1, ^^^^^^) included in the model of the transceiver 300. It is also possible to estimate ^^^^^^, ^^^^^^using a model of the transceiver 300 without thecompensation filters. In that case, (^^, ^̃^) is the same as (^^, ^̃^).With the widely linear system according to Figure 7, an output signal (^^[^^]) from the receiver in thefrequency domain (at a ^^) may be expressed as Y^^^^ = ^^^^^^ + ^^^^^^ + ^̃^ ∗^^^^ + ^^ ∗^^^^ + ^^^^^^ Here, ^^ is a natural number (including 0) and may represent a fast Fourier transform (FFT) bin. The subscript ^^represent one of the two loopback configurations (i.e., ^^ = 1 or 2). Thus, ^̃^^^1) is a widely linear system forthe first loopback configuration and is a widely linear the second loopback configuration.Furthermore, ^^ ^^^^[^^] represents content of the input signal at the baseband frequency ^^ and ^^^^^^[−^^] represents known spectral content of the input signal at the mirror frequency of the baseband frequency at frequency −^^. Note that input signal may comprise the same signal sequence for both loopbackconfigurations, i.e., ^^^^1 = ^^^^2.The output signal (^^[−^^]) at the mirror frequency of the baseband frequency ^^, may be expressed as Y= ^^ + ^^ ∗ ∗^^^^ ^^^^ ^^^^ + ^̃^^^^^ ^^^^ + ^^^^^^ + ^^^^^^ To separate ^^^^^^[^^] from ^^^^^^[^^] (and correspondingly ^^^^^^[−^^] from ^^^^^^[−^^]), at least two different loopbackconfigurations of the transceiver 300 – corresponding to widely linear systems and (^^^^2, ^̃^^^2) –should be used. This gives a total of 12 complex unknowns: ^^ ^^2, at FFT bins ^^ and −^^ when assuming that the spectral content of the input output signals are known at ^^ and −^^. Each experiment gives information about the two FFT bins ^^ and −^^ so in total 6 experiments are required to solve the for the unknowns directly from the input and output signals.For example, for the first loopback configuration and a given signal ^^^^1, changing the value in^^ [^^] makes it possible to solve for ^ ^^1^^^1[^^] value of ^^^^1[−^^] makes it possible to solve for . Doing this for both loopback configurations, it is possible to solve for the unwanted signal content ^^^^^^FFT bins ^^ and −^^. Depending on the circumstances, the number of required experiments may be reduced. One example is when the conjugate-linear frequency characteristics ^̃^^^^^is assumed to be zero. In another example, ^^^^1, ^̃^^^1, ^^^^2, ^̃^^^2 at FFT bins ^^ and −^^ are obtained from other ways, e.g., by interpolating estimates of the at nearby FFT bins (assuming that these adjacent bins are not contaminated by spurious tones). Any of DC-offset, LO-leakage, and reference clock leakage as a source of a spurious tone is frequency aligned with the sub-carrier grid of an orthogonal frequency-division multiple access (OFDMA) symbol. Thus, all the spurious energy will fall on one sub-carrier. In the context of estimating the spurious tones, it is assumed that signal levels and the system configuration are such that nonlinear distortion may be neglected. In other words, only spurious content (and possible also IQ imbalance) is considered. When estimating spurious tones from clock(s), all experiments during the first and the second loopback configurations should be started synchronously with the phase of the clock generating the spurious tone (alternatively, any phase offset may be recorded and accounted for in the computations). It is straightforward to start the experiments synchronously with spurious tones from reference clocks in the timing system since the timing system controls the alignment of the frame structure. To ensure synchrony, it is sufficient to align the experiments with the frame structure. The phase shift between the two loopback configurations does not have to be 90 degrees, as long as the conditioning of the resulting system of equations is sufficient for the desired estimation performance, which puts rather low demands on the phase shift difference. Below follows an example of estimations of spurious tones at DC.In the case of DC estimation, ^^ = 0, the number of complex unknowns to solve from Equations (4) and (5) reducesto six. By further assuming zero spectral content in the input signal (i.e., ^^^^1[^^] = ^^^^1[−^^] = ^^^^2[^^] =^^ [−^^] = 0) and that ^^ ^^^^[0] and ^̃^^^^^[0] are already obtained, then reduced^^ = ^^ ^^^^^^ + ^̃^ ^^ ∗^^^^ + ^^^^^^ ^^^^2[0] = ^^^^2[0]^^^^^^[0] + ^̃^^^2[0]^^ ∗^^^^ [0] + ^^^^^^[0]which corresponds to 4 real-valued equations with four real-valued unknowns. Note that when ^^^^1[0] ≈ ^^^^^^2[0](or ^^ [0] ≈ −^^^^ [0]) there are no issues of numerical conditioning in the s ^^2 olution of the ofThe following steps summarizes one example procedure for DC calibration (i.e., at ^^ = 0) when ^^^^1[^^] =^^ [−^^] = ^^ [^^] = ^^ ^^1 ^^2 ^^2[−^^] = 0. The input signals ^^^^1and ^^^^2may comprise spectral content at other frequencies than DC. In an example,^^ = ^^^^2= XA, XA is a discrete signal with spectral contentΩX^^ = {... , −5Δω, −3Δω, −Δω, 2Δω, 4Δω, 6Δω, ... }. It is further possible to inject a signal X^^in addition to XAfor the two loopback configurations, where X^^is a discrete signal with spectral content at ΩX^^ = {... , −6Δω, −4Δω, −2Δω, Δω, 3Δω, 5Δω, . .. }, and obtain ^^^^1[0] and ^^^^2[0] by averaging, i.e., ^^^^1[0] = (^^^^,^^1[0] + ^^^^,^^1[0]) / 2 and ^^^^2[0] =+ ^^^^,^^2[0]) / 2. Here, ^^^^,^^^^[0] is the output signal at DC for loopback configuration ^^ when XA is^^^^,^^^^[0] is the output signal at DC for loopback configuration ^^ when XBis injected.When injecting XA for the first and the second loopback configurations, it is possible to estimate and(^^ , at FFT bin ^^2 s related to the spectral content of XA for (other than DC in this notby spurious content. When injecting X^^for the first and the second loopback configurations, it ispossible to estimate and at FFT bins related to the spectral content of XB for (other thanDC in this example) content.Step 2. Obtain and . Obtaining these frequency characteristics at DC may be done in different ways. In one example, these frequencycharacteristics are obtained from interpolating estimate of and at nearby FFT bins (nearto DC in this example) that are not contaminated by Step 3. Solve the linear system of equations for ^^^^^^[0] and ^^^^^^[0]. Step 4. All steps may be done iteratively, where the corrections ^^^^^^[0] and ^^^^^^[0] are updated according to ^^^^^^[0] = ^^^^^^,^^^^^^^^[0] − (1, ^^^^^^[0])^^^^^^[0]^^^^^^ = − ^^^^^^ )−1^^^^^^ In the first iterations, previous correction values ^^^^^^,^^^^^^^^[0] and ^^^^^^,^^^^^^^^[0] may be set equal to zero. In general, the correction values may, e.g., be added in the time domain or the frequency domain during operation of the transceiver. For OFDMA based system (such as LTE, NR and potentially also 6G), a complex constant can be added to the affected sub-carrier that will cancel out (or at least suppress) the spurious tone (which may originate in time domain, in e.g., the analog interface between RFIC and DAC / ADC). More specifically, in an OFDMA system, a constant complex value may be added before the inverse FFFT (IFFT) in transmitter path and in receiver path a constant complex value is added after the FFT. The phase of the spurious tone will shift due to length com the cyclic prefix in the OFDMA system, but that is compensated by shifting the phase of the added complex constant. This phase shift is only needed to be done once per symbol. The phase shift is set by the length of the cyclic prefix and the frequency of the spurious tone. The length of the cyclic prefix is defined in corresponding standard of the OFDMA system, and the frequency of the spurious tone is known from estimation / design / production calibration. Thus, it is straight forward to estimate the phase shift of the applied complex constant needed per symbol. For the case of a DC-offset (spurious tone at DC), then there will not be any phase shift during cyclic prefix. If the cancellation is to be done in time domain, example for 3G (WCDMA) and 2G (GSM) systems (or when it is for some other reason not preferred to be implemented in frequency domain), then numerical controlled oscillators (NCOs) may be used to generate a continuous digital wave in time domain that can cancel (or at least suppress) out the spurious tone. Particularly a respective NCO per spurious tone to cancel for the transmitter and a respective NCO per spurious tone to cancel for the receiver. The amplitude and phase of the signal from the respective NCOs is then set from the compensation signals according to the present disclosure (i.e., ^^^^^^, ^^^^^^, ^^^^^^, ^^^^^^). An alternative way to do the compensation in time domain is to use an NCO to frequency shift wanted data using a complex multiplication (signal * ejw), so that the spurious is rotated down to DC frequency. A complex constant is then added, with amplitude and phase from the compensation signals according to the present disclosure (i.e., ^^^^^^, ^^^^^^, ^^^^^^, ^^^^^^). After that, the signal is rotated back with a second complex multiplication with an NCO on same but negative frequency (signal * e-jw). Figures 8A-8D – First illustration of spurious tones Figures 8A-8D illustrate how the unwanted signals ^^^^^^and ^^^^^^may be derived for a spurious tone at DC, i.e.,FFT bin zero (^^ = 0), using two experiments, viz., the first and the second loopback configurations. For illustrationpurposes, it is assumed that ^̃^^^1[0] = ^̃^^^1[0] = 0. It is also assumed that the spectral content at DC of theinput signal for the two experiments is zero, i.e., ^^^^1[0] = ^^^^2[0] = 0. It is further assumed that ^^^^2[0] =^^^^^^1[0]. In other words, the difference in phase shift (∆^^) is +90 degrees. Consequently, ^^^^1is denoted as ^^ in Figures 8A-D and in the equations below. Figure 8A shows the output signal ^^^^1[0]and its components for the first experiment. In particular, the output signal is ^^^^1[0] = ^^^^^^[0] + ^^[0]^^^^^^[0].Figure 8B shows the output signal ^^^^2[0] and its components for the second experiment. It can be seen that the spurious energy from the transmitter (^^^^^^) is rotated -90° relative to Figure 8A, and that spurious energy from the receiver (^^^^^^) maintains its phase relative to Figure 8A. In particular, the output signal for the second experiment is ^^^^2[0] = ^^^^^^[0] + ^^^^[0]^^^^^^[0].Figure 8C shows ^^^^1[0] − ^^^^2[0], which may be expressed as^^^^1[0] − ^^^^2[0] = (1 − ^^)^^[0]^^^^^^[0].Figure 8D shows ^^^^1[0] + ^^^^^^2[0], which may be expressed as^^^^1[0] + ^^^^^^2[0] = (1 + ^^)^^^^^^[0].Using the two different combinations of ^^^^1[0] and ^^^^2[0], ^^^^^^[0] and ^^^^^^[0] may be obtained from^^^^1[0] − ^^^^2[0] 9A-9B – Second illustration of tones Figures 9A-9B provide additional illustrations of the example of Figures 8A-8D. Figures 9A-9B show a zoomed in plot of the FFT for the captured output signal with an input signal X^^. More specifically, Figure 9A shows magnitude versus frequency and Figure 9B shows phase versus frequency. X^^is a discrete signal with spectral content at ΩX^^ = {... , −6Δω, −4Δω, −2Δω, Δω, 3Δω, 5Δω, . .. }. Figures 9A-9B shows the two FFT bins closest to DC. The different FFT bins are denoted SC#-2,-1,0,1,2 in the figures (where SC means sub carrier). The captured output signal has been normalized to the spectral content at SC#-2 and SC#1 (which in this example are the same). A respective image component can be seen at SC#-1 and at SC#2. At DC (i.e., SC#0), the combined spurious tones from the transmitter and receiver can be seen (i.e.,^^[0]^^^^^^[0] + ^^^^^^[0]). The frequency characteristics close to DC may be estimated from the captured outputsignal and the known signal X^^. This may, e.g., be done using the tone closest to DC (i.e., at SC#1) or by doing an interpolation of the (two) tones closest to DC (i.e., SC#-2 and SC#1). Figures 10A-10D – Loopback configurations of a transceiver Figures 10A-10D illustrate respective transceivers 300 with different ways of implementing the two loopback configurations. In Figure 10A, the first loopback configuration is provided by connecting a first signal path 1002 by means of a first switch 1071. The first signal path 1002 electrically connects the transmitter 310 at a point after the injected signal has been modulated by the modulator 320 to the receiver 340 at a point before the injected signal is demodulated by the demodulator 350. The first loopback configuration is provided by connecting a second signal path 1002 by means of a second switch 1072. The second signal path 1002 electrically connects the transmitter 310 at a point after the injected signal has been modulated by the modulator 320 to the receiver 340 at a point before the injected signal is demodulated by the demodulator 350. The second signal path provides a difference in phase shift of a signal injected during the second loopback configuration relative to a signal injected during the first loopback configuration. This difference in phase shift may, e.g., be provided by an analog phase shifter 1073. Alternatively, or in combination of, the first and the second signal paths 1002, 1003 may have different electrical lengths. Note that a first additional switch may be provided, which connects and disconnects the transmitter antenna element 330 from the transmitter 310, and a second additional switch may be provided, which connects and disconnects the receiver antenna element 360 from the receiver 340. In this way, it is possible to inject signals via the two loopback configurations without radiating energy or receiving radiated energy. In Figure 10B, the first and the second loopback configurations comprise the same signal path 1002. In Figure 10C, the first and the second loopback configurations comprise the same signal path 1004, i.e., a propagation path from the transmitter antenna element 330 to the receiver antenna element 360. In Figure 10D, the transmitter 310 and the receiver 340 share a common antenna structure 1090. A combiner 1091 connects an output of the transmitter 310 to the antenna structure 1090 and connects an input of the receiver 340 to the antenna structure 1090. The combiner 1091 may, e.g., comprise a switch configured to switch between a connection from the output of the transmitter 310 to the antenna structure 1090 or the input of the receiver 340 to the antenna structure 1090. Alternatively, the combiner 1091 may comprise a circulator. In this case, the first and the second loopback configurations comprise the same signal path 1005. In Figures 10B-10D, the difference in phase shift ∆Φmay be provided by means of using different phases of a LO signal driving the modulator and the demodulator for the two different loopback configurations. This is discussed in more detail below. As another example, the difference in phase shift ∆Φmay be provided by means of a phase shifter in the signal path, such as a phase shifter used for analog beamforming. Figure 11 – Example transceiver Figure 11 shows an example transceiver 300 in more detail. A transmitter compensation filter 411, such as a FIR filter, has been provided at the input of the transmitter 310, and a receiver compensation filter 441, such as a FIR filter, has been provided at the output of the receiver 340. In Figure 11, box 1191 represents applying a compensation signal ^^^^^^to the input of the transmitter 310 before the transmitter compensation filter 411 and box 1192 represents applying a compensation signal ^^^^^^to the output of the receiver 340 after the receiver compensation filter 441. As discussed above, it is possible to alternatively applying a compensation signal ^^^^^^to the input of the transmitter 310 after the transmitter compensation filter 411 and applying a compensation signal ^^^^^^to the output of the receiver 340 before the receiver compensation filter 441. In any case, the compensation signals ^^^^^^or ^^^^^^are for suppression of a spurious tone during a transmission by the transmitter 310 and the compensation signals ^^^^^^or ^^^^^^are for suppression of a spurious tone during a reception by the receiver 340. As is also mentioned above, it is possible to apply compensation signals without the presence of the transmitter compensation filter 411 and the receiver compensation filter 441. The compensation signal ^^^^^^or ^^^^^^is added to a transmission signal at the input of the transmitter 310 to suppress a spurious tone during the transmission of the transmission signal. The compensation signal ^^^^^^or ^^^^^^is added to a reception signal at the output of the receiver 340 to suppress a spurious tone during the reception of the reception signal. The transmission and the reception signals may be adapted for carrying traffic or user data.A digital IQ signal X (which e.g., may be XA,Φ1, X^^,Φ2, XB,Φ1, and XB,Φ2 discussed above) is injected into thetransmitter compensation filter 411 and thereafter into an input of the transmitter 310. Here, the compensation signals ^^^^^^or ^^^^^^may be omitted in a first iteration in a set of iterations. The filtered I and Q components are thereafter converted to the analog domain by respective DACs 1111. Thereafter, the analog I and Q components pass through an analog interface 1193 into respective low pass filters (LPFs). Note that such LPFs are not mandatory, i.e., some embodiment of the transceiver 300 do not comprise such LPFs. Thereafter, the filtered analog I and Q components are modulated to RF by a modulator, which in this case is a quadrature modulator 1120. The quadrature modulator 1120 comprises a mixer 1121 for each of the I and Q components. The respective mixers are driven by an LO signal from an LO 1180. A phase shifter 1122 shifts the LO signals driving the two mixers 1121 by 90° relative to each other. The respective upconverted I and Q signals are thereafter combined by a combiner 1123, which performs a summation of the upconverted I and Q signals. The resulting RF signal thereafter passes through a power amplifier (PA) 1113 and a band pass filter (BFP) 1114. Note that none of the PA and BPF are mandatory. The amplified and filtered RF signal is thereafter radiated by the transmitter antenna element 330. Also note that the transmitter 310 may comprise a chain of cascaded PAs. Other components may also be present in the transmitter 310. An example of another component may be an antenna switch configured to selectively connect either the output of the transmitter to an antenna element or the input of the receiver to the same antenna element. At the receiver 340 of the example transceiver 300 of Figure 11, an RF signal may be received by the receiver antenna element 360. The received RF signal passes through a BPF 1144 and a low noise amplifier (LNA) 1143. Note that none of the LNA and BPF are mandatory. Also note that the receiver 340 may comprise a chain of cascaded LNAs. Other components may also be present in the receiver 340. The amplified and filtered RF signal is thereafter demodulated to baseband by a demodulator, which in this case is a quadrature demodulator 1150. A splitter 1153 divides the amplified and filtered RF signal into two parts. The quadrature modulator 1150 comprises a mixer 1151 for each of the two parts. The respective mixers 1151 are driven by an LO signal from the LO 1180. A phase shifter 1152 shifts the LO signals driving the two mixers 1151 by 90° relative each other. The resulting analog I and Q components are thereafter filtered by respective LPFs 1142. Note that such LPFs are not mandatory. The filtered analog I and Q components thereafter pass through an analog interface 1194 into respective ADCs 1141 and are converted to the digital domain. The filtered analog I and Q components are then outputted from an output of the receiver 340 and are thereafter filtered by the receiver compensation filter 441. Here, the compensation signals ^^^^^^or ^^^^^^may be omitted in a first iteration in a set of iterations. The resulting IQ signal is referred to as ^^. The transceiver 300 in Figure 11 may be configured in the first and the second loopback configurations according to the discussions above. In that case, ^^ corresponds to ^^ as injected into the transmitter that is subsequently received from the receiver. As an example, the transceiver may be provided with the first and the second signal paths 1002, 1003 of Figure 10A. One or more signal paths may connect the transmitter 310 and the receiver 340 at different points along the transmitter 310 and the receiver 340. For example, the one or more signal paths may connect a point between the combiner 1123 and the PA 1113 to a point between the splitter 1153 and the LNA 1143. This is illustrated by arrow 1101. Alternatively, the one or more signal paths connect a point between the PA 1113 and the BPF 1114 to a point between the LNA 1143 and the BPF 1144. This is illustrated by arrow 1102. In another alternative, the one or more signal paths connect a point between the BPF 1114 and the transmitter antenna element 330 to a point between the BPF 1144 and the receiver antenna element 360. This is illustrated by arrow 1103. In yet another alternative, a signal path is the propagation channel between the transmitter antenna element 330 and the receiver antenna element 360. This is illustrated by arrow 1104. The different alternatives represented by 1101, 1102, 1103, 1104, enable estimation of IQ imbalance resulting from different parts of the transmitter 310 and IQ imbalance resulting from different parts of the receiver 340. For the transmitter 310, alternative 1101 allows estimation of the combined IQ imbalance resulting from DACs 1111, the LPFs 1112, and the quadrature modulator 1120. Alternative 1102 additionally includes the combined IQ imbalance resulting from the PA 1113. Alternative 1103 additionally includes the combined IQ imbalance resulting from the BPF 1114. Alternative 1104 additionally includes the combined IQ imbalance resulting from the transmitter antenna element 330. For the receiver 340, alternative 1101 allows estimation of the combined IQ imbalance resulting from ADCs 1141, the LPFs 1142, and the quadrature modulator 1150. Alternative 1102 additionally includes the combined IQ imbalance resulting from the LNA 1143. Alternative 1103 additionally includes the combined IQ imbalance resulting from the BPF 1144. Alternative 1104 additionally includes the combined IQ imbalance resulting from the receiver antenna element 360. In general, the first and the second loopback configurations may loopback a signal injected into the transmitter 310 at any point along the transmitter 310 after the injected signal has been modulated to any point along the receiver 340 before the injected signal is demodulated. Many other signal paths than the examples of by 1101, 1102, 1103, 1104 are also possible. For example, it is possible to provide a signal path from an output of the PA 1113 to the input of the splitter 1153. Figure 12 – Method for enabling suppression of spurious tones With reference to Figure 12, there is disclosed herein a method 1200 for enabling suppression of spurious tones in a transceiver 300. The transceiver 300 comprises a transmitter 310 having a modulator 320 and a receiver 340 having a demodulator 350. The transceiver 300 may be for wireless communications. The transceiver 300 may be any of the example transmitters shown in Figures 3, 4, 10A, 10B, 10C, 10D, and 11. The transceiver 300 may be comprised in a wireless device 121 or a network node 110 shown in Figure 1. The transceiver may be a homodyne transceiver. A homodyne transceiver means there is a direct upconversion from baseband to RF in the transmitter of a baseband signal injected into the transmitter, and a direct downconversion of an RF signal received by the receiver to baseband. The modulator 320 may be a quadrature modulator 1120 such as the one shown in Figure 11, and the demodulator 350 may be a quadrature demodulator 1150 such as the one shown in Figure 11. The transmitter 310 may comprises a digital-to-analog converter (DAC) physically separated from the modulator 320, and the receiver 340 may comprise an analog-to-digital converter (ADC) physically separated from the demodulator 350. Furthermore, there may be an analog interface 1193 between the DAC and the modulator 320, and there may be an analog interface 1194 between the ADC and the demodulator 350. The DAC may be implemented in a different circuit technology than the modulator 320, and the ADC may be implemented in a different circuit technology than the demodulator 350. The transceiver 300 may comprise a fist part of a first circuit technology and a second part of a second circuit technology, where in the DAC and ADC are implemented on the first part and the modulator 320 and demodulator 350 are implemented on the second part. The transceiver 300 is adapted for being put in a first and a second loopback configuration, respectively, in which respective signals injected to the transmitter 310 are modulated by the modulator 320, looped back to the receiver 340 with a difference in phase shift ∆^^, and thereafter demodulated by the demodulator 350. In other words, the first and the second loopback configurations are configured such that the first test signal is looped back from the transmitter 310 to the receiver 340 with a difference in phase relative to the second test signal as looped back from the transmitter 310 to the receiver 340. The method 1200 may comprise a number of actions, which are discussed below. Action 1210. The method 1200 comprises injecting a first test signal ^^^^1and a second test signal ^^^^2into the transmitter 310 with the transceiver 300 being in the first and the second loopback configurations, respectively. In other words, the first ^^^^1is injected during the first loopback configuration of the transceiver 300 and the second test signal ^^^^2is injected during the second loopback configuration of the transceiver 300, such that the first and the second test signals ^^^^1, ^^^^2are looped back with the difference in phase shift ∆^^relative to each other. Thus, the first and the second loopback configuration loops the first and the second test signals ^^^^1, ^^^^2from the transmitter to the receiver with a difference in phase relative to each other. Each of the first and the second loopback configurations cause an injected signal to be modulated by the modulator 320, be looped back to the receiver 340, and thereafter be demodulated by the demodulator 350. As mentioned, the first and the second loopback configurations may use the same (physical) signal path between the transmitter 310 and the receiver 340 (such as a single transmission line). Alternatively, the first and the second loopback configurations may use separate (physical) signal paths between the transmitter 310 and the receiver 340 (such as two different transmission lines). As an example, the first and the second loopback configurations provide a frequency-independent difference inphase shift for the first and the second test signals XΦ1, XΦ2 that is ±90°, i.e., ∆^^= ±90°. Other values of thedifference phase shift ∆^^is also possible, as is discussed below. In general, the first and the second test signals ^^^^1, ^^^^2are respective digital baseband signals with respective I and Q components. Each digital baseband signal comprises a sequence, i.e., of a number of samples, which are sampled at a sample rate. As an example, a signal sequence may be 64 samples long and be provided with subcarrier spacing (SCS) of 30.72 MHz. In some embodiments, the first and the second test signals ^^^^1, ^^^^2comprise the same signal sequence. However, it is possible to use different signal sequences.The first and the second test signals ^^^^1, ^^^^2 may comprise respective known spectral content at a first basebandfrequency ^^ and at a mirror frequency of the first baseband frequency −^^. Furthermore, the respective known spectral content at the first baseband frequency −^^ and at the mirror frequency of the first baseband frequency −^^ may be zero or non-zero. Each of the first and the second test signals ^^^^1, ^^^^2comprise a grid of FFT bins, and the first baseband frequency may represent one of these FFT bins.The first and the second test signals ^^^^1, ^^^^2 may additionally comprise known spectral content at otherfrequencies than the first baseband frequency ^^ and the mirror frequency of the first baseband frequency −^^. For example, ^^^^1and ^^^^2may be a signal with spectral content at ΩX^^={... , −5Δω, −3Δω, −Δω, 2Δω, 4Δω, 6Δω, ... }, or a signal with spectral content at ΩX^^ ={... , −6Δω, −4Δω, −2Δω, Δω, 3Δω, 5Δω, ... }. If the known spectral content at −^^ and at ^^ is zero, and^^^^1and ^^^^2may be a signal sequence with spectral content at ΩX^^or ΩX^^that it is punctured (i.e., has zero magnitude) at frequencies corresponding to −^^ and at ^^. The first baseband frequency may be at DC, or the first baseband frequency may be non-zero. When, the firstbaseband frequency is at DC, it follows that ^^ = −^^ = 0. Furthermore, when the first baseband frequency is atDC, it follows that parameters at ^^ and at −^^ become equal, e.g., ^^^^1[^^] = ^^^^1[−^^], ^̃^^^1[^^] = ^̃^^^1[−^^],[^^] = [^^] = [−^^] ^^ ^^^^[^^] = ^^^^^^ [^^] = ^^^^^^ . In some embodiments, the first and the second test signals ^^^^1, ^^^^2are vampire signals according to thediscussions above. As an example, ^^^^1 = ^^^^2 and has signal energy at four frequencies, where, at thecorresponding mirror frequencies of those four frequency components, the signal energy is zero. However, it is not required that each of the first and the second test signals ^^^^1, ^^^^2have respective sets of digital Fourier transform (DFT) frequencies where it has non-zero energy is perfectly disjoint from the mirror of itself. In other words, each of the first and the second test signals ^^^^1, ^^^^2may comprise a plurality of primary frequency components, where respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are zero or below a threshold. The magnitude of each frequency component of the primary frequency components is non-zero. The threshold may be relative to any of the frequency components of the plurality of primary frequency components. Furthermore, the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components may be below 20 dB, preferably below 30 dB, and more preferably below 40 dB, relative to any of the frequency components of the plurality of primary frequency components. Furthermore, a subset of the plurality of primary frequency components is above DC and the remainder of the plurality of primary frequency components is below DC. The plurality of primary frequency components may comprise at least five frequency components, and preferably at least ten frequency components. As an example, when a vampire signal formed in the frequency domain is converted to time domain, limited numerical precision in practice will add some noise to the signal. Consequently, the frequency spectrum of the converted signal will comprise added quantization noise at all frequencies. In addition, thermal noise will be present at all frequencies of the signal in practice.When the first and the second test signals ^^^^1, ^^^^2 may additionally comprise known spectral content at otherfrequencies than the first baseband frequency ^^ and the mirror frequency of the first baseband frequency −^^, andin particularly when the first and the second test signals ^^^^1, ^^^^2 are vampire signals, the estimation of thecompensation signals in the disclosed method can be based on the same type of test signals / experiments used for estimation frequency-dependent IQ (FDIQ) imbalance. Consequently, the disclosed method saves time and resources, which may further reduce the impact on traffic. As mentioned above (particularly in connection to 10A-10D), the first and the second test signals ^^^^1, ^^^^2may be looped back in different ways. As an example, the first and the second test signals ^^^^1, ^^^^2as injected, may be looped back via one or more transmission lines or via a one or more waveguides. One transmission line or one waveguide may provide a signal path for an injected signal from any point of the transmitter after the injected signal is modulated by the modulator 320 to any point of the receiver before the injected signal is demodulated by the demodulator 350. Two transmission lines or two waveguides may provide respective signal paths for two injected signals from any point of the transmitter after the two injected signals are modulated by the modulator 320 to any point of the receiver before the two injected signals are demodulated by the demodulator 350. The one or more transmission lines or the one or more waveguides may be accompanied by respective switches configured to connect and disconnect the respective signal paths. If two transmission lines or two waveguides are used, the first test signal ^^^^1may be looped back via one of the two transmission lines or two waveguides, and the second test signal ^^^^1may be looped back via the other of the two transmission lines or two waveguides. In some embodiments, the transmitter 310 comprises a transmitter antenna element 330 and the receiver 340 comprises a receiver antenna element 360. In that case, the first and the second test signals ^^^^1, ^^^^2as injected, may be looped back from the transmitter antenna element 330 to the receiver antenna element 360. Thus, the injected signals are looped back via a propagation channel between the transmitter antenna element 330 and the receiver antenna element 360. In this way, no switches need to be added to the transceiver 300. Consequently, the method may be implemented in existing transceiver without requiring any modifications of the hardware, which is advantageous in some scenarios where the test signals are allowed to be transmitted over the air. As mentioned, the difference in phase shift ∆^^may be provided in different ways. The method 1200 may comprise introducing the difference in phase shift ∆^^by providing a first signal path 602 from the transmitter 310 to the receiver 340 when injecting the first test signal ^^^^1, and by providing a second signal path 603 from the transmitter 310 to the receiver 340 when injecting the second test signal ^^^^2, where the first signal path 602 provides a different phase delay compared to the second signal path 603. If e.g., two transmission lines or two waveguides are used to provide the first signal path 602 and the second signal path 603, one of the two transmission lines or two waveguides may have a longer electrical length compared to the other of the two transmission lines or two waveguides. If, on the other hand, one transmission line or one waveguide is used, the transmission line or the waveguide may be provided by a variable phase shifter. The modulator 320 and the demodulator 350 may be driven by a local oscillator (LO) signal with an LO frequency. In some embodiments, both the modulator 320 and the demodulator 350 are driven by the same LO. In other embodiments, the modulator 320 and the demodulator 350 are driven by respective LOs. In any case, the difference in phase shift ∆^^is, according to some embodiments, different from 0° and ±180° relative to the LO frequency. The difference phase shift ∆^^preferably is 45° to 135° or –135° to –45°, more preferably 60° to 120° or –120° to –60°, and even more preferably 80° to 100° or –100° to –80°, relative to the LO frequency. Although the difference in phase shift ∆^^preferably is ±90° and frequency independent, some tolerance is acceptable. Different values than the accuracy of the estimates of the first and the second compensation signals. However, if the difference in phase shift ∆^^is 45° to 135°, preferably 60° to 120°, and more preferably 80° to 100°, or –135° to –45°, preferably –120° to –60°, and more preferably –100° to –80°, acceptable accuracies are obtained. Another way to introduce the difference in phase shift ∆^^is by means of an LO. The method 1200 may comprise introducing the difference in phase shift ∆^^by driving the modulator 320 and the demodulator 350 with the LO signal with a first LO phase when injecting the first test signal ^^^^1, and by driving the modulator and the demodulator 350 with the LO signal with a second LO phase when injecting the second test signal ^^^^2, wherein the first LO phase is shifted relative to the second LO phase. In this way, a single loopback path may be used for the first and the second test signals ^^^^1, ^^^^2(such as a transmission line or waveguide according to the discussions above, or the propagation path according to the discussions above). The different phases of the LO signal may be provided by means of a phase locked loop. As another example, the difference in phase shift ∆Φmay be provided by means of a phase shifter in the signal path, such as a phase shifter used for analog beamforming.Each of the first and the second test signals ^^^^1, ^^^^2 may have a peak to average power ratio (PAPR) ofless than 5 dB, preferably less than 4 dB, and more preferably less than 3 dB. Having such values of the PAPR reduces non-linear distortion of injected signals, which improves accuracy the estimated linear and conjugate-linear frequency characteristics. The lower the PAPR, the better. However, in practice, finding a signal sequence with a PAPR much lower than 3 dB may be challenging. As an example, a punctured Zadoff–Chu signal sequence can be configured such that it has a PAPR of less than 3 dB. The transceiver 300 may be part of a wireless communications network 100 performing Time-Division Duplex (TDD)wireless transmissions, where each of the first and second test signals ^^^^1, ^^^^2 are injected between uplink (UL)and downlink (DL) time periods in the wireless communications network 100. In this way, live traffic is not affected. Consequently, the first and / or the second compensation signals may be estimated without reducing capacity of the wireless communications network 100. As an example, the wireless communications network 100 is a new radio(NR) system, and both the first and the second test signals ^^^^1, ^^^^2 are injected at empty resources aroundSynchronization Signal Block (SSB) transmission.Action 1220. The method 1200 comprises obtaining linear frequency characteristics (^^^^1[^^], ^^^^1[−^^]) andconjugate-linear frequency characteristics (^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver 300 in the first loopback configuration and linear frequency characteristics (^^^^2[^^],^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver 300 in the second loopback configuration.The linear frequency characteristics (^^^^1[^^], ^^^^1[−^^]) of the transceiver 300 in the first loopback configurationare characteristics of the transceiver the frequency components of ^^^^1as injected to frequency content at the same frequencies (i.e., the same frequencies as the frequency of ^^^^1as injected) of ^^^^1as received from the receiver 340, when the transceiver 300 is configured in the first loopback configuration. For example, ^^^^1[^^]maps ^^^^1[^^]to ^^^^1[^^]and ^^^^1[−^^]maps ^^^^1[−^^]to ^^^^1[−^^]when the transceiver 300 is in the first loopback configuration, ^^^^1is ^^^^1as received from the receiver 340. The conjugate-linear frequency characteristics (^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver 300 in the first loopback configuration is a characteristics of the transceiver 300 that map the frequency components of ^^^^1as injected to frequency content at corresponding mirror frequencies, i.e., conjugate frequencies, (i.e. mirror frequencies to the frequency components of ^^^^1as injected) of ^^^^1as received from the receiver 340, when the transceiver 300 is configured in the first loopback configuration. For example, ^̃^^^1[^^]maps ^^^^1[−^^]to ^^^^1[^^]and ^̃^^^1[−^^]maps ^^^^1[^^]to ^^^^1when the transceiver 300 is configured in the first loopback configuration, where ^^^^1is ^^^^1as receiver 340. The linear frequency characteristics (^^^^2[^^],^^^^2[−^^]) of the transceiver 300 in the second loopback configuration are characteristics of the transceiver 300 that map the frequency components of ^^^^2as injected to frequency content at the same frequencies (i.e., same frequencies as the frequency components of ^^^^2as injected) of ^^^^2as received from the receiver 340, when the transceiver 300 is configured in the second loopback configuration. For example, ^^^^2[^^]maps ^^^^2[^^]to ^^^^2[^^]and ^^^^2[−^^]maps ^^^^2[−^^]to ^^^^2[−^^]when the transceiver 300 is configured in the second loopback configuration, where ^^^^2is ^^^^2as received from the receiver 340. The linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver 300 in the second loopback configuration are characteristics of the transceiver 300 that map the frequency components of ^^^^2as injected to frequency content at corresponding mirror frequencies, i.e., conjugate frequencies, (i.e. mirror frequencies to the frequency components of ^^^^2as injected) of ^^^^2as received from the receiver 340, when the transceiver 300 is configured in the second loopback configuration. For example, ^̃^^^2[^^]maps ^^^^2[−^^]to ^^^^2[^^]and ^̃^^^2[−^^]maps ^^^^2[^^]to ^^^^2[−^^]when the transceiver 300 is configured in the second loopback configuration, where ^^^^2is ^^^^2as received from the receiver 340.The linear frequency characteristics (^^^^1[^^], ^^^^1[−^^]) and the conjugate-linear frequency characteristics(^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver 300 in the first loopback configuration may be obtained at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^. The linear frequency characteristics (^^^^2[^^], ^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver 300 in the second loopback configuration may be obtained at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^. As mentioned, the frequency characteristics for the first and the second loopback configurations at −^^ and at −^^ may be obtained in different ways. Actions 1221 and 1222. The linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] may be obtained by: injecting ^^^^1three times with different spectral content at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^ each time (here, the transceiver is in the first loopback configuration during the three times) and of the second test signal ^^^^2three times with different spectral content at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^ each time (here, the transceiver is in the second loopback configuration during the three times); and estimating 1222 the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the firstbaseband frequency and at the mirror frequency of the first baseband frequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^],^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] based on the first and the ^^^^2 asinjected into the transmitter 310 and as received from the receiver 340 the three times. In this way, the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] may be obtained without any Here, the first test signal ^^^^1is injected three times with different spectral content at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^ each time. This means that the magnitude and / or phase at ^^ and at −^^ is different for each of the three times. The same applies for the second test signal ^^^^2. Spectral content at other frequencies than at ^^ and at −^^ may remain the same for each of the three times. Action 1223. The method 1200 may comprise obtaining the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirrorfrequency of the second baseband frequency [^^] ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]. In that case, the linear and the for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first basebandfrequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] may be estimatedfrom the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the second baseband frequency and at the mirror frequency of the second baseband frequency [^^] [−^^]. The frequency characteristics at ^^ and at −^^ may be estimated from obtained frequency characteristics at ^^ and at −^^ in various ways. For example, by assuming the same value of the frequency characteristics at ^^ and at ^^ if ^^ and ^^ are adjacent. It is also possible to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at many different frequencies different from ^^ and at −^^. In that case, it is possible to estimate the frequency characteristics at ^^ and at −^^ using interpolation or extrapolation. As mentioned, the first baseband frequency may represent at FFT bin. Similarly, the second baseband frequency may represent a different FFT bin from the FFT bin of the first baseband frequency. The frequency characteristics at frequencies different from ^^ and at −^^ may, e.g., be obtained directly from thefirst and the second test signals ^^^^1, ^^^^2 if the first and the second test signals comprise spectral content atcorresponding frequencies (different from ^^ and at −^^). In this way, the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirrorfrequency of the first baseband frequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] may be obtained with few experiments (e.g. only two experiments, i.e., signal ^^^^1once and injecting the second test signal ^^^^2once).The first and the second test signals ^^^^1, ^^^^2 may comprise respective known non-zero spectral content at thesecond baseband frequency ^^ and at the mirror frequency of the second baseband frequency −^^. In that case, the method may comprise Action 1224. Action 1224. The obtaining the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second basebandfrequency [^^] ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] may comprise: estimating1224 the characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency^^^^1[^^], ^^^^1[−^^], ^^^^2[^^], ^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] based on the first and the secondtest signals ^^ ^^1, transmitter 310 and as received from the receiver 340.Action 1230. The method 1200 comprises estimating, based on the first and second test signals ^^^^1, ^^^^2 asinjected into the transmitter 310 and as received from the receiver 340 and on the obtained linear and conjugate-linear frequency characteristics for the first and the second loopback configurations (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^],^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]), a first compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a transmission by the transmitter 310 and / or a second compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a reception by the receiver 340. The first and the second compensation signals may also be called first and second compensation values, respectively, or first and second correction values, respectively. The first and the second compensation signals may, e.g., be added in the time domain or the frequency domain during operation of the transceiver. In particular, the first compensation signal may be added to a third signal, adapted for carrying traffic or user data, at the input of the transmitter 310 to suppress a spurious tone during the transmission of the third signal. The second compensation signal may be added to a fourth signal, adapted for carrying traffic or user data, at the output of the receiver 340 to suppress a spurious tone during the reception of the third signal. Note that it is possible to estimate compensation signals with and without compensation filters comprised in the transceiver. With the compensation filters, it is possible to apply a compensation signal after the compensation filter or before the compensation filter (according to the propagation path of the signal). The first and the second compensation values may be estimated in an iterative fashion, where an updated value of a compensation signal is based on current measurements and a previous value of the same compensation signal. The first compensation signal (^^^^^^[^^],^^^^^^[−^^]) may be estimated at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ for suppression of a spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a transmission by the transmitter 310. The second compensation signal (^^^^^^[^^],^^^^^^[−^^]) may be estimated at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^, for suppression of a spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a reception by the receiver 340. Known ways of estimating compensation signals for suppressing spurious tones often contain many steps and often take significant time. Furthermore, in some known ways of estimation compensation signals, the transceiver needs to be put in calibration mode (e.g., powering down a low noise amplifier) during the estimation such that traffic cannot be upheld during the estimation. In general, it is desired to repeat the estimation concurrently during operation of the transceiver (since spurious tones may vary with, e.g., temperature). The method 1200 enables estimation of compensation signals for suppressing spurious tones quickly and with relatively few calculations. Furthermore, the method 1200 can be performed with low or no disruption of traffic. In the method 1200, compensation signals for multiple frequencies (i.e., for compensating for spurious tones at multiple frequencies) can be estimated simultaneously. Consequently, the disclosed method saves time and resources, which may further reduce the impact on traffic. The estimation of the compensation signals in the disclosed method can be based on the same type of test signals / experiments used for estimation frequency-dependent IQ (FDIQ) imbalance. Consequently, the disclosed method saves time and resources, which may further reduce the impact on traffic. In transceivers with an analog interface between RFICs and converters, the compensation of spurious tones relaxes isolation requirements between the reference clock and the analogue interface, and thereby enable integration of RF-PLL into the RFIC without extreme strict isolation requirement between the PLL reference clock (or other reference clocks in the system) and the analog interface. In other words, the method 1200 enables the use of an analog interface between RFIC and ADC / DAC, which consequently enables an architecture that integrates ADC / DAC and DFE in respective circuit technologies, which in turn leads to cost and energy savings. The linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations may be respective frequency characteristic of the transceiver 300 comprising a transmitter compensation filter 411 at the input of the transmitter and a receiver compensation filter 442 at the output of the receiver 340. In this way, the spurious tones can be estimated with compensation for FDIQ imbalance applied. The spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a transmission by the transmitter 310 and / or the spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a reception by the receiver 340 may represent any of: LO leakage, DC offset, and leakage from a reference clock of the transceiver 300. The method 1200 may further comprise Actions 1241-1244. Action 1241. The method 1200 may comprise transmitting a third signal by the transmitter 310 and / or receiving 1242 a fourth signal by the receiver 340, where the third and the fourth signals may be adapted for carrying traffic or user data. Action 1243. The method 1200 may comprise adding the first compensation signal ^^^^^^[^^],^^^^^^[−^^] to the third signal at the input of the transmitter 310 to suppress a spurious tone during the transmission of the third signal. Action 1244. The method 1200 may comprise adding the second compensation signal ^^^^^^[^^],^^^^^^[−^^] to the fourth at the output of the receiver 340 to suppress a spurious tone during the reception of the fourth signal. In this way, an improved transceiver is provided, which in turn enables an improved wireless communications system. As mentioned, the compensation signals may, e.g., be added in the time domain or the frequency domain during operation of the transceiver 300. For OFDMA based system (such as LTE, NR and potentially also 6G), a complex constant can be added to the affected sub-carrier that will cancel out (or at least suppress) the spurious tone (which may originate in time domain, in e.g., the analog interface between RFIC and DAC / ADC). In the time domain, then NCOs may be used to cancel (or at least suppress) out the spurious tone. Figure 13 – Schematic block diagram of embodiments of an apparatus Figure 13 shows a schematic block diagram of embodiments of an apparatus 110, 121 for enabling suppression of spurious tones in a transceiver 300. The apparatus 110, 121 may comprise the transceiver 300. Furthermore, the apparatus may, e.g., be a network node 110 or a wireless device 121. The transceiver 300 comprises the transmitter 310 provided with the modulator 320 and the receiver 340 provided with a demodulator 350. The transceiver 300 is adapted for being put in the first and the second loopback configuration, respectively, in which respective signals injected to the transmitter 310 are modulated by the modulator 320, looped back to the receiver 340 with a difference in phase shift ∆^^, and thereafter demodulated by the demodulator 350. The transceiver 300 may be for wireless communications. The transceiver 300 may be homodyne transceiver. In some embodiments, the modulator 320 is a quadrature modulator 1120 and wherein the demodulator 350 is a quadrature demodulator 1150. The embodiments of the apparatus 110, 121 may be considered as independent embodiments or may be considered in any combination with each other. It should also be noted that, although not shown in Figure 13, the apparatus 110, 121 may comprise known conventional features for such devices, such as a power source like a battery or mains connection. The transmitter 310 may comprises a digital-to-analog converter (DAC) physically separated from the modulator 320, and the receiver 340 may comprise an analog-to-digital converter (ADC) physically separated from the demodulator 350. Furthermore, there may be an analog interface between the DAC and the modulator 320, and there may be an analog interface between the ADC and the demodulator 350. The DAC may be implemented in a different circuit technology than the modulator 320, and the ADC may be implemented in a different circuit technology than the demodulator 350. The transceiver 300 may comprise a fist part of a first circuit technology and a second part of a second circuit technology, where in the DAC and ADC are implemented on the first part and the modulator and demodulator are implemented on the second part. The apparatus 110, 121 may comprise processing circuitry 1310 and a memory 1320. The transceiver 300 may comprise radio frequency circuitry and baseband processing circuitry capable of transmitting and receiving a radio signal in the wireless communications network 100. It should also be noted that some or all the functionality described in the embodiments above as being performed by the apparatus110, 121 may be provided by the processing circuitry 1310 executing instructions stored on a computer-readable medium, such as, e.g., the memory 1320 shown in Figures 13. Alternative embodiments of the apparatus 110, 121 may comprise additional components, such as, an injecting module 1311, obtaining module 1312, and / or an estimating module 1313, responsible for providing functionality to support the embodiments of the apparatus 110, 121 described herein. The apparatus 110, 121, the processing circuitry 1310, or the injecting module 1311 is configured to initiate injection of a first test signal ^^^^1and a second test signal ^^^^2into the transmitter 310 with the transceiver 300 being in the first and the second loopback configurations, respectively. The apparatus 110, 121, the processing circuitry 1310,or the obtaining module 1312 is configured to obtain linear frequency characteristics ^^^^1[^^], ^^^^1[−^^] and conjugate-linear frequency characteristics ^̃^^^1[^^], ^̃^^^1[−^^] of the transceiver configuration and linear frequency characteristics ^^^^2[^^],^^^^2[−^^] and conjugate-linear frequency characteristics ^̃^^^2[^^], ^̃^^^2[−^^]of the transceiver 300 in the second loopback configuration. The apparatus 110, 121, the processing circuitry 1310, or the estimating module 1313 is configured to estimate, based on the first andsecond test signals ^^^^1, ^^^^2 as injected into the transmitter 310 and as received from the receiver 340 and on theobtained linear and conjugate-linear frequency characteristics for the first and the second loopback configurations^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^], a first compensation signal ^^^^^^[^^],^^^^^^[−^^] for suppression of a spurious tone during a transmission by the transmitter 310 and / or a second compensation signal ^^^^^^[^^],^^^^^^[−^^] for suppression of a spurious tone during a reception by the receiver 340.In some embodiments, the first and the second test signals ^^^^1, ^^^^2 comprise respective known spectral contentat a first baseband frequency ^^ and at a mirror frequency of the first baseband frequency −^^.In some embodiments, the linear frequency characteristics ^^^^1[^^], ^^^^1[−^^] and the conjugate-linear frequencycharacteristics ^̃^^^1[^^], ^̃^^^1[−^^] of the transceiver 300 in the first loopback configuration are obtained at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^, and wherein the linear frequency characteristics ^^^^2[^^], ^^^^2[−^^] and conjugate-linear frequency characteristics ^̃^^^2[^^], ^̃^^^2[−^^]of the transceiver 300 in the second loopback configuration are obtained at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^. In some embodiments, the first compensation signal ^^^^^^[^^],^^^^^^[−^^] is estimated at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ for suppression of a spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a transmission by the transmitter 310, and wherein the second compensation signal ^^^^^^[^^],^^^^^^[−^^] is estimated at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^, for suppression of a spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a reception by the receiver 340.In some embodiments, for the first and the second test signals ^^^^1, ^^^^2, the respective known spectral content atthe first baseband frequency −^^ and at the mirror frequency of the first baseband frequency −^^ is zero. In some embodiments, the first baseband frequency is at DC. In other embodiments, the first baseband frequency is non-zero. In some embodiments, the apparatus 110, 121, the processing circuitry 1310, or the obtaining module 1312 is configured to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency [^^] [−^^] by: initiating injection of the first test signal ^^^^1three times with different spectral content at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^ each time, and of the second test signal ^^^^2three times with different spectral content at the first baseband frequency ^^ and at the mirror frequency of the first baseband frequency −^^ each time; and estimating the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^] based on the first and thesecond test signals ^^^^1, ^^^^2 as injected into the transmitter 310 and as received from the receiver 340 the threetimes. In some embodiments, the apparatus 110, 121, the processing circuitry 1310, or the obtaining module 1312 is configured to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second basebandfrequency [^^] [−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]. In that case, the linear andthe for the first and the second loopback configurations at the firstbaseband frequency and at the mirror frequency of the first baseband frequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^],^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]are estimated from the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the second baseband frequencyand at the mirror frequency of the second baseband frequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^].In some embodiments, the first and the second test signals ^^^^1, ^^^^2 comprise respective known non-zerospectral content at the second baseband frequency ^^ and at the mirror frequency of the second baseband frequency −^^, and the apparatus 110, 121, the processing circuitry 1310, or the obtaining module 1312 is configured to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations ata second baseband frequency and at a mirror frequency of the second baseband ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]by: estimating the frequency characteristics for the first and the second loopback configurations at a second baseband frequency andat a mirror frequency of the second baseband frequency ^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^], ^^^^2[−^^],^̃^^^2[^^], ^̃^^^2[−^^] based on the first and the second test signals ^^^^1, ^^^^2 as injected into the transmitter 310and as received from the receiver 340. In some embodiments, the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations are respective frequency characteristic of the transceiver 300 comprising a transmitter compensation filter 411 at the input of the transmitter and a receiver compensation filter 442 at the output of the receiver 340. In some embodiments, the first baseband frequency represents an FFT bin. In some embodiments, the spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a transmission by the transmitter 310 and / or the spurious tone at the first baseband frequency ^^ and / or at the mirror frequency of the first baseband frequency −^^ during a reception by the receiver 340 represents any of: LO leakage, DC offset, and leakage from a reference clock of the transceiver 300. In some embodiments, each of the first and the second test signals ^^^^1, ^^^^2comprise a plurality of primary frequency components, wherein respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are below a threshold, and wherein a subset of the plurality of primary frequency components is above DC and the remainder of the plurality of primary frequency components is below DC. In some embodiments, the threshold is relative to any of the frequency components of the plurality of primary frequency components. In some embodiments, the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are zero. In some embodiments, the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are below 20 dB, preferably below 30 dB, and more preferably below 40 dB, relative to any of the frequency components of the plurality of primary frequency components. In some embodiments, the plurality of primary frequency components comprises at least five frequency components, and preferably at least ten frequency components. In some embodiments, the modulator 320 and the demodulator 350 are driven by a local oscillator, LO, signal with an LO frequency, and wherein the difference in phase shift ∆^^is different from 0° and ±180° relative to the LO frequency, and wherein the difference in phase shift (∆^^) preferably is 45° to 135° or –135° to –45°, more preferably 60° to 120° or –120° to –60°, and even more preferably 80° to 100° or –100° to –80°, relative to the LO frequency. In some embodiments, the transceiver 300 is part of a wireless communications network 100 performing Time-Division Duplex (TDD) wireless transmissions, wherein each of the first and second test signals ^^^^1, ^^^^2 areinjected between uplink (UL) and downlink (DL) time periods in the wireless communications network In some embodiments, the apparatus 110, 121 or the processing circuitry 1310, 1311 is configured to initiate transmission of a third signal with the transmitter 310 and / or reception of a fourth signal with the receiver 340, wherein the third and the fourth signals may be adapted for carrying traffic or user data; add the first compensation signal ^^^^^^[^^],^^^^^^[−^^] to the third signal at the input of the transmitter 310 to suppress a spurious tone during the signal; and / or add the second compensation signal ^^^^^^[^^],^^^^^^[−^^] to the fourth at the output of the receiver 340 to suppress a spurious tone during the reception of the fourth signal. The methods disclosed herein may be implemented through one or more processors, such as the processing circuitry 1310 in the apparatus 110, 121 depicted in Figure 13, together with computer program code for performing the functions and actions of the embodiments herein. The computer program code may also be provided as a computer program product, for instance in the form of a data carrier carrying computer program code or code means for performing the embodiments herein when being loaded into the processing circuitry 1310 in the apparatus 110, 121. The computer program code may, e.g., be provided as pure program code in the apparatus 110, 121 or on a server and downloaded to the apparatus 110, 121. Thus, it should be noted that the modules of the apparatus 110, 121 may in some embodiments be implemented as computer programs stored in memory, e.g., in the memory modules 1320 in Figure 13, for execution by processors or processing modules, e.g., the processing circuitry 1310 of Figure 13. Those skilled in the art will also appreciate that the processing circuitry 1310 and the memory 1320 described above may refer to a combination of analog and digital circuits, and / or one or more processors configured with software and / or firmware, e.g., stored in a memory, that when executed by the one or more processors such as the processing circuitry 1310 perform as described above. One or more of these processors, as well as the other digital hardware, may be included in a single application-specific integrated circuit (ASIC), or several processors and various digital hardware may be distributed among several separate components, whether individually packaged or assembled into a system-on-a-chip (SoC). The description of the example embodiments provided herein have been presented for purposes of illustration. The description is not intended to be exhaustive or to limit example embodiments to the precise form disclosed, and modifications and variations are possible in light of the above teachings or may be acquired from practice of various alternatives to the provided embodiments. The examples discussed herein were chosen and described to explain the principles and the nature of various example embodiments and its practical application to enable one skilled in the art to utilize the example embodiments in various manners and with various modifications as are suited to the particular use contemplated. The features of the embodiments described herein may be combined in all possible combinations of methods, apparatus, modules, systems, computer programs, and computer program products. It should be appreciated that the example embodiments presented herein may be practiced in any combination with each other. It should be noted that the word “comprising” does not necessarily exclude the presence of other elements or steps than those listed and the words “a” or “an” preceding an element do not exclude the presence of a plurality of such elements. It should further be noted that any reference signs do not limit the scope of the claims, that the example embodiments may be implemented at least in part by means of both hardware and software, and that several “means”, “units” or “devices” may be represented by the same item of hardware. It should also be noted that the various example embodiments described herein are described in the general context of method steps or processes, which may be implemented in one aspect by a computer program product, embodied in a computer-readable medium, including computer-executable instructions, such as program code, executed by computers in networked environments. A computer-readable medium may include removable and non-removable storage devices including, but not limited to, Read Only Memory (ROM), Random Access Memory (RAM), compact discs (CDs), digital versatile discs (DVD), etc. Generally, program modules may include routines, programs, objects, components, data structures, etc. that perform particular tasks or implement particular abstract data types. Computer-executable instructions, associated data structures, and program modules represent examples of program code for executing steps of the methods disclosed herein. The particular sequence of such executable instructions or associated data structures represents examples of corresponding acts for implementing the functions described in such steps or processes. The embodiments herein are not limited to the above-described preferred embodiments. Various alternatives, modifications and equivalents may be used. Therefore, the above embodiments should not be construed as limiting.

Claims

CLAIMS 1. A method (1200) for enabling suppression of spurious tones in a transceiver (300), the transceiver (300) comprising a transmitter (310) having a modulator (320) and a receiver (340) having a demodulator (350), the transceiver (300) being adapted for being put in a first and a second loopback configuration, respectively, in which respective signals injected to the transmitter (310) are modulated by the modulator (320), looped back to the receiver (340) with a difference in phase shift (∆^^), and thereafter demodulated by the demodulator (350), the method comprising: injecting (1210) a first test signal (^^^^1) and a second test signal (^^^^2) into the transmitter (310) with the transceiver (300) being in the first and the second loopback configurations, respectively;obtaining (1220) linear frequency characteristics (^^^^1[^^], ^^^^1[−^^]) and conjugate-linear frequencycharacteristics (^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver (300) in the first loopback configuration and linear frequency characteristics (^^^^2[^^],^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver (300) in the second loopback configuration; andestimating (1230), based on the first and second test signals (^^^^1, ^^^^2) as injected into the transmitter (310) andas received from the receiver (340) and on the obtained linear and conjugate-linear frequency characteristics forthe first and the second loopback configurations (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^],^̃^^^2[^^], ^̃^^^2[−^^]), a first compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a transmission by the transmitter (310) and / or a second compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a reception by the receiver (340).

2. The method (1200) according to claim 1, wherein the first and the second test signals (^^^^1, ^^^^2)comprise respective known spectral content at a first baseband frequency (^^) and at a mirror frequency of the first baseband frequency (−^^).

3. The method (1200) according to claim 2, wherein the linear frequency characteristics(^^^^1[^^], ^^^^1[−^^]) and the conjugate-linear frequency characteristics (^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver(300) in the first loopback configuration are obtained at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^), and wherein the linear frequency characteristics(^^^^2[^^], ^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver (300) in the second loopback configuration are obtained at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^).

4. The method (1200) according to claim 3, wherein the first compensation signal (^^^^^^[^^],^^^^^^[−^^]) is estimated at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) for suppression of a spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a transmission by the transmitter (310), and wherein the second compensation signal (^^^^^^[^^],^^^^^^[−^^]) is estimated at the first baseband frequency (^^) and / or at the mirror frequency of thefirst baseband frequency (−^^), for suppression of a spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a reception by the receiver (340).

5. The method (1200) according to any of claims 2-4, wherein, for the first and the second test signals(^^^^1, ^^^^2 ), the respective known spectral content at the first baseband frequency (−^^) and at the mirrorfrequency of the first baseband frequency (−^^) is zero.

6. The method (1200) according to any of claims 2-5, wherein the first baseband frequency is at DC.

7. The method (1200) according to any of claims 2-5, wherein the first baseband frequency is non-zero.

8. The method (1200) according to claim 3, wherein the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirrorfrequency of the first baseband frequency (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^],^̃^^^2[^^], ^̃^^^2[−^^]) are obtained by: injecting (1221) the first test signal (^^^^1) three times with different spectral content at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^) each time, and of the second test signal (^^^^2) three times with different spectral content at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^) each time; and estimating (1222) the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) based on the first and thesecond test signals ( ^^1, ^^2) as injected into the transmitter (310) and as received from the receiver (340) thethree times.

9. The method (1200) according to claim 3 or any of claims 4-7 when dependent on claim 3, comprising obtaining (1223) the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]),wherein the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) are estimated from the linearand the conjugate-linear frequencythe second loopback configurations at the secondbaseband frequency and at the mirror frequency of the second baseband frequency (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^],^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]).

10. The method (1200) according to claim 9, wherein the first and the second test signals (^^^^1, ^^^^2)comprise respective known non-zero spectral content at the second baseband frequency (^^) and at the mirror frequency of the second baseband frequency (−^^), and wherein obtaining the linear and the conjugate-linearfrequency characteristics for the first and the second loopback configurations at a second baseband frequency andat a mirror frequency of the second baseband frequency [^^]^^ [^^],^^ [−^^], ^̃^ [^^^2 ^^2 ^^2^], ^̃^^^2[−^^]) comprises: estimating (1224) the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^], ^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) based on the first and the secondtest sgnas ( ^^1, ^^2) as njected nto the transmitter (310) and as received from the receiver (340).

11. The method (1200) according to any previous claim, wherein the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations are respective frequency characteristic of the transceiver (300) comprising a transmitter compensation filter (411) at the input of the transmitter and a receiver compensation filter (442) at the output of the receiver (340).

12. The method (1200) according to any previous claim, wherein the first baseband frequency represents an FFT bin.

13. The method (1200) according to claim 4 or any previous claim when dependent on claim 4, wherein the spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a transmission by the transmitter (310) and / or the spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a reception by the receiver (340) represents any of: LO leakage, DC offset, and leakage from a reference clock of the transceiver (300).

14. The method (1200) according to any previous claim, wherein the transmitter (310) comprises a digital-to- analog converter physically separated from the modulator (320), and wherein the receiver (340) comprises an analog-to-digital converter physically separated from the demodulator (350).

15. The method (1200) according to any previous claim, wherein each of the first and the second test signals (^^^^1, ^^^^2) comprise a plurality of primary frequency components, wherein respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are below a threshold, and wherein a subset of the plurality of primary frequency components is above DC and the remainder of the plurality of primary frequency components is below DC.

16. The method (1200) according to claim 15, wherein the threshold is relative to any of the frequency components of the plurality of primary frequency components.

17. The method (1200) according to claim 15 or 16, wherein the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are zero.

18. The method (1200) according to claim 15 or 16, wherein the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components arebelow 20 dB, preferably below 30 dB, and more preferably below 40 dB, relative to any of the frequency components of the plurality of primary frequency components.

19. The method (1200) according to any of claims 15-18, wherein the plurality of primary frequency components comprises at least five frequency components, and preferably at least ten frequency components.

20. The method (1200) according to any previous claim, wherein the modulator (320) and the demodulator (350) are driven by a local oscillator, LO, signal with an LO frequency, and wherein the difference in phase shift (∆^^) is different from 0° and ±180° relative to the LO frequency, and wherein the difference in phase shift (∆^^) preferably is 45° to 135° or –135° to –45°, more preferably 60° to 120° or –120° to –60°, and even more preferably 80° to 100° or –100° to –80°, relative to the LO frequency.

21. The method (1200) according to any previous claim, wherein the transceiver (300) is part of a wireless communications network (100) performing Time-Division Duplex, TDD, wireless transmissions, wherein each of thefirst and second test signals (^^^^1, ^^^^2) are injected between uplink, UL, and downlink, DL, time periods in thewireless communications network (100).

22. The method (1200) according to any previous claim, wherein the modulator (320) is a quadrature modulator (1120) and wherein the demodulator (350) is a quadrature demodulator (1150).

23. The method (1200) according to any previous claim, further comprising: transmitting (1241) a third signal by the transmitter (310) and / or receiving (1242) a fourth signal by the receiver (340), wherein the third and the fourth signals may be adapted for carrying traffic or user data; adding (1243) the first compensation signal (^^^^^^[^^],^^^^^^[−^^]) to the third signal at the input of the transmitter (310) to suppress a spurious tone during the transmission of the third signal; and / or adding (1244) the second compensation signal (^^^^^^[^^],^^^^^^[−^^]) to the fourth at the output of the receiver (340) to suppress a spurious tone during the reception of the fourth signal.

24. An apparatus (110, 121) for enabling suppression of spurious tones in a transceiver (300), the transceiver (300) comprising a transmitter (310) having a modulator (320) and a receiver (340) having a demodulator (350), the transceiver (300) being adapted for being put in a first and a second loopback configuration, respectively, in which respective signals injected to the transmitter (310) are modulated by the modulator (320), looped back to the receiver (340) with a difference in phase shift (∆^^), and thereafter demodulated by the demodulator (350), the apparatus (110, 121) comprising a processing circuitry (1310) configured to: initiate injection of a first test signal (^^^^1) and a second test signal (^^^^2) into the transmitter (310) with the transceiver (300) being in the first and the second loopbackobtain linear frequency characteristics (^^^^1[^^], ^^^^1[−^^]) and conjugate-linear frequency characteristics(^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver (300) in the first loopback configuration and linear frequency characteristics(^^^^2[^^],^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver (300) in the second loopback configuration; andestimate, based on the first and second test signals (^^^^1, ^^^^2) as injected into the transmitter (310) and asreceived from the receiver (340) and on the obtained linear and conjugate-linear frequency characteristics for thefirst and the second loopback configurations (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^],^̃^^^2[^^], ^̃^^^2[−^^]), a first compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a transmission by the transmitter (310) and / or a second compensation signal (^^^^^^[^^],^^^^^^[−^^]) for suppression of a spurious tone during a reception by the receiver (340).

25. The apparatus (110, 121) according to claim 24, wherein the first and the second test signals (^^^^1, ^^^^2)comprise respective known spectral content at a first baseband frequency (^^) and at a mirror frequency of the first baseband frequency (−^^).

26. The apparatus (110, 121) according to claim 25, wherein the linear frequency characteristics(^^^^1[^^], ^^^^1[−^^]) and the conjugate-linear frequency characteristics (^̃^^^1[^^], ^̃^^^1[−^^]) of the transceiver(300) in the first loopback configuration are obtained at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^), and wherein the linear frequency characteristics (^^^^2[^^], ^^^^2[−^^]) and conjugate-linear frequency characteristics (^̃^^^2[^^], ^̃^^^2[−^^]) of the transceiver (300) in the second loopback configuration are obtained at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^).

27. The apparatus (110, 121) according to claim 26, wherein the first compensation signal (^^^^^^[^^],^^^^^^[−^^]) is estimated at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) for suppression of a spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a transmission by the transmitter (310), and wherein the second compensation signal (^^^^^^[^^],^^^^^^[−^^]) is estimated at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^), for suppression of a spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a reception by the receiver (340).

28. The apparatus (110, 121) according to any of claims 25-27, wherein, for the first and the second testsignals (^^^^1, ^^^^2 ), the respective known spectral content at the first baseband frequency (−^^) and at the mirrorfrequency of the first baseband frequency (−^^) is zero.

29. The apparatus (110, 121) according to any of claims 25-28, wherein the first baseband frequency is at DC.

30. The apparatus (110, 121) according to any of claims 25-28, wherein the first baseband frequency is non- zero.

31. The apparatus (110, 121) according to claim 26, wherein the processing circuitry (1310) is configured to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) by:initiating injection of the first test signal (^^^^1) three times with different spectral content at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^) each time, and of the second test signal (^^^^2) three times with different spectral content at the first baseband frequency (^^) and at the mirror frequency of the first baseband frequency (−^^) each time; and estimating the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) based on the first and thesecond test sgnas ( ^^1, ^^2) as njected nto t e transmtter (310) and as receved from the receiver (340) thethree times.

32. The apparatus (110, 121) according to claim 26 or any of claims 29-30 when dependent on claim 26, wherein the processing circuitry (1310) is configured to: obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]),wherein the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at the first baseband frequency and at the mirror frequency of the first baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) are estimated from the linearand the conjugate-linear frequency characteristics for the first and the second loopback configurations at the secondbaseband frequency and at the mirror frequency of the second baseband frequency (^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^],^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]).

33. The apparatus (110, 121) according to claim 32, wherein the first and the second test signals (^^^^1, ^^^^2)comprise respective known non-zero spectral content at the second baseband frequency (^^) and at the mirror frequency of the second baseband frequency (- ^^), and wherein the processing circuitry (1310) is configured to obtain the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^],^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) by:estimating the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations at a second baseband frequency and at a mirror frequency of the second baseband frequency(^^^^1[^^], ^^^^1[−^^], ^̃^^^1[^^], ^̃^^^1[−^^], ^^^^2[^^], ^^^^2[−^^], ^̃^^^2[^^], ^̃^^^2[−^^]) based on the first and the secondtest sgnas ( ^^1, ^^2) as njected nto the transmitter (310) and as received from the receiver (340).

34. The apparatus (110, 121) according to any of claims 24-33, wherein the linear and the conjugate-linear frequency characteristics for the first and the second loopback configurations are respective frequency characteristic of the transceiver (300) comprising a transmitter compensation filter (411) at the input of the transmitter and a receiver compensation filter (442) at the output of the receiver (340).

35. The apparatus (110, 121) according to any of claims 24-34, wherein the first baseband frequency represents an FFT bin.

36. The apparatus (110, 121) according to claim 27 or any of claims 28-35 when dependent on claim 27, wherein the spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a transmission by the transmitter (310) and / or the spurious tone at the first baseband frequency (^^) and / or at the mirror frequency of the first baseband frequency (−^^) during a reception by the receiver (340) represents any of: LO leakage, DC offset, and leakage from a reference clock of the transceiver (300).

37. The apparatus (110, 121) according to any of claims 24-36, wherein the transmitter (310) comprises a digital-to-analog converter physically separated from the modulator (320), and wherein the receiver (340) comprises an analog-to-digital converter physically separated from the demodulator (350).

38. The apparatus (110, 121) according to any of claims 24-37, wherein each of the first and the second test signals (^^^^1, ^^^^2) comprise a plurality of primary frequency components, wherein respective magnitudes of spectral mirror frequencies of each frequency component of the plurality of primary frequency components are below a threshold, and wherein a subset of the plurality of primary frequency components is above DC and the remainder of the plurality of primary frequency components is below DC.

39. The apparatus (110, 121) according to claim 38, wherein the threshold is relative to any of the frequency components of the plurality of primary frequency components.

40. The apparatus (110, 121) according to claim 38 or 39, wherein the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are zero.

41. The apparatus (110, 121) according to claim 38 or 39, wherein the respective magnitudes of spectral content at respective mirror frequencies of each frequency component of the plurality of primary frequency components are below 20 dB, preferably below 30 dB, and more preferably below 40 dB, relative to any of the frequency components of the plurality of primary frequency components.

42. The apparatus (110, 121) according to any of claims 38-41, wherein the plurality of primary frequency components comprises at least five frequency components, and preferably at least ten frequency components.

43. The apparatus (110, 121) according to any of claims 24-42, wherein the modulator (320) and the demodulator (350) are driven by a local oscillator, LO, signal with an LO frequency, and wherein the difference in phase shift (∆^^) is different from 0° and ±180° relative to the LO frequency, and wherein the difference in phaseshift (∆^^) preferably is 45° to 135° or –135° to –45°, more preferably 60° to 120° or –120° to –60°, and even more preferably 80° to 100° or –100° to –80°, relative to the LO frequency.

44. The apparatus (110, 121) according to any of claims 24-43, wherein the transceiver (300) is part of a wireless communications network (100) performing Time-Division Duplex, TDD, wireless transmissions, whereineach of the first and second test signals (^^^^1, ^^^^2) are injected between uplink, UL, and downlink, DL, timeperiods in the wireless communications network (100).

45. The apparatus (110, 121) according to any of claims 24-44, wherein the modulator (320) is a quadrature modulator (1120) and wherein the demodulator (350) is a quadrature demodulator (1150).

46. The apparatus (110, 121) according to any of claims 24-45, wherein the processing circuitry (1310) is configured to: initiate transmission of a third signal with the transmitter (310) and / or reception of a fourth signal with the receiver (340), wherein the third and the fourth signals may be adapted for carrying traffic or user data; add the first compensation signal (^^^^^^[^^],^^^^^^[−^^]) to the third signal at the input of the transmitter (310) to suppress a spurious tone during the transmission of the third signal; and / or add the second compensation signal (^^^^^^[^^],^^^^^^[−^^]) to the fourth at the output of the receiver (340) to suppress a spurious tone during the reception of the fourth signal.

47. The apparatus (110, 121) according to any of claims 24-46, comprising the transceiver (300).

48. The apparatus (110, 121) according to claim 47, wherein the apparatus (110, 121) is a wireless device (121) or a network node (110).

49. A computer program comprising instructions which, when executed on at least one processing circuitry (1310), cause the at least one processing circuitry to carry out the method (1200) according to any of claims 1-23.

50. A computer program carrier carrying a computer program according to claim 49, wherein the computer program carrier is one of an electronic signal, optical signal, radio signal, or computer-readable storage medium.

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