Power conversion device

Abstract

This power conversion device turns on / off a semiconductor switching element (Q), thereby converting a DC voltage into an AC voltage, supplying the AC voltage to a rotating machine, and performing variable speed driving of the rotating machine. The power conversion device comprises a drive controller (10) that generates: a first command that is the voltage command of a target voltage of the AC voltage; and a gate signal for turning on / off the semiconductor switching element by comparing a second command with the carrier signal of a triangular wave. The drive controller (10) is configured to: generate the carrier signal of the triangular wave having a frequency of integer Kc times the frequency of the fundamental wave component of the first command; and to generate the second command by changing the second command in accordance with the rotational speed of the rotating machine so that a harmonic component included in the waveform of the gate signal and has a specific order with respect to the frequency of the fundamental wave component of the first command is reduced.

Description

Power Conversion Device

[0001] The present disclosure relates to a power conversion device.

[0002] Some rotating machine control devices use a PWM (Pulse Width Modulation) control system to drive and control an inverter that converts DC power to AC power. The inverter compares a sinusoidal modulation signal with a triangular carrier signal to turn on and off semiconductor switching elements to output a desired phase voltage. The on / off operation of the inverter's switching elements generates harmonic voltages in the phase voltage due to the carrier frequency. For example, harmonic voltages with frequencies of fc±2fs (frequency component A) and fc±4fs (frequency component B) are generated. Here, the frequency of the sinusoidal phase voltage is fs, and the frequency of the carrier signal is fc. This harmonic voltage generates an electromagnetic excitation force with a frequency of fc±3fs. If the frequency of the electromagnetic excitation force overlaps with the mechanical resonance frequency band of the rotating machine, noise and vibration will increase.

[0003] Therefore, in Patent Document 1, harmonic components are superimposed on a sinusoidal modulation signal to distribute the ratio between frequency component A and frequency component B contained in the phase voltage, and it is explained that this can reduce load loss.

[0004] JP 2014-072935 A

[0005] There are cases where noise or torque pulsation generated by frequency component A and frequency component B has the same frequency. In such cases, even if the ratio of frequency component A to frequency component B is varied, noise and torque pulsation cannot be reduced. For example, if the inverter has three phases and fc = 9 fs, frequency component A is 7 fs and 11 fs, and frequency component B is 5 fs and 13 fs. Considering the frequencies of noise and torque pulsation generated by frequency component A, 7 fs is a positive-phase component and is therefore 6 fs, and 11 fs is a negative-phase component and is therefore 12 fs. Similarly, for frequency component B, 5 fs is a negative-phase component and is therefore 6 fs, and 13 fs is a positive-phase component and is therefore 12 fs, resulting in noise and torque pulsation. In other words, even if the ratio of frequency component A to frequency component B is varied, noise and torque pulsation of a frequency common to both components are generated, and therefore the power conversion device of Patent Document 1 cannot reduce these.

[0006] Furthermore, the harmonic components generated by switching have different sidebands that overlap with the mechanical resonance frequency band of the rotating machine depending on the speed and carrier frequency of the rotating machine, so noise cannot be adequately suppressed for rotating machines that operate at variable speeds.

[0007] The present disclosure has been made to solve the above-mentioned problems, and provides a power conversion device that can suppress, with a simple configuration, increases in noise and vibration caused by a carrier frequency that changes depending on the rotation speed.

[0008] The power conversion device disclosed herein is a power conversion device that converts DC voltage into AC voltage by turning semiconductor switching elements on and off using PWM control, supplies the AC voltage to a rotating machine, and drives the rotating machine at a variable speed, and is equipped with a drive controller that generates a first command that is a voltage command for a target voltage of the AC voltage, and a gate signal for turning the semiconductor switching elements on and off by comparing a second command with a triangular wave carrier signal, and the drive controller is configured to generate the triangular wave carrier signal having a frequency that is an integer multiple Kc of the frequency of the fundamental wave component of the first command, and to generate the second command by changing it in accordance with the rotational speed of the rotating machine so that harmonic components of a specific order relative to the frequency of the fundamental wave component of the first command, which are contained in the waveform of the gate signal, are reduced.

[0009] According to the present disclosure, it is possible to provide a power conversion device that can suppress, with a simple configuration, increases in noise and vibration caused by a carrier frequency that changes depending on the rotation speed.

[0010] 9A is a diagram showing an example of a second command value mu of a power conversion device according to a comparative example, and FIG. 9B is a diagram showing an output voltage waveform of the power conversion device controlled by the second command value mu of FIG. 9A . FIG. 9B is a diagram showing harmonics included in the output voltage waveform output by the conventional power conversion device shown in FIG. 9B . FIG. 9C is a diagram showing an example of an output voltage waveform output by the power conversion device according to embodiment 2. FIG. 9D is a diagram showing an example of an output voltage waveform output by the power conversion device according to embodiment 2. FIG. 9E is a diagram showing harmonics included in the output voltage waveform output by the power conversion device according to embodiment 2. FIG. 9F is a diagram showing an example of an output voltage waveform output by the power conversion device according to embodiment 2. 13A is a diagram showing an example of a second command value mu of the power conversion device according to the second embodiment, and FIG. 13B is a diagram showing an output voltage waveform of the power conversion device according to the second embodiment controlled by the second command value mu of FIG. 13A. FIG. 14A is a diagram showing the magnitude of harmonic components included in the second command value mu of FIG. 13A, and FIG. 14B is a diagram showing the phase of the harmonic components included in the second command value mu of FIG. 13A. FIG. 14A is a diagram showing an example of a carrier signal when Kc of the power conversion device according to the second embodiment is an odd number. FIG. 16A and FIG. 16B are diagrams for explaining the operation of the power conversion device according to the second embodiment when Kc is an odd number. FIG. 16B is a diagram for explaining the relationship between the operation of the power conversion device according to the second embodiment and the mechanical resonance frequency of the rotating machine. FIG. 16A is a diagram showing the frequency analysis results of the phase current, phase voltage, and vibration acceleration during one operation of the power conversion device according to the second embodiment. FIG. 16B is a diagram showing the frequency analysis results of the phase current, phase voltage, and vibration acceleration during one operation of the power conversion device of the comparative example. 10 is a diagram showing the results of frequency analysis of phase currents, phase voltages, and vibration accelerations during another operation of the power conversion device according to the second embodiment. FIG.26A is a diagram showing an example of a second command value mu of a power conversion device of a comparative example, and FIG. 26B is a diagram showing an output voltage waveform of the power conversion device controlled by the second command value mu of FIG. 26A. FIG. 26B is a diagram showing harmonics included in the output voltage waveform output by the power conversion device of the comparative example shown in FIG. 26B. FIG. 26C is a diagram showing an example of an output voltage waveform output by the power conversion device of embodiment 3. FIG. 26D is a diagram showing harmonics included in the output voltage waveform output by the power conversion device of embodiment 3. FIG. 26E is a diagram showing an example of an output voltage waveform output by the power conversion device of embodiment 3. FIG. 26F is a diagram showing harmonics included in the output voltage waveform output by the power conversion device of embodiment 3. Fig. 30A is a diagram showing an example of a second command value mu of the power conversion device according to embodiment 3, and Fig. 30B is a diagram showing an output voltage waveform of the power conversion device according to embodiment 3 controlled by the second command value mu of Fig. 30A. Fig. 31A is a diagram showing the magnitude of harmonic components included in the second command value mu of Fig. 30A, and Fig. 31B is a diagram showing the phase of the harmonic components included in the second command value mu of Fig. 30A. Fig. 31B is a block diagram showing the configuration of a power conversion device according to embodiment 4. Fig. 31A is a block diagram showing the configuration of modulated voltage generating means of the power conversion device according to embodiment 4. Fig. 31B is a block diagram showing an example of the hardware configuration of the power conversion device of the present disclosure. Fig. 31B is a block diagram showing another example of the hardware configuration of the power conversion device of the present disclosure.

[0011] Embodiment 1. Figure 1 is a block diagram showing the configuration of a power conversion device 1A according to embodiment 1. The power conversion device 1A includes an inverter circuit 3 connected to a DC power supply 2 and a rotating machine 4, which is a load, and a drive controller 10 comprising voltage command generation means 5, carrier signal generation means 6A, modulated voltage generation means 7A, and gate signal generation means 8. The drive controller 10 controls the on / off of a semiconductor switching element Q of the inverter circuit 3. The DC power supply 2 applies a DC voltage vdc to the inverter circuit 3. The DC power supply 2 may be any device that outputs a DC voltage Vdc, such as a battery, a DC-DC converter, a diode rectifier, or a PWM rectifier.

[0012] The inverter circuit 3 converts DC power from the DC power source 2 into AC power and supplies the power to the rotating machine 4. Note that, in this example, the inverter circuit 3 has three phases, designated as u-phase, v-phase, and w-phase. However, the number of phases of the inverter circuit 3 does not have to be three, as long as it is the same as the number of phases of the rotating machine 4. The inverter circuit 3 is configured such that legs, each of which has two reversely conductive semiconductor switching elements Q connected in series between the positive and negative terminals of the DC power source 2, are connected in parallel in the same number as the number of phases. In this example, since the inverter circuit 3 has three phases, the inverter circuit 3 has six semiconductor switching elements Q, and as shown in FIG. 1 , the six semiconductor switching elements Q are referred to as semiconductor switching elements Qup, Qun, Qvp, Qvn, Qwp, and Qwn. Furthermore, a series connection of a positive-side semiconductor switching element Qup and a negative-side semiconductor switching element Qun corresponding to the u-phase is referred to as leg 31u, a series connection of a positive-side semiconductor switching element Qvp and a negative-side semiconductor switching element Qvn corresponding to the v-phase is referred to as leg 31v, and a series connection of a positive-side semiconductor switching element Qwp and a negative-side semiconductor switching element Qwn corresponding to the w-phase is referred to as leg 31W. The intermediate terminals of each leg are connected to the respective phases of the rotating machine 4.

[0013] Here, each semiconductor switching element Q is configured with an insulated gate bipolar transistor (IGBT) and an anti-parallel diode. Furthermore, if a metal-oxide-semiconductor field-effect transistor (MOSFET), a reverse conducting (RC) IGBT, or the like is used instead of the IGBT, the anti-parallel diode may be omitted. Furthermore, each semiconductor switching element Q may be configured with a wide bandgap semiconductor such as SiC or GaN. In the case of a wide bandgap semiconductor, in addition to being able to reduce losses in the semiconductor switching element, high-speed switching operation can shorten dead time, thereby achieving the effect of reducing errors between the switching operation command and the actual switching operation of the present disclosure.

[0014] The rotating machine 4 has a stator consisting of three-phase stator windings (u-phase, v-phase, and w-phase), and a rotor that rotates due to a rotating magnetic field generated by an AC current flowing through the stator winding. The number of phases of the stator winding does not have to be three, and the stator winding may be configured with two or more sets of windings. In this embodiment, a permanent magnet is used in the rotor of the rotating machine 4. For example, a rare earth magnet with rare earth added, such as neodymium or samarium cobalt, is used as the permanent magnet. Various types of permanent magnets, such as inexpensive ferrite magnets, may also be used. The rotating machine 4 may be a synchronous rotating machine with a field winding provided on the rotor, or an induction machine.

[0015] The voltage command generating means 5 calculates first commands vu*, vv*, vw*, which are phase voltage commands that are sinusoidal, i.e., mainly composed of fundamental sinusoidal components, as voltages to be supplied to the rotating machine 4 as shown in equation (1) based on one or more operation commands of the torque, rotational speed, position, current amplitude, phase, and frequency of the rotating machine 4, and outputs the calculated vu*, vv*, vw* to the carrier signal generating means 6A and the modulated voltage generating means 7A, respectively. In equation (1), vphp represents the amplitude of the first commands vu*, vv*, vw*, and θv represents the fundamental wave phase of the u phase.

[0016] The voltage command generating means 5 may be configured to feed back to the voltage command generating means 5 one or more detected or estimated values ​​of the torque, rotational speed, position, current amplitude, phase, and frequency of the rotating machine 4, and calculate the first commands vu*, vv*, and vw* by feedback control based on the operation command and the detected or estimated values. For example, the first commands vu*, vv*, and vw* may be calculated by well-known current feedback control, speed feedback control, or position feedback control. The first commands vu*, vv*, and vw* are not limited to sine waves, and may be waveforms that include harmonic components in a sine wave.

[0017] The carrier signal generating means 6A generates a triangular wave carrier signal c based on the first commands vu*, vv*, and vw*, and outputs the generated carrier signal c to the gate signal generating means 8.

[0018] The modulation voltage generating means 7A generates second commands mu, mv, and mw, which are phase voltage commands for modulation, based on the first commands vu*, vv*, and vw*, and outputs the generated second commands mu, mv, and mw to the gate signal generating means 8. Note that, since the amplitudes of the second commands mu, mv, and mw need to be changed according to the magnitude of the DC voltage vdc, the DC voltage vdc may be detected by a voltage sensor, and the first commands vu*, vv*, and vw* may be normalized by dividing the first commands vu*, vv*, and vw* by half the detected DC voltage vdc.

[0019] The gate signal generating means 8 compares the magnitude of the second commands mu, mv, mw with the magnitude of the carrier signal c to generate gate signals gup, gun, gvp, gvn, gwp, gwn (sometimes collectively referred to as gate signals g) that control the on / off of the semiconductor switching elements Q of the inverter circuit 3. The gate signals gup, gun, gvp, gvn, gwp, gwn correspond to the semiconductor switching elements Qup, Qun, Qvp, Qvn, Qwp, Qwn, respectively, and turn the corresponding semiconductor switching elements Q on or off. The gate signals gup, gun, gvp, gvn, gwp, gwn each take on a high "H" or low "L" value. When the value of the gate signal g is "H", the semiconductor switching element Q corresponding to the gate signal g is controlled to be ON, and when the value of the gate signal g is "L", the semiconductor switching element Q corresponding to that gate signal g is controlled to be OFF.

[0020] FIG. 2 is a diagram illustrating the operation of the gate signal generating means 8 shown in FIG. 1 . FIG. 2 shows one phase, the u phase. The gate signal generating means 8 compares the second command mu with a triangular wave carrier signal c to generate gate signals gup and gun. Specifically, when the second command mu is greater than the carrier signal c, the value of the gate signal gup is set to "H" and the value of the gate signal gun is set to "L." When the second command mu is smaller than the carrier signal c, the value of the gate signal gup is set to "L" and the value of the gate signal gun is set to "H." The gate signals gup and gun are complementary to each other. In other words, when the gate signal gup is "H," the gate signal gun is set to "L," and when the gate signal gup is "L," the gate signal gun is set to "H." When the gate signal gup is "H", the positive side semiconductor switching element Qup turns on and the output phase voltage vu becomes vdc, and when the gate signal gun is "H", the negative side semiconductor switching element Qun turns on and the output phase voltage vu becomes 0. This method of PWM control, in which the semiconductor switching elements are turned on and off by comparing the triangular wave carrier signal c with a waveform like the second command, is called triangular wave comparison type PWM (control).

[0021] 3 is a diagram showing the configuration of the carrier signal generating means 6A shown in FIG. The carrier signal generating means 6A includes a three-phase to two-phase conversion unit 601, a phase calculation unit 602, and a carrier signal calculation unit 603A. The three-phase to two-phase conversion unit 601 performs three-phase to two-phase conversion from first commands vu*, vv*, and vw* on a three-phase coordinate system to first commands vα* and vβ* on a two-phase coordinate system. Specifically, the three-phase to two-phase conversion unit 601 can perform the three-phase to two-phase conversion using equation (2).

[0022] The phase calculation unit 602 calculates the fundamental wave phase θv of the u-phase of the first commands vα* and vβ* on the two-phase coordinate system. Specifically, as shown in equation (3), the arctangent calculation is performed on the first commands vα* and vβ* on the two-phase coordinate system to obtain the phase of vα*. This phase is the same as the phase of vu*.

[0023] Note that the description here is based on the assumption that the first commands vu*, vv*, vw* and the first commands vα*, vβ* have waveforms with sufficiently few harmonic components and are the same as the fundamental wave components. If the first commands vu*, vv*, vw* and the first commands vα*, vβ* contain many harmonic components, the fundamental wave components can be extracted by passing them through a low-pass filter, for example.

[0024] In this embodiment, as an example, the carrier signal calculation unit 603A calculates a carrier signal c whose frequency is Kc times (Kc is a multiple of 3) the frequency of the fundamental wave components of the first commands vu*, vv*, and vw*. Specifically, the phase θc of the carrier signal is generated by multiplying the fundamental wave phase θv by Kc as shown in equation (4).

[0025] The carrier signal calculation unit 603A then generates a triangular wave carrier signal c as shown in FIG. 4. FIG. 4 is a diagram showing the carrier signal generated by the carrier signal calculation unit 603A shown in FIG. 3. The carrier signal c is a triangular wave whose cycle is 1 / Kc of the fundamental wave components of the first commands vu*, vv*, and vw*, and has a maximum value when the phase θc of the carrier signal is 0° and 360°, a minimum value when it is 180°, and a median value when it is 90° and 270°. In the example shown in FIG. 4, the maximum value of the triangular wave is 1, the minimum value is -1, and the median value is zero. The carrier signal calculation unit 603A outputs the carrier signal c after phase synchronization control to the gate signal generation means 8.

[0026] Next, an increase in noise and vibration due to the frequency of the carrier signal will be described. Generally, the noise and loss generated by a rotating machine depend on the rotation speed of the rotating machine. Therefore, in this embodiment, the modulation voltage is switched according to the rotation speed of the rotating machine.

[0027] The rotating machine 4 has a mechanical resonance frequency band fmr due to its structure. In this embodiment, as an example, the predetermined frequency band is the mechanical resonance frequency fmr of mode 0. It is assumed that harmonic components fc±3fs of the electromagnetic excitation force due to the frequency of the carrier signal c overlap with the mechanical resonance frequency band fmr, where fs is the frequency of the fundamental wave of the first command. At this time, the stator of the rotating machine 4 is excited by the electromagnetic excitation force, increasing vibration and therefore noise.

[0028] 5 is a diagram showing harmonic components fc±3fs of the electromagnetic excitation force when the frequency fc of the carrier signal c is set to a frequency Kc times the frequency fs of the fundamental wave component and the rotational speed of the rotating machine 4 is changed, and also showing the mechanical resonance frequency band fmr of mode 0 of the rotating machine. As shown in Fig. 5, the electromagnetic excitation force has a component (fc+3fs) that increases linearly as the frequency fc of the carrier signal c increases with an increase in the rotational speed of the rotating machine 4, and a component (fc-3fs) that also increases linearly.

[0029] When the rotation speed is N_mr1, there is a region where fc+3fs and the mechanical resonance frequency band fmr of Mode 0 overlap. Furthermore, when the rotation speed is N_mr2, there is a region where fc-3fs and the mechanical resonance frequency band fmr of Mode 0 overlap. The modulated voltage generating means in this embodiment generates a second command value so that fc+3fs, which is a harmonic of the gate signal included in the mechanical resonance frequency band fmr, changes when the rotation speed is N_mr1, and generates a second command value so that fc-3fs, which is a harmonic of the gate signal included in the mechanical resonance frequency band fmr, changes when the rotation speed is N_mr2. That is, the second command is generated by changing it in accordance with the rotation speed of the rotating machine, which is the load, so that harmonic components of a specific order relative to the frequency of the fundamental wave component of the first command, which are included in the waveform of the gate signal, are reduced.

[0030] As described above, the power conversion device according to the first embodiment uses a triangular-wave comparison PWM circuit mounted on a general-purpose controller IC to optimize the frequency distribution of harmonics generated by inverter switching with a relatively simple configuration. Furthermore, it suppresses components within a predetermined frequency band of harmonics contained in the waveform of a gate signal that changes with the rotation speed. Here, the gate signal is an on / off control signal for a semiconductor switching element, so the waveform of the output voltage output by the on / off switching of the semiconductor switching element driven by the gate signal is also the same as the waveform of the gate signal. This effectively suppresses frequency components that cause problems with load noise or torque pulsation, while simultaneously reducing load losses due to harmonic components, achieving significant, unprecedented effects.

[0031] Second Embodiment In the second embodiment, a method for calculating the carrier signal c will be described. In the present disclosure, the frequency of the carrier signal c is Kc times the fundamental wave component (Kc is a multiple of 3, the number of phases). Since the method for calculating the carrier signal differs depending on whether Kc is odd or even, they will be described separately. The configuration of the power conversion device according to the second embodiment is the same as the configuration shown in the block diagram of FIG.

[0032] FIG. 6 is a diagram showing an example of setting the frequency fc of the carrier signal c in the second embodiment. In a low rotational speed range of the rotating machine 4, a constant carrier signal frequency is set using asynchronous PWM. In a speed range where the rotational speed of the rotating machine 4 exceeds a predetermined value, synchronous PWM is used, with the carrier frequency fc being set according to the fundamental frequency fs. For synchronous PWM, a carrier signal c with a frequency that is a multiple of three of the fundamental frequency component fs is set, and as the speed increases, the frequency is switched to synchronous 21 (Kc = 21), synchronous 15 (Kc = 15), synchronous 12 (Kc = 12), and synchronous 9 (Kc = 9). Odd multiples (21, 15, 9) and even multiples (12) of the carrier signal c are used. For even higher rotational speeds, even smaller Kc may be used.

[0033] <When Kc is an Even Number> As an example of when Kc is an even number, an example will be described in which Kc is set to 6 and the phase of the carrier signal c (the median value of the triangular wave) is synchronized with the phase of the phase voltage command at 0°. In this case, the carrier signal will be as shown in FIG. 7. FIG. 7 is a diagram showing an example of the carrier signal c generated by the carrier signal calculation unit 603A shown in FIG. 3. In this embodiment, the carrier signal c is synchronized with the phase of the u-phase voltage, and a common carrier signal c is used for the three phases. For this reason, the even number Kc that determines the frequency of the carrier signal c is set to a multiple of 3, the number of phases. Note that the operation of the gate signal generation means 8 is the same as when Kc is an odd multiple.

[0034] Fig. 8 is a diagram showing the configuration of the modulated voltage generating means 7A shown in Fig. 1. The modulated voltage generating means 7A includes a three-phase to two-phase conversion unit 701, an amplitude calculation unit 702, a phase calculation unit 703, a fundamental wave signal calculation unit 704, an odd-numbered harmonic signal calculation unit 705, and an even-numbered harmonic signal calculation unit 706A.

[0035] The three-phase to two-phase conversion unit 701 performs three-phase to two-phase conversion from first commands vu*, vv*, vw*, which are sinusoidal phase voltage commands on the three-phase coordinate system, to first commands vα*, vβ* on the two-phase coordinate system, using processing similar to that of the three-phase to two-phase conversion unit 601, and outputs the first commands vα*, vβ* on the two-phase coordinate system to the amplitude calculation unit 702 and the phase calculation unit 703, respectively.

[0036] The amplitude calculation unit 702 first calculates the amplitudes vphp of the first commands vu*, vv*, and vw* from the first commands vα* and vβ* on the two-phase coordinates. For example, the amplitude calculation unit 702 can calculate the amplitudes vphp of the first commands vu*, vv*, and vw* using the following equation (5).

[0037] Next, the amplitude calculation unit 702 divides the calculated amplitude vphp of the first commands vu*, vv*, and vw* by half the DC voltage vdc, as shown in the following equation (6), to convert it into a normalized amplitude M of the first commands vu*, vv*, and vw*, and outputs the calculated amplitude M to the fundamental wave signal calculation unit 704, the odd-multiple harmonic signal calculation unit 705, and the even-multiple harmonic signal calculation unit 706A.

[0038] The phase calculation unit 703 calculates the fundamental wave phase θv of the u-phase of the first commands vα* and vβ* on the two-phase coordinate system using processing similar to that of the phase calculation unit 602, and outputs the calculated fundamental wave phase θv to the fundamental wave signal calculation unit 704, the odd-multiple harmonic signal calculation unit 705, and the even-multiple harmonic signal calculation unit 706A.

[0039] The fundamental wave signal calculation unit 704 calculates fundamental wave signals mu_1, mv_1, mw_1, which are the fundamental wave components of the first commands vu*, vv*, vw*, based on the amplitude M of the normalized first commands vu*, vv*, vw* and the fundamental wave phase θv of the u phase.

[0040] The odd-multiple harmonic signal calculation unit 705 calculates odd-multiple harmonic signals mu_odd, mv_odd, mw_odd whose frequencies are odd multiples of the fundamental wave components of the first commands vu*, vv*, vw* based on the amplitude M of the normalized first commands vu*, vv*, vw*, the fundamental wave phase θv of the u phase, and the operation command.

[0041] The even-multiple harmonic signal calculation unit 706A calculates even-multiple harmonic signals mu_even, mv_even, mw_even whose frequencies are even multiples of the fundamental wave components of the first commands vu*, vv*, vw* based on the amplitude M of the normalized first commands vu*, vv*, vw*, the fundamental wave phase θv of the u phase, and the operation command.

[0042] Then, as shown in equation (7), the modulated voltage generating means 7A adds together the fundamental signals mu_1, mv_1, mw_1, odd-multiple harmonic signals mu_odd, mv_odd, mw_odd, and even-multiple harmonic signals mu_even, mv_even, mw_even calculated by the fundamental signal calculation unit 704, odd-multiple harmonic signal calculation unit 705, and even-multiple harmonic signal calculation unit 706A, respectively, to generate second commands mu, mv, and mw, and outputs the generated second commands mu, mv, and mw to the gate signal generating means 8.

[0043] Hereinafter, we will explain in order the fundamental signals mu_1, mv_1, mw_1, odd harmonic signals mu_odd, mv_odd, mw_odd, and even harmonic signals mu_even, mv_even, mw_even calculated by the fundamental signal calculation unit 704, odd harmonic signal calculation unit 705, and even harmonic signal calculation unit 706A of the modulated voltage generating means 7A.

[0044] 9A and 9B are diagrams showing an example of the phase voltage (FIG. 9B) output by a power conversion device as a comparative example when the sinusoidal first command vu* is directly provided to the gate signal generating means 8 as the second command mu (FIG. 9A). FIGS. 9A and 9B show the waveform of the phase voltage output for the u-phase component when the amplitude M of the normalized first commands vu*, vv*, and vw* is 0.6 and the inverter circuit 3 is operated by generating the gate signal g from the second commands mu, mv, and mw using the carrier signal c and the gate signal generating means 8. FIG. 9A shows the sinusoidal second command mu and the triangular wave carrier signal c, while FIG. 9B shows the output phase voltage vu output based on the sinusoidal second command mu and the triangular wave carrier signal c. Here, α1 to α11 in FIG. 9B represent switching phases. When using the carrier signal c of this embodiment, switching occurs 11 times within one cycle of the second command mu. Of these, when the fundamental wave phase θv is 180°, the second command mu and carrier signal c always pass through 0, so the switching phase α6 is always 180°. Furthermore, since the output phase voltage vu is symmetrical in both positive and negative directions between phases 0° and 180° and between phases 180° and 360°, once the switching phases α1 to α5 are determined, the switching phases α7 to α11 are also determined. Therefore, to obtain an output phase voltage vu that reduces or increases the amplitude of predetermined frequency components of the voltage, current, torque, loss, and vibration of the rotating machine 4, and the voltage and current of the DC power supply, the switching phases α1 to α5 can be appropriately adjusted.

[0045] FIG. 10 is a diagram showing the magnitude of harmonic components contained in the output phase voltage vu shown in FIG. 9B. The horizontal axis of FIG. 10 represents the order relative to the fundamental wave, and the vertical axis represents the magnitude of the harmonic components. From FIG. 10, it can be seen that the output phase voltage vu shown in FIG. 9B contains odd- and even-multiple harmonic components. Furthermore, excluding the integer multiples of three that do not appear in the harmonic components of the phase current flowing through the rotating machine 4, it can be seen that the 11th- and 13th-order harmonics are particularly large. Therefore, in this embodiment, as an example, it is assumed that the 11th- and 13th-order harmonic components relative to the fundamental wave component of the output phase voltage vu excite mechanical resonance in the rotating machine 4, causing loud noise or torque pulsation and increasing losses in the rotating machine 4. The switching phases α1 to α5 are adjusted to reduce the 11th- and 13th-order harmonic components of the output phase voltage vu.

[0046] To find the switching phases α1 to α5 that reduce the 11th and 13th harmonic components of the output phase voltage vu, it is sufficient to solve an optimization problem so as to minimize the objective function fobj(α) expressed by, for example, equation (8). Here, the objective function fobj(α) is set to minimize the square root of the sum of squares of the nth-order harmonic voltage vn in the range from second to pth order divided by the fundamental voltage v1, i.e., the total harmonic distortion. Furthermore, kn is a weighting coefficient, and by increasing the weighting coefficient kn for a specific order, it is possible to preferentially reduce harmonics of a specific order. Note that the switching phases α1 to α5 are subject to constraints as shown in Equation (9), and the optimization problem is solved while observing these constraints.

[0047] In the second embodiment, p in equation (8) is set to 50, and the objective function fobj(α) is set so as to minimize the total harmonic distortion for the second to fiftieth harmonic voltages. In addition, by increasing the weighting coefficients of the 11th and 13th orders as shown in equation (10), the 11th and 13th harmonic components are preferentially reduced.

[0048] FIG. 11 shows an example of an output phase voltage vu obtained by adjusting the switching phases α1 to α5 to minimize the 11th- and 13th-order harmonic components. FIG. 12 shows the magnitude of the harmonic components contained in the output phase voltage vu shown in FIG. 11 . The horizontal axis of FIG. 12 represents the order relative to the fundamental wave, and the vertical axis represents the magnitude of the harmonic components. From FIG. 11 , it can be seen that the switching phases α1 to α5 of the output phase voltage vu shown in FIG. 9B are changed by solving the optimization problem to minimize the objective function fobj(α) expressed by equation (8). Furthermore, as shown in FIG. 12 , it can be seen that the 11th- and 13th-order harmonic components are reduced compared to the magnitude of the harmonic components shown in FIG. 10 . Therefore, a second command mu can be generated using a triangular-wave comparison PWM to output the output phase voltage vu shown in FIG. 11 .

[0049] 13A and 13B are diagrams showing an example of the phase voltage ( FIG. 13B ) output by the power conversion device 1A when the second command mu ( FIG. 13A ), generated so as to output the output phase voltage vu shown in FIG. 11 , is provided to the gate signal generating means 8. In the case where the amplitude M of the normalized first commands vu*, vv*, and vw* is 0.6, the waveform of the phase voltage output when the inverter circuit 3 is operated by generating the gate signal g from the second commands mu, mv, and mw using the carrier signal c and the gate signal generating means 8 is shown for the u-phase component. FIG. 13A shows the second command mu and the triangular wave carrier signal c, and FIG. 13B shows the output phase voltage vu output based on the second command mu and the triangular wave carrier signal c. By performing triangular wave comparison PWM using the second command mu shown in FIG. 13A, it is possible to output a waveform similar to the output phase voltage vu shown in FIG.

[0050] Note that, as an example of the waveform of the second command, the waveform of the second command mu is a waveform in which the value of the second command mu remains constant throughout the period in which the carrier signal c changes from its minimum value to its maximum value and the period in which the carrier signal c changes from its maximum value to its minimum value, i.e., the waveform of the second command mu is a stepped waveform. However, as long as the output phase voltage vu shown in Fig. 11 can be output, that is, as long as the phase at which the second command mu crosses the carrier signal c coincides with each switching phase α in Fig. 11, the second command mu does not need to maintain a constant value over a half cycle of the carrier signal c and may have a waveform that changes smoothly.

[0051] 14A and 14B are diagrams showing the magnitude and phase of the harmonic components of the second command mu shown in FIG. 13A. The horizontal axes of FIGS. 14A and 14B represent the order relative to the fundamental wave. The vertical axes of FIG. 14A represent the magnitude of the harmonic components, and the vertical axes of FIG. 14B represent the phase of the harmonic components relative to a sine wave. As can be seen from FIG. 14A, the second command mu is composed of a fundamental wave component, odd-multiple harmonic components, and even-multiple harmonic components. As can be seen from FIG. 14B, the fundamental wave component, odd-multiple harmonic components, and even-multiple harmonic components of the second command mu are all composed of sine wave components (sine waves in which the phase 0° of each harmonic component is synchronized with the phase 0° of the fundamental wave component).

[0052] Therefore, fundamental signal calculation unit 704, odd-multiple harmonic signal calculation unit 705, and even-multiple harmonic signal calculation unit 706A of modulated voltage generation means 7A only need to calculate fundamental signal mu_1, odd-multiple harmonic signal mu_odd, and even-multiple harmonic signal mu_even, respectively, as shown in equation (11), where q is a value set so that 2q+1 is the maximum order of harmonics to be considered, M1 is the amplitude of fundamental signal mu_1, M2n+1 is the amplitude of odd-multiple harmonic signal mu_odd at the 2n+1th order, and M2n is the amplitude of even-multiple harmonic signal mu_even at the 2nth order.

[0053] In this embodiment, the harmonic signals of each order of the odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even are preferably sine waves, and the phase zero of each harmonic signal may be synchronized with the phase zero of the fundamental wave component of the first command vu*. Furthermore, the harmonic signals of each order of the odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even are not limited to sine waves and may have distorted waveforms, as long as they have the same value at the beginning and end of one cycle of the fundamental wave component of the first command and have periodicity such that they cross zero at least once per cycle.

[0054] The odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even that constitute the second command preferably use a relatively large value for q in equation (11), i.e., are composed of a plurality of odd-multiple harmonic signals and a plurality of even-multiple harmonic signals. However, if the odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even each contain a harmonic signal of at least one order, the harmonic components generated by switching can be controlled by controlling the switching phase, as compared to conventional synchronous PWM, thereby achieving the effect of the present disclosure of reducing noise, loss, etc. of the rotating machine.

[0055] The above only describes the second command mu for the u phase, but since the waveforms of the second commands mu, mv, and mw for the three phases are waveforms in which the phases of the fundamental wave components are shifted by 120° from each other, the second commands mv and mw for the remaining two phases can also be generated in the same way as the second command mu.

[0056] <When Kc is an odd number> A case will be described where Kc is an odd number 9, the carrier signal is synchronized with the phase of the u-phase voltage, and a common carrier signal is used for three phases. Therefore, the odd number Kc that determines the frequency of the carrier signal is set to a multiple of the number of phases, 3. In this case, the carrier signal will be as shown in Fig. 15. Fig. 15 is a diagram showing an example of carrier signal c generated by carrier signal calculation unit 603A shown in Fig. 3.

[0057] FIG. 16A (modulation phase voltage command value (second command mu)) and FIG. 16B (phase voltage waveform) show examples of phase voltage waveforms for the u-phase when a gate signal is generated from a sinusoidal modulation phase voltage command value using a triangular wave carrier signal and gate signal generating means to operate the inverter circuit. Based on the concept of Fourier series expansion, the waveform of the phase voltage vu shown in FIG. 16B is expressed as the sum of sine wave components and cosine wave components of various orders. Here, since the phase voltage is positively and negatively symmetric between phases 0° to 180° and between phases 180° to 360°, even-numbered harmonic components are first removed. Next, the phase voltage is inversely symmetric between phases 0° to 90° and between phases 90° to 180°, and the cosine wave component is also removed. Therefore, when a carrier signal with an odd Kc is used, the harmonic components of the phase voltage are only odd-numbered sine wave components. Furthermore, based on this result, when optimizing a phase voltage waveform by superimposing harmonic components on a phase voltage command value or a modulating phase voltage command value, it is appropriate to superimpose only odd-order sinusoidal components on the phase voltage command value or the modulating phase voltage command value. Therefore, when Kc is odd, the even-order harmonic signal calculation unit 706A, which is provided in the configuration of the modulation voltage generating means 7A shown in FIG. 8 when Kc is even, is not necessary. Thus, when Kc is odd, the second command may be composed of an odd-order harmonic signal whose frequency is an odd multiple of the frequency of the fundamental wave component of the first command and the first command. In this case, as long as the odd-order harmonic signal includes at least one odd-order harmonic signal, it is possible to control the switching phase and control the harmonic components generated by switching, compared to conventional synchronous PWM, thereby achieving the effect of the present disclosure, which is to reduce noise, loss, etc., of the rotating machine.

[0058] Next, when a rotating machine serving as a load is driven at a variable speed, the relationship between the electromagnetic excitation force of fc±3 fs generated based on the frequency of the carrier signal and the mechanical resonance frequency band fmr of the rotating machine is shown in Figure 17. The frequency of the carrier signal is set as shown in Figure 6 in accordance with changes in rotation speed. In this figure, there is a location where the electromagnetic excitation force of fc±3 fs and the mechanical resonance frequency band fmr of the rotating machine overlap. The frequency of the carrier signal is the same as in the prior art, but in the rotation speed region where the electromagnetic excitation force caused by the carrier signal overlaps with the mechanical resonance frequency band fmr of the rotating machine, second commands mu, mv, and mw are generated to suppress components included in a specified frequency band for the harmonics of the gate signal.

[0059] When the rotation speed is N1 and the carrier signal has 15 synchronous pulses, the electromagnetic excitation force of fc+3fs overlaps with the mechanical resonance frequency band fmr of the rotating machine. Similarly, when the rotation speed is N2 and the carrier signal has 15 synchronous pulses, the electromagnetic excitation force of fc-3fs overlaps with the mechanical resonance frequency band fmr of the rotating machine. In this way, the frequency at which the electromagnetic excitation force caused by the carrier signal overlaps with the mechanical resonance frequency band fmr of the rotating machine varies depending on the rotation speed of the rotating machine.

[0060] 18 and 19 show frequency analysis results of the phase current, phase voltage, and vibration acceleration for harmonic components when the electromagnetic excitation force caused by the carrier signal overlaps with the mechanical resonance frequency band fmr of the rotating machine, with a rotational speed N1 and a carrier signal of 15 synchronous pulses. The vertical axes of the phase current and phase voltage represent values ​​obtained by dividing the fundamental frequency by the amplitudes iph1 and vph1. FIG. 18 shows the case of the power conversion device according to the second embodiment, while FIG. 19 shows, as a comparative example, a carrier synchronous PWM case in which the second command mu is the same sine wave as the voltage command, as shown in FIG. 9. As shown in FIG. 19, in the case of the carrier synchronous PWM as a comparative example, sidebands are generated in the phase current and phase voltage around the 15th order, which is the frequency of the 15-synchronous pulse carrier signal. Specifically, these sidebands are the 11th, 13th, 17th, and 19th orders. Here, the electromagnetic excitation force fc+3fs generated by the 17th and 19th orders of the phase current and phase voltage overlaps with the mechanical resonance frequency band fmr, generating an 18th-order component of vibration acceleration. A comparison of Figures 18 and 19 shows that the power conversion device of this embodiment is able to significantly reduce the sum of the amplitudes of the specific frequency components (here, the 17th and 19th-order harmonic components) that are assumed to excite mechanical resonance and cause loud noise or torque pulsation.

[0061] Regarding harmonic components when the electromagnetic excitation force caused by the carrier signal overlaps with the mechanical resonance frequency band fmr of the rotating machine, frequency analysis results of the phase current, phase voltage, and vibration acceleration when the rotational speed is N2 and the carrier signal is synchronous 15 pulses are shown in FIGS. 20 and 21 . The power conversion device of this embodiment is shown in FIG. 20 , while a comparative example, carrier-synchronous PWM, in which the second command mu is the same sine wave as the voltage command, as shown in FIG. 9 , is shown in FIG. These figures reveal that the phase current and phase voltage have sidebands centered around the 15th order, which is the frequency of the synchronous 15-pulse carrier signal. Specifically, these are the 11th, 13th, 17th, and 19th orders. Here, the electromagnetic excitation force fc-3fs generated by the 11th and 13th orders of the phase current and phase voltage overlaps with the mechanical resonance frequency band fmr, generating a 12th order component of the vibration acceleration. A comparison of Figures 20 and 21 shows that the power conversion device of this embodiment 2 is able to significantly reduce specific frequency components, in this case the 11th and 13th harmonic components, which are assumed to excite mechanical resonance and cause loud noise or torque pulsation.

[0062] The second command value is generated so that at least one of the harmonic components of the gating signal, which are generated around the frequency of the carrier signal, changes: a first harmonic component whose frequency is offset from the frequency fc of the carrier signal to the lower frequency side by an integer L1 times the frequency of the fundamental wave component of the first command; and a second harmonic component whose frequency is offset from the frequency fc of the carrier signal to the higher frequency side by an integer L2 times the frequency of the fundamental wave component of the first command.

[0063] As described above, according to the power conversion device of the second embodiment, when synchronous PWM control is performed using a carrier signal having a frequency Kc times the frequency of the fundamental wave component, it is possible to suppress harmonics of a predetermined order in voltage or current generated by PWM with a relatively simple configuration by using a triangular wave comparison PWM mounted on a general-purpose controller IC, regardless of whether the number of pulses of the output voltage in one cycle of the fundamental wave component is odd or even, thereby achieving the effect of reducing noise, torque pulsation, vibration, and loss in the rotating machine.

[0064] 22 is a block diagram showing the configuration of a power conversion device 1B according to embodiment 3. The power conversion device 1B includes an inverter circuit 3 connected to a DC power supply 2 and a rotating machine 4 as a load, a voltage command generating means 5, a carrier signal generating means 6B, a modulated voltage generating means 7B, and a gate signal generating means 8.

[0065] Power conversion device 1B has carrier signal generating means 6B having a different function from carrier signal generating means 6A of power conversion device 1A according to embodiment 2, and has modulated voltage generating means 7B having a different function from modulated voltage generating means 7A. Below, a description of the parts common to power conversion device 1A according to embodiment 2 will be omitted, and the following description will mainly focus on the parts that are different from power conversion device 1A.

[0066] Fig. 23 is a block diagram showing the configuration of the carrier signal generating means 6B shown in Fig. 22. The carrier signal generating means 6B includes a three-phase to two-phase conversion unit 601, a phase calculation unit 602, and a carrier signal calculation unit 603B. The function of the carrier signal generating means 6B is slightly different from that of the carrier signal calculation unit 603A in the carrier signal generating means 6A of the power conversion device according to the second embodiment.

[0067] <Kc is an Even Number> The function of the carrier signal calculation unit 603B is basically the same as that of the carrier signal calculation unit 603A when Kc, which determines the frequency of the carrier signal, is an even number. The following describes a case where the even number Kc, which determines the frequency of the carrier signal, is 6, as an example. In the carrier signal calculation unit 603A according to the second embodiment, the phase of the carrier signal c (90°), i.e., the median value of the triangular wave, is synchronized with the phase of the fundamental wave component of the first command vu* (0°). In the carrier signal calculation unit 603B according to the third embodiment, the phase of the carrier signal c (180°), i.e., the minimum value of the triangular wave, is synchronized with the phase of the fundamental wave component of the first command vu* (0°). In this case, the carrier signal becomes as shown in FIG. 24 . FIG. 24 is a diagram showing an example of the carrier signal c generated by the carrier signal calculation unit 603B shown in FIG. 23 . Also, in the third embodiment, as in the second embodiment, the carrier signal c is synchronized with the fundamental wave phase θv of the first command vu*, and a common carrier signal c is used for all three phases. In this case, since the phase difference between the phases is 120°, in order to generate similar output phase voltage waveforms for each phase, the even number Kc that determines the frequency of carrier signal c can be set to a multiple of 6. Note that when individual carrier signals c are used for three phases, it is not necessary to set the even number Kc that determines the frequency of carrier signal c to a multiple of 6, and Kc can be set to any even number for each phase.

[0068] 25 is a diagram showing the configuration of the modulated voltage generating means 7B shown in FIG. 22. The modulated voltage generating means 7B includes a three-phase to two-phase conversion unit 701, an amplitude calculation unit 702, a phase calculation unit 703, a fundamental wave signal calculation unit 704, an odd-multiple harmonic signal calculation unit 705, and an even-multiple harmonic signal calculation unit 706B. The even-multiple harmonic signal calculation unit 706B has a slightly different function from the even-multiple harmonic signal calculation unit 706A according to the second embodiment. The function of the even-multiple harmonic signal calculation unit 706B is basically the same as that of the even-multiple harmonic signal calculation unit 706A described in the second embodiment, but the even-multiple harmonic signals mu_even, mv_even, and mw_even that it calculates are different.

[0069] 26A and 26B are diagrams showing an example of a phase voltage (FIG. 26B) output by a power conversion device as a comparative example in which the sinusoidal first command vu* is directly provided to the gate signal generating means 8 as the second command mu (FIG. 26A). FIG. 26B shows the waveform of the phase voltage output for the u-phase component when the amplitude M of the normalized first commands vu*, vv*, and vw* is 0.6 and the gate signal generating means 8 is used to generate a gate signal g from the second commands mu, mv, and mw to operate the inverter circuit 3. FIG. 26A shows the sinusoidal second command mu and the triangular wave carrier signal c, while FIG. 26B shows the output phase voltage vu output based on the sinusoidal second command mu and the triangular wave carrier signal c. Here, α1 to α12 in FIG. 26B represent switching phases. When the carrier signal c of the third embodiment is used, switching is performed 12 times within one period of the second command mu. The output phase voltage vu is inversion symmetric between phases 0° to 90° and 90° to 180°, and between phases 180° to 270° and 270° to 360°. Therefore, once the switching phases α1 to α3 and α7 to α9 are determined, the switching phases α4 to α6 and α10 to α12 are also determined. Therefore, in order to obtain an output phase voltage vu that reduces or increases the amplitude of predetermined frequency components of the voltage, current, torque, loss, and vibration of the rotating machine 4, and the voltage and current of the DC power supply, the switching phases α1 to α3 and α7 to α9 can be appropriately adjusted.

[0070] FIG. 27 is a diagram showing the magnitude of harmonic components contained in the output phase voltage vu shown in FIG. 26B. The horizontal axis of FIG. 27 represents the order relative to the fundamental wave, and the vertical axis represents the magnitude of the harmonic components. From FIG. 27, it can be seen that the output phase voltage vu shown in FIG. 26B contains odd- and even-multiple harmonic components. Furthermore, excluding the integer multiples of three that do not appear in the harmonic components of the phase current flowing through the rotating machine 4, the 11th- and 13th-order harmonics are particularly large. However, since the second embodiment has already described a case in which the 11th- and 13th-order harmonic components are reduced, in the third embodiment, as an example, it is assumed that the 5th- and 7th-order harmonic components relative to the fundamental wave component of the output phase voltage vu excite mechanical resonance in the rotating machine 4, causing loud noise or torque pulsation and increasing losses in the rotating machine 4. The switching phases α1 to α3 and α7 to α9 are adjusted to reduce the 5th- and 7th-order harmonic components of the output phase voltage vu.

[0071] To find the switching phases α1 to α3 and α7 to α9 that reduce the fifth and seventh harmonic components of the output phase voltage vu, it is sufficient to solve an optimization problem so as to minimize the objective function fobj(α) expressed by, for example, equation (12).

[0072] Here, the objective function fobj(α) is set to minimize the square root of the sum of squares of the nth-order harmonic voltage vn in the range from second to pth order divided by the fundamental voltage v1, i.e., the total harmonic distortion. Furthermore, kn is a weighting coefficient, and by increasing the weighting coefficient kn for a specific order, it is possible to preferentially reduce harmonics of a specific order. Note that the switching phases α1 to α3 and α7 to α9 are constrained by the existence of a phase where the second command and the carrier signal cross each other every half cycle of the carrier signal, as shown in Equation (13). Therefore, the optimization problem is solved while observing these constraints.

[0073] In the third embodiment, the objective function fobj(α) is set so as to minimize the total harmonic distortion for the second to fiftieth harmonic voltages. In addition, by increasing the weighting coefficients for the fifth and seventh harmonic components as shown in equation (14), the fifth and seventh harmonic components are preferentially reduced.

[0074] FIG. 28 shows an example of an output phase voltage vu obtained by adjusting the switching phases α1 to α3 and α7 to α9 to minimize the fifth- and seventh-order harmonic components. FIG. 29 also shows the magnitude of the harmonic components contained in the output phase voltage vu shown in FIG. 28 . The horizontal axis of FIG. 29 represents the order relative to the fundamental wave, and the vertical axis represents the magnitude of the harmonic components. By solving an optimization problem to minimize the objective function fobj(α) expressed by equation (12), as shown in FIG. 28 , the switching phases α1 to α3 and α7 to α9 of the output phase voltage vu shown in FIG. 26B are changed. Furthermore, FIG. 29 shows that the fifth- and seventh-order harmonic components are reduced compared to the magnitude of the harmonic components shown in FIG. 27 (a comparative example). Therefore, a second command mu can be generated using a triangular-wave comparison PWM to output the output phase voltage vu shown in FIG. 28 .

[0075] 30A and 30B are diagrams showing an example of the phase voltage (FIG. 30B) output by the power conversion device 1B when the second command mu (FIG. 30A) generated so as to output the output phase voltage vu shown in FIG. 28 is provided to the gate signal generating means 8. FIG. 30B shows the waveform of the phase voltage output for the u-phase component when the gate signal g is generated from the second commands mu, mv, and mw using the carrier signal c and the gate signal generating means 8 to operate the inverter circuit 3, in the case where the amplitude M of the normalized first commands vu*, vv*, and vw* is 0.6. FIG. 30A shows the second command mu and the triangular wave carrier signal c, and FIG. 30B shows the output phase voltage vu output based on the second command mu and the triangular wave carrier signal c. It can be seen that the triangular wave comparison PWM using the second command mu and triangular wave carrier signal c shown in FIG. 30A can output a waveform similar to the output phase voltage vu shown in FIG. 28.

[0076] Note that, as an example of the waveform of the second command, the second command mu has been described above as a waveform in which the value of the second command mu remains constant throughout the period in which the carrier signal c changes from its minimum value to its maximum value and the period in which the carrier signal c changes from its maximum value to its minimum value, i.e., the waveform of the second command mu has a stepped shape. However, as long as the output phase voltage shown in Fig. 28 can be output, that is, as long as the second command mu is such that the phase at which it crosses the carrier signal c coincides with each switching phase α in Fig. 28, it goes without saying that the second command mu does not need to maintain a constant value over a half cycle of the carrier signal c, and may have a waveform that changes smoothly.

[0077] 31A and 31B are diagrams showing the magnitude (FIG. 31A) and phase (FIG. 31B) of the harmonic components of the second command mu shown in FIG. 30A. The horizontal axes in FIGS. 31A and 31B represent the order of the harmonic relative to the fundamental wave. The vertical axes in FIG. 31A represent the magnitude of the harmonic components, and the vertical axes in FIG. 31B represent the phase of the harmonic components relative to a sine wave. As shown in FIG. 31A, the second command mu is composed of a fundamental wave component, odd-multiple harmonic components, and even-multiple harmonic components. As shown in FIG. 31B, the fundamental wave component and odd-multiple harmonic components of the second command mu are composed of sine wave components synchronized at a phase of 0°, and the even-multiple harmonic components are composed of sine wave components synchronized at a phase minus 90° to the phase of the fundamental wave component.

[0078] Therefore, fundamental signal calculation unit 704, odd-multiple harmonic signal calculation unit 705, and even-multiple harmonic signal calculation unit 706B of modulated voltage generation means 7B only need to calculate fundamental signal mu_1, odd-multiple harmonic signal mu_odd, and even-multiple harmonic signal mu_even as shown in equation (15), where q is a value set so that 2q+1 is the maximum order considered, M1 is the amplitude of fundamental signal mu_1, M2n+1 is the amplitude of odd-multiple harmonic signal mu_odd at the 2n+1th order, and M2n is the amplitude of even-multiple harmonic signal mu_even at the 2nth order.

[0079] In this embodiment, the harmonic signals of each order of the odd-multiple harmonic signal mu_odd are sine waves, and their phases can be synchronized with the zero phase of the fundamental wave component of the first command vu*. Furthermore, if the harmonic signals of each order of the even-multiple harmonic signal mu_even, described as cosine waves in equation (15), are sine waves, the minus 90° phase of the sine wave of the harmonic signals of each order can be synchronized with the zero phase of the fundamental wave component of the first command vu*. Furthermore, the harmonic signals of each order of the odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even are not limited to sine waves and can have distorted waveforms. They can be periodic signals in which the value at the beginning and end of one cycle of the fundamental wave component of the first command is the same and which cross zero at least once per cycle. Furthermore, if the odd-multiple harmonic signal mu_odd and the even-multiple harmonic signal mu_even each contain a harmonic signal of at least one order, the switching phase can be controlled to control the harmonic components generated by switching, as compared to conventional synchronous PWM, thereby achieving the effect of the present disclosure of reducing noise, losses, etc. of the rotating machine.

[0080] In this third embodiment, only the second command mu has been described, but since the waveforms of the three-phase second commands mu, mv, and mw are waveforms in which the phases of the fundamental wave components are shifted by 120° from each other, the second commands mv and mw of the remaining two phases can also be generated in the same way as the second command mu.

[0081] As described above, in the power conversion device 1B according to the third embodiment, the phase of the carrier signal c is shifted by 90° compared to the power conversion device 1A according to the second embodiment. However, in the case of such an even number of pulses, it is possible to suppress harmonics of a predetermined order in the voltage or current generated by PWM with a relatively simple configuration by using a triangular wave comparison type PWM mounted on a general-purpose controller IC. This has the effect of reducing noise, torque pulsation, vibration, and loss in the rotating machine.

[0082] Embodiment 4 Fig. 32 is a diagram showing the configuration of a power conversion device 1C according to embodiment 3. The power conversion device 1C includes an inverter circuit 3 connected to a DC power supply 2 and a rotating machine 4 as a load, a voltage command generating means 5, a carrier signal generating means 6A, a modulated voltage generating means 7C, and a gate signal generating means 8.

[0083] The power conversion device 1C has a modulated voltage generating means 7C whose function is different from that of the modulated voltage generating means 7A of the power conversion device 1A according to embodiment 1. Below, a description of the parts common to the power conversion device 1A according to embodiment 1 will be omitted, and the following description will mainly focus on the parts that are different from the power conversion device 1A.

[0084] Fig. 33 is a block diagram showing the configuration of the modulated voltage generating means 7C shown in Fig. 32. The modulated voltage generating means 7C includes a three-phase to two-phase conversion unit 701, an amplitude calculation unit 702, a phase calculation unit 703, a storage unit 704C, and a modulated voltage calculation unit 705C.

[0085] The storage unit 704C has a first storage unit, a second storage unit, ..., and an n-th storage unit. Here, the first to n-th storage units indicate n operating points for the first command, i.e., combinations of n different parameter values ​​for various parameters such as the voltage amplitude M, the frequency, and the harmonic order to be reduced according to the rotation speed. The first storage unit, the second storage unit, ..., and the n-th storage unit each store in advance a waveform of the second command optimized by optimization calculation for each of the n operating points.

[0086] The modulation voltage calculation unit 705C includes a first modulation voltage calculation unit, a second modulation voltage calculation unit, ..., an nth modulation voltage calculation unit. The first modulation voltage calculation unit, the second modulation voltage calculation unit, ..., the nth modulation voltage calculation unit reads out a corresponding second command waveform from the second command waveforms stored in the first memory unit, the second memory unit, ..., the nth memory unit based on the operation command, the amplitude M of the first command, and the phase θv, and calculates a second command waveform to be compared with the carrier signal. At this time, the second command is generated so that, among the harmonic components of the gating signal generated around the frequency of the carrier signal, at least one of a first harmonic component having a frequency lower than the frequency of the carrier signal and a second harmonic component having a frequency higher than the frequency of the carrier signal changes.

[0087] In the fourth embodiment, the control is performed by the synchronous PWM method, and therefore, based on the operation command, calculations are performed to obtain three-phase second commands mu, mv, and mw that reduce harmonic components of a specific order that are assumed to excite mechanical resonance in the load and cause loud noise or torque pulsation.

[0088] As described above, the power conversion device according to the fourth embodiment can suppress harmonics of a predetermined order in voltage and current generated by PWM with a relatively simple configuration by using a triangular wave comparison PWM mounted on a general-purpose controller IC, regardless of whether the number of pulses is odd or even. This has the effect of reducing noise, torque pulsation, vibration, and loss in the rotating machine.

[0089] Here, the hardware configuration of power conversion devices 1A, 1B, and 1C according to embodiments 1 to 4 will be described. Here, power conversion devices 1A, 1B, and 1C are collectively referred to as power conversion device 1. The functions of drive controller 10 in power conversion device 1, i.e., voltage command generation means 5, carrier signal generation means 6A and 6B, modulated voltage generation means 7A, 7B, and 7C, and gate signal generation means 8, etc., can be realized using a processing circuit. The processing circuit may be dedicated hardware such as a dedicated processing circuit, or may be a processor and storage means.

[0090] 34 is a diagram showing a configuration example of the power conversion device 1 when dedicated hardware is used. When dedicated hardware is used, the drive controller 10 is configured with a dedicated processing circuit. The dedicated processing circuit may be a single circuit, a composite circuit, a programmed processor, a parallel programmed processor, an ASIC (Application Specific Integrated Circuit), an FPGA (Field Programmable Gate Array), or a combination thereof. Each of the multiple functions of the power conversion device 1 described above may be realized by a different dedicated processing circuit, or multiple functions of the power conversion device 1 may be realized collectively by a dedicated processing circuit.

[0091] 35 is a diagram showing a configuration example of the power conversion device 1 when a processor 15 and a storage means 16 are used as the drive controller 10. When the processor 15 and the storage means 16 are used, each function of the drive controller 10 is realized by software, firmware, or a combination of these. The software or firmware is written as a program, and the processor 15 reads and executes the program stored in the storage means 16. It can also be said that these programs cause a computer to execute the procedure or method of each function of the power conversion device 1. The processor 15 is a CPU (Central Processing Unit) and is also called a processing device, arithmetic unit, microprocessor, microcomputer, DSP (Digital Signal Processor), etc. The storage means 16 is, for example, a non-volatile or volatile semiconductor memory such as a read-only memory (ROM), an erasable programmable read-only memory (EPROM), or an electrically programmable programmable read-only memory (EEPROM), or a flexible disk, an optical disk, a compact disk, a digital versatile disk (DVD), etc. Also, some of the functions of the power conversion device 1 may be realized by dedicated hardware, and some may be realized by software or firmware.

[0092] Although various exemplary embodiments and examples are described in this disclosure, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless modifications not illustrated are contemplated within the scope of the technology of this disclosure. For example, this includes cases where at least one component is modified, added, or omitted, and even cases where at least one component is extracted and combined with components of another embodiment.

[0093] For example, in the above embodiment, the inverter circuit 3 is a three-phase inverter circuit, but it may be an inverter circuit with another number of phases, and various inverter circuits can be used, such as a multilevel inverter such as a three-level inverter or a five-level inverter.

[0094] Furthermore, in the first and second embodiments, when the carrier signal generating means 6A synchronizes the median of the triangular wave that is the carrier signal c with the phase 0° of the first commands vu*, vv*, and vw*, the carrier signal generating means 6A uses the median of the decreasing side of the triangular wave, i.e., a phase of 90° in the example of FIG. 4. However, the carrier signal generating means 6A may also synchronize the median of the increasing side of the triangular wave, i.e., a phase of 270° in the example of FIG. 4, with the phase 0° of the first commands vu*, vv*, and vw*. In this case, too, as described in the second embodiment, when the odd-multiple harmonic signals and even-multiple harmonic signals that constitute the second commands are sine waves, the zero phase of each harmonic signal is formed by a sine wave synchronized with the zero phase of the fundamental wave component of the first command.

[0095] Furthermore, in the third embodiment, the carrier signal generating means 6B synchronizes the minimum value of the triangular wave that is the carrier signal c, which is a phase of 180° in the example of FIG. 4, with the phase of 0° of the first commands vu*, vv*, and vw*. However, the carrier signal generating means 6B may also synchronize the maximum value of the triangular wave, which is a phase of 0° or 360° in the example of FIG. 4, with the phase of 0° of the first commands vu*, vv*, and vw*. In this case, as described in the third embodiment, when the odd-multiple harmonic signal and the even-multiple harmonic signal that constitute the second command mu are sine waves, the odd-multiple harmonic component is composed of a sine wave synchronized with the phase of 0° of the fundamental wave component, and the even-multiple harmonic component is composed of a sine wave whose phase -90° is synchronized with the phase of 0° of the fundamental wave component.

[0096] Furthermore, in the above embodiment, the carrier signal generating means 6A, 6B generated a carrier signal c that is common to all phases, but individual carrier signals c may be prepared for each phase. Furthermore, in the above embodiment, the carrier signal generating means 6A, 6B set the odd number Kc that determines the frequency of the carrier signal c to 9 or 15, but the odd number Kc may be an odd number other than 15. Furthermore, the even number Kc that determines the frequency of the carrier signal c was set to 6, but the even number Kc may be an even number other than 6.

[0097] In the above embodiment, the second commands mu, mv, and mw have waveforms in which the value of the second command remains constant throughout the period in which the carrier signal c changes from its minimum value to its maximum value and the period in which the carrier signal c changes from its maximum value to its minimum value, i.e., the waveform of the second command mu has a stepped shape. However, if a high-speed arithmetic device is available, the second commands mu, mv, and mw may have smoother waveforms that are updated at a sufficiently short period relative to the carrier signal c.

[0098] Furthermore, in the above embodiment, the 5th, 7th, 11th, 13th, 17th, and 19th harmonic components are reduced as frequency components, but higher harmonic components other than those exemplified above can also be reduced freely.

[0099] 10 Drive controller, c Carrier signal, mu, mv, mw Second command, Q, Qup, Qun, Qvp, Qvn, Qwp, Qwn Semiconductor switching element, vu*, vv*, vw* First command, g, gup, gun, gvp, gvn, gwp, gwn Gate signal

Claims

1. A power conversion device that converts DC voltage into AC voltage by turning semiconductor switching elements on and off using PWM control, supplies the converted AC voltage to a rotating machine, and drives the rotating machine at a variable speed, comprising a drive controller that generates a first command which is a voltage command for a target voltage of the AC voltage, and a gate signal for turning the semiconductor switching elements on and off by comparing a second command with a triangular wave carrier signal, wherein the drive controller generates the triangular wave carrier signal whose frequency is an integer Kc times the frequency of the fundamental wave component of the first command, and generates the second command by changing it in accordance with the rotational speed of the rotating machine so that a harmonic component of a specific order relative to the frequency of the fundamental wave component of the first command, which is included in the waveform of the gate signal, is reduced.

2. The power conversion device according to claim 1, wherein the carrier signal has a triangular wave whose median, minimum, or maximum value is synchronized with the zero phase of the fundamental wave component of the first command.

3. A power conversion device as described in claim 1 or 2, wherein Kc is an odd number, and the second command is composed of the first command and at least one odd-numbered harmonic signal whose frequency is an odd multiple of the frequency of the fundamental wave component of the first command.

4. A power conversion device as described in claim 1, wherein Kc is an even number, and the second command is composed of harmonic signals including at least one odd-multiple harmonic signal whose frequency is an odd multiple of the frequency of the fundamental wave component of the first command and at least one even-multiple harmonic signal whose frequency is an even multiple of the frequency of the fundamental wave component of the first command, and the first command.

5. A power conversion device as described in claim 4, wherein the carrier signal has a triangular wave whose median value is synchronized with the phase zero of the fundamental wave component of the first command, and the odd-multiple harmonic signal and the even-multiple harmonic signal constituting the second command are sine waves whose phase zero is synchronized with the phase zero of the fundamental wave component of the first command.

6. A power conversion device as described in claim 4, wherein the minimum or maximum value of the triangular wave of the carrier signal is synchronized with the phase zero of the fundamental wave component of the first command, and the even-multiple harmonic signal constituting the second command is a sine wave, and the phase minus 90° of the sine wave is synchronized with the phase zero of the fundamental wave component of the first command.

7. A power conversion device according to any one of claims 1 to 6, wherein the harmonic components of the specific order are determined based on a mechanical resonance frequency of the rotating machine.

8. A power conversion device as claimed in any one of claims 1 to 7, wherein the harmonic component of the specific order is at least one of a first harmonic component which is a harmonic component of a frequency that is an integer L1 times lower frequency than the frequency of the fundamental wave component of the first command from the frequency of the carrier signal, and a second harmonic component which is a harmonic component of a frequency that is an integer L2 times higher frequency than the frequency of the fundamental wave component of the first command from the frequency of the carrier signal.

9. A power conversion device according to any one of claims 1 to 8, wherein the waveform of the second command is a waveform that does not change and remains constant during the period in which the carrier signal changes from its maximum value to its minimum value and during the period in which the carrier signal changes from its minimum value to its maximum value.

10. A power conversion device according to any one of claims 1 to 9, comprising an inverter circuit in which a DC voltage is applied to both ends and a plurality of legs, each leg having two reverse-conducting semiconductor switching elements connected in series, are connected in parallel.

11. A power conversion device according to any one of claims 1 to 10, wherein the semiconductor switching elements are formed from wide bandgap semiconductors.

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