Multi-cross slot coupled feeding OAM antenna based on dielectric resonator

Abstract

The invention discloses a multi-cross-slot coupled feed OAM antenna based on a dielectric resonator, and the antenna comprises four parts: the dielectric resonator, a reflection plate, a dielectric substrate, and a feed network which are stacked together from top to bottom. The dielectric resonator is integrally cylindrical, a cylindrical counter bore structure is arranged in the center of the interior of a cylindrical main body, and a cross recess structure extends to the periphery by taking a cylindrical counter bore as the center; and the surface of the reflecting plate is rotationally and periodically provided with etched cross-shaped grooves. Vortex waves are generated by combining the single-dielectric resonator with multi-cross-slot coupling feeding, the feed network generates excitation phase difference by using different structures, and meanwhile, distribution of current on the surface of the antenna can be changed by combining the cross slots, so that miniaturization design of the antenna is facilitated.

Description

Technical Field

[0001] The present invention belongs to the technical field of antennas, and particularly relates to a multi-cross-slot coupled-fed OAM antenna based on a dielectric resonator. Background Art

[0002] With the iterative upgrade of mobile communication technologies, it has now entered the era of the fifth-generation mobile communication technology (5G). The commercialization of 5G has promoted the Internet of Things into a period of rapid development. Industries such as smart homes, driverless, wearable devices, and remote medical care have developed rapidly, making the connections between people, between people and things, and between things and things closer and closer. The number of mobile terminals used for communication is increasing continuously. However, as the spectrum is a non-renewable strategic resource, the available communication frequency band resources have become increasingly tense. Under this background, increasing the channel capacity of communication systems and the signal anti-crosstalk ability has become a challenge. To solve this problem, antennas that generate vortex waves have been applied to communication and radar. The vortex electromagnetic wave carrying orbital angular momentum has unique properties. The orbital angular momentum (OAM) beams of different modes are orthogonal to each other. This characteristic enables it to increase the channel capacity of communication systems and solve problems such as signal crosstalk, thereby improving the spectrum utilization rate. Theoretically speaking, based on the orthogonal OAM modes, an almost infinite number of transmission channels can be designed within the existing frequency band to achieve the simultaneous transmission of multiple pieces of information at the same frequency. However, the common OAM antenna structures are relatively complex, and there are problems such as large aperture and poor integration. When traditional array antennas excite OAM modes, a complex feeding network needs to be designed, and the number of modes is limited by the number of antenna elements. Although the size of a single patch antenna or dielectric resonator antenna is small, the number of excited OAM modes is small and the design difficulty is large, which limits the research and development of portable high-performance OAM communication systems. Summary of the Invention

[0003] The purpose of the present invention is to provide a multi-cross-slot coupled-fed OAM antenna based on a dielectric resonator, which uses a single dielectric resonator combined with multi-cross-slot coupled feeding to generate vortex waves. The feeding network uses different structures to generate excitation phase differences. At the same time, combining multiple cross slots can change the distribution of the surface current of the antenna, which is helpful for the miniaturization design of the antenna.

[0004] The present invention is realized by the following technical solutions: A multi-cross-slot coupled-fed OAM antenna based on a dielectric resonator includes four parts, which are, from top to bottom, a dielectric resonator, a reflector, a dielectric substrate, and a feeding network, and the four are stacked together;

[0005] The shape of the dielectric resonator is generally cylindrical. A cylindrical counterbore structure is provided at the central position inside the cylindrical main body, and a cross-slot structure extends from the cylindrical counterbore to the surrounding;

[0006] Etched cross slots arranged in a rotating periodic pattern on the surface of the reflector.

[0007] More preferably, the rotating periodic arrangement means that the long sides and short sides of the four cross slots are rotated at a rotation angle of 90 degrees, thereby forming a change in the arrangement pattern.

[0008] More preferably, the feeding network includes a quarter-wavelength impedance transformer, a microstrip transmission line, and a T-shaped power divider, with 5 ports. Among them, port 1 is the input port, and ports 2 - 5 are output ports. The T-shaped power divider has four branches a, b, c, and d. The T-shaped power divider is composed of three one-to-two power dividers. The left and right sides of the first one-to-two power divider are respectively connected to the second one-to-two power divider and the third one-to-two power divider through quarter-wavelength impedance transformers; the second one-to-two power divider branches out points a and b, and the third one-to-two power divider branches out points c and d. The second one-to-two power divider and the third one-to-two power divider together form 4 branches.

[0009] More preferably, at the input port is the first one-to-two power divider. One end of the microstrip transmission line at the input port is connected to one end of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line. Immediately afterwards, both the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are vertically connected to the microstrip transmission line at the input port and chamfered. The other ends of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are each connected to a vertical microstrip line.

[0010] More preferably, the two quarter-wavelength impedance transformers on both sides are perpendicularly connected to the two vertical microstrip lines of the first one-to-two power divider and chamfered. The second one-to-two power divider and the third one-to-two power divider have the same structure as the first one-to-two power divider. The second one-to-two power divider is perpendicularly connected to the quarter-wavelength impedance transformer on the left and chamfered. The first one-to-two power divider is connected to the third one-to-two power divider through another quarter-wavelength impedance transformer. The chamfering method at the connection of the third one-to-two power divider and the quarter-wavelength impedance transformer is also the same.

[0011] More preferably, microstrip line one is from point a to port 3. The starting section of microstrip line one is perpendicular to the branch of the second one-to-two power divider. Microstrip line one undergoes one 90-degree upward bend and one 90-degree right bend, and is divided into three microstrip lines;

[0012] Microstrip line two is from point b to port 4. The starting section of microstrip line two is perpendicular to the branch of the second one-to-two power divider. Microstrip line two undergoes one 90-degree upward bend and is divided into two microstrip lines;

[0013] The third microstrip line goes from point c to port 5. The starting section of the third microstrip line is parallel to the branch of the third one-to-two power divider. The third microstrip line undergoes three 90-degree bends, which are: one 90-degree bend to the right, one 90-degree bend upward, and one 90-degree bend to the left, and is divided into four sections of microstrip lines.

[0014] The fourth microstrip line goes from d to port 2. The starting section of the fourth microstrip line is perpendicular to the branch of the third one-to-two power divider. The fourth microstrip line undergoes three 90-degree bends, which are: one 90-degree bend upward, one 90-degree bend to the left, and one 90-degree bend downward, and is divided into four sections of microstrip lines.

[0015] Further preferably, in order to generate a 90° phase, the lengths between the first microstrip line, the second microstrip line, the third microstrip line, and the fourth microstrip line should differ by a quarter-wavelength length, or an n-fold wavelength length can also be added to the quarter-wavelength length. Therefore, the length difference between the third microstrip line and the second microstrip line is a quarter-wavelength, and the phase difference between the signals output at ports 4 and 5 is 90°; the length difference between the fourth microstrip line and the third microstrip line is five-quarter wavelengths, and the phase difference between the signals output at ports 2 and 5 is 90°; the length difference between the first microstrip line and the third microstrip line is a half-wavelength, and the phase difference between the signals output at ports 3 and 5 is 180°, so the phase difference between the signals output at ports 3 and 2 is 90°. Through this design, the signal amplitudes at each port can be made the same, and the phase difference between adjacent port signals is ninety degrees.

[0016] Further preferably, in order to generate a 180° phase, the lengths between the microstrip transmission lines should differ by a half-wavelength length, or an n-fold half-wavelength length can also be added to the half-wavelength length. Therefore, the length difference between the second microstrip line and the third microstrip line is a half-wavelength, and the phase difference between the signals output at ports 4 and 5 is 180°; the length difference between the fourth microstrip line and the third microstrip line is a half-wavelength, and the phase difference between the signals output at ports 2 and 5 is 180°; the length difference between the first microstrip line and the fourth microstrip line is a half-wavelength, and the phase difference between the signals output at ports 3 and 2 is 180°. Through this design, the signal amplitudes at each port can be made the same, and the phase difference between adjacent port signals is one hundred and eighty degrees.

[0017] Further preferably, the part of the physical structure of the feeding network that generates the phase has been bent and chamfered.

[0018] The beneficial effects of the present invention:

[0019] What the present invention proposes is to use a single dielectric resonator combined with a multi-cross-slot coupling feed to generate a vortex wave. This antenna realizes the excitation of the vortex wave by adopting a single dielectric resonator, breaking through the limitation in the traditional technology that relies on multiple dielectric resonator arrays, thereby significantly reducing the volume of the device and improving the integration degree.

[0020] The feeding network used utilizes different structures to generate excitation phase differences. At the same time, combining multiple cross slots can change the distribution of the surface current of the antenna, making the input impedance of the antenna closer to the characteristic impedance of the transmission line, and can optimize the radiation characteristics and expand the working bandwidth of the antenna, thus enabling good realization of antenna miniaturization. Most traditional vortex wave antennas use feeding networks that generate the same excitation phase. Therefore, phase control elements need to be added to change the excitation phase to generate vortex waves, which makes it difficult to miniaturize the antenna.

[0021] The antenna of the present invention generates different hybrid modes by adjusting the physical length of the microstrip line in the feeding network and changing its electrical length to introduce a specific phase difference between the radiation units. The electric field or magnetic field of the antenna on the xy plane shows a periodic distribution state. The higher the order of the mode that the antenna can excite, the higher the maximum modal number of the generated OAM will be.

[0022] On this basis, the present invention also studied the influence of multiple cross slots on the performance and modes of the antenna. As the number of cross slots increases, the performance of the antenna will be improved accordingly, and the antenna bandwidth and maximum gain will also have a certain increase. At the same time, the number of vortex wave modes will also increase with the increase in the number of cross slots.

[0023] Therefore, this invention and research have certain significance for the design of OAM antennas to provide miniaturization, high-performance design, and obtain high-purity multi-modal vortex waves. Brief Description of the Drawings

[0024] Figure 1 It is a perspective view of a cross-slot-fed OAM dielectric resonator antenna.

[0025] Figure 2 It is a structural diagram of the dielectric resonator and the dielectric substrate.

[0026] Figure 3 It is a structural diagram of the metal surface.

[0027] Figure 4 It is a top view of a multi-cross-slot-fed OAM dielectric resonator antenna.

[0028] Figure 5 It is a structural diagram of the feeding network.

[0029] Figure 6 It is the surface electric field distribution of the dielectric resonator antenna at 4.1 GHz

[0030] Figure 7 It is the cross-sectional magnetic field distribution of a 4-cross-slot dielectric resonator antenna.

[0031] Figure 8 It is the wavefront phase of the dielectric resonator under different phase differences.

[0032] Figure 9It is the radiation pattern of a 4-cross-slot antenna.

[0033] Figure 10 It is the structure of a multi-cross-slot fed antenna.

[0034] Figure 11 It is the wavefront phase diagram of a 6-cross-slot fed antenna.

[0035] Figure 12 It is the radiation pattern of a 6-cross-slot fed antenna at 4.1 GHz.

[0036] Figure 13 It is the wavefront phase diagram of an 8-cross-slot fed antenna.

[0037] Figure 14 It is the radiation pattern of an 8-cross-slot fed antenna at 4.1 GHz.

[0038] Figure 15 It is the scattering coefficient diagram of a multi-cross-slot antenna. Specific implementation manners

[0039] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0040] A multi-cross-slot coupled-fed OAM antenna based on a dielectric resonator, as Figure 1 shown, the main structure of the antenna consists of four parts, which are, from top to bottom, a dielectric resonator 100, a reflector 200, a dielectric substrate 300, and a feeding network 400, and the four are stacked together.

[0041] The dielectric constant of the dielectric resonator 100 is set to 12, and its shape is generally cylindrical. A cylindrical counterbore structure is provided at the center position inside the cylindrical main body, and a cross-slot structure extends from the cylindrical counterbore to the surrounding. Through the combination of the cylindrical main body, the cylindrical counterbore, and the cross-slot, a structure with electromagnetic resonance characteristics is formed.

[0042] As Figure 3 shown, the reflector 200 has etched cross slots arranged in a rotating periodic pattern. The rotating periodic arrangement means that the long slots and short slots of the four cross slots are rotated at a rotation angle of 90 degrees, thereby forming a change in the arrangement pattern. The dielectric substrate 300 uses Rogers 4003c material with a thickness of 0.508 mm (dielectric constant 3.55 and loss tangent of 0.02).

[0043] Referring to Figures 2 - 4, the parameter values of each part of the dielectric resonator 100, the reflector 200, and the dielectric substrate 300 are shown in Table 1.

[0044] Table 1 Specific parameter values of each parameter of the antenna (unit: mm)

[0045]

[0046]

[0047] Among them, W and L are the width and length of the reflector respectively, R is the diameter of the dielectric resonator, h1 is the height of the dielectric resonator, h is the height of the cylindrical counterbore of the dielectric resonator, W1 is the slot width of the cross slot of the dielectric resonator; W2 is the slot width of the cross slot of the reflector, L1 is the length of the long slot of the cross slot of the reflector, and L2 is the length of the short slot of the cross slot of the reflector.

[0048] As Figure 5 shown, the feeding network 400 includes a quarter-wavelength impedance transformer, a microstrip transmission line, and a T-shaped power divider. Among them, port 1 is the input port, and ports 2 - 5 are output ports. Below points a, b, c, and d of this feeding network is a T-shaped power divider. The T-shaped power divider has four branches a, b, c, and d. The T-shaped power divider is composed of three one-to-two power dividers. At the input port is the first one-to-two power divider. The impedance of the microstrip transmission line at the input port is about 50 ohms. The length of the microstrip transmission line at the input port is U1, and the width is W3. One end of the microstrip transmission line at the input port is connected to one end of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line. Immediately afterwards, both the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are vertically connected to the microstrip transmission line at the input port and chamfering treatment is carried out. The chamfering size is 0.4 mm. The other ends of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are each connected to a vertical microstrip line. The length of the vertical microstrip line is U3. The vertical microstrip line is perpendicular to the horizontal left microstrip line and the horizontal right microstrip line. According to the power divider principle, the characteristic impedance that needs to be matched can be obtained as 100 ohms. Therefore, the sum of the length U3 of the vertical microstrip line and the length and width of the horizontal left microstrip line can be obtained. In order to reduce signal reflection and loss, chamfering treatment is carried out at the connection of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line with the vertical microstrip line. The chamfering size is 0.6 mm. Since it is a symmetric structure, the structure on the right side of the microstrip transmission line at the input port is the same as that on the left side.

[0049] On the left and right sides of the first one-to-two power divider, the second one-to-two power divider and the third one-to-two power divider are respectively connected through quarter-wavelength impedance transformers. The length U4 of the quarter-wavelength impedance transformer is approximately a quarter wavelength. The two quarter-wavelength impedance transformers on both sides are perpendicularly connected to the two vertical microstrip lines of the first one-to-two power divider and chamfered, and the chamfer size is 1 mm. The second one-to-two power divider and the third one-to-two power divider have the same structure as the first one-to-two power divider. The second one-to-two power divider is perpendicularly connected to the left quarter-wavelength impedance transformer and chamfered, and the chamfer size is 1.2 mm. Because of the symmetric structure, the first one-to-two power divider is connected to the third one-to-two power divider through another quarter-wavelength impedance transformer, and the chamfering method at the connection of the third one-to-two power divider and the quarter-wavelength impedance transformer is the same. The second one-to-two power divider branches out points a and b, and the third one-to-two power divider branches out points c and d. The second one-to-two power divider and the third one-to-two power divider together form 4 branches, and the lengths of the microstrip lines of the 4 branches are all L3, which can generate four signals with the same amplitude and phase, but in fact there are large errors. Therefore, we still need to adjust the parameters through simulation software.

[0050] Above points a, b, c, and d of the feeding network is a part that generates phase through the physical structure. Microstrip line 1 is from point a to port 3. The starting section of microstrip line 1 is perpendicular to the branch of the second one-to-two power divider. Microstrip line 1 undergoes two 90-degree bends (one 90-degree upward bend and one 90-degree rightward bend), and is divided into three sections of microstrip lines. Starting from point a, the lengths of the three sections of microstrip lines are U6, U7, and U8 in sequence.

[0051] Microstrip line 2 is from point b to port 4. The starting section of microstrip line 2 is perpendicular to the branch of the second one-to-two power divider. Microstrip line 2 undergoes one 90-degree upward bend and is divided into two sections of microstrip lines. Starting from point b, the lengths of the two sections of microstrip lines are U9 and U 10 。

[0052] Microstrip line 3 is from point c to port 5. The starting section of microstrip line 3 is parallel to the branch of the third one-to-two power divider. Microstrip line 3 undergoes three 90-degree bends (in sequence: one 90-degree rightward bend, one 90-degree upward bend, and one 90-degree leftward bend), and is divided into four sections of microstrip lines. Starting from point c, the lengths of the four sections of microstrip lines are L8, L9, L 10 、L 11 。

[0053] The microstrip line four is connected to port 2. The starting segment of the microstrip line four is perpendicular to the branch of the third one-to-two power divider. The microstrip line four undergoes three 90-degree bends (in sequence: one 90-degree upward bend, one 90-degree left bend, and one 90-degree downward bend), and is divided into four segments of microstrip lines. Starting from point d, the lengths of the four segments of microstrip lines are L4, L5, L6, and L7 in sequence.

[0054] In order to generate a 90° phase, the lengths between the microstrip line one, microstrip line two, microstrip line three, and microstrip line four should differ by a quarter-wavelength length, or n times the wavelength length can be added to the quarter-wavelength length. Therefore, the length difference between the microstrip line three and the microstrip line two is a quarter-wavelength, and the phase difference of the output signals at ports 4 and 5 between the two is 90°; the length difference between the microstrip line four and the microstrip line three is five-quarter wavelengths, and the phase difference of the output signals at ports 2 and 5 between the two is 90°; the length difference between the microstrip line one and the microstrip line three is half a wavelength, and the phase difference of the output signals at ports 3 and 5 between the two is 180°. So the phase difference of the output signals between port 3 and port 2 is 90°. Through this design, the signal amplitude of each port can be made the same, and the phase difference between adjacent ports is ninety degrees.

[0055] In order to generate a 180° phase, the lengths between the microstrip transmission lines should differ by half a wavelength length, or n times the half-wavelength length can be added to the half-wavelength length. Therefore, the length difference between the microstrip line two and the microstrip line three is half a wavelength, and the phase difference of the output signals at ports 4 and 5 between the two is 180°; the length difference between the microstrip line four and the microstrip line three is half a wavelength, and the phase difference of the output signals at ports 2 and 5 between the two is 180°; the length difference between the microstrip line one and the microstrip line four is half a wavelength, and the phase difference of the output signals at ports 3 and 2 between the two is 180°. Through this design, the signal amplitude of each port can be made the same, and the phase difference between adjacent ports is one hundred and eighty degrees.

[0056] However, in order to make it miniaturized and avoid causing more phase errors, the part of the physical structure of the feeding network that generates the phase has been bent and chamfered. At the microstrip line two, the microstrip line two is horizontal and to the right and perpendicular to the output port b of the second one-to-two power divider. A chamfering treatment is carried out at its connection, and the chamfer size is 1 mm. Then, chamfering treatments are carried out at the outer corners of the bends at both ends of the microstrip line two, and the chamfer size is 1.2 mm. The end of the microstrip line two is the output port 4.

[0057] At the microstrip line three, chamfering treatments are carried out at the outer corners of the three 90-degree bends of the microstrip line three, and the chamfer size is 1.2 mm. The end of the microstrip line three is the output port 5.

[0058] At the four microstrip lines, the starting section of Microstrip Line 4 is horizontal and to the right, perpendicular to the output port d of the third one-to-two power divider. A chamfering treatment is performed at its connection, with a chamfer size of 1 mm. Then, chamfering treatments are performed on the outer corners of the three 90-degree bends of Microstrip Line 4, with a chamfer size of 1.2 mm. The end of Microstrip Line 4 is Output Port 2.

[0059] At the first microstrip line, the starting section of Microstrip Line 1 is horizontal and to the left, perpendicular to the output port a of the second one-to-two power divider. A chamfering treatment is performed at its connection, with a chamfer size of 1 mm. Then, chamfering treatments are performed on the outer corners of the two 90-degree bends of Microstrip Line 1, with a chamfer size of 1.2 mm. The end of Microstrip Line 1 is Output Port 3.

[0060] Finally, parameter sweeping optimization is carried out through simulation software to enable the feeding network to have better performance. The optimized 90°, 180° feeding networks are shown in Table 2 and Table 3. When the number of cross slots on the reflector of the antenna of the present invention is four, the phase difference required to obtain OAM modes 1 and 2 can be obtained by adjusting the length of the microstrip lines of the feeding network.

[0061] Table 2 Specific parameter values of the 90° feeding network (unit: mm)

[0062]

[0063]

[0064] Table 3 Specific parameter values of the 180° feeding network (unit: mm)

[0065]

[0066] By adjusting the parameters, the excitation of the hybrid electromagnetic mode can be achieved and a high-purity multi-modal vortex wave can be obtained. As Figures 6 - 8 shown, when the phase difference of the output ports is 90°, the surface electric field of the dielectric resonator is like the cross-sectional magnetic field as Figure 6 (a) shown, there is a maximum value distributed in the radial direction of the electric field. As Figure 7 (a) shown, the magnetic field distribution has 1 cycle (2 maximum values) along the circumferential direction, that is, the dielectric resonator antenna operates in HEM 11δ . Similarly, when the phase difference of the output ports is 180°, as Figure 6 (b), 7(b) shown, it can be seen that the dielectric resonator antenna operates in HEM 22δ mode. The designed feeding network with 90°, 180° phase difference is used to couple-feed the dielectric resonator cross slots, and the generated wavefront phase is as Figure 8As shown, the wavefront phases of the dielectric resonator all exhibit a spiral distribution, and the phase distribution is also relatively concentrated. There is no truncation or divergence in its spiral arms, indicating that the dielectric resonator antenna can successfully excite the vortex electromagnetic waves with OAM modes of 1 and 2. Through testing, it is known that the antenna can cover a bandwidth of 3.92 GHz - 4.53 GHz. The gain in the main radiation direction of mode 1 reaches 4.7 dBi, and the gain in the main radiation direction of mode 2 at 40° reaches 3 dBi, as Figure 9 shown. Therefore, the radiation patterns of the antenna in modes 1 and 2 both exhibit good radiation performance.

[0067] On this basis, we also studied the influence of the number of cross slots on the reflector on the antenna performance. The structure of the multi-cross slot fed antenna is as Figure 10 shown. The relationship between the number of cross slots and the number of modes that can be generated is l = N / 2 (N is the number of cross slots). Therefore, when the number of cross slots is six, three modes can be generated. At this time, the dielectric resonator antenna operates in the HEM 33δ mode, and the maximum gain reaches 7.57 dBi at 4.1 GHz, as Figure 11 and Figure 12 shown. The operating bandwidth is also correspondingly increased to the range of 3.62 GHz - 4.83 GHz. When the number of cross slots is eight, four modes can be generated. At this time, the dielectric resonator antenna operates in the HEM 44δ mode, and the maximum gain reaches 9.06 dBi at 4.1 GHz, as Figure 13 and Figure 14 shown. The operating bandwidth is also correspondingly increased to the range of 3.65 GHz - 4.93 GHz. The results show that as the number of cross slots increases, the bandwidth of the antenna gradually becomes wider. To a certain extent, the cross slots can be equivalent to array elements, significantly reducing the structural complexity and the antenna size, and the number of modes that can be excited also increases. The OAM dielectric resonator antenna of the present invention provides a new idea for compact multi-mode vortex wave generation.

[0068] The above only expresses the preferred embodiments of the present invention, and does not limit the present invention in other forms. Any person skilled in the art may use the disclosed content to make changes or modifications into equivalent embodiments with equivalent changes. However, any simple modifications, equivalent changes, and modifications made to the above embodiments based on the technical essence of the present invention without departing from the technical solution content of the present invention still fall within the protection scope of the technical solution of the present invention.

Claims

1. A multi-cross slot coupled feeding OAM antenna based on dielectric resonator, characterized by: It includes four parts, from top to bottom, dielectric resonator, reflector, dielectric substrate, and feeding network, which are stacked together; The dielectric resonator is cylindrical in shape as a whole, a cylindrical countersunk hole structure is provided at the center of the cylindrical body, and a cross groove structure extends around the cylindrical countersunk hole; The etched cross grooves are arranged periodically on the surface of the reflector.

2. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 1 is characterized in that: The rotational periodic arrangement means that the long sides and short sides of the four cross grooves are rotated at a rotation angle of 90 degrees, thereby constituting a change in the arrangement mode.

3. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 1, characterized in that: The feeding network includes a quarter-wavelength impedance transformer, a microstrip transmission line, and a T-type power divider, including 5 ports, of which port 1 is the input port and ports 2 to 5 are output ports. The T-type power divider has four branches, namely a, b, c, and d. The T-type power divider is composed of three one-to-two power dividers. The left and right sides of the first one-to-two power divider are connected to the second one-to-two power divider and the third one-to-two power divider through a quarter-wavelength impedance transformer respectively; the second one-to-two power divider divides points a and b, and the third one-to-two power divider divides points c and d. The second one-to-two power divider and the third one-to-two power divider form a total of 4 branches.

4. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 3 is characterized in that: At the input port is the first one-to-two power divider. One end of the input port microstrip transmission line is connected to one end of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line. Then the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are vertically connected to the input port microstrip transmission line and cut at an angle. The other ends of the horizontal left microstrip transmission line and the horizontal right microstrip transmission line are each connected to a vertical microstrip line.

5. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 5, characterized in that: The quarter-wavelength impedance transformers on both sides are vertically connected to the two vertical microstrip lines of the first one-to-two power divider and are cut at an angle. The second one-to-two power divider and the third one-to-two power divider have the same structure as the first one-to-two power divider. The second one-to-two power divider is vertically connected to the quarter-wavelength impedance transformer on the left and is cut at an angle. The first one-to-two power divider is connected to the third one-to-two power divider through another quarter-wavelength impedance transformer, and the cutting method at the connection between the third one-to-two power divider and the quarter-wavelength impedance transformer is also the same.

6. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 3 is characterized in that: Microstrip line 1 is from point a to port 3. The starting section of microstrip line 1 is perpendicular to the branch of the second one-to-two power divider. Microstrip line 1 is divided into three sections after a 90-degree upward bend and a 90-degree right bend. Microstrip line 2 is from point b to port 4. The starting section of microstrip line 2 is perpendicular to the branch of the second one-to-two power divider. Microstrip line 2 is bent upward by 90 degrees and is divided into two sections of microstrip line. Microstrip line 3 is from point c to port 5. The starting section of microstrip line 3 is parallel to the branch of the third one-to-two power divider. Microstrip line 3 is bent 90 degrees three times, one 90 degree bend to the right, one 90 degree bend upward, and one 90 degree bend to the left, and is divided into four sections of microstrip line. Microstrip line four is from d to port 2. The starting section of microstrip line four is perpendicular to the branch of the third one-to-two power divider. Microstrip line four undergoes three 90-degree bends, namely: a 90-degree upward bend, a 90-degree left bend, and a 90-degree downward bend, and is divided into four sections of microstrip line.

7. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 6 is characterized in that: In order to produce a 90° phase, the lengths of microstrip line one, microstrip line two, microstrip line three, and microstrip line four should differ by a quarter wavelength, or by adding n times the wavelength to a quarter wavelength; therefore, the length of microstrip line three differs by a quarter wavelength from that of microstrip line two, and the phases of their output signals at port 4 and port 5 differ by 90°; the length of microstrip line four differs by four-and-five wavelengths from that of microstrip line three, and the phases of their output signals at port 2 and port 5 differ by 90°; the length of microstrip line one differs by half a wavelength from that of microstrip line three, and the output signals at port 3 and port 5 differ by 180°, so the output signal phases of port 3 and port 2 differ by 90°. Through this design, the signal amplitude of each port is the same, and the phases of signals at adjacent ports differ by ninety degrees.

8. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 6, characterized in that: In order to produce a 180° phase, the lengths of the microstrip transmission lines should differ by half a wavelength, or by adding n times the half wavelength to the half wavelength. Therefore, the lengths of microstrip line 2 and microstrip line 3 differ by half a wavelength, and the phases of the signals output from port 4 and port 5 differ by 180°; the lengths of microstrip line 4 and microstrip line 3 differ by half a wavelength, and the phases of the signals output from port 2 and port 5 differ by 180°; the lengths of microstrip line 1 and microstrip line 4 differ by half a wavelength, and the phases of the signals output from port 3 and port 2 differ by 180°. Through this design, the amplitude of the signals at each port is the same, and the phases of the signals at adjacent ports differ by 180 degrees.

9. The multi-cross slot coupled feeding OAM antenna based on dielectric resonator according to claim 3, characterized in that: The physical structure of the feed network is bent and cut to create phases.

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